NSC LM4700

LM4700
Overture™ Audio Power Amplifier Series
30W Audio Power Amplifier with
Mute and Standby Modes
General Description
Key Specifications
The LM4700 is an audio power amplifier capable of delivering typically 30W of continuous average output power into an
8Ω load with less than 0.1% (THD + N).
n THD+N at 1 kHz at continuous average output power of
25W into 8Ω:
0.1% (max)
n THD+N from 20 Hz to 20 kHz at 30W of continuous
average output power into 8Ω:
0.08% (typ)
n Standby current:
2.1 mA (typ)
The LM4700 has an independent smooth transition fade-in/
out mute and a power conserving standby mode which can
be controlled by external logic.
The performance of the LM4700, utilizing its Self Peak Instantaneous Temperature (˚Ke) (SPiKe™) Protection Circuitry, places it in a class above discrete and hybrid amplifiers by providing an inherently, dynamically protected Safe
Operating Area (SOA). SPiKe Protection means that these
parts are completely safeguarded at the output against overvoltage, undervoltage, overloads, including thermal runaway
and instantaneous temperature peaks.
Features
n
n
n
n
n
SPiKe Protection
Minimal amount of external components necessary
Quiet fade-in/out mute function
Power conserving standby-mode
Isolated 11-lead TO-220 package
Applications
n Component stereo
n Compact stereo
Typical Application
Connection Diagram
Isolated Plastic Package
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Top View
Order Number LM4700TF
See NS Package Number TF11B
DS012369-1
*Optional components dependent upon specific design requirements. Refer
to the External Components Description section for a component functional
description.
FIGURE 1. Typical Audio Amplifier Application Circuit
SPiKe™ Protection and Overture™ are trademarks of National Semiconductor Corporation.
© 1999 National Semiconductor Corporation
DS012369
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LM4700 Overture Audio Power Amplifier Series
30W Audio Power Amplifier with Mute and Standby Modes
March 1998
Absolute Maximum Ratings (Notes 4, 5)
Junction Temperature (Note 8)
Thermal Resistance
θJC (Note 14)
θJA
Soldering Information
TF Package (10 sec.)
Storage Temperature
If Military/Aerospace specified devices are required,
please contact the National Semiconductor Sales Office/
Distributors for availability and specifications.
Supply Voltage |VCC| + |VEE|
(No Signal)
Supply Voltage |VCC| + |VEE|
(with Input and Load)
Common Mode Input Voltage
Differential Input Voltage
Output Current
Power Dissipation (Note 6)
ESD Susceptibility (Note 7)
66V
64V
(VCC or VEE) and
|VCC| + |VEE| ≤ 60V
60V
Internally Limited
62.5W
2000V
150˚C
2˚C/W
43˚C/W
260˚C
−40˚C ≤ TA ≤ +150˚C
Operating Ratings (Notes 4, 5)
Temperature Range
TMIN ≤ TA ≤ TMAX
Supply Voltage |VCC| + |VEE| (Note 1)
−20˚C ≤ TA ≤ +85˚C
20V to 64V
Electrical Characteristics
(Notes 4, 5) The following specifications are for VCC = +28V, VEE = −28V with RL = 8Ω, unless otherwise specified. Limits apply for TA = 25˚C.
Symbol
|VCC| + |VEE|
Parameter
Power Supply Voltage
Conditions
GND − VEE ≥ 9V
LM4700
Limit
(Note 9)
(Note
10)
18
20
V (min)
64
V (max)
(Note 11)
PO
Output Power
(Note 3)
(Continuous Average)
THD + N = 0.1% (max), f = 1 kHz
RL = 8Ω, |VCC| = |VEE| = 28V
30
25
W/ch
(min)
RL = 4Ω, |VCC| = |VEE| = 20V (Note 13)
22
15
W/ch
(min)
SR (Note 3)
Slew Rate
30W/ch, RL = 8Ω,
20 Hz ≤ f ≤ 20 kHz, AV = 26 dB
VIN = 1.414 Vrms, trise = 2 ns
ITOTAL
Total Quiescent Power
VCM = 0V, VO = 0V, IO = 0 mA
(Note 2)
Supply Current
THD + N
Total Harmonic Distortion
Plus Noise
Units
(Limits)
Typical
%
0.08
18
12
V/µs (min)
Standby: Off
25
40
mA (max)
Standby: On
2.1
mA
Standby Pin
VIL
Standby Low Input Voltage
Not in Standby Mode
VIH
Standby High Input Voltage
In Standby Mode
VIL
Mute Low Input Voltage
Output Not Muted
VIH
Mute High Input Voltage
Output Muted
VPIN8 = 2.5V
2.0
0.8
V (max)
2.5
V (min)
0.8
V (max)
Mute Pin
AM
Mute Attenuation
VOS (Note 2)
Input Offset Voltage
IB
Input Bias Current
IOS
Input Offset Current
IO
Output Current Limit
VCM = 0V, IO =
VCM = 0V, IO =
VCM = 0V, IO =
|VCC| = |VEE| =
2.0
2.5
V (min)
115
80
dB (min)
mV (max)
0 mA
2.0
15
0 mA
0.2
0.5
µA (max)
0 mA
0.002
0.2
µA (max)
3.5
2.9
APK (min)
1.8
2.3
V (max)
2.5
3.2
V (max)
115
85
dB (min)
110
85
dB (min)
10V, tON = 10 ms,
VO = 0V
VOD
Output Dropout Voltage
(Note 2)
(Note 12)
PSRR
Power Supply Rejection Ratio
(Note 2)
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|VCC − VO|, VCC = 20V, IO = +100 mA
|VO − VEE|, VEE = −20V, IO = −100 mA
VCC = 30V to 10V, VEE = −30V,
VCM = 0V, IO = 0 mA
VCC = 30V, VEE = −30V to −10V
VCM = 0V, IO = 0 mA
2
Electrical Characteristics
(Continued)
(Notes 4, 5) The following specifications are for VCC = +28V, VEE = −28V with RL = 8Ω, unless otherwise specified. Limits apply for TA = 25˚C.
Symbol
Parameter
Conditions
LM4700
Typical
Limit
(Note 9)
(Note
10)
110
80
Units
(Limits)
AVOL (Note 2)
Open Loop Voltage Gain
VCC = 35V to 10V, VEE = −10V to −35V,
VCM = 10V to −10V, IO = 0 mA
RL = 2 kΩ, ∆VO = 30V
110
90
dB (min)
GBWP
Gain-Bandwidth Product
fO = 100 kHz, VIN = 50 mVrms
7.5
5
MHz (min)
eIN
Input Noise
IHF — A Weighting Filter
RIN = 600Ω (Input Referred)
2.0
8
µV (max)
Signal-to-Noise Ratio
PO = 1W, A-Weighted,
98
dB
Measured at 1 kHz, RS = 25Ω
PO = 25W, A-Weighted
108
dB
CMRR
Common Mode Rejection Ratio
(Note 2)
(Note 3)
SNR
dB (min)
Measured at 1 kHz, RS = 25Ω
Note 1: Operation is guaranteed up to 64V, however, distortion may be introduced from SPiKe Protection Circuitry if proper thermal considerations are not taken into
account. Refer to the Application Information section for a complete explanation.
Note 2: DC Electrical Test; Refer to Test Circuit #1.
Note 3: AC Electrical Test; Refer to Test Circuit #2.
Note 4: All voltages are measured with respect to the GND (pin 7), unless otherwise specified.
Note 5: Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for which the device is functional, but do not guarantee specific performance limits. Electrical Characteristics state DC and AC electrical specifications under particular test conditions which guarantee specific performance limits. This assumes that the device is within the Operating Ratings. Specifications are not guaranteed for parameters where no limit is
given, however, the typical value is a good indication of device performance.
Note 6: For operating at case temperatures above 25˚C, the device must be derated based on a 150˚C maximum junction temperature and a thermal resistance of
θJC = 2˚C/W (junction to case). Refer to the section, Determining the Correct Heat Sink, in the Application Information section.
Note 7: Human body model, 100 pF discharged through a 1.5 kΩ resistor.
Note 8: The operating junction temperature maximum is 150˚C, however, the instantaneous Safe Operating Area temperature is 250˚C.
Note 9: Typicals are measured at 25˚C and represent the parametric norm.
Note 10: Limits are guarantees that all parts are tested in production to meet the stated values.
Note 11: VEE must have at least −9V at its pin with reference to ground in order for the under-voltage protection circuitry to be disabled. In addition, the voltage differential between VCC and VEE must be greater than 14V.
Note 12: The output dropout voltage, VOD, is the supply voltage minus the clipping voltage. Refer to the Clipping Voltage vs. Supply Voltage graph in the Typical Performance Characteristics section.
Note 13: For a 4Ω load, and with ± 20V supplies, the LM4700 can deliver typically 22 Watts of continuous average power per channel with less than 0.1% (THD+N).
With supplies above ± 20V, the LM4700 cannot deliver more than 22 watts into 4Ω due to current limiting of the output transistors. Thus, increasing the power supply
above ± 20V will only increase the internal power dissipation, not the possible output power. Increased power dissipation will require a larger heat sink as explained
in the Application Information section.
Note 14: Preliminary engineering evaluation of θJC for the TF package has been assessed as 2˚C/W. This is a preliminary engineering number and represents the
data to this point. Please contact your local National Semiconductor sales representative for more information.
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Test Circuit #1
(Note 2) (DC Electrical Test Circuit)
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Test Circuit #2
(Note 3) (AC Electrical Test Circuit)
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Bridged Amplifier Application Circuit
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FIGURE 2. Bridged Amplifier Application Circuit
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Single Supply Application Circuit
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FIGURE 3. Single Supply Amplifier Application Circuit
Auxillary Amplifier Application Circuit
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FIGURE 4. Auxillary Amplifier Application Circuit
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Equivalent Schematic
(Excluding Active Protection Circuitry)
DS012369-8
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External Components Description
Components
Functonal Description
1
RB
Prevents currents from entering the amplifier’s non-inverting input which may be passed through to the
load upon power down of the system due to the low input impedance of the circuitry when the
undervoltage circuitry is off. This phenomenon occurs when the supply voltages are below 1.5V.
2
RI
Inverting input resistance to provide AC gain in conjunction with RF. Also creates a highpass filter with CI
at fC = 1/(2πRICI).
Feedback resistance to provide AC gain in conjunction with RI.
3
RF
4
CI (Note 15)
Feedback capacitor which ensures unity gain at DC.
5
CS
Provides power supply filtering and bypassing. Refer to the Supply Bypassing application section for
proper placement and selection of bypass capacitors.
6
RV
(Note 15)
Acts as a volume control by setting the input voltage level.
7
RIN
(Note 15)
Sets the amplifier’s input terminals DC bias point when CIN is present in the circuit. Also works with CIN
to create a highpass filter at fC = 1/(2πRINCIN). Refer to Figure 4.
8
CIN
(Note 15)
Input capacitor which blocks the input signal’s DC offsets from being passed onto the amplifier’s inputs.
9
RSN
(Note 15)
Works with CSN to stabilize the output stage by creating a pole that reduces high frequency instabilities.
The pole is set at fC = 1/(2πRSNCSN). Refer to Figure 4.
10
CSN
(Note 15)
Works with RSN to stabilize the output stage by creating a pole that reduces high frequency instabilities.
11
L (Note 15)
12
R (Note 15)
Provides high impedance at high frequencies so that R may decouple a highly capacitive load and
reduce the Q of the series resonant circuit. Also provides a low impedance at low frequencies to short
out R and pass audio signals to the load. Refer to Figure 4.
13
RA
Provides DC voltage biasing for the transistor Q1 in single supply operation.
14
CA
Provides bias filtering for single supply operation.
15
RINP
(Note 15)
Limits the voltage difference between the amplifier’s inputs for single supply operation. Refer to the
Clicks and Pops application section for a more detailed explanation of the function of RINP.
16
RBI
Provides input bias current for single supply operation. Refer to the Clicks and Pops application section
for a more detailed explanation of the function of RBI.
17
RE
Establishes a fixed DC current for the transistor Q1 in single supply operation. This resistor stabilizes the
half-supply point along with CA.
Note 15: Optional components dependent upon specific design requirements.
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Typical Performance Characteristics
THD + N vs Frequency
THD + N vs Frequency
THD + N vs Frequency
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THD + N vs Output Power
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THD + N vs Output Power
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THD + N vs Output Power
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THD + N vs Output Power
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Clipping Voltage vs
Supply Voltage
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THD + N vs Output Power
THD + N vs Output Power
Clipping Voltage vs
Supply Voltage
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Clipping Voltage vs
Supply Voltage
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Typical Performance Characteristics
Power Dissipation vs
Output Power
(Continued)
Power Dissipation vs
Ouput Power
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Output Power vs
Load Resistance
Power Dissipation vs
Output Power
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Output Power vs
Supply Voltage
Output Mute vs
Mute Pin Voltage
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Pulse Response
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DS012369-26
Large Signal Response
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DS012369-27
Output Mute vs
Mute Pin Voltage
DS012369-29
DS012369-30
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Typical Performance Characteristics
Power Supply
Rejection Ratio
(Continued)
Common-Mode
Rejection Ratio
Open Loop
Frequency Response
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Safe Area
DS012369-32
Spike Protection Response
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Supply Current vs
Supply Voltage
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Pulse Thermal
Resistance
Pulse Thermal
Resistance
Supply Current vs
Output Voltage
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DS012369-38
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Typical Performance Characteristics
Pulse Power Limit
(Continued)
Pulse Power Limit
Supply Current vs
Case Temperature
DS012369-40
DS012369-41
DS012369-42
Standby Current (ICC) vs
Standby Pin Voltage
Supply Current (IEE) vs
Standby Pin Voltage
Input Bias Current vs
Case Temperature
DS012369-44
DS012369-43
DS012369-45
turn-off, the output of the LM4700 is brought to ground before the power supplies such that no transients occur at
power-down.
Application Information
MUTE MODE
By placing a logic-high voltage on the mute pin, the signal
going into the amplifiers will be muted. If the mute pin is left
floating or connected to a logic-low level, the amplifier will be
in a non-muted state. Refer to the Typical Performance
Characteristics section for curves concerning Mute Attenuation vs Mute Pin Voltage.
OVER-VOLTAGE PROTECTION
The LM4700 contains over-voltage protection circuitry that
limits the output current to approximately 3.5 Apk while also
providing voltage clamping, though not through internal
clamping diodes. The clamping effect is quite the same,
however, the output transistors are designed to work alternately by sinking large current spikes.
STANDBY MODE
The standby mode of the LM4700 allows the user to drastically reduce power consumption when the amplifier is idle.
By placing a logic-high voltage on the standby pin, the amplifier will go into Standby Mode. In this mode, the current
drawn from the VCC supply is typically less than 10 µA total
for both amplifiers. The current drawn from the VEE supply is
typically 2.1 mA. Clearly, there is a significant reduction in
idle power consumption when using the standby mode. Refer to the Typical Performance Characteristics section for
curves showing Supply Current vs Standby Pin Voltage for
both supplies.
SPiKe PROTECTION
The
LM4700
is
protected
from
instantaneous
peak-temperature stressing of the power transistor array.
The Safe Operating Area graph in the Typical Performance
Characteristics section shows the area of device operation
where SPiKe Protection Circuitry is not enabled. The waveform to the right of the SOA graph exemplifies how the dynamic protection will cause waveform distortion when enabled.
THERMAL PROTECTION
The LM4700 has a sophisticated thermal protection scheme
to prevent long-term thermal stress of the device. When the
temperature on the die reaches 165˚C, the LM4700 shuts
down. It starts operating again when the die temperature
drops to about 155˚C, but if the temperature again begins to
rise, shutdown will occur again at 165˚C. Therefore, the device is allowed to heat up to a relatively high temperature if
UNDER-VOLTAGE PROTECTION
Upon system power-up, the under-voltage protection circuitry allows the power supplies and their corresponding capacitors to come up close to their full values before turning
on the LM4700 such that no DC output spikes occur. Upon
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Application Information
SUPPLY BYPASSING
The LM4700 has excellent power supply rejection and does
not require a regulated supply. However, to improve system
performance as well as eliminate possible oscillations, the
LM4700 should have its supply leads bypassed with
low-inductance capacitors having short leads that are located close to the package terminals. Inadequate power
supply bypassing will manifest itself by a low frequency oscillation known as “motorboating” or by high frequency instabilities. These instabilities can be eliminated through multiple
bypassing utilizing a large tantalum or electrolytic capacitor
(10 µF or larger) which is used to absorb low frequency
variations and a small ceramic capacitor (0.1 µF) to prevent
any high frequency feedback through the power supply lines.
(Continued)
the fault condition is temporary, but a sustained fault will
cause the device to cycle in a Schmitt Trigger fashion between the thermal shutdown temperature limits of 165˚C and
155˚C. This greatly reduces the stress imposed on the IC by
thermal cycling, which in turn improves its reliability under
sustained fault conditions.
Since the die temperature is directly dependent upon the
heat sink used, the heat sink should be chosen such that
thermal shutdown will not be reached during normal operation. Using the best heat sink possible within the cost and
space constraints of the system will improve the long-term
reliability of any power semiconductor device, as discussed
in the Determining the Correct Heat Sink Section.
If adequate bypassing is not provided, the current in the supply leads which is a rectified component of the load current
may be fed back into internal circuitry. This signal causes
distortion at high frequencies requiring that the supplies be
bypassed at the package terminals with an electrolytic capacitor of 470 µF or more.
DETERMINING MAXIMUM POWER DISSIPATION
Power dissipation within the integrated circuit package is a
very important parameter requiring a thorough understanding if optimum power output is to be obtained. An incorrect
maximum power dissipation calculation may result in inadequate heat sinking causing thermal shutdown and thus limiting the output power.
Equation (1) exemplifies the theoretical maximum power dissipation point of each amplifier where VCC is the total supply
voltage.
PDMAX = VCC2/2π2RL
(1)
BRIDGED AMPLIFIER APPLICATION
One common power amplifier configuration is shown in Figure 2 and is referred to as “bridged mode” operation. Bridged
mode operation is different from the classical single-ended
amplifier configuration where one side of the output load is
connected to ground.
A bridge amplifier design has a distinct advantage over the
single-ended configuration, as it provides differential drive to
the load, thus doubling output swing for a specified supply
voltage. Consequently, theoretically four times the output
power is possible as compared to a single-ended amplifier
under the same conditions. This increase in attainable output
power assumes that the amplifier is not current limited or
clipped.
A direct consequence of the increased power delivered to
the load by a bridge amplifier is an increase in internal power
dissipation. For each operational amplifier in a bridge configuration, the internal power dissipation will increase by a
factor of two over the single ended dissipation. Since there
are two amplifiers used in a bridge configuration, the maximum system power dissipation point will increase by a factor
of four over the figure obtained by equation (1).
This value of PDMAX can be used to calculate the correct size
heat sink for a bridged amplifier application, assuming that
both IC’s are mounted on the same heatsink. Since the internal dissipation for a given power supply and load is increased by using bridged-mode, the heatsink’s θSA will have
to decrease accordingly as shown by equation (3). Refer to
the section, Determining the Correct Heat Sink, for a more
detailed discussion of proper heat sinking for a given application.
Thus by knowing the total supply voltage and rated output
load, the maximum power dissipation point can be calculated. Refer to the graphs of Power Dissipation vs Output
Power in the Typical Performance Characteristics section
which show the actual full range of power dissipation not just
the maximum theoretical point that results from equation (1).
DETERMINING THE CORRECT HEAT SINK
The choice of a heat sink for a high-power audio amplifier is
made entirely to keep the die temperature at a level such
that the thermal protection circuitry does not operate under
normal circumstances.
The thermal resistance from the die (junction) to the outside
air (ambient) is a combination of three thermal resistances,
θJC, θCS and θSA. The thermal resistance, θJC (junction to
case), of the LM4700 is 2˚C/W. Using Thermalloy Thermacote thermal compound, the thermal resistance, θCS (case to
sink), is about 0.2˚C/W. Since convection heat flow (power
dissipation) is analogous to current flow, thermal resistance
is analogous to electrical resistance, and temperature drops
are analogous to voltage drops, the power dissipation out of
the LM4700 is equal to the following:
PDMAX = (TJMAX − TAMB)/θJA
(2)
where TJMAX = 150˚C, TAMB is the system ambient temperature and θJA = θJC + θCS + θSA.
Once the maximum package power dissipation has been
calculated using equation (1), the maximum thermal resistance, θSA, (in ˚C/W) for a heat sink can be calculated. This
calculation is made using equation (3) which is derived by
solving for θSA in equation (2).
θSA = [(TJMAX−TAMB)−PDMAX(θJC+θCS)]/PDMAX (3)
Again it must be noted that the value of θSA is dependent
upon the system designer’s amplifier requirements. If the
ambient temperature that the audio amplifier is to be working
under is higher than 25˚C, then the thermal resistance for the
heat sink, given all other things are equal, will need to be
smaller.
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SINGLE-SUPPLY AMPLIFIER APPLICATION
The typical application of the LM4700 is a split supply amplifier. But as shown in Figure 3, the LM4700 can also be used
in a single power supply configuration. This involves using
some external components to create a half-supply bias
which is used as the reference for the inputs and outputs.
Thus, the signal will swing around half-supply much like it
swings around ground in a split-supply application. Along
with proper circuit biasing, a few other considerations must
be accounted for to take advantage of all of the LM4700
functions.
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Application Information
based upon a specific application loading and thus, the system designer may need to adjust these values for optimum
performance.
As shown in Figure 3, the resistors labeled RBI help bias up
the LM4700 off the half-supply node at the emitter of the
2N3904. But due to the input and output coupling capacitors
in the circuit, along with the negative feedback, there are two
different values of RBI, namely 10 kΩ and 200 kΩ. These resistors bring up the inputs at the same rate resulting in a popless turn-on. Adjusting these resistors values slightly may reduce pops resulting from power supplies that ramp
extremely quick or exhibit overshoot during system turn-on.
(Continued)
The LM4700 possesses a mute and standby function with internal logic gates that are half-supply referenced. Thus, to
enable either the mute or standby function, the voltage at
these pins must be a minimum of 2.5V above half-supply. In
single-supply systems, devices such as microprocessors
and simple logic circuits used to control the mute and
standby functions, are usually referenced to ground, not
half-supply. Thus, to use these devices to control the logic
circuitry of the LM4700, a “level shifter”, like the one shown
in Figure 5, must be employed. A level shifter is not needed
in a split-supply configuration since ground is also
half-supply.
AUDIO POWER AMPLlFIER DESIGN
Design a 25W/8Ω Audio Amplifier
Given:
Power Output
Load Impedance
Input Level
Input Impedance
25 Wrms
8Ω
1 Vrms(max)
47 kΩ
Bandwidth
20 Hz to 20 kHz ± 0.25 dB
A designer must first determine the power supply requirements in terms of both voltage and current needed to obtain
the specified output power. VOPEAK can be determined from
equation (4) and IOPEAK from equation (5).
DS012369-9
FIGURE 5. Level Shift Circuit
When the voltage at the Logic Input node is 0V, the 2N3904
is “off” and thus resistor RC pulls up mute or standby input to
the supply. This enables the mute or standby function. When
the Logic Input is 5V, the 2N3904 is “on” and consequently,
the voltage at the collector is essentially 0V. This will disable
the mute or standby function, and thus the amplifier will be in
its normal mode of operation. RSHIFT, along with CSHIFT, creates an RC time constant that reduces transients when the
mute or standby functions are enabled or disabled. Additionally, RSHIFT limits the current supplied by the internal logic
gates of the LM4700 which insures device reliability. Refer to
the Mute Mode and Standby Mode sections in the Application Information section for a more detailed description of
these functions.
(4)
(5)
To determine the maximum supply voltage, the following
conditions must be considered. Add the dropout voltage to
the peak output swing VOPEAK, to get the supply rail at a current of IOPEAK. The regulation of the supply determines the
unloaded voltage which is usually about 15% higher. The
supply voltage will also rise 10% during high line conditions.
Therefore the maximum supply voltage is obtained from the
following equation:
Max Supplies ≈ ± (VOPEAK + VOD) (1 + Regulation) (1.1)
CLICKS AND POPS
In the typical application of the LM4700 as a split-supply audio power amplifier, the IC exhibits excellent “click” and “pop”
performance when utilizing the mute and standby functions.
In addition, the device employs Under-Voltage Protection,
which eliminates unwanted power-up and power-down transients. The basis for these functions are a stable and constant half-supply potential. In a split-supply application,
ground is the stable half-supply potential. But in a
single-supply application, the half-supply needs to charge up
just like the supply rail, VCC.
For 25W of output power into an 8Ω load, the required VOPEAK is 20V. A minimum supply rail of ± 25V results from adding VOPEAK and VOD. With regulation, the maximum supplies
are ± 31.7V and the required IOPEAK is 2.5A from equation
(5). At this point it is a good idea to check the Power Output
vs Supply Voltage to ensure that the required output power is
obtainable from the device while maintaining low THD+N. In
addition, the designer should verify that with the required
power supply voltage and load impedance, that the required
heatsink value θSA is feasible given system cost and size
constraints. Once the heatsink issues have been addressed,
the required gain can be determined from equation (6).
This makes the task of attaining a clickless and popless
turn-on more challenging. Any uneven charging of the amplifier inputs will result in output clicks and pops due to the differential input topology of the LM4700.
To achieve a transient free power-up and power-down, the
voltage seen at the input terminals should be ideally the
same. Such a signal will be common-mode in nature, and
will be rejected by the LM4700. In Figure 3, the resistor RINP
serves to keep the inputs at the same potential by limiting the
voltage difference possible between the two nodes. This
should significantly reduce any type of turn-on pop, due to an
uneven charging of the amplifier inputs. This charging is
(6)
From equation (6), the minimum AV is AV ≥ 14.14.
By selecting a gain of 21, and with a feedback resistor, RF =
20 kΩ, the value of RI follows from equation (7).
(7)
RI = RF (AV − 1)
Thus with RJ = 1 kΩ a non-inverting gain of 21 will result.
Since the desired input impedance was 47 kΩ, a value of
47 kΩ was selected for RIN. The final design step is to address the bandwidth requirements which must be stated as a
pair of −3 dB frequency points. Five times away from a −3 dB
point is 0.17 dB down from passband response which is bet13
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Application Information
The high frequency pole is determined by the product of the
desired high frequency pole, fH, and the gain, AV. With a AV
= 21 and fH = 100 kHz, the resulting GBWP of 2.1 MHz is
less than the minimum GBWP of 5 MHz for the LM4700. This
will ensure that the high frequency response of the amplifier
will be no worse than 0.17 dB down at 20 kHz which is well
within the bandwidth requirements of the design.
(Continued)
ter than the required ± 0.25 dB specified. This fact results in
a low and high frequency pole of 4 Hz and 100 kHz respectively. As stated in the External Components section, RI in
conjunction with CI create a high-pass filter.
CI ≥ 1/(2π * 1 kΩ * 4 Hz) = 39.8 µF; use 39 µF.
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14
LM4700 Overture Audio Power Amplifier Series
30W Audio Power Amplifier with Mute and Standby Modes
Physical Dimensions
inches (millimeters) unless otherwise noted
Isolated TO-220 11-Lead Package
Order Number LM4700TF
NS Package Number TF11B
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