LTC1874 Dual Constant Frequency Current Mode Step-Down DC/DC Controller U FEATURES ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ DESCRIPTIO The LTC®1874 is a dual constant frequency current mode step-down DC/DC controller with excellent AC and DC load and line regulation. Each controller has an accurate undervoltage lockout that shuts down the individual controller when the input voltage falls below 2.0V. High Efficiency: Up to 94% High Output Currents Easily Achieved Wide VIN Range: 2.5V to 9.8V Constant Frequency 550kHz Operation Burst ModeTM Operation at Light Load Low Dropout: 100% Duty Cycle 0.8V Reference Allows Low Output Voltages Current Mode Operation for Excellent Line and Load Transient Response Low Quiescent Current: 270µA (Each Controller) Separate Shutdown Pin for Each Controller Shutdown Mode Draws Only 8µA Supply Current (Each Controller) ±2.5% Reference Accuracy Available in 16-Lead Narrow SSOP Each Controller Functions Independent of the Other The LTC1874 boasts ±2.5% output voltage accuracy and consumes only 270µA of quiescent current per controller. The LTC1874 is configured with Burst Mode operation, which enhances efficiency at low output current for applications where efficiency is a prime consideration. To further maximize the life of a battery source, each external P-channel MOSFET is turned on continuously in dropout (100% duty cycle). In shutdown, each controller draws a mere 8µA. High constant operating frequency of 550kHz allows the use of small external inductors. U APPLICATIO S ■ ■ ■ 1- or 2-Cell Lithium-Ion-Powered Applications Personal Information Appliances Portable Computers Distributed 3.3V, 2.5V or 1.8V Power Systems , LTC and LT are registered trademarks of Linear Technology Corporation. Burst Mode is a trademark of Linear Technology Corporation. U ■ The LTC1874 is available in a small footprint 16-lead narrow SSOP. TYPICAL APPLICATION VIN 3.5V TO 9.5V CIN 10µF 16V ×2 VOUT1 3.3V 1A C1, C2: SANYO POSCAP 6TPA47M CIN: TAIYO YUDEN CERAMIC EMK325BJ106MNT (× 2) D1, D2: MBRM120 L1, L2: COILCRAFT D01608C-472 M1, M2: Si3443DV R1, R2: DALE 0.25W 249k R1 0.04Ω 1 2 3 L1 M1 4.7µH + C1 D1 47µF 6V 4 10k 13 220pF 14 15 80.6k 16 LTC1874 8 PVIN2 VIN1 7 SENSE1– PGATE2 6 PGND2 GND1 5 ITH/RUN2 VFB1 12 VFB2 ITH/RUN1 11 PGND1 GND2 10 PGATE1 SENSE2 – 9 PVIN1 VIN2 R2 0.04Ω M2 L2 4.7µH 10k D2 220pF + C2 47µF 6V VOUT2 1.8V 1A 100k 80.6k 1874 TA01 Figure 1. LTC1874 3.5V-9.5V Input to 3.3V/1A and 1.8V/1A Dual Step-Down Converter 1 LTC1874 W U U U W W W ABSOLUTE MAXIMUM RATINGS PACKAGE/ORDER INFORMATION (Note 1) Input Supply Voltage (VIN, PVIN) ...............– 0.3V to 10V SENSE –, PGATE Voltages ............. – 0.3V to (VIN + 0.3V) VFB, ITH/RUN Voltages ..............................– 0.3V to 2.4V PGATE Peak Output Current (< 10µs) ....................... 1A Storage Ambient Temperature Range ... – 65°C to 150°C Operating Temperature Range (Note 2) .. – 40°C to 85°C Junction Temperature (Note 3) ............................. 150°C Lead Temperature (Soldering, 10 sec).................. 300°C TOP VIEW VIN1 1 16 PVIN1 SENSE1– 2 15 PGATE1 GND1 3 14 PGND1 VFB1 4 13 ITH/RUN1 ITH/RUN2 5 12 VFB2 PGND2 6 11 GND2 PGATE2 7 10 SENSE2 – PVIN2 8 9 ORDER PART NUMBER LTC1874EGN GN PART MARKING VIN2 1874 GN PACKAGE 16-LEAD NARROW PLASTIC SSOP TJMAX = 150°C, θJA = 135°C/ W Consult factory for parts specified with wider operating temperature ranges. ELECTRICAL CHARACTERISTICS All specifications apply to each controller. The ● denotes specifications that apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN = 4.2V unless otherwise specified. (Note 2) PARAMETER CONDITIONS Input DC Supply Current (Per Controller) Normal Operation Sleep Mode Shutdown UVLO Typicals at VIN = 4.2V (Note 4) 2.4V ≤ VIN ≤ 9.8V 2.4V ≤ VIN ≤ 9.8V 2.4V ≤ VIN ≤ 9.8V, VITH/RUN = 0V VIN < UVLO Threshold MIN Undervoltage Lockout Threshold VIN Falling VIN Rising ● Shutdown Threshold (at ITH/RUN) ● TYP MAX UNITS 270 230 8 6 420 370 22 10 µA µA µA µA 1.55 1.85 2.0 2.3 2.35 2.40 V V 0.15 0.35 0.55 V 0.25 0.5 0.85 µA 0.780 0.770 0.800 0.800 0.820 0.830 V V Start-Up Current Source VITH/RUN = 0V Regulated Feedback Voltage TA = 0°C to 70°C (Note 5) TA = – 40°C to 85°C (Note 5) Output Voltage Line Regulation 2.4V ≤ VIN ≤ 9.8V (Note 5) 0.05 mV/V Output Voltage Load Regulation ITH/RUN Sinking 5µA (Note 5) ITH/RUN Sourcing 5µA (Note 5) 2.5 2.5 mV/µA mV/µA VFB Input Current (Note 5) 10 50 Overvoltage Protect Threshold Measured at VFB 0.860 0.895 ● ● 0.820 Overvoltage Protect Hysteresis Oscillator Frequency 20 VFB = 0.8V VFB = 0V 500 550 120 nA V mV 650 kHz kHz Gate Drive Rise Time CLOAD = 3000pF 40 ns Gate Drive Fall Time CLOAD = 3000pF 40 ns Peak Current Sense Voltage (Note 6) 120 mV Note 1: Absolute Maximum Ratings are those values beyond which the life of a device may be impaired. Note 2: The LTC1874E is guaranteed to meet performance specifications from 0°C to 70°C. Specifications over the –40°C to 85°C operating temperature range are assured by design, characterization and correlation with statistical process controls. Note 3: TJ is calculated from the ambient temperature TA and power dissipation PD according to the following formula: TJ = TA + (PD • θJA°C/W) 2 Note 4: Dynamic supply current is higher due to the gate charge being delivered at the switching frequency. Note 5: Each controller in the LTC1874 is individually tested in a feedback loop that servos VFB to the output of the error amplifier. Note 6: Peak current sense voltage is reduced dependent upon duty cycle to a percentage of value as given in Figure 2. LTC1874 U W TYPICAL PERFORMANCE CHARACTERISTICS Reference Voltage vs Temperature 10 VIN = 4.2V 8 810 805 800 795 790 785 6 2.16 4 2.12 0 –2 –4 –6 5 25 45 65 85 105 125 TEMPERATURE (°C) 5 25 45 65 85 105 125 TEMPERATURE (°C) 2.04 2.00 1.96 1.84 –55 –35 –15 5 25 45 65 85 105 125 TEMPERATURE (°C) 1874 G02 Maximum (VIN – SENSE –) Voltage vs Duty Cycle 1874 G03 Shutdown Threshold vs Temperature 600 VIN = 4.2V TA = 25°C 120 2.08 1.88 –10 –55 –35 –15 1874 G01 130 VIN FALLING 1.92 –8 560 VIN = 4.2V 520 110 ITH/RUN VOLTAGE (mV) 775 –55 –35 –15 2.20 2 780 TRIP VOLTAGE (mV) VFB VOLTAGE (mV) 815 2.24 VIN = 4.2V TRIP VOLTAGE (V) 820 NORMALIZED FREQUENCY (%) 825 Undervoltage Lockout Trip Voltage vs Temperature Normalized Oscillator Frequency vs Temperature 100 90 80 70 440 400 360 320 280 60 50 20 480 240 30 40 50 60 70 80 DUTY CYCLE (%) 90 100 1874 G04 200 –55 –35 –15 5 25 45 65 85 105 125 TEMPERATURE (°C) 1874 G05 3 LTC1874 U U U PIN FUNCTIONS VIN1 (Pin 1): Main Supply Pin for Controller #1. This pin delivers the Input DC Supply Current (listed in the Electrical Characteristics table) plus a small amount of logic switching current. Must be connected to PVIN1 (Pin 16) and closely decoupled to GND1 (Pin 3). VIN2 (Pin 9): Main Supply Pin for Controller #2. This pin delivers the Input DC Supply Current (listed in the Electrical Characteristics table) plus a small amount of logic switching current. Must be connected to PVIN2 (Pin 8) and closely decoupled to GND2 (Pin 11). SENSE1 – (Pin 2): The Negative Input to the Current Comparator of Controller #1. SENSE2– (Pin 10): The Negative Input to the Current Comparator of Controller #2. GND1 (Pin 3): Signal Ground for Controller #1. Must be connected to PGND1 (Pin 14). GND2 (Pin 11): Signal Ground for Controller #2. Must be connected to PGND2 (Pin 6). VFB1 (Pin 4): Receives the feedback voltage from an external resistive divider across the output of Controller #1. VFB2 (Pin 12): Receives the feedback voltage from an external resistive divider across the output of Controller #2. ITH/RUN2 (Pin 5): This pin performs two functions. It serves as the error amplifier compensation point as well as the run control input of Controller #2. The current comparator threshold increases with this control voltage. Nominal voltage range for this pin is 0.7V to 1.9V. Forcing this pin below 0.35V causes Controller #2 to be shut down. In shutdown, all functions of Controller #2 are disabled and PGATE2 (Pin 7) is held high. ITH/RUN1 (Pin 13): This pin performs two functions. It serves as the error amplifier compensation point as well as the run control input of Controller #1. The current comparator threshold increases with this control voltage. Nominal voltage range for this pin is 0.7V to 1.9V. Forcing this pin below 0.35V causes Controller #1 to be shut down. In shutdown, all functions of Controller #1 are disabled and PGATE1 (Pin 15) is held high. PGND2 (Pin 6): Power Ground for Controller #2. Must be connected to GND2 (Pin 11). PGND1 (Pin 14): Power Ground for Controller #1. Must be connected to GND1 (Pin 3). PGATE2 (Pin 7): Gate Drive for the External P-Channel MOSFET of Controller #2. This pin swings from 0V to the voltage of PVIN2. PGATE1 (Pin 15): Gate Drive for the External P-Channel MOSFET of Controller #1. This pin swings from 0V to the voltage of PVIN1. PVIN2 (Pin 8): Power Supply Pin for Controller #2. This pin delivers the dynamic switching current that drives the gate of the external P-channel MOSFET of Controller #2. Must be connected to VIN2 (Pin 9) and closely decoupled to PGND2 (Pin 6). PVIN1 (Pin 16): Power Supply Pin for Controller #1. This pin delivers the dynamic switching current that drives the gate of the external P-channel MOSFET of Controller #1. Must be connected to VIN1 (Pin 1) and closely decoupled to PGND1 (Pin 14). 4 LTC1874 W FUNCTIONAL DIAGRA U VIN1 SENSE1– 1 2 Controller #1 U + PVIN1 ICMP 16 – RS1 R Q S SLOPE COMP OSC PGATE1 SWITCHING LOGIC AND BLANKING CIRCUIT 15 PGND1 14 – FREQ FOLDBACK BURST CMP + 0.3V + SHORT-CIRCUIT DETECT SLEEP – 0.15V OVP + – VREF + 60mV + VREF 0.8V VIN EAMP 0.5µA VFB1 + – 4 VIN VIN 0.3V – 0.35V + SHDN CMP VREF 0.8V VOLTAGE REFERENCE – GND1 SHDN UV 3 UNDERVOLTAGE LOCKOUT 1.2V 13 ITH/RUN1 VIN2 SENSE2 – 9 10 Controller #2 PVIN2 8 PGATE2 7 GND2 11 PGND2 CONTROLLER #2 IS THE SAME AS CONTROLLER #1 6 VFB2 12 1874FD 5 ITH/RUN2 5 LTC1874 U OPERATIO (Refer to Functional Diagram) The LTC1874 is a dual, constant frequency current mode switching regulator. The two switching regulators function identically but independent of each other. The following description of operation is written for a single switching regulator. high, turning off the external MOSFET. The sleep signal goes low when the ITH/RUN voltage goes above 0.925V and the controller resumes normal operation. The next oscillator cycle will turn the external MOSFET on and the switching cycle repeats. Main Control Loop Dropout Operation During normal operation, the external P-channel power MOSFET is turned on by the oscillator and turned off when the current comparator (ICMP) resets the RS latch. The peak inductor current at which ICMP resets the RS latch is controlled by the voltage on the ITH/RUN pin, which is the output of the error amplifier EAMP. An external resistive divider connected between VOUT and ground allows the EAMP to receive an output feedback voltage VFB. When the load current increases, it causes a slight decrease in VFB relative to the 0.8V reference, which in turn causes the ITH/RUN voltage to increase until the average inductor current matches the new load current. When the input supply voltage decreases towards the output voltage, the rate of change of inductor current during the ON cycle decreases. This reduction means that the external P-channel MOSFET will remain on for more than one oscillator cycle since the inductor current has not ramped up to the threshold set by EAMP. Further reduction in input supply voltage will eventually cause the P-channel MOSFET to be turned on 100%, i.e., DC. The output voltage will then be determined by the input voltage minus the voltage drop across the MOSFET, the sense resistor and the inductor. The main control loop is shut down by pulling the ITH/RUN pin low. Releasing ITH/RUN allows an internal 0.5µA current source to charge up the external compensation network. When the ITH/RUN pin reaches 0.35V, the main control loop is enabled with the ITH/RUN voltage then pulled up to its zero current level of approximately 0.7V. As the external compensation network continues to charge up, the corresponding output current trip level follows, allowing normal operation. Comparator OVP guards against transient overshoots greater than 7.5% by turning off the external P-channel power MOSFET and keeping it off until the fault is removed. Burst Mode Operation The controller enters Burst Mode operation at low load currents. In this mode, the peak current of the inductor is set as if VITH/RUN = 1V (at low duty cycles) even though the voltage at the ITH/RUN pin is at a lower value. If the inductor’s average current is greater than the load requirement, the voltage at the ITH/RUN pin will drop. When the ITH/RUN voltage goes below 0.85V, the sleep signal goes 6 Undervoltage Lockout To prevent operation of the P-channel MOSFET below safe input voltage levels, an undervoltage lockout is incorporated into the controller. When the input supply voltage drops below approximately 2.0V, the P-channel MOSFET and all circuitry is turned off except the undervoltage block, which draws only several microamperes. Short-Circuit Protection When the output is shorted to ground, the frequency of the oscillator will be reduced to about 120kHz. This lower frequency allows the inductor current to safely discharge, thereby preventing current runaway. The oscillator’s frequency will gradually increase to its designed rate when the feedback voltage again approaches 0.8V. Overvoltage Protection As a further protection, the overvoltage comparator in the controller will turn the external MOSFET off when the feedback voltage has risen 7.5% above the reference voltage of 0.8V. This comparator has a typical hysteresis of 20mV. LTC1874 U OPERATIO 110 Slope Compensation and Inductor’s Peak Current 100 The inductor’s peak current is determined by: VITH – 0.7 ( 10 RSENSE ) when the controller is operating below 40% duty cycle. However, once the duty cycle exceeds 40%, slope compensation begins and effectively reduces the peak inductor current. The amount of reduction is given by the curves in Figure 2. SF = IOUT/IOUT(MAX) (%) IPK = 90 80 70 60 50 IRIPPLE = 0.4IPK AT 5% DUTY CYCLE IRIPPLE = 0.2IPK AT 5% DUTY CYCLE 40 30 20 VIN = 4.2V 10 0 10 20 30 40 50 60 70 80 90 100 DUTY CYCLE (%) 1874 F02 Figure 2. Percentage of Maximum Output Current vs Duty Cycle U W U U APPLICATIO S I FOR ATIO The basic LTC1874 application circuit is shown in Figure 1. External component selection for each controller is driven by the load requirement and begins with the selection of L1 and RSENSE (= R1). Next, the power MOSFET (M1) and the output diode (D1) are selected followed by CIN and COUT (= C1). RSENSE Selection for Output Current RSENSE is chosen based on the required output current. With the current comparator monitoring the voltage developed across RSENSE, the threshold of the comparator determines the inductor’s peak current. The output current the controller can provide is given by: IOUT = 0.12V IRIPPLE − RSENSE 2 where IRIPPLE is the inductor peak-to-peak ripple current (see Inductor Value Calculation section). A reasonable starting point for setting ripple current is IRIPPLE = (0.4)(IOUT). Rearranging the above equation, it becomes: RSENSE = 1 for Duty Cycle < 40% 10 IOUT ( )( ) However, for operation that is above 40% duty cycle, slope compensation effect has to be taken into consideration to select the appropriate value to provide the required amount of current. Using Figure 2, the value of RSENSE is: RSENSE = SF (10)(IOUT )(100) where SF is the “slope factor.” Inductor Value Calculation The operating frequency and inductor selection are interrelated in that higher operating frequencies permit the use of a smaller inductor for the same amount of inductor ripple current. However, this is at the expense of efficiency due to an increase in MOSFET gate charge losses. The inductance value also has a direct effect on ripple current. The ripple current, IRIPPLE, decreases with higher inductance or frequency and increases with higher VIN or VOUT. The inductor’s peak-to-peak ripple current is given by: IRIPPLE = VIN − VOUT VOUT + VD VIN + VD fL () where f is the operating frequency. Accepting larger values of IRIPPLE allows the use of low inductances, but results in higher output voltage ripple and greater core losses. A reasonable starting point for setting ripple current is 7 LTC1874 U W U U APPLICATIO S I FOR ATIO IRIPPLE = 0.4(IOUT(MAX)). Remember, the maximum IRIPPLE occurs at the maximum input voltage. In Burst Mode operation on an LTC1874 controller, the ripple current is normally set such that the inductor current is continuous during the burst periods. Therefore, the peak-to-peak ripple current must not exceed: IRIPPLE ≤ 0.03V RSENSE This implies a minimum inductance of: LMIN = VIN − VOUT VOUT + VD 0.03 VIN + VD f RSENSE (Use VIN(MAX) = VIN) A smaller value than L MIN could be used in the circuit; however, the inductor current will not be continuous during burst periods. Inductor Core Selection Once the value of inductor is known, an off the shelf inductor can be selected. The inductor should be rated for the calculated peak current. Some manufacturers specify both peak saturation current and peak RMS current. Make sure that the RMS current meets your continuous load requirements. Also, you may want to compare the DC resistance of different inductors in order to optimize the efficiency. Inductor core losses are usually not specified and you will need to evaluate them yourself. Usually, the core losses are not a problem because the inductors operate with relatively low magnetic flux swings. The best way to evaluate the core losses is by measuring the converters efficiency. Converter efficiency will reveal the difference in both DC current losses and core losses. Off the shelf inductors are available from numerous manufacturers. Some of the most common manufacturers are Coilcraft, Coiltronics, Panasonic, Toko, Tokin, Murata and Sumida. 8 Power MOSFET Selection The main selection criteria for the power MOSFET are the threshold voltage VGS(TH), the “on” resistance RDS(ON), reverse transfer capacitance CRSS and total gate charge. Since the controller is designed for operation down to low input voltages, a logic level threshold MOSFET (RDS(ON) guaranteed at VGS = 2.5V) is required for applications that work close to this voltage. When these MOSFETs are used, make sure that the input supply to the controller is less than the absolute maximum VGS rating, typically 8V. The required minimum RDS(ON) of the MOSFET is governed by its allowable power dissipation. For applications that may operate the controller in dropout, i.e., 100% duty cycle, at its worst case the required RDS(ON) is given by: R DS(ON) DC=100% = PP (IOUT(MAX)) (1+ δp) 2 where PP is the allowable power dissipation and δp is the temperature dependency of RDS(ON). (1 + δp) is generally given for a MOSFET in the form of a normalized RDS(ON) vs temperature curve, but δp = 0.005/°C can be used as an approximation for low voltage MOSFETs. In applications where the maximum duty cycle is less than 100% and the controller is in continuous mode, the RDS(ON) is governed by: R DS(ON) ≅ PP (DC )IOUT (1+ δp) 2 where DC is the maximum operating duty cycle of the controller. Output Diode Selection The catch diode carries load current during the off-time. The average diode current is therefore dependent on the MOSFET duty cycle. At high input voltages the diode conducts most of the time. As VIN approaches VOUT the diode conducts only a small fraction of the time. The most stressful condition for the diode is when the output is LTC1874 U W U U APPLICATIO S I FOR ATIO short-circuited. Under this condition the diode must safely handle IPEAK at close to 100% duty cycle. Therefore, it is important to adequately specify the diode peak current and average power dissipation so as not to exceed the diode ratings. Under normal load conditions, the average current conducted by the diode is: V −V ID = IN OUT IOUT VIN + VD The allowable forward voltage drop in the diode is calculated from the maximum short-circuit current as: VF ≈ PD ISC(MAX) where PD is the allowable power dissipation and will be determined by efficiency and/or thermal requirements. Schottky diodes are a good choice for low forward drop and fast switching times. Remember to keep lead length short and observe proper grounding (see Board Layout Checklist) to avoid ringing and increased dissipation. CIN and COUT Selection In continuous mode, the source current of the P-channel MOSFET is a square wave of duty cycle (VOUT + VD)/ (VIN + VD). To prevent large voltage transients, a low ESR input capacitor sized for the maximum RMS current must be used. The maximum RMS capacitor current is given by: CIN Required IRMS ≈ IMAX [V (V OUT IN − VOUT This formula has a maximum at VIN = 2VOUT, where IRMS = IOUT /2. This simple worst-case condition is commonly used for design because even significant deviations do not offer much relief. Several capacitors may be paralleled to meet the size or height requirements in the design. Due to the high operating frequency of the controller, ceramic capacitors can also be used for CIN. Always consult the manufacturer if there is any question. The selection of COUT is driven by the required effective series resistance (ESR). Typically, once the ESR requirement is satisfied, the capacitance is adequate for filtering. The output ripple (∆VOUT) is approximated by: 1 ∆VOUT ≈ IRIPPLE ESR + 4 fC OUT where f is the operating frequency, COUT is the output capacitance and IRIPPLE is the ripple current in the inductor. The output ripple is highest at maximum input voltage since ∆IL increases with input voltage. Once the ESR requirement for COUT has been met, the RMS current rating generally far exceeds the IRIPPLE(P-P) requirement. Multiple capacitors may have to be paralleled to meet the ESR or RMS current handling requirements of the application. Aluminum electrolytic and dry tantalum capacitors are both available in surface mount configurations. An excellent choice of tantalum capacitors are the AVX TPS and KEMET T510 series of surface mount tantalum capacitors. )] 1/ 2 VIN 9 LTC1874 U W U U APPLICATIO S I FOR ATIO 105 Although the controller can function down to approximately 2.0V, the maximum allowable output current is reduced when VIN decreases below 3V. Figure 3 shows the amount of change as the supply is reduced down to 2V. Also shown in Figure 3 is the effect of VIN on VREF as VIN goes below 2.3V. 100 NORMALIZED VOLTAGE (%) Low Supply Operation VREF VITH 95 90 85 80 Setting Output Voltage The controller develops a 0.8V reference voltage between the feedback (VFB) terminal and ground (see Figure 4). By selecting resistor R1, a constant current is caused to flow through R1 and R2 to set the overall output voltage. The regulated output voltage is determined by: 75 2.0 2.2 3.0 2.4 2.6 2.8 INPUT VOLTAGE (V) 1874 F03 Figure 3. Line Regulation of VREF and VITH R2 VOUT = 0.8V 1 + R1 VOUT 1/2 LTC1874 For most applications, an 80k resistor is suggested for R1. To prevent stray pickup, locate resistors R1 and R2 close to the LTC1874. VFB1 R2 4 R1 GND1 3 1874F04 Foldback Current Limiting As described in the Output Diode Selection, the worstcase dissipation occurs with a short-circuited output when the diode conducts the current limit value almost continuously. To prevent excessive heating in the diode, foldback current limiting can be added to reduce the current in proportion to the severity of the fault. Foldback current limiting is implemented by adding diodes DFB1 and DFB2 between the output and the ITH/RUN pin as shown in Figure 5. In a hard short (VOUT = 0V), the current will be reduced to approximately 50% of the maximum output current. 10 Figure 4. Setting Output Voltage 1/2 LTC1874 VOUT R2 13 ITH /RUN1 VFB1 4 + DFB1 R1 GND1 3 DFB2 1874 F05 Figure 5. Foldback Current Limiting LTC1874 U W U U APPLICATIO S I FOR ATIO PC Board Layout Checklist When laying out the printed circuit board, the following checklist should be used to ensure proper operation of the LTC1874. These items are illustrated graphically for a single controller in the layout diagram in Figure 6. Check the following in your layout: 1. Is the Schottky diode closely connected between power ground (PGND) and the drain of the external MOSFET? 2. Does the (+) plate of CIN connect to the sense resistor as closely as possible? This capacitor provides AC current to the MOSFET. 3. Is the input decoupling capacitor (0.1µF) connected closely between VIN and signal ground (GND)? 5. Is the trace from SENSE – to the SENSE resistor kept short? Does the trace connect close to RSENSE? 6. Keep the switching node PGATE away from sensitive small signal nodes. 7. Does the VFB pin connect directly to the feedback resistors? The resistive divider R1 and R2 must be connected between the (+) plate of COUT and signal ground. 8. PVIN must connect to VIN and PGND must connect to GND. Isolate high current power paths from signal power and signal ground where possible in the layout. An unbroken ground plane is recommended. 4. Connect the end of RSENSE as close to VIN as possible. The VIN pin is the SENSE + of the current comparator. VIN + CIN L1 RSENSE SW VOUT M1 + R2 COUT D1 1 2 0.1µF 3 4 R1 1/2 LTC1874 VIN SENSE – GND VFB PVIN PGATE PGND ITH/RUN 16 15 14 13 RITH CITH 1874 F06 BOLD LINES INDICATE HIGH CURRENT PATHS Figure 6. LTC1874 Layout Diagram (See PC Board Layout Checklist) Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. 11 LTC1874 U TYPICAL APPLICATIO LTC1874 2.5V–8.5V Input to 3.3V/1A and 1.8V/1A Dual Converter VIN 2.5V TO 8.5V CIN 10µF 16V ×2 R1 0.03Ω 1 2 C1, C2: SANYO POSCAP 6TPA47M CIN: TAIYO YUDEN CERAMIC EMK325BJ106MNT (× 2) D1: 15MQ040N D2: MBRM120 L1: BH-ELECTRONICS BH511-1012 L2: COILTRONICS UP2B-4R7 M1, M2: Si3443DV R1, R2: DALE 0.25W VOUT1 3.3V 1A 249k L1 + • C1 47µF ×2 3 M1 4 • 10k D1 L1A 13 220pF 14 15 16 80.6k LTC1874 8 VIN1 PVIN2 7 SENSE1– PGATE2 6 GND1 PGND2 5 VFB1 ITH/RUN2 12 ITH/RUN1 VFB2 11 GND2 PGND1 10 PGATE1 SENSE2 – 9 VIN2 PVIN1 R2 0.082Ω M2 L2 4.7µH + 10k D2 220pF VOUT2 1.8V 1A C2 47µF 6V 100k 80.6k 1874 TA02 U PACKAGE DESCRIPTIO Dimensions in inches (millimeters) unless otherwise noted. GN Package 16-Lead Plastic SSOP (Narrow 0.150) (LTC DWG # 05-08-1641) 0.189 – 0.196* (4.801 – 4.978) 0.015 ± 0.004 × 45° (0.38 ± 0.10) 0.007 – 0.0098 (0.178 – 0.249) 0.053 – 0.068 (1.351 – 1.727) 0.004 – 0.0098 (0.102 – 0.249) 0.009 (0.229) REF 16 15 14 13 12 11 10 9 0° – 8° TYP 0.016 – 0.050 (0.406 – 1.270) 0.008 – 0.012 (0.203 – 0.305) 0.0250 (0.635) BSC 0.229 – 0.244 (5.817 – 6.198) * DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE ** DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE 0.150 – 0.157** (3.810 – 3.988) GN16 (SSOP) 1098 1 2 3 4 5 6 7 8 RELATED PARTS PART NUMBER DESCRIPTION COMMENTS LTC1147 Series High Efficiency Step-Down Switching Regulator Controllers 100% Duty Cycle, 3.5V ≤ VIN ≤ 16V LTC1622 Synchronizable Low Input Voltage Current Mode Step-Down DC/DC Controller VIN 2V to 10V, IOUT Up to 4.5A, Burst Mode Operation Optional, 8-Lead MSOP LTC1624 High Efficiency SO-8 N-Channel Switching Regulator Controller 8-Pin N-Channel Drive, 3.5V ≤ VIN ≤ 36V LTC1625 No RSENSETM Synchronous Step-Down Regulator 97% Efficiency, No Sense Resistor; Up to 10A LTC1626 Low Voltage, High Efficiency Step-Down DC/DC Converter Monolithic, Constant Off-Time, Low Voltage Range: 2.5V to 6V LTC1628 Dual, 2-Phase Synchronous Step-Down Controller Minimum CIN and COUT, 3.5V ≤ VIN ≤ 36V LTC1735 Single, High Efficiency, Low Noise Synchronous Switching Controller High Efficiency 5V to 3.3V Conversion at up to 15A LT1767 1.5A, 500kHz Step-Down Switching Regulators High Frequency, Small Inductor, High Efficiency LTC1772 Constant Frequency Current Mode Step-Down DC/DC Controller VIN 2.5V to 9.8V, IOUT Up to 4A, SOT-23 Package LTC1773 Synchronous Step-Down Controller VIN 2.65V to 8.5V, IOUT up to 4A LTC1877/LTC1878 Low Voltage, Monolithic Synchronous Step-Down Regulator Low Supply Voltage Range: 2.65V to 8V, IOUT = 0.5A No RSENSE is a trademark of Linear Technology Corporation. 12 Linear Technology Corporation 1874f LT/TP 0201 4K • PRINTED IN USA 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408)432-1900 ● FAX: (408) 434-0507 ● www.linear-tech.com LINEAR TECHNOLOGY CORPORATION 2000