MICREL MIC2155YML

MIC2155/2156
Two-Phase, Single-Output, PWM
Synchronous Buck Control IC
General Description
Features
The MIC2155 and MIC2156 are a family of two-phase,
single-output synchronous buck control ICs featuring small
size, high efficiency, and a high level of flexibility. The ICs
implement a 500kHz (MIC2155) and 300kHz (MIC2156)
voltage mode PWM control, with the outputs switching
180° out of phase. The result of the out-of-phase operation
is 1MHz (or 600kHz) input ripple with ripple current
cancellation, minimizing the required input filter
capacitance. A 1% output voltage tolerance allows the
maximum level of system performance. Internal drivers
with adaptive gate drive allow the highest efficiency with
the minimum external components.
Two independent enable pins and a power good output
are provided, allowing a high level of control and
sequencing capability.
The MIC215x family has a junction operating range from
-40°C to +125°C.
• Synchronous buck control ICs with outputs switching
180° out-of-phase
• Remote sensing with internal differential amplifier
• 4.5V to 14.5V input voltage range
• Adjustable output voltages down to 0.7V
• 1% output voltage accuracy
• Starts up into a pre-biased output
• 500kHz PWM operation (MIC2155)
• 300kHz PWM operation (MIC2156)
• Adaptive gate drive allows efficiencies over 95%
• Adjustable current limit with no sense resistor
• Senses low-side MOSFET current
• Internal drivers allow 25A per phase
• Power Good output allow simple sequencing
• Dual enables with micro-power shutdown and UVLO
• Programmable soft-start pin
• Output over-voltage protection
• Works with ceramic output capacitors
• Multi-input supply capability
• Single-output high-current capability with master/slave
current sharing
• External synchronization
• Small footprint 32-pin 5mm × 5mm MLF®
• Junction temperature range of -40°C to +125°C
Data sheets and support documentation can be found on
Micrel’s web site at www.micrel.com.
Applications
•
•
•
•
•
Multi-output power supplies with sequencing
DSP, FPGA, CPU and ASIC power supplies
DSL modems
Telecom and networking equipment
Servers
MLF and MicroLead Frame are registered trademarks of Amkor Technologies
Micrel Inc. • 2180 Fortune Drive • San Jose, CA 95131 • USA • tel +1 (408) 944-0800 • fax + 1 (408) 474-1000 • http://www.micrel.com
November 2009
M9999-111209-B
Micrel, Inc.
MIC2155/2156
Contents
Ordering Information ............................................................................................................................................................... 3
Typical Application .................................................................................................................................................................. 3
Pin Configuration..................................................................................................................................................................... 4
Pin Description ........................................................................................................................................................................ 4
Absolute Maximum Ratings .................................................................................................................................................... 6
Operating Ratings ................................................................................................................................................................... 6
Electrical Characteristics......................................................................................................................................................... 6
Typical Characteristics ............................................................................................................................................................ 9
Functional Diagram ............................................................................................................................................................... 11
Functional Description........................................................................................................................................................... 12
Startup .............................................................................................................................................................................. 12
Soft Start ........................................................................................................................................................................... 12
Enable............................................................................................................................................................................... 13
Supply Voltages and Internal References ........................................................................................................................ 13
UVLO ................................................................................................................................................................................ 14
Power Good ...................................................................................................................................................................... 14
Oscillator and Frequency Synchronization ....................................................................................................................... 15
MOSFET Gate Drive Circuitry .......................................................................................................................................... 15
dv/dt Induced Turn On of the Low-Side MOSFET............................................................................................................ 16
Remote Sense .................................................................................................................................................................. 16
Setting the Output Voltage................................................................................................................................................ 17
Current Limit and Overcurrent Protection ......................................................................................................................... 17
Current Limit Setting ......................................................................................................................................................... 18
The simple method ....................................................................................................................................................... 18
Accurate method .......................................................................................................................................................... 18
Example: ........................................................................................................................................................................... 18
Inductor Current Sensing.................................................................................................................................................. 19
Current Sharing................................................................................................................................................................. 19
Startup into a Pre-Biased Output...................................................................................................................................... 20
Separate Input Supplies ................................................................................................................................................... 20
Component Selection, Guidelines and Design Example .................................................................................................. 21
Output Filter Selection ...................................................................................................................................................... 21
Output Capacitor Selection............................................................................................................................................... 23
Inductor Current Sense Components ............................................................................................................................... 24
Input Capacitor Selection.................................................................................................................................................. 24
MOSFET Selection ........................................................................................................................................................... 24
RMS Current and MOSFET Power Dissipation Calculation............................................................................................. 25
External Schottky Diode ................................................................................................................................................... 26
Snubber Design ................................................................................................................................................................ 26
Compensation of the Output Voltage Loop ...................................................................................................................... 28
Error Amplifier Design Procedure ......................................................................................................................................... 30
Step 1: Decide on the crossover frequency...................................................................................................................... 30
Step 2: Determine the gain required at the crossover frequency ..................................................................................... 30
Step 3: Determine the gain boost needed at the crossover frequency (fc) ...................................................................... 30
Step 4: Determine the frequencies fz2 and fp1 ................................................................................................................ 30
Step 5: Determine the frequency for fz1........................................................................................................................... 30
Step 6: Determine the frequency for fp2........................................................................................................................... 31
Calculating Error Amplifier Component Values ................................................................................................................ 31
Compensation of the Current Sharing Loop ..................................................................................................................... 31
General Layout and Component Placement .................................................................................................................... 33
Design and Layout Checklist ............................................................................................................................................ 34
Package Information ............................................................................................................................................................. 35
November 2009
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M9999-111209-B
Micrel, Inc.
MIC2155/2156
Ordering Information
Part Number
Frequency
Voltage
Junction Temp.
(1)
Range
Package
Lead Finish
MIC2155YML
500kHz
Adj.
-40°C to +125°C
32-Pin 5mm × 5mm MLF
Pb-Free
MIC2156YML
300kHz
Adj.
-40°C to +125°C
32-Pin 5mm × 5mm MLF
Pb-Free
Typical Application
VIN
500µF
10µF
30
21
29
5
28
13
1
FDMS7672
VOUT
1.8V
40A
1.5µH
0.22µF
FDMS7660
×2
0.1µF
2
25
3
24
4
23
31
27
10
18
11
26
32
20
FDMS7672
FDMS7660 ×2
250µF
17
15
9
16
8
6
7
19
12
40A Two-Phase Converter
November 2009
3
M9999-111209-B
Micrel, Inc.
MIC2155/2156
BST2
EN1
VDD
LSD2
PGND2
PGND1
LSD1
VIN1
Pin Configuration
BST1
HSD1
SW1
CS1
EN2
SS
HSD2
SW2
N/C
VIN2
VOUT
COMP2
COMP1
FB1
FB2
EA2+
PGOOD
AGND
AVDD
N/C
SYNC
DIFFOUT
RMVOUT
RMGND
EP
®
32-Pin MLF (ML)
Pin Description
Pin Number
Pin Name
Pin Function
1
BST1
Boost 1 (Input): Provides voltage for high-side MOSFET driver 1. The gate drive
voltage is higher than the source voltage by VDD minus a diode drop.
2
HSD1
High-Side Drive 1 (Output): High-current output-driver for external high-side
MOSFET.
3
SW1
Switch Node 1 (Output): Return for HSD1
CS1
Current Sense 1 (Input). Current-limit comparator non-inverting input. Current is
sensed across the side 1 low-side FET during the off-time. Current limit is set by
the resistor in series with the CS1 pin.
5
EN2
Enable 2 (Input): Channel 2 enable. Pull high to enable. Pull low to disable.
6
SS
Soft Start (Input): Controls the turn-on time of the output voltage. Active at
Power-up, Enable, and Current Limit recovery.
7
COMP1
8
FB1
9
DIFFOUT
Output of remote sense differential amplifier.
10
RMVOUT
Remote VOUT: Connect to VOUT at the remote sense point. Input to precision
differential amplifier.
11
RMGND
12
AGND
Analog Ground
13
AVDD
Analog supply voltage (input). Connect to VDD through an RC filter network
14
N/C
15
SYNC
4
November 2009
Compensation 1 (Input): Output of the internal error amplifier for Channel 1.
Feedback 1 (Input): Negative input to the error amplifier of Channel 1.
Remote Ground: Connect to Ground at the remote sense point. Input to
precision differential amplifier.
No Connect
Sync (Input): Synchronizes switching to an external source. Leave floating when
not used.
4
M9999-111209-B
Micrel, Inc.
MIC2155/2156
Pin Description cont.
Pin Number
Pin Name
Pin Function
16
PGOOD
Power Good (Output): Asserts high when voltage on the FB pin rises above
Power Good threshold.
17
EA2+
(input) Positive input to Channel 2 (current sharing) Error Amplifier. Connect to
Channel 1 current sense.
18
FB2
Negative Input to Channel 2 (current sharing) Error Amplifier (Input). Connect to
Channel 2 current sense.
19
COMP2
20
VOUT
21
VIN2
Supply Voltage for Channel 2 (Input). Used for Channel 2 UVLO circuit.
22
N/C
No Connect
23
SW2
Switch node 2 (Output): Return for HSD2.
24
HSD2
High-Side Drive 2 (Output): High-current output-driver for the high-side
MOSFET.
25
BST2
Boost 2 (Input): Provides voltage for high-side MOSFET driver in Channel 2. The
gate drive voltage is higher than the source voltage by VDD minus a diode drop.
26
PGND2
27
LSD2
Low-Side Drive 2 (Output): High-current driver output for Channel 2 low-side
external MOSFET.
VDD
5V Internal Linear Regulator from VIN1 (Output): VDD is the ext. MOSFET gate
drive supply voltage and an internal supply bus for the IC. When VIN1 is <5V, this
regulator operates in drop-out mode. Connect external bypass capacitor.
29
EN1
Enable 1 (Input): Output enable. Turns off both sides. Pull high to enable. Pull
low to disable.
30
VIN1
Supply voltage Channel 1 (Input): Used for UVLO and VDD circuits.
31
LSD1
Low-Side Drive 1 (Output): High-current driver output for Channel 1 low-side
external MOSFET.
32
PGND1
EPAD
EP
28
November 2009
Compensation 2 (Input): Pin for external compensation of Channel 2.
Output Sense (input): Connect to output side of inductors. Used for current
sharing.
Power Ground 2. High current return for low-side driver 2.
Power Ground 1. High current return for low-side driver 1.
Exposed Pad (Power): Must make a full connection to the GND plane to
maximize thermal performance of the package.
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M9999-111209-B
Micrel, Inc.
MIC2155/2156
Absolute Maximum Ratings(1)
Operating Ratings(2)
Supply Voltage (VIN 1, 2) .................................. -0.3V to 15V
Bootstrapped Voltage (VBST) ................................... VIN + 6V
SS, FB1, RMVOUT, RMGND, AVDD, Sync, EA2+, FB2,
VOUT .................................................................... -0.3V to 6V
CS1, EN1, EN2 ................................................ -0.3V to 15V
Junction Temperature Range...............-40°C ≤ TJ ≤ +125°C
Ambient Storage Temperature...................-65°C to +150°C
ESD .................................................... 100V Machine Model
1500V Human Body Model
Lead Temperature (soldering 10sec)......................... 260°C
Supply Voltage (VIN 1, 2)............................ +4.5V to +14.5V
Output Voltage Range...................................... 0.7V to 3.6V
Package Thermal Resistance
5mm × 5mm MLF® (θJA) ....................................50°C/W
5mm × 5mm MLF® (θJC).......................................5°C/W
Electrical Characteristics(3)
TJ = 25°C; VEN = VIN1 = VIN2 =12V; unless otherwise specified. Bold values indicate -40°C≤TJ≤+125°C
Parameter
Min
Condition
Typ
Max
Units
6
10
mA
210
300
µA
VIN , VDD and VREF Supply
Total Supply Current,
IVIN1 + IVIN2
VFB = 0.8V (both O/Ps; non-switching)
PWM Mode Supply Current
Shutdown Current
VEN1 = VEN2 = 0V
CH1 VIN UVLO Start Voltage
VDD = Open
3.6
4
4.4
V
CH1 VIN UVLO Stop Voltage
VDD = Open
3.4
3.97
4.2
V
CH2 VIN UVLO Start Voltage
VDD = Open
2.5
2.7
2.9
V
CH2 VIN UVLO Stop Voltage
VDD = Open
2.3
2.5
2.7
V
VDD UVLO Start Voltage
VIN1 = VDD for VIN < 6V
3.6
VDD UVLO Stop Voltage
VIN1 = VDD for VIN < 6V
3.3
VIN UVLO Hysteresis
VDD = open
40
VEN Shutdown Threshold
(Each Channel)
VEN Hysteresis
(Each Channel)
0.6
1
mV
1.6
30
V
mV
IVDD = -75mA
4.9
5.25
5.6
IVDD = -50mA VIN = 6V
4.9
5
5.6
MIC2155
MIC2156
450
270
510
310
550
330
kHz
MIC2155
MIC2156
860
500
1200
680
kHz
Sync Level
0.5
3
V
Maximum Duty Cycle
(each Channel)
80
Internal Bias Voltage (VDD)
V
Oscillator / PWM Section
PWM Frequency
Sync Range
Minimum Headroom
between VDD and VOUT
November 2009
Sync Input is 2x PWM Frequency
Required for remote sense amplifier use
6
%
1.3
V
M9999-111209-B
Micrel, Inc.
MIC2155/2156
Electrical Characteristics(3) cont.
TJ = 25°C; VEN = VIN1 = VIN2 =12V; unless otherwise specified. Bold values indicate -40°C≤TJ≤+125°C
Parameter
Condition
Minimum On-Time
(each Channel) Note 4
Min
Typ
Max
30
Units
ns
Regulation
CH1 Feedback Voltage
Reference
(+/- 1%)
(+/- 2%)
CH1 Feedback Bias Current
VFB=0.7V
Output Voltage Line
Regulation
4.5 ≤ VIN ≤ 14.5
693
686
697
707
714
mV
30
nA
0.08
%
0.5
%
0.6
%
10
mV
DC Gain
70
dB
Output Sourcing/Sinking
Current
1
mA
70
dB
1.25
mS
Output Voltage Load
Regulation
Output Voltage Total
Regulation
4.5V ≤ VIN ≤ 14.5V; 1A ≤ IOUT ≤ 10A
(VOUT = 2.5V)
Channel Current Balancing
Asynchronous Mode VTH for
Slave Output
Error Amplifier (CH1)
Error Amplifier (CH2)
DC Gain
Transconductance
Differential Amplifier
1
Voltage Gain
Offset Voltage
Output Sourcing Current
Range
-20
+20
mV
0
500
µA
114
%Nom
Output Over Voltage Protection
VFB Threshold
(Latches LSD High)
106
109
1
Delay Blanking time
µs
Soft Start
1.25
1
Internal Soft-Start Source
Current
November 2009
7
2
2.75
3
µA
M9999-111209-B
Micrel, Inc.
MIC2155/2156
Electrical Characteristics(3) cont.
TJ = 25°C; VEN = VIN1 = VIN2 =12V; unless otherwise specified. Bold values indicate -40°C≤TJ≤+125°C
Parameter
Condition
Min
Typ
Max
Units
180
195
220
µA
-10
0
+10
mV
86
88.5
91
%Nom
0.225
0.3
V
Current Sense
CS Over Current Trip Point
Program Current
CS Comparator Sense
Threshold
(Senses drop across low-side FET)
Power Good
VFB Threshold
PGOOD Voltage Low
VFB = 0 V; IPGOOD = 1mA
Gate Drivers
Into 3000pF
Source
Sink
23
16
High-Side Drive Resistance
VDD = VIN = 5V
Source
Sink
1.6
1.7
3.5
2.5
Ω
Ω
Low-Side Drive Resistance
VDD = VIN = 5V
Source
Sink
2
1.4
3.5
2.5
Ω
Ω
Rise/Fall Time
Driver Non-Overlap Time
(adaptive)
60
ns
ns
Notes:
1. Absolute maximum ratings indicate limits beyond which damage to the component may occur. Electrical specifications do not apply when operating
the device outside of its operating ratings. The maximum allowable power dissipation is a function of the maximum junction temperature, TJ(Max), the
junction-to-ambient thermal resistance, θ JA, and the ambient temperature, TA. The maximum allowable power dissipation will result in excessive die
temperature, and the regulator will go into thermal shutdown.
2. The device is not guaranteed to function outside its operating rating.
3. Specification for packaged product only.
4. Minimum on-time before automatic cycle skipping begins. See applications section.
November 2009
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M9999-111209-B
Micrel, Inc.
MIC2155/2156
Typical Characteristics
VIN2 UVLO Threshold
V
IN
5.4
= 12V
No Switching
5.2
VIN = 12V
5.0
-4.00
-6.00
-8.00
4.8
4.6
4.4
4.2
= 5V
3.5
10 20 30 40 50 60 70
LOAD (mA)
3.0
4
6
8
10 12 14
INPUT VOLTAGE (V)
9
16
140
120
0.6980
0.6975
0.6970
0.6965
0.6960
140
IN
3.5
= 12V
120
V
4.0
IN
100
4.0
V
0.6985
80
4.5
0.6990
-20
4.5
vs. Input Voltage
No Switching
Feedback Voltage
vs. Temperature
-40
5.0
VDD (V)
5.0
100
TEMPERATURE (°C)
FEEDBACK VOLTAGE (V)
IN
5.5
= 12V
60
-40
140
120
100
80
60
40
20
0
-20
VIN = 5V
4.0
80
-10.00
VDD
V
140
vs. Temperature
TEMPERATURE (°C)
vs. Load
120
0.94 CH2 Falling
0.92
CH1 Falling
0.90
0.88
80
CH1 Rising CH2 Rising
VDD
-2.00
-40
1.04
1.02
1.00
0.98
0.96
TEMPERATURE (°C)
0.00
-14
VIN = 12V
40
16
-12.00
16
1.08
1.06
100
QUIESCENT CURRENT (mA)
140
120
8
10 12 14
INPUT VOLTAGE (V)
16
Enable Threshold
vs. Temperature
0
6
8
10 12 14
INPUT VOLTAGE (V)
20
0.04
0.02
0
4
VDD
VDD (V)
6
40
-2.00
November 2009
No Switching
60
-1.50
3.0
0
1
40
-1.00
No Switching
2
20
-0.50
IQ2
0.08
0.06
2.00
FREQUENCY (%)
FREQUENCY (%)
0.12
0.10
4.00
0.50
0.00
IQ1
ENABLE1,2 = 0V
Change in Switching Frequency
vs. Temperature
1.00
5.5
3
-20
16
Change in Switching Frequency
vs. Input Voltage
8
10 12 14
INPUT VOLTAGE (V)
4
0
0.04
0.03
0.18
0.16
0.14
Shutdown Current
vs. Input Voltage
VDD (V)
0.06
0.05
6
5
-40
0.20
0.09
0.08
0.07
-2.5
4
6
0
4
ENABLE THRESHOLD (V)
Quiescent Current 2
vs. Input Voltage
0.02
0.01
No Switching
0
4
6
8
10 12 14
INPUT VOLTAGE (V)
7
TEMPERATURE (°C)
SHUTDOWN CURRENT (mA)
QUIESCENT CURRENT (mA)
0.10
8
-20
TEMPERATURE (°C)
80
UVLO Falling
2.45
2.40
140
120
80
100
60
40
0
20
-40
3.90
-20
3.92
100
UVLO Falling
3.94
2.50
60
3.96
2.55
40
3.98
2.60
0
4.00
2.65
-20
UVLO Rising
-40
4.02
2.70
20
UVLO THRESHOLD (V)
UVLO THRESHOLD (V)
4.04
9
UVLO Rising
60
2.75
4.06
20
4.08
Quiescent Current 1
vs. Input Voltage
0
VIN1 UVLO Threshold
TEMPERATURE (°C)
M9999-111209-B
Micrel, Inc.
MIC2155/2156
Typical Characteristics (cont.)
R
1.8
RDSON (ohms)
1.7
DSON
140
120
80
100
60
40
0
-40
20
170
DSON
High Side Drive
Source/Sink
SINK
140
120
100
80
60
SOURCE
20
0.5
0
1
-20
RDSON (ohms)
VIN = 5V
2.2
V = 5V
2.1 IN
2.0
1.9
1.8
1.7
1.6
1.5
1.4
1.3
1.2
-40
VIN = 12V
TEMPERATURE (°C)
180
R
1.5
0
190
TEMPERATURE (°C)
2.5
2
VIN = 5V
200
-20
CS PIN CURRENT (µA)
3
= 12V
210
160
16
140
SOFT START CURRENT (µA)
140
120
80
100
60
TEMPERATURE (°C)
8
10 12 14
INPUT VOLTAGE (V)
Soft Start Current
vs. Temperature
3.5
VIN = 5V
40
0
IN
= 12V
20
V
6
120
16
-20
8
10 12 14
INPUT VOLTAGE (V)
Differential Amplifier Gain
vs. Temperature
-20
1.022
1.020
1.018
1.016
1.014
1.012
1.010
1.008
1.006
1.004
1.002
1.000
6
-40
GAIN
0.6950
4
192
191
190
4
80
0.6955
194
193
100
0.6960
196
195
60
0.6965
V
IN
40
0.6970
220
199
198
197
0
CS PIN CURRENT (µA)
0.6975
CS Pin Current
vs. Temperature
40
200
-40
FEEDBACK VOLTAGE (V)
0.6980
CS Pin Current
vs. Input Voltage
20
Feedback Voltage
vs. Input Voltage
TEMPERATURE (°C)
LowSide Drive
Source/Sink
VIN = 5V
SINK
1.6
1.5
1.4
SOURCE
1.3
1.2
140
120
100
80
60
40
20
0
-40
1
-20
1.1
TEMPERATURE (°C)
November 2009
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M9999-111209-B
Micrel, Inc.
MIC2155/2156
Functional Diagram
MIC2155/6 Block Diagram
November 2009
11
M9999-111209-B
Micrel, Inc.
MIC2155/2156
Functional Description
The MIC2155 and MIC2156 are two-phase, synchronous
buck controllers operating at a fixed frequency. The two
controllers differ only in switching frequency with the
MIC2155 switching at 500kHz per phase (1MHz at the
input and output) and the MIC2156 switches at 300kHz
per phase.
Some of the advantages of multi-phase operation are:
• Smaller input and output filtering components are
required because of current cancelation and higher
input and output frequency.
• Faster transient response is possible with smaller
output filter component values.
• Load current through each phase is one half the total
output current, which allows for even heat distribution
and smaller components.
• Control circuitry forces better current sharing in the
MOSFETs than paralleling FETs in a single phase
application.
The controller utilizes a voltage-mode control scheme
(VMC). Lossless current sharing is accomplished by
sensing the DC voltage across each inductor winding.
Lossless overcurrent protection is performed by sensing
the voltage across the low-side MOSFET on-resistance
during the off-time.
Other features of the controller are:
VOUT
(1V/div)
A typical output voltage and inductor current startup is
shown in Figure 2.
INDUCTOR CURRENT INDUCTOR CURRENT
(2A/div)
(2A/div)
•
•
•
•
•
•
•
•
•
Figure 1. Startup Sequence
Overvoltage protection
Soft start
UVLO
Enable
Remote sensing
Pre-biased output startup
Multiple input supplies
Power Good signal
Frequency synchronization
Channel 2
Time (4ms/div)
Figure 2. Turn On
Startup
A typical startup sequence is shown in Error! Reference
source not found. (also refer to the block diagram). The
enable pins are asserted after VIN is applied. VDD is
immediately turned on and an internal FET releases the
soft start pin. The soft start pin controls the error
amplifier voltage. As VSS ramps up, it reaches a
threshold where the gate drive is enabled and the
MOSFETs start to switch at a very low duty cycle. The
rise of the soft start voltage controls the increase in VOUT
by gradually allowing the COMP1 pin voltage to rise. A
10mV offset in the current controller keeps the Channel
2 low-side drive off when the output current is low to
prevent current from circulating between the phases.
PGOOD is asserted when VOUT reaches the PGOOD
threshold.
November 2009
Channel 1
Soft Start
The soft-start capacitor controls how fast the output
voltage rises by controlling the COMP pin risetime.
Without soft start a fast or uncontrolled turn-on requires
a higher current from the input source to charge up the
output capacitance.
The soft-start capacitor also controls the delay time
between the enable pin assertion to when VOUT starts to
rise. Figures 3 and 4 show the soft-start circuitry and
waveform timing.
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MIC2155/2156
Enable
There is an enable pin for each of the two channels.
Asserting EN1 will enable Channel 1 gate drive and
release the soft start circuit. De-asserting EN1 will
disable the gate drive, discharge CSS and disable VDD. It
will bring the controller into a low current off state.
Enable 2 only controls switching of Channel 2. Disabling
Channel 2 stops the switching of the power FETs on
Channel 2 which reduces the VDD current draw. This can
improve efficiency when operating at low output current,
especially when large MOSFETs are used.
VEN1/2
Slope = ISS/CSS
VSS
VOUT
Supply Voltages and Internal References
The MIC2155/6 is powered from a 4.5V to 14.5V supply.
The two input supply pins (VIN1 and VIN2) are
connected together in most applications. They are
powered separately in configurations with two input
supply voltages.
VIN1 supplies an internal LDO, which generates the
VDD supply voltage. VDD is used to power the gate drive
circuitry and must be externally decoupled to the power
ground pins (PGND1 and PGND2). A 10µF Ceramic
capacitor is recommended for most applications. The
AVDD pin is the supply pin for the Bandgap reference
and internal analog circuits. A small RC filter
(10Ω/0.1µF) connected to AVDD is recommended to
help attenuate switching noise from the VDD supply.
The dropout of the internal VDD regulator causes VDD
to drop if VIN1 is below 6V. When operating below 6V,
VDD may be jumpered to VIN1. This bypasses the
internal LDO and prevents VDD from dropping out.
An LDO or simple series pass regulator can be used to
limit the VDD voltage for applications with an input
voltage that spans above and below the 6V maximum
VDD limit. Figures 5 and 6 illustrate two examples of
regulating VDD with external circuitry.
Figure 3. Soft Start Waveforms
Figure 4. Soft Start Circuit
The output voltage starts to rise when VSS is
approximately 1 diode drop above ground, 0.6V.
The startup delay and output voltage risetime can be
approximated using the formula shown below:
Delay
td =
CSS × 0.6V
ISS
Risetime
C × VOUT
tR = SS
14 × ISS
C IN
1µF
C VDD
10µF
C BST
The soft start pin is discharged under the following
conditions:
•
•
•
•
EN1 pin de-asserted
UVLO on the VIN1 or VDD pins
Overcurrent
Overvoltage (latched off)
November 2009
Figure 5. LDO Regulator
13
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Micrel, Inc.
MIC2155/2156
C VDD
10µF
Using an external LDO to supply VDD (as in Figure 5)
can lower power dissipation in the controller and reduce
junction temperature by supplying VDD externally. Using
an external regulator, the power dissipated in the
controller is reduced to:
C BST
PDISS = (5V × (37nC × 4) × 500kHz = 0.37W
Careful selection and temperature rise calculations of
the external LDO should be done to prevent an
excessively high LDO junction temperature.
Figure 6. Emitter Follower Regulator
UVLO
Separate UVLO circuits monitor VIN1, VIN2 and VDD.
Switching on Channel 1 is inhibited until the voltage on
the VIN1 and VDD pins is greater than their respective
UVLO thresholds. The gate drive on Channel 2 is
inhibited until the VIN2 pin voltage exceeds its UVLO
threshold.
Individual UVLO thresholds are necessary to allow
proper operation from separate input supplies. The VIN1
threshold prevents the IC from switching if the input
voltage is too low to properly source the VDD voltage.
The VIN2 UVLO threshold is lower than VIN1 to allow
operation from a low voltage input.
Channel 1 will switch and provide a regulated output
voltage even if the VIN2 UVLO prevents Channel 2 from
switching.
The internal VDD regulator can supply up to 75mA of
current to drive the external MOSFETs. Power
dissipation inside the MIC2155/6 control IC is divided
between power dissipated in the controller’s analog
circuitry and power dissipated in the drive circuitry. Drive
circuitry power is almost always much greater than
analog circuitry power. Total regulator power dissipation
is calculated using the following formula:
PDISS = VIN1 × IIN1 = VIN1 × (fS × Qg + IQ )
Where:
Qg = total gate charge of all MOSFETs
fS = switching frequency of each stage (500kHz for
the MIC2155 and 300kHz for the MIC2156)
IQ = Controller quiescent current (non-switching
supply current)
Power Good
The power good signal asserts high when the output
voltage is greater than the power good threshold. The
power good circuit compares a portion of the reference
voltage to the voltage on the feedback pin. The output is
an open drain FET as shown in Figure 7. To assert high
it must be pulled up to AVDD through a resistor.
In some instances, power dissipation inside the control
IC may limit the controller’s maximum ambient
temperature. For example, if the MIC2155 is powered
from a 12V source and is driving 4 FETs. If each FET
has a Qg=37nC, the total power dissipation in the
MIC2155 is:
AVDD
PG Comparator
PDISS = 12V × (37nC × 4) × 500kHz = 0.888 W
PGOOD
FB1
The maximum operating ambient temperature is:
BandGap-10%
TA(MAX ) = TJ(MAX ) − PDISS × θJC
TA(MAX ) = 125°C − 0.888 W × 50 °C
Figure 7. Power Good
W
The power good signal may be connected to the enable
pin of other power supplies and used to sequence the
other outputs.
TA(MAX ) = 81°C
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MIC2155/2156
with an appropriate VGS threshold should be used in this
situation.
The voltage on the bootstrap capacitor drops each time
it delivers charge to turn on the MOSFET. The voltage
drop depends on the gate charge required by the
MOSFET. Most MOSFET specifications specify gate
charge vs. VGS voltage. Based on this information and a
recommended ΔVHB of less than 0.1V, the minimum
value of bootstrap capacitance is calculated as:
Oscillator and Frequency Synchronization
The internal oscillator free runs at a fixed frequency and
requires no external components. The oscillator
generates two clock signals that are 180° out of phase
with each other. This forces each channel of the
controller to switch 180° out of phase, which reduces
input and output ripple current.
The internal oscillator generates a clock signal and ramp
signal. The clock signal terminates the switching cycle
for each channel. The ramp voltage for Channel 1 is
compared with the output of the error amplifier and
regulates the output voltage. The ramp signal for
Channel 2 is compared with the Channel 2 error
amplifier output and forces the output current of Channel
2 to match Channel 1.
CBST ≥
QGATE
ΔVBST
Where:
QGATE = Total Gate Charge a VBST
ΔVBST = Voltage drop at the BST pin
RMP1
A minimum value of 0.1µF is required for each of the
bootstrap capacitors, regardless of the MOSFETs being
driven. Larger or paralleled MOSFETs may require
larger capacitance values for proper operation.
Placement is critical. The bypass capacitor (CBST) for the
BST supply pins must be located close between the BST
and SW1 pins. The etch connections should be short,
wide and direct. The use of a ground plane to minimize
connection impedance is recommended. Refer to the
section on layout and component placement for more
information.
A delay between the switching of the two MOSFETs is
necessary to prevent both MOSFETs from being on at
the same time and shorting VIN to ground. An adaptive
gate drive in the controller monitors the switch node
(SW1) and low-side driver (LSD1) to minimize dead time
while preventing both MOSFETs from being on at the
same time. This enables the use of a broad range of
MOSFETS without requiring excessive deadtime.
CLK1
SYNC
RMP2
CLK2
2 Phase
Oscillator
Figure 8. Oscillator and Sync Diagram
The SYNC input (pin 15) allows the MIC2155/6 to
synchronize to an external clock signal. When
synchronized, each channel switches at half of the
synchronization
frequency.
Limitations
on
the
synchronization frequency and signal amplitude are
listed in the electrical characteristics section of the spec.
When not used, the sync pin should be left open (no
connect).
MOSFET Gate-Drive Circuitry
The high-side drive circuit is designed to switch an Nchannel MOSFET. Figure 9 shows a diagram of the gate
drive and bootstrap circuit. D2 and CBST comprise the
bootstrap circuit, which is used to supply drive voltage to
the high-side FET. Bootstrap capacitor CBST is charged
through diode D2 while the low-side MOSFET is on and
the voltage on the SW pin is approximately 0V. When
the high-side MOSFET driver is turned on, energy from
CBST is used to charge the MOSFET gate, turning on the
FET. As the MOSFET turns on, the voltage on the SW
pin increases to approximately VIN. Diode D2 is reversed
biased and CBST is pulled high while continuing to keep
the high-side MOSFET on. The high-side drive voltage,
which is derived from VDD, is approximately 4.5V due the
voltage drop across D2. When operating at 4.5VIN,
without connecting VDD to VIN, the gate drive voltage to
the high-side FET could be as low as 3.2V. MOSFETs
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Micrel, Inc.
MIC2155/2156
The following steps can be taken to lower the gate drive
impedance, minimize the dv/dt induced current and
lower the FETs susceptibility to the induced glitch:
1) Chose a MOSFET with:
a) a high CGS/CGD ratio
b) a low internal gate resistance
2) Do not put a resistor between the LSD output
and the gate.
3) Insure both the gate drive and return etch are
short, low inductance connections.
4) Use a 4.5V VGS-rated MOSFET because its
higher gate threshold voltage is more immune to
glitches than a 2.5V or 3.3V rated FET.
5) Connect VDD to VIN or a 5V supply if VIN is below
6V. The RDSON of the internal driver will be lower
and a 4.5V rated MOSFET can be used.
C BST
Figure 9. Gate Drive
dv/dt Induced Turn On of the Low-Side MOSFET
As the high-side MOSFET turns on, the rising dv/dt on
the switch-node forces current through CGD of the lowside FET causing a glitch on the FET’s gate. Figure 10
illustrates the basic mechanism causing this issue. If the
glitch on the gate is greater than the FET’s turn-on
threshold, it may cause an unwanted turn-on of the lowside FET while the high-side FET is on. A short circuit
between input and ground would occur that lowers
efficiency and increases power dissipation in both FETs.
Additionally, turning on the low-side FET during the offtime could interfere with overcurrent sensing.
Remote Sense
Remote sensing provides accurate output voltage
regulation by sensing at the load. Remote sensing
makes up for losses in the power distribution path. It
uses a unity gain differential amplifier to overcome
voltage drops in both the output and return (ground)
paths. The amplifier has common mode input range from
-0.3V to 3.6V. For proper remote sense operation, VIN
must be greater than 6V. If VIN is less than 6V, the VDD
pin must be connected to VIN or externally supplied with
5V.
The output of the remote sense amplifier can source up
to 500µA. The voltage divider resistors (R1 and R4 in
Figure 12) must be chosen to insure the output current
of the amplifier does not exceed the maximum of 500µA.
VDS
R1MAX ≥
VOUT − VREF
500μA
Where:
VREF = 0.7V
A gain/phase plot of the remote sense amplifier in Figure
11 shows a typical 2MHz bandwidth. Phase lag is 45° at
1MHz.
Figure 10. dv/dt Induced Turn-On
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60
200
50
40
160
120
30
20 Phase
10
0
-10 Magnitude
-20
-30
-40
1k
10k
100k
1M
FREQUENCY (Hz)
80
40
0
-40
Setting the Output Voltage
Regardless of whether the remote sensing or local
output voltage sensing is used, the output voltage is set
with voltage divider resistors R1 and R4 (Figure 12). The
equation below is used to calculate VOUT.
PHASE (°)
MAG (dB)
Micrel, Inc.
⎡ R1 ⎤
VOUT = VREF × ⎢1 +
⎥
⎣ R4 ⎦
-80
-120
-160
-200
10M
Where:
VREF = 0.7V
Figure 11. Remote Sense Amplifier Gain/Phase Plot
A typical remote sense configuration is shown in Figure
12. The output of the remote sense amplifier feeds a
voltage divider (R1, R4), which is connected to the
Channel 1 error amplifier. The divider and compensation
network for the remote sense are the same as for a local
sense configuration. The 10Ω resistors provide an
alternate feedback path if the remote sense connections
are removed or opened. The remote sense connections
should not be shorted or the output voltage will increase
close to VIN. The OVP circuit in the controller will not
protect against this type of fault since the feedback pin
voltage will be 0V.
Current Limit and Overcurrent Protection
The MIC2155/6 uses the synchronous (low-side)
MOSFETs RDSON to sense an over current condition. The
low-side MOSFET is used because it displays lower
parasitic oscillations after switching then the upper
MOSFET. Additionally, it improves the accuracy and
reduces false tripping at lower voltage outputs and
narrow duty cycles since the off-time increases as duty
cycle decreases.
MIC2155/6
VIN
HSD
200µA
+SENSE
LOAD
CS1
Q1
–IL × RDSON
RCS
IL
L1
–SENSE
Current Limit
LSD
Q2
IL
CO
Figure 13. Overcurrent Circuit
Inductor current flows from the lower MOSFET source to
the drain during the off-time. The drain voltage becomes
negative with respect to ground as the inductor current
continues to flow from Source to Drain. This negative
voltage is proportional to instantaneous inductor current
times the MOSFET RDSON. The voltage across the lowside FET becomes even more negative as the output
current increases. The overcurrent circuit operates by
passing a known fixed current source (200µA) through a
resistor RCS. This sets up an offset voltage (ICS × RCS)
that is compared to the VDS of the low-side FET. When
ISD (Source to Drain current) × IL is equal to this voltage,
the MIC2155’s over current trigger is set, which disables
the next high side gate drive pulse. After missing the
high side pulse, the overcurrent (OC) trigger is reset. If
on the next low-side drive cycle, the current is still too
high i.e. VCS is ≤ 0V, another high side pulse is missed
and so on. This effectively reduces the overall energy
transferred to the output and VOUT starts to fall.
Figure 12. Remote Sense
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MIC2155/2156
The MIC2155/6 current limit circuit restricts the
maximum output current. If the load tries to draw
additional current the output voltage drops until it is no
longer within regulation limits. At this point (75% of
nominal output voltage) a hiccup current mode is
initiated to protect down stream loads from excessive
current during hard short circuits. This helps reduce the
overall power dissipation in the PWM converter
components during a fault.
Accurate Method
For designs where ripple current is significant when
compared to IOUT or for low duty cycle operation,
calculating the current setting resistor RCS should take
into account that we are sensing the peak inductor
current and that there is a blanking delay of
approximately 100ns.
Figure 15. Overcurrent waveform
The equations to accurately calculate the current limit
resistor value are shown below:
I
I
IPK = OUT + RIPPLE
2
2
VOUT × (1 − D)
IRIPPLE =
FS × L
Figure 14. Overcurrent Sense Waveforms
The MIC2155/6 only senses current across the low side
MOSFET of Channel 1 since both channels operate in
parallel. This means the total output current limit is
approximately twice the calculated current limit.
V
× TDLY
ISET = IPK − OUT
L
ISET × RDSON(MAX)
RCS =
ICS(MIN)
Current Limit Setting
The current limit circuit responds to the peak inductor
current flowing through the low-side FET. The value of
RCS can be estimated with the “simple” method or can be
more accurately calculated by taking the inductor ripple
current into account.
D = Duty Cycle
FS = Switching Frequency
L = Power inductor value
TDLY = Current limit blanking time ~ 100ns
ICS(min) = 180µA
The Simple Method
Current limit can be quickly estimated with the following
equation:
Example:
Consider a 12V to 3.3V @ 30A converter with 1.5µH
power inductor and 90% efficiency at full load. Each
channel will supply 15A at a 500kHz (MIC2155)
switching frequency. The on-resistance of the low side
MOSFET is 6mΩ.
Using the simple method:
RCS = IOUT/2 × RDSON(MAX)/180µA.
Where: RDSON is the maximum on-resistance of the low
side FET at the operating junction temperature
RCS
November 2009
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30 A
× 6mΩ
= 2
= 500Ω
180μA
M9999-111209-B
Micrel, Inc.
MIC2155/2156
Using the accurate method:
D=
The voltage across capacitor C1 is:
VOUT
3 .3
=
= 0 .3
VIN × Efficiency 12 × 0.9
sLo
⎤
⎡
+1 ⎥
⎢
RL
⎥
VS = IL × ⎢RL ×
sC1× R1 + 1⎥
⎢
⎥
⎢
⎦
⎣
3.3 × (1 − 0.3)
= 3.1A
500kHz × 1.5μH
30 3.1
IPK =
+
= 16.55 A
2
2
3.3 × 100ns
ISET = 16.55 −
= 16.33 A
1.5μH
16.33 × 6mΩ
RCS =
= 544Ω
180μA
IRIPPLE =
If the R1 × C1 time constant is equal to the Lo/RL time
constant, the voltage across capacitor C1 equals the RL
× IL. Figure 17 is a plot of this equation and shows the
results graphically. It assumes an inductance of 1.5µH,
RL = 0.01Ω (-40dB), C1=0.1µF and R1=1.5k. The time
constants are equal and diverge at the same rate. The
overall impedance, H(s), equals RL for all frequencies.
Using the simple method here would result in a current
limit point lower than expected.
This equation sets the minimum current limit point of the
converter, but maximum will depend on the actual
inductor value and on resistance of the MOSFET under
current limit conditions. This could be in the region of
50% higher and should be considered to ensure that all
the power components are within their thermal limits
unless thermal protection is implemented separately.
50
GAIN (dB)
40
30
-50
10
H(s)
100
RC
1k
10k 100k
FREQUENCY (Hz)
1M
Figure 17. Current Sense Gain/Phase Plot
Current Sharing
The schematic in Figure 18 illustrates the current sharing
scheme. The error amplifier in Channel 1, E/A 1,
monitors the output voltage and adjusts the duty cycle of
Channel 1 to regulate that voltage. The inputs of
transconductance error amplifier, E/A 2, are connected
to the current sense points of each channel. The error
amplifier regulates Channel 2 current by monitoring the
current sense point of Channel 1 and forcing the current
sense point of Channel 2 to be equal.
Any offset or difference in current between the two
channels is caused by tolerances in the inductance,
DCR, and tolerances of R1, C1, R2 and C2. Additionally,
voltage offset in E/A 2 may cause variations in output
current sharing. At lower currents, these variations may
force the current of Channel 2 to be 0.
A nominal 10mV offset inhibits the Channel 2 low-side
MOSFET until the output current increases to the
magnitude where the voltage across C1 is 10mV. This
prevents the low-side MOSFET of Channel 2 from
sinking current to ground during startup or during low
current operation.
Output Inductor and
Winding Resistance
Q2
Figure 16. Lossless Inductor Current Sense
November 2009
0
-10
-20
-30
-40
Inductor Current Sensing
Current sharing between the two phases is achieved by
sensing the inductor current in each phase. Lossless
inductor current sensing is used, which has the
advantages of lower power loss and lower cost – over
using a discrete resistor in series with the inductor.
The inductor sense circuit is shown in Figure 16. It
extracts the voltage drop across the inductor’s DC
winding resistance.
L/R
20
10
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Micrel, Inc.
MIC2155/2156
VIN
E/A 1
Output1 Inductor and
Winding Resistance
Q1
PWM 1
IL
RL1
L1
Driver
VREF
CO
C1
R1
Q2
+
VS
–
MIC2155/56
10mV
VIN
Q3
E/A 2
PWM 2
Driver
Output2 Inductor and
Winding Resistance
IL
RL2
R2
Q4
L2
C2
Figure 18. Current Sharing Diagram
Startup into a Pre-Biased Output
Soft start circuitry in a conventional synchronous buck
regulator forces the regulator to start up by initially
operating at a minimum duty cycle and gradually
increasing the duty cycle until the output voltage reaches
regulation. In a synchronous buck power supply, a
narrow duty cycle means the low-side MOSFET is on for
most of the switching period. If the output voltage is not
0V, the wide on time of the low-side MOSFET may
discharge the output and cause high reverse current to
flow in the inductor.
The MIC2155/6 is designed to turn on into a pre-biased
output without discharging the output. Circuitry in the
controller monitors the input and output voltage and
forces the soft start circuit to initially operate at the
proper duty cycle. This allows the output to turn on in a
controlled fashion without discharging the output. The
minimum output voltage for proper operation of the prebias startup circuitry is 0.6V. If VOUT is less than 0.6V, a
partial discharge of VOUT may occur.
November 2009
Separate Input Supplies
The MIC2155/6 can operate from two different input
supplies with different voltages. Each of the two
channels can have a different input voltage and still
share current. This allows the supply to draw power from
more than one supply.
The controller will force the output current to be equal.
Since the output voltage and currents of the two
channels are the same, the input power drawn from
each supply will approximately be the same. The input
currents will be inversely proportional to the input
voltages of each supply. For example, if the total output
power is 50W and efficiency is 91%, the total input
power from both supplies is:
PIN =
20
POUT
50
=
= 55 W
η
0. 9
M9999-111209-B
Micrel, Inc.
MIC2155/2156
will also require more output capacitance to smooth out
the larger ripple current. Smaller peak to peak ripple
currents require a larger inductance value and therefore
a larger and more expensive inductor.
Higher switching frequencies allow the use of a small
inductance but increase power dissipation in the inductor
core and MOSFET switching loss. The MIC2155
switches at 500kHz/channel and is designed to use a
smaller inductor at the expense of higher switching
losses and slightly lower efficiency. While the 300kHz
MIC2156 was optimized for higher efficiency and higher
output current but its lower switching frequency requires
a larger output inductance to maintain the same peak-topeak output ripple current.
The peak output ripple current for a two-phase converter
is shown in Figure 19. The graph shows that peak ripple
current is a function of duty cycle. Since each channel is
180° out of phase with the other, at 50% duty cycle, the
output ripple currents from each channel cancel and
output ripple current is close to zero.
Each supply contributes approximately half the power:
PIN1 = PIN2 = 27.5 W
For VIN1 = 12V and VIN2 = 3.3V
IIN1 =
27.5 W
= 2 .3 A
12V
and
IIN2 =
27.5 W
= 8. 3 A
3 .3 V
Component Selection, Guidelines and Design
Example
The following section outlines a procedure for designing
a two-phase synchronous buck converter using the
MIC2155.
This example will use the following parameters:
NORMALIZED OUTPUT RIPPLE CURRENT
VIN = 12V
VOUT = 1.8V
IOUT = 30A
Switching frequency (fS) = 500kHz/channel
(MIC2155)
Output Filter Selection
The output filter is comprised of the output capacitors
and the output inductors. The filter is designed to
attenuate the output voltage ripple to the desired value.
The output filter components also determine how well
the supply responds to output current transients. If
output transients are significant, the output capacitors
should be chosen first to meet the transient specification.
The output inductor is then selected to insure the filter
attenuates the output ripple to meet the specification.
A second, commonly used method of designing the filter
is to select the inductor value to keep the ripple current
between 20% and 30% of the output current for that
channel. Then select the output capacitance to meet the
output voltage ripple specification and output current
transient specification.
Values for inductance, peak and RMS currents are
required to choose the output inductors. The input and
output voltages and the inductance value determine the
peak to peak inductor ripple current. Output capacitor
selection requires calculation of transient current, RMS
capacitor current and output voltage.
There are several tradeoffs to be made when selecting
the output inductor. Generally, higher inductance values
are used with higher input voltages. Larger peak to peak
ripple currents will increase the power dissipation in the
inductor and MOSFETs. Larger output ripple currents
November 2009
1.0
0.9
0.8
0.7
0.6
Single Phase
0.5
0.4
0.3
0.2
0.1
2 Phase
0
0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9
DUTY CYCLE
Figure 19. 2 Phase Output Ripple Current vs. Duty Cycle
For this example, with VIN = 12V, VOUT = 1.8V and
efficiency = 88%, the duty cycle is:
D=
VOUT
1. 8 V
=
= 0.17
η × VIN 0.88 × 12
Figure 19 shows the peak-to-peak output ripple current
normalized to:
VOUT
fS × L OUT
The peak-to-peak output ripple current is less than for a
single phase conversion. If VIN varies, the input voltage
that generated the highest ripple current should be used
for the calculation.
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Micrel, Inc.
MIC2155/2156
For this example, assume the output transient loading is
small and the filter design is based on output ripple
voltage requirement.
The inductance value is calculated by the equation
below:
The peak inductor current in each channel is equal to the
average output current plus one half of the peak to peak
inductor ripple current:
VOUT × (η × VIN(MAX ) − VOUT )
The RMS inductor current is used to calculate the I2 × R
losses in the inductor:
L=
IPK = IOUT + 0.5 × IPP = 15 A + 0.5 × 3 A = 16.5 A
η × VIN(MAX ) × fS × 0.2 × IOUT
Where:
1 ⎛ IPP
⎜
IINDUCTOR(RMS) = IOUT × 1 +
12 ⎜⎝ IOUT
fS is the switching frequency
0.2 is the ratio of AC ripple current to DC output
current
VIN(MAX) is the maximum input voltage
IOUT is output current of the each channel or ½ of the
total output current
η is the converters efficiency
If another inductor value is used, the ripple current for
each channel is calculated from the formula below:
IPP =
VOUT × (η × VIN(MAX ) − VOUT )
η × VIN(MAX) × fS × L
1.8 V × (0.88 × 12V − 1.8 V )
= 3A
0.88 × 12V × 500kHz × 1μH
The output capacitors see less ripple current than each
channel because they are out of phase.
The normalizing factor is:
VOUT
1 .8 V
=
= 3 .6
fS × L OUT 500kHz × 1μH
PINDUCTOR(COPPER) = (IINDUCTOR(RMS))2 x RWINDING =
15.12 x 1.9mΩ = 0.43W
The output ripple current in the two-phase configuration
is approximately:
0.65 ×
1 ⎛ 3A ⎞
⎟ = 15.1A
⎜
12 ⎝ 15 ⎠
Maximizing efficiency requires the proper selection of
core material and minimizing the winding resistance. The
high frequency operation of the MIC2155 requires the
use of ferrite materials for all but the most cost sensitive
applications. Lower cost iron powder cores may be used
but the increase in core loss will reduce the efficiency of
the power supply. This is especially noticeable at low
output power. The inductor winding resistance
decreases efficiency at the higher output current levels.
The winding resistance must be minimized although this
usually comes at the expense of a larger inductor.
The power dissipated in the inductor is equal to the sum
of the core and copper losses. At higher output loads,
the core losses are usually insignificant and can be
ignored. At lower output currents the core losses can be
a significant contributor. Core loss information is usually
available from the magnetics vendor.
For this example a Cooper HCF1305-1R0 inductor was
chosen. Core loss for this application was taken from
the data sheet and is 15mW. Winding resistance is
1.9mΩs
Copper loss in the inductor is calculated by the equation
below:
1.8 V × (0.88 × 12 V − 1.8 V )
= 1μH
0.88 × 12V × 500kHz × 0.2 × 15 A
IPP =
2
2
IINDUCTOR(RMS) = 15 A × 1 +
For this example:
L=
⎞
⎟⎟
⎠
VOUT
1. 8 V
= 0.65 ×
= 2. 3 A
fS × L OUT
500kHz × 1μH
For the input and output voltage in this application, going
to a two-phase design decreased the total output ripple
current from 3APP to 2.3APP.
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The resistance of the copper wire, RWINDING, increases
with temperature. If so desired, a more accurate
calculation can be made if the maximum ambient
temperature and temperature rise of the inductor is
known. The value of the winding resistance at operating
temperature is calculated with the formula below:
For this example, using ΔVOPP = 10mV, the minimum
COUT is:
RWINDING(HOT) = RWINDING(20) x (1 + 0.0042 x
(TempHOT – T20)
A capacitance value this low is usually not used in high
current converters because of transient output current
requirements.
For this example, 500µF total capacitance is used. It is
split up into (4) 47µF ceramic capacitors and (2) 150µF
Aluminum Polymer capacitors
The total output ripple is a combination of the ESR and
the output capacitance. The total ripple is calculated
below:
COUT ≥
Where:
TempHOT is the temperature of the wire under operating
load
T20 is the ambient temperature
RWINDING(20) is the resistance of the winding at room
temperature, usually specified by the manufacturer.
2 .3
= 29μF
8 × 10mV × 2 × 500kHz
2
For this example, the approximate power dissipation is
0.43W. From the manufacturers data sheet this causes a
20°C rise in inductor temperature. Assuming ambient
temperature stayed at 20°C, the maximum winding
resistance would be increased from 1.9mΩs to:
ΔVOUT
To increase reliability, the recommended voltage rating
of capacitor should be twice the output voltage for a
tantalum and 20% greater for an aluminum electrolytic or
ceramic.
The output capacitor RMS current is calculated below:
R WINDING(HOT ) = 1.9mΩ × (1 + 0.0042 × ( 40°C − 20°C) = 2.06mΩ
Output Capacitor Selection
In this example, the output capacitors are chosen to
keep the output voltage ripple below a specified value.
The output ripple voltage is determined by the capacitors
ESR (equivalent series resistance) and capacitance.
Voltage rating and RMS current capability are two other
important factors in selecting the output capacitor.
Ceramic output capacitors and most polymer capacitors
have very low ESR and are recommended for use with
the MIC2155/6. The output capacitance is usually the
primary cause of output ripple in ceramic and very low
ESR capacitors. The minimum value of COUT is
calculated below:
COUT ≥
⎡
⎤
IPP
2
= ⎢
⎥ + [IPP × RESR ]
⎣ 8 × COUT × 2 × fS ⎦
I
2. 3 A
ICOUT(RMS) = PP =
= 0.66 A
12
12
The power dissipated in the output capacitors can be
calculated by the equation below:
(
)
PDISS(COUT ) = ICOUT(RMS) 2 × RESR
IPP
8 × ΔVOPP × 2 × fS
Where:
ΔVOPP is the peak-to-peak output voltage ripple
IPP is the peak-to-peak ripple current as see by the
capacitors
fS is the per channel switching frequency
Notice the calculation is performed at 2x the switching
frequency since the capacitors see ripple current from
both phases.
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The power dissipated in the input capacitor is:
Inductor Current Sense Components
The RC circuit values that sense current across the
inductor can be calculated once the inductor is selected.
The circuit is shown in Figure 20.
(
)
MORMALIZED RMS INPUT CURRENT
PDISS(CIN) = ICIN(RMS) 2 × RESR
Output Inductor and
Winding Resistance
Q2
Figure 20. Inductor Current Sense
0.6
Single Phase
0.5
0.4
0.3
2 Phase
0.2
0.1
0
0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9
DUTY CYCLE
Figure 21. RMS Input Current vs. Duty Cycle
The inductor has the following values:
MOSFET Selection
External N-channel logic level power MOSFETs must be
used for the high and low side switches. The MOSFET
gate to source drive voltage of the MIC2155 is regulated
by an internal 5V VDD regulator. Logic level MOSFETs,
whose operation is specified at VGS = 4.5V must be
used. This resistance is used to calculate the losses
during the MOSFET’s conduction time. If operating at
4.5VIN, without connecting VDD to VIN, the gate drive
voltage to the high-side FET could be as low as 3.2V.
MOSFETs with low VGS enhanced gates should be used
in this situation.
It is important to note the on-resistance of a MOSFET
increases at high junction temperature. A 75°C rise in
junction temperature will increase the channel resistance
of the MOSFET by 40% to 75% of the resistance
specified at 25°C. This change in resistance must be
accounted for when calculating MOSFET power
dissipation.
Total gate charge is the charge required to turn the
MOSFET on and off under specified operating conditions
(VDS and VGS). The gate charge is supplied by the
MIC2155 gate drive circuit. Gate charge can be a
significant source of power dissipation in the controller
due to the high switching frequencies and generally
large MOSFETs that are driven. At low output load this
power dissipation is noticeable as a reduction in
efficiency.
L = 1.0µH, R L= 1.9mΩ
Proper sensing of the DC voltage across the inductor
requires the RL/L time constant be equal to the R1×C1
time constant:
L
= C1× R1
RL
A good range of values for C1 is 0.1µF to 1µF. For this
example C1 is chosen as 0.22µF. R1 is:
R1 =
L
1μH
=
= 2.39k
RL × C1 1.9mΩ × 0.22μF
Input Capacitor Selection
In addition to high-frequency ceramic capacitors, a larger
bulk capacitance, either ceramic or Al. El. should be
used to help attenuate ripple on the input and to supply
current to the input during large output current
transients. The input capacitors must be rated for the
RMS input current of the power supply. RMS input
capacitor current is determined at the maximum output
current. The graph in Figure 21 shows the normalized
RMS input ripple current vs. duty cycle. Data is
normalized to the output current.
For a two-phase converter operating at 17% duty cycle,
the input RMS current is determined from the graph:
ICIN _ RMS ≈ IOUT × 0.24 = 7.2A
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Making the assumption the turn-on and turn-off transition
times are equal, the total AC switching loss is:
The average current required to drive the MOSFETs is:
IDD = Qg × fs
PAC = ( VIN +VD ) × ISW _ PEAK × Tt × fS
Where:
Where:
Qg is the total gate charge for all high and low side
MOSFETs. This information should be obtained from
the manufacturer’s data sheet with a 5V VGS.
Since the current from the gate drive comes from the
input voltage, the power dissipated in the MIC2155 due
to gate drive is:
Tt is the switching transition time (typically 15ns to
30ns)
fS it the switching frequency of each phase
RMS Current and MOSFET Power Dissipation
Calculation
Under normal operation, the high side MOSFET’s RMS
current is greatest when VIN is low (maximum duty
cycle). The low side MOSFET’s RMS current is greatest
when VIN is high (minimum duty cycle). However, the
MOSFET sees maximum stress during short circuit
conditions, where the output current is equal to the
maximum overcurrent level. The calculations below are
for normal operation. To calculate the stress under short
circuit conditions, substitute the maximum overcurrent
level for IOUT(max).
The RMS value of the high side switch current is:
PGATE _ DRIVE = Qg × fS × VIN
A convenient figure of merit for switching MOSFETs is
the on-resistance times the total gate charge (RDSON ×
Qg). Lower numbers translate into higher efficiency. Low
gate charge, logic level MOSFETs are a good choice for
use with the MIC2155. The internal LDO that supplies
VDD is rated for 75mA. Exceeding this value could
damage the regulator or cause excessive power
dissipation in the IC. Refer to the “Supply Voltages and
Internal Regulator” section of this specification for
additional information.
Parameters that are important to MOSFET switch
selection are:
I 2
ISW _ RMS(HIGH _ SIDE ) = D (IOUT(max)2 + PP )
12
• Voltage rating
• On resistance
• Total Gate Charge
The VDS voltage rating of the MOSFETs is essentially
equal to the input voltage. A safety factor of 20% should
be added to the VDS(max) of the MOSFETs to account for
voltage spikes due to circuit parasitics.
The power dissipated in the switching transistor is the
sum of the conduction losses during the on-time
(PCONDUCTION) and the switching losses that occur during
the period of time when the MOSFETs turn on and off
(PAC).
I 2
ISW _ RMS(LOW _ SIDE ) = (1 − D) (IOUT(max)2 + PP )
12
Where:
D is the duty cycle of the converter
IPP is the individual inductor ripple current
V
D = OUT
η × VIN
and η is the efficiency of the converter.
Converter efficiency also depends on other component
parameters that have not yet been selected. For design
purposes, an efficiency estimate of 85% – 90% can be
used. The efficiency can be more accurately calculated
once the design is complete. If the assumed efficiency is
grossly inaccurate, a second iteration through the design
procedure should be made.
For the high-side switch, the maximum DC power
dissipation is:
PSW = PCONDUCTION + PAC
Where:
PCONDUCTION = ISWITCH(rms )2 × RSWITCH
PAC = PAC(off ) + PAC(on)
RSWITCH is the on resistance of the MOSFET switch.
(
)
PSWITCH1(DC) = RDSON1 × ISW1(rms ) 2
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This power dissipation is calculated below:
For the low-side switch, the DC power dissipation is:
(
)
PSWITCH 2(DC) = RDSON2 × ISW 2(rms ) 2
ID(ave ) = IOUT × 2 × t d × fS
The switching loss for each of the high-side MOSFETs
is:
Where:
td is the dead time when both MOSFETs are off.
The reverse voltage requirement of the diode is:
PAC = VIN × ISW (peak ) × Tt × fS
VDIODE _ RRM = VIN
The total power dissipation for each MOSFET is:
The power dissipated by the diode is:
PFET _ total = PSWITCH1(DC) + PAC
External Schottky Diode
A freewheeling diode in parallel with the low-side FET is
needed to keep the inductor current flow continuous
while both MOSFETs are turned off (dead time). Dead
time is necessary to prevent current from flowing
unimpeded through both MOSFETs. An external
Schottky diode is not necessary for circuit operation
since the low-side MOSFET contains a parasitic body
diode. An external diode will improve efficiency due to its
lower forward voltage drop as compared to the internal
parasitic diode in the FET. It may also decrease high
frequency noise because the schottky diode junction
does not suffer from reverse recovery.
If the MOSFET body diode is used, it must be rated to
handle the peak and average current. The body diode
may have a relatively slow reverse recovery time and a
relatively high forward voltage drop. The power lost in
the diode is proportional to the forward voltage drop of
the diode. As the high-side MOSFET starts to turn on,
the body diode becomes a short circuit for the reverse
recovery period, dissipating additional power. The diode
recovery and the circuit inductance will cause ringing
during the high-side MOSFET turn on. If the internal FET
diode is used, power dissipated during the dead time
should be added to the PDISS of the low-side MOSFET.
An external Schottky diode conducts at a lower forward
voltage preventing the body diode in the MOSFET from
turning on. The lower forward voltage drop dissipates
less power than the body diode. The lack of a reverse
recovery mechanism in a Schottky diode causes is less
ringing and power loss. Depending on the circuit
components and operating conditions, an external
Schottky diode may give a ½% to 1% improvement in
efficiency.
PDIODE = ID _ AVE × VF
Where:
VF is the forward voltage at the peak diode current.
Snubber Design
A snubber is used to damp out high frequency ringing
caused by parasitic inductance and capacitance in the
buck converter circuit. A snubber is needed for each of
the two phases in the converter. Figure 22 shows a
simplified schematic of one of the buck converter
phases. Stray capacitance consists mostly of the two
MOSFET’s output capacitance (COSS). The stray
inductance is mostly package and etch inductance. The
arrows show the resonant current path when the high
side MOSFET turns on. This ringing causes stress on
the semiconductors in the circuit as well as increased
EMI.
COSS1
+
LSTRAY1
LSTRAY2
L
Q1
CIN
VDC
LSTRAY3
Sync_buck
Controller
Q2
COSS2
COUT
LSTRAY4
–
Figure 22. Output Parasitics
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One method of reducing the ringing is to use a resistor
and capacitor to lower the Q of the resonant circuit. The
circuit in Figure 23 shows the resistor in between the
switch node and ground. Capacitor Cs is used to block
DC and minimize the power dissipation in the resistor.
This capacitor value should be between 5 and 10 times
the parasitic capacitance of the MOSFET COSS. A
capacitor that is too small will have high impedance and
prevent the resistor from damping the ringing. A
capacitor that is too large causes unnecessary power
dissipation in the resistor, which lowers efficiency.
The snubber components should be placed as close as
possible to the low-side MOSFET and/or external
schottky diode since in contributes to most of the stray
capacitance. Placing the snubber too far from the FET or
using etch that is too long or thin will add inductance to
the snubber and diminishes its effectiveness.
A proper snubber design requires the parasitic
inductance and capacitance be known. A method of
determining these values and calculating the damping
resistor value is outlined below.
1. Measure the ringing frequency at the switch node
which is determined by parasitic LP and CP. Define this
frequency as f1.
2. Add a capacitor CS (normally at least 3 times as big as
the COSS of the FET) from the switch node to ground and
measure the new ringing frequency. Define this new
(lower) frequency as f2. LP and CP can now be solved
using the values of f1, f2 and CS.
Define f’ as:
f
f' = 1
f2
Combining the equations for f1, f2 and f’ to derive CP, the
parasitic capacitance:
CP =
LP =
Q=
× CP × ( f1)2
1
RS
LP
=1
CS
Solving for RS
RS =
LP
CS
Figure 23 shows the snubber in the circuit and the
damped switch node waveform.
1
2π LP × CP
LSTRAY1
Where:
CP and LP are the parasitic capacitance and
inductance
LSTRAY2
RDS
LSTRAY3
Step 2: Add a capacitor, CS, in parallel with the
synchronous MOSFET, Q2. The capacitor value should
be approximately 3 times the COSS of Q2. Measure the
frequency of the switch node ringing, f2:
November 2009
(2π)
1
2
Step 3: Calculate the damping resistor.
Critical damping occurs at Q = 1:
damping.
Step 1: First measure the ringing frequency on the
switch node voltage when the high-side MOSFET turns
on. This ringing is characterized by the equation:
f2 =
2 × ( f ' )2 − 1
LP is solved by re-arranging the equation for f1:
3. Add a resistor RS in series with C S to generate critical
f1 =
CS
RS
COSS2
CS
LSTRAY4
1
2π Lp × (Cs + Cp)
Figure 23. Snubber Circuit
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Compensation is necessary to insure the control loop
has adequate bandwidth and phase margin to properly
respond to input voltage and output current transients.
High gain at DC and low frequencies is needed for
accurate output voltage regulation. Attenuation near the
switching frequency prevents switching frequency noise
from interfering with the control loop.
For this analysis, the LC filter in the two phase design is
combined into one. The inductances are in parallel and
the output capacitance is the total sum of all COUT. The
ESR is the parallel combination of all ESRs. The output
load is represented by a resistor R.
The output filter contains a complex double pole formed
by the capacitor and inductor and a zero from the output
capacitor and it’s ESR. The transfer function of the filter
is:
The snubber capacitor, CS, is charged and discharged
each switching cycle. The energy stored in CS is
dissipated by the snubber resistor, RS, two times per
switching period. This power is calculated in the
equation below:
PSNUBBER = fS × CS × VIN2
Where:
fS is the switching frequency for each phase
VIN is the DC input voltage
Compensation of the Output Voltage Loop
The voltage regulation, filter and power stage section is
shown in Figure 24. The error amplifier for Channel 1 is
used to regulate the output voltage and compensate the
voltage regulation loop. It is a voltage output op amp that
is designed to use type III (PID) compensation. Type III
compensation has two compensating zeros, two poles
and a pole at the origin. The figure also shows the
transfer function for each section.
1+
Gfilter(s) =
1+
s
ωz
s
⎛ s ⎞
+⎜
⎟
2 × Q × ω o ⎝ ωo ⎠
2
Where:
ωz =
ωo =
1
CO × Re sr
1
CO × L O
Q = R×
CO
L
C1
E/A
R2
C2
C3
R3
VIN
Modulator
Q1
R1
Filter
L1 || L2
Driver
CO
R4
VREF = 0.7V
Q2
RESR
R
Figure 24. Voltage Loop and Transfer Functions
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The Modulator Gain is proportional to the input voltage
and inversely proportional to the internal ramp voltage
generated by the oscillator. The MIC2155/6 peak-peak
ramp voltage is 1V:
As the frequency approaches the zero frequency (Fz),
formed by Co and it’s ESR, the slope of the gain curve
changes from -40db/dec. to -20db/dec and the phase
increases. The zero causes a 90° phase boost. Ceramic
capacitors, with their smaller values of capacitance and
ESR, push the zero and its phase boost out to higher
frequencies, which allow the phase lag from the LC filter
to drop closer to -180°. The system will be close to being
unstable if the overall open loop gain crosses 0dB while
the phase is close to -180°. If the output capacitance
and/or ESR is high, the zero moves lower in frequency
and helps to boost the phase, leading to a more stable
system.
The error amplifier is a type III which has two zeros, two
poles and a pole at the origin. This type of error amplifier
works well when Ceramic output capacitors make up the
majority of COUT because it introduces an extra zero that
helps improve phase margin:
⎛ VIN ⎞
⎟⎟
GMOD = ⎜⎜
⎝ VRAMP ⎠
The output voltage divider attenuates VOUT and feeds it
back to the error amplifier. The divider gain is:
H=
R4
V
= REF
R1 + R4 VOUT
The modulator, filter and voltage divider gains can be
multiplied together to show the open loop gain of these
parts:
s ⎞⎛
s ⎞
⎛
⎜1 +
⎟⎜1 +
⎟
ω
z
1
ω
z2 ⎠
⎝
⎠
⎝
Gea(s) =
s ⎛
s ⎞⎛
s ⎞
⎜1 +
⎟⎜1 +
⎟
ωpo ⎜⎝
ωp1 ⎟⎠⎜⎝
ωp2 ⎟⎠
G VD (s) = GFILTER (s) × H × GMOD
This transfer function is plotted in Figure 25. At low
frequency, the transfer function gain equals the
modulator gain times the voltage divider gain. As the
frequency increases toward the LC filter resonant
frequency, the gain starts to peak. The increase in the
gain’s amplitude equals Q. Just above the resonant
frequency, the gain drops at a -40db/decade rate. The
phase quickly drops from 0° to almost 180° before the
phase boost of the zero brings it back up to -90°. Higher
values of Q will cause the phase to drop quickly. In a
well damped, low Q system the phase will change more
slowly.
Where:
1
R2 × C2
1
ωz 2 =
(R1 + R3) ⋅ C3
1
ωpo =
R1× C1
1
ωp1 =
C1× C2
⋅ R2
C1 + C2
1
ωp 2 =
R3 × C3
ωz1 =
90
GAIN/PHASE
45
0
Gain
Figure 26 shows the bode plot of the error amplifier
transfer function.
-45
Phase
-90
-135
-180
10
100
1k
10k 100k
FREQUENCY
1M
Figure 25. GVD Transfer Function
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Z2
1
Z0 =
2 ⋅ π ⋅ R1 ⋅ C 2
1
2 . π . ( R1
P1
R3 ) . C3
Step 3: Determine the gain boost needed at the
crossover frequency (fc)
Typically, 50° of phase margin can be used for most
applications. This is a good tradeoff between an
overdamped system (slower response to transients) and
an underdamped system (overshoot or unstable
response to transients). It also allows some margin for
component tolerances and variations due to ambient
temperature changes. The phase margin at the
crossover frequency (fc) can be determined by plotting
the GVD(s) phase on a bode plot or can be estimated
with the following formula:
1
C2 . C1 .
R2
2.π .
C2 C1
R2
20. log
R1
Z1
1
2 . π . R2 . C2
P2
1
2 . π . R3 . C3
ϕM
⎡
⎤
fc
⎢
⎥
⎡ fc ⎤
−1 ⎢ Q × fo ⎥
= tan ⎢
+ tan −1 ⎢ ⎥
2⎥
⎣ fz ⎦
⎢1 − ⎛⎜ fc ⎞⎟ ⎥
⎢⎣ ⎝ fo ⎠ ⎥⎦
The additional phase boost required from the error
amplifier is:
Figure 26. Type III Error Amplifier Gain/Phase
Error Amplifier Design Procedure
ϕBoost = 52° − ϕM
Step 1: Decide on the crossover frequency
To maximize transient response, the open loop
bandwidth should be made reasonably high. Initially, the
bandwidth can be selected to be 1/10 of the output
switching frequency. This may be improved once the
design is built and measurements are made. An initial
bandwidth of 100kHz for the 2155 and 60kHz for the
2156 are good choices.
Step 4: Determine the frequencies fz2 and fp1
The frequencies for the zero and pole (fz2 and fp1) are
calculated for the desired amount of phase boost at the
crossover frequency (fc):
Step 2: Determine the gain required at the crossover
frequency
GBoost is how much gain boost is needed so the open
loop transfer function crosses 0dB at the pre-determined
crossover frequency. This can be measured by plotting
the GVD(s) transfer function or can be estimated with the
following formula:
GBoost =
1 − sin[ϕBoost ]
1 + sin[ϕBoost ]
fp1 = fc ×
1 + sin[ϕBoost ]
1 − sin[ϕBoost ]
Step 5: Determine the frequency for fz1
The low-frequency zero, fz1, is initially set to one-fifth of
the LC resonant frequency. If it is set too low, it will force
the low frequency gain to be low and impact transient
response. If set too high, it will not add enough phase
boost at the LC resonant frequency. This could cause
conditional stability, which when the phase drops below 180° before the gain crosses 0dB. If the DC gain should
drop in this situation, this may lead to an unstable
system.
1
2
H × VIN ⎛ fo ⎞ ⎛ fc ⎞
×⎜ ⎟ ×⎜ ⎟
VM
⎝ fc ⎠ ⎝ fz ⎠
Where: fo = LC filter resonant frequency
fc= open loop bandwidth chosen in Step 1
fz = zero formed by COUT and its ESR
H = voltage divider attenuation
VM=amplitude of the internal sawtooth ramp (VM=1)
VIN= Input voltage to the power supply
November 2009
fz2 = fc ×
fz2 =
30
fo
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MIC2155/2156
Based on the amount of gain necessary at the crossover
frequency the mid-band gain and R2 value is calculated
using the following formula:
Step 6: Determine the frequency for fp2
This is the high frequency pole, which is useful in
additional attenuation of the switching frequency. It
should initially be set at half of the switching frequency. If
it is set too low, it will lower the phase margin at the
crossover frequency, making it difficult to achieve the
proper phase margin. If set too high, it will not provide
attenuation of the switching frequency, which could lead
to jitter of the switching waveform or instability under
certain conditions.
fp2 =
2
Vm
fz2
⎛ fo ⎞ ⎛ fc ⎞
×⎜ ⎟ ×⎜ ⎟×
H × Vin ⎝ fc ⎠ ⎝ fz ⎠
fp1
R2 = R1× GCO
GCO =
The other component values are calculated as follows:
fs
2
Calculating Error Amplifier Component Values
Once the pole and zero frequencies have been fixed, the
error amplifier’s resistor and capacitor values are
calculated.
R1: This value is chosen first. All other component
values are calculated from R1. A value of 10K is
suggested. If R1 is chosen too high, R2 may be very
large and the high impedances could be sensitive to
noise. If the remote sense amplifier is used, R1 must be
large enough so than not more than 500µA of current is
drawn from the amplifier.
R2: The value of R2 is determined from the mid-band
gain of the error amplifier. This gain depends on the
frequencies of the poles, zeros and LC filter resonant
frequency.
1
2 × π × fz1× R2
C3 =
1
2 × π × fz2 × R1
C3 =
C2
2 × π × ( fp1× C2 × R2 − 1)
1
2 × π × fp2 × C3
R3 =
Compensation of the Current Sharing Loop
The control circuitry for Channel 2 forces the channel’s
output current to match the current in Channel 1. The
Channel 2 error amplifier compares the inductor currents
in the two channels and adjusts the duty cycle of
Channel 2 to control its output current. A block diagram
is shown in Figure 27.
Filter
VIN
Modulator
Channel 2
Transconductance E/A
C2 =
C2
R2
Q1
CZ1
Ramp
L2
RL2
Driver
RZ1
Q2
Channel 1 IL
RL1
R1
L1
C1
CO
Figure 27. Current Sharing Loop and Transfer Functions
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Unlike the voltage output amplifier used for Channel 1
compensation, a transconductance amplifier is used for
the Channel 2 compensation since only a pole/zero
combination is required for compensation. The
transconductance amplifier transfer function is:
Gea(s) = gm ×
The loop is inherently stable because the phase shift is
only 90 degrees. The error amplifier pole and zero is
selected to achieve a desired crossover frequency. In
this example, the desired crossover frequency is 50kHz.
The transfer function of the filter, modulator and
feedback is plotted in Figure 28.
1 + s × Rz1× Cz1
s × Cz1
50
GAIN/PHASE
Where:
Rz1 and Cz1 are the external components connected to
the COMP2 pin
gm is the transconductance of the internal amplifier.
The pole and zero frequencies are:
0
Gain
-50
Phase
-100
10
gm
fPOLE =
2 × π × Cz1
1
fZERO =
2 × π × Rz1× Cz1
100
1k
10k 100k
FREQUENCY
1M
Figure 28. Current Sharing Loop Gain/Phase
The gain boost required at 50kHz is 28dB which is a
gain of 25. The gain for frequencies above the zero is:
The gain of the modulator is:
gMOD =
VIN = 12V
VOUT = 1.8V
L = 1.5µH
GMID = Rz1× gm
1
VM
For a typical gm = 1.25mS, solving for Rz1:
Rz1 =
Where:
VM is the peak-to-peak amplitude of the internal
sawtooth.
The gain of the feedback circuit is output current divide
by VEA
GMID
25
=
= 20kΩ
gm
1.25mS
Set the zero frequency to be 1/5 of the crossover
frequency:
H = RL 2
1
2 × π × Rz1× fz1
1
Cz1 =
= 800pF
2 × π × 20k × 10k
The filter transfer function is the output current over the
applied voltage:
Cz1 =
V − VOUT
GFILTER( s) = IN
s × L2
The compensated open loop gain/phase plot is shown in
Figure 29.
The open loop transfer function is:
GOL 2( s) = GEA ( s) × GMOD × H × GFILTER( s) =
gm × (1 + s × Rz1× Cz1) × RL 2 × (VIN − VOUT )
(s × Cz1) × Vm × (s × L2)
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200
fZERO = 10kHz
fC = 50kHz
Phase margin = 80°
134.17466
bb(f)
Gain
-100
-179.9424
-200
10
Cin
0
HSFET
GAIN/PHASE
100
Phase
100
1k
10k 100k
FREQUENCY
1M
Figure 29. Compensated Current Sharing Loop Gain/Phase
Figure 31. Layout
General Layout and Component Placement
There are three basic types of currents in a switching
power supply – high di/dt, moderate di/dt and DC.
Examples of each are shown in Figure 30.
Moderate di/dt currents flow in the inductor and output
capacitor. Although layout is not as critical, it is still
important to minimize inductance by using short, wide
traces and a ground plane. Figure 31 shows the etch
connecting the inductor to the output is shaped to force
current to flow past the output capacitor before reaching
the output terminal (or output load). This minimizes the
series inductance between the inductor and the
capacitor, which improves the ability of the capacitor to
filter ripple. Additionally, the inductor current has a large
DC component and requires a wide trace to minimize
voltage drop and power dissipation.
DC currents in a high-current buck converter require
wide etch paths to minimize voltage drop and power
dissipation. The input and output current are mainly DC.
At or near maximum output power, the inductor current
is also predominately DC and requires ample etch to
reduce copper loss, reduce temperature rise and
improve efficiency. Minimizing voltage drops in the
output and ground path helps improve output voltage
regulation for configurations without remote voltage
sensing.
The gate drive connections to both the high-side and
low-side MOSFETs must each have their own return
current path. The high-side MOSFET’s source is
connected to the switch node and returns back to the
controller’s SW1 or SW2 pin. The high-side gate drive
and return (switch node) traces should be routed on top
of each other on adjacent layers to minimize inductance.
These traces swing between VIN and ground and should
be routed away from low-voltage and noise-sensitive
analog etch or components. The low-side MOSFET
return path is power ground. High di/dt currents flow in
the low-side gate drive and return paths. These must be
kept away from noise sensitive signal traces and signal
ground planes.
Figure 30. Current Diagram
In a buck converter, high di/dt currents in the 0.5A/ns
range are generated by MOSFETs switching on and off.
These fast switching currents flow in the high and lowside MOSFETs, external freewheeling schottky diode
and the input capacitor. Fast switching currents also flow
in the gate drive and return etch between the controller
and the power FETs. At that switching speed at 10nH
piece of etch generates 5V across itself. Therefore,
attention to proper layout techniques is essential. Traces
that have high di/dt currents must be kept short and
wide. Additionally a power ground plane should be used
on an adjacent layer to help minimize etch inductance.
Figure 31 shows a layout example that minimizes
inductance.
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• These analog signals should be referenced or
decoupled to the analog ground plane:
AVDD, SYNC, EN, SS, PGOOD, COMP1, COMP2,
FB2, EA2, VOUT, FB1, AGND
• Place the current sharing RC components (that
connect across the inductor) and any related filtering
components close to the FB2, EA2+ and VOUT pins
(18, 17, 20). The traces connecting the inductors and
these components should be routed close together to
minimize pickup or EMI radiation.
• Place the overcurrent sense resistor close to the CS1
pin (pin 4). The trace coming from the switch node to
this resistor has high dv/dt and should be routed away
from other noise sensitive components and traces.
• The remote sense traces must be routed close
together or on adjacent layers to minimize noise
pickup. The traces should be routed away from the
switch node, inductors, MOSFETs and other high
dv/dt or di/dt sources.
Ceramic capacitors are recommended for most
decoupling and filtering applications because of their low
impedance and small size. Depending on the
application, most dielectrics (X5R, X7R, NPO) are
acceptable, however, Z5U type ceramic capacitor
dielectrics are not recommended due to their large
change in capacitance over temperature and voltage.
Design and Layout Checklist
• Ceramic capacitor placed between the HS FET drain
and the LSFET source.
• MOSFET gate drive traces must be low inductance
and routed away from noise sensitive analog signals,
components and ground planes.
• The signal and power ground planes must be
separated to prevent high current and fast switching
signals from interfering with the low level, noise
sensitive analog signals. These planes should be
connected at only 1 point, next to the MIC2155/6
controller.
• The following signals and their components should be
decoupled or referenced to the power ground plane:
VIN1, VIN2, VDD, PGND1, PGND2
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Package Information
32-Pin MLF® (ML)
MICREL, INC. 2180 FORTUNE DRIVE SAN JOSE, CA 95131 USA
TEL +1 (408) 944-0800 FAX +1 (408) 474-1000 WEB http://www.micrel.com
The information furnished by Micrel in this data sheet is believed to be accurate and reliable. However, no responsibility is assumed by Micrel for its
use. Micrel reserves the right to change circuitry and specifications at any time without notification to the customer.
Micrel Products are not designed or authorized for use as components in life support appliances, devices or systems where malfunction of a product
can reasonably be expected to result in personal injury. Life support devices or systems are devices or systems that (a) are intended for surgical implant
into the body or (b) support or sustain life, and whose failure to perform can be reasonably expected to result in a significant injury to the user. A
Purchaser’s use or sale of Micrel Products for use in life support appliances, devices or systems is a Purchaser’s own risk and Purchaser agrees to fully
indemnify Micrel for any damages resulting from such use or sale.
© 2009 Micrel, Incorporated.
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