MICREL MIC2168ABMM

MIC2168A
1MHz PWM Synchronous Buck Control IC
General Description
Features
The MIC2168A is a high-efficiency, simple to use 1MHz PWM
synchronous buck control IC housed in a small MSOP-10
package. The MIC2168A allows compact DC/DC solutions
with a minimal external component count and cost.
The MIC2168A operates from a 3V to 14.5V input, without
the need of any additional bias voltage. The output voltage
can be precisely regulated down to 0.8V. The adaptive all
N-Channel MOSFET drive scheme allows efficiencies over
95% across a wide load range.
The MIC2168A senses current across the high-side NChannel MOSFET, eliminating the need for an expensive
and lossy current-sense resistor. Current limit accuracy is
maintained by a positive temperature coefficient that tracks
the increasing RDS(ON) of the external MOSFET. Further cost
and space are saved by the internal in-rush-current limiting
digital soft-start.
The MIC2168A is available in a 10-pin MSOP package, with
a wide junction operating range of –40°C to +125°C.
All support documentation can be found on Micrel’s web site
at www.micrel.com.
•
•
•
•
•
•
•
•
•
•
•
•
•
•
3V to 14.5V input voltage range
Adjustable output voltage down to 0.8V
Up to 95% efficiency
1MHz PWM operation
Adjustable current-limit senses high-side N-Channel
MOSFET current
No external current sense resistor
Adaptive gate drive increases efficiency
Ultra-fast response with hysteretic transient recovery
mode
Overvoltage protection protects the load in fault
conditions
Dual mode current limit speeds up recovery time
Hiccup mode short-circuit protection
Internal soft-start
Dual function COMP and EN pin allows low-power shutdown
Small size MSOP 10-lead package
Applications
•
•
•
•
•
•
•
Point-of-load DC/DC conversion
Set-top boxes
Graphic cards
LCD power supplies
Telecom power supplies
Networking power supplies
Cable modems and routers
Typical Application
VIN = 5V
SD103BWS
100µF
0.1µF
VDD
VIN
100nF
100pF
4kΩ
BST
CS
HSD
MIC2168A
VSW
COMP/EN
GND
LSD
85
80
1kΩ
2Ω
1000nF
MIC2168 Efficiency
100
95
90
4.7µF
IRF7821
1.2µH
75
3.3V
10kΩ
IRF7821
3.24kΩ
V IN = 5V
V OUT = 3.3V
60
150µF x 2
FB
70
65
55
50
0
2
4
6
I LOAD (A)
8
10
MIC2168A Adjustable Output 1MHz Converter
Micrel, Inc. • 2180 Fortune Drive • San Jose, CA 95131 • USA • tel + 1 (408) 944-0800 • fax + 1 (408) 474-1000 • http://www.micrel.com
April 2005
1
M9999-062205
MIC2168A
Micrel, Inc.
Ordering Information
Part Number
Standard
Pb-Free
MIC2168ABMM
MIC2168AYMM
Frequency
Junction Temp. Range
Package
1MHz
-40°C to +125°C
10-Lead MSOP
Pin Configuration
VIN 1
10 BST
VDD 2
9 HSD
CS 3
8 VSW
COMP/EN 4
7 LSD
FB 5
6 GND
10-Pin MSOP (MM)
Pin Description
Pin Number
Pin Name
Pin Function
1
VIN
Supply Voltage (Input): 3V to 14.5V.
2
VDD
5V Internal Linear Regulator (Output): VDD is the external MOSFET gate
drive supply voltage and an internal supply bus for the IC. When VIN is <5V,
this regulator operates in dropout mode.
3
CS
Current Sense / Enable (Input): Current-limit comparator noninverting input.
The current limit is sensed across the MOSFET during the ON time. The current can be set by the resistor in series with the CS pin.
4
COMP/EN
5
FB
6
GND
Ground (Return).
7
LSD
Low-Side Drive (Output): High-current driver output for external synchronous MOSFET.
8
VSW
Switch (Return): High-side MOSFET driver return.
9
HSD
10
BST
High-Side Drive (Output): High-current output-driver for the high-side
MOSFET. When VIN is between 3.0V to 5V, 2.5V threshold-rated MOSFETs
should be used. At VIN > 5V, 5V threshold MOSFETs should be used.
M9999-062205
Compensation (Input): Dual function pin. Pin for external compensation.
If this pin is pulled below 0.2V, with the reference fully up the device shuts
down (50µA typical current draw).
Feedback (Input): Input to error amplifier. Regulates error amplifier to 0.8V.
Boost (Input): Provides the drive voltage for the high-side MOSFET driver.
The gate-drive voltage is higher than the source voltage by VIN minus a
diode drop.
2
April 2005
MIC2168A
Micrel, Inc.
Absolute Maximum Ratings(1)
Operating Ratings(2)
Supply Voltage (VIN) ................................................... 15.5V
Booststrapped Voltage (VBST) .................................VIN +5V
Junction Temperature (TJ) ...................–40°C ≤ TJ ≤+125°C
Storage Temperature (TS) ........................ –65°C to +150°C
Supply Voltage (VIN) ..................................... +3V to +14.5V
Output Voltage Range .......................... 0.8V to VIN × DMAX
Package Thermal Resistance
θJA 10-lead MSOP ............................................. 180°C/W
Electrical Characteristics(3)
TJ = 25°C, VIN = 5V, unless otherwise specified. Bold values indicate –40°C < TJ < +125°C
Parameter
Condition
Min
Typ
Max
Units
Feedback Voltage Reference
(± 1%)
0.792
0.8
0.808
V
Feedback Voltage Reference
(± 2% over temp)
0.784
0.8
0.816
V
30
100
nA
Feedback Bias Current
Output Voltage Line Regulation
0.03
%/V
Output Voltage Load Regulation
0.5
%
0.6
%
Output Voltage Total Regulation
Oscillator Section
3V ≤ VIN ≤ 14.5V; 1A ≤ IOUT ≤ 10A; (VOUT =
2.5V)(4)
Oscillator Frequency
900
Maximum Duty Cycle
90
Minimum
On-Time(4)
Input and VDD Supply
1000
1100
kHz
%
30
60
ns
PWM Mode Supply Current
VCS = VIN –0.25V; VFB = 0.7V (output switching but excluding
external MOSFET gate current.)
1.6
3
mA
Shutdown Quiescent Current
VCOMP/EN = 0V
50
150
µA
0.25
0.4
VCOMP Shutdown Threshold
VCOMP Shutdown Blanking
Period
CCOMP = 100nF
Digital Supply Voltage (VDD)
VIN ≥ 6V
0.1
4
4.7
5
V
ms
5.3
V
Notes:
1. Absolute maximum ratings indicate limits beyond which damage to the component may occur. Electrical specifications do not apply when operating
the device outside of its operating ratings. The maximum allowable power dissipation is a function of the maximum junction temperature, TJ(max),
the junction-to-ambient thermal resistance, θJA, and the ambient temperature, TA. The maximum allowable power dissipation will result in excessive
die temperature, and the regulator will go into thermal shutdown.
2. Devices are ESD sensitive, handling precautions required.
3. Specification for packaged product only.
4. Guaranteed by design.
April 2005
3
M9999-062205
MIC2168A
Micrel, Inc.
Electrical Characteristics(5)
Parameter
Condition
Min
Typ
Max
Units
Error Amplifier
DC Gain
70
dB
Transconductance
1
ms
8.5
µA
Soft-Start
Soft-Start Current
After timeout of internal timer. See “Soft-Start” section.
Current Sense
CS Over Current Trip Point
VCS = VIN –0.25V
160
Temperature Coefficient
200
240
µA
+1800
ppm/°C
+3
%
Lower Threshold, VFB_UVT
(relative to VFB)
(relative to VFB)
–3
%
Rise/Fall Time
Into 3000pF at VIN > 5V
30
Output Fault Correction Thresholds
Upper Threshold, VFB_OVT
Gate Drivers
Output Driver Impedance
6
Ω
6
Ω
Source, VIN = 3V
10
Ω
Sink, VIN = 5V
Driver Non-Overlap Time
ns
Source, VIN = 5V
Sink, VIN = 3V
10
Note 6
10
20
Ω
ns
Notes:
5. Specification for packaged product only.
6. Guaranteed by design.
M9999-062205
4
April 2005
MIC2168A
Micrel, Inc.
Typical Characteristics
VIN = 5V
PWM Mode Supply Current
vs. Temperature
2.9
2.7
2.5
2.3
2.1
1.9
1.7
1.5
1.3
1.1
0.9
0.7
0.5
-40 -20 0 20 40 60 80 100 120 140
TEMPERATURE (°C)
2.0
PWM Mode Supply Current
vs. Supply Voltage
0.815
0.810
1.5
0.805
0.800
0.795
1.0
0.790
0.785
0.5
0
V FB vs. Temperature
0.820
5
10
SUPPLY VOLTAGE (V)
15
0.780
0
5
10
15
V IN (V)
V DD Line Regulation
6
0.815
V FB Line Regulation
0.820
V DD Load Regulation
5.01
4.99
5
0.810
4.97
4
0.805
0.800
4.95
4.93
3
0.795
4.91
2
0.790
4.89
1
0.785
0.780
-60 -30
0 30 60 90 120 150
TEMPERATURE (°C)
V DD Line Regulation
vs. Temperature
5.0
0
0
5
10
15
1050
0.5
1000
0
2.0
950
-0.5
900
-1.0
850
0.5
0.0
-60 -30
0 30 60 90 120 150
TEMPERATURE (°C)
Current Limit Foldback
4
800
-60 -30
240
30
1.0
3.0
2.5
1.0
5
10 15 20 25
LOAD CURRENT (mA)
Oscillator Frequency
vs. Supply Voltage
1.5
1100
1.5
0
Oscillator Frequency
vs. Temperature
1150
3.5
4.85
V IN (V)
1200
4.5
4.0
4.87
0 30 60 90 120 150
TEMPERATURE (°C)
-1.5
0
5
10
SUPPLY VOLTAGE (V)
15
Overcurrent Trip Point
vs. Temperature
220
3
200
180
2
160
1
0
April 2005
0
Top MOSFET = Si4800
140
R CS = 1kΩ
120
2
4
6
I LOAD (A)
8
10
100
-60 -30
0 30 60 90 120 150
TEMPERATURE (°C)
5
M9999-062205
MIC2168A
Micrel, Inc.
Functional Diagram
CIN
RCS
VIN
CS
VDD
5V LDO
D1
Current Limit
Comparator
VDD
5V
High-Side
Driver
5V
Bandgap
Reference
BG Valid
SW
Clamp &
Startup
Current
Ramp
Clock
CBST
4W
RSW
Driver
Logic
5V
Soft-Start &
Digital Delay
Counter
Q1
BOOST
Current Limit
Reference
0.8V
HSD
L1
VOUT
COUT
5V
Low-Side
Driver
LSD
Q2
PWM
Comparator
Enable
Error
Loop
0.8V
VREF +3%
VREF 3%
Error
Amp
FB
Hys
Comparator
R3
R2
MIC2168A
COMP
C2
GND
C1
R1
MIC2168A Block Diagram
Functional Description
version of VOUT to be slightly less than the reference voltage
causing the output voltage of the error amplifier to go high.
This will cause the PWM comparator to increase tON time of
the top side MOSFET, causing the output voltage to go up
and bringing VOUT back in regulation.
Soft-Start
The COMP/EN pin on the MIC2168A is used for the following
three functions:
1. Disables the part by grounding this pin
2. External compensation to stabilize the voltage
control loop
3. Soft-start
For better understanding of the soft-start feature, let’s assume VIN = 12V, and the MIC2168A is allowed to power-up
by un-grounding the COMP/EN pin. The COMP pin has an
internal 8.5µA current source that charges the external compensation capacitor. As soon as this voltage rises to 180mV
(t = Cap_COMP × 0.18V/8.5µA), the MIC2168A allows the
internal VDD linear regulator to power up and as soon as it
crosses the undervoltage lockout of 2.6V, the chip’s internal
oscillator starts switching. At this point in time, the COMP
pin current source increases to 40µA and an internal 11-bit
counter starts counting which takes approximately 2ms to
complete. During counting, the COMP voltage is clamped
at 0.65V. After this counting cycle the COMP current source
The MIC2168A is a voltage mode, synchronous step-down
switching regulator controller designed for high output power
without the use of an external sense resistor. It includes an
internal soft-start function which reduces the power supply
input surge current at start-up by controlling the output voltage rise time, a PWM generator, a reference voltage, two
MOSFET drivers, and short-circuit current limiting circuitry
to form a complete 1MHz switching regulator.
Theory of Operation
The MIC2168A is a voltage mode step-down regulator. The
figure above illustrates the block diagram for the voltage
control loop. The output voltage variation due to load or line
changes will be sensed by the inverting input of the transconductance error amplifier via the feedback resistors R3, and
R2 and compared to a reference voltage at the non-inverting input. This will cause a small change in the DC voltage
level at the output of the error amplifier which is the input to
the PWM comparator. The other input to the comparator is
a 0 to 1V triangular waveform. The comparator generates
a rectangular waveform whose width tON is equal to the
time from the start of the clock cycle t0 until t1, the time the
triangle crosses the output waveform of the error amplifier.
To illustrate the control loop, let us assume the output voltage drops due to sudden load turn-on, this would cause the
inverting input of the error amplifier which is divided down
M9999-062205
6
April 2005
MIC2168A
Micrel, Inc.
is reduced to 8.5µA and the COMP pin voltage rises from
0.65V to 0.95V, the bottom edge of the saw-tooth oscillator.
This is the beginning of 0% duty cycle and it increases slowly
causing the output voltage to rise slowly. The MIC2168A has
two hysteretic comparators that are enabled when VOUT is
within ±3% of steady state. When the output voltage reaches
97% of programmed output voltage, then the gm error amplifier
is enabled along with the hysteretic comparator. This point
onwards, the voltage control loop (gm error amplifier) is fully
in control and will regulate the output voltage.
Soft-start time can be calculated approximately by adding
the following four time frames:
t1 = Cap_COMP × 0.18V/8.5µA
t2 = 12 bit counter, approx 2ms
t3 = Cap_COMP × 0.3V/8.5µA
The current limiting resistor RCS is calculated by the following equation:
RCS = RDS(ON) Q1 × LL
200µA
I L = I LOAD =
VOUT =
VIN
Q1
MOSFET N
0
VOUT
L1 Inductor
RCS
CS
LSD
Q2
MOSFET N
C1
COUT
200µA
Figure 1. The MIC2168A Current Limiting Circuit
April 2005
(VIN - VOUT )
VIN × F SWITCHING × L
FSWITCHING = 1MHz
200µA is the internal sink current to program the
MIC2168A current limit.
The MOSFET RDS(ON) varies 30% to 40% with temperature;
therefore, it is recommended to add a 50% margin to the load
current (ILOAD) in the above equation to avoid false current
limiting due to increased MOSFET junction temperature rise.
It is also recommended to connect RCS resistor directly to
the drain of the top MOSFET Q1, and the RSW resistor to the
source of Q1 to accurately sense the MOSFETs RDS(ON). A
0.1µF capacitor in parallel with RCS should be connected to
filter some of the switching noise.
Internal VDD Supply
The MIC2168A controller internally generates VDD for self biasing and to provide power to the gate drives. This VDD supply
is generated through a low-dropout regulator and generates
5V from VIN supply greater than 5V. For supply voltage less
than 5V, the VDD linear regulator is approximately 200mV
in dropout. Therefore, it is recommended to short the VDD
supply to the input supply through a 10Ω resistor for input
supplies between 2.9V to 5V.
MOSFET Gate Drive
The MIC2168A high-side drive circuit is designed to switch an
N-Channel MOSFET. The block diagram in Figure 2 shows a
bootstrap circuit, consisting of D2 and CBST, supplies energy
to the high-side drive circuit. Capacitor CBST is charged while
the low-side MOSFET is on and the voltage on the VSW pin
is approximately 0V. When the high-side MOSFET driver is
turned on, energy from CBST is used to turn the MOSFET
on. As the MOSFET turns on, the voltage on the VSW pin
increases to approximately VIN. Diode D2 is reversed biased
and CBST floats high while continuing to keep the high-side
MOSFET on. When the low-side switch is turned back on,
CBST is recharged through D2. The drive voltage is derived
from the internal 5V VDD bias supply. The nominal low-side
gate drive voltage is 5V and the nominal high-side gate drive
voltage is approximately 4.5V due the voltage drop across D2.
An approximate 20ns delay between the high- and low-side
driver transitions is used to prevent current from simultaneously flowing unimpeded through both MOSFETs.
MOSFET Selection
The MIC2168A controller works from input voltages of 3V to
13.2V and has an internal 5V regulator to provide power to
turn the external N-Channel power MOSFETs for high- and
low-side switches. For applications where VIN < 5V, the internal
Soft-Start Time(Cap_COMP=100nF) = t1 + t2 + t3 +
t4 = 2.1ms + 2ms + 3.5ms + 1.8ms = 10ms
Current Limit
The MIC2168A uses the RDS(ON) of the top power MOSFET
to measure output current. Since it uses the drain to source
resistance of the power MOSFET
MOSFET, it is not very accurate. This
scheme is adequate to protect the power supply and external
components during a fault condition by cutting back the time
the top MOSFET is on if the feedback voltage is greater than
0.67V. In case of a hard short when feedback voltage is less
than 0.67V, the MIC2168A discharges the COMP capacitor
to 0.65V, resets the digital counter and automatically shuts
off the top gate drive, and the gm error amplifier and the
–3% hysteretic comparators are completely disabled and the
soft-start cycles restarts. This mode of operation is called the
“hiccup mode” and its purpose is to protect the down stream
load in case of a hard short. The circuit in Figure 1 illustrates
the MIC2168A current limiting circuit.
HSD
1
2(Inductor Ripple Current)
where:
Inductor Ripple Current =
 V

Cap_COMP
t4 -  OUT  x 0.5 x
8.5µA
 V IN 
C2
CIN
Equation (1)
7
M9999-062205
MIC2168A
Micrel, Inc.
VDD regulator operates in dropout mode, and it is necessary
that the power MOSFETs used are low threshold and are in
full conduction mode for VGS of 2.5V. For applications when
VIN > 5V; logic-level MOSFETs, whose operation is specified
at VGS = 4.5V must be used.
It is important to note the on-resistance of a MOSFET increases with increasing temperature. A 75°C rise in junction
temperature will increase the channel resistance of the MOSFET by 50% to 75% of the resistance specified at 25°C. This
change in resistance must be accounted for when calculating
MOSFET power dissipation and in calculating the value of
current-sense (CS) resistor. Total gate charge is the charge
required to turn the MOSFET on and off under specified
operating conditions (VDS and VGS). The gate charge is supplied by the MIC2168A gate drive circuit. At 1MHz switching
frequency and above, the gate charge can be a significant
source of power dissipation in the MIC2168A. At low output
load, this power dissipation is noticeable as a reduction in
efficiency. The average current required to drive the highside MOSFET is:
IG[high - side](avg) = Q G × fS
of the conduction losses during the on-time (PCONDUCTION)
and the switching losses that occur during the period of time
when the MOSFETs turn on and off (PAC).
PSW = P CONDUCTION + P AC
where:
PCONDUCTION
(
 VO 
D = duty cycle 

 V IN 
Making the assumption the turn-on and turn-off transition times
are equal; the transition times can be approximated by:
C
× V GS + C OSS × VIN
tT = ISS
IG
where:
CISS and COSS are measured at VDS = 0
IG = gate-drive current (1A for the MIC2168A)
The total high-side MOSFET switching loss is:
P AC = (V
(VIN +VD ) × IPK × tT × fS
where:
tT = switching transition time (typically 20ns to 50ns)
VD = freewheeling diode drop, typically 0.5V
fS it the switching frequency, nominally 1MHz
The low-side MOSFET switching losses are negligible and
can be ignored for these calculations.
Inductor Selection
Values for inductance, peak, and RMS currents are required
to select the output inductor. The input and output voltages
and the inductance value determine the peak-to-peak inductor ripple current. Generally, higher inductance values are
used with higher input voltages. Larger peak-to-peak ripple
currents will increase the power dissipation in the inductor
and MOSFETs. Larger output ripple currents will also require
more output capacitance to smooth out the larger ripple current. Smaller peak-to-peak ripple currents require a larger
inductance value and therefore a larger and more expensive
inductor. A good compromise between size, loss and cost is
to set the inductor ripple current to be equal to 20% of the
maximum output current. The inductance value is calculated
by the equation below.
)
A convenient figure of merit for switching MOSFETs is the on
resistance times the total gate charge RDS(ON) × QG. Lower
numbers translate into higher efficiency. Low gate-charge
logic-level MOSFETs are a good choice for use with the
MIC2168A.
Parameters that are important to MOSFET switch selection
are:
• Voltage rating
• On-resistance
• Total gate charge
The voltage ratings for the top and bottom MOSFET are
essentially equal to the input voltage. A safety factor of 20%
should be added to the VDS(max) of the MOSFETs to account
for voltage spikes due to circuit parasitics.
The power dissipated in the switching transistor is the sum
M9999-062205
× R SW
RSW = on-resistance of the MOSFET switch
Since the current from the gate drive comes from the input
voltage, the power dissipated in the MIC2168A due to gate
drive is:
= VIN IG[high -side](avg)
- side](avg) + IG[low -side](avg)
- side](avg)
2
P AC = P AC(off) + P AC(on)
where:
IG[high-side](avg) = average high-side MOSFET gate
current.
QG = total gate charge for the high-side MOSFET taken from
manufacturer’s data sheet for VGS = 5V.
The low-side MOSFET is turned on and off at VDS = 0
because the freewheeling diode is conducting during this
time. The switching loss for the low-side MOSFET is usually negligible. Also, the gate-drive current for the low-side
MOSFET is more accurately calculated using CISS at
VDS = 0 instead of gate charge.
For the low-side MOSFET:
IG[low -side](avg)
side](avg) = C ISS × V GS × fS
PGATEDRIVE
= ISW(rms)
L=
VOUT × (V
(V IN (max
max ) − VOUT )
VIN (max
max ) × fS × 0.2 × IOUT (max )
where:
fS = switching frequency, 1MHz
0.2 = ratio of AC ripple current to DC output current
VIN(max) = maximum input voltage
The peak-to-peak inductor current (AC ripple current) is:
8
April 2005
MIC2168A
IPP =
Micrel, Inc.
The output capacitor ESR also affects the overall voltage
feedback loop from stability point of view. See “Feedback Loop
Compensation” section for more information. The maximum
value of ESR is calculated:
∆V OUT
R ESR ≤
IPP
VOUT × (V
(V IN (max
max ) − VOUT )
VIN (max
max ) × fS × L
The peak inductor current is equal to the average output current
plus one half of the peak-to-peak inductor ripple current.
IPK = IOUT (max ) + 0.5 × IPP
The RMS inductor current is used to calculate the I2 × R
losses in the inductor.
IINDUCTOR(rms)

1
IP
= IOUT (max
max ) × 1 + 

3  IOUT (max
max ) 
where:
VOUT = peak-to-peak output voltage ripple
IPP = peak-to-peak inductor ripple current
The total output ripple is a combination of the ESR output
capacitance. The total ripple is calculated below:
2
Maximizing efficiency requires the proper selection of core
material and minimizing the winding resistance. The high
frequency operation of the MIC2168A requires the use of
ferrite materials for all but the most cost sensitive applications. Lower cost iron powder cores may be used but the
increase in core loss will reduce the efficiency of the power
supply. This is especially noticeable at low output power. The
winding resistance decreases efficiency at the higher output
current levels. The winding resistance must be minimized
although this usually comes at the expense of a larger inductor. The power dissipated in the inductor is equal to the sum
of the core and copper losses. At higher output loads, the
core losses are usually insignificant and can be ignored. At
lower output currents, the core losses can be a significant
contributor. Core loss information is usually available from the
magnetics vendor. Copper loss in the inductor is calculated
by the equation below:
PINDUCTOR Cu = IINDUCTOR(rms) 2 × R WINDING
2
∆V OUT =
(
)
2
COUT = output capacitance value
fS = switching frequency
The voltage rating of capacitor should be twice the voltage for
a tantalum and 20% greater for an aluminum electrolytic.
The output capacitor RMS current is calculated below:
I
IC
= PP
OUT(rms)
12
The power dissipated in the output capacitor is:
P DISS(C
× R ESR(C )
)= I C
OUT
OUT(rms) 2
OUT
Input Capacitor Selection
The input capacitor should be selected for ripple current rating
and voltage rating. Tantalum input capacitors may fail when
subjected to high inrush currents, caused by turning the input
supply on. Tantalum input capacitor voltage rating should
be at least 2 times the maximum input voltage to maximize
reliability. Aluminum electrolytic, OS-CON, and multilayer
polymer film capacitors can handle the higher inrush currents
without voltage derating. The input voltage ripple will primarily
depend on the input capacitor’s ESR. The peak input current
is equal to the peak inductor current, so:
∆V
VIN = IINDUCTOR(peak) × R ESR(C )
)
where:
THOT = temperature of the wire under operating load
T20°C = ambient temperature
RWINDING(20°C) is room temperature winding resistance (usually specified by the manufacturer)
Output Capacitor Selection
The output capacitor values are usually determined capacitors ESR (equivalent series resistance). Voltage and RMS
current capability are two other important factors selecting
the output capacitor. Recommended capacitors tantalum,
low-ESR aluminum electrolytics, and POSCAPS. The output
capacitor’s ESR is usually the main cause of output ripple.
April 2005
(
where:
D = duty cycle
The resistance of the copper wire, RWINDING, increases with
temperature. The value of the winding resistance used should
be at the operating temperature.
R WINDING(hot) = R WINDING(20 °C) × 1 + 0.0042
0.0042 × (T HOT − T 20 °C )
 I × (1
(1 − D
D) 
PP

 + IPP × R ESR
 C OUT × fS 
IN
The input capacitor must be rated for the input current ripple.
The RMS value of input capacitor current is determined at
the maximum output current. Assuming the peak-to-peak
inductor ripple current is low:
9
M9999-062205
MIC2168A
Micrel, Inc.
VF = forward voltage at the peak diode current
The external Schottky diode, D1, is not necessary for circuit
operation since the low-side MOSFET contains a parasitic
body diode. The external diode will improve efficiency and
decrease high frequency noise. If the MOSFET body diode
is used, it must be rated to handle the peak and average current. The body diode has a relatively slow reverse recovery
time and a relatively high forward voltage drop. The power
lost in the diode is proportional to the forward voltage drop
of the diode. As the high-side MOSFET starts to turn on, the
body diode becomes a short circuit for the reverse recovery
period, dissipating additional power. The diode recovery and
the circuit inductance will cause ringing during the high-side
MOSFET turn-on. An external Schottky diode conducts at
a lower forward voltage preventing the body diode in the
MOSFET from turning on. The lower forward voltage drop
dissipates less power than the body diode. The lack of a
reverse recovery mechanism in a Schottky diode causes
less ringing and less power loss. Depending on the circuit
components and operating conditions, an external Schottky
diode will give a 1/2% to 1% improvement in efficiency.
Feedback Loop Compensation
The MIC2168A controller comes with an internal transconductance error amplifier used for compensating the voltage
feedback loop by placing a capacitor (C1) in series with a
resistor (R1) and another capacitor C2 in parallel from the
COMP pin to ground. See “Functional Block Diagram.”
Power Stage
The power stage of a voltage mode controller has an inductor, L1, with its winding resistance (DCR) connected to the
output capacitor, COUT, with its electrical series resistance
(ESR) as shown in Figure 3. The transfer function G(s), for
such a system is:
IC IN (rms) ≈ IOUT (max
max ) × D × (1 − D)
The power dissipated in the input capacitor is:
PDISS(C ) = IC (rms) 2 × R ESR(C )
IN
IN
IN
Voltage Setting Components
The MIC2168A requires two resistors to set the output voltage as shown in Figure 2.
R1
Error
Amp
FB
7
R2
VREF
0.8V
MIC2168 [adj.]
Figure 2. Voltage-Divider Configuration
Where:
VREF for the MIC2168A is typically 0.8V
The output voltage is determined by the equation:
 R1 
V O = VREF × 1 +

 R2 
A typical value of R1 can be between 3kΩ and 10kΩ. If R1 is
too large, it may allow noise to be introduced into the voltage
feedback loop. If R1 is too small, in value, it will decrease the
efficiency of the power supply, especially at light loads. Once
R1 is selected, R2 can be calculated using:
V
×R
R1
R2 = REF
V O − VREF
L
External Schottky Diode
An external freewheeling diode is used to keep the inductor
current flow continuous while both MOSFETs are turned off.
This dead time prevents current from flowing unimpeded
through both MOSFETs and is typically 15ns. The diode
conducts twice during each switching cycle. Although the
average current through this diode is small, the diode must
be able to handle the peak current.
VO
ESR
COUT
Figure 3. The Output LC Filter in a Voltage Mode
Buck Converter


(1 + ESR × s × C)
G(S) - 
2

DCR × s × C + s × L × C + 1 + ESR × s × C 
The reverse voltage requirement of the diode is:
VDIODE(rrm) = VIN
Plotting this transfer function with the following assumed values
(L=2 µH, DCR=0.009Ω, COUT=1000µF, ESR=0.050Ω) gives
lot of insight as to why one needs to compensate the loop by
adding resistor and capacitors on the COMP pin. Figures 4
and 5 show the gain curve and phase curve for the above
transfer function.
The power dissipated by the Schottky diode is:
PDIODE = ID(avg) × VF
where:
M9999-062205
DCR
10
April 2005
MIC2168A
30
Micrel, Inc.
30
7.5
GAIN
0
0
50
P
15
100
37.5
150
60 60
100
100
180
3
1.10
5
4
1 .10
f
6
1 .10
1 .10
1000000
100
100
P
50
100
150
3
1.10
100
100
4
1 .10
f
5
1 .10
6
1 .10
1000000
Figure 5. Phase Curve for G(s)
It can be seen from the transfer function G(s) and the gain
curve that the output inductor and capacitor create a two pole
system with a break frequency at:
fC =
1
2 × π √ L × COUT
Therefore, fLC = 3.6kHz
By looking at the phase curve, it can be seen that the output
capacitor ESR (0.050Ω) cancels one of the two poles (LCOUT)
system by introducing a zero at:
fZERO =
Error Amplifier (z) - gm ×
1+ R1 × S × C1

C1 × C2 × S
s × (C1 + C2)1 + R1 ×
C1 + C2 

The above equation can be simplified by assuming
C2<<C1,


1+ R1 × S × C1
Error Amplifier (z) - gm ×
 s × (C1)(1+ R1 × C2 × S) 


1
2 × π × ESR × C OUT
Therefore, FZERO = 6.36kHz.
From the point of view of compensating the voltage loop, it is
recommended to use higher ESR output capacitors since they
provide a 90° phase gain in the power path. For comparison
purposes, Figure 6, shows the same phase curve with an
ESR value of 0.002Ω.
April 2005
6
1 .10
1000000
5
1 .10
It can be seen from Figure 5 that at 50kHz, the phase is
approximately –90° versus Figure 6 where the number is
–150°. This means that the transconductance error amplifier has to provide a phase boost of about 45° to achieve a
closed loop phase margin of 45° at a crossover frequency
of 50kHz for Figure 4, versus 105° for Figure 6. The simple
RC and C2 compensation scheme allows a maximum error
amplifier phase boost of about 90°. Therefore, it is easier to
stabilize the MIC2168A voltage control loop by using high
ESR value output capacitors.
gm Error Amplifier
It is undesirable to have high error amplifier gain at high
frequencies because high frequency noise spikes would be
picked up and transmitted at large amplitude to the output,
thus, gain should be permitted to fall off at high frequencies.
At low frequency, it is desired to have high open-loop gain to
attenuate the power line ripple. Thus, the error amplifier gain
should be allowed to increase rapidly at low frequencies.
The transfer function with R1, C1, and C2 for the internal
gm error amplifier can be approximated by the following
equation:
0
180
4
1 .10
f
Figure 6. The Phase Curve with ESR = 0.002Ω
Figure 4. The Gain Curve for G(s)
0
3
1.10
From the above transfer function, one can see that R1 and
C1 introduce a zero and R1 and C2 a pole at the following
frequencies:
Fzero= 1/2 π × R1 × C1
Fpole = 1/2 π × C2 × R1
[email protected] = 1/2 π × C1
Figures 7 and 8 show the gain and phase curves for the above
transfer function with R1 = 9.3k, C1 = 1000pF, C2 = 100pF,
and gm = .005Ω–1. It can be seen that at 50kHz, the error
amplifier exhibits approximately 45° of phase margin.
11
M9999-062205
MIC2168A
Micrel, Inc.
100
71.607
OPEN LOOP GAIN MARGIN
60
60
40
E
20
.001
3
1 .10
1000
4
1 .10
5
6
1 .10
f
1 .10
0
42.933
50
100
100
7
1 .10
10000000
3
1.10
OPEN LOOP PHASE MARGIN
220
240
E
3
1.10
4
f
1 .10
5
1 .10
1 .10
1000000
300
350
360
260
100
6
5
1 .10
250
269.097
200
215.856
270
4
1 .10
f
Figure 9. Open-Loop Gain Margin
Figure 7. Error Amplifier Gain Curve
10
10
50
10
10
6
1 .10
1000000
100
3
1.10
4
f
1 .10
5
1 .10
6
1 .10
1000000
Figure 10. Open-Loop Phase Margin
Figure 8. Error Amplifier Phase Curve
Total Open-Loop Response
The open-loop response for the MIC2168A controller is easily
obtained by adding the power path and the error amplifier
gains together, since they already are in Log scale. It is
desirable to have the gain curve intersect zero dB at tens of
kilohertz, this is commonly called crossover frequency; the
phase margin at crossover frequency should be at least 45°.
Phase margins of 30° or less cause the power supply to have
substantial ringing when subjected to transients, and have
little tolerance for component or environmental variations.
Figures 9 and 10 show the open-loop gain and phase margin.
It can be seen from Figure 9 that the gain curve intersects
the 0dB at approximately 50kHz, and from Figure 10 that at
50kHz, the phase shows approximately 50° of margin.
M9999-062205
12
April 2005
MIC2168A
Micrel, Inc.
Design Example
Layout and Checklist:
1. Connect the current limiting (CS) resistor directly
to the drain of top MOSFET Q1.
2. Connect the VSW pin directly to the source of top
MOSFET Q1 thru a 4Ω to 10Ω resistor. The purpose
of this resistor is to filter the switch node.
3. The feedback resistors R1 and R2 should be placed
close to the FB pin. The top side of R1 should
connect directly to the output node. Run this trace
away from the switch node (junction of Q1, Q2, and
L1). The bottom side of R1 should connect to the
GND pin on the MIC2168A.
4. The compensation resistor and capacitors should
be placed right next to the COMP/EN pin and the
other side should connect directly to the GND pin
on the MIC2168A rather than going to the plane.
5. The input bulk capacitors should be placed close
to the drain of the top MOSFET.
6. The 1µF ceramic capacitor should be placed right
on the VIN pin of the MIC2168A.
7. The 4.7µF to 10µF ceramic capacitor should be
placed right on the VDD pin.
8. The source of the bottom MOSFET should connect
directly to the input capacitor GND with a thick
trace. The output capacitor and the input capacitor
should connect directly to the GND plane.
9. Place a 0.1µF ceramic capacitor in parallel with
the CS resistor to filter any switching noise.
April 2005
13
M9999-062205
MIC2168A
Micrel, Inc.
Package Information
Rev.. 00
10-Pin MSOP (MM)
MICREL INC.
TEL
2180 FORTUNE DRIVE
SAN JOSE, CA 95131
USA
+ 1 (408) 944-0800 FAX + 1 (408) 474-1000 WEB http://www.micrel.com
This information furnished by Micrel in this data sheet is believed to be accurate and reliable. However no responsibility is assumed by Micrel for its use.
Micrel reserves the right to change circuitry and specifications at any time without notification to the customer.
Micrel Products are not designed or authorized for use as components in life support appliances, devices or systems where malfunction of a product can
reasonably be expected to result in personal injury. Life support devices or systems are devices or systems that (a) are intended for surgical implant into
the body or (b) support or sustain life, and whose failure to perform can be reasonably expected to result in a significant injury to the user. A Purchaser's
use or sale of Micrel Products for use in life support appliances, devices or systems is a Purchaser's own risk and Purchaser agrees to fully indemnify
Micrel for any damages resulting from such use or sale.
© 2005 Micrel Incorporated
M9999-062205
14
April 2005