19-3289; Rev 1; 6/05 KIT ATION EVALU LE B A IL A AV 3V to 28V Input, Low-Cost, Hysteretic Synchronous Step-Down Controllers The MAX8576–MAX8579 synchronous PWM buck controllers use a hysteretic voltage-mode control algorithm to achieve a fast transient response without requiring loop compensation. The MAX8576/MAX8577 contain an internal LDO regulator allowing the controllers to function from only one 3V to 28V input supply. The MAX8578/MAX8579 do not contain the internal LDO and require a separate supply to power the IC when the input supply is higher than 5.5V. The MAX8576– MAX8579 output voltages are adjustable from 0.6V to 0.9 x VIN at loads up to 15A. Nominal switching frequency is programmable over the 200kHz to 500kHz range. High-side MOSFET sensing is used for adjustable hiccup current-limit and short-circuit protection. The MAX8576/MAX8578 can start up into a precharged output without pulling the output voltage down. The MAX8577/MAX8579 have startup output overvoltage protection (OVP), and will pull down a precharged output. Applications Features ♦ 3V to 28V Supply Voltage Range ♦ 1.2% Accurate Over Temperature ♦ Adjustable Output Voltage Down to 0.6V ♦ 200kHz to 500kHz Switching Frequency ♦ Adjustable Temperature-Compensated Hiccup Current Limit ♦ Lossless Peak Current Sensing ♦ Monotonic Startup into Prebias Output (MAX8576/MAX8578) ♦ Startup Overvoltage Protection (MAX8577/MAX8579) ♦ Enable/Shutdown ♦ Adjustable Soft-Start Ordering Information TEMP RANGE PIN-PACKAGE MAX8576EUB -40°C to +85°C 10 µMAX® AGP and PCI-Express Power Supplies MAX8577EUB -40°C to +85°C 10 µMAX Graphic-Card Power Supplies MAX8578EUB -40°C to +85°C 10 µMAX Set-Top Boxes MAX8579EUB -40°C to +85°C 10 µMAX Motherboard Power Supplies PART Point-of-Load Power Supplies Typical Operating Circuit INPUT UP TO 28V FB OCSET SS VL GND DL IN MAX8576 MAX8577 DH OUTPUT 0.6V TO 0.9 x VIN LX BST µMAX is a registered trademark of Maxim Integrated Products, Inc. Pin Configurations appear at end of data sheet. ________________________________________________________________ Maxim Integrated Products For pricing, delivery, and ordering information, please contact Maxim/Dallas Direct! at 1-888-629-4642, or visit Maxim’s website at www.maxim-ic.com. 1 MAX8576–MAX8579 General Description MAX8576–MAX8579 3V to 28V Input, Low-Cost, Hysteretic Synchronous Step-Down Controllers ABSOLUTE MAXIMUM RATINGS IN to GND (MAX8576/MAX8577) ...........................-0.3V to +30V VL to GND (MAX8576/MAX8577).............................-0.3V to +6V IN to VL (MAX8576/MAX8577) ...............................-0.3V to +30V VCC to GND (MAX8578/MAX8579) ..........................-0.3V to +6V SS to GND (MAX8576/MAX8577) ...............-0.3V to (VVL + 0.3)V SS to GND (MAX8578/MAX8579)...............-0.3V to (VCC + 0.3)V DL to GND (MAX8576/MAX8577) ...............-0.3V to (VVL + 0.3)V DL to GND (MAX8578/MAX8579) ..............-0.3V to (VCC + 0.3)V BST to GND ............................................................-0.3V to +36V BST to LX..................................................................-0.3V to +6V LX to GND .....................-1V (-2.5V for <50ns Transient) to +30V DH to LX..................................................-0.3V to +(VBST + 0.3)V FB to GND ................................................................-0.3V to +6V EN to GND (MAX8578/MAX8679EUB) .....................-0.3V to +6V OCSET to GND (MAX8576/MAX8677) ........-0.3V to (VIN + 0.3)V OCSET to GND (MAX8578/MAX8679) ...................-0.3V to +30V OCSET to LX (MAX8576/MAX8677) ............-0.6V to (VIN + 0.3)V OCSET to LX (MAX8578/MAX8679) .......................-0.6V to +30V DH and DL Continuous Current ............................±250mA RMS Continuous Power Dissipation (TA = +70°C) 10-Pin µMAX (derate 5.6mW/°C above +70°C) ...........444mW Operating Temperature Range ...........................-40°C to +85°C Junction Temperature ......................................................+150°C Storage Temperature Range .............................-65°C to +150°C Lead Temperature (soldering, 10s) ................................+300°C Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability. ELECTRICAL CHARACTERISTICS (VIN = 12V (MAX8576/MAX8577 only), 4.7µF capacitor from VL (MAX8576/MAX8577 only) or VCC (MAX8578/MAX8579 only) to GND; VCC = VEN = 5V (MAX8578/MAX8579 only); 0.01µF capacitor from SS to GND; VFB = 0.65V; VBST = 5V; VLX = VGND = 0V; VOCSET = 11.5V; DH = unconnected; DL = unconnected; TA = 0°C to +85°C, unless otherwise noted. Typical values are at TA = +25°C.) PARAMETER CONDITIONS MIN TYP MAX UNITS SUPPLY VOLTAGES MAX8576/MAX8577 5.5 28.0 IN = VL (MAX8576/MAX8577) 3.0 5.5 VCC Input Voltage MAX8576/MAX8577 3.0 5.5 VL Output Voltage IVL = 10mA (MAX8576/MAX8577) 4.75 VL Maximum Output Current MAX8576/MAX8577 IN Supply Voltage VL or VCC Undervoltage Lockout (UVLO) 5.0 5.25 20 2.75 2.8 2.90 Falling 2.4 2.45 2.5 No switching, VFB = 0.65V (MAX8576/MAX8577) Supply Current VEN = 0V or VFB = 0.65V, no switching (MAX8578/MAX8579) V V mA Rising Hysteresis V 350 V mV VIN = 12V 0.6 2 VIN = VVL = 5V 1.1 3 VIN = VVL = 3.3V 0.6 2 VCC = 5V 0.6 2 VCC = 3.3V 0.6 2 0.6 0.607 V 20 28.0 mV mA REGULATOR Output Regulation Accuracy Output Regulation Hysteresis FB Propagation Delay VFB peak 0.593 (Note 1) 12.5 FB falling to DL falling 50 FB rising to DH falling 70 Overvoltage-Protection (OVP) Threshold High-Side Current-Sense Program Current (Note 2) 2 0.70 TA = +85°C TA = +25°C 0.75 ns 0.80 60 42.5 50 _______________________________________________________________________________________ 57.5 V µA 3V to 28V Input, Low-Cost, Hysteretic Synchronous Step-Down Controllers (VIN = 12V (MAX8576/MAX8577 only), 4.7µF capacitor from VL (MAX8576/MAX8577 only) or VCC (MAX8578/MAX8579 only) to GND; VCC = VEN = 5V (MAX8578/MAX8579 only); 0.01µF capacitor from SS to GND; VFB = 0.65V; VBST = 5V; VLX = VGND = 0V; VOCSET = 11.5V; DH = unconnected; DL = unconnected; TA = 0°C to +85°C, unless otherwise noted. Typical values are at TA = +25°C.) PARAMETER High-Side Current-Sense Overcurrent Trip Adjustment Range CONDITIONS VIN - VOCSET MIN TYP 0.05 Soft-Start Internal Resistance 45 Fault Hiccup Internal SS Pulldown VLX < VOCSET and VFB < VSS Current 80 MAX UNITS 0.40 V 125 kΩ 250 nA DRIVER SPECIFICATIONS DH Driver Resistance DL Driver Resistance Dead Time Sourcing current 2.6 4.0 Sinking current 1.9 3.0 Sourcing current 2.6 4.0 Sinking current 1.1 2.0 DH low to DL high and DL low to DH high (adaptive) 40 140 245 Normal operation 120 220 Current fault 580 VBST - VLX = 5.5V, VLX = 28V, VFB < VSS 1.65 DH Minimum On-Time DL Minimum On-Time BST Current Ω Ω ns ns ns mA EN Input Voltage Low VCC = 3V (MAX8578/MAX8579) Input Voltage High VCC = 5.5V (MAX8578/MAX8579) 0.7 1.5 V V THERMAL SHUTDOWN Thermal Shutdown Rising temperature, hysteresis = 20°C (typ) °C +160 ELECTRICAL CHARACTERISTICS (VIN = 12V (MAX8576/MAX8577 only), 4.7µF capacitor from VL (MAX8576/MAX8577 only) or VCC (MAX8578/MAX8579 only) to GND; VCC = VEN = 5V (MAX8578/MAX8579 only); 0.01µF capacitor from SS to GND; VFB = 0.65V; VBST = 5V; VLX = VGND = 0V; VOCSET = 11.5V; DH = unconnected; DL = unconnected; TA = -40°C to +85°C, unless otherwise noted. Note 3) PARAMETER CONDITIONS MIN TYP MAX UNITS SUPPLY VOLTAGES MAX8576/MAX8577 5.5 28.0 IN = VL, MAX8576/MAX8577 3.0 5.5 VCC Input Voltage MAX8576/MAX8577 3.0 5.5 VL Output Voltage IVL = 10mA, MAX8576/MAX8577 4.75 5.25 VL Maximum Output Current MAX8576/MAX8577 IN Supply Voltage 20 V V V mA _______________________________________________________________________________________ 3 MAX8576–MAX8579 ELECTRICAL CHARACTERISTICS (continued) MAX8576–MAX8579 3V to 28V Input, Low-Cost, Hysteretic Synchronous Step-Down Controllers ELECTRICAL CHARACTERISTICS (continued) (VIN = 12V (MAX8576/MAX8577 only), 4.7µF capacitor from VL (MAX8576/MAX8577 only) or VCC (MAX8578/MAX8579 only) to GND; VCC = VEN = 5V (MAX8578/MAX8579 only); 0.01µF capacitor from SS to GND; VFB = 0.65V; VBST = 5V; VLX = VGND = 0V; VOCSET = 11.5V; DH = unconnected; DL = unconnected; TA = -40°C to +85°C, unless otherwise noted. Note 3) PARAMETER VL or VCC Undervoltage Lockout (UVLO) CONDITIONS MIN TYP MAX Rising 2.75 2.90 Falling 2.40 2.55 No switching, VFB = 0.65V (MAX8576/MAX8577) Supply Current VIN = 12V V 2 VIN = VVL = 5V 3.5 VIN = VVL = 3.3V VEN = 0V or VFB = 0.65V, no switching (MAX8578/MAX8579) UNITS 2 VCC = 5V 2 VCC = 3.3V 2 mA REGULATOR Output Regulation Accuracy VFB peak Overvoltage-Protection (OVP) Threshold High-Side Current-Sense OverCurrent Trip Adjustment Range VIN - VOCSET 0.591 0.607 V 0.70 0.80 V 0.05 0.40 V DRIVER SPECIFICATIONS DH Driver Resistance DL Driver Resistance Sourcing current 4 Sinking current 3.0 Sourcing current 4.0 Sinking current 2.0 DH Minimum On-Time Ω Ω 245 ns Normal operation 220 ns Input Voltage Low VCC = 3V, MAX8578/MAX8579 0.7 V Input Voltage High VCC = 5.5V, MAX8578/MAX8579 DL Minimum On-Time EN 1.5 Note 1: Guaranteed by design. Note 2: This current linearly compensates for the MOSFET temperature coefficient. Note 3: Specifications to -40°C are guaranteed by design and not production tested. 4 _______________________________________________________________________________________ V 3V to 28V Input, Low-Cost, Hysteretic Synchronous Step-Down Controllers EFFICIENCY vs. LOAD CURRENT (CIRCUIT OF FIGURE 3) EFFICIENCY (%) 60 50 40 60 40 20 20 10 10 0.1 1 1.82 1.81 1.80 1.79 1.78 1.77 1.76 VIN = 12V 1.75 0.1 10 1 5 0 10 15 LOAD CURRENT (A) LOAD CURRENT (A) LOAD CURRENT (A) LINE REGULATION (CIRCUIT OF FIGURE 2) LINE REGULATION (CIRCUIT OF FIGURE 3) SWITCHING FREQUENCY vs. INPUT VOLTAGE (CIRCUIT OF FIGURE 3) 0A LOAD 1.83 1.82 1.81 15A LOAD NO LOAD 1.82 1.80 1.78 1.76 5A LOAD 1.74 1.80 1.72 10 15 20 500 450 400 350 300 200 5 25 550 250 1.70 1.79 MAX8576-79 toc06 1.84 600 SWITCHING FREQUENCY (kHz) 1.84 MAX8576-79 toc05 1.86 MAX8576-79 toc04 1.85 5 1.83 0 100 10 VOUT = 1.5V 50 30 VIN = 12V VOUT = 1.8V VOUT = 2.5V 70 30 0 OUTPUT VOLTAGE (V) 80 1.84 MAX8576-79 toc03 VOUT = 1.8V VOUT = 1.5V 70 VOUT = 3.3V 90 OUTPUT VOLTAGE (V) EFFICIENCY (%) 80 100 OUTPUT VOLTAGE (V) VOUT = 2.5V VOUT = 3.3V 90 MAX8576-79 toc01 100 LOAD REGULATION (CIRCUIT OF FIGURE 2) MAX8576-79 toc02 EFFICIENCY vs. LOAD CURRENT (CIRCUIT OF FIGURE 2) 10 INPUT VOLTAGE (V) 15 20 25 10 5 INPUT VOLTAGE (V) LOAD TRANSIENT (CIRCUIT OF FIGURE 2) 15 20 25 INPUT VOLTAGE (V) LOAD TRANSIENT (CIRCUIT OF FIGURE 3) MAX8576-79 toc07 MAX8576-79 toc08 5A IOUT 2.5A 12A IOUT 6A 50mV/div AC-COUPLED VOUT 40µs/div 50mV/div AC-COUPLED VOUT 40µs/div _______________________________________________________________________________________ 5 MAX8576–MAX8579 Typical Operating Characteristics (TA = +25°C, unless otherwise noted.) MAX8576–MAX8579 3V to 28V Input, Low-Cost, Hysteretic Synchronous Step-Down Controllers Typical Operating Characteristics (continued) ( TA = +25°C, unless otherwise noted.) POWER-UP VIN (CIRCUIT OF FIGURE 3) POWER-UP VCC (CIRCUIT OF FIGURE 3) MAX8576-79 toc09 VIN MAX8576-79 toc10 10V/div 10V/div VIN 0 VCC 5V/div VOUT 1V/div 5A/div ILX 0 5V/div VCC 1V/div VOUT 5A/div ILX 0 0 400µs/div 400µs/div POWER-DOWN VCC (CIRCUIT OF FIGURE 3) POWER-UP (CIRCUIT OF FIGURE 2) MAX8576-79 toc11 MAX8576-79 toc12 10V/div VIN 10V/div VIN 0 0 5V/div VCC 1V/div VOUT VOUT 1V/div ILX 5A/div 0 0 10A/div ILX 400µs/div 0 1ms/div POWER-DOWN (CIRCUIT OF FIGURE 2, MAX8576) STARTUP AND SHUTDOWN (CIRCUIT OF FIGURE 3) MAX8576-79 toc13 MAX8576-79 toc14 5V/div VIN VEN 2V/div 0 0 VOUT 1V/div VDL 10V/div 0 VOUT 1V/div 0 0 10A/div ILX 5A/div ILX 0 0 4ms/div 6 400µs/div _______________________________________________________________________________________ 3V to 28V Input, Low-Cost, Hysteretic Synchronous Step-Down Controllers STARTUP AND SHUTDOWN (CIRCUIT OF FIGURE 2) MONOTONIC OUTPUT-VOLTAGE RISE (CIRCUIT OF FIGURE 2, MAX8576) MAX8576-79 toc15 MAX8576-79 toc16 10V/div 0 VGS(Q3) 0.5V/div 0 2V/div 0 VSS VOUT VIN 10V/div VOUT 1.5V 0.5V/div 20V/div VLX 10A/div 0 ILX 5V/div 0 VDL 4ms/div 1ms/div NONMONOTONIC OUTPUT-VOLTAGE RISE (CIRCUIT OF FIGURE 2, MAX8577) SHORT CIRCUIT AND RECOVERY (CIRCUIT OF FIGURE 2) MAX8576-79 toc17 MAX8576-79 toc18 VIN 10V/div VOUT 1.5V 0.5V/div 2A/div IIN 20V/div VLX 10V/div VIN IOUT 10A/div VOUT 2V/div 0 5V/div 0 VDL 1ms/div 10ms/div OUTPUT OVERVOLTAGE PROTECTION (CIRCUIT OF FIGURE 2) MAX8576-79 toc19 VDH 20V/div 0 VDL 5V/div 0 VOUT 1V/div 0 VFB 0.5V/div 0 200µs/div _______________________________________________________________________________________ 7 MAX8576–MAX8579 Typical Operating Characteristics (continued) ( TA = +25°C, unless otherwise noted.) MAX8576–MAX8579 3V to 28V Input, Low-Cost, Hysteretic Synchronous Step-Down Controllers Pin Description PIN MAX8576/ MAX8577 MAX8578/ MAX8579 NAME 1 1 FB Feedback Input. Regulates at VFB = 0.59V. Connect FB to a resistor-divider to set the output voltage. See the Setting the Output Voltage section. 2 2 SS Soft-Start. Use an external capacitor (CSS) to adjust the soft-start time. An internal 80kΩ resistor gives approximately 4ms soft-start time for a 0.01µF external capacitor. An internal 250nA current sink in hiccup mode gives approximately 10% duty cycle during fault conditions. 3 — VL Internal 5V Linear-Regulator Output. Bypass with a 4.7µF or larger ceramic capacitor. Must be connected to IN for operation from a 3.3V to 5.5V input. — 3 VCC Supply Input (3V to 5.5V). Bypass with a 4.7µF or larger ceramic capacitor to GND. 4 4 GND Ground 5 5 DL Low-Side Gate-Drive Output. Drives the synchronous-rectifier MOSFET. 6 6 BST Boost-Capacitor Connection for High-Side Gate-Drive Output. Connect a 0.1µF ceramic capacitor from BST to LX and a Schottky or switching diode and a 4.7Ω series resistor from BST to VL (MAX8576/MAX8577) or VCC (MAX8578/MAX8579). See Figure 4. 7 7 LX External Inductor Connection. Connect LX to the junctions of the MOSFETs and inductor. 8 8 DH High-Side Gate-Drive Output. Drives the high-side MOSFET. 9 — IN Supply Voltage Input of the Internal Linear Regulator (3V to 28V). Connect to VL for operation from 3V to 5.5V input. Connect a 0.47µF or larger ceramic capacitor from IN to GND. — 9 EN Enable Input. A logic low on EN shuts down the converter and discharges the soft-start capacitor. Drive high or connect to VCC for normal operation. 10 8 10 OCSET FUNCTION Overcurrent-Limit Set. Programs the high-side peak current-limit threshold by setting the maximum-allowed VDS voltage drop across the high-side MOSFET. Connect a resistor from IN to OCSET; an internal 50µA current sink sets the maximum voltage drop relative to VIN. See the Setting the Current Limit section. _______________________________________________________________________________________ 3V to 28V Input, Low-Cost, Hysteretic Synchronous Step-Down Controllers MAX8576–MAX8579 BST FAULT OCSET LX MAX8578/ MAX8579 EN GND 0.3 DHI DH FB DRIVERS OVP LOGIC 0.75V SS SS RAMP LX DLI DL 0.05V IN MAX8576 MAX8577 VL REG VLOK POK MAX8576–MAX8579 VL VCC MAX8578 MAX8579 REF REFOK GND Figure 1. Functional Diagram _______________________________________________________________________________________ 9 MAX8576–MAX8579 3V to 28V Input, Low-Cost, Hysteretic Synchronous Step-Down Controllers INPUT 9V TO 24V C1 C11 C2 R1 1 FB OCSET 10 2 SS IN R2 C4 C3 9 C8 C12 C5 OFF ON R4 3 Q3 MAX8576 MAX8577 VL C6 DH Q1 8 R7 R3 OUTPUT 1.8V/12A L1 R5 4 GND 5 DL LX 7 BST 6 C7 C9 C13 Q2 R6 R8 D1 12V INPUT, 1.8V/12A OUTPUT (fS = 300kHz) CIRCUIT IS TARGETED FOR 10.8V TO 13.2V INPUT. HOWEVER, INPUT RANGE OF 9V TO 24V IS POSSIBLE FOR IC EVALUATION. 30V RATED MOSFET MUST BE INSTALLED IF INPUT IS RAISED ABOVE 16V. ALL OTHER COMPONENTS CAN REMAIN UNCHANGED. Figure 2. MAX8576/MAX8577 Typical Application Circuit INPUT 9V TO 24V C21 ON OFF R9 1 FB C14 OCSET 10 R10 C16 2 SS 3 VCC 3V TO 5.5V C17 MAX8578 MAX8579 EN 9 DH 8 C15 C19 Q4 R11 VOUT 1.8V/5A L2 4 GND 5 DL LX 7 BST 6 C18 C20 C23 Q5 R12 R13 D2 12V INPUT, 1.8V/5A OUTPUT (fS = 500kHz, ALL CERAMIC) CIRCUIT IS TARGETED FOR 10.8V TO 13.2V INPUT. HOWEVER, INPUT RANGE OF 9V TO 24V IS POSSIBLE FOR IC EVALUATION. 30V RATED MOSFET MUST BE INSTALLED IF INPUT IS RAISED ABOVE 16V. ALL OTHER COMPONENTS CAN REMAIN UNCHANGED. Figure 3. MAX8578/MAX8579 Typical Application Circuit 10 ______________________________________________________________________________________ C22 C10 3V to 28V Input, Low-Cost, Hysteretic Synchronous Step-Down Controllers COMPONENTS QTY C1, C2 2 DESCRIPTION/VENDOR PART NUMBER MAX8578/MAX8579 External Component List DESCRIPTION/VENDOR PART NUMBER COMPONENT QTY 470µF, 35V aluminum electrolytic capacitors Sanyo 35MV470WX C14 1 10µF, 25V X5R ceramic capacitor C15 1 1µF, 25V X5R ceramic capacitor C16 1 4700pF, 10V X7R ceramic capacitor C3 1 10µF, 25V X7R ceramic capacitor C17 1 4.7µF, 6.3V X5R ceramic capacitor C4 1 0.01µF, 10V X7R ceramic capacitor C18 1 0.1µF, 10V X7R ceramic capacitor C5 1 1µF, 35V X7R ceramic capacitor C19 1 0.01µF, 25V X7R ceramic capacitor C6 1 4.7µF, 6.3V X5R ceramic capacitor C7, C12 2 0.1µF, 10V X7R ceramic capacitors C20 1 C8 1 0.027µF, 25V X7R ceramic capacitor 47µF, 6.3V, ESR = 5mΩ, ceramic capacitor Taiyo Yuden JMK432476MM 1 0.01µF, 25V X5R ceramic capacitor 2 2200µF, 6.3V aluminum electrolytic capacitors Rubycon 6.3MBZ2200M10X20 C21 C9, C10 C22 0 Optional (47µF, 6.3V, ESR = 5mΩ ceramic capacitor Taiyo Yuden JMK432476MM) C11 1 0.01µF, 25V X5R ceramic capacitor C13 1 3300pF, 6.3V X5R ceramic capacitor C23 1 1000pF, 25V X5R ceramic capacitor D1 1 High-speed diode, 100V, 250mA Philips BAS316 (SOD-323) D2 1 High-speed diode, 100V, 250mA Philips BAS316 (SOD-323) L1 1 1.8µH, 14A, 3.48mΩ Panasonic ETQP2H1R8BFA L2 1 2.2µH, 7.3A, 9.8mΩ Sumida CDEP104L-2R2 Q1 1 30V, 12.5mΩ (max), SO-8 International Rectifier IRF7821 Q4 1 30V, 18mΩ (max), SO-8 International Rectifier IRF7807Z Q2 1 30V, 3.7mΩ, SO-8 International Rectifier IRF7832 Q5 1 30V, 9.5mΩ, SO-8 International Rectifier IRF7821 Q3 1 2N7002 SOT-23 R9 1 6.04kΩ ±1% resistor R1 1 6.04kΩ ±1% resistor R10 1 2.49kΩ ±1% resistor R2 1 5.11 kΩ ±1% resistor R11 1 12.4kΩ ±1% resistor R3 1 12.4kΩ ±1% resistor R12 1 2Ω ±5% resistor R4 1 1kΩ ±5% resistor R13 1 4.7Ω ±5% resistor R5 1 20kΩ ±5% resistor R6 1 2Ω ±5% resistor R7 1 10Ω ±5% resistor R8 1 4.7Ω ±5% resistor Detailed Description The MAX8576–MAX8579 synchronous PWM buck controllers use Maxim’s proprietary hysteretic voltagemode control algorithm to achieve fast transient response without any loop-compensation requirement. The controller drives a pair of external n-channel power MOSFETs to improve efficiency and cost. The MAX8576/MAX8577 contain an internal linear lowdropout (LDO) regulator allowing the controller to operate from a single 3V to 28V input supply. The MAX8578/MAX8579 do not contain the internal LDO and require a separate supply to power the IC when the input supply is higher than 5.5V. The MAX8576– MAX8579 output voltages are adjustable from 0.6V to 0.9 x VIN at loads up to 15A. ______________________________________________________________________________________ 11 MAX8576–MAX8579 MAX8576/MAX8577 External Component List MAX8576–MAX8579 3V to 28V Input, Low-Cost, Hysteretic Synchronous Step-Down Controllers Nominal switching frequency is programmable over the 200kHz to 500kHz range. High-side MOSFET sensing is used for adjustable hiccup current-limit and short-circuit protection. The MAX8576/MAX8578 can start up into a precharged output without pulling the output voltage down. The MAX8577/MAX8579 have startup output overvoltage protection (OVP). The MAX8578/MAX8579 have a logic-enable input to turn on and off the output. The MAX8576/MAX8577 are turned off by pulling SS low with an external small n-channel MOSFET (see Figure 2). DC-DC Converter Control Architecture A proprietary hysteretic-PWM control scheme ensures high efficiency, fast switching, and fast transient response. This control scheme is simple: when the output voltage falls below the regulation threshold, the error comparator begins a switching cycle by turning on the high-side switch. This switch remains on until the minimum on-time expires and the output voltage is in regulation or the current-limit threshold is exceeded. Once off, the high-side switch remains off until the minimum off-time expires and the output voltage falls below the regulation threshold. During this period, the lowside synchronous rectifier turns on and remains on until the voltage at FB drops below its regulation threshold. The internal synchronous rectifier eliminates the need for an external Schottky diode. Voltage-Positioning Load Regulation As seen in Figures 2 and 3, the MAX8576–MAX8579 use a unique feedback network. By taking feedback from the LX node through R3 (R11 for the MAX8578/MAX8579), the usual phase lag due to the output capacitor does not exist, making the loop stable for either electrolytic or ceramic output capacitors. This configuration causes the output voltage to shift by the inductor DC resistance multiplied by the load current. This voltage-positioning load regulation greatly reduces overshoot during load transients, which effectively halves the peak-to-peak outputvoltage excursions compared to traditional step-down converters. See the Load Transient graphs in the Typical Operating Characteristics. Internal 5V Linear Regulator All MAX8576/MAX8577 functions are powered from the on-chip, low-dropout 5V regulator with the input connected to IN. Bypass the regulator’s output (VL) with a 1µF or greater ceramic capacitor. The capacitor must have an equivalent series resistance (ESR) of no greater than 10mΩ. When VIN is less than 5.5V, short VL to IN. The MAX8578/MAX8579 do not have the onchip 5V regulator and must use a separate external 12 supply from 3V to 5.5V connected to VCC if the input voltage is greater than 5.5V. Undervoltage Lockout If VL (MAX8576/MAX8577) or VCC (MAX8578/MAX8579) drops below 2.45V (typ), the MAX8576–MAX8579 assume that the supply voltage is too low for proper circuit operation, so the UVLO circuitry inhibits switching and forces the DL and DH gate drivers low for the MAX8576/MAX8578, and DH low and DL high for the MAX8577/MAX8579. After VIN rises above 2.8V (typ), the controller goes into the startup sequence and resumes normal operation. Output Overvoltage Protection The MAX8576–MAX8579 output overvoltage protection is provided by a glitch-resistant comparator on FB with a trip threshold of 750mV (typ). The overvoltage-protection circuit is latched by an OVP fault, terminating the run cycle and setting DH low and DL high. The fault is cleared by toggling EN or UVLO. Output OVP is active whenever the internal reference is in regulation. Startup and Soft-Start The soft-start sequence is initiated upon initial powerup, recovering from UVLO, or driving EN (MAX8578/ MAX8579) high from a low state, or releasing SS (MAX8576/MAX8577) from a low state. The external soft-start capacitor (CSS) is connected to an internal resistor-divider that exponentially charges the capacitor to 0.6V, with an SS ramp interval of 5 x RC or 4ms per 0.01µF. SS is one input to the internal voltage error comparator, while FB is the other input. The output voltage fed back to FB tracks the rising SS voltage. Switching commences immediately if VFB is initially less than VSS; if VFB is greater than VSS, DH remains low until V FB is less than V SS . DL remains low in the MAX8576/MAX8578. This prevents the converter from operating in reverse. However, DL is high before startup in the MAX8577/MAX8579 to enable OVP protection in case the high-side MOSFET is shorted. Enable Connecting EN to GND or logic low places the MAX8578/MAX8579 in shutdown mode. In shutdown, DH and DL are forced low, and the voltage at SS is discharged with a 250nA current, resulting in a ramp-down interval of approximately 10x the soft-start ramp-up interval. VSS must fall to within 50mV of GND before another cycle can commence. SS (MAX8576/ MAX8577) or EN (MAX8578/MAX8579) do not need to be cycled after an overcurrent event. Connect EN to VCC or logic high for normal operation. To shut down the MAX8576/MAX8577, use an external circuit connected ______________________________________________________________________________________ 3V to 28V Input, Low-Cost, Hysteretic Synchronous Step-Down Controllers Synchronous-Rectifier Driver (DL) Synchronous rectification reduces conduction losses in the rectifier by replacing the normal Schottky catch diode with a low-resistance MOSFET switch. The MAX8576–MAX8579 also use the synchronous rectifier to ensure proper startup of the boost gate-driver circuit. The DL low-side waveform is always the complement of the DH high-side drive waveform (with controlled dead time to prevent cross-conduction or shoot-through). A dead-time circuit monitors the DL output and prevents the high-side MOSFET from turning on until DL is fully off. For the dead-time circuit to work properly, there must be a low-resistance, low-inductance path from the DL driver to the MOSFET gate. Otherwise, the sense circuitry in the MAX8576–MAX8579 may interpret the MOSFET gate as off when gate charge actually remains. Use very short, wide traces (50 mils to 100 mils wide if the MOSFET is 1in from the device). The dead time at the other edge (DH turning off) is also determined through gate sensing. High-Side Gate-Drive Supply (BST) Gate-drive voltage for the high-side n-channel switch is generated by a flying-capacitor boost circuit (Figure 4). The capacitor between BST and LX is charged from the IN supply up to VIN minus the diode drop while the lowside MOSFET is on. When the low-side MOSFET is switched off, the stored voltage of the capacitor is stacked above LX to provide the necessary turn-on voltage (VGS) for the high-side MOSFET. The controller then closes an internal switch between BST and DH to turn the high-side MOSFET on. Current-Limit Circuit Current limit is set externally with a resistor from OCSET to the drain of the high-side n-channel MOSFET that is normally connected to the input supply. The resistor programs the high-side peak current limit by setting the maximum-allowed V DS(ON) voltage drop across the high-side MOSFET. An internal 50µA current sink sets the maximum voltage drop relative to V IN . If V FB < 300mV, any overcurrent event (VDS of the high-side n-channel MOSFET is larger than the limit programmed at OCSET) immediately sets DH low and terminates the run cycle. If VFB > 300mV and an overcurrent event is detected, DH is immediately set low and four sequential overcurrent events terminate the run cycle. Once the run cycle is terminated, the SS capacitor is slowly discharged through the internal 250nA current sink to provide a hiccup current-limit effect. Choosing the proper value resistor is discussed in the Setting the Current Limit section. Switching Frequency Nominal switching frequency is programmable over the 200kHz to 500kHz range. This allows tradeoffs in efficiency, switching frequency, inductor value, and component size. Faster switching frequency allows for smaller inductor values but does result in some efficiency loss. Inductor-value calculations are provided in the Inductor Value section. The switching frequency is tuned by the selection of the feed-forward capacitor (CFF). See the Feed-Forward Capacitor section. Thermal-Overload Protection Thermal-overload protection limits total power dissipation in the MAX8576–MAX8579. When the junction temperature exceeds T J = +160°C, an internal thermal sensor shuts down the IC, allowing the IC to cool. The thermal sensor turns the IC on again after the junction temperature cools to +140°C, resulting in a pulsed output during continuous thermal-overload conditions. Design Procedures IN Setting the Output Voltage BST DH MAX8576– MAX8579 N LX DL N Select an output voltage between 0.6V and 0.9 x VIN by connecting FB to a resistive voltage-divider between LX and GND (see Figures 2 and 3). Choose R1 for approximately 50µA to 150µA bias current in the resistive divider. A wide range of resistor values is acceptable, but a good starting point is to choose R1 as 6.04kΩ. Then, R3 is given by: Figure 4. DH Boost Circuit ______________________________________________________________________________________ 13 MAX8576–MAX8579 to SS. See Figure 2 for details. The maximum on-resistance of the small external n-channel MOSFET should be less than 40Ω so that the SS voltage is below 10mV. MAX8576–MAX8579 3V to 28V Input, Low-Cost, Hysteretic Synchronous Step-Down Controllers ⎛ VOUT + 0.01V + (RDC × 0.5 × IOUTMAX ) ⎞ R3 = R1 × ⎜ − 1⎟ VFB ⎝ ⎠ where VFB = 0.590V, RDC is the DC resistance of the output inductor, IOUTMAX is the maximum output current. The term 0.01V is to reflect 1/2 of the feedbackthreshold hysteresis. Inductor Value The inductor value is bounded by two operating parameters: the switching frequency and the inductor peakto-peak ripple current. The peak-to-peak ripple current is typically in the range of 20% to 40% of the maximum output current. The equation below defines the inductance value: ⎛ ⎞ VOUT × (VIN − VOUT ) L= ⎜ ⎟ ⎜ VIN × fS × ILOAD(MAX ) × LIR ⎟ ⎝ ⎠ where LIR is the ratio of inductor current ripple to DC load current and fS is the switching frequency. A good compromise between size, efficiency, and cost is an LIR of 30%. The selected inductor must have a saturated current rating above the sum of the maximum output current and half of the peak-to-peak ripple current. The DC current rating of the inductor must be above the maximum output current to keep the temperature rise within the desired range. In addition, the DC resistance of the inductor must meet the requirement below: RDC ≤ ∆VOUT IOUTMAX where ∆VOUT is the maximum-allowed output-voltage drop from no load to full load (IOUTMAX). Setting the Current Limit Resistor R2 (R7 for the MAX8577/MAX8579) of Figure 2 (Figure 3 for the MAX8577/MAX8579) sets the current limit and is connected between OCSET and the drain of the high-side n-channel MOSFET. An internal 50µA current sink sets the maximum voltage drop across the high-side n-channel MOSFET relative to VIN. The maximum VDS drop needs to be determined. This is calculated by: VDS(ON)MAX = IDS(MAX) × RDS(ON)MAX IDS(MAX) must be equal or greater than the maximum peak inductor current at the maximum output current. Use RDS(ON)MAX at the junction temperature of +25°C. The current limit is temperature compensated. 14 ROCSET is calculated using the VDS(ON)MAX with the following formula: VDS(ON)MAX ROCSET = 50µA A 0.01µF ceramic capacitor is required in parallel with ROCSET to decouple high-frequency noise. MOSFET Selection The MAX8576–MAX8579 drive two external, logic-level, n-channel MOSFETs as the circuit switching elements. The key selection parameters are: 1) On-resistance (RDS(ON)): the lower, the better. 2) Maximum drain-to-source voltage (VDSS): should be at least 20% higher than the input supply rail at the high-side MOSFET’s drain. 3) Gate charges (Qg, Qgd, Qgs): the lower, the better. For a 3.3V input application, choose a MOSFET with a rated RDS(ON) at VGS = 2.5V. For a 5V input application, choose the MOSFETs with rated RDS(ON) at VGS ≤ 4.5V. For a good compromise between efficiency and cost, choose the high-side MOSFET (N1) that has conduction losses equal to switching loss at nominal input voltage and output current. The selected high-side MOSFET (N1) must have RDS(ON) that satisfies the current-limit-setting condition above. For N2, make sure that it does not spuriously turn on due to dV/dt caused by N1 turning on as this results in shoot-through current degrading the efficiency. MOSFETs with a lower Qgd / Qgs ratio have higher immunity to dV/dt. For proper thermal-management design, the power dissipation must be calculated at the desired maximum operating junction temperature, maximum output current, and worst-case input voltage (for the low-side MOSFET, worst case is at VIN(MAX); for the high-side MOSFET, it could be either at VIN(MAX) or VIN(MIN)). N1 and N2 have different loss components due to the circuit operation. N2 operates as a zero-voltage switch; therefore, major losses are: the channel-conduction loss (P N2CC ) and the body-diode conduction loss (PN2DC). ⎛ ⎞ V PN2CC = ⎜1 − OUT ⎟ × ILOAD2 × RDS(ON) VIN ⎠ ⎝ Use RDS(ON) at TJ(MAX). PN2DC = 2 × ILOAD × VF × t dt × fS where VF is the body-diode forward-voltage drop, tDT is the dead time between N1 and N2 switching transitions (40ns typ), and fS is the switching frequency. ______________________________________________________________________________________ 3V to 28V Input, Low-Cost, Hysteretic Synchronous Step-Down Controllers ⎛V ⎞ PN1CC = ⎜ OUT ⎟ × ILOAD2 × RDS(ON) ⎝ VIN ⎠ Input Capacitor The input filter capacitor reduces peak currents drawn from the power source and reduces noise and voltage ripple on the input caused by the circuit’s switching. The input capacitor must meet the ripple-current requirement (IRMS) imposed by the switching currents defined by the following equation: IRMS = ILOAD × VOUT × (VIN − VOUT ) VIN Use RDS(ON) at TJ(MAX). ⎛ Qgs + Qgd ⎞ PN1SW = VIN × ILOAD × ⎜ ⎟ × fS ⎝ IGATE ⎠ where IGATE is the average DH driver output-current capability determined by: IGATE ≅ 0.5 × VL RDH + RGATE where RDH is the high-side MOSFET driver’s on-resistance (2Ω typ) and RGATE is the internal gate resistance of the MOSFET (approximately 2Ω). RGATE PN1DR = Qg × VGS × fS × RGATE + RDH where VGS is approximately equal to VL. In addition to the losses above, allow about 20% more for additional losses due to MOSFET output capacitances and N2 body-diode reverse-recovery charge dissipated in N1 that exists, but is not well defined in the MOSFET data sheet. Refer to the MOSFET data sheet for thermal-resistance specification to calculate the PC board area needed to maintain the desired maximum operating junction temperature with the above calculated power dissipations. To reduce EMI caused by switching noise, add 0.1µF ceramic capacitor from the high-side switch drain to the low-side switch source or add resistors in series with DH and DL to slow down the switching transitions. However, adding series resistors increases the power dissipation of the MOSFET, so be sure this does not overheat the MOSFET. The minimum load current must exceed the high-side MOSFET’s maximum leakage current over temperature if fault conditions are expected. I RMS has a maximum value when the input voltage equals twice the output voltage (VIN = 2 x VOUT), so IRMS(MAX) = ILOAD / 2. Ceramic capacitors are recommended due to their low ESR and ESL at high frequency, with relatively lower cost. Choose a capacitor that exhibits less than 10°C temperature rise at the maximum operating RMS current for optimum long-term reliability. Output Capacitor The key selection parameters for the output capacitor are the actual capacitance value, the ESR, the equivalent series inductance (ESL), and the voltage-rating requirements. These parameters affect the overall stability, output voltage ripple, and transient response. The output ripple has three components: variations in the charge stored in the output capacitor, the voltage drop across the capacitor’s ESR, and the ESL caused by the current into and out of the capacitor. The maximum output ripple voltage can be estimated by: VRIPPLE = VRIPPLE(ESR) + VRIPPLE(C) + VRIPPLE(ESL) The output voltage ripple as a consequence of the ESR and output capacitance is: VRIPPLE(ESR) = IP−P × ESR VRIPPLE(C) = IP−P COUT × fS ⎛V ⎞ VRIPPLE(ESL) = ⎜ IN ⎟ × ESL ⎝ L ⎠ ⎛ V − VOUT ⎞ ⎛ VOUT ⎞ IP−P = ⎜ IN ⎟ × ⎜ V ⎟ fS × L ⎝ ⎠ ⎝ IN ⎠ where IP-P is the peak-to-peak inductor current (see the Inductor Value section). These equations are suitable for initial capacitor selection, but final values should be ______________________________________________________________________________________ 15 MAX8576–MAX8579 N1 operates as a duty-cycle control switch and has the following major losses: the channel-conduction loss (PN1CC), the VL overlapping switching loss (PN1SW), and the drive loss (PN1DR). N1 does not have bodydiode conduction loss because the diode never conducts current. MAX8576–MAX8579 3V to 28V Input, Low-Cost, Hysteretic Synchronous Step-Down Controllers chosen based on a prototype or evaluation circuit. As a general rule, a smaller current ripple results in less output voltage ripple. Since the inductor ripple current is a factor of the inductor value and input voltage, the output voltage ripple decreases with larger inductance and increases with higher input voltages. For reliable and safe operation, ensure that the capacitor’s voltage and ripple-current ratings exceed the calculated values. The response of the MAX8576–MAX8579 to a load 1transient depends on the selected output capacitors. After a load transient, the output voltage instantly changes by ESR times ∆ILOAD. Before the controller can respond, the output voltage deviates further depending on the inductor and output capacitor values. The controller responds immediately as the output voltage deviates from its regulation limit (see the Typical Operating Characteristics). The MAX8576–MAX8579 are compatible with both aluminum electrolytic and ceramic output capacitors. Due to the limited capacitance of a ceramic capacitor, it is typically used for a higher switching frequency and lower output current. Aluminum electrolytic is more applicable to frequencies up to 300KHz and can support higher output current with its much higher capacitance value. Due to the much higher ESL and ESR of the aluminum electrolytic capacitor, an RC filter (R7 and C12 of Figure 2) is required to prevent excessive ESL and ESR ripple from tripping the feedback threshold prematurely. MOSFET Snubber Circuit Fast-switching transitions cause ringing because of resonating circuit parasitic inductance and capacitance at the switching nodes. This high-frequency ringing occurs at LX’s rising and falling transitions and can interfere with circuit performance and generate EMI. To dampen this ringing, a series RC snubber circuit is added across each switch. Below is the procedure for selecting the value of the series RC circuit: 1) Connect a scope probe to measure VLX to GND, and observe the ringing frequency, fR. 2) Find the capacitor value (connected from LX to GND) that reduces the ringing frequency by half. The circuit parasitic (CPAR) at LX is then equal to 1/3 the value of the added capacitance above. The circuit parasitic inductance (LPAR) is calculated by: LPAR = 16 1 (2πfR )2 × CPAR The resistor for critical dampening (RSNUB) is equal to 2π x fR x LPAR. Adjust the resistor value up or down to tailor the desired damping and the peak voltage excursion. The capacitor (CSNUB) should be at least 2 to 4 times the value of CPAR to be effective. The power loss of the snubber circuit is dissipated in the resistor (PRSNUB) and can be calculated as: PRSNUB = CSNUB × (VIN )2 × fSW where VIN is the input voltage and fSW is the switching frequency. Choose an RSNUB power rating that meets the specific application’s derating rule for the power dissipation calculated. Feed-Forward Capacitor The feed-forward capacitor, C8 (Figure 2, MAX8576/ MAX8577 with aluminum electrolytic output capacitor), or C19 (Figure 3, MAX8578/MAX8579 with ceramic output capacitor), dominantly affects the switching frequency. Choose a ceramic X7R capacitor with a value given by: C8 = ⎛ 1 ⎛ V ⎞ 1 V ⎞ ×⎜ − 120ns × IN ⎟ × 49.5 × ⎜1− OUT ⎟ RFB ⎝ FS VOUT ⎠ VIN ⎠ ⎝ C19 = ⎛ V ⎞ ⎛ 1 1 V ⎞ ×⎜ − 120ns × IN ⎟ × 39.5 × ⎜1− OUT ⎟ RFB ⎝ FS VOUT ⎠ VIN ⎠ ⎝ or where FS is the desired switching frequency, and RFB is the parallel combination of the two feedback dividerresistors (R1 and R3 of Figure 2, and R9 and R11 of Figure 3). Select the closest standard value to C8 and C19 as possible. The output inductor and output capacitor also affect the switching frequency, but to a much lesser extent. The equations for C8 and C19 above should yield within ±30% of the desired switching frequency for most applications. The values of C8 and C19 can be increased to lower the frequency, or decreased to raise the frequency for better accuracy. Application Information PC Board Layout Guidelines Careful PC board layout is critical to achieve low switching losses and clean, stable operation. The switching power stage requires particular attention. Follow these guidelines for good PC board layout: ______________________________________________________________________________________ 3V to 28V Input, Low-Cost, Hysteretic Synchronous Step-Down Controllers 5) Place the MOSFET as close as possible to the IC to minimize trace inductance. If parallel MOSFETs are used, keep the gate connection to both gates equal. 6) Connect the drain leads of the power MOSFET to a large copper area to help cool the device. Refer to the power MOSFET data sheet for the recommended copper area. 2) For output current > 10A, a four-layer PC board is recommended. Pour a ground plane in the second layer underneath the IC to minimize noise coupling. 7) Place the feedback components as close to the IC pins as possible. The feedback divider-resistor from FB to the output inductor should be connected directly to the inductor and not sharing with other connections to this node. 3) Input, output, and VL capacitors are connected to the power ground plane with the exception of C12 and C22. These capacitors and all other capacitors are connected to the analog ground plane. 4) Make the connection from the current-limit setting resistor directly to the high-side MOSFET’s drain to minimize the effect of PC board trace resistance and inductance. 8) Refer to the EV kit for further guidelines. Suggested External Component Manufacturers MANUFACTURER COMPONENT Central Semiconductor Panasonic WEBSITE PHONE Diodes www.centralsemi.com 631-435-1110 Inductors www.panasonic.com 402-564-3131 Sumida Inductors www.sumida.com 847-956-0666 International Rectifier MOSFETs www.irf.com 800-341-0392 Kemet Capacitors www.kemet.com 864-963-6300 Taiyo Yuden Capacitors www.t-yuden.com 408-573-4150 TDK Capacitors www.component.tdk.com 888-835-6646 Rubycon Capacitors www.rubycon.com 408-467-3864 Pin Configurations TOP VIEW FB 1 SS 2 VL 3 GND 4 DL 5 10 OCSET MAX8576 MAX8577 µMAX FB 1 10 OCSET 9 IN SS 2 8 DH VCC 3 7 LX GND 4 7 LX 6 BST DL 5 6 BST MAX8578 MAX8579 9 EN 8 DH µMAX Chip Information TRANSISTOR COUNT: 2087 PROCESSS: BICMOS ______________________________________________________________________________________ 17 MAX8576–MAX8579 1) Place IC decoupling capacitors as close to IC pins as possible. Place the input ceramic decoupling capacitor directly across and as close as possible to the high-side MOSFET’s drain and the low-side MOSFET’s source. This is to help contain the high switching current within this small loop. Package Information (The package drawing(s) in this data sheet may not reflect the most current specifications. For the latest package outline information, go to www.maxim-ic.com/packages.) e 10LUMAX.EPS MAX8576–MAX8579 3V to 28V Input, Low-Cost, Hysteretic Synchronous Step-Down Controllers 4X S 10 10 INCHES H Ø0.50±0.1 0.6±0.1 1 1 0.6±0.1 BOTTOM VIEW TOP VIEW D2 MILLIMETERS MAX DIM MIN 0.043 A 0.006 A1 0.002 A2 0.030 0.037 0.120 D1 0.116 0.118 D2 0.114 E1 0.116 0.120 0.118 E2 0.114 0.199 H 0.187 L 0.0157 0.0275 L1 0.037 REF b 0.007 0.0106 e 0.0197 BSC c 0.0035 0.0078 0.0196 REF S α 0° 6° MAX MIN 1.10 0.05 0.15 0.75 0.95 2.95 3.05 2.89 3.00 2.95 3.05 2.89 3.00 4.75 5.05 0.40 0.70 0.940 REF 0.177 0.270 0.500 BSC 0.090 0.200 0.498 REF 0° 6° E2 GAGE PLANE A2 c A b A1 α E1 D1 L L1 FRONT VIEW SIDE VIEW PROPRIETARY INFORMATION TITLE: PACKAGE OUTLINE, 10L uMAX/uSOP APPROVAL DOCUMENT CONTROL NO. 21-0061 REV. I 1 1 Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are implied. Maxim reserves the right to change the circuitry and specifications without notice at any time. 18 ____________________Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600 © 2005 Maxim Integrated Products Printed USA is a registered trademark of Maxim Integrated Products, Inc.