MICROCHIP AN912

AN912
Designing LF Talkback for a Magnetic Base Station
Author:
Ruan Lourens
Microchip Technology Inc.
INTRODUCTION
This application note builds on application note AN232
Low Frequency Magnetic Transmitter Design
(DS00232). It covers the design process to implement
LF Talkback functionality. AN232 covers some of the
magnetism basics and design principles to implement
the drive circuitry. LF Talkback generally refers to the
process in which a transponder can communicate back
to a magnetic transmitter base station by loading the
generated magnetic field. By measuring the small
changes in the transmitter coil's voltage, used to generate the field, the communications’ data is extracted. LF
Talkback is commonly used in RFID, automotive
transponders, active transponders, and many other
bidirectional LF communications topologies.
MAIN BUILDING BLOCKS
Figure 1 shows the main building blocks that make up
the LF Talkback system described in this document.
The base station generates a strong magnetic field by
setting up resonance in a serial resonant tank. The
circulating energy in the resonant tank typically generates 300V peak-to-peak voltage across the transmitting
antenna coil at 125 kHz. The transponder, whether
active or passive, is magnetically coupled to the base
station’s transmitting coil and the transponder’s
magnetic loading has a small effect on the quality factor
(Q) of the transmitter resonant tank. Talkback is
accomplished by changing or modulating the magnetic
loading and can be observed as small voltage changes
across the base station's resonant transmitter coil. The
difficulty is to detect a few mV of modulation on the
300V peak-to-peak carrier.
This document will cover the different stages needed to
implement a typical LF Talkback system and explain
the process in choosing the different stage characteristics. It explains the various performance and cost tradeoffs made for the reference design and how it can be
adapted to better suit the readers needs.
FIGURE 1:
Base Station
Transponder
Peak
Detector
DC
Decouple
Low Pass
Filter
Data
Slicer
VREF
M
Data
 2004 Microchip Technology Inc.
Data
DS00912A-page 1
AN912
A high voltage peak detector is used to extract the
basic envelope of the base station's resonant tank. The
output of the peak detector will be 150 VDC with about
2V peak-to-peak of carrier ripple at 125 kHz and then
about 2 mV of modulated signal. The modulation signal
strength is mostly dependent on the distance between
the transponder and the transmitter coil as the
magnetic coupling decreases to the third power of the
distance between the two devices. The next stage is a
passive high-pass filter to decouple or block the high
DC voltage. The DC extracted voltage is then fed into a
low-pass filter, leaving the required modulating signal.
The last stage is the data slicer that compares the
modulating signal to some reference to extract the
original signal sent by the transponder.
LF Talkback receiver can be thought of as detecting
and decoding an amplitude modulation (AM) signal that
has a very low modulation index on a relatively large
carrier.
SYSTEM ASSUMPTIONS
The LF Talkback system designed in this document is
targeted for a LF base station that has the following
characteristics and is based on the design as per
AN232:
• The LF Talkback signal is amplitude modulated at
200 µs multiples. This is also referred to as the
basic pulse element period or TE.
• The tank is driven by a 12V half-bridge driver.
• The tank inductance is 162 µH and the resonant
capacitor is 10 nF with a resonant frequency of
125 kHz.
• The tank Q is 25. As a result, the tank or carrier
voltage is 300V peak-to-peak or 150V 0-to-peak.
• Transponder induced modulation of 2 mV in
magnitude needs to be detected.
To get an understanding of the impedances involved,
lets consider the following: using Equation 1, the
equivalent parallel resistance of the tank is 3.18 kΩ.
The additional parallel impedance that a transponder
represents to induce a 2 mV signal on the tank is in the
order of 500 MΩ. What the LF Talkback system detects
is the result of a 500 MΩ resistor being switched in and
out in parallel with the tank at the data rate. Therefore,
it is very important that the peak detector have a highimpedance at the data rate to maintain good sensitivity.
EQUATION 1:
THE EFFECTIVE PARALLEL
IMPEDANCE OF A
RESONANT TANK
RPARALLEL
THE PEAK DETECTOR
There are a number of aspects to consider in designing
a peak detector for this application:
1.
2.
3.
4.
5.
6.
The peak detector has to be able to operate at
the high voltages of the resonant tank.
Maintain a good tank Q or, in other words, it
should not add unnecessary loading on the main
resonant tank. If it does load the tank, it will
result in a lower modulation voltage induced by
the transponder.
Reduce carrier ripple as far as possible.
Maintain the modulation signal.
Have a fast large swing dynamic response and
be able to settle quickly after the field is turned
on.
Cost of the system.
Some of the peak detector requirements are conflicting
and as a result, the designer has to find an acceptable
compromise with the final system performance in mind.
One can sacrifice a specific parameter and make up for
it in a later stage where optimization of that aspect is
easily accomplished.
As an example to optimize requirement 3, one needs to
increase the size of the capacitor C2 (Figure 2), but
that will negatively affect requirements 2, 4 and 5 if a
passive peak detector is used. An active peak detector
could have solved the conflict, but at the 600V swing,
one has little choice but to use a passive peak detector
while maintaining a low-cost design. A relatively low
capacitance value is chosen for C1 of 1 nF. This
maintains the dynamic response requirement for
settling quickly after the field is applied and does not
load the tank unnecessarily. Capacitor C2 should have
at least a 300 VDC peak rating and a high tolerance
capacity is acceptable to save cost. An ultra fast diode
is required in the peak detector with a 400V or better
rating and low junction capacitance. A UF1005 diode
was chosen, it has a 600V rating and 10 pF of junction
capacitance.
FIGURE 2:
HV-Env
D1
C1
L1
C2
R1
= 2πLFCQ
L = Tank inductance in H = 162 µH
Fc = Center frequency of tank = 125 kHz
Q = Tank quality factor = 25
DS00912A-page 2
 2004 Microchip Technology Inc.
AN912
The envelope detector with only D1 and C2 has a
greatly different response to increasing and decreasing
voltage amplitudes of the resonant tank. The voltage
designated by the signal HV_Env (Figure 2) rises
quickly with increasing tank amplitudes because D1
has a low-impedance in forward conduction. The tank
voltage decreases slower when the tank amplitude is
lowered because C2 can only discharge through D1,
which has a high-impedance in the reverse direction.
The situation can be remedied to some extent by the
introduction of R1 which helps to discharge C2, but the
value of R1 should be high enough to maintain a good
tank Q as per requirement 2 above. A 10 MΩ value for
R1 works well, but note that R1 needs to be
implemented as a series of two resistors. This is done
to stay within the safe voltage range of 0805 resistors
are used.
The 125 kHz carrier ripple voltage, without R1, is about
2V peak-to-peak and is due to the junction capacitance
and reverse leakage of D1. The addition of R1 has little
effect on the ripple voltage, but does improve the
detectors dynamic performance at the data rate. The
carrier ripple voltage will be filtered out at a later stage
where a more effective solution can be implemented.
THE DC DECOUPLER CONFLICTS
The HV_Env signal (Figure 2) consists of three main
components:
1.
2.
3.
A 150V DC signal, as a result of the peak
detector.
2V peak-to-peak ripple voltage at the carrier
frequency.
The modulated data signal at a TE of 200 µs and
a 2 mV peak-to-peak amplitude, highest fundamental harmonic content is at 2.5 kHz [1/(2*200
uS)], irrespective of the modulation scheme
used (i.e., Manchester, PWM etc.).
The aim of the decoupling stage is to reject the high DC
voltage without adding unnecessary loading to the tank
via the peak detector. It should also have a fast dynamic
response and stabilize quickly after the tank is energized. The dynamic response of the LF Talkback system
is the major design hurdle to overcome as far as the
decoupling stage is concerned. The problem is aggravated when the transponder needs to communicate on
the LF link soon after the base station communicated
with the transponder.
FIGURE 3:
HV-Env
LP Filter
C
R
The system can be simplified as shown in Figure 3.
The output of the peak detector can be simplified as the
step response source with a 150V amplitude that also
has the carrier and data signals superimposed on it as
described earlier. The output response of the
decoupling stage is given by Equation 2. This is also
the input signal to the low-pass filter.
EQUATION 2:
V = 150e-t/τ
τ = RC
It is useful to think in terms of τ (RC time constant)
because the voltage across the resistor reduces by a
factor of 0.368 as every τ second elapses. The
exponential decay curve, for the voltage across R, is
shown in Figure 4 and indicates that the initial voltage
decays rapidly, but settles out slower as the voltage is
reduced across the resistor. The system must be
allowed to settle for a long enough period so that the
step response voltage has reduced to a voltage that is
smaller than the modulation voltage.
The required value for RC, or τ, can be calculated using
Equation 3, based on the following assumptions:
• The system needs to be able to start LF communications 200 µs after the resonant tank has
stabilized.
• The decoupler should settle to at least half the
data modulation voltage.
The base station typically uses On Off Keying (OOK)
modulation to communicate to the transponder. This
means the tank resonance is completely halted and
then started up to transfer data via the magnetic link.
The decoupling stage experiences large “step”
responses as data is transmitted to the transponder.
The tank can ramp up to its full resonant amplitude in
100 µs to 400 µs depending on the drive system used.
 2004 Microchip Technology Inc.
DS00912A-page 3
AN912
FIGURE 4:
160
140
120
Voltage
100
80
60
40
20
0
0
1
2
EQUATION 3:
τ =
4
5
EQUATION 4:
tSETTLE
ln(Vo/VSETTLE)
tSETTLE = 200 µS
VO = 150V
VSETTLE = 1 mV
Using Equation 3, τ was calculated to be 16.78 µs. The
question now is how will the data signal be affected by
the decoupling stage? The decoupling stage, shown in
Figure 3, is also a high-pass filter and it was calculated
that the RC time constant needs to be 16.78 µs to
satisfy the transient response requirement. The 3 dB
cutoff frequency, for a τ, of 16.78 µs is calculated as
9.48 kHz using Equation 4. This means that the
decoupling stage will only pass one quarter of the
original data signal at 2.5 kHz, which is not desirable
from a signal-to-noise ratio perspective.
DS00912A-page 4
τ
3
FC =
1
2πτ
From a data signal conservation, or high-pass filter
point-of-view, τ should be at least 64 µs. The conflicting
τ requirement shows that a basic high-pass filter is not
sufficient as a decoupler unless either dynamic
response or data signal strength is sacrificed.
 2004 Microchip Technology Inc.
AN912
AN IMPROVED DECOUPLER
FIGURE 6:
From the previous section, it is clear that a high-pass
filter is needed with either a controllable τ or a nonlinear
τ that is based on the voltage across the output of the
decoupler. Both approaches will be covered and the
latter solution is shown in Figure 5.
2.5V
HV-Env
S1
C
FIGURE 5:
R1
2.5V
HV-Env
-2.5V
R2
LP Filter
-2.5V
C
R
LP Filter
The addition of the two diodes, shown in Figure 5,
results in a nonlinear τ with respect to voltage because
it effectively lowers the R component of τ whenever the
voltage is either above 3.1V or below -3.1V. In a practical circuit, the diodes will start conducting when the
tank is turned on and the voltage, across the resistor R,
is around 3 volts, after the tank has stabilized.
Previously, τ was calculated with an initial voltage of
150 VDC, but if the calculation is repeated with an initial
voltage of 3 VDC, then the required τ comes to 25 µs.
The RC time constant is improved by a significant
factor from 16.78 µs to 25 µs with the additional diodes,
however, it is still not in the 64 µs ball park. The diodes
have the additional advantage in that they protect the
low-pass filter from the large positive and negative
voltages that develop across the resistor during tank
transient periods.
The final part to solving the time constant problem is to
add an additional resistor via a switch, as shown in
Figure 6. The switch is closed to reduce τ from 64 µs to
25 µs during transient periods and opened while data is
received via the LF Talkback link.
 2004 Microchip Technology Inc.
The final part of the decoupling stage is to lower the
output impedance by adding an active buffer in the
form of an inverting amplifier that has an input resistance equal to R1. The use of an inverting amplifier has
the additional advantages that it can add gain and a
single order low-pass filter to the decoupler, as shown
in Figure 7. The gain is equal to the ratio of R3/R1, and
the low pass cutoff frequency is set by R3 and C2, as
per Equation 4. The low pass cutoff frequency should
be chosen at least two decades above the main lowpass filter, otherwise it will have an undesirable effect
on the envelope response. For single ended 5V
designs, the gain should be limited to about 10 dB to
avoid amplifier saturation due to carrier ripple and data
modulation.
FIGURE 7:
C2
HV-Env
C1
S1
R1
R3
R2
Output
+
DS00912A-page 5
AN912
THE LOW PASS FILTER STAGE
The output signal from the decoupling stage consists of
the 125 kHz carrier ripple and the modulated data
signal, if one ignores the dynamic response signal. The
carrier ripple is about 300 mV peak-to-peak. The data
is 4 mV peak-to-peak with 6 dB of gain of the decoupler
and a cutoff frequency at about 10 kHz. The aim of the
low-pass filter stage is to amplify the data signal at
2.5 kHz and to filter out the carrier ripple in the most
effective manner.
The three most common active filter topologies used
are the Chebyshev, Butterworth and Bessel filters. The
Chebyshev filter has the steepest transition from pass
band to stop band, but has ripple in the pass band. The
Butterworth filters have the flattest pass band
response, but does not have such a steep transition as
the Chebyshev. The Bessel filter has a linear phase
response with a smooth transition from pass to stop
band. It seems the Chebyshev filter would best be
suited for this application, but the frequency response
does not tell the whole story.
The data signal is amplitude modulated and the tank
has steep transient response dynamics. As a result, the
filter should have a stable and flat transient response.
The Chebyshev filter has a very sharp frequency cutoff
response, but has the worst transient response of the
three filter topologies. The Chebyshev filter also has an
underdamped step response with overshoot and
ringing. The Butterworth filter has a better transient
response, but still some overshoot. The Bessel filter
has the worst response from a frequency perspective,
but has the best transient response as a result of its
linear phase characteristics. There are of course other
active filter topologies such as elliptical, state variable,
biquad and more, but a Bessel filter has adequate
performance for the application.
The filter gain is the final aspect to specifying the
Bessel filter. Using Microchip's FilterLab® program,
one can get the response for a unity gain – 2.5 kHz, 3d
order Bessel filter. At 125 kHz, the filter has 93 dB of
attenuation and the input ripple amplitude is 300 mV
peak-to-peak. Assuming the filter should have an
output ripple of no more then 1 mV peak-to-peak with
12 dB of headroom for noise, coupled through the
supply line, then one needs at least 62 dB of attenuation. This leaves 31 dB of allowable gain from the third
order filter. For the design, a gain of 20 or 26 dB was
chosen, leaving some additional headroom for ripple
rejection. The 3d order low-pass Bessel filter is shown
in Figure 8 and has a Fc = 2.5 kHz and 26 dB of gain.
Please note that the circuit shown in Figure 8 has a
fairly high output impedance at the data rate, but the
output of the filter will be driving a high-impedance
load, and this is therefore acceptable.
FIGURE 8:
78.7k
Input
16.5k
3.92k
10 nF
150 pF
+
Output
4.87k
10 nF
The data signal, in this example, has maximum
modulation frequency of 2.5 kHz or a TE of 200 µs. A
Bessel filter, with a cutoff frequency of 1/(2.2TE) =
2.27 kHz, would be ideal from a noise rejection point of
view, but a 2.5 kHz cutoff was chosen to minimize symbol overlap. The target is to design a filter with sufficient
performance using a single operational amplifier in
order to reduce the system cost. A dual operational
amplifier can then be used because the decoupling
stage also uses an amplifier. A third order Bessel filter
can now be implemented with the remaining amplifier.
DS00912A-page 6
 2004 Microchip Technology Inc.
AN912
DATA SLICER
AN EXAMPLE SYSTEM
The data slicer is essentially a comparator with some
input hysteresis voltage to reduce the influence of
noise. The overall system gain of the decoupler and the
low-pass filter, at 2.5 kHz, is about 29 dB or a factor of
28, and the system should be able to detect the 2 mV
data signal. The headroom between the hysteresis and
data signal was chosen to be about 9 dB or a factor of
2.8. This means that the minimum input voltage to
overcome the data slicer hysteresis is about 700 µV.
This translates to 20 mV of hysteresis for the data
slicer. Most comparators have some deliberate hysteresis to improve noise stability and this amount should
be extracted from the required hysteresis when calculating the amount of required feedback. Figure 9 shows
a typical hysteresis circuit and Equation 5 can be used
to calculate the amount of hysteresis for a single-ended
circuit.
A complete circuit with layout, based on the foregoing
design study, is shown in this section. The circuit
diagram is shown in Figure 11. The top and bottom
layout for the printed circuit board is shown in
Figure 12. The PIC16F648A was chosen for the
application, it has two comparators, a USART,
EEPROM and 4k of Flash program memory. The
PIC16F648A can be substituted with its smaller
program memory equivalents, the PIC16F627A or
PIC16F628A. The filter examples have been converted
to operate from a single 5 VDC supply. The 2.5 VDC
virtual ground is provided by the voltage divider
consisting of R23 and R24, shown in Figure 11. The
Reference voltage does not have to be actively
buffered, it is lightly loaded. A 0.1 µF decoupling
capacitor C10 is sufficient for noise reduction.
EQUATION 5:
R1
VHYST =
R2
VDD
FIGURE 9:
VREF
Input
-
Output
+
R1
R2
For example, if a comparator with 10 mV of offset and
hysteresis is used, then an additional 10 mV of hysteresis should be added. The resistor R2 is calculated to
be 5 MΩ for a VDD of 5 VDC and R1 = 10 kΩ.
 2004 Microchip Technology Inc.
A TC4422 FET driver, U1, drives the resonant tank
consisting of L1 and C2. The tank generates a strong
magnetic field and the voltage at the test pin TP1 can
reach 320V peak-to-peak. The main antenna, L1, is an
air-cored inductor with a 25 mm radius and 41 turns of
26-gage wire, and has a 162 µH inductance. The
inductor L2 and capacitors C3 and C4 are not populated and are added to the printed circuit board to test
alternative antennas. The peak detector consists of D1,
C5, R1, and R2, and is connected to the decoupling
stage via C6. The RC time constant of the decoupling
tank is set by C6 and R4 to 177 µs, which is substantially longer than the minimum filter requirement of
64 µs. Resistor R3 is used to change the decoupler's
time constant to 11 µs by changing RB7 from a highimpedance input to an output.
The decoupler buffer, U2:A, has a gain of 6 dB and a
low pass cutoff frequency at 9.8 kHz, set by R5 and C8.
The R22 resistor is used to ensure the proper DC bias
of the stage, but does not have a significant effect on
the overall sensitivity. The output of the decoupler is
connected to the input of the low pass Bessel filter and
one of the PIC16F648's comparators. The remaining
op amp, U2:B, is used for the Bessel filter. U2 is a dual
MCP6002 op amp that has a GBWP of 1 MHz. The
filter components should have better tolerances than
the high voltage components and 1% resistors. The 5%
NP0 capacitors are recommended.
DS00912A-page 7
AN912
The PIC16F648A has various comparator options.
Figure 10 shows the topology that was chosen for this
application. The main filter output signal “ENV_IN” is
connected to comparator C1 via RA0. Resistor R10
was placed in series with the output of the filter to have
10 kΩ impedance. Together with the 4.99 MΩ resistor,
R11 adds an additional 10 mV of hysteresis. The
comparator has a combined offset and hysteresis of
10 mV, in the worst case, making for a total of 20 mV of
hysteresis, in the worst case, and about 15 mV on
average. It should be noted that the output of comparator C1 has to be inverted by setting bit C1INV, in the
CMCON register. The output inversion is needed to
result in positive feedback, via R11, as is shown in
Figure 9. At first glance, it seems as if R10 can be
removed and R11 changed to a 2.43 MΩ resistor, but
the capacitor C12 will cause delay and that can lead to
instability.
FIGURE 10:
Two common Reference Comparators with Outputs
CM2:CM0 = 110
RA0/AN0
A
RA3/AN3/CMP1
D
RA1/AN1
A
RA2/AN2/VREF
A
RA4/T0CKI/CMP2
DS00912A-page 8
VIN-
VIN+ C1
+
C1VOUT
VIN- VIN+ C2
+
C2VOUT
Open Drain
There are additional aspects around the decoupler that
need to be explained for the system as it is implemented. The port pin RB7 is essentially an open circuit
when it is configured as an input and the input voltage
is between VDD (5V) and ground. All the general
purpose I/O pins have internal ESD protection diodes
that become conductive when a pin voltage is forced
outside the VDD to ground range. This has the effect
that the RC time constant for the decoupling stage is
reduced to 11 µs from 177 µs whenever the “BIAS”
signal is about 0.6V above VDD, or below ground even
if RB7 is configured as an input. The addition of R3
works well, but keep in mind, the stable DC voltage for
signal “A”, shown in Figure 11, is 2.5 VDC and the signal
“BIAS” is either 5V or ground. One can implement one
of two approaches to correctly bias the signal at point
“A”.
The first solution is to toggle RB7 between high and low
with a 50% duty cycle at 20 kHz or more. This is
equivalent to connecting the “BIAS” signal to the
desired 2.5 VDC. This is only done for a short period
after the tank is turned on or off, to force the decoupler
to stabilize faster than it would with just R4. The second
approach is to force the signal “A” in the required
direction. The voltage at “A” will go above VDD if the
tank is turned on after it has been turned off for some
time. The “BIAS” signal can be grounded during the
turn-on transient period until the voltage at point “A”
reaches the desired 2.5 VDC or VREF. By monitoring
either of the comparator output signals, it is possible to
detect when the voltage at point A goes through VREF.
Pin RB7 can be turned into an input as soon as the
cross over is detected resulting in a decoupler RC time
constant of 177 µs. The filters introduce delay that
cause some overshoot of the voltage at point “A”. The
overshoot can be resolved by allowing some additional
stabilizing time with R4, before LF communication is
interpreted as data.
 2004 Microchip Technology Inc.
AN912
SYSTEM MODIFICATIONS
INCREASED DATA SENSITIVITY
The system can be modified to better suit the user's
requirements. The first aspect is to change the Bessel
filter for a different LF Talkback TE. The rule of thumb is
to set the filter's 3 dB cutoff frequency to Fc = 1/(2*TE).
The new values for the Bessel filter, with a 400 µs TE,
is given in Table 1.
Increasing the system’s sensitivity to the modulated
data signal can increase the LF Talkback range. A solution has been partly described in the previous section;
increase TE from 200 µs to 400 µs and then increase
the gain by up to 18 dB. The component values for a
system with a 400 µs TE, or a center frequency of
1.25 kHz, and a gain of 100, or a 14 dB increase, is
described in Table 2 below. This approach decreases
the dynamic range that may or may not be used
depending on how well the transponder loads the
resonant tank.
TABLE 1:
R6 = 3.57k
R7 = 15.0k
R8 = 71.5k
R10 = 5.62k
C9 = C12 = 22 nF
C11 = 330 pF
R9 = 4.42k
In addition to changing the filter cutoff frequency for a
TE of 400 µs, it is possible to increase the gain up to
18 dB and still maintain the carrier rejection chosen. It
is also possible to increase C6 to a 4.7 nF capacitor,
but please note that this will increase the transient
response period. Increasing C6 will not have a
dramatic influence on the overall system performance
and it is not recommended.
LONGER TRANSIENT STABILIZING
PERIOD
The example system was designed with the requirement that LF Talkback communications should be able
to start 200 µS after the resonant tank has stabilized.
The tank itself takes 100 µS to 400 µs to stabilize
sufficiently, depending on the drive mechanism. The
example circuit should be able to start LF Talkback
communications with 2 mV of data modulation after
350 µs to 450 µs, from when the tank is turned. The
exact period depends on the residual charge in the
peak detector from previous transmissions.
The system can be simplified and improved if the
system allows for a longer transient stabilizing period
before LF Talkback communications are initiated. The
peak detector capacitor, C5, can be increased proportionally to the longer stabilizing period, but not by more
than a factor of about 3, otherwise it can influence data
modulation sensitivity. R1 and R2 tank resistors should
be reduced if C5 is increased, but not proportionally, it
will effect sensitivity. The combined value for R1 and
R2 should be no less then 4 MΩ.
Capacitor C6 can also be increased, but it will not have
a dramatic performance increase. The biggest advantage of a longer transient stabilizing period is that bias
resistor R3 can be increased. Increasing the value of
R3 will result in a slower change in the signal at point
“A”, which means the tank can be controlled more
accurately during the transient period.
 2004 Microchip Technology Inc.
TABLE 2:
R6 = 2.26k
R7 = 10.5k
R10 = 5.62k
C9 = 33 nF
R8 = 226k
R9 = 4.42k
C11 = 100 pF C12 = 22 nF
Another solution is to remove resistor R11 to get the
maximum sensitivity from the comparator, but this will
also increase noise in the data. Another quick solution
is to increase the gain of the decoupler buffer by up to
10 dB and lower the decoupler cutoff frequency by
about half the gain increase ratio.
The existing design makes use of a 3d order Bessel
filter. For improved noise reduction, increase the order
of the filter and add more gain per stage. This would
typically be done if a TE of 200 µs or 100 µs, is desired
with more sensitivity than can be reliably obtained with
the example system.
DRIVE SYSTEM
The example circuit uses a half-bridge driver based on
the TC range of FET drivers from Microchip. To
increase the transient response period of the tank, start
the tank in Full-bridge mode until the desired tank
amplitude is reached and the tank oscillation is maintained in Half-bridge mode. This method is described in
AN232 Low Frequency Magnetic Transmitter Design.
CONCLUSION
This LF Talkback Design application note can be used
to implement a cost-effective system to be used in
RFID, passive keyless entry and other bidirectional
transponder based technologies. The example circuit
can be used as a basis for further hardware and firmware development to suit the user's requirements.
DS00912A-page 9
DS00912A-page 10
R22
4.99M
2
VREF
80.6K
R4
4.99K
R3
GND
IN
4
VREF
TP3
5
U1
TC4422
OUT
BIAS
GND
VDD
6
3 +IN
2 -IN
A
4
VSS
VDD
8
R5
C2
+5V
R24
49.9K
R23
49.9K
C3
.200LS
TP4
VREF
3.92K
R6
COARSE_ENV_IN
A
HIGH-VOLTAGE SECTION
10 nF
400V
P3476-ND
MCP6002/SN
1
C7
0.1 uF
U2:A
OUT
+5V
100 pF
162K
C8
DO5022P
L2
10-00189
L1
TP1
10 nF
D1
R7
C10
0.1 uF
VREF
UF1005
16.5K
C4
.200LS
C9
A
TP2
5 +IN
6 -IN
78.7K
R8
C5
1.0 nF
500V
1412PH-ND
C11
OUT
150 pF
R2
4.99M
R1
4.99M
500V
U2:B
MCP6002/SN
7
2.2 nF
C6
R9
4.87K
A
TP5
C12
10 nF
5.11K
R10
4.99M
R11
TP6
TP7
ENV_IN
ENV_OUT
COARSE_ENV_OUT
Items labeled with B are socketed and
populated.
Items labeled with A are unpopulated.
Device names and numbers shown here
are for reference only and may differ from
the actual number.
FIGURE 11:
A
1
3
VDD
NOTES:
Unless otherwise specified;
Resistance values are in ohms.
Resistors are 1% tolerance.
Capacitance values are in uF.
SMT resistors are size 1206 and 1/8W.
APPENDIX A:
PWM
C1
0.1 uF
+5V
AN912
LF BASE STATION
SCHEMATICS
 2004 Microchip Technology Inc.
6
4
2
C17
1 uF
C18
1 uF
1K
R15
1K
R14
1
7
5
3
J2
4P-DIN
POWER DYNAMIC
MDC-034
+12V
RX
TX
C20
V+
R2OUT
R1OUT
6
V-
3 C1-
T1OUT
7
5
4
8
13
3 IN
C2-
C2+
R2IN
R1IN
T2OUT
14
U5
MAX232CPE
C13
560 uF
25V
P11220-ND
1 C1+
9
12
11 T1IN
10
T2IN
2
VCC 16
GND
 2004 Microchip Technology Inc.
15
GND
2
VR1
78L05
1 uF
C21
OUT
C19
1 uF
1
J1
9
8
7
6
C14
47 uF
10V
P11180-ND
DE-9S (FEM)
5
4
3
1
2
2
1
3
4
+5V
+5V
4.7K
D2
POWER
GRN
R12
270
MOM-NO
RESET
SW2
R20
+5V
PWM
TX
RX
RB0
MCLR
COARSE_ENV_OUT
ENV_OUT
VREF
RA2
B
U3
RB4
RB5
RB6
RB7
VDD
OSC2
OSC1
470
R21
RF_IN
C16
0.1 uF
MCLR
10
11
12
13
14
15
16
RA1 18
17
RA0
PIC16F648A/P
RB3
RB2
RB1
RBO/INT
VSS
MCLR
RA4/TO
C15
10 uF
6.3V
9
8
7
6
5
4
3
2 RA3
1
1
1K
R13
ANTENNA
WIRE
6.8"
A1
RF_IN
RB6
RB5
BIAS
OSC2
OSC1
2
1
NC
3
4
C24
0.1 uF
MOM-NO
SW1
+5V
COARSE_ENV_IN
ENV_IN
U4
DATA_IN
RF_GND
RF_+VCC
470
R18
AF_+VCC
15
RR8
433.92 MHz
AF_+VCC
13 TP
14 DATA_OUT
12
10 AF_+VCC
11 AF_GND
7 RF_GND
3
2
1
R19
4.7K
+5V
C22
20 pF
OSC2
OSC1
+5V
RB0
RB5
D3
RED
R16
270
BIAS
RB6
RB5
TX
RX
RBO
311-1153-1-ND
20.0 MHz
Y1
RB6
D4
GRN
R17
270
A J3
+5V
6
5
4
3
2
1
20 pF
C23
FIGURE 12:
1 uF
+5V
AN912
LF BASE STATION (Continued)
DS00912A-page 11
AN912
FIGURE 13:
BOTTOM SIDE
05-01 XXXX REV. A BOTTOM SIDE
FIGURE 14:
TOP SIDE
05-01 XXXX REV. A TOP SIDE
DS00912A-page 12
 2004 Microchip Technology Inc.
AN912
FIGURE 15:
TOP MASK
D2 J2
HIGH VOLTAGE
SECTION
U1
R12
J1
D1
D5
R15
R14
C17
C21 C20
U5
C13
C14
VR1
R19
TP1
U2
J3
R4
R3
L2
C3
C12
C7
C9
RB0
RB1
RB2
RB5
RB6
RB7
D4 R17
U4
A1
R18
U3 R9
C11
R8
R7
R6
R5
TP4
C8
TP3
TP6
TP5
C4
C5
R2
C10 R22
RB0
R11
R10
RESET R20
L1
TP7
R23 R24
C22 C23
C15
C16
R13 D3 R16
R21
POWER
FILTER
C2
C18 C19 RS232
C24
D1
R1
TP2
C6
05-01 XXXX REV. A TOP MASK
 2004 Microchip Technology Inc.
DS00912A-page 13
AN912
NOTES:
DS00912A-page 14
 2004 Microchip Technology Inc.
Note the following details of the code protection feature on Microchip devices:
•
Microchip products meet the specification contained in their particular Microchip Data Sheet.
•
Microchip believes that its family of products is one of the most secure families of its kind on the market today, when used in the
intended manner and under normal conditions.
•
There are dishonest and possibly illegal methods used to breach the code protection feature. All of these methods, to our
knowledge, require using the Microchip products in a manner outside the operating specifications contained in Microchip's Data
Sheets. Most likely, the person doing so is engaged in theft of intellectual property.
•
Microchip is willing to work with the customer who is concerned about the integrity of their code.
•
Neither Microchip nor any other semiconductor manufacturer can guarantee the security of their code. Code protection does not
mean that we are guaranteeing the product as “unbreakable.”
Code protection is constantly evolving. We at Microchip are committed to continuously improving the code protection features of our
products. Attempts to break Microchip’s code protection feature may be a violation of the Digital Millennium Copyright Act. If such acts
allow unauthorized access to your software or other copyrighted work, you may have a right to sue for relief under that Act.
Information contained in this publication regarding device
applications and the like is intended through suggestion only
and may be superseded by updates. It is your responsibility to
ensure that your application meets with your specifications.
No representation or warranty is given and no liability is
assumed by Microchip Technology Incorporated with respect
to the accuracy or use of such information, or infringement of
patents or other intellectual property rights arising from such
use or otherwise. Use of Microchip’s products as critical
components in life support systems is not authorized except
with express written approval by Microchip. No licenses are
conveyed, implicitly or otherwise, under any intellectual
property rights.
Trademarks
The Microchip name and logo, the Microchip logo, Accuron,
dsPIC, KEELOQ, MPLAB, PIC, PICmicro, PICSTART,
PRO MATE and PowerSmart are registered trademarks of
Microchip Technology Incorporated in the U.S.A. and other
countries.
AmpLab, FilterLab, microID, MXDEV, MXLAB, PICMASTER,
SEEVAL, SmartShunt and The Embedded Control Solutions
Company are registered trademarks of Microchip Technology
Incorporated in the U.S.A.
Application Maestro, dsPICDEM, dsPICDEM.net,
dsPICworks, ECAN, ECONOMONITOR, FanSense,
FlexROM, fuzzyLAB, In-Circuit Serial Programming, ICSP,
ICEPIC, microPort, Migratable Memory, MPASM, MPLIB,
MPLINK, MPSIM, PICkit, PICDEM, PICDEM.net, PICtail,
PowerCal, PowerInfo, PowerMate, PowerTool, rfLAB, rfPIC,
Select Mode, SmartSensor, SmartTel and Total Endurance
are trademarks of Microchip Technology Incorporated in the
U.S.A. and other countries.
Serialized Quick Turn Programming (SQTP) is a service mark
of Microchip Technology Incorporated in the U.S.A.
All other trademarks mentioned herein are property of their
respective companies.
© 2004, Microchip Technology Incorporated, Printed in the
U.S.A., All Rights Reserved.
Printed on recycled paper.
Microchip received ISO/TS-16949:2002 quality system certification for
its worldwide headquarters, design and wafer fabrication facilities in
Chandler and Tempe, Arizona and Mountain View, California in October
2003. The Company’s quality system processes and procedures are for
its PICmicro® 8-bit MCUs, KEELOQ® code hopping devices, Serial
EEPROMs, microperipherals, nonvolatile memory and analog
products. In addition, Microchip’s quality system for the design and
manufacture of development systems is ISO 9001:2000 certified.
 2004 Microchip Technology Inc.
DS00912A-page 15
WORLDWIDE SALES AND SERVICE
AMERICAS
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Korea
Corporate Office
Unit 706B
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Tel: 82-2-554-7200 Fax: 82-2-558-5932 or
82-2-558-5934
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Technical Support: 480-792-7627
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01/26/04
DS00912A-page 16
 2004 Microchip Technology Inc.