LINER LTC3547B

LTC3547B
Dual Monolithic
300mA Synchronous
Step-Down Regulator
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DESCRIPTIO
FEATURES
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High Efficiency (Up to 96%) Dual Step-Down
Outputs
300mA Output Current per Channel at VIN = 3V
2.25MHz Constant-Frequency Operation
2.5V to 5.5V Input Voltage Range
Low Dropout Operation: 100% Duty Cycle
No Schottky Diodes Required
Internally Compensated for All Ceramic Capacitors
Independent Internal Soft-Start for Each Channel
Available in Fixed Output Versions
Current Mode Operation for Excellent Line and Load
Transient Response
0.6V Reference Allows Low Output Voltages
Short-Circuit Protected
Ultralow Shutdown Current: IQ < 1μA
Low Profile (0.75mm) 8-Lead 3mm × 2mm
DFN Package
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APPLICATIO S
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The LTC®3547B is a dual, 2.25MHz, constant-frequency,
synchronous step-down DC/DC converter in a tiny 3mm ×
2mm DFN package. 100% duty cycle provides low dropout operation, extending battery life in portable systems.
Low output voltages are supported with the 0.6V feedback reference voltage. Each regulator can supply 300mA
continuous output current.
The input voltage range is 2.5V to 5.5V, making it ideal
for Li-Ion polymer and USB powered applications. Supply
current is only 1μA in shutdown.
An internally set 2.25MHz switching frequency allows the
use of tiny surface mount inductors and capacitors. Internal soft-start reduces inrush current during start-up. All
outputs are internally compensated to work with ceramic
capacitors. The LTC3547B is available in a low profile
(0.75mm) 3mm × 2mm DFN package. The LTC3547B
is also available in a fixed output voltage configuration
selected via internal resistor dividers (see Table 2).
, LT, LTC and LTM are registered trademarks of Linear Technology Corporation.
Burst Mode is a registered trademark of Linear Technology Corporation. All other
trademarks are the property of their respective owners. Protected by U.S. Patents,
including 6580258, 5481178, 6304066, 6127815, 6498466, 6611131.
Cellular Telephones
Digital Still Cameras
Wireless and DSL Modems
Portable Media Players
PDAs/Palmtop PCs
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TYPICAL APPLICATIO
Efficiency vs Output Current
for VOUT = 2.5V
100
Dual Monolithic Buck Regulator in 8-Lead 3mm × 2mm DFN
1
90
VIN
2.5V TO 5.5V
RUN2 VIN RUN1
4.7μH
VOUT2
1.8V AT
300mA
LTC3547B
SW2
4.7μH
SW1
10pF
10pF
VFB2
4.7μF
475k
237k
GND
VOUT1
2.5V AT
300mA
70
0.1
60
50
40
0.01
30
20
VFB1
150k
475k
4.7μF
VIN = 2.7V
VIN = 3.6V
VIN = 4.2V
10
0
3547b TA01
POWER LOSS (W)
4.7μF
EFFICIENCY (%)
80
1
10
100
OUTPUT CURRENT (mA)
0.001
1000
3547b TA01b
3547bfb
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LTC3547B
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ABSOLUTE
AXI U RATI GS
PIN CONFIGURATION
(Note 1)
TOP VIEW
VIN
LTC3547BE .............................................. –0.3V to 6V
LTC3547BI ............................................... –0.3V to 7V
VFB1, VFB2 ......................................... –0.3V to VIN +0.3V
RUN1, RUN2 ..................................... –0.3V to VIN +0.3V
SW1, SW2 (DC) ................................ –0.3V to VIN +0.3V
P-Channel Switch Source Current (DC)................500mA
N-Channel Switch Sink Current (DC) ...................500mA
Peak SW Sink and Source Current (Note 5) .........700mA
Ambient Operating Temperature Range (Note 2)
LTC3547BE .......................................... –40°C to 85°C
LTC3547BI ......................................... –40°C to 125°C
Maximum Junction Temperature........................... 125°C
Storage Temperature Range................... –65°C to 125°C
8 VFB2
VFB1 1
RUN1 2
VIN 3
9
SW1 4
7 RUN2
6 SW2
5 GND
DDB PACKAGE
8-LEAD (3mm × 2mm) PLASTIC DFN
TJMAX = 125°C, θJA = 76°C/W
EXPOSED PAD (PIN 9) IS GND
MUST BE SOLDERED TO PCB
ORDER INFORMATION
LEAD FREE FINISH
TAPE AND REEL
PART MARKING*
PACKAGE DESCRIPTION
TEMPERATURE RANGE
LTC3547BEDDB#PBF
LTC3547BEDDB#TRPBF
LCPD
8-Lead (3mm × 2mm) Plastic DFN
–40°C to 85°C
LTC3547BEDDB-1#PBF
LTC3547BEDDB-1#TRPBF LCPF
8-Lead (3mm × 2mm) Plastic DFN
–40°C to 85°C
LTC3547BIDDB #PBF
LTC3547BIDDB #TRPBF
8-Lead (3mm × 2mm) Plastic DFN
–40°C to 125°C
LCPD
Consult LTC Marketing for parts specified with wider operating temperature ranges. *Temperature grades are identified by a label on the shipping container.
Consult LTC Marketing for information on non-standard lead based finish parts.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
ELECTRICAL CHARACTERISTICS
The ● denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 3.6V, unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
VIN
VIN Operating Voltage
LTC3547BE, LTC3547BE-1,
LTC3547BI
●
●
VUV
VIN Undervoltage Lockout
VIN Low to High
●
IFB
Feedback Pin Input Current
LTC3547BE, VFB = VFBREG
LTC3547BE-1, VFB = VFBREG
●
●
VFBREG1
Regulated Feedback Voltage (VFB1)
VFBREG2
Regulated Feedback Voltage (VFB2)
LTC3547BE, 0°C ≤ TA ≤ 85°C
LTC3547BE, –40°C ≤ TA ≤ 85°C
LTC3547BI, –40°C ≤ TA ≤ 125°C
LTC3547BE-1, 0°C ≤ TA ≤ 85°C
LTC3547BE-1, –40°C ≤ TA ≤ 85°C
LTC3547BE, 0°C ≤ TA ≤ 85°C
LTC3547BE, –40°C ≤ TA ≤ 85°C
LTC3547BI, –40°C ≤ TA ≤ 125°C
LTC3547BE-1, 0°C ≤ TA ≤ 85°C
LTC3547BE-1, –40°C ≤ TA ≤ 85°C
MIN
●
●
●
●
●
●
TYP
2.5
2.5
MAX
UNITS
5.5
6.5
V
V
2.2
2.5
V
3
±30
6
nA
μA
0.590
0.588
0.588
1.770
1.764
0.600
0.600
0.600
1.800
1.800
0.610
0.612
0.612
1.830
1.836
V
V
V
V
V
0.590
0.588
0.588
1.180
1.176
0.600
0.600
0.600
1.200
1.200
0.610
0.612
0.612
1.220
1.224
V
V
V
V
V
3547bfb
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LTC3547B
ELECTRICAL CHARACTERISTICS
The ● denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VCC = 3.6V, unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
TYP
MAX
ΔVLINEREG
ΔVLOADREG
Reference Voltage Line Regulation
VIN = 2.5V to 5.5V
MIN
0.3
0.5
Output Voltage Load Regulation
ILOAD = 50mA to 300mA
0.5
IS
Input DC Supply Current
Active Mode (Note 4)
Shutdown
VFB1 = VFB2 = 0.95 × VFBREG
RUN1 = RUN2 = 0V, VIN = 5.5V
450
0.1
700
1
μA
μA
fOSC
Oscillator Frequency
VFB = 0.6V
1.8
2.25
2.7
MHz
ILIM
Peak Switch Current Limit
Channel 1 (300mA)
Channel 2 (300mA)
VIN = 2.5V, VFB < VFBREG , Duty Cycle < 35%
400
400
550
550
RDS(ON)
Channel 1 (Note 3)
Top Switch On-Resistance
Bottom Switch On-Resistance
Channel 2 (Note 3)
Top Switch On-Resistance
Bottom Switch On-Resistance
●
Ω
Ω
VIN = 3.6V, ISW = 100mA
VIN = 3.6V, ISW = 100mA
0.8
0.75
1.05
1.05
Ω
Ω
0.01
1
μA
0.650
0.850
ms
1
1.2
V
0.01
1
μA
tSOFTSTART
Soft-Start Time
VFB From 10% to 90% Full-Scale
VRUN
RUN Threshold High
●
IRUN
RUN Leakage Current
●
0.450
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TYPICAL PERFOR A CE CHARACTERISTICS
TA = 25°C unless otherwise specified.
Efficiency vs Input Voltage
VOUT
50mV/DIV
70
0.4
Note 4: Dynamic supply current is higher due to the internal gate charge
being delivered at the switching frequency.
Note 5: Guaranteed by long-term current density limitations.
Note 6: This IC includes overtemperature protection that is intended
to protect the device during momentary overload conditions. Junction
temperature will exceed 125°C when overtemperature protection is active.
Continuous operation above the specified maximum operating junction
temperature may impair device reliability.
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: The LTC3547BE is guaranteed to meet specified performance
from 0°C to 85°C. Specifications over the –40°C and 85°C operating
temperature range are assured by design, characterization and correlation
with statistical process controls. The LTC3547BI is guaranteed over the
–40°C to 125°C operating temperature range.
Note 3: The DFN switch on-resistance is guaranteed by correlation to
wafer level measurements.
90
Supply Current vs Temperature
400
VOUT = 1.8V
ILOAD = 0A
350
60
IOUT = 0.1mA
IOUT = 1mA
IOUT = 10mA
IOUT = 100mA
IOUT = 300mA
50
40
30
3547b G01
VIN = 3.6V
VOUT = 1.8V
ILOAD = 1mA
10μs/DIV
SUPPLY CURRENT (μA)
EFFICIENCY (%)
80
IL
50mA/DIV
mA
mA
1.05
1.05
VIN = 5V, VRUN = 0V
SW,
AC-COUPLED
5V/DIV
%
0.8
0.75
Switch Leakage Current
100
%/V
VIN = 3.6V, ISW = 100mA
VIN = 3.6V, ISW = 100mA
ISW(LKG)
Pulse Skip Mode Operation
UNITS
VIN = 5.5V
300
250
VIN = 2.7V
200
150
20
10
2.5
3
3.5
4.5
4
VIN (V)
5
5.5
3547b G02
100
–50
–25
50
25
0
TEMPERATURE (ºC)
75
100
3547b G03
3547bfb
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LTC3547B
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TYPICAL PERFOR A CE CHARACTERISTICS
Oscillator Frequency
vs Temperature
Switch Leakage vs Temperature
2.6
Switch Leakage vs Input Voltage
50
500
40
400
FREQUENCY (MHz)
2.3
LEAKAGE CURRENT (nA)
VIN = 4.2V
2.4
VIN = 3.6V
2.2
VIN = 2.7V
2.1
2.0
30
SYNCHRONOUS
SWITCH
20
10
75
50
25
TEMPERATURE (°C)
0
100
0
–50 –25
125
0
50
75
25
TEMPERATURE (°C)
300
SYNCHRONOUS
SWITCH
200
100
MAIN SWITCH
1.9
1.8
–50 –25
LEAKAGE CURRENT (pA)
2.5
MAIN SWITCH
100
0
2.5
125
3
3.5
4.5
4
VIN (V)
5
5.5
3547b G05
3547b G04
Reference Voltage
vs Temperature
3547b G06
RDS(ON) vs Input Voltage
1.5
RDS(ON) vs Temperature
1.0
1.3
MAIN SWITCH
1.0
0.9
0.5
0.8
6
VIN = 2.7V
1.2
0
0.7
–0.5
0.6
–1.0
0.5
1.0
RDS(ON) (Ω)
RDS(ON) (Ω)
VFB (% ERROR)
1.1
SYNCHRONOUS
SWITCH
VIN = 3.6V
0.9
0.8
0.7
0.6
–1.5
–50
0.5
–25
0
50
25
TEMPERATURE (°C)
75
0.4
2.5
100
3
3.5
4.5
4
VIN (V)
5
3547b G07
5.5
VOUT = 1.2V
90
80
80
70
70
60
50
40
10
0
0.1
125
3547b G09
VOUT = 1.8V
60
50
40
30
30
20
100
Efficiency vs Load Current
100
EFFICIENCY (%)
EFFICIENCY (%)
90
75
50
25
TEMPERATURE (°C)
0
3547b G08
Efficiency vs Load Current
100
0.4
–50 –25
6
MAIN SWITCH
SYNCHRONOUS
SWITCH
VIN = 4.2V
VIN = 2.7V
VIN = 3.6V
VIN = 4.2V
1
10
100
OUTPUT CURRENT (mA)
1000
3547b G10
20
10
0
0.1
VIN = 2.7V
VIN = 3.6V
VIN = 4.2V
10
100
1
OUTPUT CURRENT (mA)
1000
3547b G11
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LTC3547B
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TYPICAL PERFOR A CE CHARACTERISTICS
Efficiency vs Load Current
90
Load Regulation
1.2
VOUT = 2.5V
Line Regulation
VIN = 3.6V
1.O
VOUT ERROR (%)
EFFICIENCY (%)
80
70
60
50
40
30
0.6
VOUT = 1.2V
VOUT = 1.8V
VOUT = 2.5V
0.8
0.6
PULSE SKIP MODE
OPERATION
0.4
0.2
20
VIN = 2.7V
VIN = 3.6V
VIN = 4.2V
10
0
0.1
10
100
1
OUTPUT CURRENT (mA)
0.2
0
–0.2
–0.4
0
–0.6
2.5
–0.2
1000
VOUT = 1.8V
ILOAD = 100mA
0.4
VOUT ERROR (%)
100
0
50
3547b G12
150 200 250
LOAD CURRENT (mA)
100
300
350
3
3.5
4
VIN (V)
3547b G13
Start-Up From Shutdown
RUN
2V/DIV
VOUT
1V/DIV
VOUT
1V/DIV
IL
100mA/DIV
IL
200mA/DIV
5.5
Load Step
VOUT,
AC-COUPLED
100mV/DIV
IL
200mA/DIV
ILOAD
200mA/DIV
3547b G15
VIN = 3.6V
VOUT = 1.8V
ILOAD = 0A
5
3547b G14
Start-Up From Shutdown
RUN
2V/DIV
4.5
3547b G16
200μs/DIV
VIN = 3.6V
VOUT = 1.8V
ILOAD = 300mA
250μs/DIV
Load Step
3547b G17
10μs/DIV
VIN = 3.6V
VOUT = 1.8V
ILOAD = 0mA TO 300mA
Load Step
VOUT,
AC-COUPLED
100mV/DIV
VOUT,
AC-COUPLED
100mV/DIV
IL
200mA/DIV
IL
200mA/DIV
ILOAD
200mA/DIV
ILOAD
200mA/DIV
3547b G18
10μs/DIV
VIN = 3.6V
VOUT = 1.8V
ILOAD = 20mA TO 300mA
3547b G19
10μs/DIV
VIN = 3.6V
VOUT = 1.8V
ILOAD = 50mA TO 300mA
3547bfb
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LTC3547B
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PI FU CTIO S
VFB1 (Pin 1): Regulator 1 Output Feedback. Receives
the feedback voltage from the external resistor divider
across the regulator 1 output. Nominal voltage for this
pin is 0.6V.
RUN1 (Pin 2): Regulator 1 Enable. Forcing this pin to
VIN enables regulator 1, while forcing it to GND causes
regulator 1 to shut down.
VIN (Pin 3): Main Power Supply. Must be closely decoupled to GND.
SW1 (Pin 4): Regulator 1 Switch Node Connection to the
Inductor. This pin swings from VIN to GND.
GND (Pin 5): Ground. Connect to the (–) terminal of COUT,
and the (–) terminal of CIN.
SW2 (Pin 6): Regulator 2 Switch Node Connection to
the Inductor. This pin swings from VIN to GND.
RUN2 (Pin 7): Regulator 2 Enable. Forcing this pin to
VIN enables regulator 2, while forcing it to GND causes
regulator 2 to shut down.
VFB2 (Pin 8): Regulator 2 Output Feedback. Receives
the feedback voltage from the external resistor divider
across the regulator 2 output. Nominal voltage for this
pin is 0.6V.
Exposed Pad (Pin 9): Electrically Connected to GND.
Must be soldered to the PCB for optimum thermal
performance.
FUNCTIONAL DIAGRAM
REGULATOR 1
3 VIN
SLOPE
COMP
VFB1
–
–
1
EA
0.6V
SLEEP
ITH
VSLEEP
+
–
+
5Ω
ICOMP
+
PULSE SKIP
S
Q
RS
LATCH
SOFT-START
R
Q
SWITCHING
LOGIC
AND
BLANKING
CIRCUIT
ANTI
SHOOTTHRU
4 SW1
+
IRCMP
–
SHUTDOWN
5 GND
3547 FD
RUN1
2
RUN2
7
0.6V REF
OSC
OSC
VFB2
8
REGULATOR 2 (IDENTICAL TO REGULATOR 1)
6 SW2
3547bfb
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LTC3547B
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OPERATIO (Refer to Functional Diagram )
The LTC3547B uses a constant-frequency current mode
architecture. The operating frequency is set at 2.25MHz.
Both channels share the same clock and run in-phase.
The output voltage is set by an external resistor divider
returned to the VFB pins. An error amplifier compares the
divided output voltage with a reference voltage of 0.6V and
regulates the peak inductor current accordingly.
Main Control Loop
During normal operation, the top power switch (P-channel
MOSFET) is turned on at the beginning of a clock cycle
when the VFB voltage is below the reference voltage. The
current into the inductor and the load increases until the
peak inductor current (controlled by ITH) is reached. The
RS latch turns off the synchronous switch and energy
stored in the inductor is discharged through the bottom
switch (N-channel MOSFET) into the load until the next
clock cycle begins, or until the inductor current begins to
reverse (sensed by the IRCMP comparator).
The peak inductor current is controlled by the internally
compensated ITH voltage, which is the output of the error amplifier. This amplifier regulates the VFB pin to the
internal 0.6V reference by adjusting the peak inductor
current accordingly.
At very low load currents, the LTC3547B automatically
transitions from constant frequency operation to discontinuous operation, where the regulator begins skipping
pulses. Even though the total power loss decreases with
load, the efficiency of the switching regulator will drop,
because the power delivered to the output becomes very
small. For applications where light load efficiency is a
priority, consider using the LTC3547 instead.
Dropout Operation
When the input supply voltage decreases toward the output voltage the duty cycle increases to 100%, which is the
dropout condition. In dropout, the PMOS switch is turned
on continuously with the output voltage being equal to the
input voltage minus the voltage drops across the internal
P-channel MOSFET and the inductor.
An important design consideration is that the RDS(ON)
of the P-channel switch increases with decreasing input
supply voltage (see Typical Performance Characteristics).
Therefore, the user should calculate the worst-case power
dissipation when the LTC3547B is used at 100% duty cycle
with low input voltage (see Thermal Considerations in the
Applications Information Section).
Soft-Start
In order to minimize the inrush current on the input bypass capacitor, the LTC3547B slowly ramps up the output
voltage during start-up. Whenever the RUN1 or RUN2 pin is
pulled high, the corresponding output will ramp from zero
to full-scale over a time period of approximately 650μs. This
prevents the LTC3547B from having to quickly charge the
output capacitor and thus supplying an excessive amount
of instantaneous current.
Short-Circuit Protection
When either regulator output is shorted to ground, the
corresponding internal N-channel switch is forced on for
a longer time period for each cycle in order to allow the
inductor to discharge, thus preventing current runaway.
This technique has the effect of decreasing switching
frequency. Once the short is removed, normal operation
resumes and the regulator output will return to its nominal
voltage.
3547bfb
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LTC3547B
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APPLICATIO S I FOR ATIO
A general LTC3547B application circuit is shown in
Figure 1. External component selection is driven by the
load requirement, and begins with the selection of the
inductor L. Once the inductor is chosen, CIN and COUT
can be selected.
Inductor Selection
Although the inductor does not influence the operating frequency, the inductor value has a direct effect on
ripple current. The inductor ripple current ΔIL decreases
with higher inductance and increases with higher VIN
or VOUT :
IL =
VOUT
fO • L
V
• 1 OUT VIN (1)
Inductor Core Selection
Different core materials and shapes will change the
size/current and price/current relationship of an inductor. Toroid or shielded pot cores in ferrite or permalloy
materials are small and do not radiate much energy, but
generally cost more than powdered iron core inductors
with similar electrical characteristics. The choice of which
style inductor to use often depends more on the price vs
size requirements, and any radiated field/EMI requirements,
than on what the LTC3547B requires to operate. Table 1
shows some typical surface mount inductors that work
well in LTC3547B applications.
Table 1. Representative Surface Mount Inductors
MANUFACTURER
Accepting larger values of ΔIL allows the use of low
inductances, but results in higher output voltage ripple,
greater core losses, and lower output current capability.
A reasonable starting point for setting ripple current
is 40% of the maximum output load current. So, for a
300mA regulator, ΔIL = 120mA (40% of 300mA).
LTC3547B
L2
CF2
COUT2
R4
R3
SW2
SW1
VFB2
VFB1
GND
R1
R2
VOUT1
COUT1
HEIGHT
2.2μH
2.2μH
3.3μH
510mA
530mA
410mA
0.13Ω 1.6mm
0.33Ω 1.25mm
0.27Ω 1.6mm
Panasonic
ELT5KT4R7M
4.7μH
950mA
0.2Ω
1.2mm
Sumida
CDRH2D18/LD
4.7μH
630mA
0.086Ω
2mm
Murata
LQH32CN4R7M23
4.7μH
450mA
0.2Ω
2mm
Taiyo Yuden
NR30102R2M
NR30104R7M
2.2μH
4.7μH
1100mA
750mA
0.1Ω
0.19Ω
1mm
1mm
FDK
FDKMIPF2520D
FDKMIPF2520D
FDKMIPF2520D
4.7μH
3.3μH
2.2μH
1100mA
1200mA
1300mA
0.11Ω
0.1Ω
0.08Ω
1mm
1mm
1mm
TDK
VLF3010AT4R7MR70
VLF3010AT3R3MR87
VLF3010AT2R2M1RD
4.7μH
700mA
0.24Ω
1mm
3.3μH
870mA
0.17Ω
1mm
2.2μH
1000mA
0.12Ω
1mm
L1
CF1
DCR
CB2016T2R2M
CB2012T2R2M
CB2016T3R3M
RUN2 VIN RUN1
VOUT2
MAX DC
VALUE CURRENT
Taiyo Yuden
VIN
2.5V TO 5.5V
C1
PART NUMBER
3547b F01
Figure 1. LTC3547B General Schematic
3547bfb
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LTC3547B
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APPLICATIO S I FOR ATIO
Input Capacitor (CIN) Selection
In continuous mode, the input current of the converter
is a square wave with a duty cycle of approximately
VOUT/VIN. To prevent large voltage transients, a low equivalent series resistance (ESR) input capacitor sized for
the maximum RMS current must be used. The maximum RMS capacitor current is given by:
IRMS ≈ IMAX
VOUT (VIN − VOUT )
VIN
(2)
Where the maximum average output current IMAX equals
the peak current minus half the peak-to-peak ripple current, IMAX = ILIM – ΔIL/2.
This formula has a maximum at VIN = 2VOUT, where IRMS
= IOUT/2. This simple worst-case is commonly used to
design because even significant deviations do not offer
much relief. Note that capacitor manufacturer’s ripple current ratings are often based on only 2000 hours lifetime.
This makes it advisable to further derate the capacitor,
or choose a capacitor rated at a higher temperature than
required. Several capacitors may also be paralleled to meet
the size or height requirements of the design. An additional 0.1μF to 1μF ceramic capacitor is also recommended
on VIN for high frequency decoupling when not using an
all-ceramic capacitor solution.
Output Capacitor (COUT) Selection
The selection of COUT is driven by the required effective
series resistance (ESR). Typically, once the ESR requirement for COUT has been met, the RMS current rating
generally far exceeds the IRIPPLE(P-P) requirement. The
output ripple ΔVOUT is determined by:
1 VOUT IL ESR +
8fCOUT (3)
where f = operating frequency, COUT = output capacitance
and ΔIL = ripple current in the inductor. For a fixed output
voltage, the output ripple is highest at maximum input
voltage since ΔIL increases with input voltage.
If tantalum capacitors are used, it is critical that the capacitors are surge tested for use in switching power supplies.
An excellent choice is the AVX TPS series of surface mount
tantalum. These are specially constructed and tested for low
ESR so they give the lowest ESR for a given volume. Other
capacitor types include Sanyo POSCAP, Kemet T510 and
T495 series, and Sprague 593D and 595D series. Consult
the manufacturer for other specific recommendations.
Using Ceramic Input and Output Capacitors
Higher values, lower cost ceramic capacitors are now
becoming available in smaller case sizes. Their high ripple
current, high voltage rating and low ESR make them
ideal for switching regulator applications. Because the
LTC3547B control loop does not depend on the output
capacitor’s ESR for stable operation, ceramic capacitors
can be used freely to achieve very low output ripple and
small circuit size.
However, care must be taken when ceramic capacitors are
used at the input. When a ceramic capacitor is used at the
input and the power is supplied by a wall adapter through
long wires, a load step at the output can induce ringing at
the input, VIN. At best, this ringing can couple to the output
and be mistaken as loop instability. At worst, a sudden
inrush of current through the long wires can potentially
cause a voltage spike at VIN, large enough to damage the
part. For more information, see Application Note 88.
When choosing the input and output ceramic capacitors,
choose the X5R or X7R dielectric formulations. These
dielectrics have the best temperature and voltage characteristics of all the ceramics for a given value and size.
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LTC3547B
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Setting the Output Voltage
The LTC3547B regulates the VFB1 and VFB2 pins to 0.6V
during regulation. Thus, the output voltage is set by a
resistive divider according to the following formula:
R2 VOUT = 0.6V 1+ R1
(4)
Keeping the current small (< 5μA) in these resistors maximizes efficiency, but making it too small may allow stray
capacitance to cause noise problems or reduce the phase
margin of the error amp loop.
To improve the frequency response of the main control
loop, a feedback capacitor (CF) may also be used. Great
care should be taken to route the VFB line away from noise
sources, such as the inductor or the SW line.
Fixed output versions of the LTC3547B (e.g. LTC3547B-1)
include an internal resistive divider, eliminating the need
for external resistors. The resistor divider is chosen such
that the VFB input current is 3μA. For these versions, the
VFB pin should be connected directly to VOUT. Table 2 lists
the fixed output voltages available for the LTC3547B.
Table 2. Fixed Output Voltage Versions
Part Number
LTC3547B
LTC3547B-1
VOUT1
VOUT2
Adjustable
Adjustable
1.8V
1.2V
Checking Transient Response
The regulator loop response can be checked by looking
at the load transient response. Switching regulators take
several cycles to respond to a step in load current. When
a load step occurs, VOUT immediately shifts by an amount
equal to ΔILOAD • ESR, where ESR is the effective series
resistance of COUT. ΔILOAD also begins to charge or discharge COUT generating a feedback error signal used by the
regulator to return VOUT to its steady-state value. During
this recovery time, VOUT can be monitored for overshoot
or ringing that would indicate a stability problem.
The initial output voltage step may not be within the bandwidth of the feedback loop, so the standard second-order
overshoot/DC ratio cannot be used to determine the phase
margin. In addition, feedback capacitors (CF1 and CF2)
can be added to improve the high frequency response, as
shown in Figure 1. Capacitor CF provides phase lead by
creating a high frequency zero with R2 which improves
the phase margin.
The output voltage settling behavior is related to the stability
of the closed-loop system and will demonstrate the actual
overall supply performance. For a detailed explanation of
optimizing the compensation components, including a review of control loop theory, refer to Application Note 76.
In some applications, a more severe transient can be
caused by switching in loads with large (>1μF) input capacitors. The discharged input capacitors are effectively
put in parallel with COUT, causing a rapid drop in VOUT.
No regulator can deliver enough current to prevent this
problem if the switch connecting the load has low resistance
and is driven quickly. The solution is to limit the turn-on
speed of the load switch driver. A Hot Swap™ controller
is designed specifically for this purpose and usually incorporates current limiting, short-circuit protection, and
soft-starting.
Hot Swap is a trademark of Linear Technology Corporation.
3547bfb
10
LTC3547B
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Efficiency Considerations
The percent efficiency of a switching regulator is equal to
the output power divided by the input power times 100%.
It is often useful to analyze individual losses to determine
what is limiting the efficiency and which change would
produce the most improvement. Percent efficiency can
be expressed as:
% Efficiency = 100% – (L1 + L2 + L3 + ...)
where L1, L2, etc., are the individual losses as a percentage of input power.
Although all dissipative elements in the circuit produce
losses, four sources usually account for the losses in
LTC3547B circuits: 1) VIN quiescent current, 2) switching
losses, 3) I2R losses, 4) other system losses.
1) The VIN current is the DC supply current given in the
Electrical Characteristics which excludes MOSFET
driver and control currents. VIN current results in a
small (<0.1%) loss that increases with VIN, even at
no load.
2) The switching current is the sum of the MOSFET driver
and control currents. The MOSFET driver current results from switching the gate capacitance of the power
MOSFETs. Each time a MOSFET gate is switched from
low to high to low again, a packet of charge dQ moves
from VIN to ground. The resulting dQ/dt is a current out
of VIN that is typically much larger than the DC bias current. In continuous mode, IGATECHG = fO(QT + QB), where
QT and QB are the gate charges of the internal top and
bottom MOSFET switches. The gate charge losses are
proportional to VIN and thus their effects will be more
pronounced at higher supply voltages.
3) I2R losses are calculated from the DC resistances
of the internal switches, RSW, and external inductor,
RL. In continuous mode, the average output current
flows through inductor L, but is “chopped” between
the internal top and bottom switches. Thus, the series
resistance looking into the SW pin is a function of both
top and bottom MOSFET RDS(ON) and the duty cycle
(DC) as follows:
RSW = (RDS(ON)TOP) • (DC) + (RDS(ON)BOT) • (1– DC)
(5)
The RDS(ON) for both the top and bottom MOSFETs can
be obtained from the Typical Performance Characteristics curves. Thus, to obtain I2R losses:
I2R losses = IOUT2 • (RSW + RL)
4) Other “hidden” losses, such as copper trace and internal battery resistances, can account for additional
efficiency degradations in portable systems. It is very
important to include these “system” level losses in
the design of a system. The internal battery and fuse
resistance losses can be minimized by making sure that
CIN has adequate charge storage and very low ESR at
the switching frequency. Other losses, including diode
conduction losses during dead-time, and inductor
core losses, generally account for less than 2% total
additional loss.
3547bfb
11
LTC3547B
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Thermal Considerations
PC Board Layout Considerations
In a majority of applications, the LTC3547B does not dissipate much heat due to its high efficiency. In the unlikely
event that the junction temperature somehow reaches
approximately 150°C, both power switches will be turned
off and the SW node will become high impedance.
When laying out the printed circuit board, the following
checklist should be used to ensure proper operation of
the LTC3547B. These items are also illustrated graphically
in the layout diagrams of Figures 2 and 3. Check the following in your layout:
The goal of the following thermal analysis is to determine
whether the power dissipated causes enough temperature
rise to exceed the maximum junction temperature (125°C)
of the part. The temperature rise is given by:
1. Does the capacitor CIN connect to the power VIN (Pin 3)
and GND (Pin 5) as closely as possible? This capacitor
provides the AC current of the internal power MOSFETs
and their drivers.
TRISE = PD • θJA
(6)
Where PD is the power dissipated by the regulator and
θJA is the thermal resistance from the junction of the die
to the ambient temperature.
The junction temperature, TJ, is given by:
TJ = TRISE + TAMBIENT
(7)
As a worst-case example, consider the case when the
LTC3547B is in dropout on both channels at an input
voltage of 2.7V with a load current of 300mA and an ambient temperature of 70°C. From the Typical Performance
Characteristics graph of Switch Resistance, the RDS(ON)
of the main switch is 0.9Ω. Therefore, power dissipated
by each channel is:
PD = IOUT2 • RDS(ON) = 81mV
Given that the thermal resistance of a properly soldered
DFN package is approximately 76°C/W, the junction
temperature of an LTC3547B device operating in a 70°C
ambient temperature is approximately:
TJ = (2 • 0.081W • 76°C/W) + 70°C = 82.3°C
which is well below the absolute maximum junction temperature of 125°C.
2. Are the respective COUT and L closely connected?
The (–) plate of COUT returns current to GND and the
(–) plate of CIN.
3. The resistor divider, R1 and R2, must be connected
between the (+) plate of COUT1 and a ground sense line
terminated near GND (Pin 5). The feedback signals VFB1
and VFB2 should be routed away from noisy components
and traces, such as the SW lines (Pins 4 and 6), and
their trace length should be minimized.
4. Keep sensitive components away from the SW pins if
possible. The input capacitor CIN and the resistors R1,
R2, R3 and R4 should be routed away from the SW
traces and the inductors.
5. A ground plane is preferred, but if not available, keep
the signal and power grounds segregated with small
signal components returning to the GND pin at a single
point. These ground traces should not share the high
current path of CIN or COUT.
6. Flood all unused areas on all layers with copper. Flooding with copper will reduce the temperature rise of
power components. These copper areas should be
connected to VIN or GND.
3547bfb
12
LTC3547B
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APPLICATIO S I FOR ATIO
VIN
2.5V TO 5.5V
C1
RUN2 VIN RUN1
LTC3547B
L2
VOUT2
SW2
L1
VOUT1
SW1
CF2
CF1
VFB2
R4
COUT2
R3
GND
VFB1
R2
R1
COUT1
3547b F02
BOLD LINES INDICATE HIGH CURRENT PATHS
Figure 2. LTC3547B Layout Diagram (See Board Layout Checklist)
R1
VIA TO GND
VFB1
R3
VFB2
R2
R4
CF2
CF2
VIA TO GND
VIA TO VOUT1
VIA TO VOUT2
VIA TO VIN
VIN
SW1
L1
SW2
L2
CIN
GND
COUT2
VOUT2
COUT1
3547b F03
VOUT1
Figure 3. LTC3547B Suggested Layout
3547bfb
13
LTC3547B
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Design Example
The feedback resistors program the output voltage. To
maintain high efficiency at light loads, the current in these
resistors should be kept small. Choosing 2μA with the
0.6V feedback voltage makes R1~300k. A close standard
1% resistor is 280k, using Equation 4.
As a design example, consider using the LTC3547B in
a portable application with a Li-Ion battery. The battery
provides a VIN ranging from 2.8V to 4.2V. The load on each
channel requires a maximum of 300mA in active mode
and 2mA in standby mode. The output voltages are VOUT1
= 2.5V and VOUT2 = 1.8V.
V
R2 = OUT 1 • R1= 887k
0.6
Start with channel 1. First, calculate the inductor value
for about 40% ripple current (120mA in this example) at
maximum VIN. Using a derivation of Equation 1:
An optional 10pF feedback capacitor (CF1) may be used
to improve transient response.
2.5V
2.5V = 3.75μH
L1=
• 1
2.25MHz • (120mA) 4.2V Using the same analysis for channel 2 (VOUT2 = 1.8V),
the results are:
L2 = 3.81μH
For the inductor, use the closest standard value of 4.7μH.
A 4.7μF capacitor should be more than sufficient for this
output capacitor. As for the input capacitor, a typical value
of CIN = 4.7μF should suffice, as the source impedance of
a Li-Ion battery is very low.
VIN
2.5V TO 5.5V
C1
4.7μF
R4 = 560k
Figure 4 shows the complete schematic for this example,
along with the efficiency curve and transient response.
RUN2 VIN RUN1
L2
4.7μH
VOUT2
1.8V AT 300mA
R3 = 280k
L1
4.7μH
LTC3547B
SW2
SW1
CF2, 10pF
CF1, 10pF
VFB2
COUT2
4.7μF
R4
562k
R3
280k
GND
VOUT1
2.5V AT 300mA
VFB1
R1
280k
R2
887k
COUT1
4.7μF
3547b F04a
C1, C2, C3: TAIYO YUDEN JMK316BJ475ML
L1, L2: MURATA LQH32CN4R7M33
Figure 4a. Design Example Circuit
100
100
VOUT = 1.8V
90
80
80
70
70
EFFICIENCY (%)
EFFICIENCY (%)
90
60
50
40
30
20
10
0
0.1
VOUT = 2.5V
60
50
40
30
VIN = 2.7V
VIN = 3.6V
VIN = 4.2V
1
10
100
OUTPUT CURRENT (mA)
1000
20
VIN = 2.7V
VIN = 3.6V
VIN = 4.2V
10
0
0.1
10
100
1
OUTPUT CURRENT (mA)
1000
3547b F04b
Figure 4b. Efficiency vs Output Current
3547bfb
14
LTC3547B
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APPLICATIO S I FOR ATIO
VOUT,
AC-COUPLED
100mV/DIV
VOUT,
AC-COUPLED
100mV/DIV
IL
200mA/DIV
IL
200mA/DIV
ILOAD
200mA/DIV
ILOAD
200mA/DIV
3547b F04c
10μs/DIV
VIN = 3.6V
VOUT = 1.8V
ILOAD = 20mA TO 300mA
10μs/DIV
VIN = 3.6V
VOUT = 2.5V
ILOAD = 20mA TO 300mA
Figure 4c. Transient Response
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PACKAGE DESCRIPTIO
DDB Package
8-Lead Plastic DFN (3mm × 2mm)
(Reference LTC DWG # 05-08-1702 Rev B)
0.61 ±0.05
(2 SIDES)
3.00 ±0.10
(2 SIDES)
R = 0.115
TYP
5
R = 0.05
TYP
0.40 ± 0.10
8
0.70 ±0.05
2.55 ±0.05
1.15 ±0.05
PACKAGE
OUTLINE
0.25 ± 0.05
0.50 BSC
2.20 ±0.05
(2 SIDES)
PIN 1 BAR
TOP MARK
(SEE NOTE 6)
0.200 REF
2.00 ±0.10
(2 SIDES)
0.56 ± 0.05
(2 SIDES)
0.75 ±0.05
0 – 0.05
RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS
4
0.25 ± 0.05
1
PIN 1
R = 0.20 OR
0.25 × 45°
CHAMFER
(DDB8) DFN 0905 REV B
0.50 BSC
2.15 ±0.05
(2 SIDES)
BOTTOM VIEW—EXPOSED PAD
NOTE:
1. DRAWING CONFORMS TO VERSION (WECD-1) IN JEDEC PACKAGE OUTLINE M0-229
2. DRAWING NOT TO SCALE
3. ALL DIMENSIONS ARE IN MILLIMETERS
4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE
MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE
5. EXPOSED PAD SHALL BE SOLDER PLATED
6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON THE TOP AND BOTTOM OF PACKAGE
3547bfb
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
15
LTC3547B
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TYPICAL APPLICATIO
Dual 300mA Buck Converter with Pulse Skip Mode
VIN
2.5V TO 5.5V
C1
4.7μF
VOUT2
1.8V AT 300mA
VIN
2.5V TO 5.5V
L2
4.7μH
CF2, 10pF
R3
280k
VOUT1
2.5V AT 300mA
SW1
LTC3547B
VFB2
R4
562k
L1
4.7μH
RUN2 VIN RUN1
SW2
COUT2
4.7μF
1.8V/1.2V Dual 300mA Buck Converter with Pulse Skip Mode
GND
CF1, 10pF
L2
4.7μH
L1
4.7μH
RUN2 VIN RUN1
SW2
VOUT1
1.8V AT 300mA
SW1
LTC3547B-1
VFB1
R1
280k
VOUT2
1.2V AT 300mA
C1
4.7μF
R2
887k
COUT2
4.7μF
COUT1
4.7μF
VFB2
GND
VFB1
COUT1
4.7μF
3547b TA02
C1, COUT1, COUT2: TAIYO YUDEN JMK316BJ475ML
L1, L2: MURATA LQH32CN4R7M33
3547b TA03
C1, COUT1, COUT2: TAIYO YUDEN JMK316BJ475ML
L1, L2: MURATA LQH32CN4R7M33
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ThinSOT is a trademark of Linear Technology Corporation
3547bfb
16 Linear Technology Corporation
LT 0807 REV B • PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 ● FAX: (408) 434-0507
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