NSC LM3478_09

LM3478/LM3478Q
High Efficiency Low-Side N-Channel Controller for
Switching Regulator
General Description
Features
The LM3478 is a versatile Low-Side N-Channel MOSFET
controller for switching regulators. It is suitable for use in
topologies requiring a low side MOSFET, such as boost, flyback, SEPIC, etc. Moreover, the LM3478 can be operated at
extremely high switching frequency in order to reduce the
overall solution size. The switching frequency of the LM3478
can be adjusted to any value between 100kHz and 1MHz by
using a single external resistor. Current mode control provides superior bandwidth and transient response, besides
cycle-by-cycle current limiting. Output current can be programmed with a single external resistor.
The LM3478 has built in features such as thermal shutdown,
short-circuit protection, over voltage protection, etc. Power
saving shutdown mode reduces the total supply current to
5µA and allows power supply sequencing. Internal soft-start
limits the inrush current at start-up.
■ LM3478Q is AEC-Q100 qualified and manufactured on an
Key Specifications
■
■
■
■
Wide supply voltage range of 2.97V to 40V
100kHz to 1MHz Adjustable clock frequency
±2.5% (over temperature) internal reference
10µA shutdown current (over temperature)
Automotive Grade Flow
8-lead Mini-SO8 (MSOP-8) package
Internal push-pull driver with 1A peak current capability
Current limit and thermal shutdown
Frequency compensation optimized with a capacitor and
a resistor
■ Internal softstart
■ Current Mode Operation
■ Undervoltage Lockout with hysteresis
■
■
■
■
Applications
■
■
■
■
■
Distributed Power Systems
Battery Chargers
Offline Power Supplies
Telecom Power Supplies
Automotive Power Systems
Typical Application Circuit
10135501
Typical High Efficiency Step-Up (Boost) Converter
© 2009 National Semiconductor Corporation
101355
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LM3478/LM3478Q High Efficiency Low-Side N-Channel Controller for Switching Regulator
March 30, 2009
LM3478/LM3478Q
Connection Diagram
10135502
8 Lead Mini SO8 Package (MSOP-8 Package)
Package Marking and Ordering Information
Order Number
Package Type
LM3478MM
LM3478MMX
LM3478QMM
LM3478QMMX
Package Marking
MSOP-8
S14B
MSOP-8
SSFB
Supplied As:
Feature
1000 units on Tape and Reel
3500 units on Tape and Reel
1000 units on Tape and Reel
3500 units on Tape and Reel
AEC-Q100 (Grade 1) qualified.
Automotive Grade Production Flow*
* Automotive Grade (Q) product incorporates enhanced manufacturing and support processes for the automotive market, including defect detection methodologies.
Reliability qualification is compliant with the requirements and temperature grades defined in the AEC-Q100 standard. Automotive grade products are identified
with the letter Q. For more information go to http://www.national.com/automotive.
Pin Descriptions
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Pin Name
Pin Number
Description
ISEN
1
Current sense input pin. Voltage generated across an external sense
resistor is fed into this pin.
COMP
2
Compensation pin. A resistor, capacitor combination connected to this
pin provides compensation for the control loop.
FB
3
Feedback pin. The output voltage should be adjusted using a resistor
divider to provide 1.26V at this pin.
AGND
4
Analog ground pin.
PGND
5
Power ground pin.
DR
6
Drive pin. The gate of the external MOSFET should be connected to
this pin.
FA/SD
7
Frequency adjust and Shutdown pin. A resistor connected to this pin
sets the oscillator frequency. A high level on this pin for longer than
30 µs will turn the device off. The device will then draw less than 10µA
from the supply.
VIN
8
Power Supply Input pin.
2
If Military/Aerospace specified devices are required,
please contact the National Semiconductor Sales Office/
Distributors for availability and specifications.
Input Voltage
FB Pin Voltage
FA/SD Pin Voltage
Peak Driver Output Current (<10µs)
Power Dissipation
Storage Temperature Range
Junction Temperature
ESD Susceptibilty
Human Body Model (Note 2)
45V
-0.4V < VFB < 7V
-0.4V < VFA/SD < 7V
1.0A
Internally Limited
−65°C to +150°C
+150°C
215°C
260°C
−0.4V ≤ VDR ≤ 8V
500mV
ISEN Pin Voltage
Operating Ratings
(Note 1)
2.97V ≤ VIN ≤ 40V
Supply Voltage
Junction Temperature
Range
Switching Frequency
−40°C ≤ TJ ≤ +125°C
100kHz ≤ FSW ≤ 1MHz
2kV
Electrical Characteristics
Specifications in Standard type face are for TJ = 25°C, and in bold type face apply over the full Operating Temperature
Range. Unless otherwise specified, VIN = 12V, RFA = 40kΩ
Symbol
VFB
Parameter
Feedback Voltage
Conditions
VCOMP = 1.4V,
Typical
Limit
Units
1.2416/1.228
1.2843/1.292
V
V(min)
V(max)
1.26
2.97 ≤ VIN ≤ 40V
ΔVLINE
Feedback Voltage Line
Regulation
2.97 ≤ VIN ≤ 40V
0.001
%/V
ΔVLOAD
Output Voltage Load
Regulation
IEAO Source/Sink
±0.5
%/V (max)
VUVLO
Input Undervoltage Lock-out
VUV(HYS)
Fnom
2.85
Input Undervoltage Lock-out
Hysteresis
170
Nominal Switching Frequency RFA = 40KΩ
400
2.97
V
V(max)
130
210
mV
mV (min)
mV (max)
350
440
kHz
kHz(min)
kHz(max)
RDS1 (ON)
Driver Switch On Resistance
(top)
IDR = 0.2A, VIN= 5V
16
Ω
RDS2 (ON)
Driver Switch On Resistance
(bottom)
IDR = 0.2A
4.5
Ω
VDR (max)
Maximum Drive Voltage
Swing(Note 6)
VIN < 7.2V
VIN
V
VIN ≥ 7.2V
7.2
Dmax
Maximum Duty Cycle(Note 7)
100
Tmin (on)
Minimum On Time
325
ISUPPLY
IQ
VSENSE
VSC
Supply Current (nonswitching)
(Note 9)
Quiescent Current in
Shutdown Mode
VFA/SD = 5V (Note 10),
VIN = 5V
Current Sense Threshold
Voltage
VIN = 5V
Short-Circuit Current Limit
Sense Voltage
VIN = 5V
2.7
%
210
600
nsec
nsec(min)
nsec(max)
3.3
mA
mA (max)
10
µA
µA (max)
135/ 125
180/ 190
mV
mV (min)
mV (max)
250
415
mV
mV (min)
mV (max)
5
156
343
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LM3478/LM3478Q
Lead Temperature
MM Package
Vapor Phase (60 sec.)
Infared (15 sec.)
DR Pin Voltage
Absolute Maximum Ratings (Note 1)
LM3478/LM3478Q
Symbol
VSL
Parameter
Conditions
Typical
Internal Compensation Ramp VIN = 5V
Voltage
92
VOVP
Output Over-voltage
Protection (with respect to
feedback voltage) (Note 8)
VCOMP = 1.4V
50
VOVP(HYS)
Output Over-Voltage
VCOMP = 1.4V
Protection Hysteresis(Note 8)
60
Gm
Error Ampifier
Transconductance
VCOMP = 1.4V
IEAO = 100µA (Source/Sink)
800
AVOL
Error Amplifier Voltage Gain
VCOMP = 1.4V
IEAO = 100µA (Source/Sink)
38
IEAO
Units
52
132
mV
mV(min)
mV(max)
32/ 25
78/ 85
20
110
Error Amplifier Output Current Source, VCOMP = 1.4V, VFB = 0V
(Source/ Sink)
Sink, VCOMP = 1.4V, VFB = 1.4V
VEAO
Limit
Error Amplifier Output Voltage Upper Limit
Swing
VFB = 0V
COMP Pin = Floating
Lower Limit
VFB = 1.4V
600/ 365
1000/ 1265
mV
mV(min)
mV(max)
mV
mV(min)
mV(max)
µmho
µmho (min)
µmho (max)
26
44
V/V
V/V (min)
V/V (max)
80/ 50
140/ 180
µA
µA (min)
µA (max)
−100/ −85
−180/ −185
µA
µA (min)
µA (max)
1.8
2.4
V
V(min)
V(max)
0.2
1.0
V
V(min)
V(max)
110
−140
2.2
0.56
TSS
Internal Soft-Start Delay
VFB = 1.2V, VCOMP = Floating
4
msec
Tr
Drive Pin Rise Time
Cgs = 3000pf, VDR = 0 to 3V
25
ns
Tf
Drive Pin Fall Time
Cgs = 3000pf, VDR = 0 to 3V
25
ns
VSD
Shutdown threshold (Note 5)
Output = High
1.27
Output = Low
ISD
Shutdown Pin Current
1.4
V
V (max)
0.3
V
V (min)
0.65
VSD = 5V
−1
VSD = 0V
+1
µA
IFB
Feedback Pin Current
15
nA
TSD
Thermal Shutdown
165
°C
Tsh
Thermal Shutdown Hysteresis
10
°C
θJA
Thermal Resistance
200
°C/W
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MM Package
4
Note 2: The human body model is a 100 pF capacitor discharged through a 1.5kΩ resistor into each pin.
Note 3: All limits are guaranteed at room temperature (standard type face) and at temperature extremes (bold type face). All room temperature limits are 100%
tested. All limits at temperature extremes are guaranteed via correlation using standard Statistical Quality Control (SQC) methods. All limits are used to calculate
Average Outgoing Quality Level (AOQL).
Note 4: Typical numbers are at 25°C and represent the most likely norm.
Note 5: The FA/SD pin should be pulled to VIN through a resistor to turn the regulator off. The voltage on the FA/SD pin must be above the maximum limit for
Output = High to keep the regulator off and must be below the limit for Output = Low to keep the regulator on.
Note 6: The voltage on the drive pin, VDR is equal to the input voltage when input voltage is less than 7.2V. VDR is equal to 7.2V when the input voltage is greater
than or equal to 7.2V.
Note 7: The limits for the maximum duty cycle can not be specified since the part does not permit less than 100% maximum duty cycle operation.
Note 8: The over-voltage protection is specified with respect to the feedback voltage. This is because the over-voltage protection tracks the feedback voltage.
The overvoltage protection threshold is given by adding the feedback voltage, VFB to the over-voltage protection specification.
Note 9: For this test, the FA/SD pin is pulled to ground using a 40K resistor.
Note 10: For this test, the FA/SD pin is pulled to 5V using a 40K resistor.
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LM3478/LM3478Q
Note 1: Absolute Maximum Ratings are limits beyond which damage to the device may occur. Operating Ratings are conditions under which operation of the
device is intended to be functional. For guaranteed specifications and test conditions, see the Electrical Characteristics.
LM3478/LM3478Q
Typical Performance Characteristics
Unless otherwise specified, VIN = 12V, TJ = 25°C.
IQ vs Input Voltage (Shutdown)
ISupply vs Input Voltage (Non-Switching)
10135503
10135534
ISupply vs VIN (Switching)
Switching Frequency vs RFA
10135535
10135504
Drive Voltage vs Input Voltage
Frequency vs Temperature
10135554
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10135505
6
LM3478/LM3478Q
Current Sense Threshold vs Input Voltage
COMP Pin Voltage vs Load Current
10135545
10135562
Efficiency vs Load Current (3.3V In and 12V Out)
Efficiency vs Load Current (5V In and 12V Out)
10135559
10135558
Efficiency vs Load Current (9V In and 12V Out)
Efficiency vs Load Current (3.3V In and 5V Out)
10135560
10135553
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LM3478/LM3478Q
Error Amplifier Gain
Error Amplifier Phase
10135555
10135556
COMP Pin Source Current vs Temperature
Short Circuit Sense Voltage vs Input Voltage
10135557
10135536
Compensation Ramp vs Compensation Resistor
Shutdown Threshold Hysteresis vs Temperature
10135546
10135551
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LM3478/LM3478Q
Duty Cycle vs Current Sense Voltage
10135594
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LM3478/LM3478Q
Functional Block Diagram
10135506
the external MOSFET in on during the blanking interval is
more than what is delivered to the load. An over-voltage comparator inside the LM3478 prevents the output voltage from
rising under these conditions. The over-voltage comparator
senses the feedback (FB pin) voltage and resets the RS latch.
The latch remains in reset state until the output decays to the
nominal value.
Functional Description
The LM3478 uses a fixed frequency, Pulse Width Modulated
(PWM) current mode control architecture. The block diagram
above shows the basic functionality. In a typical application
circuit, the peak current through the external MOSFET is
sensed through an external sense resistor. The voltage
across this resistor is fed into the Isen pin. This voltage is fed
into the positive input of the PWM comparator. The output
voltage is also sensed through an external feedback resistor
divider network and fed into the error amplifier negative input
(feedback pin, FB). The output of the error amplifier (COMP
pin) is added to the slope compensation ramp and fed into the
negative input of the PWM comparator. At the start of any
switching cycle, the oscillator sets the RS latch using the
switch logic block. This forces a high signal on the DR pin
(gate of the external MOSFET) and the external MOSFET
turns on. When the voltage on the positive input of the PWM
comparator exceeds the negative input, the RS latch is reset
and the external MOSFET turns off.
The voltage sensed across the sense resistor generally contains spurious noise spikes, as shown in Figure 2. These
spikes can force the PWM comparator to reset the RS latch
prematurely. To prevent these spikes from resetting the latch,
a blank-out circuit inside the IC prevents the PWM comparator
from resetting the latch for a short duration after the latch is
set. This duration is about 325ns and is called the blanking
interval and is specified as minimum on-time in the Electrical
Characteristics section. Under extremely light-load or no-load
conditions, the energy delivered to the output capacitor when
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OVER VOLTAGE PROTECTION
The LM3478 has over voltage protection (OVP) for the output
voltage. OVP is sensed at the feedback pin (pin 3). If at anytime the voltage at the feedback pin rises to VFB+ VOVP, OVP
is triggered. See ELECTRICAL CHARACTERISTICS section
for limits on VFB and VOVP.
OVP will cause the drive pin to go low, forcing the power
MOSFET off. With the MOSFET off, the output voltage will
drop. The LM3478 will begin switching again when the feedback voltage reaches VFB + (VOVP - VOVP(HYS)). See
ELECTRICAL CHARACTERISTICS for limits on VOVP(HYS).
OVP can be triggered if the unregulated input voltage crosses
7.2V, the output voltage will react as shown in Figure 1. The
internal bias of the LM3478 comes from either the internal
LDO as shown in the block diagram or the voltage at the Vin
pin is used directly. At Vin voltages lower than 7.2V the internal IC bias is the Vin voltage and at voltages above 7.2V the
internal LDO of the LM3478 provides the bias. At the
switchover threshold at 7.2V a sudden small change in bias
voltage is seen by all the internal blocks of the LM3478. The
control voltage shifts because of the bias change, the PWM
comparator tries to keep regulation. To the PWM comparator,
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LM3478/LM3478Q
the scenario is identical to a step change in the load current,
so the response at the output voltage is the same as would
be observed in a step load change. Hence, the output voltage
overshoot here can also trigger OVP. The LM3478 will regulate in hysteretic mode for several cycles, or may not recover
and simply stay in hysteretic mode until the load current drops
or Vin is not crossing the 7.2V threshold anymore. Note that
the output is still regulated in hysteretic mode.
Depending on the requirements of the application there is
some influence one has over this effect. The threshold of 7.2V
can be shifted to higher voltages by adding a resistor in series
with Vin. In case Vin is right at the threshold of 7.2V it can
happen that the threshold is crossed over and over due to
some slight ripple on Vin. To minimize the effect on the output
voltage one can filter the Vin pin with an RC filter.
10135511
FIGURE 1. The Feedback Voltage Experiences an
Oscillation if the Input Voltage crosses the 7.2V Internal
Bias Threshold
10135507
FIGURE 2. Basic Operation of the PWM Comparator
SLOPE COMPENSATION RAMP
The LM3478 uses a current mode control scheme. The main
advantages of current mode control are inherent cycle-by-cycle current limit for the switch and simpler control loop characteristics. It is also easy to parallel power stages using
current mode control since current sharing is automatic. However, current mode control has an inherent instability for duty
cycles greater than 50%, as shown in Figure 3.
A small increase in the load current causes the switch current
to increase by ΔI0. The effect of this load change is ΔI1.
The two solid waveforms shown are the waveforms compared
at the internal pulse width modulator, used to generate the
MOSFET drive signal. The top waveform with the slope Se is
the internally generated control waveform VC. The bottom
waveform with slopes Sn and Sf is the sensed inductor current
waveform VSEN.
10135512
FIGURE 3. Sub-Harmonic Oscillation for D>0.5 and
Compensation Ramp to Avoid Sub-Harmonic Oscillation
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LM3478/LM3478Q
It is a good design practice to only add as much slope compensation as needed to avoid subharmonic oscillation. Additional slope compensation minimizes the influence of the
sensed current in the control loop. With very large slope compensation the control loop characteristics are similar to a
voltage mode regulator which compares the error voltage to
a saw tooth waveform rather than the inductor current.
Sub-harmonic Oscillation can be easily understood as a geometric problem. If the control signal does not have slope, the
slope representing the inductor current ramps up until the
control signal is reached and then slopes down again. If the
duty cycle is above 50%, any perturbation will not converge
but diverge from cycle to cycle and causes sub-harmonic oscillation.
It is apparent that the difference in the inductor current from
one cycle to the next is a function of Sn, Sf and Se as follows:
Hence, if the quantity (Sf - Se)/(Sn + Se) is greater than 1, the
inductor current diverges and subharmonic oscillation results.
This counts for all current mode topologies. The LM3478 has
some internal slope compensation VSL which is enough for
many applications above 50% duty cycle to avoid subharmonic oscillation .
For boost applications, the slopes Se, Sf and Sn can be calculated with the formulas below:
10135513
Se = VSL x fs
FIGURE 4. Adding External Slope Compensation
Sf = (VOUT - VIN)/L
Sn = VIN/L
When Se increases then the factor which determines if subharmonic oscillation will occur decreases. When the duty
cycle is greater than 50%, and the inductance becomes less,
the factor increases.
For more flexibility slope compensation can be increased by
adding one external resistor, RSL, in the Isens path. Figure 4
shows the setup. The externally generated slope compensation is then added to the internal slope compensation of the
LM3478. When using external slope compensation, the formula for Se becomes:
10135595
Se = (VSL + (K x RSL)) x fs
FIGURE 5. External Slope Compensation
ΔVSL vs RSL
A typical value for factor K is 40 µA.
The factor changes with switching frequency. Figure 5 is used
to determine the factor K for individual applications and the
formula below gives the factor K.
FREQUENCY ADJUST/SHUTDOWN
The switching frequency of the LM3478 can be adjusted between 100kHz and 1MHz using a single external resistor. This
resistor must be connected between FA/SD pin and ground,
as shown in Figure 6. To determine the value of the resistor
required for a desired switching frequency refer to the typical
performance characteristics.
K = ΔVSL / RSL
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10135514
FIGURE 6. Frequency Adjust
10135516
FIGURE 7. Shutdown Operation in Frequency Adjust Mode
SHORT-CIRCUIT PROTECTION
When the voltage across the sense resistor measured on the
Isen pin exceeds 343 mV, short circuit current limit protection
gets activated. A comparator inside the LM3478 reduces the
switching frequency by a factor of 5 and maintains this condition until the short is removed. In normal operation the
sensed current will trigger the power MOSFET to turn off.
During the blanking interval the PWM comparator will not react to an over current so that this additional 343 mV current
limit threshold is implemented to protect the device in a short
circuit or severe overload condition.
Typical Applications
The LM3478 may be operated in either the continuous (CCM)
or the discontinuous current conduction mode (DCM). The
following applications are designed for the CCM operation.
This mode of operation has higher efficiency and usually lower EMI characteristics than the DCM.
BOOST CONVERTER
The boost converter converts a low input voltage into a higher
output voltage. The basic configuration for a boost converter
is shown in Figure 8. In the CCM (when the inductor current
never reaches zero at steady state), the boost regulator operates in two states. In the first state of operation, MOSFET
Q is turned on and energy is stored in the inductor. During this
state, diode D is reverse biased and load current is supplied
by the output capacitor, Cout.
In the second state, MOSFET Q is off and the diode is forward
biased. The energy stored in the inductor is transferred to the
load and the output capacitor. The ratio of the switch on time
to the total period is the duty cycle D:
D = 1 - (Vin / Vout)
Including the voltage drop across the MOSFET and the diode
the definition for the duty cycle is:
D = 1 - ((Vin - Vq)/(Vout + Vd))
Vd is the forward voltage drop of the diode and Vq is the voltage drop across the MOSFET when it is on.
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LM3478/LM3478Q
The FA/SD pin also functions as a shutdown pin. If a high
signal (>1.35V) appears on the FA/SD pin, the LM3478 stops
switching and goes into a low current mode. The total supply
current of the IC reduces to less than 10 uA under these conditions. Figure 7 shows implementation of the shutdown function when operating in frequency adjust mode. In this mode a
high signal for more than 30us shuts down the IC. However,
the voltage on the FA/SD pin should be always less than the
absolute maximum of 7V to avoid any damage to the device.
LM3478/LM3478Q
10135522
FIGURE 8. Simplified Boost Convertert
A. First Cycle Operation B. Second Cycle of Operation
POWER INDUCTOR SELECTION
The inductor is one of the two energy storage elements in a
boost converter. Figure 9 shows how the inductor current
varies during a switching cycle. The current through an inductor is quantified by the following relationship of L, IL and
VL :
The important quantities in determining a proper inductance
value are IL (the average inductor current) and ΔIL (the inductor current ripple). If ΔIL is larger than IL, the inductor
current will drop to zero for a portion of the cycle and the converter will operate in the DCM. All the analysis in this
datasheet assumes operation in the CCM. To operate in the
CCM, the following condition must be met:
Choose the minimum Iout to determine the minimum inductance value. A common choice is to set ΔIL to 30% of IL.
Choosing an appropriate core size for the inductor involves
calculating the average and peak currents expected through
the inductor. In a boost converter the peak inductor current is:
ILPEAK = Average IL(max) + ΔIL(max)
Average IL(max) = Iout / (1-D)
ΔIL(max) = D x Vin / (2 x fs x L)
An inductor size with ratings higher than these values has to
be selected. If the inductor is not properly rated, saturation will
occur and may cause the circuit to malfunction.
The LM3478 can be set to switch at very high frequencies.
When the switching frequency is high, the converter can be
operated with very small inductor values. The LM3478 senses
the peak current through the switch which is the same as the
peak inductor current as calculated above.
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10135524
FIGURE 9. Inductor Current and Diode Current
14
LM3478/LM3478Q
The expression for RSEN is:
PROGRAMMING THE OUTPUT VOLTAGE AND OUTPUT
CURRENT
The output voltage can be programmed using a resistor divider between the output and the FB pin. The resistors are
selected such that the voltage at the FB pin is 1.26V. Pick
RF1 (the resistor between the output voltage and the feedback
pin) and RF2 (the resistor between the feedback pin and
ground) can be selected using the following equation,
The numerator of the above equation is the VCS and the denominator is the peak current.
RF2 = (1.26V x RF1) / (Vout - 1.26V)
A 100pF capacitor may be connected between the feedback
and ground pins to reduce noise.
CURRENT LIMIT WITH THE INTERNAL SLOPE
COMPENSATION PLUS ADDITIONAL EXTERNAL SLOPE
COMPENSATION
If an external slope compensation resistor RSL is used, the
internal command signal will be modified and this will have an
effect on the current limit. The command signal then includes
the external slope ΔVSL:
CURRENT LIMIT WITH ONLY THE INTERNAL SLOPE
COMPENSATION
The maximum amount of current that can be delivered at the
output is controlled by the sense resistor, R SEN. Current limit
occurs when the voltage that is generated across the sense
resistor equals the current sense threshold voltage, VSENSE.
Limits for VSENSE have been specified in the electrical characteristics section. This can be expressed as:
VCS = VSENSE – (D x (VSL + ΔVSL))
Where ΔVSL is the additional slope compensation generated
as discussed in the section slope compensation ramp. This
changes the equation for RSEN to:
VSENSE = ISW(PEAK) x RSEN
VSENSE represents the maximum value of the control signal
VCS as shown in Figure 10. This control signal, however, is
not a constant value and changes over the course of a period
as a result of the internal compensation ramp. Therefore the
current limit will also change as a result of the internal compensation ramp. The actual VCS can be better expressed as
a function of the sense voltage (VSENSE) and the internal compensation ramp:
Note that RSL which defines ΔVSL is an additional way of setting the current limit.
In some designs RSL can also help to filter noise to keep the
ISEN pin quiet.
VCS = VSENSE − (D x VSL)
POWER DIODE SELECTION
Observation of the boost converter circuit shows that the average current through the diode is the average load current,
and the peak current through the diode is the peak current
through the inductor. The diode should be rated to handle
more than its peak current. The peak diode current can be
calculated using the formula:
ID(Peak) = IOUT/ (1−D) + ΔIL
Thermally the diode must be able to handle the maximum average current delivered to the output. The peak reverse voltage for boost converters is equal to the regulated output
voltage. The diode must be capable of handling this voltage.
To improve efficiency, a low forward drop schottky diode is
recommended.
10135552
POWER MOSFET SELECTION
The drive pin of the LM3478 must be connected to the gate
of an external MOSFET. The drive pin (DR) voltage depends
on the input voltage (see typical performance characteristics).
In most applications, a logic level MOSFET can be used. For
very low input voltages, a sub logic level MOSFET should be
used. The selected MOSFET has a great influence on the
system efficiency. The critical parameters for selecting a
MOSFET are:
1. Minimum threshold voltage, VTH(MIN)
2. On-resistance, RDS(ON)
3. Total gate charge, Qg
4. Reverse transfer capacitance, CRSS
5. Maximum drain to source voltage, VDS(MAX)
FIGURE 10. Current Sense Voltage vs Duty Cycle
Figure 10 shows how VCS changes with duty cycle. The curve
shows the ramp Se which is defined by the voltage VSENSE (at
0% dutycycle) and by the internally generated slope VSL
which changes VCS with duty cycle. The dotted line shows
VSENSE. At 100% duty cycle, the current sense voltage will be
VSENSE minus VSL.
The graph also shows the increased current limit of 343 mV
(typical) during the 325 ns (typical) blank out time. For different frequencies this fixed blank out time obviously occupies
more duty cycle, percentage wise.
The peak current through the switch is equal to the peak inductor current.
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LM3478/LM3478Q
The off-state voltage of the MOSFET is approximately equal
to the output voltage. Vds(max) must be greater than the output voltage. The power losses in the MOSFET can be categorized into conduction losses and switching losses. Rds(on)
is needed to estimate the conduction losses, Pcond:
OUTPUT CAPACITOR SELECTION
The output capacitor in a boost converter provides all the output current when the inductor is charging. As a result it sees
very large ripple currents. The output capacitor should be capable of handling the maximum rms current. The rms current
in the output capacitor is:
Pcond = I2 x RDS(ON) x D x fS
The temperature effect on the RDS(ON) usually is quite significant. Assume 30% increase at hot.
For the current I in the formula above the average inductor
current may be used.
Especially at high switching frequencies the switching losses
may be the largest portion of the total losses.
The switching losses are very difficult to calculate due to
changing parasitics of a given MOSFET in operation. Often
the individual MOSFETS datasheet does not give enough information to yield a useful result. The following formulas give
a rough idea how the switching losses are calculated:
Where
The ESR and ESL of the capacitor directly control the output
ripple. Use capacitors with low ESR and ESL at the output for
high efficiency and low ripple voltage. Surface mount tantalums, surface mount polymer electrolytic, polymer tantalum,
or multi-layer ceramic capacitors are recommended at the
output.
For applications that require very low output voltage ripple, a
second stage LC filter often is a good solution. Most of the
time it is lower cost to use a small second Inductor in the
power path and an additional final output capacitor than to
reduce the output voltage ripple by purely increasing the output capacitor without an additional LC filter.
INPUT CAPACITOR SELECTION
Due to the presence of an inductor at the input of a boost
converter, the input current waveform is continuous and triangular as shown in figure 9. The inductor ensures that the
input capacitor sees fairly low ripple currents. However, as the
input capacitor gets smaller, the input ripple goes up. The rms
current in the input capacitor is given by:
LAYOUT GUIDELINES
Good board layout is critical for switching controllers. First the
ground plane area must be sufficient for thermal dissipation
purposes and second, appropriate guidelines must be followed to reduce the effects of switching noise. Switching
converters are very fast switching devices. In such devices,
the rapid increase of input current combined with the parasitic
trace inductance generates unwanted Ldi/dt noise spikes.
The magnitude of this noise tends to increase as the output
current increases. This parasitic spike noise may turn into
electromagnetic interference (EMI), and can also cause problems in device performance. Therefore, care must be taken
in layout to minimize the effect of this switching noise. The
current sensing circuit in current mode devices can be easily
affected by switching noise. This noise can cause duty cycle
jittering which leads to increased spectral noise. Although the
LM3478 has 325ns blanking time at the beginning of every
cycle to ignore this noise, some noise may remain after the
blanking time.
The most important layout rule is to keep the AC current loops
as small as possible. Figure 12 shows the current flow of a
boost converter. The top schematic shows a dotted line which
represents the current flow during on-state and the middle
schematic shows the current flow during off-state. The bottom
schematic shows the currents we refer to as AC currents.
They are the most critical ones since current is changing in
very short time periods. The dotted lined traces of the bottom
schematic are the once to make as short as possible.
The input capacitor should be capable of handling the rms
current. Although the input capacitor is not as critical in a
boost application, low values can cause impedance interactions. Therefore a good quality capacitor should be chosen in
the range of 100µF to 200µF. If a value lower than 100µF is
used, then problems with impedance interactions or switching
noise can affect the LM3478. To improve performance, especially with Vin below 8 volts, it is recommended to use a 20
Ohm resistor at the input to provide an RC filter. The resistor
is placed in series with the VIN pin with only a bypass capacitor attached to the VIN pin directly (see figure 11). A 0.1µF
or 1µF ceramic capacitor is necessary in this configuration.
The bulk input capacitor and inductor will connect on the other
side of the resistor at the input power supply.
10135593
FIGURE 11. Reducing IC Input Noise
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16
Since the LM3478 controls a low-side N-Channel MOSFET,
it can also be used in SEPIC (Single Ended Primary Inductance Converter) applications. An example of a SEPIC using
the LM3478 is shown in Figure 13. Note that the output voltage can be higher or lower than the input voltage. The SEPIC
uses two inductors to step-up or step-down the input voltage.
The inductors L1 and L2 can be two discrete inductors or two
windings of a coupled inductor since equal voltages are applied across the inductor throughout the switching cycle. Using two discrete inductors allows use of catalog magnetics, as
opposed to a custom inductor. The input ripple can be reduced along with size by using the coupled windings for L1
and L2.
Due to the presence of the inductor L1 at the input, the SEPIC
inherits all the benefits of a boost converter. One main advantage of a SEPIC over a boost converter is the inherent
input to output isolation. The capacitor CS isolates the input
from the output and provides protection against a shorted or
malfunctioning load. Hence, the SEPIC is useful for replacing
boost circuits when true shutdown is required. This means
that the output voltage falls to 0V when the switch is turned
off. In a boost converter, the output can only fall to the input
voltage minus a diode drop.
The duty cycle of a SEPIC is given by:
10135520
FIGURE 12. Current Flow In A Boost Application
The PGND and AGND pins have to be connected to the same
ground very close to the IC. To avoid ground loop currents,
attach all the grounds of the system only at one point.
A ceramic input capacitor should be connected as close as
possible to the Vin pin and grounded close to the GND pin.
For a layout example please see Application Note1204. For
more information about layout in switch mode power supplies
please refer to Application Note 1229.
In the above equation, VQ is the on-state voltage of the MOSFET, Q, and VDIODE is the forward voltage drop of the diode.
COMPENSATION
For detailed explanation on how to select the right compensation components to attach to the compensation pin for a
boost topology please see Application Note 1286.
10135544
FIGURE 13. Typical SEPIC Converter
to source voltage, VDS(MAX). The peak switch voltage in a
SEPIC is given by:
POWER MOSFET SELECTION
As in a boost converter, parameters governing the selection
of the MOSFET are the minimum threshold voltage, VTH
(MIN), the on-resistance, RDS(ON), the total gate charge, Qg, the
reverse transfer capacitance, CRSS, and the maximum drain
VSW(PEAK) = VIN + VOUT + VDIODE
The selected MOSFET should satisfy the condition:
VDS(MAX) > VSW(PEAK)
17
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LM3478/LM3478Q
Designing SEPIC Using the LM3478
LM3478/LM3478Q
The peak switch current is given by:
IL1PK must be lower than the maximum current rating set by
the current sense resistor.
The value of L1 can be increased above the minimum recommended to reduce input ripple and output ripple. However,
once DIL1 is less than 20% of IL1AVE, the benefit to output ripple
is minimal.
By increasing the value of L2 above the minimum recommended, ΔIL2 can be reduced, which in turn will reduce the
output ripple voltage:
The rms current through the switch is given by:
POWER DIODE SELECTION
The Power diode must be selected to handle the peak current
and the peak reverse voltage. In a SEPIC, the diode peak
current is the same as the switch peak current. The off-state
voltage or peak reverse voltage of the diode is VIN + VOUT.
Similar to the boost converter, the average diode current is
equal to the output current. Schottky diodes are recommended.
where ESR is the effective series resistance of the output capacitor.
If L1 and L2 are wound on the same core, then L1 = L2 = L.
All the equations above will hold true if the inductance is replaced by 2L.
SENSE RESISTOR SELECTION
The peak current through the switch, ISW(PEAK) can be adjusted using the current sense resistor, RSEN, to provide a certain
output current. Resistor RSEN can be selected using the formula:
SELECTION OF INDUCTORS L1 AND L2
Proper selection of inductors L1 and L2 to maintain continuous current conduction mode requires calculations of the
following parameters.
Average current in the inductors:
Sepic Capacitor Selection
IL2AVE = IOUT
The selection of the SEPIC capacitor, CS, depends on the
rms current. The rms current of the SEPIC capacitor is given
by:
Peak to peak ripple current, to calculate core loss if necessary:
The SEPIC capacitor must be rated for a large ACrms current
relative to the output power. This property makes the SEPIC
much better suited to lower power applications where the rms
current through the capacitor is relatively small (relative to
capacitor technology). The voltage rating of the SEPIC capacitor must be greater than the maximum input voltage.
There is an energy balance between CS and L1, which can
be used to determine the value of the capacitor. The basic
energy balance equation is:
Maintaining the condition IL > ΔiL/2 to ensure continuous current conduction yields:
where
Peak current in the inductor, to ensure the inductor does not
saturate:
is the ripple voltage across the SEPIC capacitor, and
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18
is the ripple current through the inductor L1. The energy balance equation can be solved to provide a minimum value for
CS:
Output Capacitor Selection
Input Capacitor Selection
The output capacitor of the SEPIC sees very large ripple currents (similar to the output capacitor of a boost converter). The
rms current through the output capacitor is given by:
Similar to a boost converter, the SEPIC has an inductor at the
input. Hence, the input current waveform is continuous and
triangular. The inductor ensures that the input capacitor sees
fairly low ripple currents. However, as the input capacitor gets
smaller, the input ripple goes up. The rms current in the input
capacitor is given by:
The ESR and ESL of the output capacitor directly control the
output ripple. Use low capacitors with low ESR and ESL at
the output for high efficiency and low ripple voltage. Surface
mount tantalums, surface mount polymer electrolytic and
polymer tantalum, Sanyo- OSCON, or multi-layer ceramic capacitors are recommended at the output for low ripple.
The input capacitor should be capable of handling the rms
current. Although the input capacitor is not as critical in a
Other Application Circuit
10135543
FIGURE 14. Typical Flyback Circuit
19
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LM3478/LM3478Q
boost application, low values can cause impedance interactions. Therefore a good quality capacitor should be chosen in
the range of 100µF to 200µF. If a value lower than 100µF is
used than problems with impedance interactions or switching
noise can affect the LM3478. To improve performance, especially with VIN below 8 volts, it is recommended to use a
20Ω resistor at the input to provide a RC filter. The resistor is
placed in series with the VIN pin with only a bypass capacitor
attached to the VIN pin directly (see Figure 11). A 0.1µF or 1µF
ceramic capacitor is necessary in this configuration. The bulk
input capacitor and inductor will connect on the other side of
the resistor with the input power supply.
LM3478/LM3478Q
Physical Dimensions inches (millimeters) unless otherwise noted
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20
LM3478/LM3478Q
Notes
21
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LM3478/LM3478Q High Efficiency Low-Side N-Channel Controller for Switching Regulator
Notes
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