LTC3639 High Efficiency, 150V 100mA Synchronous Step-Down Regulator Features n n n n n n n n n n n n n n Wide Operating Input Voltage Range: 4V to 150V Synchronous Operation for Highest Efficiency Internal High Side and Low Side Power MOSFETs No Compensation Required Adjustable 10mA to 100mA Maximum Output Current Low Dropout Operation: 100% Duty Cycle Low Quiescent Current: 12µA Wide Output Range: 0.8V to VIN 0.8V ±1% Feedback Voltage Reference Precise RUN Pin Threshold Internal or External Soft-Start Programmable 1.8V, 3.3V, 5V or Adjustable Output Few External Components Required Programmable Input Overvoltage Lockout Thermally Enhanced High Voltage MSOP Package Applications Industrial Control Supplies Medical Devices n Distributed Power Systems n Portable Instruments n Battery-Operated Devices nAutomotive nAvionics n n The LTC®3639 is a high efficiency step-down DC/DC regulator with internal high side and synchronous power switches that draws only 12μA typical DC supply current while maintaining a regulated output voltage at no load. The LTC3639 can supply up to 100mA load current and features a programmable peak current limit that provides a simple method for optimizing efficiency and for reducing output ripple and component size. The LTC3639’s combination of Burst Mode® operation, integrated power switches, low quiescent current, and programmable peak current limit provides high efficiency over a broad range of load currents. With its wide input range of 4V to 150V and programmable overvoltage lockout, the LTC3639 is a robust regulator suited for regulating from a wide variety of power sources. Additionally, the LTC3639 includes a precise run threshold and soft-start feature to guarantee that the power system start-up is well-controlled in any environment. A feedback comparator output enables multiple LTC3639s to be connected in parallel for higher current applications. The LTC3639 is available in a thermally enhanced high voltage-capable 16-lead MSE package with four missing pins. L, LT, LTC, LTM, Burst Mode, Linear Technology and the Linear logo are registered trademarks of Linear Technology Corporation. All other trademarks are the property of their respective owners. Typical Application Efficiency and Power Loss vs Load Current 100 90 5V to 150V Input to 5V Output, 100mA Step-Down Regulator 470µH 1µF 200V VIN SW LTC3639 RUN VFB OVLO VPRG2 SS VPRG1 GND 3639 TA01a VOUT 5V 10µF 100mA EFFICIENCY 70 50 40 30 1000 VIN = 12V VIN = 36V VIN = 72V VIN = 150V 60 100 POWER LOSS 10 POWER LOSS (mW) VIN 5V TO 150V VOUT = 5V 80 EFFICIENCY (%) n Description 20 10 0 0.1 1 1 10 LOAD CURRENT (mA) 100 3639 TA01b 3639f For more information www.linear.com/LTC3639 1 LTC3639 Absolute Maximum Ratings Pin Configuration (Note 1) VIN Supply Voltage.................................... –0.3V to 150V RUN Voltage............................................. –0.3V to 150V SS, FBO, OVLO, ISET Voltages....................... –0.3V to 6V VFB, VPRG1, VPRG2 Voltages.......................... –0.3V to 6V Operating Junction Temperature Range (Notes 2, 3) LTC3639E, LTC3639I.......................... –40°C to 125°C LTC3639H........................................... –40°C to 150°C LTC3639MP........................................ –55°C to 150°C Storage Temperature Range................... –65°C to 150°C Lead Temperature (Soldering, 10 sec).................... 300°C TOP VIEW SW 1 16 GND VIN 3 FBO VPRG2 VPRG1 GND 17 GND 5 6 7 8 14 RUN 12 11 10 9 OVLO ISET SS VFB MSE PACKAGE VARIATION: MSE16 (12) 16-LEAD PLASTIC MSOP TJMAX = 150°C, θJA = 40°C/W, θJC = 10°C/W EXPOSED PAD (PIN 17) IS GND, MUST BE SOLDERED TO PCB Order Information LEAD FREE FINISH TAPE AND REEL PART MARKING* PACKAGE DESCRIPTION TEMPERATURE RANGE LTC3639EMSE#PBF LTC3639EMSE#TRPBF 3639 16-Lead Plastic MSOP –40°C to 125°C LTC3639IMSE#PBF LTC3639IMSE#TRPBF 3639 16-Lead Plastic MSOP –40°C to 125°C LTC3639HMSE#PBF LTC3639HMSE#TRPBF 3639 16-Lead Plastic MSOP –40°C to 150°C LTC3639MPMSE#PBF LTC3639MPMSE#TRPBF 3639 16-Lead Plastic MSOP –55°C to 150°C Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container. Consult LTC Marketing for information on non-standard lead based finish parts. For more information on lead free part marking, go to: http://www.linear.com/leadfree/ For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/ Electrical Characteristics The l denotes the specifications which apply over the specified operating junction temperature range, otherwise specifications are at TA = 25°C (Note 2). VIN = 12V, unless otherwise noted. SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS 150 V Input Supply (VIN) VIN Input Voltage Operating Range VOUT Output Voltage Operating Range UVLO VIN Undervoltage Lockout IQ DC Supply Current (Note 4) Active Mode Sleep Mode Shutdown Mode 4 0.8 VIN Rising VIN Falling Hysteresis l l 3.5 3.3 No Load VRUN = 0V 1.17 1.06 VIN V 3.75 3.5 250 4.0 3.8 V V mV 150 12 1.4 350 22 6 µA µA µA 1.21 1.10 110 1.25 1.14 V V mV VRUN RUN Pin Threshold RUN Rising RUN Falling Hysteresis IRUN RUN Pin Leakage Current RUN = 1.3V –10 0 10 nA VOVLO OVLO Pin Threshold OVLO Rising OVLO Falling Hysteresis 1.17 1.06 1.21 1.10 110 1.25 1.14 V V mV 3639f 2 For more information www.linear.com/LTC3639 LTC3639 Electrical Characteristics The l denotes the specifications which apply over the specified operating junction temperature range, otherwise specifications are at TA = 25°C (Note 2). VIN = 12V, unless otherwise noted. SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS V V Output Supply (VFB) Feedback Comparator Threshold (Adjustable Output) VFB Rising, VPRG1 = VPRG2 = 0V LTC3639E, LTC3639I LTC3639H, LTC3639MP l l 0.792 0.788 0.800 0.800 0.808 0.812 VFBH Feedback Comparator Hysteresis (Adjustable Output) VFB Falling, VPRG1 = VPRG2 = 0V l 3 5 9 mV IFB Feedback Pin Current VFB = 1V, VPRG1 = VPRG2 = 0V VFB(FIXED) Feedback Comparator Thresholds (Fixed Output) VFB(ADJ) –10 0 10 nA VFB Rising, VPRG1 = SS, VPRG2 = 0V VFB Falling, VPRG1 = SS, VPRG2 = 0V l l 4.94 4.91 5.015 4.985 5.09 5.06 V V VFB Rising, VPRG1 = 0V, VPRG2 = SS VFB Falling, VPRG1 = 0V, VPRG2 = SS l l 3.26 3.24 3.31 3.29 3.36 3.34 V V VFB Rising, VPRG1 = VPRG2 = SS VFB Falling, VPRG1 = VPRG2 = SS l l 1.78 1.77 1.81 1.80 1.84 1.83 V V ISET Floating 100k Resistor from ISET to GND ISET Shorted to GND l l l 200 100 17 230 120 25 260 140 30 mA mA mA Operation IPEAK Peak Current Comparator Threshold RON Power Switch On-Resistance Top Switch Bottom Switch ISW = –50mA ISW = 50mA 4.2 2.2 ILSW Switch Pin Leakage Current VIN = 150V, SW = 0V ISS Soft-Start Pin Pull-Up Current VSS < 2.5V tINT(SS) Internal Soft-Start Time SS Pin Floating Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Note 2: The LTC3639 is tested under pulsed load conditions such that TJ ≈ TA. The LTC3639E is guaranteed to meet performance specifications from 0°C to 85°C. Specifications over the –40°C to 125°C operating junction temperature range are assured by design, characterization and correlation with statistical process controls. The LTC3639I is guaranteed over the –40°C to 125°C operating junction temperature range, the LTC3639H is guaranteed over the –40°C to 150°C operating junction temperature range and the LTC3639MP is tested and guaranteed over the –55°C to 150°C operating junction temperature range. High junction temperatures degrade operating lifetimes; operating lifetime is derated for junction temperatures greater than 125°C. Note that the 4 Ω Ω 0.1 1 μA 5 6 μA 1 ms maximum ambient temperature consistent with these specifications is determined by specific operating conditions in conjunction with board layout, the rated package thermal impedance and other environmental factors. Note 3: The junction temperature (TJ, in °C) is calculated from the ambient temperature (TA, in °C) and power dissipation (PD, in Watts) according to the formula: TJ = TA + (PD • θJA) where θJA is 40°C/W for the MSOP package. Note that the maximum ambient temperature consistent with these specifications is determined by specific operating conditions in conjunction with board layout, the rated package thermal impedance and other environmental factors. Note 4: Dynamic supply current is higher due to the gate charge being delivered at the switching frequency. See Applications Information. 3639f For more information www.linear.com/LTC3639 3 LTC3639 Typical Performance Characteristics Efficiency vs Load Current, VOUT = 5V Efficiency vs Load Current, VOUT = 3.3V 90 80 80 80 70 70 70 60 50 40 ISET OPEN FIGURE 14 CIRCUIT VIN = 12V VIN = 36V VIN = 72V VIN = 150V 30 10 0 0.1 1 10 LOAD CURRENT (mA) 60 50 40 ISET OPEN FIGURE 14 CIRCUIT VIN = 12V VIN = 36V VIN = 72V VIN = 150V 30 20 10 0 0.1 100 1 10 LOAD CURRENT (mA) 3639 G01 60 50 40 30 0 75 50 100 VIN VOLTAGE (V) 25 125 800 799 798 –55 150 95 5 35 65 TEMPERATURE (°C) 125 155 PEAK CURRENT (mA) 50 1.16 1.14 1.12 75 100 125 150 175 200 RISET (kΩ) 3639 G07 FALLING 1.10 1.08 250 200 150 RISET = 100kΩ 100 0 –55 65 35 95 5 TEMPERATURE (°C) 125 155 3639 G06 Peak Current Trip Threshold vs Input Voltage ISET OPEN 50 25 1.20 1.18 300 250 200 50 RISING 1.22 1.06 –55 –25 300 250 0 1.24 Peak Current Trip Threshold vs Temperature 100 100 3639 G03 3639 G05 Peak Current Trip Threshold vs RISET 0 –25 3639 G04 150 1 10 LOAD CURRENT (mA) RUN and OVLO Comparator Threshold vs Temperature 801 ILOAD = 100mA ILOAD = 10mA ILOAD = 1mA 0 0 0.1 100 RUN OR OVLO THRESHOLD VOLTAGE (V) EFFICIENCY (%) 70 ISET OPEN FIGURE 14 CIRCUIT VIN = 12V VIN = 36V VIN = 72V VIN = 150V 30 10 802 THRESHOLD VOLTAGE (V) 80 10 40 Feedback Comparator Trip Threshold vs Temperature ISET OPEN FIGURE 14 CIRCUIT 20 50 3639 G02 Efficiency vs Input Voltage, VOUT = 5V 90 60 20 PEAK CURRENT (mA) 100 EFFICIENCY (%) 100 90 EFFICIENCY (%) 100 90 EFFICIENCY (%) 100 20 PEAK CURRENT TRIP THRESHOLD (mA) Efficiency vs Load Current, VOUT = 1.8V 65 95 5 35 TEMPERATURE (°C) 200 150 RISET = 100kΩ 100 50 ISET = GND –25 ISET OPEN 125 155 3639 G08 0 ISET = GND 0 30 60 90 VIN VOLTAGE (V) 120 150 3639 G09 3639f 4 For more information www.linear.com/LTC3639 LTC3639 Typical Performance Characteristics Quiescent Supply Current vs Temperature Quiescent Supply Current vs Input Voltage 30 5 30 60 90 VIN VOLTAGE (V) 20 15 SLEEP 10 5 SHUTDOWN 0 7 –25 65 35 5 95 TEMPERATURE (°C) VIN = 150V 6 5 4 3 2 SW = 150V 1 0 SW = 0V –1 SHUTDOWN 0 –55 150 120 SWITCH LEAKAGE CURRENT (µA) 10 0 8 VIN = 150V 25 SLEEP VIN SUPPLY CURRENT (µA) VIN SUPPLY CURRENT (µA) 15 Switch Leakage Current vs Temperature 125 155 –2 –55 –25 95 65 35 TEMPERATURE (°C) 5 3639 G10 125 155 3639 G12 3639 G11 Switch On-Resistance vs Temperature 7 8 6 7 SWITCH ON-RESISTANCE (Ω) SWITCH ON-RESISTANCE (Ω) Switch On-Resistance vs Input Voltage 5 TOP 4 3 BOTTOM 2 1 Load Step Transient Response OUTPUT VOLTAGE 50mV/DIV 6 TOP 5 LOAD CURRENT 50mA/DIV 4 BOTTOM VIN = 48V 200µs/DIV VOUT = 3.3V 1mA TO 100mA LOAD STEP FIGURE 15 CIRCUIT 3 2 0 30 60 90 VIN VOLTAGE (V) 150 120 1 –55 –25 65 95 35 TEMPERATURE (°C) 3639 G13 Operating Waveforms, VIN = 48V OUTPUT VOLTAGE 50mV/DIV SWITCH VOLTAGE 20V/DIV SWITCH VOLTAGE 50V/DIV INDUCTOR CURRENT 200mA/DIV INDUCTOR CURRENT 200mA/DIV 3639 G16 125 155 3639 G14 Operating Waveforms, VIN = 150V OUTPUT VOLTAGE 50mV/DIV VIN = 48V 10µs/DIV VOUT = 3.3V IOUT = 100mA FIGURE 15 CIRCUIT 5 3639 G15 Short-Circuit and Recovery OUTPUT VOLTAGE 1V/DIV INDUCTOR CURRENT 100mA/DIV VIN = 150V 10µs/DIV VOUT = 3.3V IOUT = 50mA FIGURE 15 CIRCUIT 3639 G17 500µs/DIV FIGURE 15 CIRCUIT 3639 G18 3639f For more information www.linear.com/LTC3639 5 LTC3639 Pin Functions SW (Pin 1): Switch Node Connection to Inductor. This pin connects to the drains of the internal power MOSFET switches. VIN (Pin 3): Main Supply Pin. A ceramic bypass capacitor should be tied between this pin and GND. FBO (Pin 5): Feedback Comparator Output. Connect to the VFB pins of additional LTC3639s to combine the output current. The typical pull-up current is 20µA. The typical pulldown impedance is 70Ω. See Applications Information. VPRG2, VPRG1 (Pins 6, 7): Output Voltage Selection. Short both pins to ground for a resistive divider programmable output voltage. Short VPRG1 to SS and short VPRG2 to ground for a 5V output voltage. Short VPRG1 to ground and short VPRG2 to SS for a 3.3V output voltage. Short both pins to SS for a 1.8V output voltage. GND (Pin 8, 16, Exposed Pad Pin 17): Ground. The exposed pad must be soldered to the PCB ground plane for rated thermal performance. VFB (Pin 9): Output Voltage Feedback. When configured for an adjustable output voltage, connect to an external resistive divider to divide the output voltage down for comparison to the 0.8V reference. For the fixed output configuration, directly connect this pin to the output. ISET (Pin 11): Peak Current Set Input. A resistor from this pin to ground sets the peak current comparator threshold. Leave floating for the maximum peak current (230mA typical) or short to ground for minimum peak current (25mA typical). The maximum output current is one-half the peak current. The 5µA current that is sourced out of this pin when switching is reduced to 1µA in sleep. Optionally, a capacitor can be placed from this pin to GND to trade off efficiency for light load output voltage ripple. See Applications Information. OVLO (Pin 12): Overvoltage Lockout Input. Connect to the input supply through a resistor divider to set the overvoltage lockout level. A voltage on this pin above 1.21V disables the internal MOSFET switches. Normal operation resumes when the voltage on this pin decreases below 1.10V. Exceeding the OVLO lockout threshold triggers a soft-start reset, resulting in a graceful recovery from an input supply transient. RUN (Pin 14): Run Control Input. A voltage on this pin above 1.21V enables normal operation. Forcing this pin below 0.7V shuts down the LTC3639, reducing quiescent current to approximately 1.4µA. Optionally, connect to the input supply through a resistor divider to set the undervoltage lockout. SS (Pin 10): Soft-Start Control Input. A capacitor to ground at this pin sets the output voltage ramp time. A 50µA current initially charges the soft-start capacitor until switching begins, at which time the current is reduced to its nominal value of 5µA. The output voltage ramp time from zero to its regulated value is 1ms for every 6.25nF of capacitance from SS to GND. If left floating, the ramp time defaults to an internal 1ms soft-start. 3639f 6 For more information www.linear.com/LTC3639 LTC3639 Block Diagram 1.3V 11 ACTIVE: 5µA SLEEP: 1µA ISET VIN + 3 CIN PEAK CURRENT COMPARATOR + – 14 12 RUN + 1.21V – OVLO LOGIC AND SHOOTTHROUGH PREVENTION – SW VOUT COUT GND 1.21V L1 1 + 16 + – 5V REVERSE CURRENT COMPARATOR 20µA 5 FEEDBACK COMPARATOR FBO + + – 70Ω 8 17 VOLTAGE REFERENCE START-UP: 50µA NORMAL: 5µA 0.800V R1 R2 GND GND 5V VPRG2 VPRG1 GND GND SS SS GND SS GND SS VOUT ADJUSTABLE 5V FIXED 3.3V FIXED 1.8V FIXED R1 VFB VPRG1 VPRG2 R2 1.0M ∞ 4.2M 800k 2.5M 800k 1.0M 800k SS 10 9 7 6 IMPLEMENT DIVIDER EXTERNALLY FOR ADJUSTABLE VERSION 3639 BD 3639f For more information www.linear.com/LTC3639 7 LTC3639 Operation (Refer to Block Diagram) The LTC3639 is a synchronous step-down DC/DC regulator with internal power switches that uses Burst Mode control, combining low quiescent current with high switching frequency, which results in high efficiency across a wide range of load currents. Burst Mode operation functions by using short “burst” cycles to switch the inductor current through the internal power MOSFETs, followed by a sleep cycle where the power switches are off and the load current is supplied by the output capacitor. During the sleep cycle, the LTC3639 draws only 12µA of supply current. At light loads, the burst cycles are a small percentage of the total cycle time which minimizes the average supply current, greatly improving efficiency. Figure 1 shows an example of Burst Mode operation. The switching frequency is dependent on the inductor value, peak current, input voltage and output voltage. SLEEP CYCLE BURST CYCLE SWITCHING FREQUENCY INDUCTOR CURRENT BURST FREQUENCY OUTPUT VOLTAGE ∆VOUT 3639 F01 Figure 1. Burst Mode Operation Main Control Loop The LTC3639 uses the VPRG1 and VPRG2 control pins to connect internal feedback resistors to the VFB pin. This enables fixed outputs of 1.8V, 3.3V or 5V without increasing component count, input supply current or exposure to noise on the sensitive input to the feedback comparator. External feedback resistors (adjustable mode) can be used by connecting both VPRG1 and VPRG2 to ground. In adjustable mode the feedback comparator monitors the voltage on the VFB pin and compares it to an internal 800mV reference. If this voltage is greater than the reference, the comparator activates a sleep mode in which the power switches and current comparators are disabled, reducing the VIN pin supply current to only 12µA. As the load current discharges the output capacitor, the voltage on the VFB pin decreases. When this voltage falls 5mV below the 800mV reference, the feedback comparator trips and enables burst cycles. At the beginning of the burst cycle, the internal high side power switch (P-channel MOSFET) is turned on and the inductor current begins to ramp up. The inductor current increases until either the current exceeds the peak current comparator threshold or the voltage on the VFB pin exceeds 800mV, at which time the high side power switch is turned off and the low side power switch (N-channel MOSFET) turns on. The inductor current ramps down until the reverse current comparator trips, signaling that the current is close to zero. If the voltage on the VFB pin is still less than the 800mV reference, the high side power switch is turned on again and another cycle commences. The average current during a burst cycle will normally be greater than the average load current. For this architecture, the maximum average output current is equal to half of the peak current. The hysteretic nature of this control architecture results in a switching frequency that is a function of the input voltage, output voltage, and inductor value. This behavior provides inherent short-circuit protection. If the output is shorted to ground, the inductor current will decay very slowly during a single switching cycle. Since the high side switch turns on only when the inductor current is near zero, the LTC3639 inherently switches at a lower frequency during start-up or short-circuit conditions. 3639f 8 For more information www.linear.com/LTC3639 LTC3639 Operation (Refer to Block Diagram) Start-Up and Shutdown If the voltage on the RUN pin is less than 0.7V, the LTC3639 enters a shutdown mode in which all internal circuitry is disabled, reducing the DC supply current to 1.4µA. When the voltage on the RUN pin exceeds 1.21V, normal operation of the main control loop is enabled. The RUN pin comparator has 110mV of internal hysteresis, and therefore must fall below 1.1V to disable the main control loop. An internal 1ms soft-start function limits the ramp rate of the output voltage on start-up to prevent excessive input supply droop. If a longer ramp time and consequently less supply droop is desired, a capacitor can be placed from the SS pin to ground. The 5µA current that is sourced out of this pin will create a smooth voltage ramp on the capacitor. If this ramp rate is slower than the internal 1ms soft-start, then the output voltage will be limited by the ramp rate on the SS pin instead. The internal and external soft-start functions are reset on start-up and after an undervoltage or overvoltage event on the input supply. Peak Inductor Current Programming The peak current comparator nominally limits the peak inductor current to 230mA. This peak inductor current can be adjusted by placing a resistor from the ISET pin to ground. The 5µA current sourced out of this pin through the resistor generates a voltage that adjusts the peak current comparator threshold. During sleep mode, the current sourced out of the ISET pin is reduced to 1µA. The ISET current is increased back to 5µA on the first switching cycle after exiting sleep mode. The ISET current reduction in sleep mode, along with adding a filtering capacitor, CISET, from the ISET pin to ground, provides a method of reducing light load output voltage ripple at the expense of lower efficiency and slightly degraded load step transient response. For applications requiring higher output current, the LTC3639 provides a feedback comparator output pin (FBO) for combining the output current of multiple LTC3639s. By connecting the FBO pin of a master LTC3639 to the VFB pin of one or more slave LTC3639s, the output currents can be combined to source 100mA times the number of LTC3639s. Dropout Operation When the input supply decreases toward the output supply, the duty cycle increases to maintain regulation. The P-channel MOSFET top switch in the LTC3639 allows the duty cycle to increase all the way to 100%. At 100% duty cycle, the P-channel MOSFET stays on continuously, providing output current equal to the peak current, which is twice the maximum load current when not in dropout. Input Undervoltage and Overvoltage Lockout The LTC3639 additionally implements protection features which inhibit switching when the input voltage is not within a programmable operating range. By use of a resistive divider from the input supply to ground, the RUN and OVLO pins serve as a precise input supply voltage monitor. Switching is disabled when either the RUN pin falls below 1.1V or the OVLO pin rises above 1.21V, which can be configured to limit switching to a specific range of input supply voltage. Furthermore, if the input voltage falls below 3.5V typical (3.8V maximum), an internal undervoltage detector disables switching. When switching is disabled, the LTC3639 can safely sustain input voltages up to the absolute maximum rating of 150V. Input supply undervoltage or overvoltage events trigger a soft-start reset, which results in a graceful recovery from an input supply transient. 3639f For more information www.linear.com/LTC3639 9 LTC3639 Applications Information The basic LTC3639 application circuit is shown on the front page of this data sheet. External component selection is determined by the maximum load current requirement and begins with the selection of the peak current programming resistor, RISET. The inductor value L can then be determined, followed by capacitors CIN and COUT. Peak Current Resistor Selection The peak current comparator has a maximum current limit of at least 200mA, which guarantees a maximum average current of 100mA. For applications that demand less current, the peak current threshold can be reduced to as little as 20mA. This lower peak current allows the efficiency and component selection to be optimized for lower current applications. The peak current threshold is linearly proportional to the voltage on the ISET pin, with 100mV and 1V corresponding to 20mA and 200mA peak current respectively. This pin may be driven by an external voltage source to modulate the peak current, which may be beneficial in some applications. Usually, the peak current is programmed with an appropriately chosen resistor (RISET) between the ISET pin and ground. The voltage generated on the ISET pin by RISET and the internal 5µA current source sets the peak current. The value of resistor for a particular peak current can be computed by using Figure 2 or the following equation: The internal 5μA current source is reduced to 1μA in sleep mode to maximize efficiency and to facilitate a tradeoff between efficiency and light load output voltage ripple, as described in the Optimizing Output Voltage Ripple section. The peak current is internally limited to be within the range of 20mA to 200mA. Shorting the ISET pin to ground programs the current limit to 20mA, and leaving it floating sets the current limit to the maximum value of 200mA. When selecting this resistor value, be aware that the maximum average output current for this architecture is limited to half of the peak current. Therefore, be sure to select a value that sets the peak current with enough margin to provide adequate load current under all conditions. Selecting the peak current to be 2.2 times greater than the maximum load current is a good starting point for most applications. Inductor Selection The inductor, input voltage, output voltage, and peak current determine the switching frequency during a burst cycle of the LTC3639. For a given input voltage, output voltage, and peak current, the inductor value sets the switching frequency during a burst cycle when the output is in regulation. Generally, switching at a frequency between 50kHz and 200kHz yields high efficiency, and 100kHz is a good first choice for many applications. The inductor value can be determined by the following equation: RISET = IPEAK • 106 where 20mA < IPEAK < 200mA. The variation in switching frequency during a burst cycle with input voltage and inductance is shown in Figure 3. For lower values of IPEAK, multiply the frequency in Figure 3 by 230mA/IPEAK. 250 CURRENT (mA) 200 TYPICAL PEAK INDUCTOR CURRENT 150 An additional constraint on the inductor value is the LTC3639’s 150ns minimum on-time of the high side switch. Therefore, in order to keep the current in the inductor well-controlled, 100 MAXIMUM LOAD CURRENT 50 0 V V L = OUT • 1– OUT VIN f •IPEAK 0 25 50 75 100 125 150 175 200 RISET (kΩ) 3639 F02 Figure 2. RISET Selection 3639f 10 For more information www.linear.com/LTC3639 LTC3639 Applications Information 10000 ISET OPEN 120 100 L = 100µH 80 L = 220µH INDUCTOR VALUE (µH) SWITCHING FREQUENCY (kHz) 140 60 40 L = 330µH 1000 20 0 0 30 60 90 120 VIN INPUT VOLTAGE (V) 150 3639 F03 Figure 3. Switching Frequency for VOUT = 3.3V the inductor value must be chosen so that it is larger than a minimum value which can be computed as follows: L> VIN(MAX) • tON(MIN) IPEAK 100 10 •1.2 where VIN(MAX) is the maximum input supply voltage when switching is enabled, tON(MIN) is 150ns, IPEAK is the peak current, and the factor of 1.2 accounts for typical inductor tolerance and variation over temperature. For applications that have large input supply transients, the OVLO pin can be used to disable switching above the maximum operating voltage VIN(MAX) so that the minimum inductor value is not artificially limited by a transient condition. Inductor values that violate the above equation will cause the peak current to overshoot and permanent damage to the part may occur. Although the previous equation provides the minimum inductor value, higher efficiency is generally achieved with a larger inductor value, which produces a lower switching frequency. For a given inductor type, however, as inductance is increased DC resistance (DCR) also increases. Higher DCR translates into higher copper losses and lower current rating, both of which place an upper limit on the inductance. The recommended range of inductor values for small surface mount inductors as a function of peak current is shown in Figure 4. The values in this range are a good compromise between the trade-offs discussed above. For applications where board area is not a limiting factor, inductors with larger cores can be used, which extends the recommended range of Figure 4 to larger values. 100 PEAK INDUCTOR CURRENT (mA) 300 3639 F04 Figure 4. Recommended Inductor Values for Maximum Efficiency Inductor Core Selection Once the value for L is known, the type of inductor must be selected. High efficiency regulators generally cannot afford the core loss found in low cost powdered iron cores, forcing the use of the more expensive ferrite cores. Actual core loss is independent of core size for a fixed inductor value but is very dependent of the inductance selected. As the inductance increases, core losses decrease. Unfortunately, increased inductance requires more turns of wire and therefore copper losses will increase. Ferrite designs have very low core losses and are preferred at high switching frequencies, so design goals can concentrate on copper loss and preventing saturation. Ferrite core material saturates “hard,” which means that inductance collapses abruptly when the peak design current is exceeded. This results in an abrupt increase in inductor ripple current and consequently output voltage ripple. Do not allow the core to saturate! Different core materials and shapes will change the size/ current and price/current relationship of an inductor. Toroid or shielded pot cores in ferrite or permalloy materials are small and do not radiate energy but generally cost more than powdered iron core inductors with similar characteristics. The choice of which style inductor to use mainly depends on the price versus size requirements and any radiated field/EMI requirements. New designs for surface mount inductors are available from Coiltronics, Coilcraft, TDK, Toko, and Sumida. 3639f For more information www.linear.com/LTC3639 11 LTC3639 Applications Information CIN and COUT Selection The input capacitor, CIN, is needed to filter the trapezoidal current at the source of the top high side MOSFET. CIN should be sized to provide the energy required to magnetize the inductor without causing a large decrease in input voltage (∆VIN). The relationship between CIN and ∆VIN is given by: CIN > L •IPEAK 2 2 • VIN • ∆VIN To prevent large ripple voltage, a low ESR input capacitor sized for the maximum RMS current should be used. RMS current is given by: VIN –1 VOUT This formula has a maximum at VIN = 2VOUT, where IRMS = IOUT/2. This simple worst-case condition is commonly used for design because even significant deviations do not offer much relief. Note that ripple current ratings from capacitor manufacturers are often based only on 2000 hours of life which makes it advisable to further derate the capacitor, or choose a capacitor rated at a higher temperature than required. Several capacitors may also be paralleled to meet size or height requirements in the design. The output capacitor, COUT, filters the inductor’s ripple current and stores energy to satisfy the load current when the LTC3639 is in sleep. The output ripple has a lower limit of VOUT/160 due to the 5mV typical hysteresis of the feedback comparator. The time delay of the comparator adds an additional ripple voltage that is a function of the load current. During this delay time, the LTC3639 continues to switch and supply current to the output. The output ripple can be approximated by: COUT ≥ It is recommended to use a larger value for CIN than calculated by the previous equation since capacitance decreases with applied voltage. In general, a 1µF X7R ceramic capacitor is a good choice for CIN in most LTC3639 applications. V IRMS =IOUT(MAX) • OUT • VIN The output ripple is a maximum at no load and approaches lower limit of VOUT/160 at full load. Choose the output capacitor COUT to limit the output voltage ripple ∆VOUT using the following equation: I 4 • 10 –6 VOUT ∆VOUT ≈ PEAK –ILOAD • + 2 COUT 160 IPEAK • 2 •10 –6 V ∆VOUT – OUT 160 The value of the output capacitor must also be large enough to accept the energy stored in the inductor without a large change in output voltage during a single switching cycle. Setting this voltage step equal to 1% of the output voltage, the output capacitor must be: 2 100% L I COUT > • PEAK • 2 VOUT 1% Typically, a capacitor that satisfies the voltage ripple requirement is adequate to filter the inductor ripple. To avoid overheating, the output capacitor must also be sized to handle the ripple current generated by the inductor. The worst-case ripple current in the output capacitor is given by IRMS = IPEAK/2. Multiple capacitors placed in parallel may be needed to meet the ESR and RMS current handling requirements. Dry tantalum, special polymer, aluminum electrolytic, and ceramic capacitors are all available in surface mount packages. Special polymer capacitors offer very low ESR but have lower capacitance density than other types. Tantalum capacitors have the highest capacitance density but it is important only to use types that have been surge tested for use in switching power supplies. Aluminum electrolytic capacitors have significantly higher ESR but can be used in cost-sensitive applications provided that consideration is given to ripple current ratings and longterm reliability. Ceramic capacitors have excellent low ESR characteristics but can have high voltage coefficient and audible piezoelectric effects. The high quality factor (Q) of ceramic capacitors in series with trace inductance can also lead to significant input voltage ringing. 3639f 12 For more information www.linear.com/LTC3639 LTC3639 Applications Information Input Voltage Steps If the input voltage falls below the regulated output voltage, the body diode of the internal high side MOSFET will conduct current from the output supply to the input supply. If the input voltage falls rapidly, the voltage across the inductor will be significant and may saturate the inductor. A large current will then flow through the high side MOSFET body diode, resulting in excessive power dissipation that may damage the part. If rapid voltage steps are expected on the input supply, put a small silicon or Schottky diode in series with the VIN pin to prevent reverse current and inductor saturation, shown below as D1 in Figure 5. The diode should be sized for a reverse voltage of greater than the regulated output voltage, and to withstand repetitive currents higher than the maximum peak current of the LTC3639. LTC3639 INPUT SUPPLY D1 VIN SW L VOUT COUT CIN 3639 F05 Figure 5. Preventing Current Flow to the Input Ceramic Capacitors and Audible Noise Higher value, lower cost ceramic capacitors are now becoming available in smaller case sizes. Their high ripple current, high voltage rating, and low ESR make them ideal for switching regulator applications. However, care must be taken when these capacitors are used at the input and output. When a ceramic capacitor is used at the input and the power is supplied by a wall adapter through long wires, a load step at the output can induce ringing at the input, VIN. At best, this ringing can couple to the output and be mistaken as loop instability. At worst, a sudden inrush of current through the long wires can potentially cause a voltage spike at VIN large enough to damage the part. For applications with inductive source impedance, such as a long wire, a series RC network may be required in parallel with CIN to dampen the ringing of the input supply. Figure 6 shows this circuit and the typical values required to dampen the ringing. Refer to Application Note 88 for additional information on suppressing input supply transients. LIN LTC3639 VIN R= LIN CIN CIN 3639 F06 4 • CIN Figure 6. Series RC to Reduce VIN Ringing Ceramic capacitors are also piezoelectric. The LTC3639’s burst frequency depends on the load current, and in some applications the LTC3639 can excite the ceramic capacitor at audio frequencies, generating audible noise. This noise is typically very quiet to a casual ear; however, if the noise is unacceptable, use a high performance tantalum or electrolytic capacitor at the output. Output Voltage Programming The LTC3639 has three fixed output voltage modes and an adjustable mode that can be selected with the VPRG1 and VPRG2 pins. The fixed output modes use an internal feedback divider which enables higher efficiency, higher noise immunity, and lower output voltage ripple for 5V, 3.3V, and 1.8V applications. To select the fixed 5V output voltage, connect VPRG1 to SS and VPRG2 to GND. For 3.3V, connect VPRG1 to GND and VPRG2 to SS. For 1.8V, connect both VPRG1 and VPRG2 to SS. For any of the fixed output voltage options, directly connect the VFB pin to VOUT. 3639f For more information www.linear.com/LTC3639 13 LTC3639 Applications Information For the adjustable output mode (VPRG1 = VPRG2 = GND), the output voltage is set by an external resistive divider according to the following equation: VOUT LTC3639 R1 5V 4.2M R1 VOUT = 0.8V • 1+ R2 R2 0.8V 800k The resistive divider allows the VFB pin to sense a fraction of the output voltage as shown in Figure 7. The output voltage can range from 0.8V to VIN. Be careful to keep the divider resistors very close to the VFB pin to minimize noise pick-up on the sensitive VFB trace. 0.8V SS VPRG1 VPRG2 3639 F08 Figure 8. Setting the Output Voltage with External and Internal Resistors chosen to be less than 200k to keep the output voltage variation less than 1% due to the tolerance of the LTC3639’s internal resistor. VOUT VFB LTC3639 VPRG1 VPRG2 VFB R1 RUN Pin and Overvoltage/Undervoltage Lockout R2 3639 F07 Figure 7. Setting the Output Voltage with External Resistors To minimize the no-load supply current, resistor values in the megohm range may be used; however, large resistor values should be used with caution. The feedback divider is the only load current when in shutdown. If PCB leakage current to the output node or switch node exceeds the load current, the output voltage will be pulled up. In normal operation, this is generally a minor concern since the load current is much greater than the leakage. To avoid excessively large values of R1 in high output voltage applications (VOUT ≥ 10V), a combination of external and internal resistors can be used to set the output voltage. This has an additional benefit of increasing the noise immunity on the VFB pin. Figure 8 shows the LTC3639 with the VFB pin configured for a 5V fixed output with an external divider to generate a higher output voltage. The internal 5M resistance appears in parallel with R2, and the value of R2 must be adjusted accordingly. R2 should be The LTC3639 has a low power shutdown mode controlled by the RUN pin. Pulling the RUN pin below 0.7V puts the LTC3639 into a low quiescent current shutdown mode (IQ ~ 1.4µA). When the RUN pin is greater than 1.21V, switching is enabled. Figure 9 shows examples of configurations for driving the RUN pin from logic. The RUN and OVLO pins can alternatively be configured as precise undervoltage (UVLO) and overvoltage (OVLO) lockouts on the VIN supply with a resistive divider from VIN to ground. A simple resistive divider can be used as shown in Figure 10 to meet specific VIN voltage requirements. The current that flows through the R3-R4-R5 divider will directly add to the shutdown, sleep, and active current of the LTC3639, and care should be taken to minimize the impact of this current on the overall efficiency of the application circuit. Resistor values in the megohm range may be required to keep the impact on quiescent shutdown and sleep currents low. To pick resistor values, the sum total of R3 + R4 + R5 (RTOTAL) should be chosen first based on the allowable DC current that can be drawn from VIN. 3639f 14 For more information www.linear.com/LTC3639 LTC3639 Applications Information VIN SUPPLY 4.7M LTC3639 RUN LTC3639 RUN 3639 F09 Figure 9. RUN Pin Interface to Logic Soft-start is implemented by ramping the effective reference voltage from 0V to 0.8V. To increase the duration of the reference voltage soft-start, place a capacitor from the SS pin to ground. An internal 5µA pull-up current will charge this capacitor. The value of the soft-start capacitor can be calculated by the following equation: R3 RUN LTC3639 OVLO R5 R5 VIN(MAX) • < 6V R3 +R4 +R5 Soft-Start VIN R4 Be aware that the OVLO pin cannot be allowed to exceed its absolute maximum rating of 6V. To keep the voltage on the OVLO pin from exceeding 6V, the following relation should be satisfied: 3639 F10 Figure 10. Adjustable UV and OV Lockout The individual values of R3, R4 and R5 can then be calculated from the following equations: R5 = R TOTAL • 1.21V Rising VIN OVLO Threshold R4 = R TOTAL • 1.21V –R5 Rising VIN UVLO Threshold R3 = R TOTAL –R5 –R4 For applications that do not need a precise external OVLO, the OVLO pin can be tied directly to ground. The RUN pin in this type of application can be used as an external UVLO using the previous equations with R5 = 0Ω. Similarly, for applications that do not require a precise UVLO, the RUN pin can be tied to VIN. In this configuration, the UVLO threshold is limited to the internal VIN UVLO thresholds as shown in the Electrical Characteristics table. The resistor values for the OVLO can be computed using the previous equations with R3 = 0Ω. CSS = Soft-Start Time • 5µA 0.8V The minimum soft-start time is limited to the internal soft-start timer of 1ms. When the LTC3639 detects a fault condition (input supply undervoltage or overvoltage) or when the RUN pin falls below 1.1V, the SS pin is quickly pulled to ground and the internal soft-start timer is reset. This ensures an orderly restart when using an external soft-start capacitor. Note that the soft-start capacitor may not be the limiting factor in the output voltage ramp. The maximum output current, which is equal to half of the peak current, must charge the output capacitor from 0V to its regulated value. For small peak currents or large output capacitors, this ramp time can be significant. Therefore, the output voltage ramp time from 0V to the regulated VOUT value is limited to a minimum of Ramp Time ≥ 2COUT V IPEAK OUT 3639f For more information www.linear.com/LTC3639 15 LTC3639 Applications Information Optimizing Output Voltage Ripple After the peak current resistor and inductor have been selected to meet the load current and frequency requirements, an optional capacitor, CISET can be added in parallel with RISET to reduce the output voltage ripple dependency on load current. At light loads the output voltage ripple will be a maximum. The peak inductor current is controlled by the voltage on the ISET pin. The current out of the ISET pin is 5µA while the LTC3639 is active and is reduced to 1µA during sleep mode. The ISET current will return to 5µA on the first switching cycle after sleep mode. Placing a parallel RC network to ground on the ISET pin filters the ISET voltage as the LTC3639 enters and exits sleep mode, which in turn will affect the output voltage ripple, efficiency, and load step transient performance. Higher Current Applications For applications that require more than 100mA, the LTC3639 provides a feedback comparator output pin (FBO) for driving additional LTC3639s. When the FBO pin of a master LTC3639 is connected to the VFB pin of one or more slave LTC3639s, the master controls the burst cycle of the slaves. Figure 11 shows an example of a 5V, 200mA regulator using two LTC3639s. The master is configured for a 5V fixed output with external soft-start and VIN UVLO/OVLO levels set by the RUN and OVLO pins. Since the slave is directly controlled by the master, its SS pin should be floating, RUN should be tied to VIN, and OVLO should be tied to ground. Furthermore, the slave should be configured for a 1.8V fixed output (VPRG1 = VPRG2 = SS) to set the VFB pin threshold at 1.8V. The inductors L1 and L2 do not necessarily have to be the same, but should both meet the criteria described in the Inductor Selection section. Efficiency Considerations The efficiency of a switching regulator is equal to the output power divided by the input power times 100%. It is often VIN CIN R3 R4 R5 VIN SW LTC3639 (MASTER) VFB RUN SS VPRG1 OVLO VPRG2 VOUT 5V COUT 200mA L1 CSS FBO VFB VIN LTC3639 (SLAVE) SW RUN SS VPRG1 OVLO VPRG2 FBO L2 3639 F11 Figure 11. 5V, 200mA Regulator useful to analyze individual losses to determine what is limiting the efficiency and which change would produce the most improvement. Efficiency can be expressed as: Efficiency = 100% – (L1 + L2 + L3 + ...) where L1, L2, etc. are the individual losses as a percentage of input power. Although all dissipative elements in the circuit produce losses, two main sources usually account for most of the losses: VIN operating current and I2R losses. The VIN operating current dominates the efficiency loss at very low load currents whereas the I2R loss dominates the efficiency loss at medium to high load currents. 1. The VIN operating current comprises two components: The DC supply current as given in the electrical characteristics and the internal MOSFET gate charge currents. The gate charge current results from switching the gate capacitance of the internal power MOSFET switches. Each time the gate is switched from high to low to high again, a packet of charge, ∆Q, moves from VIN to ground. The resulting ∆Q/dt is the current out of VIN that is typically larger than the DC bias current. 3639f 16 For more information www.linear.com/LTC3639 LTC3639 Applications Information 2.I2R losses are calculated from the resistances of the internal switches, RSW and external inductor RL. When switching, the average output current flowing through the inductor is “chopped” between the high side PMOS switch and the low side NMOS switch. Thus, the series resistance looking back into the switch pin is a function of the top and bottom switch RDS(ON) values and the duty cycle (DC = VOUT/VIN) as follows: The junction temperature is given by: RSW = (RDS(ON)TOP)DC + (RDS(ON)BOT)(1 – DC) As an example, consider the LTC3639 in dropout at an input voltage of 5V, a load current of 230mA and an ambient temperature of 85°C. From the Typical Performance graphs of Switch On-Resistance, the RDS(ON) of the top switch at VIN = 5V and 100°C is approximately 7.5Ω. Therefore, the power dissipated by the part is: The RDS(ON) for both the top and bottom MOSFETs can be obtained from the Typical Performance Characteristics curves. Thus, to obtain the I2R losses, simply add RSW to RL and multiply the result by the square of the average output current: I2R Loss = IO2(RSW + RL) Other losses, including CIN and COUT ESR dissipative losses and inductor core losses, generally account for less than 2% of the total power loss. Thermal Considerations In most applications, the LTC3639 does not dissipate much heat due to its high efficiency. But, in applications where the LTC3639 is running at high ambient temperature with low supply voltage and high duty cycles, such as dropout, the heat dissipated may exceed the maximum junction temperature of the part. To prevent the LTC3639 from exceeding the maximum junction temperature, the user will need to do some thermal analysis. The goal of the thermal analysis is to determine whether the power dissipated exceeds the maximum junction temperature of the part. The temperature rise from ambient to junction is given by: TR = PD • θJA Where PD is the power dissipated by the regulator and θJA is the thermal resistance from the junction of the die to the ambient temperature. TJ = TA + TR Generally, the worst-case power dissipation is in dropout at low input voltage. In dropout, the LTC3639 can provide a DC current as high as the full 230mA peak current to the output. At low input voltage, this current flows through a higher resistance MOSFET, which dissipates more power. PD = (ILOAD)2 • RDS(ON) = (230mA)2 • 7.5Ω = 0.4W For the MSOP package the θJA is 40°C/W. Thus, the junction temperature of the regulator is: 40°C TJ = 85°C+ 0.4W • W = 101°C which is below the maximum junction temperature of 150°C. Pin Clearance/Creepage Considerations The LTC3639 MSE package has been uniquely designed to meet high voltage clearance and creepage requirements. Pins 2, 4, 13, and 15 are omitted to increase the spacing between adjacent high voltage solder pads (VIN, SW, and RUN) to a minimum of 0.657mm which is sufficient for most applications. For more information, refer to the printed circuit board design standards described in IPC2221 (www.ipc.org). Design Example As a design example, consider using the LTC3639 in an application with the following specifications: VIN = 36V to 72V (48V nominal), VOUT = 12V, IOUT = 100mA, f = 200kHz, and that switching is enabled when VIN is between 30V and 90V. 3639f For more information www.linear.com/LTC3639 17 LTC3639 Applications Information First, calculate the inductor value based on the switching frequency: 12V 12V L= • 1– ≅ 196µH 200kHz • 0.23A 48V Choose a 220µH inductor as a standard value. Next, verify that this meets the LMIN requirement at the maximum input voltage: LMIN = 90V •150ns •1.2 = 70µH 0.23A Therefore, the minimum inductor requirement is satisfied and the 220μH inductor value may be used. Next, CIN and COUT are selected. For this design, CIN should be sized for a current rating of at least: IRMS = 100mA • 12V 36V • – 1≅ 47mARMS 36V 12V The value of CIN is selected to keep the input from drooping less than 360mV (1%) at low line: CIN > 220µH • 0.23A 2 ≅ 0.45µF 2 • 36V • 360mV COUT also needs an ESR that will satisfy the output voltage ripple requirement. The required ESR can be calculated from: ESR < 120mV ≅ 522mΩ 0.23A A 10µF ceramic capacitor has significantly less ESR than 522mΩ. The output voltage can now be programmed by choosing the values of R1 and R2. Since the output voltage is higher than 10V, the LTC3639 should be set for a 5V fixed output with an external divider to divide the 12V output down to 5V. R2 is chosen to be less than 200k to keep the output voltage variation to less than 1% due to the internal 5M resistor tolerance. Set R2 = 196k and calculate R1 as: R1= 12V – 5V • (196kΩ 5MΩ ) = 264kΩ 5V Choose a standard value of 267k for R1. The undervoltage and overvoltage lockout requirements on VIN can be satisfied with a resistive divider from VIN to the RUN and OVLO pins (refer to Figure 10). Choose R3 + R4 + R5 = 2.5M to minimize the loading on VIN. Calculate R3, R4 and R5 as follows: Since the capacitance of capacitors decreases with DC bias, a 1µF capacitor should be chosen. R5 = COUT will be selected based on a value large enough to satisfy the output voltage ripple requirement. For a 1% output ripple (120mV), the value of the output capacitor can be calculated from: 1.21V • 2.5MΩ = 33.6k VIN _ OV(RISING) R4 = 1.21V • 2.5MΩ –R5 = 67.2k VIN _ UV(RISING) 0.23A • 2 •10 –6 COUT ≥ ≅ 10µF 12V 120mV – 160 R3 = 2.5MΩ –R4 –R5 = 2.4M Since specific resistor values in the megohm range are generally less available, it may be necessary to scale R3, R4, and R5 to a standard value of R3. For this example, 3639f 18 For more information www.linear.com/LTC3639 LTC3639 Applications Information choose R3 = 2.2M and scale R4 and R5 by 2.2M/2.4M. Then, R4 = 61.6k and R5 = 30.8k. Choose standard values of R3 = 2.2M, R4 = 62k, and R5 = 30.9k. Note that the falling thresholds for both UVLO and OVLO will be 10% less than the rising thresholds, or 27V and 81V respectively. The ISET pin should be left open in this example to select maximum peak current (230mA). Figure 12 shows a complete schematic for this design example. 2.Connect the (+) terminal of the input capacitor, CIN, as close as possible to the VIN pin. This capacitor provides the AC current into the internal power MOSFETs. 3.Keep the switching node, SW, away from all sensitive small signal nodes. The rapid transitions on the switching node can couple to high impedance nodes, in particular VFB, and create increased output ripple. L1 220µH VIN 36V TO 72V VIN SW LTC3639 2.2M ISET VPRG1 30.9k GND FBO OVLO SS CSS 196k COUT ISET VPRG1 VPRG2 GND R5 SS OVLO VFB RUN 10µF VOUT R1 LTC3639 CIN FBO 62k R3 SW R4 VFB RUN 1µF 267k VOUT 12V 100mA VIN VIN R2 RISET VPRG2 3639 F12 Figure 12. 36V to 72V Input to 12V Output, 100mA Regulator L1 GND PC Board Layout Checklist When laying out the printed circuit board, the following checklist should be used to ensure proper operation of the LTC3639. Check the following in your layout: 1.Large switched currents flow in the power switches and input capacitor. The loop formed by these components should be as small as possible. A ground plane is recommended to minimize ground impedance. CIN VOUT COUT VIN R4 R2 R1 R3 R5 RISET CSS GND VIAS TO GROUND PLANE VIAS TO INPUT SUPPLY (VIN) VIAS TO OUTPUT SUPPLY (VOUT) OUTLINE OF LOCAL GROUND PLANE 3639 F13 Figure 13. Example PCB Layout 3639f For more information www.linear.com/LTC3639 19 LTC3639 Typical Applications Efficiency vs Input Voltage L1 1000µH VOUT* 5V 100mA SW VIN LTC3639 VFB RUN FBO CIN 1µF 250V COUT 10µF 10V SS ISET VPRG1 OVLO VPRG2 GND VOUT = 5V 85 80 VOUT = 3.3V 75 70 3639 F14 CIN: TDK C5750X7R2E105K COUT: TDK C3216X7R1C106M L1: TDK SLF12555T-102MR34 IOUT = 50mA 90 EFFICIENCY (%) VIN 4V TO 150V 95 VOUT = 1.8V 65 *VOUT = VIN FOR VIN < 5V 60 0 30 120 60 90 VIN INPUT VOLTAGE (V) 150 3639 F14b Figure 14. High Efficiency 100mA Regulator L1 150µH VIN 4V TO 150V VIN VOUT 3.3V 100mA SW LTC3639 RUN VFB FBO CIN 1µF 250V 470nF Soft-Start Waveform SS ISET VPRG1 OVLO VPRG2 GND 220k 220pF COUT 22µF 6.3V X5R OUTPUT VOLTAGE 500mV/DIV 10ms/DIV 3639 F15 CIN: TDK C5750X7R2E105K L1: COILCRAFT LPS6235-154ML 3639 F15b Figure 15. Low Output Voltage Ripple 100mA Regulator with 75ms Soft-Start Maximum Load Current vs Input Voltage 4V to 135V Input to –15V Output Positive-to-Negative Regulator CIN 1µF 200V VIN 100 SW LTC3639 RUN 200k VFB FBO SS ISET VPRG1 OVLO VPRG2 GND 102k COUT 10µF 25V VOUT –15V VIN I MAXIMUM LOAD CURRENT ≈ • PEAK VIN + VOUT 2 CIN: VISHAY VJ2225Y105KXCA COUT: AVX 12103C106KAT L1: SUMIDA CDRH105RNP-221NC MAXIMUM LOAD CURRENT (mA) VIN 4V TO 135V L1 220µH VOUT = –5V 90 80 VOUT = –15V 70 60 50 40 30 20 0 30 120 60 90 VIN INPUT VOLTAGE (V) 150 3639 TA04b 3639 TA04a 3639f 20 For more information www.linear.com/LTC3639 LTC3639 Typical Applications 4V to 90V Input to 12V/200mA Output Regulator with Overvoltage Lockout L1 100µH VIN 4V TO 90V UP TO 150V TRANSIENT VIN LTC3639 (MASTER) VFB RUN 1M 267k OVLO CIN1 1µF 200V SS VPRG1 VPRG2 GND 13.7k Low Dropout Startup and Shutdown VOUT* 12V 200mA SW COUT 22µF 16V ISET FBO VIN VIN/VOUT 5V/DIV VOUT L1 CURRENT 200mA/DIV 196k L2 CURRENT 200mA/DIV 3639 TA05b 1s/DIV L2 100µH VIN SW LTC3639 (SLAVE) VFB RUN Overvoltage Lockout Operation OVLO CIN2 1µF 200V SS VPRG1 VPRG2 GND L2 CURRENT 200mA/DIV CIN1/CIN2: VISHAY VJ2225Y105KXCA COUT: TDK C3225X7R1C226M L1/L2: TDK SLF7045T-101MR60-1 *VOUT = VIN FOR VIN < 12V SW 100k RUN VOUT 1.2V 50mA VFB FBO SS ISET VPRG1 OVLO VPRG2 GND COUT 100µF 100k CSET 1nF 200k 3639 TA03a CIN: AVX 2225PC105MAT1A COUT: KEMET C1210C107M9PAC L1: COOPER SD25-331 PEAK-TO-PEAK OUTPUT VOLTAGE RIPPLE (mV) L1 330µH LTC3639 CIN 1µF 250V 3639 TA05c 100ms/DIV Output Voltage Ripple vs Load Current 4V to 150V Input to 1.2V/50mA Output Regulator with Low Output Voltage Ripple VIN 72V L1 CURRENT 200mA/DIV 3639 TA05a VIN 4V TO 150V TRANSIENT TO 150V VIN 50V/DIV VOUT 10V/DIV ISET FBO 45 VIN = 36V 40 COUT = 47µF CSET = OPEN 35 30 25 COUT = 47µF CSET = 1nF 20 15 COUT = 100µF CSET = 1nF 10 5 0 0 10 40 20 30 LOAD CURRENT (mA) 50 3639 TA03b 3639f For more information www.linear.com/LTC3639 21 LTC3639 Typical Applications Maximum Load and Input Current vs Input Voltage 40V to 150V Input to 36V/100mA Output with 25mA Input Current Limit SW VIN LTC3639 R1 715k CIN 1µF 250V R2 5k 221k 120 VOUT 36V 100mA* MAXIMUM CURRENT (mA) L1 220µH VIN 40V TO 150V VFB RUN ISET FBO OVLO GND SS VPRG1 VPRG2 35.7k COUT 2.2µF 50V 3639 TA06a INPUT CURRENT LIMIT = 100 MAXIMUM LOAD CURRENT 80 60 40 MAXIMUM INPUT CURRENT 20 VOUT R2 5µA •R1 VOUT R2 • • 1+ • ≈ 10 R1+R2 10 R1+R2 VIN 0 VIN • 25mA ≤ 100mA 36V CIN: MURATA GRM55DR72E105KW01L COUT: TDK C3225X7R1H225M L1: WÜRTH 744 778 922 2 *MAXIMUM LOAD CURRENT = 40 50 60 70 80 90 100 110 120 130 140 150 VIN INPUT VOLTAGE (V) 3639 TA06b Burst Frequency vs Load Current 100 VIN = 36V BURST FREQUENCY (kHz) WITH BURST FREQUENCY LIMIT 5V to 150V Input to 5V/100mA Output with 20kHz Minimum Burst Frequency VIN 5V TO 150V CIN 1µF 250V L1 220µH VIN SW 953k LTC3639 RUN VFB ISET FBO VPRG2 VPRG1 OVLO SS GND V+ LTC6994-1 IN OUT DIV 100k 30Ω 2N7000 VOUT 5V 100mA 10 1 WITHOUT BURST FREQUENCY LIMIT 0.1 0.1 COUT 10µF 10V 100 3639 TA08b SET GND 1 10 LOAD CURRENT (mA) Input Current vs Load Current 200k 100 VIN = 36V CIN: KEMET C2225C105KARACTU COUT: MURATA GRM40X5R106K10H520 L1: BOURNS SRR1005-221KCT-ND INPUT CURRENT (mA) 3639 TA08a 10 WITH BURST FREQUENCY LIMIT 1 0.1 WITHOUT BURST FREQUENCY LIMIT 0.01 0.1 1 10 LOAD CURRENT (mA) 100 3639 TA08c 3639f 22 For more information www.linear.com/LTC3639 LTC3639 Package Description Please refer to http://www.linear.com/designtools/packaging/ for the most recent package drawings. MSE Package Variation: MSE16 (12) 16-Lead Plastic MSOP with 4 Pins Removed Exposed Die Pad (Reference LTC DWG # 05-08-1871 Rev C) BOTTOM VIEW OF EXPOSED PAD OPTION 2.845 ±0.102 (.112 ±.004) 5.23 (.206) MIN 2.845 ±0.102 (.112 ±.004) 0.889 ±0.127 (.035 ±.005) 8 1 1.651 ±0.102 (.065 ±.004) 1.651 ±0.102 3.20 – 3.45 (.065 ±.004) (.126 – .136) 16 0.305 ±0.038 (.0120 ±.0015) TYP 0.50 (.0197) 1.0 BSC (.039) BSC RECOMMENDED SOLDER PAD LAYOUT 0.254 (.010) 0.35 REF 4.039 ±0.102 (.159 ±.004) (NOTE 3) 0.12 REF DETAIL “B” CORNER TAIL IS PART OF DETAIL “B” THE LEADFRAME FEATURE. FOR REFERENCE ONLY 9 NO MEASUREMENT PURPOSE 0.280 ±0.076 (.011 ±.003) REF 16 14 121110 9 DETAIL “A” 0° – 6° TYP 3.00 ±0.102 (.118 ±.004) (NOTE 4) 4.90 ±0.152 (.193 ±.006) GAUGE PLANE 0.53 ±0.152 (.021 ±.006) DETAIL “A” 1.10 (.043) MAX 0.18 (.007) SEATING PLANE 0.17 – 0.27 (.007 – .011) TYP 1 0.50 (.0197) BSC 3 567 8 1.0 (.039) BSC NOTE: 1. DIMENSIONS IN MILLIMETER/(INCH) 2. DRAWING NOT TO SCALE 3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS. MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE 4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS. INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE 5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX 6. EXPOSED PAD DIMENSION DOES INCLUDE MOLD FLASH. MOLD FLASH ON E-PAD SHALL NOT EXCEED 0.254mm (.010") PER SIDE. 0.86 (.034) REF 0.1016 ±0.0508 (.004 ±.002) MSOP (MSE16(12)) 0911 REV C 3639f Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. For more information www.linear.com/LTC3639 23 LTC3639 Typical Application 12V/100mA Automotive Supply L1 470µH VIN SW 90 SS GND VPRG1 VPRG2 196k 60 50 100 40 30 POWER LOSS 10 20 3639 TA07 10 *VOUT = VIN FOR VIN < 12V CIN: AVX 2225PC105MAT1A COUT: KEMET C1206C475K4RAC L1: COILCRAFT MSS1048T-474KL VIN = 24V 1000 VIN = 48V VIN = 120V 70 POWER LOSS (mW) ISET FBO OVLO COUT 4.7µF 16V X7R EFFICIENCY 80 VFB RUN CIN 1µF 250V X7R 267k LTC3639 100 VOUT 12V* 100mA EFFICIENCY (%) VIN 4V TO 150V Efficiency and Power Loss vs Load Current 0 0.1 1 1 10 LOAD CURRENT (mA) 100 3639 TA07b Related Parts PART NUMBER DESCRIPTION COMMENTS LTC3630 65V, 500mA Synchronous Step-Down DC/DC Converter VIN: 4V to 65V, VOUT(MIN) = 0.8V, IQ = 12µA, ISD < 3µA, 3mm × 5mm DFN16, MSOP16E Packages LTC3642 45V (Transient to 60V), 50mA Synchronous StepDown DC/DC Converter VIN: 4.5V to 45V, VOUT(MIN) = 0.8V, IQ = 12µA, ISD < 3µA, 3mm × 3mm DFN8, MSOP8 Packages LTC3631/LTC3631-3.3 45V (Transient to 60V), 100mA Synchronous StepLTC3631-5 Down DC/DC Converter VIN: 4.5V to 45V, VOUT(MIN) = 0.8V, IQ = 12µA, ISD < 3µA, 3mm × 3mm DFN8, MSOP8 Packages LTC3632 50V (Transient to 60V), 20mA Synchronous StepDown DC/DC Converter VIN: 4.5V to 45V, VOUT(MIN) = 0.8V, IQ = 12µA, ISD < 3µA, 3mm × 3mm DFN8, MSOP8 Packages LT®3990/LT3990-3.3/ LT3990-5 62V, 350mA 2.2MHz High Efficiency Micropower Step-Down DC/DC Converter with IQ = 2.5µA VIN: 4.2V to 62V, VOUT(MIN) = 1.21V, IQ = 2.5µA, ISD < 1µA, 3mm × 2mm DFN10, MSOP10 Packages LT3970/LT3970-3.3 LT3970-5 40V, 350mA 2.2MHz High Efficiency Micropower Step-Down DC/DC Converter with IQ = 2.5µA VIN: 4.2V to 40V, VOUT(MIN) = 1.21V, IQ = 2.5µA, ISD < 1µA, 3mm × 2mm DFN10, MSOP10 Packages LTC3810 100V Synchronous Step-Down DC/DC Controller VIN: 6.4V to 100V, VOUT(MIN) = 0.8V, IQ = 2mA, ISD < 240µA, SSOP28 Package LTC3891 60V Synchronous Step-Down DC/DC Controller with VIN: 4V to 60V, VOUT(MIN) = 0.8V, IQ = 50µA, ISD < 14µA, 3mm × 4mm QFN20, TSSOP20E Packages Burst Mode Operation 3639f 24 Linear Technology Corporation 1630 McCarthy Blvd., Milpitas, CA 95035-7417 For more information www.linear.com/LTC3639 (408) 432-1900 ● FAX: (408) 434-0507 ● www.linear.com/LTC3639 LT 0413 • PRINTED IN USA LINEAR TECHNOLOGY CORPORATION 2013