LM1572
1.5A, 500kHz Step-down Voltage Regulator
General Description
Features
The LM1572 is a 500kHz step-down (buck) switching voltage regulator capable of driving up to 1.5A in to a load while
occupying a very small PCB area. Current Mode Control
results in superior transient response and regulation over a
wider range of operating conditions. National’s advanced
analog bipolar, CMOS plus DMOS process enables high
efficiency at high switching frequency, and the internal
150mΩ MOSFET switch provides more power from a
smaller package.
The LM1572 has programmable soft-start and frequency
foldback to limit the inrush current, and a TTL compatible
shutdown for easy sequencing. It draws 2.3mA of supply
current in standby mode, and only 26µA in shutdown mode.
The LM1572 is available in a TSSOP-16 package with an
adjustable output or fixed outputs of 5V and 3.3V. The adjustable version can be set between 2.42V and 5V.
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500kHz clock allows small, surface mount components
150mΩ MOSFET switch
Guaranteed load current of 1.5A
Current mode control
Programmable soft-start
Internally set slope compensation
TTL compatible shutdown
Fixed 5V, 3.3V or adjustable output
Low shutdown supply current of 26µA
Cycle-by-cycle current limit
Short-circuit protection and thermal protection
TSSOP-16 package
Applications
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LCD Monitors and TVs
Set-Top Boxes
Cable Modems
Down conversion from 12V in local/distributed systems
Typical Applications (Fixed/Adjustable Voltage Parts)
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© 2002 National Semiconductor Corporation
DS200333
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LM1572 1.5A, 500kHz Step-down Voltage Regulator
July 2002
LM1572
Typical Applications (Fixed/Adjustable Voltage Parts)
(Continued)
20033314
Connection Diagram
20033315
16-Lead TSSOP
NS Package Number MTC16
adjustable part. Note that the NPN signal transistor shown,
must have a guaranteed hfe greater than 400. Damage can
occur to Pin 1 if it is connected (via a diode) to any external
voltage source greater than 6V.
AVIN (Pin 2) - This is the Analog VIN and provides the supply
to the internal control circuitry, the (Power) VIN pins (Pins
3,4) providing the supply to the internal power stage. In the
simplest layout scheme, the Analog VIN pin can simply be
connected to the VIN pins directly on the pads where the IC
is mounted. But for better noise rejection the trace to Pin 2
can be routed separately from the (Power) VIN trace, starting
from the positive terminal of the input capacitor. A simple RC
filter solution can also be used for better results, particularly
at low input voltages. This consists of a 10Ω resistor connected between Analog VIN and VIN, and a 0.47µF capacitor
between Analog VIN and Ground. Note that if this RC filter is
used, a 1MΩ resistor between Pin 1 and Ground is also
required.
Pin Description
BOOT (Pin 1) - Bootstrap pin. It provides the upper rail for
the floating driver stage of the internal MOSFET switch, the
lower rail being the switching node (Pins 5 and 6). A small
decoupling capacitor (typically 0.1µF-0.22µF) is therefore
connected between the Bootstrap pin and the switching
node. This capacitor should be 0.18µF-0.22µF for applications with an output voltage greater than 3.3V, if the minimum load (including the current drawn by internal/external
feedback resistor divider) is less than 1mA. Additional drive
voltage is provided by connecting this pin directly to the 5V
output rail via a diode as shown in the Typical Application
Circuit for the fixed voltage part. The same method can be
used for an adjustable part provided the part is adjusted for
an output of 5V. For other output voltages (between 2.42V to
5V) a more general method of providing this external drive
voltage is illustrated in the Typical Application Circuit for the
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NC (Pins 10,11) - No Internal Connection.
SS (Pin 12) - Softstart pin. A small capacitor connected from
this pin to ground programs the amount of softstart. This
capacitor charges up by means of an internal 4.5µA current
source, during power-up, and also whenever the output of
the converter is enabled. The allowed duty cycle increases
slowly as the capacitor charges, reaching the maximum
allowed when the voltage on this pin approaches 2V. The
capacitor continues to charge, finally reaching 6V, at which
level it is internally clamped. This pin is internally forced to
ground (to discharge the softstart capacitor and to reset the
softstart function) whenever the shutdown pin is taken below
2.38V. If the softstart feature is not required, the softstart pin
can be left floating.
FB (Pin 15) - This is the feedback pin for the IC and is used
to set the output of the converter to regulate to the desired
value. For the fixed voltage part this pin is normally connected directly to the output. For the adjustable part, a
resistive divider is used between the output and ground, so
that the voltage on this pin is 2.42V when the output is at the
required level. For fixed voltage parts, the internal divider
draws about 0.5mA, a consideration possibly affecting the
choice of the bootstrap capacitor (see description of Pin 1
above).
COMP (Pin 16) - This is the output of the error transconductance amplifier and is used for frequency compensation of
the feedback loop. A small capacitor from this pin to ground
(about 3.3nF to 6.8nF) provides the simplest loop compensation, but a series resistor-capacitor combination (R between 1k to 1.5k) may also be used to improve the phase
margin/crossover frequency of the loop. The voltage on this
pin is at about 1V at very light loads. Under very heavy loads
or under output short-circuit, the voltage on this pin clamps
to 2V, and the converter enters protective foldback. The IC
automatically recovers from this mode when the load is
reduced.
(Continued)
VIN (Pins 3,4) - This is the input supply to the power stage
(connected to the Drain of the switching MOSFET). To aid
thermal dissipation from the die, two pins are used for this
function. Both these pins must be connected together, very
close to the IC, onto a large PCB copper plane.
SW (Pins 5,6) - This is the Source of the internal switching
MOSFET and forms the ’switching node’ of the buck converter. These two pins should be connected together on the
PCB close to the IC. The length of the trace from this node to
the cathode of the catch diode, and from the anode of the
diode to the IC ground must be kept very small. The maximum inductance connected to the switching node (for any
application) is recommended to be 15µH. See the Inductor
Selection procedure for more details.
GND (Pins 7,9,13,14) - This is the Ground for the IC and for
the input and output rails of the buck converter. To aid
thermal dissipation from the die, four pins are used for this
function. Connect as many as possible of these ground pins
together, close to the IC onto a large PCB copper plane. A
two-sided PCB with one side serving as a ’ground plane’ is
strongly recommended. The ground pins must then connect
to the ground plane very close to the IC through several vias.
The vias also serve to transfer heat to the other side of the
board for better thermal management.
SD (Pin 8) - Shutdown/Standby/UVLO Pin. This pin actually
has two thresholds. If it is taken below 2.38V (typical), the
switch turns off and the output of the converter falls to
zero.This is the ’standby mode’. The internal circuitry of the
IC remains active, continuing to draw about 2.3mA from the
input. If the voltage on this pin is lowered below 1V (typical),
the IC enters ’shutdown mode’ drawing only 26 µA from the
input. Above 2.38V, the switching action resumes, and so
this pin can also be used to set an undervoltage lockout
threshold (UVLO) for the input rail. If this pin is not intended
to be used actively, it can be left floating to allow continuous
switching. The voltage on this pin should not exceed 7V to
avoid damage.
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LM1572
Pin Description
LM1572
Block Diagram
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(Note 1)
Junction Temperature
If Military/Aerospace specified devices are required,
please contact the National Semiconductor Sales Office/
Distributors for availability and specifications.
ESD Tolerance (Note 2)
Input Voltage
SD Pin Voltage
FB Pin Voltage (All Options)
Storage Temp. Range
150˚C
Operating Ratings
2kV
Supply Voltage (VIN) (Note 3)
8.5V to 16V
17V
Junction Temperature Range
−40˚C to +125˚C
7V
Package Thermal Resistance
(TSSOP-16) (Note 4)
130˚C/W
7V
−65˚C to 150˚C
Electrical Characteristics
Unless otherwise specified, all limits are guaranteed for TA = 25˚C, VIN = 15V, VCOMP = 1.5V, VSD = 5V, ILOAD = 0A, unless
otherwise noted. Boldface apply over the temperature extremes. ’VFB low (high)’ is 0.95 (1.05) times the nominal value at
regulation.
Symbol
Parameter
Min
(Note 5)
Conditions
Typ
(Note
6)
Max
(Note 5)
Units
VFB_ADJ
Voltage on Feedback pin
(Adjustable version in regulation)
2.37
2.35
2.42
2.49
2.5
V
VFB_5
Voltage on Feedback pin (Fixed
5V version in regulation)
4.85
4.8
5.0
5.15
5.2
V
VFB_3.3
Voltage on Feedback pin (Fixed
3.3V version in regulation)
3.22
3.16
3.3
3.4
3.44
V
∆VFB/VIN
Feedback Voltage Line
Regulation
VIN = 8.5V to VIN = 16V
−0.05
0
0.05
%/V
IFB_REG
Feedback Pin Bias Current
(Adjustable Part)
VFB at regulation
0
0.5
1.5
µA
AVERROR
Error Amplifier Voltage Gain
(Note 7)
gmEA
Error Amplifier
Transconductance (Note 7)
2700
3200
µMho
gmCOMP_SW
Comp Pin to Switch Current
Transconductance
IEA_SOURCE
Error Amplifier Source Current
VFB low
IEA_SINK
Error Amplifier Sink Current
VFB high
2.4
mA
VCOMP_TH
Comp Pin Switching Threshold
Duty Cycle = 0
0.9
V
VCOMP_LIM
Comp Pin High Clamp
ICLIM
Switch Current Limit
350
1100
800
2000
2
50
200
A/V
300
2
VBOOT = VSW + 5V,
Comp Open,
VFB low
2.0
2.7
3.2
D = 0.8
1.75
2.4
3
0.15
0.4
0.5
Switch ON Resistance
ISW = 1.5A, VBOOT = VIN + 5V
DMAX
Maximum Duty Cycle
(Note 8)
Comp Open,
VFB low
fSW
Switch Frequency
VFB low, VCOMP = 1V,
86
94
Full Temp.
Range
400
500
-20˚C ≤ TJ ≤
125˚C
440
fREG
Switch Frequency Line
Regulation
VIN = 8.5V and
VIN = 16V,VFB low, VCOMP = 1V
∆fFOLDBACK
Foldback Frequency shift
(Adjustable part)
VFB = 0.8V, VCOMP = 1V
5
V
D≤ 0.5
RDS
A
Ω
%
570
kHz
560
0.01
20
µA
90
%/V
160
kHz
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LM1572
Absolute Maximum Ratings
LM1572
Electrical Characteristics
(Continued)
Unless otherwise specified, all limits are guaranteed for TA = 25˚C, VIN = 15V, VCOMP = 1.5V, VSD = 5V, ILOAD = 0A, unless
otherwise noted. Boldface apply over the temperature extremes. ’VFB low (high)’ is 0.95 (1.05) times the nominal value at
regulation.
Symbol
Parameter
Conditions
Min
(Note 5)
Typ
(Note
6)
2.5
Max
(Note 5)
Units
ISS
Softstart Pin Current
VSS = 1V,VFB=0V
4.5
8
µA
ISD
Shutdown Supply Current
VSD = 0V,VCOMP = 1V,VFB low
26
52
75
µA
ISTDBY
Standby Supply Current
VSD = 1.5V, Comp Open
2.3
4
4.3
mA
VUVLO
Undervoltage Lockout Threshold
Comp Open, VFB low
2.2
2.38
2.5
V
VSD
Shutdown Threshold
Comp Open, VCOMP = 1V, VFB low
0.75
1.0
1.28
V
Note 1: Absolute maximum ratings indicate limits beyond which damage to the device may occur. Operating ratings indicate conditions for which the device is
intended to be functional, but specific performance is not guaranteed. For guaranteed specifications and the test conditions, see the Electrical Characteristics.
Note 2: This is for the human body model, which is a 100pF capacitor discharged through a 1.5k resistor into each pin.
Note 3: Minimum input voltage is defined as the voltage where internal bias lines are still regulated so that the reference voltage and oscillator remain constant.
Actual minimum input voltage to maintain output in regulation depends on output voltage and load current. In particular, the required duty cycle must be less than
the lowest possible upper duty cycle limit of the controller (DMAX = 0.86). The maximum input voltage will also depend on output voltage and load current. In
particular, the required duty cycle must be greater than the lowest possible duty cycle limit of the controller (DMIN = 0.15), estimated from the typical minimum
on-time, which is about 300ns.
Note 4: Junction to Ambient thermal resistance with the TSSOP-16 package soldered on a 1oz. printed circuit board with copper area of approximately 1in2.
Note 5: All limits guaranteed at room temperature (standard face type) and at temperature extremes (bold face type). All room temperature limits are 100%
production tested. All limits at temperature extremes are guaranteed via correlation using Statistical Quality Control (SQC) methods. All limits are used to calculate
Average Outgoing Quality Level (AOQL).
Note 6: Typical numbers are at 25˚C and represent the most likely norm.
Note 7: Transconductance and voltage gain refer to the internal amplifier, excluding any voltage divider as is present on the fixed voltage parts. To calculate the gain
and transconductance for the fixed voltage parts, divide values shown in table by the ratio VFB_5/2.42 = 2.07 for the 5V part and by VFB_3.3/2.42 = 1.36 for the 3.3V
part.
Note 8: To ensure stable operation, the maximum recommended operating duty cycle is 80%.
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LM1572
Typical Performance Characteristics
Efficiency (3.3VOUT)
Characteristics of Switch
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Efficiency (5VOUT)
Supply Current vs Load
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20033303
Supply Current vs Voltage on Shutdown Pin
Characteristics of Shutdown Pin
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LM1572
Typical Performance Characteristics
(Continued)
Shutdown Pin Current at Shutdown Threshold
Shutdown Supply Current vs Input Voltage
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20033307
Frequency Foldback
Characteristics of Feedback Pin
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20033310
Operating Feedback Pin Current vs Input Voltage
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Resistive Divider Calculation
In (peak) current mode control, the main additional consideration is the phenomenon of subharmonic instability (also
called alternate cycle or half-frequency oscillations). This is
fundamental to the topology, and no amount of ’tweaking’ the
compensation resistor/capacitor values will circumvent it.
The well known solution is to add a certain amount of ’slope
compensation’, the value of which is directly related to the
inductance being used. Higher inductance requires smaller
slope compensation. If the slope compensation is fixed, as
for the LM1572, it is the inductance that needs to be controlled. Then higher slope compensation requires smaller
inductance. This defines a ’minimum’ value of inductance
required to avoid subharmonic instability. The value can
therefore be exceeded. If for example the first priority in a
given application is not the size of the inductor, but the
reduction of output ripple, a higher than the minimum inductance may be selected. But too high an inductance, for a
given slope compensation (or equivalently too much of slope
compensation for a given inductance), will cause the loop
response to become more and more that of voltage mode
control, eventually making it slower and harder to compensate. For any LM1572 design therefore, the maximum recommended inductance is 15µH, irrespective of input or output conditions.
For the adjustable part, the voltage on the feedback pin is set
to 2.42V under regulation. This is achieved by means of a
resistive divider, as indicated in the Typical Applications for
the adjustable part. Designating the upper resistor as ’R2’
(connected to the output) and the lower resistor as ’R1’
(connected to ground), the following equation relates R1, R2
and the output voltage level VO :
Setting the lower resistor to 2.21k (which is a standard
resistance value), the upper resistor is chosen as 806 ohms
for a 3.3V output and as 2.37k for a 5V output. This should
suffice for most applications. However the more experienced
designer may like to know more about the rather overlooked
intricacy of selecting resistors especially in regard to the
resultant error in the output voltage. It is also helpful to
consider the other factors affecting the tolerance of the
output voltage. This is disussed under ’Tolerance of set
Output Voltage’ at the end of ’Application Information’. Note
that if the the lower resistor is set to 2.21k, the divider current
is greater than 1mA, so a 0.1µF boostrap will always suffice
(see Pin Descriptions for Pin 1 and Pin 15 above).
For the LM1572, the slope compensation can vary (from
device to device) over the range 0.42 to 0.75 A/µs. A little
thought will lead to the conclusion that any calculation for the
minimum inductance (required to avoid subharmonic instability), must be carried out at its ’worst-case’: which is the
lower limit of the slope compensation (i.e. 0.42 A/µs). This
also happens to be the value used for peak power calculation since it corresponds to the lower limit of current limit
(2A). The value of 0.75 A/µs can be used to check if the
slope compensation is not ’excessive’ in the sense discussed above.
The effective current limit, ’ICLIM’ (see Electrical Characteristics) is the sum of two terms. The first is the basic preset
current limit (the flat part) , which we call ’ICL’ here, and is the
value given for ’ICLIM’ for D ≤ 0.5). Superimposed on this is
the effect of slope compensation. This causes the current
limit to fall (almost linearly) for D > 0.5. In general, the slope
compensation can be expressed as ’mC’ in units of A/µs.
From D = 0.5 to a projected value of D = 1 (a time interval of
1µs), the current limit would therefore fall exactly by mC
Amps. At D = 0.8 the current limit falls by 3/5th of this i.e. by
mC*0.6. So the current limit at D = 0.8 would be ICL −
(0.6*mC). This value (’ICLIM’ for D = 0.8 ) is also given in the
Electrical Characteristics tables.
As mentioned, the inductance must be chosen to be higher
than the minimum value corresponding to the condition of
peak calculated switch current equal to the current limit. The
worst case must be used here: i.e. the ’min’ of current limit
values in the Electrical Characteristics (not ’typ’). Further, it
should be confirmed over the entire input voltage range (or
duty cycle) that the peak current does not attempt to exceed
the effective current limit. This is easily carried out using the
same general strategy: by calculating the minimum inductance at both input voltage extremes, and then choosing the
greater of the two calculated ’minimum’ inductances.
It should also be remembered that subharmonic instability
can only occur when several conditions are simultaneously
satisfied: (peak) current mode control, duty cycle greater
than (or around) 0.5, and continuous conduction mode. Sub-
Inductor Selection
Inductor selection for buck converters is discussed in great
detail in AN-1197, to which the reader can refer to for a
deeper understanding. It must be understood that though the
scope of the above Application Note is limited to buck converters that rely on voltage mode control, all the considerations contained therein also apply to buck converters relying on current mode control, such as the LM1572. In fact,
with current mode control, there are additional considerations that may apply which need to be discussed here.
The basic requirement for any converter is that it should be
able to deliver the required power without hitting the current
limit of the switch. This is ensured by having an inductance
large enough to limit the peak current (this is obviously not
feasible if the required load current is very close to or larger
than the current limit!). In the LM1572, a ’slope compensation’ ramp is also summed-in with the switch current ramp,
for duty cycles greater than 0.5. The reason for this slope
compensation will be explained later below, here it suffices
to realize that it affects the effective current limit for duty
cycles greater than 0.5. From the Electrical Characteristics it
can be seen that the current limit ICLIM is stated as two
terms: one for D less than (or equal to) 0.5, and one for D =
0.8. Since the current limit falls off at high duty cycles/low
input voltage due to the slope compensation, a peak power
calculation should generally be done both at the highest and
the lowest input voltage, so as to ensure that the inductance
is large enough to cover the entire desired operating input
voltage range.
The overall strategy here is to determine various ’minimum
inductances’ based on all different considerations (as applicable), and to then pick the largest of all the ’minimums’ so
as to satisfy each of the conditions.
It is noted here that there can also be an ’optimum’ value for
the inductance, one which offers a compromise solution for
reducing the overall size of the power converter, the magnetics and capacitors included. However, since the primary
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LM1572
reason for going to higher switching frequencies is to reduce
the size of the magnetics alone, ’optimization’ may be relegated to a lower priority, as in the example to follow.
Application Information
LM1572
Application Information
LMIN_CL_CCM = 5.2µH
(Continued)
There is one more possible influence on the value of minimum inductance, which is now discussed.
Step 3: Subharmonic Instability
harmonic instability is not of concern if any one or more of
the above conditions are not true. And in that case, the
inductor selection considerations become identical to those
for voltage mode control. Note that the worst case condition
to check for subharmonic instability, and/or to choose an
inductance large enough to prevent these oscillations, is at
the lowest desired input voltage.
The LM1572 is current mode controlled. If the minimum input
voltage is less than roughly twice the output voltage, the duty
cycle is close to or greater than 0.5. That makes two of the
three conditions required for subharmonic instability. Therefore also assuming continuous conduction mode, the value
of inductance must be large enough to prevent these oscillations (occurring at f/2). The relevant equation is
Designing for discontinuous conduction mode is clearly an
attractive option for some experienced designers. One reason for this is that subharmonic instability is then of no
concern. However discontinuous mode is possible only if the
maximum load is less than half the current limit. Further, the
design procedure is rather complicated and iterative too.
Therefore it is considered out of the scope of this section.
On the left is the minimum inductance required. The right
side contains ’Q, which is the quality factor of the half frequency peaking prior to the outbreak of subharmonic oscillations. Q should typically not be greater than 2, or subharmonic oscillations become increasingly likely. Further, to
avoid voltage mode control type of response (excessive
slope compensation), neither must Q be less than about 0.2.
Very large values of L lead to smaller and smaller Q. So
acceptable values of Q usually lie within the range 0.2 to 2.
This calculation should always be done at the worst case:
i.e. the minimum input voltage.
The required equation (for Continuous Conduction Mode,
’CCM’) is
where the duty cycle is
VSW and VD are the forward drops across the switch and the
diode respectively. A sample calculation follows.
Example: The input voltage range is 8.5-16V. The output is
5V @ 1.5A. It can be assumed that both the switch forward
drop and diode drop are 0.5V, by default.
Step 1: Current Limit at maximum input
LMIN_f/2 = 6.2µH
But what is the optimum value? This is examined next.
Step 4: Optimum Inductance
The designer should be clear that choosing smaller inductance values may not necessarily lead to smaller sized inductors. The relationship between inductance and inductor
size is not always intuitive. The size of an inductor is determined by its energy handling requirement which is 1⁄2*L*IP2.
So very small inductors may lead to excessively high peak
currents, which can also increase the size of the input and
output capacitors. The reader is referred to AN-1197 for
further details. There it is shown that ’r’ , the current ripple
ratio, should be around 0.4. Using this as the yardstick, the
’optimum’ value of inductance is
The duty cycle at 16V input is
DCCM = 0.344
ICL = 2A (min value), so the required minimum inductance is
LMIN_CL_CCM = 7.2µH
Step 2: Current Limit at maximum input
The calculation is repeated at the lowest desired input voltage of 8.5V. The duty cycle at this point is
This calculation must always be done at the maximum input
voltage, which is 16V for this example
The current limit at this duty cycle is
ICLIM = ICL − {(mC • T) • (D − 0.5)} A
ICLIM = 2 − {(0.42 • 2) • (0.65 − 0.5)} A
ICLIM = 1.87A
So the minimum inductance at this point is
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LOPT = 12µH
Step 5: Conclusions
Several ’minimum inductances’ were calculated: 7.2, 5.2 and
6.2µHs. The optimum value is 12µH. A standard value higher
than all the ’minimums can be picked. Here, a standard
8.2µH/1.5A was chosen so as to provide the smallest sized
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overshoot was only about 10% (plus the waveform was that
of a well damped system). The designer should also be
aware that some older DC power supplies actually exacerbate the problem, while apparently trying to ’correct’ the
output voltage. The situation gets even worse if the DC
power supply has a remote-sense which is being used to
apparently ’correct’ the input voltage at the input of the
converter. Therefore, it is always a good idea to try out
another available DC power supply to see how severe the
problem is in reality, or whether it is just a ’bad’ lab supply.
(Continued)
inductor for the application. 10µH is a more widely available
standard value, and very close to the optimum value too, and
would therefore be a good choice too. Note that inductances
larger than 15µH are not recommended in general.
To more accurately predict how the selected off-the-shelf
part will actually perform in the real application, the designer
is referred to AN-1197. The procedure contained therein
could greatly help in correctly choosing the lowest acceptable current/energy rating of the inductor, and thereby reducing its size further.
If because of size constraints the designer must use tantalums, a minimum capacitance of 22µF is recommended for
any application, irrespective of input/output conditions. This
’softens’ up the input dV/dt significantly and reduces the
ringing.
The basic electrical criterion for selecting an input capacitor
is the input RMS current. The equation for this is
Input Capacitor Selection
At the input, the first requirement is a high frequency (preferably ceramic) decoupling capacitor, of value 0.1µF, placed
very close to, and between the VIN pins and the Ground Pins
of the IC. This provides the triangular pulsed current waveform that flows through the switch. In addition, a bulk capacitor is also required, which replenishes the decoupling capacitor, and may be placed slightly further away if necessary.
The rating and selection of this capacitor is discussed below.
In general a standard low-esr aluminum electrolytic is recommended at the input (’esr’ refers to the equivalent series
resistance hereafter). There are several reasons for this.
Firstly, tantalum capacitors have inherent input surge-current
limitations. So when the input surge current comes from a
very low impedance source (such as a high current lab DC
power supply), there is a chance that the capacitor may not
survive several such repeated high dV/dt events. In any
case, even using ’surge-tested’ tantalums (like TPS series
from AVX) a 50% voltage derating is recommended in such
conditions. Therefore hypothetically, a 35V tantalum must be
used for the preceding example, in which the maximum input
was 16V. The second reason for avoiding very low esr input
capacitors is that there is a possibility of severe input oscillations. The elements involved in this resonance are the
inductance of the input leads, the input capacitance and the
(negative) input impedance of the switching stage. It is
known that the esr of the input capacitor actually serves a
useful purpose in damping out these oscillations.
These oscillations can only be seen clearly under lab conditions if the output of the lab DC power supply is ON/OutputEnabled and then the lead from the converter stage is physically connected to the output terminals of the DC power
supply. Just turning the DC power supply ON/OFF (or with
an Output-Enable button) does not generate the high dV/dt
required to provoke these oscillations. Under a real situation,
input oscillations can become severe enough to cause the
maximum voltage rating of the IC to be exceeded. The
ringing can in turn, also feed in to the Analog sections of the
LM1572, causing strange behavior and possibly device failure.
The designer needs to therefore monitor the input ramp
close to the input of the converter, preferably with a digitizing
oscilloscope set to about 10-20µs/div and using the single
acquisition mode. Once the ramp is being captured, it will be
seen that large input capacitances ’slow’ the dV/dt considerably, thereby reducing the overshoot and the input oscillations. However, besides the capacitance itself, the esr of the
input capacitor is a major contributor too. Therefore, in a
typical comparison of a 10µF aluminum electrolytic vs. a
10µF tantalum electrolytic (tantalum has lower esr), it was
seen that there was an almost 50% overshoot in the peak
input voltage for the tantalum capacitor (accompanied by
severe ringing), whereas for the aluminum capacitor, the
where ’r’ is the current ripple ratio. It is given by
where L is in µH and f in Hz. This calculation should be done
at the worst case condition for this parameter, which corresponds to 50% duty cycle. If the application never ’sees’ 50%
duty cycle over its entire operating range, then the worst
case is simply the closest duty cycle to 50%. This can, in
general, occur at either of the input voltage extremes, and
therefore both ends must then be examined. In the example,
it was seen that the duty cycle varies from 34.4% to 65%. So
it is clear that there does exist an input voltage point within
the range, at which the duty cycle is 50%. At this worst-case
condition, for the chosen inductor, ’r’ at D=0.5 is
r = 0.45
At this point the input RMS current in the capacitor is
IIN = 0.76A
Therefore, this is also the minimum required RMS current
rating of any input capacitor to be used. Now, a typical 25V
aluminum capacitor would need to be around 470-1000µF
just to be able to handle this current. It would also take up
valuable space on the board. Therefore for the example, the
choice is tantalum 22µF/35V TPS series AVX capacitor, Part
Number TPSE226K035S0200, rated for 0.812A at 85˚C.
Though it is also possible to use a Panasonic surface mount
aluminum
470µF/25V
FK
series,
Part
Number
EEVFK1E471P, rated for 0.85A at 105˚C.
11
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LM1572
Application Information
LM1572
Application Information
than tantalums for the same esr), softstart is recommended
so as to prevent startup problems.In addition, very low esr
(irrespective of whether capacitor is aluminimum, tantalum
etc.), can lead to loop instability and therefore a Bode plot is
recommended to ensure adequate phase margin.
(Continued)
Output Capacitor Selection
In voltage mode control, the esr of the output capacitor plays
an important role in the feedback loop. Therefore in such
cases, it is usually cautioned against reducing the esr too
much. But keeping the esr high enough to guarantee loop
stability has several ’side-effects’: it prevents the use of
ceramic capacitors at the output, it also keeps the dissipation
in the output capacitor ’high’ (since this is IP2*esr), and it also
keeps the output voltage ripple ’high’ (which too is proportional to esr). Note that a post LC filter therefore becomes
necessary with voltage mode controllers, if really low output
voltage ripple is required.
With current mode control, the feedback loop is different,
and so the output esr can be reduced significantly. Therefore
the main criterion for selection of the output capacitor is
based on the acceptable output voltage ripple. In the example, assuming that ± 75mV of ripple is acceptable (i.e.
150mV peak to peak), the peak to peak current is
IPP = IO • r
Sequencing
This section may be skipped if the SD pin is floating, or tied
high. It is of concern only if the Designer intends to use the
Shutdown pin in an active manner.
The following scenario explains the situation: if the input
voltage is applied and the converter has been running for
some time (SD pin high), the bootstrap capacitor is (as is
normal) charged up to about 5V. Now if the input is disconnected, and then reconnected immediately, while holding the
SD pin low, the following can happen: the output which is
expected to be zero, may go ’high’ (no regulation). It returns
to regulation only when the SD pin is taken high (over
2.38V). This mode occurs only under the above set of conditions, and only if the applied input ramp has an extremely
high slope. Then the dV/dt of the ramp injects stray charge
through the Drain-Gate capacitance of the internal Fet drivers, causing the gate voltage to go high, and may eventually
cause the switching Fet to turn on spuriously. The switch will
then stay in full conduction, till the next level shift command
comes from the SD pin. Several options exist so as to avoid
this:
1. The SD pin must not be held low during the instant that
the reapplied input voltage is ramping up across the
input of the converter.
2. Or the input dV/dt must be kept low. One way is to
increase the input capacitance (and/or esr), as mentioned earlier. Therefore, it is recommended that if the
SD pin is expected to be used actively (not floating or
high), the input capacitor should always be an aluminum
electrolytic. This will automatically lead to a larger capacitance value and esr, as desired. Further, the oscillations and overshoot at the input, described earlier,
which are also contributory factors to this spurious turnon, will also be suppressed.
3. Or the Bootstrap capacitor must be discharged. Now,
since the voltage across the bootstrap capacitor happens to be the supply for the internal driver, if this
capacitor is discharged before the input is reapplied,
there will be no problem: no supply, no drive! To implement this, it is recommended that the bootstrap capacitance used is reduced to 0.01µF and in addition, a
1M-4.7M resistor placed from the bootstrap pin to
ground. This provides a discharge path for the bootstrap
capacitor. The RC time constant is about 10-50ms, and
so a ’wait period’ of like amount is recommended before
input power is reapplied. This will allow sufficient time for
the bootstrap capacitor to discharge, and the spurious
turn-on will be prevented.
The worst case condition for this parameter is at minimum
duty cycle (max input). At this point, with the chosen inductor
r = 0.59
So peak to peak current is
IPP = 1.5 • 0.59 = 0.88A
For a maximum 150mV ripple the esr must be less than
esr = 0.150/0.88 = 0.17Ω
The RMS current capability should also be checked. The
RMS output current is
The worst case condition for this parameter is at highest
input voltage. So
Therefore a close fit is tantalum 100µF/10V TPS series AVX
capacitor, Part Number TPSY010K010S0150, esr of 0.15Ω,
rated for RMS current 0.822A at 85˚C. An alternative is
Panasonic surface mount aluminum 330µF/10V FK series,
Part Number EEVFK1A331P, esr of 0.16Ω, rated for RMS
0.6A at 105˚C. Note that the esr (and allowed output voltage
ripple) played the dominant rule in the selection here. If very
low output ripple is demanded, it would point in the direction
of larger and larger capacitances. However, it must be kept
in mind that a very large output capacitance can lead to
startup problems, because of the huge charging current (and
its duration). The choice of tantalum at the output will permit
a much lower capacitance to be used, which leads to a
smaller energy inrush (1⁄2*C*V2) and no startup problems.
Therefore when using ’low cost’ aluminum capacitors at the
output, (which always end up having a larger capacitance
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Overload Protection
The LM1572 incorporates a useful protection feature called
’frequency foldback’. When the voltage on the feedback
node starts falling to zero below a certain threshold, the IC
commands a progressive reduction in switching frequency
from 500kHz to 100kHz. The reader is referred to the relevant curve in Typical Performance Characteristics of this
datasheet. The pulse width also decreases to the minimum
width of 300ns (typical). These actions help protect not only
the IC, but also the external power components and the load.
12
remained at 500 kHz, the duty cycle would have been 15%,
and this would have led to a calculated output voltage of
(16*0.15)-0.5=2V, though we are forcing the output to zero.
This therefore represents a ’struggle’, which manifests itself
as an overstress condition. In this condition parasitics like
inductor winding resistance etc. will be called upon to control
the situation, and to stabilize the situation. For the lower
frequency case, with a duty cycle of 3%, since the calculated
output voltage is commensurate with the external condition
of a short-circuit on the output, the converter does not
’struggle’ to maintain this condition. However even with the
foldback protection as it is present on the LM1572, the
Designer is cautioned that the actual load current which can
flow with a short-circuit on the output, depends on various
factors. For example, high current Schottky diodes will be
found to lead to higher short-circuit currents than modestly
rated diodes. This is because it can be shown that if the
diode drop is lower than ’typical’ (as is the case for for high
current diodes), it requires a duty cycle even lower than 3%
to keep the calculated output voltage really close to ’zero’.
Therefore it may not be a good idea for example, to use say
a 5A/30V Schottky diode for a 1.5A application. The selected
diode in the typical application circuit is correctly sized to be
a 2A/30V Schottky from IRF.
Under startup, the frequency foldback effectively limits the
inrush current spike. The soft start feature, acting on its own,
cannot suppress the current spike at all. The role of softstart
is to gradually raise the duty cycle and thereby to bring up
the output rail slowly. But the inrush spike, which is mainly
the initial charging current of the output capacitors, occurs at
the moment of application of input power, even before the
voltage across the output capacitors has really started to rise
significantly. At this instant, soft start would call out for minimum on time, but as seen above, this is just not enough to
limit the current. However, with foldback of frequency to
100kHz, the startup duty cycle falls from 15% to 3%. This
leaves enough off time for the current to subside every cycle,
as explained above, and there is no cumulative current
buildup, or ’staircasing’.
(Continued)
It can be shown that this protection feature is vital to avoiding
overstress during overload and even under normal startup/
powerup.
Consider what happens if the output of a buck converter is at
zero volts with the maximum input voltage applied at the
input. This ’zero output volts’ condition represents the natural
initial condition at normal powerup/startup but could also be
a forced condition in the form of an output short. Then for the
LM1572, almost the full input of 16V can find iself across the
inductor during the on time of the switch. During the off time,
the voltage across the inductor reverses but the magnitude
of this voltage is only 0.5V (which comes from the ’typical’
Schottky forward drop).This leads to the problem: depite
having a ’current limit’, in fact there is absolutely no effective
current limit in this condition. Because if the switch turns on,
it has a minimum pulse width of about 300ns before it can
actually respond to any information about having exceeded
the current limit. This minimum pulse width is unavoidable
due to various internal delays, propagation intervals, and
also the internal blanking time carefully set for rejection of
transition noise, as required with current mode control.
Therefore using V=L*dI/dt is can be shown that for a switching frequency of 500kHz, and say with an inductance of
8.2µH, the current ramps up by about 0.58A during the
minimum switch on time of 0.3µs. During the off time of
1.7µs, it ramps down, but only by about 0.1A. Therefore the
current peak will incrementally increase or staircase upwards by a net 0.48A every cycle. And in a few cycles this
could blow the switch. Increasing the inductance will not
help, as it will only affect the rate of the current staircasing,
not necessarily its peak. In the absence of any other effective measure, the only way out in the current situation is to
’hope and pray’ that the the output voltage rises fast enough
before damage occurs. For a normal power up, the output
rail would rise eventually, at a rate which would be dependent on the value of the output capacitance. However for a
short on the output, it would never rise. In either case we
have a potentially destructive situation. Now it can also be
shown that if the frequency was immediately reduced to
100kHz following the ’zero output volts’ condition, the off
time is increased to 10-0.3=9.7µs. This will cause the calculated current ramp-down to be 0.59A instead of 0.1A. Since
this is greater than the current ramp-up of 0.58A the current
will actually return to zero every cycle, and there will be no
staircasing. This is how the LM1572 frequency foldback
protection works, thus avoiding this potentially dangerous
condition altogether. We consider the two possible situations
for ’zero output volts condition’ in more detail below, to
understand it better.
By definition, an ’overload’ is where the switch current limit
has been reached, and then any attempt to increase the load
further, causes the output voltage rail to ’droop’, though the
load current remains virtually constant during this time. If an
attempt is made to increase the load even further, the voltage on the feedback pin will fall low enough to cause the
LM1572 to start lowering its switching frequency. At the
same time the output of the error amplifier clamps high (at
2V), and this causes the LM1572 to suddenly reduce the
on-time to the minimum pulse width. The foldback frequency
is now 100 kHz, and with this minimum pulse width, the
effective duty cycle is 3%. It can be easily shown by calculation that at the highest input voltage (worst case), assuming a typical Schottky catch diode drop, a 3% duty cycle
produces a very low (almost zero) output voltage (ignoring
switch voltage drop here). However if the frequency had
Layout Guidelines
Refer to the sample PCB layout provided. The Bill of Material
is also provided. The board is based on the schematic in
’Typical Applications’ for the fixed voltage part. The design is
based on the worked example presented in this datasheet.
The input voltage can vary between 8.5V to 16V. The output
rail is 5V and the peak current is 1.5A. The inductor is
however sized to handle only 1A continuous current. If
higher continuous rating is required (this depends on ambient temperature range too), an appropriately rated inductor,
possibly a higher series from the same vendor (keeping
inductance unchanged) can be selected.
Considering the critical aspects of the layout, it is recommended that the routing and positioning of the 0.1µF input
decoupling capactor, C2, be kept the same as shown, and
also the catch diode D1. The rest are not critical, and may be
changed. Note however that the trace to the feedback pin is
routed through the quiet ground plane on the bottom side.
This helps prevent noise pickup and maintain correct output
voltage. Note that vias are provided, for example directly
below the IC to the ground plane, and this helps not only in
transferring heat to the other side of the board, but references the IC ground directly to the ground plane.
13
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LM1572
Application Information
LM1572
Application Information
resistor’ between feedback pin and ground. Its value would
be 2.42V/0.5µA = 4.84MΩ . In fact this would have been
acceptable had the current been a constant. But as seen
from the Electrical Characteristics, this current can be as
high as 1.5µA. This would mean that this internal resistor can
actually be 3 times lower i.e. 4.84/3 = 1.6MΩ. Therefore, it is
not the feedback pin current itself, but its variability that
poses the problem: a single fixed known value of feedback
pin current can be easily modeled, not a spread of values.
(Continued)
Loop Compensation
Since the LM1572 uses current mode control, the loop response does not involve the inductor. The error amplifier can
be modeled as a transconductance amplifier with a large
output impedance of 200k of resistance in parallel with 12pF
of output capacitance. In practical applications, the impedance of the external compensation network from the Comp
pin to ground dominates completely, and the error amplifier
characteristics do not contribute any significant phase shift
to the loop. Therefore the error amplifier can for all practical
purposes be considered simply as a 2000µMhos of
transconductance, the loop phase/gain being determined
externally.
The most direct approach to the problem is as follows: to
continue to use the basic (’ideal’) resistive divider equation
as the basic design equation, but to also use the equation
which ’models in’ a 1.6MΩ resistor as representing the worst
case error, then to compare these two equations to calculate
what the error is. Conclusions on how to reduce this error
would follow.
The simplest recommended compensation is a 3.3nF capacitor from comp pin to ground. This provides a pole at
240Hz. The overall loop then has a low frequency gain of
about 62dB at 1.5A, with a crossover at about 15kHz, and a
phase margin of about 33˚. A resistor may be added in series
with this capacitor to improve the loop phase
margin/crossover frequency. Recommended values for this
are 1k to 1.5k. If the loop response needs to be further
improved, by increasing the value of this resistor, then a
small capacitor of about 470pF is required across the RC. Its
purpose is to limit the ripple on the Comp pin to within
100mVPP, which can otherwise cause problems with the
behavior of the LM1572.
So proceeding in this manner, the worst-case resistive divider equation (modified to include 1.6MΩ resistor in parallel
with R1) is
Comparing this with the basic equation provides the following error equation
Loop compensation must be further validated by a bench
measurement, using standard Bode plot/spectrum analyzer
equipment. Step load transient response can also be tested
and should not reveal excessive ringing on the output of the
converter.
Note that the right hand side depends on R1 (not R2, nor the
output voltage directly). So for a given maximum allowed
error, first R1 is calculated. Then the basic resistive divider
equation is invoked and used to calculate R2.
The effect of R1 is now considered.
If R1 was say 5k, the error is about 0.3%. This error is in the
’+’ direction as mentioned earlier, for the case of current
flowing into the pin. The error is reduced to 0.14% for the
value R1 = 2.21k as used in the Typical Application circuit. In
general it can be concluded that to restrict the error to within
0.25%, R1should be 4k (or less).
If R1 is 4.02k (a standard value), R2 as calculated from the
basic resistive divider design equation is 4.286k for a 5V
output and is 1.462k for a 3.3V output.
There is an alternative way of stating the error, in terms of
current rather than resistance. Since if R1=4k, the current
flowing through the resistive divider is 2.42/4k=0.6mA, therefore it can also be stated that the divider current should be
0.6mA or less. This will restrict the error (due to the feedback
pin current and its variation) to less than 0.25%. This ’thumbrule’ does not depend on input or output conditions, and is
typical for most applications.
The other related problem is that to get an exact value for R1
or R2 from standard resistor values may not be easy. Any
one of the two resistors can of course be selected to be a
standard value, but the other value as calculated from the
equation, will more likely than not, not correspond to any
standard value. The so called ’EIA standard values’ are the
E6, E12, E24, E48, E96 and E192 series, listed in most
resistor catalogs. E12 for example has 12 values in every
decade of resistance. The reader can for example do a
search within a typical vendor’s index home page e.g. http://
www.vishay.com/ with the keyword E96 for the table of standard values.
Tolerance of set Output Voltage
This section may be skipped altogether, unless the designer
wants to get a more precise understanding of the possible
variation or ’spread’ on the output voltage and how this can
be controlled better.
This ’basic resistive divider design equation’ seems to suggest that R2 is always a certain fixed ’ratio’ to R1, for a given
output voltage. For example, referring to the Typical Application circuit, where the values shown are R2=806 and R1=
2.21k, it may have been thought that using the following
values: R2=8.06k and R1= 22.1k, would have been equally
acceptable. But the simple equation is just that: an ’ideal’
equation that unrealistically assumes zero current into or out
of the feedback pin. It can be easily shown that the effect of
any ’real’ current, flowing into the feedback pin for example,
is to raise the output voltage slightly from the ’ideal’ calculation. This is considered to be an output voltage ’error’, and
this needs to be understood and quantified.
Now, as mentioned, had the selection been: R2=8.06k and
R1= 22.1k, (possibly with the intention of reducing the dissipation in the resistive divider by a factor of 10), it would also
have increased the error in the output voltage by almost the
same factor. A compromise can always be considered if
efficiency at light loads is a key concern, but first it must be
understood how to actually design the resistive divider for a
certain (maximum) error.
As can be seen from the Typical Performance curves and
tables of Electrical Characteristics for the LM1572, a current
of about 0.5µA (typical value) flows into the feedback pin at
regulation (IFB_REG). Since VFB = 2.42V, it may have been
thought appropriate to ’modify’ the basic resistive divider
equation by simply modeling this current in by an ’internal
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14
Characteristics tables). Another significant source of error to
consider is the tolerance of the resistors themselves. For a
majority of applications these are generally chosen to be of
1% tolerance. But they can be chosen to be 0.5% or 0.1% in
critical applications, or even 2% in less sensitive applications.
The percentage error in the output voltage is 2*(VO-VFB)*Tol/
VO. So for example with 1% resistors being used to set the
output to 5V, the maximum error on the output on account of
the resistor tolerance is 2*(5-2.42)*1/5=1%. (This is actually
+,-1% since the tolerance of the resistors to start with was
also +,-1%). For a 3.3V output, the error is about +,-0.5%
with 1% resistors. Similarly, the output error (on this account)
is reduced by a factor of 10, if 0.1% resistors are used
instead.
As for the shunt resistor of 560k (in parallel to R2), it is not
really necessary to have a very tight tolerance for this resistor. Since the entire effect of this resistor is to add a slight
’trim’ to the output voltage, the effect of its tolerance on the
output is proportionally small too. The proportionality factor
here is the ratio R2/560k. Therefore, the 560k can be a 5%
resistor in almost all applications. This was actually kept in
mind well in advance, when the value 560k was initially
proposed. Because 560k happens to be a standard value in
the 5% series (E24), though it does not exist in the higher
(expensive) series. Cost was clearly of concern here.
Concluding this discussion, there is a last observation to
make concerning the fixed voltage part. Here the output is
normally connected to the feedback pin directly. The resistive divider is therefore internal. The relevant design information here is that for the 5V part the effective resistance
(R1 + R2) from feedback pin to ground is 10k. For the 3.3V
part this resistance is 6.6k. This gives a current of 0.5mA
passing through the divider. This is a satisfactory choice,
since it was seen to limit the contribution of the feedback
node current on the output to less than 0.3%. The error due
to the ’tolerance’ of the resistive divider is almost negligible
for fixed voltage parts. To understand this, the reason for the
error from an external divider, as used in an adjustable part,
must first be clarified. There the worst case situation is
where one resistor is at the lower end of its tolerance band,
while at the same time the other resistor is at the upper end
of the tolerance band. This gives the worst case error on the
output. However if both the upper and lower resistors were
simultaneously say ’x%’ higher (or lower) than their nominal,
their ’ratio’ would remain unchanged. It was earlier indicated
that if the ratio of the resistances in a resistive divider is
maintained, then theoretically there is no change in the
output voltage. This is the situation in the case of the internal
resistive divider. Because the two resistors are in the same
package, any drift or tolerance will affect both of them almost
equally. It is expected, and borne out, that their ’relative
tolerance’ is typically almost 10 times better than the (absolute) tolerance of each. Therefore, the effect of the tolerance
of the resistors in an internal resistive divider, as in fixed
voltage parts, can be ignored.
(Continued)
The available tolerances should also be checked out as they
govern what can be considered a ’standard’ value for a
certain requirement. Therefore for example, if 1% resistors
are required, it will be almost impossible to find such a
resistor in the E12 or E24 series, for which 10% and 5%
respectively are more commonly available tolerances. E48 is
normally 2%, E96 is 1%, and E192 is 0.5%/ 0.1%. Further,
each series is usually ’devoted’ to its tolerance.Therefore a
value like 221Ω which exists in E96 and E192 may not be
readily available as a 2% resistor, simply because in E48
(which is usually for 2% resistors), the nearest standard
values are 215Ω and 226Ω. One could always pay more for
better resistors than required, but the question is one of
’optimum’ design here. ’Optimum’ means a correct
consideration/compromise between several factors: cost,
tolerance of the resistors, and the output error (from all
related sources). All these indicate that the task of correctly
selecting a resistive divider is usually under-estimated or
down-played.
Looking at the standard value series, it should also be noted
that every series is a subset of the next higher series.
However there are two distinct sets. Therefore E6 is a subset
of E12 and E24, and E48 is a subset of E96 and E192.
However no E24 value can be found in E48 (or higher). As
an example, the ’well-known’ resistance value of 220Ω is
available in E6, E12 and E24. But this value does not exist in
the higher (more modern) series.There the closest ’standard’
values are 215Ω and 226Ω in E48, and also 221Ω in the
more expensive series as discussed above. The following
example will make these considerations clearer.
Example: It is required to set an output voltage of 5V using
the adjustable part. R1 is taken to be a 4.02k (E48 series)
standard resistor value (for a 0.25% error expected on account of the feedback pin current). So using the basic divider
equation
This is, as expected, not a standard resistance value. The
closest is 4.32k (E96). Using that value for R2 would give an
additional error of (4.32−4.286)/4.286 = 0.8% (on top of the
errors due to other causes). If this is unacceptable, an
additional resistor can be placed in parallel with the 4.32k.
Checking to see if a ’standard’ 560k resistor would do the
job: this gives an effective value of (4.32*560)/(4.32 + 560) =
4.287k, which is almost exactly the required value of 4.286k!
More on the choice of the 560k will follow below.
Final recommended values here are
R1 = 4.02k, and R2 = 4.32k \ 560k
Note that the effect of other tolerances have not been considered, including the possible spread on VFB itself (this
adds another +,- 1.25% of error as indicated by the Electrical
Bill of Material for LM1572 Evaluation Board
Designator
Description
U1
LM1572-5.0
D1
2A/30V Schottky
D2
SS diode
L1
8.2µH
C1
22µF/35V
Manufacturer
National Semiconductor
Part Number
Qty.
LM1572-5.0
1
International Rectifier
20BQ030
1
General Semiconductor
1N4448W
1
SMP3013-821K*
1
595D226X0035R2T
1
Gowanda
Vishay-Sprague
15
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LM1572
Application Information
LM1572
Application Information
(Continued)
Bill of Material for LM1572 Evaluation Board
Manufacturer
(Continued)
Designator
Description
Part Number
Qty.
C2, C3, C5**
0.1µF/50V
Vishay-Vitramon
VJ0805Y104KXAAR
3
C4
3.3nF/50V
Vishay-Vitramon
VJ0805Y332KXAAR
1
C6
100µF/10V
Vishay-Sprague
594D107X0010C2T
1
*Rated continuous 1A load. Use SMP5025-821K (or equivalent) for higher continuous load rating. **Increase C5 to 0.18µF for regulation below 0.5mA external load.
PCB Layout
20033341
Top Layer
20033342
Bottom Layer
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16
LM1572 1.5A, 500kHz Step-down Voltage Regulator
Physical Dimensions
inches (millimeters) unless otherwise noted
TSSOP-16 Pin Package
NS Package Number MTC16
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2. A critical component is any component of a life
support device or system whose failure to perform
can be reasonably expected to cause the failure of
the life support device or system, or to affect its
safety or effectiveness.
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Asia Pacific Customer
Response Group
Tel: 65-2544466
Fax: 65-2504466
Email: [email protected]
National Semiconductor
Japan Ltd.
Tel: 81-3-5639-7560
Fax: 81-3-5639-7507
National does not assume any responsibility for use of any circuitry described, no circuit patent licenses are implied and National reserves the right at any time without notice to change said circuitry and specifications.