IL33153

TECHNICAL DATA
IL33153
Single IGBT Gate Driver
The IL33153 is specifically designed as an IGBT driver for high power
applications that include ac induction motor control, brushless dc motor
control and uninterruptable power supplies. Although designed for driving discrete and module IGBTs, this device offers a cost effective solution for driving power MOSFETs and Bipolar Transistors. Device protection features include the choice of desaturation or overcurrent sensing
and undervoltage detection. These devices are available in dual-inline
and surface mount packages and include the following features:
SOP-8
DIP-8
FEATURES
• High Current Output Stage: 1.0 A Source/2.0 A Sink
• Protection Circuits for Both Conventional and Sense IGBTs
• Programmable Fault Blanking Time
• Protection against Overcurrent and Short Circuit
• Undervoltage Lockout Optimized for IGBT's
• Negative Gate Drive Capability
• Cost Effectively Drives Power MOSFETs and Bipolar Transistors
ORDERING INFORMATION
Device
IL33153N
IL33153DT
IL33153D
Operating Temperature Range
TA = -40°C ~+105°C
for all packages
Package
DIP-8
SOP-8
SOP-8
Shipping
Tube
Tape& Reel
Tube
Block Diagram
2011, February, Rev. 04
IL33153
Absolute Maximum Ratings
Rating
Power Supply Voltage
VCC to VEE
Kelvin Ground to VEE (Note 1 )
Logic Input
Current Sense Input
Blanking/Desaturation Input
Gate Drive Output
Source Current
Sink Current
Diode Clamp Current
Fault Output
Source Current
Sink Curent
Power Dissipation and Thermal Characteristics
D Suffix SO-8 Package, Case 751
Maximum Power Dissipation @ TA = 5OC Thermal Resistance, Junction-to-Air
P Suffix DIP-8 Package, Case 626
Maximum Power Dissipation @ TA = 5OC Thermal Resistance, Junction-to-Air
Operating Junction Temperature
Operating Ambient Temperature
Storage Temperature Range
Symbol
VCC-VEE
KGnd - VEE
Vin
VS
VBD
Value
20
20
VEE-0.3 to VCC
-0.3 to Vcc
-0.3 to Vcc
IO
Unit
V
V
V
V
A
1.0
2.0
1.0
IFO
mA
25
10
PD
RJA
0.56
180
W
C/W
PD
RJA
1.0
100
W
C/W
TJ
TA
Tstg
+150
-40 to +105
-65 to +150
C
C
C
* Stresses beyond those listed under “absolute maximum ratings” may cause permanent damage to the device.
These are stress ratings only and functional operation of the device at these or any other conditions beyond those indicated under “recommended operating conditions” is not implied.
Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
2011, February, Rev. 04
IL33153
ELECTRICAL CHARACTERISTICS
(Vcc=15V, VEE=0V, Kelvin Gnd connected to VEE. For typical values TA=25C, for min/max values TA is the operating ambient
temperature range that applies (Note 2), unless otherwise noted.)
Characteristic
Symbol
Min
Typ
Max
Unit
LOGIC INPUT
Input Threshold Voltage ]
High State (Logic 1 )
Low State (Logic 0)
Input Current
High State (VIH = 3.0 V)
Low State (Vii. = 1.2 V)
V
VIH
VIL
1.2
2.70
2.30
3.2
-
-
130
50
500
100
12
-
2.0
13.9
100
2.5
200
12
0.2
13.3
1.0
-
A
IIH
IIL
DRIVE OUTPUT
Output Voltage
Low State (Isink = 1.0 A)
High State (Isource = 500 mA)
Output Pull-Down Resistor
V
VOL
VOH
RPD
k
FAULT OUTPUT
Output voltage
Low Slate (Isink = 5.0 mA)
High State (Isource = 20 mA)
V
VFL
VFH
SWITCHING CHARACTERISTICS
Propagation Delay (50% Input to 50% Output CL = 1.0 nF)
Logic Input to Drive Output Rise
Logic Input to Drive Output Fall
tPLH(in/out) tPHL (in/out)
Drive Output Rise Time (10% to 90%) CL = 1.0 nF
tr
Drive Output Fall Time (90% to 10%) CL= 1.0 nF
tf
Propagation Delay
Current Sense Input to Drive Output
tp(OC)
Fault Blanking/Desaturation Input to Drive Output
tp(FLT)
-
ns
80
120
17
17
300
300
55
55
0.3
0.3
1.0
1.0
11.3
10.4
12
11
12.6
11.7
V
V
50
100
6.0
-
65
130
6.5
-1.4
80
160
7.0
-10
mV
mV
V
uA
-270
2.5
-300
-
uA
mA
7.2
7.9
14
20
ns
ns
s
UVLO
Startup Voltage
Disable Voltage
VSS Start
VSS dis
COMPARATORS
Overcurrent Threshold Voltage (Vpin8 > 7,0 V)
VSOC
Short Circuit Threshold Voltage (V pine8> 7,0 V)
VSSC
Fault Blanking/Desaturation Threshold (Vpin1 > 100 mV)
Vth(FLT)
Current Sense Input Current (Vsi = 0 V)
ISI
FAULT BLANKING/DESATURATION INPUT
Current Source (Vpjn8 = 0 V, Vpin4 = 0 V)
Ichg
-200
Discharge Current (Vpin8 = 15 V, Vpin4 = 5.0 V)
Idschg
1.0
Power Supply Current
Standby (Vpin 4 = VCC, Output Open)
Operating (CL= 1.0 nF, f= 20 kHz)
TOTAL DEVICE
ICC
mA
-
NOTES: 1. Kelvin Ground must always be between VEE and VCC.
2.Low duty cycle pulse techniques are used during test to maintain the junction temperature as close to ambient as possible.
Tlow = -40°C for IL33153
Thigh = +105C for IL33153
2011, February, Rev. 04
IL33153
Typical Characteristics
Figure 2. Input Current versus Input Voltage
Figure 3. Output Voltage versus Input Voltage
Figure 4. Input Threshold Voltage
versus Temperature
Figure 5. Input Threshold Voltage
versus Supply Voltage
Figure 6. Drive Output Low State Voltage
versus Temperature
Figure 7. Drive Output Low State Voltage
versus Sink Current
2011, February, Rev. 04
IL33153
Figure 8. Drive Output High State Voltage
versus Temperature
Figure 9. Drive Output High State Voltage
versus Source Current
Figure 10. Drive Output Voltage
versus Current Sense Input Voltage
Figure 11. Fault Output Voltage
versus Current Sense Input Voltage
Figure 12. Overcurrent Protection Threshold
Voltage versus Temperature
Figure 13. Overcurrent Protection Threshold
Voltage versus Supply Voltage
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IL33153
Figure 14. Short Circuit Comparator Threshold
Voltage versus Temperature
Figure 15. Short Circuit Comparator Threshold
Voltage versus Supply Voltage
Figure 16. Current Sense Input Current
versus Voltage
Figure 17. Drive Output Voltage versus Fault
Blanking/Desaturation Input Voltage
Figure 18. Fault Blanking/Desaturation Comparator
Threshold Voltage versus Temperature
Figure 19. Fault Blanking/Desaturation Comparator
Threshold Voltage versus Supply Voltage
2011, February, Rev. 04
IL33153
Figure 20. Fault Blanking/Desaturation Current
Source versus Temperature
Figure 21. Fault Blanking/Desaturation Current
Source versus Supply Voltage
Figure 22. Fault Blanking/Desaturation
Current Source versus Input Voltage
Figure 23. Fault Blanking/Desaturation Discharge
Current versus Input Voltage
Figure 24. Fault Output Low State Voltage
versus Sink Current
Figure 25. Fault Output High State Voltage
versus Source Current
2011, February, Rev. 04
IL33153
Figure 26. Drive Output Voltage
versus Supply Voltage
Figure 27. UVLO Thresholds
versus Temperature
Figure 28. Supply Current
versus Supply Voltage
Figure 29. Supply Current
versus Temperature
Figure 30. Supply Current versus Input Frequency
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IL33153
OPERATING DESCRIPTION
GATE DRIVE
Controlling Switching Times
The most important design aspect of an IGBT gate
drive is optimization of the switching characteristics.
The switching characteristics are especially important in
motor control applications in which PWM transistors are
used in a bridge configuration. In these applications, the
gate drive circuit components should be selected to optimize turn−on, turn−off and off−state impedance. A
single resistor may be used to control both turn−on and
turn−off as shown in Figure 31. However, the resistor
value selected must be a compromise in turn−on abruptness and turn−off losses. Using a single resistor is normally suitable only for very low frequency PWM. An
optimized gate drive output stage is shown in Figure 32.
This circuit allows turn−on and turn−off to be optimized
separately. The turn−on resistor, Ron, provides control
over the IGBT turn−on speed. In motor control circuits,
the resistor sets the turn−on di/dt that controls how fast
the free−wheel diode is cleared. The interaction of the
IGBT and free−wheeling diode determines the turn−on
dv/dt. Excessive turn−on dv/dt is a common problem in
half−bridge circuits. The turn−off resistor, Roff, controls
the turn−off speed and ensures that the IGBT remains
off under commutation stresses. Turn−off is critical to
obtain low switching losses. While IGBTs exhibit a
fixed minimum loss due to minority carrier recombination, a slow gate drive will dominate the turn−off losses.
This is particularly true for fast IGBTs. It is also possible to turn−off an IGBT too fast. Excessive turn−off
speed will result in large overshoot voltages. Normally,
the turn−off resistor is a small fraction of the turn−on
resistor.
The IL33153 contains a bipolar totem pole output
stage that is capable of sourcing 1.0 amp and sinking 2.0
amps peak. This output also contains a pull down resistor to ensure that the IGBT is off whenever there is insufficient VCC to the IL33153.
In a PWM inverter, IGBTs are used in a half−bridge
configuration. Thus, at least one device is always off.
Whilethe IGBT is in the off−state, it will be subjected to
changes in voltage caused by the other devices. This is
particularly a problem when the opposite transistor turns
on.
When the lower device is turned on, clearing the upper diode, the turn−on dv/dt of the lower device appears
across the collector emitter of the upper device. To eliminate shoot−through currents, it is necessary to provide a
low sink impedance to the device that is in the off−state.
In most applications the turn−off resistor can be made
small enough to hold off the device that is under commutation without causing excessively fast turn−off
speeds.
Figure 31. Using a Single Gate Resistor
Figure 32. Using Separate Resistors
for Turn−On and Turn−Off
A negative bias voltage can be used to drive the
IGBT into the off−state. This is a practice carried over
from bipolar Darlington drives and is generally not required for IGBTs. However, a negative bias will reduce
the possibility of shoot−through. The IL33153 has separate pins for VEE and Kelvin Ground. This permits operation using a +15/−5.0 V supply.
INTERFACING WITH OPTOISOLATORS
Isolated Input
The IL33153 may be used with an optically isolated
input. The optoisolator can be used to provide level
shifting, and if desired, isolation from ac line voltages.
An optoisolator with a very high dv/dt capability should
be used, such as the Hewlett Packard HCPL4053. The
IGBT gate turn−on resistor should be set large enough
to ensure that the opto’s dv/dt capability is not exceeded.
Like most optoisolators, the HCPL4053 has an active
low open−collector output. Thus, when the LED is on,
the output will be low. The IL33153 has an inverting
input pin to interface directly with an optoisolator using
a pullup resistor. The input may also be interfaced directly to 5.0 V CMOS logic or a microcontroller.
2011, February, Rev. 04
IL33153
Optoisolator Output Fault
The IL33153 has an active high fault output. The
fault output may be easily interfaced to an optoisolator.
While it is important that all faults are properly reported,
it is equally important that no false signals are propagated. Again, a high dv/dt optoisolator should be used.
The LED drive provides a resistor programmable
current of 10 to 20 mA when on, and provides a low
impedance path when off. An active high output, resistor,
and small signal diode provide an excellent LED driver.
This circuit is shown in Figure 33.
Figure 33. Output Fault Optoisolator
UNDERVOLTAGE LOCKOUT
It is desirable to protect an IGBT from insufficient
gate voltage. IGBTs require 15 V on the gate to achieve
the rated on−voltage. At gate voltages below 13 V, the
on−voltage increases dramatically, especially at higher
currents. At very low gate voltages, below 10 V, the
IGBT may operate in the linear region and quickly
overheat. Many PWM motor drives use a bootstrap supply for the upper gate drive. The UVLO provides protection for the IGBT in case the bootstrap capacitor discharges.
The IL33153 will typically start up at about 12 V.
The UVLO circuit has about 1.0 V of hysteresis and will
disable the output if the supply voltage falls below about
11 V.
The output characteristics of an IGBT are similar to
a Bipolar device. However, the output current is a function of gate voltage instead of current. The maximum
current depends on the gate voltage and the device type.
IGBTs tend to have a very high transconductance and a
much higher current density under a short circuit than a
bipolar device. Motor control IGBTs are designed for a
lower current density under shorted conditions and a
longer short circuit survival time.
The best method for detecting desaturation is the use
of a high voltage clamp diode and a comparator. The
IL33153 has a Fault Blanking/Desaturation Comparator
which senses the collector voltage and provides an output indicating when the device is not fully saturated.
Diode D1 is an external high voltage diode with a rated
voltage comparable to the power device. When the
IGBT is “on” and saturated, D1 will pull down the voltage on the Fault Blanking/Desaturation Input. When the
IGBT pulls out of saturation or is “off”, the current
source will pull up the input and trip the comparator.
The comparator threshold is 6.5 V, allowing a maximum
on−voltage of about 5.8 V.
A fault exists when the gate input is high and VCE is
greater than the maximum allowable VCE(sat). The output of the Desaturation Comparator is ANDed with the
gate input signal and fed into the Short Circuit and
Overcurrent Latches. The Overcurrent Latch will
turn−off the IGBT for the remainder of the cycle when a
fault is detected. When input goes high, both latches are
reset. The reference voltage is tied to the Kelvin Ground
instead of the VEE to make the threshold independent of
negative gate bias. Note that for proper operation of the
Desaturation Comparator and the Fault Output, the Current Sense Input must be biased above the Overcurrent
and Short Circuit Comparator thresholds. This can be
accomplished by connecting Pin 1 to VCC.
PROTECTION CIRCUITRY
Desaturation Protection
Bipolar Power circuits have commonly used what is
known as “Desaturation Detection”. This involves monitoring the collector voltage and turning off the device if
this voltage rises above a certain limit. A bipolar transistor will only conduct a certain amount of current for a
given base drive. When the base is overdriven, the device is in saturation. When the collector current rises
above the knee, the device pulls out of saturation. The
maximum current the device will conduct in the linear
region is a function of the base current and the dc current gain (hFE) of the transistor.
Figure 34. Desaturation Detection
The IL33153 also features a programmable fault
blanking time. During turn−on, the IGBT must clear the
opposing free−wheeling diode. The collector voltage
will remain high until the diode is cleared. Once the
diode hasbeen cleared, the voltage will come down
quickly to the VCE(sat) of the device. Following turn−on,
there is normally considerable ringing on the collector
due to the COSS capacitance of the IGBTs and the para-
2011, February, Rev. 04
IL33153
sitic wiring inductance. The fault signal from the Desaturation Comparator must be blanked sufficiently to allow the diode to be cleared and the ringing to settle out.
The blanking function uses an NPN transistor to clamp
the comparator input when the gate input is low. When
the input is switched high, the clamp transistor will turn
“off”, allowing the internal current source to charge the
blanking capacitor. The time required for the blanking
capacitor to charge up from the on−voltage of the internal NPN transistor to the trip voltage of the comparator
is the blanking time.
If a short circuit occurs after the IGBT is turned on
and saturated, the delay time will be the time required
for the current source to charge up the blanking capacitor from the VCE(sat) level of the IGBT to the trip voltage of the comparator. Fault blanking can be disabled by
leaving Pin 8 unconnected.
Sense IGBT Protection
Another approach to protecting the IGBTs is to
sense the emitter current using a current shunt or Sense
IGBTs. This method has the advantage of being able to
use high gain IGBTs which do not have any inherent
short circuit capability. Current sense IGBTs work as
well as current sense MOSFETs in most circumstances.
However, the basic problem of working with very low
sense voltages still exists. Sense IGBTs sense current
through the channel and are therefore linear with respect to the collector current. Because IGBTs have a
very low incremental on−resistance, sense IGBTs behave much like low−on resistance current sense
MOSFETs. The output voltage of a properly terminated
sense IGBT is very low, normally less than 100 mV.
The sense IGBT approach requires fault blanking to
prevent false tripping during turn−on. The sense IGBT
also requires that the sense signal is ignored while the
gate is low. This is because the mirror output normally
produces large transient voltages during both turn−on
and turn−off due to the collector to mirror capacitance.
With non−sensing types of IGBTs, a low resistance current shunt (5.0 to 50 mΩ) can be used to sense the emitter current. When the output is an actual short circuit,
the inductance will be very low. Since the blanking circuit provides a fixed minimum on−time, the peak current under a short circuit can be very high. A short circuit discern function is implemented by the second
comparator which has a higher trip voltage. The short
circuit signal is latched and appears at the Fault Output.
When a short circuit is detected, the IGBT should be
turned−off for several milliseconds allowing it to cool
down before it is turned back on. The sense circuit is
very similar to the desaturation circuit. It is possible to
build a combination circuit that provides protection for
both Short Circuit capable IGBTs and Sense IGBTs.
2011, February, Rev. 04
TECHNICAL DATA
APPLICATION INFORMATION
Figure 35 shows a basic IGBT driver application.
When driven from an optoisolator, an input pull up resistor is required. This resistor value should be set to
bias the output transistor at the desired current. A decoupling capacitor should be placed close to the IC to
minimize switching noise.
A bootstrap diode may be used for a floating supply.
If the protection features are not required, then both the
Fault Blanking/Desaturation and Current Sense Inputs
should both be connected to the Kelvin Ground (Pin 2).
When used with a single supply, the Kelvin Ground and
VEE pins should be connected together. Separate gate
resistors are recommended to optimize the turn−on and
turn−off drive.
Kelvin Ground. The Current Sense Input should be tied
high because the two comparator outputs are ANDed
together. Although the reverse voltage on collector of
the IGBT is clamped to the emitter by the free−wheeling
diode, there is normally considerable inductance within
the package itself. A small resistor in series with the
diode can be used to protect the IC from reverse voltage
transients.
Figure 37. Desaturation Application
Figure 35. Basic Application
When using sense IGBTs or a sense resistor, the
sense voltage is applied to the Current Sense Input. The
sense trip voltages are referenced to the Kelvin Ground
pin. The sense voltage is very small, typically about 65
mV, and sensitive to noise. Therefore, the sense and
ground return conductors should be routed as a differential pair. An RC filter is useful in filtering any high frequency noise. A blanking capacitor is connected from
the blanking pin to VEE. The stray capacitance on the
blanking pin provides a very small level of blanking if
left open. The blanking pin should not be grounded
when using current sensing, that would disable the sense.
The blanking pin should never be tied high, that would
short out the clamp transistor.
Figure 36. Dual Supply Application
When used in a dual supply application as in Figure
36, the Kelvin Ground should be connected to the emitter of the IGBT. If the protection features are not used,
then both the Fault Blanking/Desaturation and the Current Sense Inputs should be connected to Ground. The
input optoisolator should always be referenced to VEE.
If desaturation protection is desired, a high voltage
diode is connected to the Fault Blanking/Desaturation
pin. The blanking capacitor should be connected from
the Desaturation pin to the VEE pin. If a dual supply is
used, the blanking capacitor should be connected to the
Figure 38. Sense IGBT Application
2011, February, Rev. 04
TECHNICAL DATA
Package Dimension
N SUFFIX PLASTIC DIP
(MS – 001BA)
A
Dimension, mm
5
8
B
1
4
F
Symbol
MIN
MAX
A
8.51
10.16
B
6.1
7.11
C
L
C
5.33
D
0.36
0.56
F
1.14
1.78
-T- SEATING
PLANE
N
G
K
D
0.25 (0.010) M
M
H
J
T
NOTES:
1. Dimensions “A”, “B” do not include mold flash or protrusions.
Maximum mold flash or protrusions 0.25 mm (0.010) per side.
G
2.54
H
7.62
J
0°
10°
K
2.92
3.81
L
7.62
8.26
M
0.2
0.36
N
0.38
2011, February, Rev. 04
0.25 (0.010) M
T
NOTES:
1. Dimensions “A”, “B” do not include mold flash or protrusions.
Maximum mold flash or protrusions 0.25 mm (0.010) per side.
K
2.92
3.81
L
7.62
8.26
M
0.2
N
0.38
0.36
IL33153
D SUFFIX SOIC
(MS - 012AA)
Dimension, mm
A
8
5
B
H
1
G
P
4
D
K
MIN
MAX
A
4.8
5
B
3.8
4
C
1.35
1.75
D
0.33
0.51
F
0.4
1.27
R x 45
C
-T-
Symbol
SEATING
PLANE
J
F
0.25 (0.010) M T C M
NOTES:
1. Dimensions A and B do not include mold flash or protrusion.
2. Maximum mold flash or protrusion 0.15 mm (0.006) per side
for A; for B ‑ 0.25 mm (0.010) per side.
M
G
1.27
H
5.72
J
0°
8°
K
0.1
0.25
M
0.19
0.25
P
5.8
6.2
R
0.25
0.5
2011, February, Rev. 04