Application Note 9016
February, 2001
IGBT Basics 1
by K.S. Oh
CONTENTS
1. Introduction........................................................................................................
2. Device structure and operation ........................................................................
2-1. Structure.....................................................................................................
2-2. Operation ...................................................................................................
3. Basic Characteristics.........................................................................................
3-1. Advantages, Disadvantages and Characteristics Comparison with
BJT and MOSFET .....................................................................................
3-2. Latch-up .....................................................................................................
3-3. Static Blocking Characteristics ..................................................................
3-4. Leakage Current .......................................................................................
3-5. Forward Conduction Characteristics .........................................................
3-6. Switching Characteristics ..........................................................................
3-7 SOA (Safe-Operating-Area) .......................................................................
4. Explanation of Data Sheet Parameters ............................................................
4-1. Absolute Maximum Ratings ......................................................................
4-2. Thermal Characteristics ............................................................................
4-3. Electrical Characteristics of IGBT .............................................................
4-4. Electrical Characteristics of DIODE ..........................................................
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Rev. A2, February 2001
1. Introduction
Prior to the development of the IGBTs (insulated gate bipolar transistor), power MOSFETs
were used in medium or low voltage applications which require fast switching. Whereas bipolar power transistors, thyristors and GTOs were used in medium to high voltage applications
which require high current conduction. A power MOSFET allows for simple gate control circuit
design and has excellent fast switching capability. On the other hand, at 200V or higher, it has
the disadvantage of rapidly increasing on-resistance as the breakdown voltage increases. The
bipolar power transistor has excellent on-state characteristics due to the low forward voltage
drop, but its base control circuit is complex, and fast switching operation is difficult as compared with the MOSFET. The IGBT developed in the early 1980s has the combined advantages of the above two devices. It has a MOS gate input structure, which has a simple gate
control circuit design and is capable of fast switching up to 100kHz. Additionally, because the
IGBT output has a bipolar transistor structure, its current conduction capability is superior to a
bipolar power transistor. Based upon these excellent characteristics, the IGBT has been
extensively used in applications exceeding 300V voltage as an alternative to power MOSFETs
and bipolar power transistors. Its area of application continues to increase. The IGBT is
becoming more modular as its use increases in applications that require higher current conduction capability.
History of the Fairchild IGBT
Fairchild Semiconductor began developing the IGBT in 1992. This was later than its competitors for a power semiconductor company. However, Fairchild Semiconductor was able to catch
up with its leading competitors with the development of the third generation 600V IGBT and
1500V ultra-fast IGBT for 220V power IH applications in 1995. In 1996, Fairchild Semiconductor developed the 600V rugged type RUF series with its own stripe pattern. This has strengthened short circuit withstanding capability, which makes it suitable for motor control
applications such as inverters. Following this development, Fairchild Semiconductor used
trench technology in 1998 to develop the 400V IGBT for camera strobes and the 900V IGBT
for 110V power IH applications. Both of these require low-loss high current conduction capabilities. These design achievements indicated that Fairchild Semiconductor now possessed both
planar and trench technologies. In particular, Fairchild has achieved world-class quality in
1998 by developing 1200V IGBT using SDB (silicon direct bonding) technology. Fairchild
began research on SDB in 1996. Unlike existing technology, which uses an epi-grown wafer,
SDB technology binds P+ substrate and N- substrate directly to allow easier manipulation of
the thickness of the N- substrate. This enables easier fabrication of high voltage IGBTs. Specifically, as the formation of a high density N+ buffer layer is possible, fast switching characteristics can be obtained without high density electron irradiation, which increases the leakage
current and decreases reliability. Hence, it enables the production of high speed, highly efficient and reliable IGBTs. It is also suitable for large capacity drives, as it has the same temperature characteristics as the NPT IGBT, which is suitable for a parallel drive. In the year 2000,
Fairchild has applied this technology to develop 1500V and 1700V IGBTs. These can be used
in 220V 1φ IH applications. Now, Fairchild is in the process of developing IPMs (Intelligent
Power Modules), which are IGBT Inverter Modules that combine the control ICs in order to
provide a lot of intelligent functions. The IPMs will drastically change the three-phase AC/DC
Motor Speed Control arenas paving the way for reliable, compact and high performance
designs.
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Rev. A2, February 2001
2. Device structure and operation
2-1. Structure
The IGBT combines the advantages of a power MOSFET and a bipolar power transistor. Similarly its structure is a combination of the two devices. As shown in Fig. 1, the input has a MOS
gate structure, and the output is a wide base PNP transistor. The base drive current for the
PNP transistor is fed through the input channel. Besides the PNP transistor, there is an NPN
transistor, which is designed to be inactivated by shorting the base and the emitter to the
MOSFET source metal. The 4 layers of PNPN, which comprises the PNP transistor and the
NPN transistor form a thyristor structure, which causes the possibility of a latch-up. Unlike the
power MOSFET, it does not have an integral reverse diode that exists parasitically, and
because of this it needs to be connected with the appropriate fast recovery diode when
needed.
GATE
EMITTER
EMITTER
.
N-CH.
P
-
+
N
J3
RS
.
GATE
NPN
RS
P
NPN
+
RMODULATION
N-CHANNEL
-
J2
-
N epi. (N drift)
PNP
PNP
+
N buffer
J1
+
P substrate
.
COLLECTOR
COLLECTOR
Figure 1. Equivalent Circuit for the IGBT & a Cross Section of the IGBT Structure (PT & N-Channel)
PT & NPT
An IGBT is called a PT (punch-through) or asymmetrical when there is an N+ buffer layer
between the P+ substrate and N- drift region. Otherwise, it is called an NPT (non-punchthrough) IGBT or symmetrical IGBT. The N+ buffer layer improves turn-off speed by reducing
the minority carrier injection quantity and by raising the recombination rate during the switching transition. In addition, latch-up characteristics are also improved by reducing the current
gain of the PNP transistor. The problem is that the on-state voltage drop increases. However,
the thickness of the N- drift region can be reduced with the same forward voltage blocking
capability because the N+ buffer layer improves the forward voltage blocking capability. As a
result, on-state voltage drop can be lowered. Hence, the PT-IGBT has superior trade-off characteristics as compared to the NPT-IGBT in switching speed and forward voltage drop. Currently, most commercialized IGBTs (Fairchild IGBTs) are PT-IGBTs. Section (3-3) about static
blocking characteristics illustrates that IGBT forward and reverse blocking capability are
approximately equal because both are determined by the same N- drift layer thickness and
resistance. The reverse-blocking voltage of PT-IGBTs that contain N+ buffer layer between P+
substrate and N- drift region is lowered to tens of volts due to the existence of a heavy doping
region on both sides of J1.
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Rev. A2, February 2001
2-2. Operation
Turn-on
When the device is in the forward blocking mode, and if the positive gate bias (threshold voltage), which is enough to invert the surface of P-base region under the gate, is applied, then an
n-type channel forms and current begins to flow. At this time the anode-cathode voltage must
be above 0.7V (potential barrier) so that it can forward bias the P+ substrate / N- drift junction
(J1). The electron current, which flows from the N+ emitter via the channel to the N- drift
region, is the base drive current of the vertical PNP transistor. It induces the injection of hole
current from the P+ region to the N- base region. The conductivity modulation improves
because of this high level injection of the minority carrier (hole). This increases the conductivity of the drift region by a factor varying from ten to hundred. This conductivity modulation
enables IGBTs to be used in high voltage applications by significantly reducing the drift region
resistance. There are two kinds of currents flowing into the emitter electrode. One is the electron current (MOS current) flowing through the channel, and the other is the hole current (bipolar current) flowing through the P+ body / N- drift junction (J2).
Turn-off
The gate must be shorted to the emitter or a negative bias must be applied to the gate. When
the gate voltage falls below the threshold voltage, the inversion layer cannot be maintained,
and the supply of electrons into the N- drift region is blocked, at which point, the turn-off process begins. However, the turn-off cannot be quickly completed due to the high concentration
minority carrier injected into the N- drift region during forward conduction. First, the collector
current rapidly decreases due to the termination of the electron current through the channel,
and then the collector current gradually reduces, as the minority carrier density decays due to
recombination.
3. Basic Characteristics
3-1. Advantages, Disadvantages and Characteristic Comparison with
BJT and MOSFET
Advantages
(1) High forward conduction current density and low on-state voltage drop:
The IGBT has a very low on-state voltage drop due to conductivity modulation and has
superior on-state current density compared to the power MOSFET and bipolar
transistor. So a smaller chip size is possible and the cost can be reduced.
(2) Low driving power and a simple drive circuit due to the input MOS gate structure:
The IGBT can easily be controlled as compared to current controlled devices
(thyristor, BJT) in high voltage and high current applications.
(3) Wide SOA:
With respect to output characteristics, the IGBT has superior current conduction capability
compared with the bipolar transistor. It also has excellent forward and reverse blocking
capabilities.
Disadvantages
(1) Switching speed (less than 100kHz) is inferior to that of the power MOSFETs, but it is
superior to that of the BJT. The collector current tailing due to the minority carrier causes
the turn-off speed to be slow.
(2) There is the possibility of latch-up due to the internal PNPN thyristor structure.
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Rev. A2, February 2001
The Characteristics Comparison with a BJT and a MOSFET
Table 1: The IGBT Characteristics Comparison with BJT, MOSFET
Features
BJT
MOSFET
IGBT
Drive Method
Current
Voltage
Voltage
Drive Circuit
Complex
Simple
Simple
Input Impedance
Low
High
High
Drive Power
High
Low
Low
Switching Speed
Slow (µs)
Fast (ns)
Middle
Operating Frequency
Low (less than 100kHz) Fast (less than 1MHz)
Middle
S.O.A.
Narrow
Wide
Wide
Saturation Voltage
Low
High
Low
3-2. Latch-up
The IGBT contains a parasitic PNPN thyristor structure between the collector and the emitter.
A latch-up means the turning on of the thyristor. When there is action by a thyristor, the IGBT
current is no longer controlled by the MOS gate. The IGBT would be destroyed because of
excessive power dissipation produced by the amount of current over the rated value between
the collector and the emitter.
Causes of latch-up
(1) Static latch-up mode:
Since the conductivity of the drift region under the gate electrode is increased by the
introduction of electron current through the channel, most of the holes injected into the
drift region are injected at the P body region under the channel and flow to the source
metal along the bottom of N+ source. Due to this, the lateral voltage drops across the
shunting resistance (RS, refer to Fig. 1) of the P body layer. If this voltage drop becomes
greater than the potential barrier of the N+ source / P body layer junction (J3), electrons
are injected from the N+ source to the P body layer, and the parasitic NPN transistor
(N+ source, P body and N- drift) is turned-on. If the sum of the two NPN, PNP parasitic
transistors’ current gain becomes 1 (αNPN + αPNP≥1), latch-up occurs.
(2) Dynamic latch-up mode:
When the IGBT is switched off, the depletion layer of the N- drift / P body junction (J2) is
abruptly extended, and the IGBT latches up at a current lower than 1/2 of the static
latch-up current due to the displacement current. And because of this, the safe operating
area is limited.
Avoidance of latch-up
The first method is to prevent the NPN parasitic transistor from turning on, and the second is
to keep the sum of the two NPN, PNP parasitic transistors’ current gain less than 1 (αNPN +
αPNP<1) when the NPN parasitic transistor is turned on. In the latter case, the two transistors’
current gain must be reduced. As the current gain is proportional to the base transport factor
and the injection efficiency of an emitter-base junction, both must be reduced. In the PNP transistor, electron irradiation and the inserting of a N+ buffer layer between the P+ substrate and
N- drift region can be used, but it is difficult to reduce the current gain in the NPN transistor. So
the following representative methods to improve latch-up characteristics are focused on preventing the turn-on of the NPN parasitic transistor.
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Rev. A2, February 2001
(1) Application of P+ body
This method is applied to most MOS gate power devices. In addition to the body, a
highly doped P+ region is formed in the middle of the body, and it covers most of the
bottom part of source. When the doping concentration of the P+ region is excessively
raised to improve latch-up characteristics, the P+ region affects the threshold voltage by
diffusing the channel region and debase the forward characteristics.
(2) Cushion structure p+ Double Implantation:
When using a highly doped P+ body, there are limitations in lowering the resistance of the
bottom part of N+ source as the doping concentration of the P+ body reduces
horizontally due to diffusion. Ion is first injected to form the N+ source. Then high energy
p+ ion is injected, and it is put through a thermal process to form P+region at the bottom
of the N+ source. Fairchild is producing latch-up proofed IGBTs by applying
this method and EBR (Emitter Ballast Resistor: further explained in (6)).
(3) Application of short N+ source:
This method reduces the resistance at the bottom of the N+ source.
(4) Reduction of the hole current:
The voltage drop across the bottom of the N+ source can be reduced by reducing the
hole current passing through the P body. Inserting the highly doped N+ buffer, reducing
the space between P bodies, and reducing the current gain by electron irradiation are
the methods to reduce the hole current.
(5) Minority Carrier By-Pass:
The minority carrier by-pass design has two hole current paths. One is a normal path
and the other flows to the emitter contact directly without passing through the bottom of
the N+ emitter. So the hole current flowing through the bottom of the N+ emitter is
reduced to half, and the latching current density doubles. In spite of an increase in the
on-state voltage drop, this method is widely used because it enables safe operation
without latch-up phenomenon at higher current and temperature.
(6) Layout:
The latch-up characteristics can be improved by changing the N+ source’s cell structure
on the surface of the device. Generally, a linearly structured cell has worse forward
current conduction characteristics than a cellularly structured cell, but shows better latchup characteristics. A MSS (multiple surface short) structure with better latch-up
characteristics as compared to the linear structure has also been developed. There is
another method (EBR) used in Fairchild IGBTs, which increases the voltage drop level
across the bottom of N+ source that forward biases the emitter (N+ source region)-base
(P body region) junction (J3). This is done by making the voltage drop inside the N+
source by changing the length of the N+ source.
(7) Others:
There is also a method to build in a sense IGBT cell and protection devices. This limits
the current by reducing the gate voltage automatically, when the current flows over a
certain limit. If a large series gate resistance RG is used when the IGBT is applied to a
certain set, the turn-off speed is slowed and the diffusion speed of the depletion region of
the drift region also slows down. Hence, the dynamic latch-up can be prevented. IGBTs
currently under production have the latch-up proof characteristics.
High temperature characteristics (latching current density)
With a rise in temperature, the current gains of the NPN and PNP transistors increase. This
decreases the latching current. The effect is aggravated by an increase in the resistance of the
P base region due to a decrease in hole mobility.
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Rev. A2, February 2001
3-3. Static Blocking Characteristics
Reverse Blocking Capability
When negative voltage is introduced to the collector as shown in Fig. 1, the P+ substrate / Ndrift junction (J1) is reverse biased, and the depletion layer generally expands to the N- drift
region. As such, securing an optimal design in resistivity and thickness for the N- drift region is
essential in obtaining desirable reverse blocking capability. As a general guideline, the width of
the N-drift region is equivalent to the sum of depletion width at maximum operating voltage
and minority carrier diffusion length. It is important to optimize the breakdown voltage while
maintaining a narrow N- drift region width, as the forward voltage drop increases with an
increase in N- drift region width. The following is an equation for calculating the N- drift region
width:
d1 =
2εV m
--------------- + L p
qND
where, d1: N-drift region width
Vm: maximum blocking voltage
ND: doping concentration
Lp: minority carrier diffusion length
Forward Blocking Capability
When the gate is shorted at the emitter, and a positive voltage is introduced at the collector,
the P base / N- drift region junction (J2) is reverse biased, and it is supported upto the rated
voltage by the depletion region formed at the N- drift region.
3-4. Leakage Current
Leakage current is divided into two types. One is leakage current from the depleted drift
region, and the other flows on the surface of the junction termination. Since the IGBT uses a
P+ substrate and is irradiated with electrons to improve switching speed, the amount of leakage current from the drift region is greater than that of the power MOSFET. If there is a high
voltage between the anode and cathode when the IGBT is off, the depletion region widens to
nearly the entire drift region from the P+ body / N- drift junction (J2), and the entire region
becomes depleted. The electron hole created by the heat of the depletion region is manifested
as leakage current according to the area of the depletion region and minority carrier lifetime.
This increases with the increase of the rated voltage and current. In particular, leakage current
increases as the minority carrier lifetime shortens. As such, leakage current tends to increase
with higher speed IGBTs.
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Rev. A2, February 2001
3-5. Forward Conduction Characteristics
Due to its structure, the IGBT is sometimes viewed as a serial connection of the MOSFET and
PiN diode, and sometimes it is seen as a wide base PNP transistor driven by the MOSFET in
Darlington configuration. The former description can be used to understand the behavior of
the device, but the latter better describes the IGBT.
25
Common Emitter
TC = 25℃
20V
15V
IC [A]
Increasing VGE
Collector Current, I
C
[A]
20
15
12V
10
VGE = 10V
5
0
0
2
4
6
8
VCE [V]
Collector - Emitter Voltage, VCE [V]
Figure 2. Static Characteristics of the IGBT
Fig. 2 is a graph of the IGBT’s static characteristics. Even if a MOSFET channel of the input
side is formed, the collector current does not flow if the anode-cathode forward voltage drop
does not exceed approximately 0.7V as in the PiN diode. In addition, the current is saturated
when the voltage across the MOSFET channel is greater than (VGE – Vth) and has an infinite
output resistance, as in a power MOSFET. However, in a symmetrical IGBT, the collector current increases with the increase in collector voltage, and the rate of increase in the collector
current also increases with the increase in collector voltage. Such finite output resistance is
due to a shortening of the channel due to an increase in the collector voltage, and a secondary
decrease in the drain output resistance due to bipolar transistor current flow. In order to
increase the collector output resistance, an asymmetrical structure with a N+ buffer layer
between the N- drift region and P+ substrate is used to prevent an increase in the bipolar transistor’s current gain with the increase in the collector voltage. In an asymmetrical structure, the
width of the undepleted N- drift region does not change rapidly with the increase in the collector voltage due to the high concentration of the buffer layer, but it remains the same width as
the N+ buffer layer for all collector voltages. This results in a constant value of the PNP transistor’s current gain. In addition to this, the N+ buffer layer reduces the injection efficiency of the
P+ substrate / N+ buffer junction (J1). This reduces the current gain of the PNP transistor. As
such, an IGBT with an asymmetrical structure has much superior output characteristics than a
symmetrical type. In addition, collector output resistance can be increased with electron irradiation to shorten the minority carrier lifetime, which reduces the diffusion length. The following
is the equation for obtaining the saturated collector current of the IGBT:
µ ns C ox Z
1
2
I C ,sat = ---------------------------- ----------------------- ( V GE – V th )
( 1 – α PNP ) 2L CH
where, µns: surface mobility of electrons
Cox: gate-oxide capacitance per unit area
Z: channel width
LCH: channel length
Vth: threshold voltage
VGE: applied gate voltage
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Rev. A2, February 2001
Transconductance at the active region can be obtained by differentiating the IC,sat with respect
to VGE.
µ ns C ox Z
1
g fe = ---------------------------- ----------------------- ( V GE – V th )
( 1 – α PNP ) L CH
The IGBT’s saturated collector current and transconductance are higher than those of the
power MOSFETs of the same aspect ratio (Z/LCH). This is because the PNP transistor’s current gain (αPNP) is less than 1 (0.2 to 0.3 in general).
On-state voltage drop
Forward current-voltage characteristics and the conduction loss of a MOSFET are described
as on-resistance. On the other hand, the characteristics of the IGBT are described as voltage
drop at rated current, as is the case with the bipolar power transistor. On-state voltage drop is
comprised of voltage drop of the forward biased P+ substrate / N- drift junction (J1), the voltage
drop of conductivity modulated N- drift region and the voltage drop of MOSFET. Cut-in voltage
for forward biased J1 is about 0.7V at room temperature. Cut-in voltage decreases due to a
sharp increase in intrinsic carrier concentration as the temperature rises. The voltage drop of
the N drift region can be obtained by integrating the electric field of the entire drift region, and
it is generally less than 0.1V due to a strong conductivity modulation caused by injected holes
from J1. The voltage drop of a MOSFET is the sum of the voltage drops from the channel
region, JFET region and accumulation layer. Due to a decrease in drift layer resistance, the
portion of JFET resistance and channel resistance is increased in the voltage drop between
on-state collector-emitter. Hence, low JFET and channel resistance design are important factors in obtaining the best performance in an IGBT. The voltage drop at the channel is proportional to the channel length, gate oxide thickness. And it is inversely proportional to channel
width, electron mobility and gate bias. The channel width can be increased by increasing the
concentration of circuits by decreasing the size of each unit cell. But because of this the JFET
resistance increases significantly, so the optimal size of the unit cell exists for each voltage rating. The IGBT decreases the minority carrier lifetime with electron irradiation in order to
improve the switching speed, and this increases the on-state voltage drop. Even in IGBTs with
the same structure, the IGBT with a fast switching speed has a larger voltage drop, and the
IGBT with a slower switching speed has a smaller on-state voltage drop depending on the
condition of electron irradiation, which takes place after device fabrication.
High temperature characteristics
One must be aware of the changes in characteristics from changes in temperature, as the
IGBT’s input characteristics are similar to a MOSFET, and output characteristics are similar to
bipolar transistors. As temperature rises, the energy barrier of the P+ substrate / N- drift region
junction (J1: emitter-base junction of PNP transistor) decreases, which leads to a lower cut-in
voltage, and the threshold voltage decreases as in a MOSFET. As channel resistance
increases, the amount of electron current (MOS current) decreases, which is injected to the Ndrift region. However, current gain, which is the ratio of the hole current (bipolar current) to the
electron current, increases. In addition, N- drift region (base of the PNP transistor) resistance
increases. Due to these characteristics, changes in cut-in voltage of J1 are larger than those in
channel resistance and N- drift region resistance at low collector current level, so the IGBT has
negative temperature coefficient similar to the bipolar transistor. On the other hand, channel
resistance and N- drift region resistance determine the on-state voltage at high collector current, which results in a positive temperature coefficient similar to a power MOSFET. The crossover point for the two characteristics is different for each product, and the collector-emitter
voltage drop is independent from temperature at the crossover point. In real applications, it is
used in areas with negative temperature coefficient, and these factors must be considered in
parallel application. Figs. 3~5 illustrate these characteristics with graphs from the data sheet.
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Rev. A2, February 2001
4.0
20
Common Emitter
VGE = 15V
3.5
CE
12
Collector - Emitter Voltage, V
Collector Current, I C [A]
16
[V]
Common Emitter
VGE = 15V
TC = 25℃ ━━
TC = 125℃ ------
PTC characteristics
8
4
Crossover
10A
3.0
2.5
5A
2.0
IC = 3A
1.5
NTC characteristics
1.0
0
1
-50
10
0
20
150
20
Common Emitter
TC = 25℃
[V]
Common Emitter
TC = 125℃
16
16
Collector - Emitter Voltage, V
CE
CE
[V]
100
Figure 4. Saturation Voltage vs. Case
Temperature at variant Current Level
Figure 3. Typical Saturation Voltage Characteristics
Collector - Emitter Voltage, V
50
Case Temperature, TC [℃]
Collector - Emitter Voltage, VCE [V]
12
8
10A
4
5A
12
8
10A
4
5A
IC = 3A
IC = 3A
0
0
0
4
8
12
16
20
0
4
8
12
16
20
Gate - Emitter Voltage, VGE [V]
Gate - Emitter Voltage, VGE [V]
Figure 5. Saturation Voltage vs. VGE due to Collector Current, Case Temperature
3-6. Switching Characteristics
Turn-on
When the device is in the forward blocking mode, and if the positive gate bias (threshold voltage), which is enough to invert the surface of P base region under the gate, is applied, then an
n-type channel forms and the current begins to flow. At this time the anode-cathode voltage
must be above 0.7V (potential barrier), so that it can forward bias the P+ substrate / N- drift
junction (J1). The electron current flowing from the N+ emitter to the N- drift region through the
channel is the base drive current of the vertical PNP transistor, and it induces a minority carrier
(hole) injection from the P+ region to the N- base region. The current that flows to the emitter
electrode are divided into the electron current (MOS current) flowing through the channel and
bipolar current flowing through the P body / N- drift junction (J2). When gate bias falls to near
the threshold voltage at on-state, the inversion layer conductivity is reduced, and significant
voltage drop that arises from electron current flow occurs across the region as in a MOSFET.
When the voltage drop is equal to the difference between the gate bias and threshold voltage
(VGE - Vth), then the channel is pinched off. At this point, the electron current becomes saturated. Since this limits the base drive current of the PNP transistor, the hole current flowing
through the PNP transistor is also limited. As a result, the device operates with saturated current at the active region (gate controlled output current).
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Rev. A2, February 2001
Turn-off
The gate must be shorted to the emitter or a negative bias must be applied to the gate. When
the gate voltage falls below the threshold voltage, the inversion layer cannot be maintained,
and the supply of electrons into the N- drift region is blocked. At this point the turn-off process
begins. As illustrated in Fig. 6, the collector current (ICO) falls to zero in two stages. As the
electron current supplied through the MOSFET channel during the on-state is stopped, collector current suffers an initial abrupt fall (ICD). After that, the tail current (ICT) comes from the
minority carrier (hole) that was injected through the N- drift region from the P+ substrate during
the on-state. The tail current of the IGBT lowers switching characteristics and increases
switching loss. Since N- drift region is the base of the PNP transistor, it cannot be approached
from outside, so it is not possible to control the tail current from outside. But it can be controlled with the amount of minority carrier (hole) injected through the N- drift region and recombination rate when it is off. In order to reduce the amount of injected minority carrier and
increase the recombination rate when it is off, the concentration and the thickness of the N+
buffer layer between the P+ substrate and N- drift region must increase, as well as the dose of
electron irradiation (in FSC, electron irradiation is applied to above 600V class except 400V)
that takes place after device fabrication. However, improving the switching speed of the IGBT
generally accompanies reduced current handling capability. As such, the trade-off between
switching speed, which is related to switching loss, and forward voltage drop, which in turn is
related to conduction loss, is important. The asymmetric structure is superior in such trade-offs
as compared to a symmetric structure, and it can be improved by increasing the doping concentration in the buffer layer. In terms of power loss, the power MOSFET is better suited for
lower blocking voltage and high operating frequency applications, while the IGBT is better
suited for higher blocking voltage and lower operating frequency.
IC
ICO
ICD
MOSFET current (electron current) => ≒90%
ICT
BJT current (hole current) : tail current => ≒10%
t
Figure 6. Collector Current During Turn-off
High temperature characteristics
The minority carrier lifetime in the drift region increases as the temperature increases. This not
only delays recombination process (tail current) of the minority carrier, but it also increases the
PNP transistor gain. So the portion of the initial abrupt fall (ICD) in the overall collector current
reduces. As such, tf(fall time) of the spec is lengthened, and turn-off time increases with an
increase in temperature, and the asymmetric structure has a lower rate of increase than the
symmetric structure.
3-7. SOA (Safe-Operating-Area)
The SOA of an IGBT can be divided into the following three boundaries based on the stress
conditions of voltage and current.
(1) High voltage, low current: Maximum voltage is limited by breakdown voltage.
(2) High current, low voltage: Maximum current is limited by latch-up of parasitic thyristor.
(3) Simultaneous high current and voltage: Limited by the rise in temperature caused by high
power dissipation.
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Rev. A2, February 2001
If the high current and voltage are introduced simultaneously for a short time, then the SOA is
no longer limited by power dissipation. The following explanation illustrates the safe operating
area for short duration simultaneous application of high current and voltage stress at the time
of turn-on and turn-off of the IGBT with inductive load with FBSOA and RBSOA.
FBSOA (Forward Biased Safe-Operating-Area)
FBSOA indicates the safe operating region when both electron and hole current flow takes
place along the high voltage on the device as positive gate bias is introduced on the IGBT during turn-on transient (Fig. 7). This is an important characteristic of an application with inductive
load, and the IGBT has superior FBSOA without a snubber. The FBSOA of an IGBT is the
maximum voltage the device can withstand without failure when the collector current is saturated. The limit can be obtained with the following equation:
α PNP M = 1
where, αPNP: current gain of the PNP transistor
1
α PNP = --------------------------------cosh ( 1 ⁄ L a )
where, l: undepleted N base width
La: ambipolar diffusion length
M: multiplication coefficient
n
V
M = 1 – -------------------
BV SOA
–1
where, V: applied reverse bias supported by the junction
BVSOA: avalanche breakdown voltage
n: 6 for a p+n junction
It can be seen from the above formula that avalanche breakdown takes place at lower collector bias when the collector current increases. By reducing the doping concentration at the drift
region, BVSOA(breakdown voltage) can be raised, and this is possible in an asymmetrical
structure, which does not have the danger of reach-through breakdown.
RBSOA (Reverse Biased Safe-Operating-Area, Turn-Off SOA)
RBSOA shows the safe operating area during the turn-off transient period, when the gate bias
with positive value switches to zero or a negative value, which leaves the device with high voltage and hole current transport (refer to Fig. 8). Unlike the FBSOA, the snubber circuit design
is important in the IGBT for safe operation during turn-off. A wider RBSOA can be obtained by
reducing the current gain of the PNP transistor. The RBSOA of p-channel and n-channel
IGBTs are compared under an identical doping profile and cell structure. The n-channel has a
wider safe operating area for collector voltage (avalanche induced SOA limit) but a narrower
safe operating area for collector current (current induced latch-up limit). It is the opposite for pchannel. The avalanche induced SOA limit is affected by the impact ionization coefficient of
the carrier transported through the depletion layer. The n-channel IGBT has a superior avalanche induced SOA limit because the impact ionization coefficient of the hole transported in
the n-channel IGBT is lower than that of the electron. Since N base region sheet resistance of
the p-channel IGBT is about 2.5 times lower than the P base region sheet resistance of the nchannel IGBT, the p-channel IGBT is superior to the n-channel IGBT in current induced latchup limit.
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Rev. A2, February 2001
50
40
Ic MAX. (Pulsed)
50us
100us
Ic MAX. (Continuous)
C
[A]
1㎳
Collector Current, I
Collector Current, I C [A]
10
DC Operation
1
0.1
0.01
0.1
Single Nonrepetitive
Pulse T C = 25℃
Curves must be derated
linearly with increase
in temperature
1
10
Safe Operating Area
VGE = 20V, T C = 100℃
1
10
100
1
1000
Collector-Emitter Voltage, VCE [V]
10
100
1000
Collector-Emitter Voltage, VCE [V]
Figure 8. Turn-off SOA Characteristics
Figure 7. SOA Characteristic
SCSOA (Short Circuit Safe-Operating-Area)
The system must be shut down to prevent the destruction of the device when the load shorts
and allows excessive current, high voltage and high current to simultaneously put stress on
the device. SCSOA defines the area of safe operation for the device under such fault condition.
4. Explanation of Data Sheet Parameters
4-1. Absolute Maximum Ratings
VCES (Collector-Emitter Voltage)
It is the maximum allowable voltage between the collector and the emitter when the gate and
the emitter are shorted. If this limit is exceeded, the device can be destroyed due to the breakdown of the junction between the collector and the emitter.
VGES (Gate-Emitter Voltage)
It is the maximum voltage that can be introduced between the gate and the emitter. It is specified at 20V or 25V depending on the thickness and the characteristics of the gate oxide layer
(or by product).
IC (Collector Current @TC=25°C and @TC=100°C)
It is the maximum DC current the device can flow at the specified case temperature, and the
calculation of IC can be obtained from the following formula for calculating PD.
T J ( max ) – T C
= V ce × I c
P D = --------------------------------R θJC
[W]
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Rev. A2, February 2001
The above formula can be restated with respect to Ic.
2
R θJC × VTO max + 4 × R CE × ( T J ( max ) – T C ) VTO max
I c = ------------------------------------------------------------------------------------------------------------------------ – ----------------------2 × R CE
2 × R θJC × R CE
[A]
where, RθJC: junction-to-case thermal resistance
TC: case temperature
TJ(max): maximum junction temperature
VCE, RCE, VTOmax: Refer to Fig. 9
V ce = VTO + R CE × I c
∆V ce
R CE = ------------∆I c
VTO max = VTO + [ V ce ( sat ) ,( max ) – V ce ( sat ) ,( typ ) ]
IC
∆Ic
∆Vce
0
VTO
VCE
Figure 9. Linear Interpolation of output characteristics at Tj = 150°C
The specification is influenced by the device’s ability to remove heat (heat resistance), and the
current specification at the case temperature of 25°C represents the current rating of the
device. The current specification at the case temperature of 100°C is more usable in real
applications.
ICM (Pulsed Collector Current)
* Repetitive rating: Pulse width limited by max. junction temperature
It is the peak current the device can flow above the IC specification under the maximum junction temperature. It varies with the current pulse width, duty cycle and the heat dissipation conditions.
IF (Diode Continuous Forward Current @TC=100°C)
It is the maximum DC current the diode can flow in the forward direction at the specified case
temperature.
IFM (Diode Maximum Forward Current)
It is the peak current the diode can flow above the IF specification under the maximum junction
temperature.
TSC (Short Circuit Withstand Time @TC=100°C)
Refer to the TSC explanation of Electrical Characteristics of IGBT.
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Rev. A2, February 2001
PD (Maximum Power Dissipation @TC=25°C, @TC=100°C)
The maximum power dissipation value with the assumption that the junction temperature rises
to the maximum rating at case temperatures of 25°C and 100°C.
T J ( max ) – T C
P D = --------------------------------R θJC
[W]
TJ (Operating Junction Temperature)
The industry standard range is -55°C ~ 150°C, and the maximum junction temperature is
150°C.
Tstg (Storage Temperature Range)
It is the range of temperature for storage or transportation for the device, and it must be
between -55°C ~ 150°C.
TL (Maximum Lead Temp. for Soldering Purposes at 1/8 from case for 5 seconds)
It is the maximum lead temperature during soldering. The lead temperature must not exceed
300°C for 5 seconds at 1/8” from the case.
4-2. Thermal Characteristics
The power loss from the device turns into heat and increases the junction temperature inside
the chip. This degrades the characteristics of the device and shortens its life. It is important to
allow the heat produced from the chip junction to escape outside to lower the junction temperature. The thermal impedance Zthjc(t) measures the ability of the device to dissipate heat.
R θJC
PD
Sink
Case
Junction
R θSA
R θCS
TJ
Ambient
TA
TS
TC
Figure 10. An Equivalent Circuit Based on Thermal Resistances
Sink
Case
R θJC
TJ
R θSA
R θCS
DIODE
IGBT
Ambient
R θJC
Junction
Junction
PD
PD
TC
TS
TA
TJ
Figure 11. An Equivalent Circuit Based on Thermal Resistances (co-pak)
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Rev. A2, February 2001
If the movement of heat is considered the same as current and it is changed to an electric circuit, the heat dissipation channel can be described as shown in Figs. 10 and 11 after considering thermal resistance. However, this is the case in DC operation, and most IGBT applications
involve switching operations with a certain duty factor. In such cases, an equivalent circuit can
be described with thermal impedance with consideration for thermal capacitance as well as
thermal resistance (refer to the Thermal Response Characteristics section). The entire thermal
resistance from the chip junction to ambient is denoted as RθJA(Thermal Resistance, Junction–to–Ambient), and the following formula is for an equivalent circuit.
R θJA = R θJC + R θCS + R θSA
[ °C ⁄ W ]
RθJC (Thermal Resistance, Junction-to-Case)
It is the internal thermal resistance from the chip junction to the package case. This thermal
resistance of a pure package is determined by the package design and lead frame materials.
RθJC is measured under TC = 25[°C] condition and can be described by the following formula.
TJ – TC
R θJC = ------------------PD
[ °C ⁄ W ]
TC = 25[°C] is a condition with infinite heat sink.
* Infinite heat sink: Temperature of the package case is the same as the ambient temperature,
it is a heat sink that can achieve TC = TA.
Co-pak products have specified the RθJC of IGBT and the RθJC of the diode, respectively.
RθCS (Thermal Resistance, Case–to–Sink)
It is the thermal resistance from the package case to the heat sink. It varies with the package,
type of insulation sheet, the type of thermal grease and the thickness of its application, and the
mounting method to the heat sink.
RθSA (Thermal Resistance, Sink–to–Ambient)
It is the thermal resistance from the heat sink to ambient, and it is determined by the geometric
structure, surface area, cooling method and the quality of the heat sink.
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Rev. A2, February 2001
Thermal Response Characteristics
Fig. 12 is derived from the Fig. 17 graph in the data sheet, and it illustrates Zthjc(t) (junction–
to–case thermal impedance) with respect to changes in rectangular pulse duration under several duty factors. Zthjc(t) determines the peak junction temperature under the equation for peak
TJ = Pdm × Zthjc + TC (Power dissipation during the conduction period is assumed to be constant at Pdm from Fig. 17 in the data sheet). As it approaches DC operation with a duty factor
of D=1, and the rectangular pulse duration lengthens, Zthjc(t) is saturated at the maximum
value of RθJC. Fig. 13 illustrates that the junction temperature increases as the duty factor
increases.
Thermal Response, Zthjc [℃/W]
10
1
0.5
0.2
0.1
0.05
0.1
0.02
0.01
single pulse
0.01
10
-5
-4
10
10
-3
-2
10
10
-1
0
10
10
1
Rectangular Pulse Duration [sec]
Figure 12. Transient Thermal Impedance of IGBT
Pdm
t 11
t 12
t2
Junction Temperature
TJ12
( Tj )
TJ11
TC
0
t 11
t 12
Time ( t )
Figure 13. Changes in junction temperature respect to conduction time
A single pulse curve determines the thermal resistance for repetitive power pulses having a
constant duty factor (D) as shown in the following equation.
Z thjc ( t ) = R θJC ⋅ D + ( 1 – D ) ⋅ S thjc ( t )
where, Zthjc(t): Thermal impedance for repetitive power pulse with duty factor D
Sthjc(t): Thermal impedance for single pulse
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Rev. A2, February 2001
(4-3) Electrical Characteristics of the IGBT
Off Characteristics
BVCES (Collector-Emitter Breakdown Voltage)
It is the breakdown voltage between collector and when the gate is shorted to emitter and a
specified current flows.
∆BVCES / ∆TJ (Temperature Coefficient of Breakdown Voltage)
Collector-emitter breakdown voltage has positive temperature coefficient characteristics, and it
generally increases by 0.6 [V/°C].
ICES (Collector Cut-Off Current)
It is the maximum collector-emitter leakage current with the gate and the emitter shorted and
with the introduction of rated collector-emitter voltage.
IGES (G-E Leakage Current)
It is the maximum gate-emitter leakage current with the collector and the emitter shorted and
with the introduction of rated gate-emitter voltage.
On Characteristics
VGE(th) (G-E Threshold Voltage)
It is the gate-emitter voltage at which collector current begins to flow. In the data sheet, it
means the gate-emitter voltage that make the specified collector current flow.
VCE(SAT) (Collector to Emitter Saturation Voltage)
It is an important factor in determining the device’s conduction loss, and it is the collector-emitter voltage drop when the collector current is at the rated value (IC @TC=25° and
@TC=100°C) and the gate-emitter voltage is VGE=15V. When the collector current is low, it
has negative temperature coefficient characteristics, and when the current is high, it has positive temperature coefficient characteristics.
Dynamic Characteristics
The capacitance of an IGBT is generally measured under specific conditions, and its decrease
is inversely proportional to the bias voltage introduced in the collector-emitter as shown in Fig.
15. The datasheet specifies typical values under VGE = 0V, f = 1MHz and VCE = 30V.
Cies (Input Capacitance)
The input capacitance when the collector is shorted with the emitter.
Coes (Output Capacitance)
The output capacitance when the gate is shorted with the emitter.
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Rev. A2, February 2001
Cres (Reverse Transfer Capacitance)
The capacitance between the collector and the gate terminal.
Collector
O
700
Cgc
Common Emitter
VGE = 0V, f = 1MHz
TC = 25℃
600
Gate O
Capacitance [pF]
Cce
Cge
O
Emitter
500
400
Cies
300
200
Coes
100
Cies = Cge + Cgc , Cce is shorted
Cres
0
Cres = Cgc
1
10
Collector - Emitter Voltage, VCE [V]
Coes = Cce + Cgc
Figure 15. Capacitance Characteristics
Figure 14. Equivalent Circuit Showing Parasitic
Capacitance
Switching Characteristics
The diode-clamped inductive load circuit shown in Fig. 16 is a circuit commonly encountered
in power electronics. It is a test circuit for switching characteristics. So with this circuit, we
examine the IGBT’s turn-on and turn-off behavior. Fig. 17 is a realistic switching waveform
considering the characteristics of diode recovery and stray inductance (LS). First, we set the
condition to be in the constant steady state current, which initially flows through the inductive
load and then flows through the ideal diode (freewheeling diode) connected in parallel with the
inductive load.
LS
.
+
LOAD
DIODE
LS
LS
VCC
CBANK
2
RG
IGBT
3
VGG±
1
.
_
.
Figure 16. Diode-Clamped inductive load
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Rev. A2, February 2001
vGE( t
vGE(t)
VGE,Io
VGG+
VGE(th)
0
0
VGG-
VCC
IO
vCE(t)
vCE(t)
iC(t)
iC(t)
MOSFET current
VCE(sat
0
t0
t1 t2 t3 t4
BJT current
0
t5
t6
(a)
t7
t8
t9
(b)
Figure 17. VGE(t), VCE(t), IC(t) Switching waveforms
(a) Turn-on
(b) Turn-off
Since the input side of the IGBT has a MOS gate structure, its on and off state transition is
very similar to that of a power MOSFET.
The analysis of the Turn-on transition
Turn-on switching time is determined by the charging speed of input capacitance (Cies).
(1) t0 section: It is the section where vGE rises to VGE(th) while iG (gate current) charges the
parasitic input capacitance Cge, Cgc. The vGE increase pattern is shown to be linear, but it is
actually an exponential curve with time constant RG(Cge+Cgc). The vCE is maintained at the
VCC value, and iC remains at zero. The delay time is generally defined as the time when the
gate voltage is 10% of VGG+ to when iC reaches 10% of IO value. As such, most of the turn-on
delay falls under this section.
(2) t1 section: vGE continues to increase exponentially past VGE(th) as it does in t0 section. As
vGE increases, iC begins to increase to reach IO which is the full load current. In the t1 and t2
section, vCE appears to be shaved off than VCC. This is due to the voltage induced to LS in Fig.
16 during the increase in iC, VLS = LS*diC/dt. The amount it is shaved off is proportional to the
magnitude of diC/dt and LS, and the shape is determined by iC.
(3) t2, t3 sections: In iD, the diode current which begins to decrease from t1 section does not
immediately fall to zero, but there is a reverse recovery current where it flows in the reverse
direction. This current is added to iC current to take the same shape as iC in the t2 and t3 section. At this time, the diode voltage recovers and increases, and vCE decreases, and when vCE
is high, it decreases rapidly as Cgc takes a small value. In t3 section, Cgc discharges by
absorbing the gate drive current and the discharged current from Cge. At the end of t3 section,
reverse recovery of the diode stops.
(4) t4 section: In this section, iG continues to charge Cgc. vGE maintains VGE,Io value, and iC
maintains a constant full load current (IO). On the other hand, vCE falls at a rate of {(VGGVGE,Io)/(RGCgc)}. At this time, vCE has already fallen significantly, and when vCE is low, the
value of Cgc is large. Slow charging causes a voltage tail.
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Rev. A2, February 2001
(5) t5 section: In this section, vGE increases exponentially to VGG+ with a time constant
RG(Cge+Cgc,miller). At this point, Cgc,miller is Cgc which increases in the low vCE value due to
the Miller effect. In this section, vCE decreases slowly to collector-to-emitter on-state voltage
and becomes fully saturated. In addition to the effect from Cgc,miller, it is because the speed of
the IGBT PNP transistor portion to cross the active region to the on-state (hard saturation) is
slower than that of the MOSFET portion.
(6) t6 section: This is the section that includes most of the td(off) (Turn off delay time). vGE falls
from the injected VGG+ to VGE,Io. At this time, there is no change in the values of vCE or iC.
(7) t7 section: This is the section where vCE (collector voltage) increases according to the following equation, and the rate of increase can be controlled with RG(gate resistance).
V GE ,I O
dv CE
------------= ----------------------dt
C res ⋅ R G
(8). t8 section: vCE maintains VCC value, and iC decreases at the rate of the following equation,
and its rate of increase can also be controlled with RG.
V GE ,IO
di
-------c- = g fe ⋅ ---------------------C ies ⋅ R G
dt
As with the turn-on transient period, there is over-voltage as VLS = LS*diC/dt that is induced to
stray inductance from the effect of diC/dt is added to IGBT C-E terminals in sections t7 and 8.
The MOSFET current disappears at t8, which is the first of the two sections where iC
decreases.
(9) t9 section: PNP transistor current, out of the IGBT iC, disappears in this section, and this
current is commonly called the tail current. It takes place due to the recombination of the
minority carrier (hole), which has been injected into the N- drift region during the on-state. Due
to the existence of this region, the IGBT’s switching characteristics are inferior as compared
with a power MOSFET
SWITCHING TIME
Switching time is divided into 4 sections as shown in Fig. 18, and they are specified at
TC=25°C and 125°C for VCC=VCES/2, IC= IC,MAX(@TC=100°C), VGE=15V and under inductive
load conditions. Data for TC=125°C are provided to the user because the temperature of the
devices in the system rise during operation. Switching time generally rises due to increases in
RG (Gate Resistance), IC (Collector Current) and TC (Case Temperature). Figs. 19 and 20
show detailed changes in switching time with changes in RG, IC and TC. These data are not
absolute values, but they are included in the data sheet as a reference for design purposes.
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Rev. A2, February 2001
VGG+
90% VGG+
10% VGG+
0
VCC , IO
Gate-emitter voltage
VCC
IO
VCC
90% IO
90% IO
10% IO
10% VCC
Collector-emitter voltage
10% IO
0
Collector current
td(on) tr
td(off)
tf
Figure 18. The Switching Waveforms of Gate-Emitter Voltage,
Collector-Emitter Voltage and Collector Current
td(on) (Turn-On Delay Time)
The time it takes for the collector current to reach 10% of IO from the time a pulse is injected
into the gate.
tr (Rise Time)
The time it takes for the collector current to reach 90% of IO from 10%. Any fall in the collector
current after that point is not considered.
td(off)(Turn-Off Delay Time)
The time it takes for the collector-emitter voltage to reach 10% of VCC from the time a pulse is
removed from the gate (90% of VGG+).
tf(Fall Time)
The time it takes for the collector current to reach from 90% of IO to 10%. We ignore the section where the collector voltage is rising and the tail current where the current is less than 10%
of IO.
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Rev. A2, February 2001
Ton
Switching Time [ns]
100
Common Emitter
VCC = 300V, VGE = ± 15V
IC = 5A
TC = 25℃ ━━
TC = 125℃ ------
Switching Time [ns]
Common Emitter
VCC = 300V, VGE = ± 15V
IC = 5A
T C = 25℃ ━━
T C = 125℃ ------
Tr
Toff
Tf
Toff
Tf
100
10
10
10
100
Gate Resistance, RG [Ω ]
100
Gate Resistance, RG [Ω ]
Figure 19. Turn-On / Turn-Off Characteristics vs. Gate Resistance
Common Emitter
VGE = ± 15V, RG = 40Ω
T C = 25℃ ━━
T C = 125℃ ------
1000
Switching Time [ns]
Switching Time [ns]
Common Emitter
VGE = ± 15V, RG = 40Ω
TC = 25℃ ━━
TC = 125℃ -----100
Ton
Tr
Tf
Toff
Toff
Tf
100
10
3
4
5
6
7
8
9
10
3
Collector Current, IC [A]
4
5
6
7
8
9
10
Collector Current, IC [A]
Figure 20. Turn-On / Turn-Off Characteristics vs. Collector Current
SWITCHING ENERGY
It is not enough to calculate the switching loss only with the switching time, as there is a region
where some switching loss occurs although it is not specifically included in the switching time.
As such, a switching energy specification is indicated for system designers in calculating
switching loss. Indicating Eoff specification would allow designers to compensate for the region
where the collector-emitter voltage rises and the collector tail current that is outside the tf
region during turn-off, while Eon specification would compensate for the region where the collector-emitter voltage falls during turn-on. Eon, Eoff and Ets are specified at TC=25°C and
125°C for VCC=VCES/2, IC= IC,MAX(@TC=100°C), VGE=15V and under inductive load conditions. Data for TC=125°C are provided to the user because the temperature of the devices in
the system rise during operation. Switching energy generally rises due to increases in RG
(Gate Resistance), IC (Collector Current) and TC (Case Temperature). Figs. 21 and 22 show
detailed changes in switching energy with changes in RG, IC and TC. These data are not absolute values, but they are included in the data sheet as a reference for design purposes.
23
Rev. A2, February 2001
Eon (Turn-On Switching Loss)
It is the amount of total energy lost during turn-on under inductive load, and it includes the loss
from the diode reverse recovery. In practice, it is measured from the point where the collector
current begins to flow to the point where the collector-emitter voltage completely falls to zero in
order to exclude any conduction loss.
Eoff (Turn-Off Switching Loss)
It is the amount of total energy lost during turn-off under inductive load. In practice, it is measured from the point where the collector-emitter voltage begins to rise from zero to the point
where the collector current falls completely to zero.
Ets (Total Switching Loss)
It only includes the energy loss in the switching region, and it is expressed as the sum of Eon
and Eoff.
1000
1000
Eoff
Switching Loss [uJ]
Switching Loss [uJ]
Common Emitter
VGE = ± 15V, RG = 40Ω
TC = 25℃ ━━
TC = 125℃ ------
Eon
Eoff
100
Common Emitter
VCC = 300V, VGE = ± 15V
IC = 5A
T C = 25℃ ━━
T C = 125℃ ------
10
10
100
Eoff
100
Eon
3
Gate Resistance, RG [Ω ]
4
5
6
7
8
9
10
Collector Current, IC [A]
Figure 21. Switching Loss vs. Gate Resistance
Figure 22. Switching Loss vs. Collector Current
TSC (Short Circuit Withstand Time)
Fig. 23 shows an IGBT’s short circuit modes in a motor drive circuit such as an error in the
gate drive signal (1), shorting at the collector terminal (2), shorting at grounding (3). When it is
short circuited, current, which is several times the rated IC, flows due to the introduction of the
gate bias under a large voltage between the collector and the emitter of the IGBT. If the protection circuit is not functional at this time, it would lead to the destruction of the device. It commonly takes some time (normally 3~7[µs]) for the protection circuit to become active after the
over current is detected in an ordinary system. Hence, the device must be designed to withstand at least this period of time with a little margin for safety. The RUF-series, which is a short
circuit rated IGBT (IGBT for use in a motor drive), is protected for a minimum of 10 [µs] under
VCC=VCES/2, VGE=15V at TC=100°C conditions.
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Rev. A2, February 2001
2
2
2
1
1
1
3
3
3
MOTOR
2
2
2
(1)
1
(2)
1
(3)
1
3
3
3
Figure 23. Short Circuit Mode of IGBT
GATE CHARGE
It is the total gate charge necessary to fully turn on the IGBT. Test criteria are set at VCC=
VCES/2, VGE=15[V], IC= rated IC (@TC=100°C), which is similar to the conditions in actual
operation to give useful specifications for setting the switching speed or drive circuit design.
Fig. 24 shows the gate charge test circuit, and Fig. 25 shows the characteristics of the gate
charge by VCC. In the test circuit, a constant current source is used as the input source for the
gate drive, and resistance load has been connected to make changes in IC according to
changes in VCC.
Gate - Emitter Voltage, VGE [V]
15
RL
1
2
VCC
1
2
DUT
Common Emitter
RL = 60Ω
T C = 25℃
12
300V
200V
VCC = 100V
9
6
Qge
Qgc
3
3
Qg
0
0
3
6
9
12
15
18
Gate Charge, Qg [nC]
Figure 24. Gate Charge Test Circuit
Figure 25. Gate Charge Characteristics by VCC
Qg (Total Gate Charge)
It is the total gate charge necessary to raise the gate-emitter voltage to 15V, which is necessary to fully turn on the IGBT.
Qge (Gate-Emitter Charge)
The amount of gate charge necessary to charge the gate-emitter voltage to allow specified
collector current.
Qgc (Gate-Collector Charge)
It is the amount of gate charge charged in the gate-collector capacitor during the flat region. In
this region, the collector-emitter voltage falls from VCC to on-state collector-emitter voltage,
and the collector current maintains the specified collector current value.
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Rev. A2, February 2001
Le (Internal Emitter Inductance)
This is the stray inductance between the collector and emitter at a distance of 5mm from the
package. The value changes depending on the length of the bonding wires between the chip
and the emitter pad of the package lead frame. Typical values are 7.5[nH] for TO-220/F package, 14[nH] for TO-3P/F package and 18[nH] for TO-264 package.
4-4. Electrical Characteristics of a DIODE
VFM (Diode Forward Voltage)
Diode forward voltage is specified under the maximum IF (Diode Continuous Forward Current
@TC=100°C) and at case temperatures of 25°C and 100°C.
100
Forward Current, I
F
[A]
TC = 25℃ ━━
TC = 100℃ ------
10
1
0.1
0
1
2
3
Forward Voltage Drop, VF [V]
Figure 26. Forward Characteristics
Trr(Diode Reverse Recovery Time) &
Irr(Diode Peak Reverse Recovery Current) &
Qrr(Diode Reverse Recovery Charge)
Irr and Trr are defined as shown in Fig. 28 in the reverse recovery region. The amount of Qrr,
which is charged during diode conduction and is discharged during reverse recovery region,
can be calculated with Irr and Trr using the following equation:
1
Q rr = --- × I rr × T rr
2
Values of Trr, Irr and Qrr can vary according to the test conditions, but they are generally specified at TC=25°C and 100°C for rated IF (Diode Continuous Forward Current @TC=100°C) and
diF/dt = 200A/µs conditions. As shown in Fig. 27, Trr, Irr and Qrr generally increase with a rise
in case temperature. On the other hand, Irr and Qrr increase and Trr decreases when the diF/dt
slope rises.
26
Rev. A2, February 2001
100
[nC]
400
rr
Stored Recovery Charge, Q
[A]
rr
Reverse Recovery Current, I
500
VR = 200V
IF = 8A
TC = 25℃ ━━
TC = 100℃ ------
10
1
100
VR = 200V
IF = 8A
TC = 25℃ ━━
TC = 100℃ ------
300
200
100
0
100
1000
di/dt [A/us]
1000
di/dt [A/us]
80
VR = 200V
IF = 8A
TC = 25℃ ━━
TC = 100℃ ------
Reverce Recovery Time, t
rr
[ns]
100
60
40
20
0
100
1000
di/dt [A/us]
Figure 27. Trr, Irr, Qrr Characteristics with Changes in Case Temperature, diF/dt
27
Rev. A2, February 2001
Fig. 28 illustrates the test circuit and waveforms of each part for diode reverse recovery characteristics.
2
DUT
1
+
VCE
3
_
IS
L
1
.
Driver
2
RG
1
VR
2
Same Type as DUT
3
VGE
.
VGE
15[V]
Gate Pulse Width
D=
Gate Pulse Period
(Driver)
IF, Body Diode Forward Current
IS
(DUT)
trr
diF/dt
10[%] of Irr
Body Diode Reverse Current, Irr
ta
tb
Body Diode Recovery dvCE/dt
VCE
(DUT)
VFM
VR
Body Diode Forward Voltage Drop
Figure 28. Diode Reverse Recovery Characteristics Test Circuit & Waveforms
28
Rev. A2, February 2001
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Advance Information
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This datasheet contains the design specifications for
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©2001 Fairchild Semiconductor Corporation
Rev. G