Product Folder Sample & Buy Support & Community Tools & Software Technical Documents LM5017 SNVS783H – JANUARY 2012 – REVISED DECEMBER 2014 LM5017 100-V, 600-mA Constant On-Time Synchronous Buck Regulator 1 Features 3 Description • • The LM5017 is a 100-V, 600-mA synchronous stepdown regulator with integrated high side and low side MOSFETs. The constant on-time (COT) control scheme employed in the LM5017 requires no loop compensation, provides excellent transient response, and enables very high step-down ratios. The on-time varies inversely with the input voltage resulting in nearly constant frequency over the input voltage range. A high voltage startup regulator provides bias power for internal operation of the IC and for integrated gate drivers. 1 • • • • • • • • • • • • • Wide 7.5-V to 100-V Input Range Integrated 100-V High-Side, and Low-Side Switches No Schottky Required Constant On-Time Control No Loop Compensation Required Ultra-Fast Transient Response Nearly Constant Operating Frequency Intelligent Peak Current Limit Adjustable Output Voltage From 1.225 V Precision 2% Feedback Reference Frequency Adjustable to 1 MHz Adjustable Undervoltage Lockout (UVLO) Remote Shutdown Thermal Shutdown Packages: – WSON-8 – SO PowerPAD™-8 A peak current limit circuit protects against overload conditions. The undervoltage lockout (UVLO) circuit allows the input undervoltage threshold and hysteresis to be independently programmed. Other protection features include thermal shutdown and bias supply undervoltage lockout (VCC UVLO). The LM5017 device is available in WSON-8 and HSOP PowerPAD-8 plastic packages. Device Information(1) PART NUMBER 2 Applications • • • • LM5017 Smart Power Meters Telecommunication Systems Automotive Electronics Isolated Bias Supply PACKAGE BODY SIZE (NOM) SO PowerPAD (8) 4.89 mm × 3.90 mm WSON (8) 4.00 mm × 4.00 mm (1) For all available packages, see the orderable addendum at the end of the data sheet. Typical Application LM5017 CIN 2 + 4 RUV2 RON SD 3 BST VIN SW RON + 8 CBST L1 VOUT CVCC VCC UVLO FB RUV1 7 RTN 1 + 7.5V-100V VIN 6 RFB2 5 RC + RFB1 COUT 1 An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications, intellectual property matters and other important disclaimers. PRODUCTION DATA. LM5017 SNVS783H – JANUARY 2012 – REVISED DECEMBER 2014 www.ti.com Table of Contents 1 2 3 4 5 6 7 Features .................................................................. Applications ........................................................... Description ............................................................. Revision History..................................................... Pin Configuration and Functions ......................... Specifications......................................................... 1 1 1 2 3 4 6.1 6.2 6.3 6.4 6.5 6.6 6.7 4 4 4 4 5 5 6 Absolute Maximum Ratings ..................................... ESD Ratings.............................................................. Recommended Operating Conditions....................... Thermal Information ................................................. Electrical Characteristics........................................... Timing Requirements ................................................ Typical Characteristics .............................................. Detailed Description .............................................. 8 7.1 Overview ................................................................... 8 7.2 Functional Block Diagram ......................................... 8 7.3 Feature Description................................................... 9 7.4 Device Functional Modes ....................................... 13 8 Application and Implementation ........................ 14 8.1 Application Information............................................ 14 8.2 Typical Application ................................................. 14 9 Power Supply Recommendations...................... 23 10 Layout................................................................... 23 10.1 Layout Guidelines ................................................. 23 10.2 Layout Example .................................................... 23 11 Device and Documentation Support ................. 24 11.1 11.2 11.3 11.4 Documentation Support ........................................ Trademarks ........................................................... Electrostatic Discharge Caution ............................ Glossary ................................................................ 24 24 24 24 12 Mechanical, Packaging, and Orderable Information ........................................................... 24 4 Revision History Changes from Revision G (December 2013) to Revision H Page • Added package designators to pin out drawings. ................................................................................................................. 3 • Changed Thermal Information table. ..................................................................................................................................... 4 • Added D1 to Figure 12 ......................................................................................................................................................... 13 • Updated the calculation for K from 10-10 to 10-11 .................................................................................................................. 15 • Changed Series Ripple Resistor RC section to Type III Ripple Circuit ................................................................................ 16 Changes from Revision F (September 2013) to Revision G Page • Changed formatting throughout document to the TI standard ............................................................................................... 1 • Changed minimum operating input voltage from 9 V to 7.5 V in Features ........................................................................... 1 • Changed minimum operating input voltage from 9 V to 7.5 V in Typical Application ........................................................... 1 • Changed minimum operating input voltage from 9 V to 7.5 V in Pin Descriptions ............................................................... 3 • Added Maximum Junction Temperature................................................................................................................................. 4 • Changed minimum operating input voltage from 9 V to 7.5 V in Recommended Operating Conditions .............................. 4 Changes from Revision E (July 2013) to Revision F • 2 Page Added SW to RTN (100 ns transient) to Absolute Maximum Ratings ................................................................................... 4 Submit Documentation Feedback Copyright © 2012–2014, Texas Instruments Incorporated Product Folder Links: LM5017 LM5017 www.ti.com SNVS783H – JANUARY 2012 – REVISED DECEMBER 2014 5 Pin Configuration and Functions 8-Pin SO PowerPAD DDA Package Top View RTN 1 VIN 2 UVLO 3 RON 4 SO PowePAD-8 Exp Pad 8 SW 7 BST 6 VCC 5 FB 8-Pin WSON NGU Package Top View RTN 1 VIN 2 UVLO 3 RON 4 8 SW WSON-8 Exp Pad 7 BST 6 VCC 5 FB Pin Functions PIN NO. NAME I/O DESCRIPTION APPLICATION INFORMATION 1 RTN — Ground Ground connection of the integrated circuit. 2 VIN I Input Voltage Operating input range is 7.5 V to 100 V. Resistor divider from VIN to UVLO to GND programs the undervoltage detection threshold. An internal current source is enabled when UVLO is above 1.225 V to provide hysteresis. When UVLO pin is pulled below 0.66 V externally, the regulator is in shutdown mode. 3 UVLO I Input Pin of Undervoltage Comparator 4 RON I On-Time Control A resistor between this pin and VIN sets the buck switch on-time as a function of VIN. Minimum recommended on-time is 100 ns at max input voltage. 5 FB I Feedback This pin is connected to the inverting input of the internal regulation comparator. The regulation level is 1.225 V. 6 VCC O Output from the Internal High Voltage Series Pass Regulator. Regulated at 7.6 V The internal VCC regulator provides bias supply for the gate drivers and other internal circuitry. A 1.0 μF decoupling capacitor is recommended. 7 BST I Bootstrap Capacitor An external capacitor is required between the BST and SW pins (0.01-μF ceramic). The BST pin capacitor is charged by the VCC regulator through an internal diode when the SW pin is low. 8 SW O Switching Node Power switching node. Connect to the output inductor and bootstrap capacitor. EP — Exposed Pad Exposed pad must be connected to the RTN pin. Solder to the system ground plane on application board for reduced thermal resistance. Submit Documentation Feedback Copyright © 2012–2014, Texas Instruments Incorporated Product Folder Links: LM5017 3 LM5017 SNVS783H – JANUARY 2012 – REVISED DECEMBER 2014 www.ti.com 6 Specifications 6.1 Absolute Maximum Ratings (1) MIN MAX UNIT VIN, UVLO to RTN –0.3 100 V SW to RTN –1.5 VIN +0.3 V –5 VIN +0.3 V BST to VCC 100 V BST to SW 13 V SW to RTN (100 ns transient) RON to RTN –0.3 100 V VCC to RTN –0.3 13 V FB to RTN –0.3 5 V 200 °C 150 °C 150 °C Lead Temperature (2) Maximum Junction Temperature (3) Storage temperature, Tstg (1) (2) (3) –55 Absolute Maximum Ratings are limits beyond which damage to the device may occur. Recommended Operating Conditions are conditions under which operation of the device is intended to be functional. For ensured specifications and test conditions, see the Electrical Characteristics . The RTN pin is the GND reference electrically connected to the substrate. For detailed information on soldering plastic SO PowerPAD package, refer to Absolute Maximum ratings for Soldering (SNOA549). Maximum solder time not to exceed 4 seconds. High junction temperatures degrade operating lifetimes. Operating lifetime is de-rated for junction temperatures greater than 125°C. 6.2 ESD Ratings VALUE Human-body model (HBM), per ANSI/ESDA/JEDEC JS-001 V(ESD) (1) (2) Electrostatic discharge (1) UNIT ±2000 Charged-device model (CDM), per JEDEC specification JESD22C101 (2) V ±750 JEDEC document JEP155 states that 500-V HBM allows safe manufacturing with a standard ESD control process. JEDEC document JEP157 states that 250-V CDM allows safe manufacturing with a standard ESD control process. 6.3 Recommended Operating Conditions over operating free-air temperature range (unless otherwise noted) (1) VIN Voltage (1) Operating Junction Temperature (1) (2) (2) MIN MAX UNIT 7.5 100 V –40 125 °C Recommended Operating Conditions are conditions under the device is intended to be functional. For specifications and test conditions, see Electrical Characteristics . High junction temperatures degrade operating lifetimes. Operating lifetime is de-rated for junction temperatures greater than 125°C 6.4 Thermal Information LM5017 THERMAL METRIC (1) NGU DDA 8 PINS 8 PINS 41.3 41.1 RθJA Junction-to-ambient thermal resistance RθJCbot Junction-to-case (bottom) thermal resistance 3.2 2.4 ΨJB Junction-to-board thermal characteristic parameter 19.2 24.4 RθJB Junction-to-board thermal resistance 19.1 30.6 RθJCtop Junction-to-case (top) thermal resistance 34.7 37.3 ΨJT Junction-to-top thermal characteristic parameter 0.3 6.7 (1) 4 UNIT °C/W For more information about traditional and new thermal metrics, see the IC Package Thermal Metrics application report (SPRA953). Submit Documentation Feedback Copyright © 2012–2014, Texas Instruments Incorporated Product Folder Links: LM5017 LM5017 www.ti.com SNVS783H – JANUARY 2012 – REVISED DECEMBER 2014 6.5 Electrical Characteristics Typical values correspond to TJ = 25°C. Minimum and maximum limits apply over –40°C to 125°C junction temperature range, unless otherwise stated. VIN = 48 V unless otherwise stated. See (1) PARAMETER TEST CONDITIONS MIN TYP MAX 6.25 7.6 8.55 UNIT VCC SUPPLY VCC Reg VCC Regulator Output VIN = 48 V, ICC = 20 mA VCC Current Limit VIN = 48 V (2) VCC Undervoltage Lockout Voltage (VCC increasing) –40 ≤ TJ ≤ 125 26 4.15 VCC Undervoltage Hysteresis V mA 4.5 4.9 300 V mV VCC Drop Out Voltage VIN = 9 V, ICC = 20 mA 2.3 IIN Operating Current Non-Switching, FB = 3 V V 1.75 mA IIN Shutdown Current UVLO = 0 V 50 225 µA Buck Switch RDS(ON) ITEST = 200 mA, BST-SW = 7 V 0.8 1.8 Ω Synchronous RDS(ON) ITEST = 200 mA 0.45 1 Ω Gate Drive UVLO VBST − VSW Rising 3 3.6 SWITCH CHARACTERISTICS 2.4 Gate Drive UVLO Hysteresis 260 V mV CURRENT LIMIT Current Limit Threshold –40°C ≤ TJ ≤ 125°C Current Limit Response Time Time to Switch Off OFF-Time Generator (Test 1) OFF-Time Generator (Test 2) 0.7 1.02 1.3 A 150 ns FB = 0.1 V, VIN = 48 V 12 µs FB = 1.0 V, VIN = 48 V 2.5 µs REGULATION AND OVERVOLTAGE COMPARATORS FB Regulation Level Internal Reference Trip Point for Switch ON FB Overvoltage Threshold Trip Point for Switch OFF 1.2 FB Bias Current 1.225 1.25 V 1.62 V 60 nA UNDERVOLTAGE SENSING FUNCTION UV Threshold UV Rising 1.19 1.225 1.26 V UV Hysteresis Input Current UV = 2.5 V –10 –20 –29 µA Remote Shutdown Threshold Voltage at UVLO Falling 0.32 0.66 V 110 mV 165 °C 20 °C Remote Shutdown Hysteresis THERMAL SHUTDOWN Tsd Thermal Shutdown Temperature Thermal Shutdown Hysteresis (1) (2) All hot and cold limits are specified by correlating the electrical characteristics to process and temperature variations and applying statistical process control. VCC provides self bias for the internal gate drive and control circuits. Device thermal limitations limit external loading. 6.6 Timing Requirements Typical values correspond to TJ = 25°C. Minimum and maximum limits apply over –40°C to 125°C junction temperature range unless otherwise stated. VIN = 48 V unless otherwise stated. MIN NOM MAX UNIT ON-TIME GENERATOR TON Test 1 VIN = 32 V, RON = 100 k 270 350 460 ns TON Test 2 VIN = 48 V, RON = 100 k 188 250 336 ns TON Test 3 VIN = 75 V, RON = 250 k 250 370 500 ns TON Test 4 VIN = 10 V, RON = 250 k 1880 3200 4425 ns MINIMUM OFF-TIME Minimum Off-Timer FB = 0 V 144 ns Submit Documentation Feedback Copyright © 2012–2014, Texas Instruments Incorporated Product Folder Links: LM5017 5 LM5017 SNVS783H – JANUARY 2012 – REVISED DECEMBER 2014 www.ti.com 6.7 Typical Characteristics 6 Figure 1. Efficiency at 200 kHz, 10 V Figure 2. VCC vs VIN Figure 3. VCC vs ICC Figure 4. ICC vs External VCC Figure 5. TON vs VIN and RON Figure 6. TOFF (ILIM) vs VFB and VIN Submit Documentation Feedback Copyright © 2012–2014, Texas Instruments Incorporated Product Folder Links: LM5017 LM5017 www.ti.com SNVS783H – JANUARY 2012 – REVISED DECEMBER 2014 Typical Characteristics (continued) Figure 7. IIN vs VIN (Operating, Non-Switching) Figure 8. IIN vs VIN (Shutdown) Figure 9. Switching Frequency vs VIN Submit Documentation Feedback Copyright © 2012–2014, Texas Instruments Incorporated Product Folder Links: LM5017 7 LM5017 SNVS783H – JANUARY 2012 – REVISED DECEMBER 2014 www.ti.com 7 Detailed Description 7.1 Overview The LM5017 step-down switching regulator features all the functions needed to implement a low cost, efficient, buck converter capable of supplying up to 0.6 A to the load. This high voltage regulator contains 100-V, Nchannel buck and synchronous switches, is easy to implement, and is provided in thermally enhanced HSOP PowerPAD-8 and WSON-8 packages. The regulator operation is based on a constant on-time control scheme using an on-time inversely proportional to VIN. This control scheme does not require loop compensation. The current limit is implemented with a forced off-time inversely proportional to VOUT. This scheme ensures short circuit protection while providing minimum foldback. The LM5017 can be applied in numerous applications to efficiently regulate down higher voltages. This regulator is well suited for 48-V telecom and automotive power bus ranges. Protection features include: thermal shutdown, Undervoltage Lockout (UVLO), minimum forced off-time, and an intelligent current limit. 7.2 Functional Block Diagram LM5017 START-UP REGULATOR VIN VCC V UVLO 20 µA 4.5V UVLO THERMAL SHUTDOWN UVLO 1.225V SD VDD REG BST 0.66V SHUTDOWN BG REF VIN DISABLE ON/OFF TIMERS RON 1.225V SW COT CONTROL LOGIC FEEDBACK FB ILIM COMPARATOR OVER-VOLTAGE 1.62V CURRENT LIMIT ONE-SHOT RTN 8 Submit Documentation Feedback + VILIM Copyright © 2012–2014, Texas Instruments Incorporated Product Folder Links: LM5017 LM5017 www.ti.com SNVS783H – JANUARY 2012 – REVISED DECEMBER 2014 7.3 Feature Description 7.3.1 Control Overview The LM5017 buck regulator employs a control principle based on a comparator and a one-shot on-timer, with the output voltage feedback (FB) compared to an internal reference (1.225 V). If the FB voltage is below the reference the internal buck switch is turned on for the one-shot timer period, which is a function of the input voltage and the programming resistor (RON). Following the on-time the switch remains off until the FB voltage falls below the reference, but never before the minimum off-time forced by the minimum off-time one-shot timer. When the FB pin voltage falls below the reference and the minimum off-time one-shot period expires, the buck switch is turned on for another on-time one-shot period. This will continue until regulation is achieved and the FB voltage is approximately equal to 1.225 V (typ). In a synchronous buck converter, the low side (sync) FET is ‘on’ when the high side (buck) FET is ‘off’. The inductor current ramps up when the high side switch is ‘on’ and ramps down when the high side switch is ‘off’. There is no diode emulation feature in this IC, and therefore, the inductor current may ramp in the negative direction at light load. This causes the converter to operate in continuous conduction mode (CCM) regardless of the output loading. The operating frequency remains relatively constant with load and line variations. The operating frequency can be calculated as shown in Equation 1. VOUT gSW = K x RON (1) Where: K = 9 x 10–11 The output voltage (VOUT) is set by two external resistors (RFB1, RFB2). The regulated output voltage is calculated as shown in Equation 2. RFB2 + RFB1 VOUT = 1.225V x RFB1 (2) This regulator regulates the output voltage based on ripple voltage at the feedback input, requiring a minimum amount of ESR for the output capacitor (COUT). A minimum of 25 mV of ripple voltage at the feedback pin (FB) is required for the LM5017. In cases where the capacitor ESR is too small, additional series resistance may be required (RC in Figure 10). For applications where lower output voltage ripple is required the output can be taken directly from a low ESR output capacitor, as shown in Figure 10. However, RC slightly degrades the load regulation. L1 VOUT SW LM5017 RFB2 FB RC + RFB1 COUT VOUT (low ripple) Figure 10. Low Ripple Output Configuration 7.3.2 VCC Regulator The LM5017 device contains an internal high voltage linear regulator with a nominal output of 7.6 V. The input pin (VIN) can be connected directly to the line voltages up to 100 V. The VCC regulator is internally current limited to 30 mA. The regulator sources current into the external capacitor at VCC. This regulator supplies current to internal circuit blocks including the synchronous MOSFET driver and the logic circuits. When the voltage on the VCC pin reaches the undervoltage lockout (VCC UVLO) threshold of 4.5 V, the IC is enabled. An internal diode connected from VCC to the BST pin replenishes the charge in the gate drive bootstrap capacitor when SW pin is low. Submit Documentation Feedback Copyright © 2012–2014, Texas Instruments Incorporated Product Folder Links: LM5017 9 LM5017 SNVS783H – JANUARY 2012 – REVISED DECEMBER 2014 www.ti.com Feature Description (continued) At high input voltages, the power dissipated in the high voltage regulator is significant and can limit the overall achievable output power. As an example, with the input at 48 V and switching at high frequency, the VCC regulator may supply up to 7 mA of current resulting in 48 V × 7 mA = 336 mW of power dissipation. If the VCC voltage is driven externally by an alternate voltage source between 8.55 V and 14 V, the internal regulator is disabled. This reduces the power dissipation in the IC. 7.3.3 Regulation Comparator The feedback voltage at FB is compared to an internal 1.225 V reference. In normal operation, when the output voltage is in regulation, an on-time period is initiated when the voltage at FB falls below 1.225 V. The high side switch will stay on for the on-time, causing the FB voltage to rise above 1.225 V. After the on-time period, the high side switch will stay off until the FB voltage again falls below 1.225 V. During start-up, the FB voltage will be below 1.225 V at the end of each on-time, causing the high side switch to turn on immediately after the minimum forced off-time of 144 ns. The high side switch can be turned off before the on-time is over if the peak current in the inductor reaches the current limit threshold. 7.3.4 Overvoltage Comparator The feedback voltage at FB is compared to an internal 1.62 V reference. If the voltage at FB rises above 1.62 V the on-time pulse is immediately terminated. This condition can occur if the input voltage and/or the output load changes suddenly. The high side switch will not turn on again until the voltage at FB falls below 1.225 V. 7.3.5 On-Time Generator The on-time for the LM5017 device is determined by the RON resistor and is inversely proportional to the input voltage (VIN), resulting in a nearly constant frequency as VIN is varied over the operating range. The on-time for the LM5017 can be calculated using Equation 3. 10-10 x RON TON = VIN (3) See Figure 5. RON should be selected for a minimum on-time (at maximum VIN) greater than 100 ns for proper operation. This requirement limits the maximum switching frequency for high VIN. 7.3.6 Current Limit The LM5017 device contains an intelligent current limit off-timer. If the current in the buck switch exceeds 1.02 A, the present cycle is immediately terminated, and a non-resetable off-timer is initiated. The length of the off-time is controlled by the FB voltage and the input voltage VIN. As an example, when FB = 0 V and VIN = 48 V, the offtime is set to 16 μs. This condition occurs when the output is shorted and during the initial part of start-up. This VIN dependent off-time ensures safe short circuit operation up to the maximum input voltage of 100 V. In cases of overload where the FB voltage is above zero volts (not a short circuit) the current limit off-time is reduced. Reducing the off-time during less severe overloads reduces the amount of foldback, recovery time, and start-up time. The off-time is calculated from Equation 4. 0.07 x VIN Ps TOFF(ILIM) = VFB + 0.2V (4) The current limit protection feature is peak limited. The maximum average output current will be less than the peak. 7.3.7 N-Channel Buck Switch and Driver The LM5017 device integrates an N-Channel Buck switch and associated floating high voltage gate driver. The gate driver circuit works in conjunction with an external bootstrap capacitor and an internal high voltage diode. A 0.01 uF ceramic capacitor connected between the BST pin and the SW pin provides the voltage to the driver during the on-time. During each off-time, the SW pin is at approximately 0 V, and the bootstrap capacitor charges from VCC through the internal diode. The minimum off-timer, set to 144 ns, ensures a minimum time each cycle to recharge the bootstrap capacitor. 10 Submit Documentation Feedback Copyright © 2012–2014, Texas Instruments Incorporated Product Folder Links: LM5017 LM5017 www.ti.com SNVS783H – JANUARY 2012 – REVISED DECEMBER 2014 Feature Description (continued) 7.3.8 Synchronous Rectifier The LM5017 provides an internal synchronous N-Channel MOSFET rectifier. This MOSFET provides a path for the inductor current to flow when the high-side MOSFET is turned off. The synchronous rectifier has no diode emulation mode, and is designed to keep the regulator in continuous conduction mode even with light loads which would otherwise result in discontinuous operation. 7.3.9 Undervoltage Detector The LM5017 device contains a dual level undervoltage lockout (UVLO) circuit. A summary of threshold voltages and operational states is provided in Device Functional Modes . When the UVLO pin voltage is below 0.66 V, the regulator is in a low current shutdown mode. When the UVLO pin voltage is greater than 0.66V but less than 1.225 V, the regulator is in standby mode. In standby mode the VCC bias regulator is active while the regulator output is disabled. When the VCC pin exceeds the VCC undervoltage threshold and the UVLO pin voltage is greater than 1.225 V, normal operation begins. An external set-point voltage divider from VIN to GND can be used to set the minimum operating voltage of the regulator. UVLO hysteresis is accomplished with an internal 20-μA current source that is switched on or off into the impedance of the set-point divider. When the UVLO threshold is exceeded, the current source is activated to quickly raise the voltage at the UVLO pin. The hysteresis is equal to the value of this current times the resistance RUV2. If the UVLO pin is connected directly to the VIN pin, the regulator will begin operation once the VCC undervoltage is satisfied. VIN CIN 2 VIN + RUV2 LM5017 3 UVLO RUV1 Figure 11. UVLO Resistor Setting 7.3.10 Thermal Protection The LM5017 device should be operated so the junction temperature does not exceed 150°C during normal operation. An internal Thermal Shutdown circuit is provided to protect the LM5017 in the event of a higher than normal junction temperature. When activated, typically at 165°C, the regulator is forced into a low power reset state, disabling the buck switch and the VCC regulator. This feature prevents catastrophic failures from accidental device overheating. When the junction temperature falls below 145°C (typical hysteresis = 20°C), the VCC regulator is enabled, and normal operation is resumed. 7.3.11 Ripple Configuration LM5017 uses Constant-On-Time (COT) control in which the on-time is terminated by an on-timer and the off-time is terminated by the feedback voltage (VFB) falling below the reference voltage (VREF). Therefore, for stable operation, the feedback voltage must decrease monotonically, in phase with the inductor current during the offtime. Furthermore, this change in feedback voltage (VFB) during off-time must be larger than any noise component present at the feedback node. Table 1 shows three different methods for generating appropriate voltage ripple at the feedback node. Type 1 and Type 2 ripple circuits couple the ripple at the output of the converter to the feedback node (FB). The output voltage ripple has two components: 1. Capacitive ripple caused by the inductor current ripple charging/discharging the output capacitor. 2. Resistive ripple caused by the inductor current ripple flowing through the ESR of the output capacitor. Submit Documentation Feedback Copyright © 2012–2014, Texas Instruments Incorporated Product Folder Links: LM5017 11 LM5017 SNVS783H – JANUARY 2012 – REVISED DECEMBER 2014 www.ti.com Feature Description (continued) The capacitive ripple is not in phase with the inductor current. As a result, the capacitive ripple does not decrease monotonically during the off-time. The resistive ripple is in phase with the inductor current and decreases monotonically during the off-time. The resistive ripple must exceed the capacitive ripple at the output node (VOUT) for stable operation. If this condition is not satisfied unstable switching behavior is observed in COT converters, with multiple on-time bursts in close succession followed by a long off-time. Type 3 ripple method uses Rr and Cr and the switch node (SW) voltage to generate a triangular ramp. This triangular ramp is ac coupled using Cac to the feedback node (FB). Since this circuit does not use the output voltage ripple, it is ideally suited for applications where low output voltage ripple is required. See AN-1481 Controlling Output Ripple and Achieving ESR Independence in Constant On-Time (COT) Regulator Designs (SNVA166) for more details for each ripple generation method. Table 1. Ripple Configuration TYPE 1 LOWEST COST CONFIGURATION TYPE 2 REDUCED RIPPLE CONFIGURATION VOUT TYPE 3 MINIMUM RIPPLE CONFIGURATION VOUT L1 VOUT L1 L1 R FB2 Cac R FB2 RC To FB C OUT COUT R FB2 GND R FB1 GND 25 mV VOUT x ûIL(MIN) VREF Cr Cac To FB R FB1 RC > Rr RC C OUT To FB R FB1 GND C> (5) 5 gsw(RFB2||RFB1) 25 mV RC > ûIL(MIN) (6) Cr = 3300 pF Cac = 100 nF (VIN(MIN) - VOUT) x TON RrCr < 25 mV (7) 7.3.12 Soft-Start A soft-start feature can be implemented with the LM5017 using an external circuit. As shown in Figure 12, the soft-start circuit consists of one capacitor, C1, two resistors, R1 and R2, and a diode, D. During the initial start-up, the VCC voltage is established prior to the VOUT voltage. Capacitor C1 is discharged and D is thereby forward biased to pull up the FB voltage. The FB voltage exceeds the reference voltage (1.225 V) and switching is therefore disabled. As capacitor C1 charges, the voltage at node B gradually decreases and switching commences. VOUT will gradually rise to maintain the FB voltage at the reference voltage. Once the voltage at node B is less than a diode drop above FB voltage, the soft-start is finished and D is reverse biased. During the initial part of the start-up, the FB voltage can be approximated as follows. Please note that the effect of R1 has been ignored to simplify the calculation shown in Equation 8. RFB1 x RFB2 VFB = (VCC - VD) x R2 x (RFB1 + RFB2) + RFB1 x RFB2 (8) C1 is charged after the first start up. Diode D1 is optional and can be added to discharge C1 when the input voltage experiences a momentary drop to initialize the soft-start sequence. To achieve the desired soft-start, the following design guidance is recommended: (1) R2 is selected so that VFB is higher than 1.225 V for a VCC of 4.5 V, but is lower than 5 V when VCC is 8.55 V. If an external VCC is used, VFB should not exceed 5 V at maximum VCC. (2) C1 is selected to achieve the desired start-up time that can be determined from Equation 9. RFB1 x RFB2 ) tS = C1 x (R2 + RFB1 + RFB2 12 Submit Documentation Feedback (9) Copyright © 2012–2014, Texas Instruments Incorporated Product Folder Links: LM5017 LM5017 www.ti.com SNVS783H – JANUARY 2012 – REVISED DECEMBER 2014 (3) R1 is used to maintain the node B voltage at zero after the soft-start is finished. A value larger than the feedback resistor divider is preferred. Note that the effect of R1 is ignored in the previous equations. Based on the schematic shown in Figure 13, selecting C1 = 1 uF, R2 = 1 kΩ, R1 = 30 kΩ results in a soft-start time of about 2 ms. VOUT VCC C1 RFB2 R2 To FB D D1 B RFB1 R1 Figure 12. Soft-Start Circuit 7.4 Device Functional Modes Table 2. UVLO Modes UVLO VCC Regulator MODE DESCRIPTION < 0.66 V Disabled Shutdown VCC regulator disabled. Switching disabled. 0.66 V – 1.225 V Enabled Standby VCC regulator enabled Switching disabled. VCC < 4.5 V Standby VCC regulator enabled. Switching disabled. VCC > 4.5 V Operating > 1.225 V VCC enabled. Switching enabled. Submit Documentation Feedback Copyright © 2012–2014, Texas Instruments Incorporated Product Folder Links: LM5017 13 LM5017 SNVS783H – JANUARY 2012 – REVISED DECEMBER 2014 www.ti.com 8 Application and Implementation NOTE Information in the following applications sections is not part of the TI component specification, and TI does not warrant its accuracy or completeness. TI’s customers are responsible for determining suitability of components for their purposes. Customers should validate and test their design implementation to confirm system functionality. 8.1 Application Information The LM5017 device is step-down dc-dc converter. The device is typically used to convert a higher dc voltage to a lower dc voltage with a maximum available output current of 650 mA. Use the following design procedure to select component values for the LM5017 device. Alternately, use the WEBENCH® software to generate a complete design. The WEBENCH software uses an iterative design procedure and accesses a comprehensive database of components when generating a design. This section presents a simplified discussion of the design process. 8.2 Typical Application 8.2.1 Application Circuit: 12.5-V to 95-V Input and 10-V, 600-mA Output Buck Converter The application schematic of a buck supply is shown in Figure 13. For output voltage (VOUT) more than one diode drop above the maximum regulation threshold of VCC (8.55 V, see Electrical Characteristics ), the VCC pin can be connected to VOUT through a diode (D2), as shown in Figure 13, for higher efficiency and lower power dissipation in the IC. The design example below uses equations from the Feature Description with component names provided in the Typical Application. Corresponding component designators from Figure 13 are also provided for each selected value. 12.5 V±95 V VIN (TP1) SW (TP6) LM5017 2 C5 + C4 + R5 127 N 2.2 F 0.47 F GND (TP2) 4 R3 499 N 3 BST VIN SW RON UVLO R7 14 N FB EXP 220 H L1 R4 46.4 N VCC (TP4) UVLO/SD 7 0.01 F + C1 8 RTN 1 6 D2 5 + U1 C7 1 F 0 R8 VOUT (TP3) R2 0 C6 3300 pF C8 0.1 F R1 6.98 N R6 1 N + C9 22 F GND (TP5) Figure 13. Final Schematic for 12.5-V to 95-V Input, and 10-V, 600-mA Output Buck Converter 8.2.1.1 Design Requirements Selection of external components is illustrated through a design example. The design example specifications are shown in Table 3. Table 3. Buck Converter Design Specifications 14 DESIGN PARAMETERS VALUE Input voltage range 12.5 V to 95 V Output voltage 10 V Maximum Load current 600 mA Switching Frequency ≈ 225 kHz Submit Documentation Feedback Copyright © 2012–2014, Texas Instruments Incorporated Product Folder Links: LM5017 LM5017 www.ti.com SNVS783H – JANUARY 2012 – REVISED DECEMBER 2014 8.2.1.2 Detailed Design Procedure 8.2.1.2.1 RFB1, RFB2 VOUT = VFB x (RFB2/RFB1 + 1), and since VFB = 1.225 V, the ratio of RFB2 to RFB1 calculates as 7:1. Standard values are chosen with RFB2 = R1 = 6.98 kΩ and RFB1 = R6 = 1.00 kΩ. Other values could be used as long as the 7:1 ratio is maintained. 8.2.1.2.2 Frequency Selection At the minimum input voltage, the maximum switching frequency of LM5017 is restricted by the forced minimum off-time (TOFF(MIN)) as given by Equation 10. 1 - DMAX 1 - 10/12.5 = = 1 MHz gSW(MAX) = 200 ns TOFF(MIN) (10) Similarly, at maximum input voltage, the maximum switching frequency of LM5017 is restricted by the minimum TON as given by Equation 11. DMIN 10/95 gSW(MAX) = = = 1.05 MHz TON(MIN) 100 ns (11) Resistor RON sets the nominal switching frequency based on Equation 12. VOUT gSW = K x RON (12) Where: K = 9 x 10–11 Operation at high switching frequency results in lower efficiency while providing the smallest solution. For this example a conservative 225 kHz was selected, resulting in RON = 493 kΩ. A standard value for RON = R3 = 499 kΩ is selected. 8.2.1.2.3 Inductor Selection The minimum inductance is selected to limit the output ripple to 15 to 40 percent of the maximum load current. In addition, the peak inductor current at maximum load should be smaller than the minimum current limit as given in Electrical Characteristics table. The inductor current ripple is given by Equation 13. VIN - VOUT VOUT x ûIL = VIN L1 x gSW (13) The maximum ripple is observed at maximum input voltage. Substituting VIN = 95 V and ΔIL = 40 percent × IOUT (max) results in L1 = 198 μH. The next higher standard value of 220 μH is chosen. The peak-to-peak minimum and maximum inductor current ripple are 40 mA and 181 mA at the minimum and maximum input voltages respectively. The peak inductor and switch current is given by Equation 14. ûIL(MAX) ILI(peak) = IOUT + = 690 mA (14) 2 690 mA is less than the minimum current limit threshold of 0.7 A. The selected inductor should be able to withstand the maximum current limit of 1.3 A during startup and overload conditions without saturating. 8.2.1.2.4 Output Capacitor The output capacitor is selected to minimize the capacitive ripple across it. The maximum ripple is observed at maximum input voltage and is given by: ûIL COUT = 8 x gsw x ûVripple (15) Where: ΔVripple is the voltage ripple across the capacitor Submit Documentation Feedback Copyright © 2012–2014, Texas Instruments Incorporated Product Folder Links: LM5017 15 LM5017 SNVS783H – JANUARY 2012 – REVISED DECEMBER 2014 www.ti.com Assuming VIN = 95 V and substituting ΔVripple = 10 mV gives COUT = 10.1 μF. A 22-μF standard value is selected for COUT = C9. An X5R or X7R type capacitor with a voltage rating 16 V or higher should be selected. 8.2.1.2.5 Type III Ripple Circuit Type III ripple circuit as described in Ripple Configuration is chosen for this example. For a constant on-time converter to be stable, the injected in-phase ripple should be larger than the capacitive ripple on COUT. Using the type III ripple circuit equation, the target ripple will be greater than the capacitive ripple generated at the primary output if the following condition is satisfied: Cr = C6 = 3300 pF Cac = C8 = 100 nF (VIN(MIN) - VOUT ) ´ TON(VINMIN) Rr £ (25mV ´ Cr ) (16) For TON, refer to Equation 3. Ripple resistor Rr is calculated to be 57.6 kΩ. This value provides the minimum ripple for stable operation. A smaller resistance should be selected to allow for variations in TON, COUT, and other components. Rr = R4 = 46.4 kΩ is selected for this example application. 8.2.1.2.6 VCC and Bootstrap Capacitor The VCC capacitor provides charge to bootstrap capacitor as well as internal circuitry and low side gate driver. The Bootstrap capacitor provides charge to high side gate driver. The recommended value for CVCC = C7 = 1 μF. A good value for CBST = C1 = 0.01 μF. 8.2.1.2.7 Input Capacitor Input capacitor should be large enough to limit the input voltage ripple as shown in Equation 17. IOUT(MAX) CIN > 8 x gSW x ûVIN (17) Choosing a ΔVIN = 0.5 V gives a minimum CIN = 0.66 μF. A standard value of 2.2 μF is selected for CIN = C4 . The input capacitor should be rated for the maximum input voltage under all conditions. A 100-V, X7R dielectric should be selected for this design. The input capacitor should be placed directly across VIN and RTN (pin 1 and 2) of the IC. If it is not possible to place all of the input capacitor close to the IC, a 0.47-μF capacitor should be placed near the IC to provide a bypass path for the high frequency component of the switching current. 8.2.1.2.8 UVLO Resistors The UVLO resistors RFB1 and RFB2 set the UVLO threshold and hysteresis according to the relationship shown in Equation 18 and Equation 19. VIN(HYS) = IHYS x RUV2 (18) VIN (UVLO,rising) = 1.225V x ( RUV2 + 1) RUV1 (19) Where: IHYS = 20 μA Setting UVLO hysteresis of 2.5 V and UVLO rising threshold of 12 V results in RUV1 = 14.53 kΩ and RUV2 = 125 kΩ. Selecting standard values of RUV1 = R7 = 14 kΩ and RUV2 = R5 = 127 kΩ results in UVLO threshold and hysteresis of 12.4 V and 2.5 V respectively. 16 Submit Documentation Feedback Copyright © 2012–2014, Texas Instruments Incorporated Product Folder Links: LM5017 LM5017 www.ti.com SNVS783H – JANUARY 2012 – REVISED DECEMBER 2014 8.2.1.3 Application Curves Figure 14. Efficiency vs Load Current Figure 15. Frequency vs Input Voltage Figure 16. Typical Switching Waveform (Vin = 48 V, Iout = 200 mA 8.2.2 Isolated DC-DC Converter Using LM5017 An isolated supply using LM5017 is shown in Figure 17. Inductor (L) in a typical buck circuit is replaced with a coupled inductor (X1). A diode (D1) is used to rectify the voltage on a secondary output. The nominal voltage at the secondary output (VOUT2) is given by Equation 20. N VOUT2 = VOUT1 x S - VF NP (20) Where: VF is the forward voltage drop of D1 NP and NS are the number of turns on the primary and secondary of coupled inductor X1. For output voltage (VOUT1) more than one diode drop above the maximum VCC (8.55 V), the VCC pin can be diode connected to VOUT1 for higher efficiency and low dissipation in the IC. See AN-2204 (SNVA611) for a complete isolated bias design with LM5017. Submit Documentation Feedback Copyright © 2012–2014, Texas Instruments Incorporated Product Folder Links: LM5017 17 LM5017 SNVS783H – JANUARY 2012 – REVISED DECEMBER 2014 www.ti.com VOUT2 D1 + N2 X1 1:1 LM5017 BST VIN 20V-95V 0.01 µF + CBST N1 33 µH VOUT1 SW 46.4 N 1 nF Rr Cr VIN + CIN 2.2 µF COUT2 1 µF CBYP 0.47 µF + RON RUV2 127 N RON 130 N 0.1 µF RUV1 8.25 N RTN COUT1 1 µF RFB2 VCC UVLO + Cac D2 7.32 N CVCC 1 µF RFB1 1 N FB + Figure 17. Typical Isolated Application Schematic 8.2.2.1 Design Requirements DESIGN PARAMETERS VALUE Input Voltage Range 20 V – 100 V Primary Output Voltage 10 V Secondary (Isolated) Output Voltage 9.5 V Maximum Load Current (Primary + Secondary) 300 mA Maximum Power Output 3W Nominal Switching Frequency 750 kHz 8.2.2.2 Detailed Design Procedure 8.2.2.2.1 Transformer Turns Ratio The transformer turns ratio is selected based on the ratio of the primary output voltage to the secondary (isolated) output voltage. In this design example, the two outputs are nearly equal and a 1:1 turns ratio transformer is selected. Therefore, N2 / N1 = 1. If the secondary (isolated) output voltage is significantly higher or lower than the primary output voltage, a turns ratio less than or greater than 1 is recommended. The primary output voltage is normally selected based on the input voltage range such that the duty cycle of the converter does not exceed 50% at the minimum input voltage. This condition is satisfied if VOUT1 < VIN_MIN / 2. 8.2.2.2.2 Total IOUT The total primary referred load current is calculated by multiplying the isolated output load(s) by the turns ratio of the transformer as shown in Equation 21. N2 IOUT(MAX) IOUT1 IOUT2 u 0.3 A (21) N1 18 Submit Documentation Feedback Copyright © 2012–2014, Texas Instruments Incorporated Product Folder Links: LM5017 LM5017 www.ti.com SNVS783H – JANUARY 2012 – REVISED DECEMBER 2014 8.2.2.2.3 RFB1, RFB2 The feedback resistors are selected to set the primary output voltage. The selected value for RFB1 is 1 kΩ. RFB2 can be calculated using the following equations to set VOUT1 to the specified value of 10 V. A standard resistor value of 7.32 kΩ is selected for RFB2. VOUT1 = 1.225V x (1 + RFB2 ) RFB1 (22) VOUT1 - 1) x RFB1 = 7.16 k: :RFB2 = ( 1.225 (23) 8.2.2.2.4 Frequency Selection Equation 24 is used to calculate the value of RON required to achieve the desired switching frequency. VOUT1 f SW = .x RON (24) –11 Where K = 9 × 10 For VOUT1 of 10 V and fSW of 750 kHz, the calculated value of RON is 148 kΩ. A lower value of 130 kΩ is selected for this design to allow for second order effects at high switching frequency that are not included in Equation 24. 8.2.2.2.5 Transformer Selection A coupled inductor or a flyback-type transformer is required for this topology. Energy is transferred from primary to secondary when the low-side synchronous switch of the buck converter is conducting. The maximum inductor primary ripple current that can be tolerated without exceeding the buck switch peak current limit threshold (0.7 A minimum) is given by Equation 25. N2 · § 'IL1 ¨ 0.7 IOUT1 IOUT2 u u 2 0.8 A N1 ¸¹ © (25) Using the maximum peak-to-peak inductor ripple current ΔIL1 from Equation 25, the minimum inductor value is given by Equation 26. VIN(MAX) VOUT V L1 14.9 PH u OUT VIN(MAX) 'IL1 u fSW (26) A higher value of 33 µH is selected to insure the high-side switch current does not exceed the minimum peak current limit threshold. With this inductance, the inductor current ripple is ΔIL1= 0.36 A at the maximum VIN. 8.2.2.2.6 Primary Output Capacitor In a conventional buck converter the output ripple voltage is calculated as shown in Equation 27. f 'VOUT = 'IL1 x f x COUT1 (27) To limit the primary output ripple voltage ΔVOUT1 to approximately 50 mV, an output capcitor COUT1 of 1.2 µF would be required for a conventional buck. Figure 18 shows the primary winding current waveform (IL1) of a Fly-Buck™ converter. The reflected secondary winding current adds to the primary winding current during the buck switch off-time. Because of this increased current, the output voltage ripple is not the same as in conventional buck converter. The output capacitor value calculated in Equation 27 should be used as the starting point. Optimization of output capacitance over the entire line and load range must be done experimentally. If the majority of the load current is drawn from the secondary isolated output, a better approximation of the primary output voltage ripple is given by Equation 28. N2 ö æ ç I OUT2 ´ N1 ÷ ´ TON(MAX) ø DVOUT1 = è » 67mV COUT1 (28) Submit Documentation Feedback Copyright © 2012–2014, Texas Instruments Incorporated Product Folder Links: LM5017 19 LM5017 SNVS783H – JANUARY 2012 – REVISED DECEMBER 2014 www.ti.com TON(MAX) x IOUT2 x N2/N1 IL1 IOUT2 IL2 TON(MAX) x IOUT2 Figure 18. Current Waveforms for COUT1 Ripple Calculation A standard 1-µF, 25 V capacitor is selected for this design. If lower output voltage ripple is required, a higher value should be selected for COUT1 and/or COUT2. 8.2.2.2.7 Secondary Output Capacitor A simplified waveform for secondary output current (IOUT2) is shown in Figure 19. IOUT2 IL2 TON(MAX) x IOUT2 Figure 19. Secondary Current Waveforms for COUT2 Ripple Calculation The secondary output current (IOUT2) is sourced by COUT2 during on-time of the buck switch, TON. Ignoring the current transition times in the secondary winding, the secondary output capacitor ripple voltage can be calculated using Equation 29. IOUT2 x TON (MAX) 'VOUT2 = COUT2 (29) For a 1:1 transformer turns ratio, the primary and secondary voltage ripple equations are identical. Therefore, COUT2 is chosen to be equal to COUT1 (1 µF) to achieve comparable ripple voltages on primary and secondary outputs. If lower output voltage ripple is required, a higher value should be selected for COUT1 and/or COUT2. 8.2.2.2.8 Type III Feedback Ripple Circuit Type III ripple circuit as described in Ripple Configuration is required for the Fly-Buck topology. Type I and Type II ripple circuits use series resistance and the triangular inductor ripple current to generate ripple at VOUT and the FB pin. The primary ripple current of a Fly-Buck is the combination or primary and reflected secondary currents as illustrated in Figure 18. In the Fly-Buck topology, Type I and Type II ripple circuits suffer from large jitter as the reflected load current affects the feedback ripple. 20 Submit Documentation Feedback Copyright © 2012–2014, Texas Instruments Incorporated Product Folder Links: LM5017 LM5017 www.ti.com SNVS783H – JANUARY 2012 – REVISED DECEMBER 2014 VOUT L1 Rr Cac C OUT Cr R FB2 GND To FB R FB1 Figure 20. Type III Ripple Circuit Selecting the Type III ripple components using the equations from Ripple Configuration will ensure that the FB pin ripple is be greater than the capacitive ripple from the primary output capacitor COUT1. The feedback ripple component values are chosen as shown in Equation 30. Cr = 1000 pF Cac = 0.1 PF RrCr d (VIN (MIN) - VOUT) x TON 50 mV (30) The calculated value for Rr is 66 kΩ. This value provides the minimum ripple for stable operation. A smaller resistance should be selected to allow for variations in TON, COUT1 and other components. For this design, Rr value of 46.4 kΩ is selected. 8.2.2.2.9 Secondary Diode The reverse voltage across secondary-rectifier diode D1 when the high-side buck switch is off can be calculated using Equation 31. VD1 = N2 VIN N1 (31) For a VIN_MAX of 95 V and the 1:1 turns ratio of this design, a 100 V Schottky is selected. 8.2.2.2.10 VCC and Bootstrap Capacitor A 1-µF capacitor of 16 V or higher rating is recommended for the VCC regulator bypass capacitor. A good value for the BST pin bootstrap capacitor is 0.01-µF with a 16 V or higher rating. 8.2.2.2.11 Input Capacitor The input capacitor is typically a combination of a smaller bypass capacitor located near the regulator IC and a larger bulk capacitor. The total input capacitance should be large enough to limit the input voltage ripple to a desired amplitude. For input ripple voltage ΔVIN, CIN can be calculated using Equation 32. I OUT(MAX) CIN ³ 4 ´ f ´ DVIN (32) Choosing a ΔVIN of 0.5 V gives a minimum CIN of 0.2 μF. A standard value of 0.47 μF is selected for CBYP in this design. A bulk capacitor of higher value reduces voltage spikes due to parasitic inductance between the power source to the converter. A standard value of 2.2 μF is selected for for CIN in this design. The voltage ratings of the two input capacitors should be greater than the maximum input voltage under all conditions. Submit Documentation Feedback Copyright © 2012–2014, Texas Instruments Incorporated Product Folder Links: LM5017 21 LM5017 SNVS783H – JANUARY 2012 – REVISED DECEMBER 2014 www.ti.com 8.2.2.2.12 UVLO Resistors UVLO resistors RUV1 and RUV2 set the undervoltage lockout threshold and hysteresis according to Equation 33 and Equation 34. VIN (HYS) = IHYS x RUV2 (33) VIN (UVLO, rising) = 1.225V x R ( RUV2 + 1) (34) UV1 Where IHYS = 20 μA, typical. For a UVLO hysteresis of 2.5 V and UVLO rising threshold of 20 V, Equation 33 and Equation 34 require RUV1 of 8.25 kΩ and RUV2 of 127 kΩ and these values are selected for this design example. 8.2.2.2.13 VCC Diode Diode D2 is an optional diode connected between VOUT1 and the VCC regulator output pin. When VOUT1 is more than one diode drop greater than the VCC voltage, the VCC bias current is supplied from VOUT1. This results in reduced power losses in the internal VCC regulator which improves converter efficiency. VOUT1 must be set to a voltage at least one diode drop higher than 8.55 V (the maximum VCC voltage) if D2 is used to supply bias current. 8.2.2.3 Application Curves Figure 21. Steady State Waveform (VIN=48 V, IOUT1=100 mA, IOUT2=200 mA Figure 22. Step Load Response (VIN=48 V, IOUT1=0, Step Load on IOUT2=100 mA to 200 mA) Figure 23. Efficiency at 750 kHz, VOUT1=10 V 22 Submit Documentation Feedback Copyright © 2012–2014, Texas Instruments Incorporated Product Folder Links: LM5017 LM5017 www.ti.com SNVS783H – JANUARY 2012 – REVISED DECEMBER 2014 9 Power Supply Recommendations LM5017 is a power management device. The power supply for the device is any dc voltage source within the specified input range. 10 Layout 10.1 Layout Guidelines A proper layout is essential for optimum performance of the circuit. In particular, the following guidelines should be observed: 1. CIN: The loop consisting of input capacitor (CIN), VIN pin, and RTN pin carries switching currents. Therefore, the input capacitor should be placed close to the IC, directly across VIN and RTN pins and the connections to these two pins should be direct to minimize the loop area. In general it is not possible to accommodate all of input capacitance near the IC. A good practice is to use a 0.1-μF or 0.47-μF capacitor directly across the VIN and RTN pins close to the IC, and the remaining bulk capacitor as close as possible (see Figure 24). 2. CVCC and CBST: The VCC and bootstrap (BST) bypass capacitors supply switching currents to the high and low side gate drivers. These two capacitors should also be placed as close to the IC as possible, and the connecting trace length and loop area should be minimized (see Figure 24). 3. The Feedback trace carries the output voltage information and a small ripple component that is necessary for proper operation of LM5017. Therefore, care should be taken while routing the feedback trace to avoid coupling any noise to this pin. In particular, feedback trace should not run close to magnetic components, or parallel to any other switching trace. 4. SW trace: The SW node switches rapidly between VIN and GND every cycle and is therefore a possible source of noise. The SW node area should be minimized. In particular, the SW node should not be inadvertently connected to a copper plane or pour. 10.2 Layout Example RTN 1 VIN 2 UVLO 3 RON 4 8 SW 7 BST 6 VCC 5 FB CIN SO PowerPAD8 CVCC Figure 24. Placement of Bypass Capacitors Submit Documentation Feedback Copyright © 2012–2014, Texas Instruments Incorporated Product Folder Links: LM5017 23 LM5017 SNVS783H – JANUARY 2012 – REVISED DECEMBER 2014 www.ti.com 11 Device and Documentation Support 11.1 Documentation Support 11.1.1 Related Documentation • • • Absolute Maximum Ratings for Soldering (SNOA549) AN-2204 LM5017 Isolated Supply Evaluation Board (SNVA611) AN-1481 Controlling Output Ripple and Achieving ESR Independence in Constant On-Time (COT) Regulator Designs (SNVA166) 11.2 Trademarks PowerPAD, Fly-Buck are trademarks of Texas Instruments. WEBENCH is a registered trademark of Texas Instruments. All other trademarks are the property of their respective owners. 11.3 Electrostatic Discharge Caution These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam during storage or handling to prevent electrostatic damage to the MOS gates. 11.4 Glossary SLYZ022 — TI Glossary. This glossary lists and explains terms, acronyms, and definitions. 12 Mechanical, Packaging, and Orderable Information The following pages include mechanical, packaging, and orderable information. This information is the most current data available for the designated devices. This data is subject to change without notice and revision of this document. For browser-based versions of this data sheet, refer to the left-hand navigation. 24 Submit Documentation Feedback Copyright © 2012–2014, Texas Instruments Incorporated Product Folder Links: LM5017 PACKAGE OPTION ADDENDUM www.ti.com 2-Sep-2014 PACKAGING INFORMATION Orderable Device Status (1) Package Type Package Pins Package Drawing Qty Eco Plan Lead/Ball Finish MSL Peak Temp (2) (6) (3) Op Temp (°C) Device Marking (4/5) LM5017MR/NOPB ACTIVE SO PowerPAD DDA 8 95 Green (RoHS & no Sb/Br) CU SN Level-3-260C-168 HR L5017 MR LM5017MRE/NOPB ACTIVE SO PowerPAD DDA 8 250 Green (RoHS & no Sb/Br) CU SN Level-3-260C-168 HR L5017 MR LM5017MRX/NOPB ACTIVE SO PowerPAD DDA 8 2500 Green (RoHS & no Sb/Br) CU SN Level-3-260C-168 HR L5017 MR LM5017SD/NOPB ACTIVE WSON NGU 8 1000 Green (RoHS & no Sb/Br) CU SN Level-1-260C-UNLIM L5017 LM5017SDX/NOPB ACTIVE WSON NGU 8 4500 Green (RoHS & no Sb/Br) CU SN Level-1-260C-UNLIM L5017 (1) The marketing status values are defined as follows: ACTIVE: Product device recommended for new designs. LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect. NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design. PREVIEW: Device has been announced but is not in production. Samples may or may not be available. OBSOLETE: TI has discontinued the production of the device. (2) Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check http://www.ti.com/productcontent for the latest availability information and additional product content details. TBD: The Pb-Free/Green conversion plan has not been defined. Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes. Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and package, or 2) lead-based die adhesive used between the die and leadframe. The component is otherwise considered Pb-Free (RoHS compatible) as defined above. Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame retardants (Br or Sb do not exceed 0.1% by weight in homogeneous material) (3) MSL, Peak Temp. - The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature. (4) There may be additional marking, which relates to the logo, the lot trace code information, or the environmental category on the device. (5) Multiple Device Markings will be inside parentheses. Only one Device Marking contained in parentheses and separated by a "~" will appear on a device. If a line is indented then it is a continuation of the previous line and the two combined represent the entire Device Marking for that device. Addendum-Page 1 Samples PACKAGE OPTION ADDENDUM www.ti.com 2-Sep-2014 (6) Lead/Ball Finish - Orderable Devices may have multiple material finish options. Finish options are separated by a vertical ruled line. Lead/Ball Finish values may wrap to two lines if the finish value exceeds the maximum column width. Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. 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Addendum-Page 2 PACKAGE MATERIALS INFORMATION www.ti.com 2-Sep-2014 TAPE AND REEL INFORMATION *All dimensions are nominal Device Package Package Pins Type Drawing SPQ Reel Reel A0 Diameter Width (mm) (mm) W1 (mm) B0 (mm) K0 (mm) P1 (mm) W Pin1 (mm) Quadrant LM5017MRE/NOPB SO Power PAD DDA 8 250 178.0 12.4 6.5 5.4 2.0 8.0 12.0 Q1 LM5017MRX/NOPB SO Power PAD DDA 8 2500 330.0 12.4 6.5 5.4 2.0 8.0 12.0 Q1 LM5017SD/NOPB WSON NGU 8 1000 178.0 12.4 4.3 4.3 1.3 8.0 12.0 Q1 LM5017SDX/NOPB WSON NGU 8 4500 330.0 12.4 4.3 4.3 1.3 8.0 12.0 Q1 Pack Materials-Page 1 PACKAGE MATERIALS INFORMATION www.ti.com 2-Sep-2014 *All dimensions are nominal Device Package Type Package Drawing Pins SPQ Length (mm) Width (mm) Height (mm) LM5017MRE/NOPB SO PowerPAD DDA LM5017MRX/NOPB SO PowerPAD DDA 8 250 213.0 191.0 55.0 8 2500 367.0 367.0 35.0 LM5017SD/NOPB WSON LM5017SDX/NOPB WSON NGU 8 1000 210.0 185.0 35.0 NGU 8 4500 367.0 367.0 35.0 Pack Materials-Page 2 MECHANICAL DATA DDA0008B MRA08B (Rev B) www.ti.com MECHANICAL DATA NGU0008B SDC08B (Rev A) www.ti.com IMPORTANT NOTICE Texas Instruments Incorporated and its subsidiaries (TI) reserve the right to make corrections, enhancements, improvements and other changes to its semiconductor products and services per JESD46, latest issue, and to discontinue any product or service per JESD48, latest issue. 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