INTERSIL ISL3873IK

ISL3873
TM
Data Sheet
February 2001
File Number
4868.2
Wireless LAN Integrated Medium Access
Controller with Baseband Processor
New Features of the ISL3873
The Intersil ISL3873 Wireless LAN
Integrated Medium Access Controller
with Integrated Baseband Processor
is part of the PRISM® 2.4GHz radio
chip set. The ISL3873 directly interfaces with the Intersil’s IF
QMODEM (HFA3783). Adding Intersil’s RF/IF Converter
(ISL3685) and Intersil’s Power Amp (HFA3983) offers the
designer a complete end-to-end WLAN Chip Set solution.
Protocol and PHY support are implemented in firmware
thus, supporting customization of the WLAN solution.
• New Start Up Modes Allow the PCMCIA Card Information
Structure to be Initialized From a Serial EEPROM. This
Allows Firmware to be Downloaded from the Host,
Eliminating the Parallel Flash Memory Device
Firmware implements the full IEEE 802.11 Wireless LAN
MAC protocol. It supports BSS and IBSS operation under
DCF, and operation under the optional Point Coordination
Function (PCF). Low level protocol functions such as
RTS/CTS generation and acknowledgment, fragmentation
and de-fragmentation, and automatic beacon monitoring are
handled without host intervention. Active scanning is
performed autonomously once initiated by host command.
Host interface command and status handshakes allow
concurrent operations from multi-threaded I/O drivers.
Additional firmware functions specific to access point
applications are also available.
• Improvements to Debug Mode Support Tracing Execution
From on Chip Memory
The ISL3873 has on-board A/Ds and D/A for analog I and Q
inputs and outputs, for which the HFA3783 IF QMODEM is
recommended. Differential phase shift keying modulation
schemes DBPSK and DQPSK, with data scrambling
capability, are available along with Complementary Code
Keying to provide a variety of data rates. Both Receive and
Transmit AGC functions with 7-bit AGC control obtain
maximum performance in the analog portions of the
transceiver.
• USB Host Interface Supports USB V1.1 at 12Mbps.
• Firmware Can be Loaded from Serial Flash Memory
• Zero Glue Connection to 16-Bit Wide SRAM Devices
• Low Frequency Crystal Oscillator to Maintain Time and
Allow Baseband Clock Source to Power off During Sleep
Mode
• Improved Performance of Internal WEP Engine
• Programmable MBUS Cycle Extension Allows Accessing
of Slow Memory Devices Without Slowing the Clock
• Complete DSSS Baseband Processor
• RAKE Receiver with Decision Feedback Equalizer
• Processing Gain . . . . . . . . . . . . . . . . . . . . FCC Compliant
• Programmable Data Rate. . . . . . . . 1, 2, 5.5, and 11Mbps
• Ultra Small Package . . . . . . . . . . . . . . . . . . . . . 14 x 14mm
• Single Supply Operation. . . . . . . . . . . . . . . . . 2.7V to 3.6V
• Modulation Methods . . . . . . . . DBPSK, DQPSK, and CCK
• Supports Full or Half Duplex Operations
• On-Chip A/D and D/A Converters for I/Q Data (6-Bit,
22MSPS), AGC, and Adaptive Power Control (7-Bit)
• Targeted for Multipath Delay Spreads 125ns at 11Mbps,
250ns at 5.5Mbps
• Supports Short Preamble and Antenna Diversity
Applications
• PC Card Wireless LAN Adapters
Built-in flexibility allows the ISL3873 to be configured
through a general purpose control bus, for a range of
applications. The ISL3873 is housed in a thin plastic BGA
package suitable for PCMCIA board applications.
• USB PCMCIA Wireless LAN Adapters
The ISL3873 is designed to provide maximum performance
with minimum power consumption. External pin layout is
organized to provide optimal PC board layout to all user
interfaces including PCMCIA and USB.
• Wireless LAN Access Points and Bridge Products
Ordering Information
• ISA, ISA PNP WLAN Cards
PART
NUMBER
TEMP.
RANGE (oC)
PACKAGE
• PCN / Wireless PBX / Wireless Local Loop
• High Data Rate Wireless LAN Systems Targeting IEEE
802.11b Standard
• Spread Spectrum WLAN RF Modems
• TDMA or CSMA Packet Protocol Radios
• PCI Wireless LAN Cards (Using Ext. Bridge Chip)
PART
NUMBER
ISL3873IK
-40 to 85
192 BGA
ISL3873IK96
-40 to 85
Tape and Reel 1000 Units /Reel
V192.14x14
Microsoft® and Windows® are registered trademarks of Microsoft Corporation.
PRISM® is a registered trademark of Intersil Americas Inc.
PRISM and design is a trademark of Intersil Americas Inc.
1
CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures.
1-888-INTERSIL or 321-724-7143 | Intersil and Design is a trademark of Intersil Americas Inc.
Copyright © Intersil Americas Inc. 2001, All Rights Reserved
ISL3873
Simplified Block Diagram
HOST
COMPUTER
DATA
ADDRESS
CONTROL
USB
ISL3873
USB
HOST
INTERFACE
PC CARD
HOST
INTERFACE
ANT_SEL
RX_RF_AGC
1
1
AGC
CTL
MICROPROGRAMMED
MAC ENGINE
PRISM RADIO
RF SECTION
THRESH.
DETECT
7
IF
DAC
RX_IF_DET
RX_IF_AGC
RXI±
6
I ADC
DEMOD
WEP
ENGINE
ON-CHIP
ROM
ON-CHIP
RAM
MEMORY
CONTROLLER
6
Q ADC
PHY
INTERFACE
(MDI)
SERIAL
CONTROL
(MMI)
RXQ±
DATA I/O
VREF
I/O
TXI±
6
I DAC
TXQ±
MOD
6
TX
ALC
7
TX
DAC
6
TX
ADC
Q DAC
TX_IF_AGC
TX_AGC_IN
RADIO AND SYNTH
SERIAL CONTROL
MEDIUM ACCESS
CONTROLLER
ADDRESS
BASEBAND PROCESSOR
44MHz CLOCK
SOURCE †
DATA
SELECT
EXTERNAL
SRAM AND
FLASH
MEMORY
† THE ISL3873 MUST BE SUPPLIED WITH A
SEPARATE CLOCK WHEN USB IS USED.
2
ISL3873
ISL3873 Signal Descriptions
HOST INTERFACE PINS
PIN NAME
PIN I/O TYPE
DESCRIPTION
HA0-9
5V tol, CMOS, Input, 50K Pull Down
Host PC Card Address Input, Bits 0 to 9
HCE1-
5V tol, CMOS, Input, 50K Pull Up
Host PC Card Select, Low Byte
HCE2-
5V tol, CMOS, Input, 50K Pull Up
Host PC Card Select, High Byte
HD0-15
5V tol, BiDir, 2mA, 50K Pull Down
Host PC Card Data Bus, Bit 0 to 15
HINPACK-
CMOS Output, 2mA
Host PC Card I/O Decode Confirmation
HIORD-
5V tol, CMOS, Input, 50K Pull Up
Host PC Card I/O Space Read Strobe
HIOWR-
5V tol, CMOS, Input, 50K Pull Up
Host PC Card I/O Space Write Strobe
HRDY/HIREQ-
CMOS Output, 4mA
Host PC Card interrupt Request (I/O Mode), also used as PC Card
Ready (Memory Mode) output which is asserted to indicate card
initialization is complete
HOE-
5V tol, CMOS, Input, 50K Pull Up
Host PC Card Memory Attribute Space Output Enable
HREG-
5V tol, CMOS, Input, 50K Pull Up
Host PC Card Attribute Space Select
RESET
5V tol, CMOS, ST Input, 50K Pull Up
Hardware Reset. Self-asserted by internal pull-up at power-on. Clock
signal CLKIN or XTALIN must be available before negation of Reset.
Value of MD[15..0] copied to MDIR[15..0] and various control register
bits on the first MCLK following release of Reset
HSTSCHG-
CMOS Output, 4mA
Host PC Card Status Change
HWAIT-
CMOS Output, 4mA
Host Wait, asserted to indicate data transfer not complete and to force
force host bus wait states
HWE-
5V tol, CMOS Input, 50K Pull Up
Host PC Card Memory Attribute Space Write Enable
USB INTERFACE PINS
PIN NAME
PIN I/O TYPE
DESCRIPTION
USB+
CMOS BiDir, 2mA, (Also USB Transceiver)
USB, MBUS Address Bit 20, or I/O as PL5
USB-
CMOS BiDir, 2mA, (Also USB Transceiver)
USB, MBUS Address Bit 21, or I/O as PL6
USB_DETECT
Input, 5V tolerant, pull-down
Sense USB VBUS to indicate cable attachment
MEMORY INTERFACE PINS
PIN NAME
PIN I/O TYPE
DESCRIPTION
MUBE- / MA0 /
MWEH-
CMOS TS Output, 2mA
MBUS Upper Byte Enable for x16 Memory; MBUS Address Bit 0 (byte)
for x8 Memory; High Byte Write Enable for 2 x8 Memories
MA1-18
CMOS TS Output, 2mA
MBUS Address Bits 1 to 18
PL4-MA19
CMOS BiDir, 2mA
MBUS Address Bit 19
MLBE-
CMOS TS Output, 2mA, 50K Pull Up
MBUS Lower Byte Enable, or I/O as PM2
MOE-
CMOS TS Output, 2mA
Memory Output Enable
MWE- / MWEL-
CMOS TS Output, 2mA
Low (or only) Byte Memory Write Enable
RAMCS-
CMOS TS Output, 2mA
RAM Select
NVCS-
CMOS TS Output, 2mA
NV Memory Select
MD0-7
5V tol, CMOS, BiDir, 2mA, 100K Pull Up
MBUS Low Data Byte, Bits 0 to 7
MD8-15
5V tol, CMOS, BiDir, 2mA
50K Pull-Downs on MD15, MD14, MD13, MD11,
MD10, MD09
50K Pull-Ups MD12, MD08
MBUS High Data Byte, Bits 8 to 15
Default power up states are defined by pull-up and pull-down internal
resistors as shown. Device defaults to external EEPROM for boot up
mode. Using external 10K resistors, configure these pins according to
Table 4 to change power-up configuration
3
ISL3873
MAC RADIO INTERFACE AND GENERAL PURPOSE PORT PINS
PIN NAME
DESCRIPTION OF FUNCTION
(IF OTHER THAN I/O PORT)
PIN I/O TYPE
PJ4
CMOS BiDir, 2mA
PE1
PJ5
CMOS BiDir, 2mA, 50K Pull Up
LE_IF
PJ6
CMOS BiDir, 2mA
LED1
PJ7
CMOS BiDir, 2mA, 50K Pull Up
RADIO_PE
PK0
CMOS BiDir, 2mA, ST, 50K Pull Down
LE_RF
PK1
CMOS BiDir, 2mA, 50K Pull Down
SYNTHCLK
PK2
CMOS BiDir, 2mA, 50K Pull Down
SYNTHDATA
PK3
CMOS BiDir, 2mA
PA_PE
PK4
CMOS BiDir, 2mA
PE2
PK7
CMOS BiDir, 2mA
CAL_EN
PL3
CMOS BiDir, 2mA
TR_SW_BAR
PL7
CMOS BiDir, 2mA, Pull Down
TR_SW
SERIAL EEPROM PORT PINS
PIN NAME
PIN I/O TYPE
DESCRIPTION
PJ0
CMOS BiDir
SCLK, Serial Clock
PJ1
CMOS BiDir, 50K Pull Down
SD, Serial Data Out
PJ2
CMOS BiDir, 50K Pull Down
MISO, Serial Data IN
TCLKIN (CS_)
CMOS BiDir
CS_, Chip Select
CLOCKS PORT PINS
PIN NAME
PIN I/O TYPE
DESCRIPTION
CLKIN
CMOS Input, 50K Pull Down
External Clock Input to MCLK prescaler (at >= 2X Desired MCLK
Frequency, Typically 44-48MHz)
XTALIN
Analog Input
32.768kHz Crystal Input
XTALOUT
CMOS Output, 2mA
32.768kHz Crystal Output
CLKOUT
CMOS, TS Output, 2mA
Internal Clock Output (Selectable as MCLK, TCLK, or TOUT0)
BBP_CLK
Input
Baseband Processor Clock. The nominal frequency for this clock is
44MHz. This is used internally to generate divide by 2 and 4 for the
transceiver clock
BASEBAND PROCESSOR RECEIVER PORT PINS
PIN NAME
PIN I/O TYPE
DESCRIPTION
RX_IF_AGC
O
Analog drive to the IF AGC control
RX_RF_AGC
O
Drive to the RF AGC stage attenuator. CMOS digital
RX_IF_DET
I
Analog input to the receive power A/D converter for AGC control
RXI, ±
I
Analog input to the internal 6-bit A/D of the In-phase received data. Balanced differential 10+/11-
RXQ, ±
I
Analog input to the internal 6-bit A/D of the Quadrature received data. Balanced differential 13+/14BASEBAND PROCESSOR TRANSMITTER PORT PINS
PIN NAME
PIN I/O TYPE
DESCRIPTION
TX_AGC_IN
I
Input to the transmit power A/D converter for transmit AGC control
TX_IF_AGC
O
Analog drive to the transmit IF power control
TXI ±
O
TX Spread baseband I digital output data. Data is output at the chip rate. Balanced differential 23+/24-
TXQ ±
O
TX Spread baseband Q digital output data. Data is output at the chip rate. Balanced differential
29+/30-
4
ISL3873
MISC CONTROL PORT PINS
PIN NAME
PIN I/O TYPE
DESCRIPTION
ANTSEL
O
The antenna select signal changes state as the receiver switches from antenna to
antenna during the acquisition process in the antenna diversity mode. This is a
complement for ANTSEL (pin 40) for differential drive of antenna switches
ANTSEL
O
The antenna select signal changes state as the receiver switches from antenna to
antenna during the acquisition process in the antenna diversity mode. This is a
complement for ANTSEL (pin 39) for differential drive of antenna switches
TestMode
I/O
Factory level test pin. This pin must be pulled low with a 10K resistor.
CompCap1
I
Compensation Capacitor
CompCap2
I
Compensation Capacitor
CompRes1
I
Compensation Resistor
CompRes2
I
Compensation Resistor
DBG(0-4)
I/O
Debug factory test signals. Do not connect
POWER PORT PINS
PIN NAME
PIN I/O TYPE
VDDA
DESCRIPTION
Power
DC Power Supply 2.7 - 3.6V (Not Hardwired Together on Chip)
VDD
Power
DC Power Supply 2.7 - 3.6V
SUPPLY5V
Power
5V Tolerant DC Power Supply
VSSA
Ground
Analog Ground
Vsub
Ground
Analog Ground
GND
Ground
Digital Ground
VREF
Input
Voltage Reference for A/D’s and D/A’s
IREF
Input
Current Reference for internal ADC and DAC devices. Requires 12K resistor to ground.
ST = Schmitt Trigger (Hysteresis), TS = Three-State. Signals ending with “-” are active low.
ISL3873 PIN NUMBER ASSIGNMENTS
SIGNAL NAME
PIN NUMBER
SIGNAL NAME
PIN NUMBER
SIGNAL NAME
PIN NUMBER
SIGNAL NAME
A1
PIN NUMBER
NC
C7
HD4
F4
MA5
K16
VDD
A2
MA10
C8
HD6
F13
HD9
A3
MA13
C9
HD14
F14
HD10
L1
MD8
A4
MA16
C10
HD11
F15
HA2
L2
MD7
A5
GND
C11
HD7
F16
HA1
L3
MD10
A6
PL4_MA19
C12
HA7
L4
MD9
A7
DBG2
C13
GND
G1
MD12
L13
GND
A8
VDD
C14
DBG3
G2
MD14
L14
RX_RF_AGC
A9
HD3
C15
NC
G3
VDD
L15
ANT_SEL
A10
HCE2
C16
RESET
G4
MA2
L16
ANT_SEL
A11
GND
G13
GND
A12
HD15
D1
MA3
G14
HSTSCHG
M1
MD5
A13
HA9
D2
MA8
G15
HD0
M2
VDD
A14
VDD
D3
MA7
G16
BBP_CLK
M3
GND
A15
HA6
D4
MA14
M4
MD6
A16
NC
D5
MA17
H1
VDD
M13
VDDA
D6
DBG0
H2
MLBE
M14
COMPCAP1
D7
GND
H3
MD11
M15
GND
B1
VDD
5
ISL3873
ISL3873 PIN NUMBER ASSIGNMENTS (CONTINUED)
PIN NUMBER
B2
SIGNAL NAME
PIN NUMBER
SIGNAL NAME
PIN NUMBER
SIGNAL NAME
PIN NUMBER
SIGNAL NAME
NC
D8
HD5
H4
MD13
M16
VDD
B3
MA9
D9
HIREQ
H13
HD2
B4
MA12
D10
HIOWR
H14
HD1
N1
MD4
B5
VDD
D11
HOE
H15
HA0
N2
MD0
H16
HD8
B6
MA18
D12
NC
B7
DBG1
D13
HA5
N3
MD3
N4
MD2
B8
HD12
D14
HWAIT
J1
XTALIN
N5
NC
B9
HCE1
D15
SUPPLY5V
J2
XTALOUT
N6
PJ7
(RADIO_PE)
B10
VDD
D16
HREG
J3
RAMCS
N7
PK2
(SYNTHDATA)
B11
HIORD
B12
HA8
J4
NVCS
N8
VDDA
E1
GND
J13
USB_DET
N9
VSSA
B13
HWE
E2
MA4
J14
VDD
N10
VSUB
B14
HA4
E3
GND
J15
USB-
N11
VDD
B15
NC
E4
NC
J16
USB+
N12
IREF
B16
DBG4
E13
HA3
N13
VSSA
E14
VDD
K1
CLKIN
N14
NC
C1
MA6
E15
HINPACK
K2
MOE
N15
RX_IF_AGC
C2
NC
E16
GND
K3
MWEL
N16
TX_IF_AGC
C3
MA11
K4
GND
C4
MA15
F1
K13
TESTMODE
C5
CLKOUT
F2
MA1
K14
GND
C6
HD13
F3
MWEH_MA0
K15
GND
P1
MD1
R1
PJ1
(SDATA)
T1
PJ0
(SCLK)
P2
PJ2
(MISO)
R2
NC
T2
VDD
P3
TCLKIN
R3
NC
T3
PJ6
(LED1)
P4
PJ5
(LE_IF)
R4
PJ4
(PE1)
T4
PK1
(SYNTHCLK)
P5
GND
R5
PK0
(LE_RF)
T5
PK4
(PE2)
P6
PL7
(TR_SW)
R6
PK3
(PA_PE)
T6
PL3
(TR_SW_BAR)
P7
PK7
(CAL_EN)
R7
RXI+
T7
RXI-
P8
VDDA
R8
VDDA
T8
VDDA
P9
GND
R9
RXQ+
T9
RXQ-
P10
VSUB
R10
RX_IF_DET
T10
TX_AGC_IN
P11
VREF
R11
VDDA
T11
VSSA
P12
VDDA
R12
TXI+
T12
TXI-
P13
COMPRES2
R13
COMCAP2
T13
VSSA
P14
NC
R14
TXQ+
T14
TXQ-
P15
NC
R15
NC
T15
COMPRES1
P16
NC
R16
NC
T16
NC
6
MD15
ISL3873
Absolute Maximum Ratings
Thermal Information
Supply Voltage, VCC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3.6V
Input, Output or I/O Voltage . . . . . . . . . . . . GND -0.5V to VCC +0.5V
ESD Classification . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Class 2
Thermal Resistance (Typical, Note 1)
θJA (oC/W)
BGA Package. . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
55
Maximum Storage Temperature Range . . . . . . . . . . -65oC to 150oC
Maximum Junction Temperature . . . . . . . . . . . . . . . . . . . . . . .100oC
Maximum Lead Temperature (Soldering 10s) . . . . . . . . . . . . .300oC
(Lead Tips Only)
Operating Conditions
Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . +3.3V
Ambient Temperature Range . . . . . . . . . . . . . . . . . . . -40oC to 85oC
CAUTION: Stresses above those listed in “Absolute Maximum Ratings” may cause permanent damage to the device. This is a stress only rating and operation of the
device at these or any other conditions above those indicated in the operational sections of this specification is not implied.
NOTE:
1. θJA is measured with the component mounted on an evaluation PC board in free air.
DC Electrical Specifications
PARAMETER
SYMBOL
MIN
TYP
MAX
UNITS
-
-
175
mA
VCC = Max, Input = 0V or VCC
-10
1
10
mA
IO
VCC = Max, Input = 0V or VCC
-10
1
10
mA
Power Supply Current
ICCOP
Input Leakage Current
II
Output Leakage Current
TEST CONDITIONS
VCC = 3.6V, CLK Frequency 44MHz
Logical One Input Voltage
VIH
VCC = Max, Min
0.7VCC
-
-
V
Logical Zero Input Voltage
VIL
VCC = Min, Max
-
-
0.3V
V
Logical One Output Voltage
VOH
IOH = -1mA, VCC = Min
0.9VCC
-
-
V
Logical Zero Output Voltage
VOL
IOL = 2mA, VCC = Min
-
0.1
0.1VCC
V
Input Capacitance
CIN
CLK Frequency 1MHz. All measurements
referenced to GND. TA = 25oC
-
5
10
pF
COUT
CLK Frequency 1MHz. All measurements
referenced to GND. TA = 25oC
-
5
10
pF
Output Capacitance
NOTE: All values in this table have not been measured and are only estimates of the performance at this time.
AC Electrical Specifications
PARAMETER
SYMBOL
MIN
TYP
MAX
UNITS
tCYC
20
20.8
200
ns
High Period
tH1
10
10.4
-
-
Low Period
tL1
10
10.4
-
-
CLOCK SIGNAL TIMING
OSC Clock Period (Typ. 44MHz)
EXTERNAL MEMORY READ INTERFACE
MOE-Setup Time from RAMCS_
tS1
0
-
-
ns
MOE_Setup Time from MA (17..0)
tS2
0
-
-
ns
MA (17..1) Hold Time from MOE_ Rising Edge
tH1
20
-
-
ns
RAMCS_ Hold from MOE_ Rising Edge
tH2
20
-
-
ns
MD (15..0) Enable from MOE_ Falling
tE1
5
-
-
ns
MO (15..0) Disable from MOE_ Rising Edge
tD1
-
-
100
ns
tS3
0
0
0
ns
EXTERNAL MEMORY WRITE INTERFACE
MA (17..0) Setup to MWE_ Falling Edge
RAMCS_ Setup to MWE
tS4
0
-
-
ns
MA (17..0) Hold from MWE_ Rising Edge
tH3
15
-
-
ns
RAMCS _ Hold from MWE_ Rising Edge
tH4
15
-
-
ns
MD (15..0) Setup to MWE_ Rising Edge
tS5
40
-
-
ns
MD (15..0) Hold from MWE_ Rising Edge
tH5
15
-
-
ns
tCYC
83
-
4,000
ns
SYNTHESIZER
SYNTHCLK(PK1) Period
7
ISL3873
AC Electrical Specifications
(Continued)
SYMBOL
MIN
TYP
MAX
UNITS
SYNTHCLK(PK1) Width Hi
PARAMETER
tH1
tCYC /2 - 10
-
tCYC /2 + 10
ns
SYNTHCLK(PK1) Width Lo
tL1
tCYC /2 - 10
-
tCYC /2 + 10
ns
SERIAL PORT
SYNTHCLK(PK1) Clock Period
tCYC
83ns
-
4000
ns
tH1 , tL1
tCYC/2 -10
-
tCYC/2 + 10
ns
tCD
-
10
-
ns
tDRS
15
-
-
ns
Hold Time of SYTHNDATA(PK2) Read from SYTHNCLK(PK1) Falling Edge
tDRH
0
-
-
Hold Time of SYTHNDATA(PK2) Write from SYTHNCLK(PK1) Falling Edge
tDWH
0
-
-
Low Width
Delay from Clock Falling Edge to SPCSx, SPAS, SPREAD,
SYNTHDATA(PK2) Outputs
Setup Time of SYTHNDATA(PK2) Read to SYTHNCLK(PK1) Falling Edge
SYSTEM INTERFACE - PC CARD IO READ 16
Data Delay After HIORD-
tDIORD
-
-
100
Data Hold Following HIORD-
tHIORD
0
-
-
ns
ns
HIORD- Width Time
tWIORD
165
-
-
ns
Address Setup Before HIORD-
tSUA
70
-
-
ns
Address Hold Following HIORD-
tHA
20
-
-
ns
HCE(1,2)- Setup Before HIORD-
tSUCE
5
-
-
ns
HCE(1,2)- Hold After HIORD-
tHCE
20
-
-
ns
HREG- Setup Before HIORD-
tSUREG
5
-
-
ns
HREG- Hold Following HIORD-
tHREG
0
-
-
ns
HINPACK- Delay Falling from HIORD-
tDFINPACK
0
-
45
ns
HINPACK- Delay Rising from HIORDN
dDRINPACK
30
-
45
ns
HWAIT-
tDFWT
-
-
35
ns
Data Delay from HWAIT- Rising
tDRWT
-
-
0
ns
HWAIT- Width Time
tWWT
-
-
12,000
ns
tSUIOWR
30
-
92
ns
SYSTEM INTERFACE - PC CARD IO WRITE 16
Data Setup Before HIOWRData Hold Following HIOWR-
tHIOWR
20
-
-
ns
HIOWRN- Width Time
tWIOWR
165
-
-
ns
tSUA
70
-
-
ns
Address Hold Following HIOWR-
tHA
20
-
-
ns
HCE(1,2)- Setup Before HIOWR-
tSUCE
5
-
-
ns
Address Setup Before HIOWR-
HCE(1,2)- Hold Following HIOWR-
tHCE
20
-
-
ns
HREG- Setup Before HIOWR-
tSUREG
5
-
-
ns
HREG- Hold Following HIOWR-
tHREG
0
-
-
ns
HWAIT- Delay Falling from HIOWR-
tDFWT
-
-
35
ns
HWAIT- Width Time
tWWT
-
-
12,000
ns
tDRIOWR
0
-
-
ns
0.25
0.50
1.0
V
Input Bandwidth (-0.5dB)
-
20
-
MHz
Input Capacitance
-
5
-
pF
Input Impedance (DC)
5
-
-
kΩ
FS (Sampling Frequency)
-
-
22
MHz
HIOWRN High from HWAIT- High
BASEBAND SIGNALS
Full Scale Input Voltage (VP-P)
8
ISL3873
Waveforms
ADDRESS
MA(17..1)
tH1
RAMCS_
tH2
tS1
MOE_
tS2
tD1
tE1
MD(15..0)
FIGURE 1. EXTERNAL MEMORY READ TIMING
ADDRESS
MA(17..1)
tH3
RAMCS_
tS4
tH4
MWE_
tH5
tS3
MD(15..0)
tS5
FIGURE 2. EXTERNAL MEMORY WRITE TIMING
SYNTHCLK
tH1
SYNLE
SPCSPWR
tD3
tCYC
tD1
SYNTHDATA
tL1
tD2
D[n]
D[n -1]
D[n -2]
D[2]
FIGURE 3. SYNTHESIZER
9
D[1]
D[0]
ISL3873
Waveforms
(Continued)
HA[15:0]
tSUREG
tHREG
HREGISUCE
tHCE
HCE(1, 2) tWIORD
tHA
tDIORD
HIORDtSUA
tDRINPACK
tDFINPACK
HINPACK-
HWAITtWWT
tDFWT
tDRWT
tHIORD
HD[15:0]
FIGURE 4. PC CARD IO READ 16
HA[15:0]
tHREG
tSUREG
HREGN-
tHCE
tSUCE
HCE (1, 2) tSUA
tWIOWR
tHA
HIOWRtDRINPACK
tDRIOWR
HWAIT-
tDFWT
tSUIOWR
tWWT
HD[15:0]
FIGURE 5. PC CARD IO WRITE 16
10
tHIOWR
ISL3873
I
ISL3873 MAC System Overview
ISL3873
FLASH
128Kx8
MD0..15
MD0..7
MA1..17
MA0..16
NVCS_
CS_
MOE_
OE_
SRAM
128Kx8
SRAM
128Kx8
MD0..7
MA1..17
OE_
MD8..15
MWEL_
WE_
MA1..17
MA0/MWEH_
CS_
OE_
WE_
CS_
RAMCS_
FIGURE 6. 8-BIT MEMORY INTERFACE REQUIREMENTS FOR ISL3873
FLASH
128Kx16
ISL3873
MA1..17
ADDR(0..16)
MD0..15
DATA(0..15)
NVCS-
CEOE-
MA0/MWEH-
WE
SRAM
128Kx16
ADDR(0..16)
DATA(0..15)
UBMLBE-
LB-
RAMCS-
CE-
MOE-
OE
MWEL-
WE
FIGURE 7. 16-BIT MEMORY INTERFACE REQUIREMENTS FOR ISL3873
11
ISL3873
LARGE SERIAL EEPROM
SMALL SERIAL EEPROM
PULLUP
MISO (PJ2)
AO
SD (PJ1)
SI
ISL3873
SCLK (PJ0)
ISL3873
SO
SCK
PULLUP
CS# (TCLKIN)
SDA
SCLK (PJ0)
SCL
RESET#
CS# (TCLKIN)
CS
45DB011
A2
WP
WP#
PULLUP
A1
24C08 (NOTE)
NOTE: Must operate at 400kHz AT 3.3VDC
FIGURE 8. SERIAL EEPROM INTERFACE
External Memory Interface
The ISL3873 provides separate external chip selects for
code space and data storage space. Code space is
accessible as data space through an overlay mechanism,
except for an internal ROM. Refer to Figures 6, 7 and 8 for
ISL3873 memory configuration detail examples.
The maximum possible memory space size is 4Mbytes. If
USB is the host interface, this is reduced to 1Mbyte.
Most of the data store space is reserved for storage of
received and transmitted data, with some areas reserved for
use by firmware. However, a portion of the data store may be
allocated as code store. This permits higher speed
instruction execution, by using fast RAMs, than is possible
from Flash memories. The maximum size of this overlay is
the full code space address range, 128Kbytes, and is
allocated in independent sections of 16KBytes each, on
16Kbyte boundaries, ranging from the highest address of the
actual physical memory space and extending down.
Mapping code execution to RAM requires the RAM to have
code written into it. Typically, this is done by placing code in a
non-volatile memory such as a Flash in the code space. At
initialization, the code in the non-volatile memory transfers itself
to RAM, maps the appropriate blocks of the code space to the
RAM, and then branches to begin execution from RAM. This
allows low cost, slow Flash devices to hold an entire code
image, which can be executed much faster from RAM. If code
is not placed in an external non-volatile memory as described
here, it must be transferred to the RAM via the Host Interface.
Slow memories are not dynamically sensed. Following reset,
the instruction clock operates with a slower cycle while the
Flash is copied to RAM. Once code has been copied from
Flash to RAM, execution transfers to RAM and the clock is
raised to the normal operating frequency.
As mentioned above, it is feasible to operate without a code
image in a non-volatile memory. In such a system, the
12
firmware must be downloaded to RAM through the host
interface before operation can commence.
The external SRAM memory must be organized in a 16-bit
width to provide adequate performance to implement the
802.11 protocol at 11Mb/s rates. Systems designed for lower
performance applications may be able to use 8-bit wide
memory.
The minimum external memory is 128Kbytes of SRAM,
organized 8 or 16 bits wide. Typical applications, including
802.11 station designs, use 256Kbytes organized 128K x 16.
An access point application could make use of the full address
space of the device with 4Mbytes organized a 2M x 16.
The ISL3873 supports 8 or 16-bit code space, and 8 or 16-bit
data space. Code space is typically populated with the lease
expensive Flash memory available, usually an 8-bit device.
Data space is usually populated with high-speed RAMs
configured as a 16-bit space. This mixing of 8/16 bit spaces is
fully supported, and may be done in any combination desired
for code and data space.
The ISL3873 supports direct control of single chip 16-bit
wide SRAMs with high/low byte enables, as well as direct
control of a 16-bit space constructed from 8-bit wide SRAMs.
The type of memory configuration is specified via the
appropriate MD pin, sensed when the ISL3873 is reset.
ISL3873 pin MUBE-/MA0/MWEH- functions as Address 0 for
8-bit access, (such as Flash) as MWEH (High Byte Write
Enable) when two x8 memories are configured as a single
x16 space, and as the upper Byte Enable when a single x16
memory is used. No external logic is required to generate
the required signals for both types of memory configurations,
even when both exist together; all that is required is for the
ISL3873 code to configure the ISL3873 memory controller to
generate the proper signals for the particular address space
being accessed.
ISL3873
For 8-bit spaces, the ISL3873 dynamically configures pin
MUBE-/MA0/MWEH- cycle-by-cycle as the address LSB.
MWEL-/MWE- is the only write control, and MOE- is the read
output enable.
HA[9:6] are ignored when the internal HAMASK register is
set to the defaults used by the standard firmware. During
attribute memory accesses HA[9:1] are used.
For 16-bit spaces constructed from 8-bit memories, the
ISL3873 dynamically configures pin MUBE-/MA0/MWEHcycle-by-cycle as the high byte write enable, MWEL- as the low
write enable signal, and MOE- as the read output enable.
The host interface is primarily designed for word accesses,
although all byte access modes are fully supported. See
HCE1-, HCE2- for a further description. Note that attribute
memory is specified for and operates with even bytes accesses
only.
For 16-bit spaces constructed from single-chip x16
memories (such as SRAMs), the ISL3873 dynamically
configures pin MUBE_/MA0/MWEH- cycle-by-cycle as the
upper byte enable. Pin MLBE- is connected as the low byte
enable, MWEL-/MWE- is the write control, and MOE- is the
read output enable.
These memory implementations require no external logic.
The memory spaces may each be constructed from any type
of memory desired. The only restriction is that a single
memory space must be constructed from the same type of
memory; for example, data space may not use both x8 and
x16 memories, it must be all x8, or all x16. This restriction
does not apply across memory spaces; e.g., code space may
use a x8 memory and data space a single x16 memory, or
code space two x8 memories and data space a single x8
memory.
Serial EEPROM Interface
The ISL3873 contains a small on-chip ROM firmware which
was added to allow the CIS or CIS plus firmware image to be
transferred from an off-chip serial non-volatile memory device
to RAM after a system reset. This allows a system configuration
without a parallel Flash device. The operating frequency of the
serial port is 400kHz with a voltage of 3.3V. Refer to Figure 8 for
additional details on configuring the serial memory to the
ISL3873. The Power On Reset Configuration section in this
document provides additional details on memory selection and
control after a Reset condition.
PC Card Interface
PC Card Physical Interface
The Host interface is compatible to the PC Card 95 Standard
(PCMCIA v2.1). The ISL3873 Host Interface pins connect
directly to the correspondingly named pins on the PC Card
connector with no external components (other than resistors)
required. The ISL3873 operates as an I/O card using less
than 64 octet locations. Reads and writes to internal registers
and buffer memory are performed by I/O accesses. Attribute
memory (256 octets) is provided for the CIS table which is
located in external memory. Common memory is not used.
The following describes specific features of various pins:
HD[15:0]
HCE1-, HCE2The PC Card cycle type and width are controlled with the CE
signals. Word and Byte wide accesses are supported, using
the combinations of HCE1-, HCE2-, and HA0 as specified in
the PC Card standard.
HWE-, HOEHOE- and HWE- are only used to access attribute memory.
Common Memory, as specified in the PC Card standard, is
not used in the ISL3873. HOE- is the strobe that enables an
attribute memory read cycle. HWE- is the corresponding
strobe for the attribute memory write cycle. The attribute
space contains the Card Information Structure (CIS) as well
as the Function Configuration Registers (FCR).
HIORD-, HIOWRHIORD- and HIOWR- are the enabling strobes for register
access cycles to the ISL3873. These cycles can only be
performed once the initialization procedure is complete and
the ISL3873 has been put into IO mode.
HREGThis signal must be asserted for I/O or attribute cycles. A
cycle where HREG- is not asserted will be ignored as the
ISL3873 does not support common memory.
HINPACKThis signal is asserted by the ISL3873 whenever a valid I/O
read cycle takes place. A valid cycle is when HCE1-, HCE2-,
HREG-, and HIORD- are asserted, once the initialization
procedure is complete.
HWAITWait states are inserted in accesses using HWAIT-. The host
interface synchronizes all PC Card cycles to the internal
ISL3873 clock. The following wait states should be expected:
Direct Read or Write to Hardware Register
• 1/2 to 1 MCLK assertion of HWAIT- for internal
synchronization.
Write to Memory Mapped Register, Buffer Access Path,
or Attribute Space (Post-Write)
HA[9:0]
• The data required for the write cycle will be latched and
therefore only the synchronizing wait state will occur.
Decoding of the system address space is performed by the
HCEx-. During I/O accesses HA[5:0] decode the register.
• Until the queued cycle has actually written to the memory,
any subsequent access by the Host will result in a WAIT.
13
ISL3873
• WAIT will assert until the memory arbitration and access
have completed.
Note: All register cycles, including hardware registers, incur
a short wait state on the PC Card bus to insure the host
cycle is synchronized with the ISL3873's internal MCLK.
Buffer Access Paths, BAP0 and BAP1
Memory Mapped Registers in Data RAM (MM)
• An internal Pre-Read cycle to memory is initiated by a
host Buffer Read cycle, after the internal address pointer
has auto-incremented. If the next host cycle is a read to
the same buffer, the data will be available without a
memory arbitration delay.
• 1 to 1 correspondence.
Read to Attribute Space and Memory Mapped Registers
• A single register holds the pre-read data. Thus, any read
access to any other memory-mapped register (or the other
buffer access path) will result in the pre-read data
becoming invalidated.
• Requires memory arbitration, since registers are actually
locations in ISL3873 memory.
• Attribute memory access is mapped into RAM as Baseaddress + 0x400.
• AUX port provides host access to any location in ISL3873
RAM (reserved).
Buffer Access Path (BAP)
• If another read cycle has invalidated the pre-read, then a
memory arbitration delay will occur on the next buffer
access path read cycle.
• No 1 to 1 correspondence between register address and
memory address (due to indirect access through buffer
address pointer registers).
HIREQ-
• Auto increment of pointer registers after each access.
Immediately after reset, the HIREQ- signal serves as the
RDY/BSY (per the PC Card standard). Once the ISL3873
firmware initialization procedure is complete, HIREQ- is
configured to operate as the interrupt to the PC Card socket
controller. Both Level Mode and Pulse Mode interrupts are
supported. By default, Level mode interrupts are used, so
the interrupt source must be specifically acknowledged or
disabled before the interrupt will be removed.
RESET
When reset is de-asserted, the CIS table is initialized and,
once complete, HIREQ- is set high (HIREQ- acts as
RDY/BSY from reset and is set high to indicate the card is
ready for use). The CIS table resides in Flash memory and is
copied to RAM during firmware initialization. The host
system can then initialize the card by reading the CIS
information and writing to the configuration register.
ISA PNP
The ISL3873 can be connected to the ISA bus and operate
in a Plug and Play environment with an additional chip such
as the Fujitsu MB86703, Texas Instruments TL16PNP200A,
or Fairchild Semiconductor NM95MS15. See the Application
Note AN9874, “ISA Plug and Play with the HFA3841” for
more details.
Register Interface
The logical view of the ISL3873 from the host is a block of 32
word wide registers. These appear in IO space starting at
the base address determined by the socket controller. There
are three types of registers.
Hardware Registers (HW)
• 1 to 1 correspondence between addresses and registers.
• No memory arbitration delay, data transfer directly to/from
registers.
• AUX base and offset are write-only, to set up access
through AUX data port.
14
• Require memory arbitration since buffers are located in
ISL3873 memory.
• Buffer access may incur additional delay for Hardware
Buffer Chaining.
Buffer Access Paths
The ISL3873 has two independent buffer access paths, which
permits concurrent read and write transfers. The firmware
provides dynamic memory allocation between Transmit and
Receive, allowing efficient memory utilization. On-the-fly
allocation of (128-byte) memory blocks as needed for reception
wastes minimal space when receiving fragments. The ISL3873
hides management of free memory from the driver, and allows
fast response and minimum data copying for low latency. The
firmware provides direct access to TX and RX buffers based on
Frame ID (FID). This facilitates Power Management queuing,
and allows dynamic fragmentation and de-fragmentation by the
controller. Simple Allocate/De-allocate commands ensure low
host CPU overhead for memory management.
Hardware buffer chaining provides high performance while
reading and writing buffers. Data is transferred between the
host driver and the ISL3873 by writing or reading a single
register location (the Buffer Access Path, or BAP). Each
access increments the address in the buffer memory.
Internally, the firmware allocates blocks of memory as needed
to provide the requested buffer size. These blocks may not be
contiguous, but the firmware builds a linked list of pointers
between them. When the host driver is transferring data
through a buffer access path and reaches the end of a
physical memory block, hardware in the host interface follows
the linked list so that the buffer access path points to the
beginning of the next memory block. This process is
completely transparent to the host driver, which simply writes
or reads all buffer data to the same register. If the host driver
attempts to access beyond the end of the allocated buffer,
subsequent writes are ignored, and reads will be undefined.
ISL3873
FID
BUFFER DESCRIPTOR
ACCESS (FIRMWARE)
ALLOCATE/
DEALLOCATE
REQUEST
BLOCK
BUFFER
MEMORY
VIRTUAL
FRAME BUFFER
STATUS
A
OFFSET CENTER
OFFSET
HEADER
HOST
BUS
DATA PORT
PRE-READ/
POST-WRITE
D
DATA
FIGURE 9. BLOCK DIAGRAM OF A BUFFER ACCESS PATH
USB Port
The USB interface implemented in the ISL3873 complies
with the Universal Serial Bus Specification Revision 1.1.
dated September 23, 1998, which is available from the USB
Implementers’ Forum at http://www.usb.org/.
The USB supports 4 endpoints.
• One Communications Class control endpoint for interface
management;
• One Communications Class interrupt endpoint for
signalling interrupts to the host; and,
• Two Bulk endpoints for transfer of encapsulated NDIS
functions to and from the host.
The USB along with USB support firmware provides an
alternate host interface for attaching an 802.11{b} WLAN
adapter to a host computer. This interface does not provide
“wireless USB” where USB packets are sent on the wireless
medium due to timing constraints in the USB protocol.
USB+ and USB- are the differential pair signals provided for
the user. These signals are capable of directly driving a USB
cable.
USB_DETECT is a 5V tolerant input to the ISL3873 device.
It is used to signal the MAC processor that a USB cable is
attached to the unit.
Complete details on the USB firmware for controlling this
port can be obtained by contacting the factory directly.
Power Sequencing
The ISL3873 provides a number of firmware controlled port
pins that are used for controlling the power sequencing and
other functions in the front end components of the radio.
Packet transmission requires precise control of the radio.
Ideally, energy at the antenna ceases after the last symbol of
information has been transmitted. Additionally, the
15
transmit/receive switch must be controlled properly to protect
the receiver. It's also important to apply appropriate
modulation to the PA while it's active.
Signaling sequences for the beginning and end of normal
transmissions are illustrated in Figure 10. Table 1 lists
applicable delays associated with these control signals.
A transmission begins with PE2 as shown in Figure 10. Next,
the transmit/receive switch is configured for transmission via
the differential pair TR_SW and TR_SW_BAR. This is
followed by a transmit enable (TX_ENABLE) to the
Baseband processor inside the ISL3873. This enable
activates the transmit state machine in the BBP. Lastly,
PA_PE activates the PA. Delays for these signals related to
the initiation of transmission are referenced to PE2.
Immediately after the final data bit has been clocked out of the
MAC the Baseband processor is disabled. The MAC then waits
for a control signal (TX_READY) from the Baseband processor
to go inactive, signaling that the BBP has modulated the final
information-rich symbol. It then immediately de-asserts PA_PE
followed by placing the transmit/receive switch in the receive
position and ending with PE2 going high. Delays for these
signals related to the termination of transmission are
referenced to the rising edge of PE2.
TABLE 1. TRANSMIT CONTROL TIMING SPECIFICATIONS
PARAMETER
SYMBOL
DELAY
TOLERANCE UNITS
PE2 to TR Switch
tD1
2
±0.1
µs
PE2 to PA_PE
tD3
3
±0.1
µs
PA_PE to PE2
tD4
3
±0.1
µs
TR Switch to PE2
tD5
2
±0.1
µs
PE1 and PE2 encoding details are found in Table 2.
Note that during normal receive and transmit operation that
PE1 is static and PE2 toggles for receive and transmit
states.
ISL3873
PE1
PE2
TR_SW
TR_SW_BAR
tD5
tD1
PA_PE
tD3
tD4
FIGURE 10. TRANSMIT CONTROL SIGNAL SEQUENCING
tuning-fork type watch crystal to permit accurate timekeeping
with very low power consumption during sleep state.
TABLE 2. POWER ENABLE STATES
PE1
PE2
PLL_PE
Power Down State
0
0
1
Receive State
1
1
1
Transmit State
1
0
1
PLL Active State
0
1
1
PLL Disable State
X
X
0
PLL_PE is controlled via the serial interface, and can be used to
disable the internal synthesizer, the actual synthesizer control is
an AND function of PLL_PE, and a result of the OR function of
PE1 and PE2. PE1 and PE2 will directly control the power enable
functionality of the LO buffer(s)/phase shifter.
Master Clock
Prescaler
The ISL3873 contains a clock prescaler to provide flexibility in
the choice of clock input frequencies. For 11Mb/s operation, the
internal master clock, MCLK, must be between 11MHz and
16MHz. The clock generator itself requires an input from the
prescaler that is twice the desired MCLK frequency. Thus the
lowest oscillator frequency that can be used for an 11MHz
MCLK is 22MHz. The prescaler can divide by integers and 1/2
steps (IE 1, 1.5, 2, 2.5). Another way to look at it is that the
divisor ratio between the external clock source and the internal
MCLK may be integers between 2 and 14.
Typically, the 44MHz baseband clock is used as the input, and
the prescaler is set to divide by 2. Another useful configuration
is to set the prescaler to divide by 1.5 (resulting in 44MHz ÷3)
for an MCLK of 14.67MHz. Contact the factory for further
details on setting the clock prescaler register in the ISL3873.
Low-Frequency Crystal
The ISL3873 MAC controller can accept the same clock signal
as the PHY baseband processor (typically 44MHz), thereby
avoiding the need for a separate, MAC-specific oscillator. The
ISL3873 input has a low-frequency oscillator. This lowfrequency oscillator is intended for use with a 32.768KHz,
16
If a 32.768KHz crystal is connected, the resulting LF clock is
supplied to an interval timer to permit measuring sleep
intervals as well as providing a programmable wake-up time.
In addition, the clock generator can operate either from CLKIN
or (very slowly) from the LF clock. Glitch-free switching
between these two clock sources, under firmware control, is
provided by two, non-architectural Strobe functions (“FAST”
and “SLOW”). In addition, during hardware reset, the clock
generator source is set to the LF clock if no edges are
detected on CLKIN for two cycles of the LF clock (roughly 61
microseconds). This allows proper initialization with omission
of either clock source, since without the LF crystal attached
there will not be cycles of the LF clock to activate the detection
circuit. The ability to initialize the ISL3873 using the LF
oscillator to generate MCLK allows the high-frequency (PHY)
oscillator to be powered down during sleep state. If this is
done, firmware can turn on power to the PHY oscillator upon
wake-up, and use the interval timer to measure the start-up
and stabilization period before switching to use CLKIN.
Clock Generator
The ISL3873 can operate with MCLK frequencies up to at least
25MHz and CLKIN frequencies of at least 50MHz. The MCLK
prescaler generates MCLK (and QCLK) from the external clock
provided at the CLKIN input, or from the output of the LF
oscillator. The MCLK prescaler divides the selected input clock
by any integer value between 2 and 16, inclusive.
• When using a 44MHz CLKIN, as is typical for 802.11 or
802.11b controllers with a PC Card Host Interface, common
divisors are 3 (14.67MHz), 4 (11MHz), or 5 (8.8MHz)
• When using a 48MHz CLKIN, as is typical for 802.11 or
802.11b controllers with a USB host interface, common
divisors are 3 (16MHz), 4 (12MHz), or 6 (8MHz)
The MCLK prescaler is set to divide by 16 at hardware reset
to allow initialization firmware to be executed from slow
ISL3873
memory devices at any CLKIN frequency. The MCLK
prescaler generates glitch free output when the divisor is
changed. This allows firmware to change the MCLK
frequency during operation, which is especially useful to
selectively reduce operating speed, thereby conserving
power, when full speed processing is not required.
22pF
XTALIN
C1
X1
Option Register (COR, bit 7). RESET originates from the
HOST system which applies RESET for at least 0.01ms after
VCC has reached 90% of its end value (see PC-Card
standard, Vol. 2, Ch. 4.12.1).
The MD[15:8] pin values are sampled during RESET or
Software Reset (SRESET). These pins have internal 50K
resistors. External pull-up or pull-down resistors (typically
10kΩ) are used for bits which need to be configured
differently than the default.
Table 3 summarizes the effect per pin. Table 4 provides the
MD15 and MD14 bit values required to allow the ISL3873 to
use Serial EEPROM option.
10MΩ
C2
XTALOUT
4700pF
FIGURE 11. 32.768kHz CRYSTAL
Power On Reset Configuration
Power On Reset is issued to the ISL3873 with the RESET
pin or via the soft reset bit, SRESET, in the Configuration
MD[11], StrIdle, has no equivalent functionality in any control
register. When asserted at reset, it will inhibit firmware
execution. This is used to allow the initial download of
firmware in “Genesis Mode”. See the Hardware Reference
Manual for more details. The latch is cleared when the
Software Reset, SRESET, COR(7) is active.
TABLE 3. INITIALIZATION STRAPPING OPTIONS ON MBUS DATA PINS
BITS
NAME
DEFAULT
FUNCTION
30
Indicates type of serial NV memory to be read by initialization firmware in on-chip ROM.
Up to 8 NV device types can be encoded with (StrIdle or NVtype). If StrIdle = 0, NV memory holds a firmware
image, and NVtype identifies 1 of 4 “large” (. = 128Kb) types. If StrIdle = 1, the NV memory just holds the CIS,
and NVtype identifies 1 of 4 “small” (< = 8Kb) types.
15:14 NVtype[1:0]
13
SHIenable
0
Use the Serial Host Interface (USB), and disable all PC Card functions except attribute space, for access to the
COR and HCR for firmware debugging support. When = 0, use the Parallel Host Interface (PC Card or ISA).
12
4Wire
1
Use 4-wire interface to SRAM (CS-, OE-, WEH-, WEL-) the ISL3873 x8 SRAMs. When = 0 selects 5-wire
interface for use with x16 SRAM (CS-, OE-, WE-, UBE-, LBE-).
11
StrIdle
0
Start idle (wait for download from PC Card host interface).
10
Mem16
0
RAM and NV space at startup is x 16. When = 0 RAM and NV space at startup is x 8. If starting from off-chip NV
memory this setting must indicate the width of the startup Flash Memory. During initialization, firmware can set
separate widths or RAM and NV space in the Memory Control Register.
9
NVds
0
Disable mapping of off-chip control store to NV space (hence map off-chip control store to RAM space). When
= 0 off-chip control store is mapped to NV memory
8
ROMds
1
Disable on-chip control store ROM. When = 0 enable on-chip control store ROM.
7
ISAmode
0
Set host interface control signals and address decoding for PC card. When = 1 set host interface signals and
address decoding is for ISA bus, with all registers in I/O space and attribute space disabled. To use ISA mode,
PHIenable must be = 1 to enable a parallel host interface.
6
FCRinIO
0
Enable I/O space decoding for the physical FCRs. When = 1, the COR, CSR, and PRR registers are accessible
at I/O space offsets 0x40, 0x42, and 0x44 respectively. When = 0 these registers are only accessible in attribute
space. This bit is ignored when PHIenable = 0, and is overridden (forced = 1) when ISAmode =1. FCRinIO = 1
is useful for PC Card operation (PHIenable = 1, ISAmode = 0) to allow non-OS software to access the COR/HCR
in OS environments where the system software does not permit application software to access attribute space.b
5:0
Spare
0 x 00
Not assigned.
a. FCRinIO = 1 forces HAMASK [0] = 1 to expand I/O space decoding from 0 x 40 to 0 x 80 bytes.
TABLE 4. SERIAL EEPROM SELECTION
MD15
MD14
0
0
AT45DB011
DEVICE TYPE
Large Serial Device used to transfer firmware to SRAM
FUNCTION
0
1
24C08 (Note)
Small Serial Device which contains only CIS. MAC goes idle after loading CIS and waits for host.
1
X
None
Modes not supported in firmware at this time. Consult factory for additional device types added.
NOTE: The operating frequency of the serial port is 400kHz with a voltage of 3.3V.
17
ISL3873
Baseband Processor
The Baseband Processor operation is controlled by the
ISL3873 firmware. Detailed information on programming the
Baseband Processor can be obtain by contacting the factory.
BBP Packet Reception
The receive demodulator scrutinizes I and Q for packet
activity. When a packet arrives at a valid signal level the
demodulator acquires and tracks the incoming signal. It then
sifts through the demodulator data for the Start Frame
Delimiter (SFD). After SFD is detected, The BBP picks off
the needed header fields from the real-time demodulated
bitstream.
Assuming all is well with the header, the BBP decodes the
signal field in the header and switches to the appropriate
data rate. If the signal field is not recognized, or the CRC16
is in error, the demodulator will return to acquisition mode
looking for another packet. If all is well with the header, and
after the demodulator has switched to the appropriate data
rate, then the demodulator will continue to provide data to
the MAC in the ISL3873 indefinitely.
of the receiver when it is needed most at low signal level. At
IF, the gain control is linear and covers the bulk of the gain
control range of the receiver.
The AGC loop is partially digital which allows for holding the
gain fixed during a packet. The AGC sensing mechanism uses
a combination of the I and Q A/D converters and the detected
signal level in the IF to determine the gain settings. The A/D
outputs are monitored in the ISL3873 for the desired nominal
level. When it is reached, by adjusting the receiver gain, the
gain control is locked for the remainder of the packet.
RX_AGC_IN Interface
The signal level in the IF stage is monitored to determine
when to impose the 30dB gain reduction in the RF stage.
This maximizes the dynamic range of the receiver by
keeping the RF stages out of saturation at high signal levels.
When the IF circuits’ sensor output reaches 0.5VDD, the
ISL3873 comparator switches in the 30dB pad and also
adds 30dB of gain to the IF AGC amplifier. This
compensates the IF AGC and RSSI measures.
TX I/Q DAC Interface
RX I/Q A/D Interface
The PRISM baseband processor chip (ISL3873) includes
two 6-bit Analog to Digital converters (A/Ds) that sample the
balanced differential analog input from the IF down converter
device (HFA3783). The I/Q A/D clock, samples at twice the
chip rate with a nominal sampling rate of 22MHz.
The interface specifications for the I and Q A/Ds are listed in
Table 5. The ISL3873 is designed to be DC coupled to the
HFA3783.
TABLE 5. I, Q, A/D SPECIFICATIONS
PARAMETER
MIN
TYP
MAX
0.90
1.00
1.10
Input Bandwidth (-0.5dB)
-
11MHz
-
Input Capacitance (pF)
-
2
-
Input Impedance (DC)
5kΩ
-
-
-
22MHz
-
Full Scale Input Voltage (VP-P)
fS (Sampling Frequency)
The voltages applied to pin 16, VREF and pin 21, IREF set
the references for the internal I and Q A/D converters. In
addition, For a nominal I/Q input of 400mVP-P, the
suggested VREF voltage is 1.2V.
AGC Circuit
The AGC circuit as shown in Figure 12 is designed to adjust
for signal level variations and optimize A/D performance for
the I and Q inputs by maintaining the proper headroom on
the 6-bit converters. There are two gain stages being
controlled. At RF, the gain control is a 30dB step change.
This RF gain control optimizes the receiver dynamic range
when the signal level is high and maintains the noise figure
18
The transmit section outputs balanced differential analog
signals from the transmit DACs to the HFA3783. These are
DC coupled and digitally filtered.
Transmitter Description
The ISL3873 transmitter is designed as a Direct Sequence
Spread Spectrum Phase Shift Keying (DSSS PSK)
modulator which is capable of handling data rates of up to
11Mbps (refer to AC and DC specifications). The various
modes of the modulator are Differential Binary Phase Shift
Keying (DBPSK) for 1Mbps, Differential Quaternary Phase
Shift Keying (DQPSK) for 2Mbps, and Complementary Code
Keying (CCK) for 5.5Mbps and 11Mbps.
CCK is essentially a quadra-phase form of M-ARY Orthogonal
Keying. A description of that modulation can be found in
Chapter 5 of: “Telecommunications System Engineering”, by
Lindsey and Simon, Prentiss Hall publishing.
The implemented data rates using a clock rate of 44MHz are
shown in Table 6 and the modulation schemes are indicated
in Figure 13. The major functional blocks of the transmitter
include a Processor Interface, Modulator, Data Scrambler,
Preamble/Header Generator, TX Filter, AGC Control, and
ADC and DAC circuits. Figure 17 provides a basic block
diagram of the DSSS Baseband Processor with an
emphasis on the transmitter section. Figure 19 provides a
basic block diagram of the DSSS Baseband Processor with
an emphasis on the receive section.
The preamble is always transmitted as the DBPSK waveform
while the header can be configured to be either DBPSK, or
DQPSK, and data packets can be configured for DBPSK,
DQPSK, or CCK. The preamble is used by the receiver to
ISL3873
Header/Packet Description
achieve initial Pseudo Noise (PN) synchronization while the
header includes the necessary data fields of the
communications protocol to establish the physical layer link.
The transmitter generates the synchronization preamble and
header and knows when to make the DBPSK to DQPSK or
CCK switchover, as required.
The ISL3873 is designed to handle packetized Direct
Sequence Spread Spectrum (DSSS) data transmissions. The
ISL3873 generates its own preamble and header information.
It uses two packet preamble and header configurations. The
first is backwards compatible with the existing IEEE 802.111997 1 and 2Mbps modes and the second is the optional
shortened mode which maximizes throughput at the expense
of compatibility with legacy equipment.
For the 1 and 2Mbps modes, the transmitter accepts data
from the external source, scrambles it, differentially encodes
it as either DBPSK or DQPSK, and spreads it with the BPSK
PN sequence. The baseband digital signals are then output
to the external IF modulator.
In the long preamble mode, the device uses a
synchronization preamble of 128 symbols along with a
header that includes four fields. The preamble is all 1’s
(before entering the scrambler) plus a Start Frame Delimiter
(SFD). The actual transmitted pattern of the preamble is
randomized by the scrambler. The preamble is always
transmitted as a DBPSK waveform (1Mbps). The duration of
the long preamble and header is 192µs.
For the CCK modes, the transmitter inputs the data and
partitions it into nibbles (4 bits) or bytes (8 bits). At 5.5Mbps,
it uses two of those bits to select one of 4 complex spread
sequences from a table of CCK sequences and then QPSK
modulates that symbol with the remaining 2 bits. Thus, there
are 4 possible spread sequences to send at four possible
carrier phases, but only one is sent. This sequence is then
modulated on the I and Q outputs. The initial phase
reference for the data portion of the packet is the phase of
the last bit of the header. At 11Mbps, one byte is used as
above where 6 bits are used to select one of 64 spread
sequences for a symbol and the other 2 are used to QPSK
modulate that symbol. Thus, the total possible number of
combinations of sequence and carrier phases is 256. Of
these only one is sent.
In the short preamble mode, the modem uses a
synchronization field of 56 zero symbols along with an SFD
transmitted at 1Mbps. The short header is transmitted at
2Mbps. The synchronization preamble is all 0’s to distinguish
it from the long header mode and the short preamble SFD is
the time reverse of the long preamble SFD. The duration of
the short preamble and header is 96µs.
Start Frame Delimiter (SFD) Field (16 Bits)
Bit rates for the ISL3873 are defined in Table 6. This table
provides information on bit rates, data rates and symbol
rates for an MCLK of 44MHz clock. Figure 13 shows the
modulation schemes for the different bits rates. The
modulator is completely independent from the demodulator,
allowing the PRISM baseband processor to be used in full
duplex operation.
This field is used to establish the link frame timing. The
ISL3873 will not declare a valid data packet, even if it PN
acquires, unless it detects the SFD. The ISL3873 receiver
auto-detects if the packet is long or short preamble and sets
SFD time-out. The timer starts counting after initialization of
the de-scrambler is complete.
RX_RF_AGC
RX_IF_DET
RX_IF_AGC
RX_I±
HFA3683
HFA3783
RX_Q±
1
THRESH.
DETECT
1
7
AGC
CTL
IF
DAC
I ADC
Q ADC
6
6
DEMOD
DATA I/O
I/O
ISL3873
FIGURE 12. AGC CIRCUIT
TABLE 6. BIT RATE TABLE EXAMPLES FOR MCLK = 44MHz
DATA
MODULATION
A/D SAMPLE CLOCK
(MHz)
TX SETUP CR 5
BITS 1, 0
RX SIGNAL CR 63
BITS 7, 6
DATA RATE
(Mbps)
SYMBOL RATE
(MSPS)
DBPSK
22
00
00
1
1
DQPSK
22
01
01
2
1
CCK
22
10
10
5.5
1.375
CCK
22
11
11
11
1.375
19
ISL3873
802.11 DSSS BPSK
1Mbps
BARKER
802.11 DSSS QPSK
2Mbps
BARKER
5.5Mbps CCK
COMPLEX
SPREAD FUNCTIONS
11Mbps CCK
COMPLEX
SPREAD FUNCTIONS
DATA
1 BIT ENCODED TO
ONE OF 2 CODE
WORDS
(TRUE-INVERSE)
2 BITS ENCODED
TO ONE OF
4 CODE WORDS
4 BITS ENCODED
TO ONE OF 16
COMPLEX CCK
CODE WORDS
8 BITS ENCODED
TO ONE OF 256
COMPLEX CCK
CODE WORDS
IOUT
QOUT
CHIP
RATE
SYMBOL
RATE
11 CHIPS
11 CHIPS
8 CHIPS
8 CHIPS
11 MC/S
11 MC/S
11 MC/S
11 MC/S
1 MS/S
1 MS/S
1.375 MS/S
1.375 MS/S
I vs. Q
FIGURE 13. MODULATION MODES
PREAMBLE (SYNC)
128/56 BITS
Start FRAME DELIMITER
16 BITS
SIGNAL FIELD
8 BITS
SERVICE FIELD
8 BITS
LENGTH FIELD
16 BITS
CRC16
16 BITS
HEADER
PREAMBLE
FIGURE 14. 802.11 PREAMBLE/HEADER
Header Field
The header field is defined by four fields which are shown in
Figure 14. These fields are Signal Field, Service Field,
Length Field and CITT-CRC16 Field. They are further
defined by the following:
Signal Field (8 Bits) - This field indicates what data rate the
data packet that follows the header will be. The ISL3873
receiver looks at the signal field to determine whether it
needs to switch from DBPSK demodulation into DQPSK, or
CCK demodulation at the end of the preamble and header
fields.
Service Field (8 Bits) - The MSB of this field is used to
indicate the correct length when the length field value is
ambiguous at 11Mbps. See IEEE STD 802.11 for definition
of the other bits. Bit 2 is used by the ISL3873 to indicate that
the carrier reference and the bit timing references are
derived from the same oscillator (locked oscillators).
Length Field (16 Bits) - This field indicates the number of
microseconds it will take to transmit the payload data
(PSDU). The external controller (MAC) will check the length
field in determining when it needs to de-assert RX_PE.
20
CCITT - CRC 16 Field (16 Bits) - This field includes the
16-bit CCITT - CRC 16 calculation of the three header fields.
This value is compared with the CCITT - CRC 16 code
calculated at the receiver. The ISL3873 receiver will indicate
a CCITT - CRC 16 error via CR24 bit 2 and will lower
MD_RDY and reset the receiver to the acquisition mode if
there is an error.
The CRC or cyclic Redundancy Check is a CCITT CRC-16
FCS (Frame Check Sequence). It is the ones complement of
the remainder generated by the modulo 2 division of the
protected bits by the polynomial:
x16 + x12 + x5 + 1
The protected bits are processed in transmit order. All CRC
calculations are made ahead of data scrambling. A shift
register with two taps is used for the calculation. It is preset
to all ones and then the protected fields are shifted through
the register. The output is then complemented and the
residual shifted out MSB first.
The following Configuration Registers (CR) are used to
program the preamble/header functions, more programming
details about these registers can be found in the Control
Registers section of this document:
ISL3873
CR 3 - Defines the short preamble length minus the SFD in
symbols. The 802.11 protocol requires a setting of 56d = 38h
for the optional short preamble.
CR 4 - Defines the long preamble length minus the SFD in
symbols. The 802.11 protocol requires a setting of
128d = 80h for the mandatory long preamble.
CR 5 Bits 0, 1 - These bits of the register set the Signal field
to indicate what modulation is to be used for the data portion
of the packet.
CR 6 - The value to be used in the Service field.
CR 7 and 8 - Defines the value of the transmit data length
field. This value includes all symbols following the last
header field symbol and is in microseconds required to
transmit the data at the chosen data rate.
The packet consists of the preamble, header and MAC Protocol
Data Unit (MPDU). The data is transmitted exactly as received
from the control processor. Some dummy bits will be appended
to the end of the packet to ensure an orderly shutdown of the
transmitter. This prevents spectrum splatter. At the end of a
packet, the external controller is expected to de-assert the
TX_PE line to shut the transmitter down.
For the 1Mbps DBPSK data rates and for the header in all rates
using the long preamble, the data coder implements the
desired DBPSK coding by differential encoding the serial data
from the scrambler and driving both the I and Q output
channels together. For the 2Mbps DQPSK data rate and for the
header in the short preamble mode, the data coder implements
the desired coding as shown in the DQPSK Data Encoder
Table 7. This coding scheme results from differential coding of
dibits (2 bits). Vector rotation is counterclockwise although bits
6 and 7 of configuration register CR 1 can be used to reverse
the rotation sense of the TX or RX signal if desired.
Spread Spectrum Modulator Description
The modulator is designed to generate DBPSK, DQPSK, and
CCK spread spectrum signals. The modulator is capable of
automatically switching its rate where the preamble is
DBPSK modulated, and the data and/or header are
modulated differently. The modulator can support date rates
of 1, 2, 5.5 and 11Mbps. Quadraphase (I/Q) modulation is
used at the baseband for all modulation modes. Further
information on the programming details required to set up
the modulator can be obtained by contacting the factory.
TABLE 7. DQPSK DATA ENCODER
Scrambler and Data Encoder Description
The modulator has a data scrambler that implements the
scrambling algorithm specified in the IEEE 802.11 standard.
This scrambler is used for the preamble, header, and data in
all modes. The data scrambler is a self synchronizing circuit.
It consists of a 7-bit shift register with feedback from
specified taps of the register. Both transmitter and receiver
use the same scrambling algorithm. The scrambler can be
disabled by setting CR32 bit 2 to 1.
NOTE: Be advised that the IEEE 802.11 compliant scrambler in the
ISL3873 has the property that it can lock up (stop scrambling) on
random data followed by repetitive bit patterns. The probability of this
happening is 1/128. The patterns that have been identified are all
zeros, all ones, repeated 10s, repeated 1100s, and repeated
111000s. Any break in the repetitive pattern will restart the scrambler.
To ensure that this does not cause any problem, the CCK waveform
uses a ping pong differential coding scheme that breaks up repetitive
0’s patterns.
Scrambling is done by division with a prescribed polynomial
as shown in Figure 15. A shift register holds the last quotient
and the output is the exclusive or of the data and the sum of
taps in the shift register. The transmit scrambler seed for the
long preamble or for the short preamble can be set with
CR48 or CR49.
SERIAL DATA
IN
XOR
SERIAL
DATA OUT
Z-1 Z-2 Z-3 Z-4
Z-5 Z-6 Z-7
XOR
0
00
+90
01
+180
11
-90
10
In the 1Mbps DBPSK mode, the I and Q Channels are
connected together and driven with the output of the scrambler
and differential encoder. The I and Q Channels are then both
multiplied with the 11-bit Barker word at the spread rate. The I
and Q signals go to the Quadrature upconverter (HFA3724) to
be modulated onto a carrier. Thus, the spreading and data
modulation are BPSK modulated onto the carrier.
For the 2Mbps DQPSK mode, the serial data is formed into
dibits or bit pairs in the differential encoder as detailed
above. One of the bits from the differential encoder goes to
the I Channel and the other to the Q Channel. The I and Q
Channels are then both multiplied with the 11-bit Barker
word at the spread rate. This forms QPSK modulation at the
symbol rate with BPSK modulation at the spread rate.
CCK Modulation
For the CCK modes, the spreading code length is 8 complex
chips and based on complementary codes. The chipping rate is
11Mchip/s. The following formula is used to derive the CCK
code words that are used for spreading both 5.5 and 11Mbps:
 j ( ϕ1 + ϕ2 + ϕ3 + ϕ4 ) j ( ϕ1 + ϕ3 + ϕ4 ) j ( ϕ1 + ϕ2 + ϕ4 )
,e
,e
c = e
,

FIGURE 15. SCRAMBLING PROCESS
–e
21
DIBIT PATTERN (d0, d1)
d0 IS FIRST IN TIME
PHASE SHIFT
j ( ϕ1 + ϕ4 )
,e
j ( ϕ1 + ϕ2 + ϕ3 )
,e
j ( ϕ1 + ϕ3 )
, –e
j ( ϕ1 + ϕ2 )
,e
jϕ 1 


ISL3873
CCK symbol. All odd numbered symbols of the MPDU are
given an extra 180 degree (π) rotation in accordance with the
DQPSK modulation as shown in Table 8. Symbol numbering
starts with “0” for the first symbol of the MPDU.
(LSB to MSB), where c is the code word.
The terms: ϕ1, ϕ2, ϕ3, and ϕ4 are defined below for
5.5Mbps and 11Mbps.
This formula creates 8 complex chips (LSB to MSB) that are
transmitted LSB first. The coding is a form of the generalized
Hadamard transform encoding where the phase ϕ1 is added
to all code chips, ϕ2 is added to all odd code chips, ϕ3 is
added to all odd pairs of code chips and ϕ4 is added to all
odd quads of code chips.
The phase ϕ1 modifies the phase of all code chips of the
sequence and is DQPSK encoded for 5.5 and 11Mbps. This
will take the form of rotating the whole symbol by the
appropriate amount relative to the phase of the preceding
symbol. Note that the last chip of the symbol defined above
is the chip that indicates the symbol’s reference phase.
For the 5.5Mbps CCK mode, the output of the scrambler is
partitioned into nibbles. The first two bits are encoded as
differential symbol phase modulation in accordance with Table
8. All odd numbered symbols of the MPDU are given an extra
180 degree (π) rotation in addition to the standard DQPSK
modulation as shown in the table. The symbols of the MPDU
shall be numbered starting with “0” for the first symbol for the
purposes of determining odd and even symbols. That is, the
MPDU starts on an even numbered symbol. The last data dibits
d2, and d3 CCK encode the basic symbol as specified in Table
9. This table is derived from the CCK formula above by setting
ϕ2 = (d2*pi)+ pi/2, ϕ3 = 0, and ϕ4 = d3*pi. In Table 9 d2 and d3
are in the order shown and the complex chips are shown LSB
to MSB (left to right) with LSB transmitted first.
The data dibits: (d2, d3), (d4, d5), (d6, d7) encode ϕ2, ϕ3,
and ϕ4 respectively based on QPSK as specified in Table
10. Note that this table is binary, not Grey, coded.
Transmit Filter Description
To minimize the requirements on the analog transmit filtering,
the transmit section shown in Figure 17 has an output digital
filter. This filter is a Finite Impulse Response (FIR) style filter
whose passband shape is set by tap coefficients. This filter
shapes the spectrum to meet the radio spectral mask
requirements while minimizing the peak to average amplitude
on the output. To meet the particular spread spectrum
processing gain regulatory requirements in Japan on channel
14, an extra FIR filter shape has been included that has a
wider main lobe. This increases the 90% power bandwidth
from about 11MHz to 14MHz. It has the unavoidable side
effect of increasing the amplitude modulation, so the available
transmit power is compromised by 2dB when using this filter
(CR 11 bit 5).
TABLE 10. QPSK ENCODING TABLE
DIBIT PATTERN (d(i), d(i+1))
d(i) IS FIRST IN TIME
00
01
10
11
TABLE 8. DQPSK ENCODING TABLE
EVEN SYMBOLS ODD SYMBOLS
DIBIT PATTERN (d(0), d(1)) PHASE CHANGE PHASE CHANGE
d(0) IS FIRST IN TIME
(+jω)
(+jω)
0
π
π/2
π
3π/2 (-π/2)
3π/2 (-π/2)
00
01
11
10
0
π/2
TABLE 9. 5.5Mbps CCK ENCODING TABLE
d2, d3
0
π/2
π
3π/2 (-π/2)
TX Power Control
The transmitter power can be controlled via two registers.
The first register, CR58, contains the results of power
measurements digitized by the ISL3873. By comparing this
measurement to what is needed for transmit power, a
determination is made whether to raise or lower the transmit
power. It does this by writing the power level desired to
register CR31.
Clear Channel Assessment (CCA) and
Energy Detect (ED) Description
CHIPS
00
1j
1
1j
-1
1j
1
-1j
1
01
-1j
-1
-1j
1
1j
1
-1j
1
10
-1j
1
-1j
-1
-1j
1
1j
1
11
1j
-1
1j
1
-1j
1
1j
1
At 11Mbps, 8 bits (d0 to d7; d0 first in time) are transmitted
per symbol.
The first dibit (d0, d1) encodes the phase ϕ1 based on
DQPSK. The DQPSK encoder is specified in Table 8 above.
The phase change for ϕ1 is relative to the phase ϕ1 of the
preceding symbol. In the case of rate change, the phase
change for ϕ1 is relative to the phase ϕ1 of the preceding
22
PHASE
The Clear Channel Assessment (CCA) circuit implements the
carrier sense portion of a Carrier Sense Multiple Access
(CSMA) networking scheme. The Clear Channel Assessment
(CCA) monitors the environment to determine when it is clear
to transmit. The CCA circuit in the ISL3873 can be
programmed to be a function of RSSI (energy detected on the
channel), CS1, SQ1, or various combinations. The CCA is
used by the Media Access Controller (MAC) in the ISL3873.
The MAC decides on transmission based on traffic to send
and the CCA indication. The CCA indication can be ignored,
allowing transmissions independent of any channel
conditions. The CCA in combination with the visibility of the
ISL3873
various internal parameters (i.e., Energy Detection
measurement results), can assist the MAC in executing
algorithms that can adapt to the environment. These
algorithms can increase network throughput by minimizing
collisions and reducing transmissions liable to errors.
There are three measures that can be used in the CCA
assessment. The Receive Signal Strength Indication (RSSI)
which indicates the energy at the antenna, CS1 and carrier
sense (SQ1). CS1 becomes active anytime the AGC portion
of the circuit becomes unlocked, which is likely at the onset of
a signal that is strong enough to support 11Mbps, but may not
occur with the onset of a signal that is only strong enough to
support 1 or 2MBps. CS1 stays active until the AGC locks and
a SQ1 assessment is done, if SQ1 is false, then CS1 is
cleared, which deasserts CCA. If SQ1 is true, then tracking is
begun, and CCA continues to show the channel busy. CS1
may occur at any time during acquisition as the AGC state
machine runs asynchronously with respect to slot times.
SQ1 becomes active only when a spread signal with the
proper PN code has been detected, and the peak correlation
amplitude to sidelobe ratio exceeds a set threshold, so it
may not be adequate in itself.
A SQ1 evaluation occurs whenever the AGC has remained
locked for the entire data ingest period. When this happens,
SQ1 is updated between 8 and 9µs into the 10µs dwell. If
CS1 is not active, two consecutive SQ1’s are required to
advance the part to tracking.
The state of CCA is not guaranteed from the time RX_PE
goes high until the first CCA assessment is made. At the end
of a packet, after RXPE has been deasserted, the state of
CCA is also not guaranteed.
The Receive Signal Strength Indication (RSSI) measurement is
derived from the state of the AGC circuit. ED is the comparison
result of RSSI against a threshold. The threshold may be set to
an absolute power value, or it may be set to be N dB above the
measured noise floor. See CR 35. The ISL3873 measures and
stores the RSSI level when it detects no presence of BPSK or
QPSK signals. The average value of a 256 value buffer is taken
to be the noise floor. Thus, the value of the noise floor will adapt
to the environment. A separate noise floor value is maintained
for each antenna. An initial value of the noise floor is
established within 50µs of the chip being active and is refined
as time goes on. Deasserting RX_PE does not corrupt the
learned values. If the absolute power metric is chosen, this
threshold is normally set to between -70 and -80dBm.
If desired, ED may be used in the acquisition process as well
as CCA. ED may be used to mask (squelch) weak signals
and prevent radio reception of signals too weak to support
the high data rates, signals from adjacent cells, networks, or
buildings.
The Configuration registers effecting the CCA algorithm
operation are summarized below (more programming details
23
on these registers can be found under the Control Registers
section of this document).
The CCA output from pin 60 of the device can be defined as
active high or active low through CR 1 (bit 2).
CR9(6:5) allows CCA to be programmed to be a function of ED
only, the logical operation of (CS1 OR SQ1), the logical function
of (ED AND (CS1 OR SQ1)), or (ED OR (CS1 OR SQ1)).
CR9(7) lets the user select from sampled CCA mode, which
means CCA will not glitch, is updated once per symbol and is
valid for reading at 15.8µs or 18.7µs. In non-sampled mode,
CCA may change at any time, potentially several times per slot,
as ED and CS1 operate asynchronously to slot times.
In a typical system CCA will be monitored to determine when
the channel is clear. Once the channel is detected busy,
CCA should be checked periodically to determine if the
channel becomes clear. Once MD_RDY goes active, CCA
should be ignored for the remainder of the message. Failure
to monitor CCA until MD_RDY goes active (or use of a timeout circuit) could result in a stalled system as it is possible for
the channel to be busy and then become clear without an
MD_RDY occurring.
AGC Description
The AGC system consists of the 3 chips handling the receive
signal, the RF to IF downconverter HFA3683, the IF to
baseband converter HFA3783, and the baseband processor
(BBP) section of the ISL3873. The AGC loop (Figure 11) is
digitally controlled by the BBP. Basically it operates as follows:
Initially, the receiver is set for high gain. The percent of time
that the A/D converters in the baseband processor are
saturated is monitored along with signal amplitude and the
gain is adjusted down until the amplitude is what will
optimize the demodulator’s performance. If the amount of
saturation is great, the initial gain adjust steps are large. If
the signal overload is small, they are less. When the gain is
about right and the A/Ds’ outputs are within the lock window
(CR19), the BBP declares AGC lock and stops adjusting for
the duration of the packet. If the signal level then varies more
than a preset amount (CR20, CR29), the AGC is declared
unlocked and the gain again allowed to readjust.
The BBP looks for the locked state following an unlocked
state (CS1) as one indication that a received signal is on the
antenna. This starts the receive process of looking for PN
correlation (SQ1). Once PN correlation and AGC lock are
found, the processor begins acquisition.
For large signals, the power level in the RF stage output is
also monitored and if it is large, the LNA stage is shut down.
This removes 30dB of gain from the receive chain which is
compensated for by replacing 30dB of gain in the IF AGC
stage. There is some hysteresis in this operation and once
the AGC locks, it is locked as well. This improves the
receiver dynamic range.
ISL3873
RX_RF_AGC Pad Operation
30dB Pad Engaging (RF Chip Low Gain):
If the AGC is not locked onto a packet, a '1' on the
ifCompDet input will engage in the 30dB attenuation pad.
This causes the AGC to go out of lock and also forces the
attenuation accumulator to be set to the programmed value
of CR27. The AGC then attempts to lock on the signal.
If the AGC is locked on a packet, ifCompDet is ignored.
30DB PAD RELEASING (RF CHIP HIGH GAIN):
If the AGC is not locked onto a packet and the attenuation
accumulator sum falls below the programmable threshold
(CR27), the pad will release. This is for the case where a
noise spike kicked in the 30dB pad and the pad should
release when the noise spike ends. Since the noise floor is
different for different environments, it is possible that in many
cases CR27’s programmed value will be below the noise floor
and the pad will not be removed except by RXPE going low.
There is a recommended value to program CR27 (24dB), but
that depends on what environment the radio is in.
During a packet (after AGC lock), the 30dB pad is held
constant and the CR27 threshold is ignored.
RXPE low forces the pad to release whether in the middle of
a packet or not. At the end of a packet, RXPE always goes
low, forcing the pad to release.
Notes: The attenuation accumulator is basically about equal to
the current RSSI value.
The accumulator output, after going through the interpolator
lookup table, feeds the AGC D/A.
The pad value is programmable (CR17), but is
recommended to be set to 30dB.
ifCompDet is a signal from the HFA3783 chip. A '1' indicates
its inputs are near saturation and it needs the RF chip to
switch from high gain to low gain.
RX_IF_Det is the input to the ISL3873 chip which is
connected to ifCompDet on the HFA3783.
RX_RF_AGC is the output of the ISL3873 chip and '1' is high
gain, '0' is low gain.
Demodulator Description
The receiver portion of the baseband processor, performs A/D
conversion and demodulation of the spread spectrum signal.
It correlates the PN spread symbols, then demodulates the
DBPSK, DQPSK, or CCK symbols. The demodulator includes
a frequency tracking loop that tracks and removes the carrier
frequency offset. In addition, it tracks the symbol timing, and
differentially decodes and descrambles the data. The data is
output through the RX Port to the external processor.
The PRISM baseband processor in the ISL3873 uses
differentially coherent demodulation. The ISL3873 is
designed to achieve rapid settling of the carrier tracking loop
during acquisition. Rapid phase fluctuations are handled
24
with a relatively wide loop bandwidth which is then stepped
down as the packet progresses. Coherent processing
improves the BER performance margin as opposed to
differentially coherent processing for the CCK data rates.
The baseband processor uses time invariant correlation to
strip the Barker code spreading and phase processing to
demodulate the resulting signals in the header and
DBPSK/DQPSK demodulation modes. These operations are
illustrated in Figure 18 which is an overall block diagram of
the receiver processor.
In processing the DBPSK header, input samples from the I and
Q A/D converters are correlated to remove the spreading
sequence. The peak position of the correlation pulse is used to
determine the symbol timing. The sample stream is decimated
to the symbol rate and corrected for frequency offset prior to
PSK demodulation. Phase errors from the demodulator are fed
to the NCO through a lead/lag filter to maintain phase lock. The
carrier is de-rotated by the carrier tracking loop. The
demodulated data is differentially decoded and descrambled
before being sent to the header detection section.
In the 1Mbps DBPSK mode, data demodulation is performed
the same as in header processing. In the 2Mbps DQPSK
mode, the demodulator demodulates two bits per symbol
and differentially decodes these bit pairs. The bits are then
serialized and descrambled prior to being sent to the output.
In the CCK modes, the receiver removes carrier frequency
offsets and uses a bank of correlators to detect the
modulation. A biggest picker finds the largest correlation in
the I and Q Channels and determines the sign of those
correlations. For this to happen, the demodulator must know
the starting phase which is determined by referencing the
data to the last bit of the header. Each symbol demodulated
determines 1 or 2 nibbles of data. This is then serialized and
descrambled before being passed to the output.
Carrier tracking is via a lead/lag filter using a digital Costas
phase detector. Chip tracking in the CCK modes is chip
decision directed or slaved to the carrier tracking depending
on whether or not the locked oscillator design is utilized in
the radio.
Acquisition Description
A projected worst case time line for the acquisition of a
signal with a short preamble and header is shown. The
synchronization part of the preamble is 56 symbols long
followed by a 16-bit SFD. The receiver must monitor the
antenna to determine if a signal is present. The timeline is
broken into 10µs blocks (dwells) for the scanning process.
This length of time is necessary to allow enough integration
of the signal to make a good acquisition decision. This worst
case time line example assumes that the signal arrives part
way into the first dwell such as to just barely catch detection.
The signal and the scanning process are asynchronous and
the signal could start anywhere. In this timeline, it is
assumed that the signal is present in the first 10µs dwell, but
was missed due to power amplifier ramp up.
ISL3873
TX
POWER
RAMP
SFD
56 SYMBOL SYNC
2
20 SYMBOLS
20 SYMBOLS
7 SYM
AGC SETTLE AND LOCK
AND INITIAL DETECTION
VERIFY AND CIR/FREQUENCY
ESTIMATION AND CMF/NCO
JAMMING
16 SYMBOLS
SFD DET
START DATA
SEED
DESCRAMBLER
START SFD SEARCH
FIGURE 16. ACQUISITION TIMELINE, NON DIVERSITY
VDDA (ANALOG)
GND (ANALOG)
VDD (DIGITAL)
GND (DIGITAL)
TX AGC
CONTROL
TX_IF_AGC
6-BIT
DAC
ANTSEL
ANTSEL
REGISTER
TRANSMIT
FILTER
PREAMBLE/HEADER
CRC-16
GENERATOR
TXI+/DAC
TXQ+/DAC
INTERNAL
SIGNALS
TRANSMIT
PORT
6-BIT
ADC
OUTPUT MUX
TX_AGC_IN
TEST CONTROL
VREF
OUTPUT MUX
IREF
TX_RDY
TXCLK
MODULATOR,
BARKER/CCK
TX_DATA
SCRAMBLER
TXD
RXCLK
TIMING
GENERATOR
MCLK
TX_PE
CCA
PROCESSOR
INTERFACE
TX
STATE
CONTROL
MAC
CONTROL
SIGNALS
MCLK
FIGURE 17. DSSS BASEBAND PROCESSOR, TRANSMIT SECTION
Meanwhile signal quality and signal frequency
measurements are made simultaneous with symbol timing
measurements. A CS1 followed by SQ1 active, or two
consecutive SQ1s will cause the part to finish the acquisition
phase and enter the tracking phase.
Prior to initial acquisition the NCO is inactive (0Hz) and
carrier phase measurement are done on a symbol by symbol
basis. After acquisition, coherent DPSK demodulation is in
effect. After a brief setup time as illustrated on the timeline,
the signal begins to emerge from the demodulator.
25
It takes 7 more symbols to seed the descrambler before valid
data is available. This occurs in time for the SFD to be received.
At this time the demodulator is tracking and in the coherent
PSK demodulation mode so it will no longer acquire new
signals. If a much larger signal overrides the signal being
demodulated (a collision), the demodulator will abort the
tracking process and attempt to acquire the new signal. Failure
to find an SFD within the SFD timeout interval will result in a
receiver reset and return to acquisition mode.
ISL3873
Channel Matched Filter (CMF) Description
The receive section shown in Figure 19 operates on the
RAKE receiver principle which maximizes the SNR of the
signal by combining the energy of multipath signal
components. The RAKE receiver is implemented with a
Channel Matched Filter (CMF) using a FIR filter structure with
16 taps. The CMF is programmed by calculating the Channel
Impulse Response (CIR) of the channel and mathematically
manipulating that to form the tap coefficients of the CMF.
Thus, the CMF is set to compensate the channel
characteristics that distort the signal. Since the calculation of
the CIR is inaccurate at low SNR or in the presence of strong
CW interference, the chip has thresholds (CR 36 to 39) that
are set to substitute a default CMF shape under those
conditions. This default CMF shape is designed to
compensate only the known transmit and receive non linearity.
PN Correlators Description
There are two types of correlators in the ISL3873 baseband
processor. The first is a parallel matched filter correlator that
correlates for the Barker sequence used in preamble,
header, and PSK data modes. This Barker code correlator is
designed to handle BPSK spreading with carrier offsets up
to ±50ppm and 11 chips per symbol. Since the spreading is
BPSK, the correlator is implemented with two real
correlators, one for the I and one for the Q Channel. The
same Barker sequence is always used for both I and Q
correlators.
These correlators are time invariant matched filters
otherwise known as parallel correlators. They use one
sample per chip for correlation although two samples per
chip are processed. The correlator despreads the samples
from the chip rate back to the original symbol rate giving
10.4dB processing gain for 11 chips per symbol. While
despreading the desired signal, the correlator spreads the
energy of any non correlating interfering signal.
The second form of correlator is the parallel correlator bank
used for detection of the CCK modulation. For the CCK
modes, the 64 wide bank of parallel correlators is
implemented with a Fast Walsh Transform to correlate the 4
or 64 code possibilities. This greatly simplifies the circuitry
of the correlation function. It is followed by a biggest picker
which finds the biggest of 4 or 64 correlator outputs
depending on the rate. This is translated into 2 or 6 data
bits. The detected output is then processed through the
differential phase decoder to demodulate the last two bits
of the symbol.
Data Demodulation and Tracking
Description (DBPSK and DQPSK Modes)
The signal is demodulated from the correlation peaks
tracked by the symbol timing loop (bit sync) as shown in
Figure 18. The frequency and phase of the signal is
corrected using the NCO that is driven by the phase locked
26
loop. Averaging the phase errors over 10 symbols gives the
necessary frequency information for seeding the NCO
operation.
Data Decoder and Descrambler
Description
The data decoder that implements the desired DQPSK
coding/decoding as shown in Table 11. The data is formed
into pairs of bits called dibits. The left bit of the pair is the first
in time. This coding scheme results from differential coding
of the dibits. Vector rotation is counterclockwise for a positive
phase shift, but can be reversed with bit 7 or 6 of CR 1.
For DBPSK, the decoding is simple differential decoding.
TABLE 11. DQPSK DATA DECODER
PHASE SHIFT
DIBIT PATTERN (D0, D1)
D0 IS FIRST IN TIME
0
00
+90
01
+180
11
-90
10
The data scrambler and de-scrambler are self synchronizing
circuits. They consist of a 7-bit shift register with feedback of
some of the taps of the register. The scrambler is designed
to ensure smearing of the discrete spectrum lines produced
by the PN code.
One thing to keep in mind is that both the differential
decoding and the descrambling cause error extension or
burst errors. This is due to two properties of the processing.
First, the differential decoding process causes errors to
occur on pairs of symbols. When a symbol’s phase is in
error, the next symbol will also be decoded wrong since the
data is encoded in the change in phase from one symbol to
the next. Thus, two errors are made on two successive
symbols. Therefore up to 4 bits may be wrong although on
the average only 2 are. In QPSK mode, these may occur
next to one another or separated by up to 2 bits. In the CCK
mode, when a symbol decision error is made, up to 6 bits
may be in error although on average only 3 bits will be in
error. Secondly, when the bits are processed by the
descrambler, these errors are further extended. The
descrambler is a 7-bit shift register with two taps exclusive
or’ed with the bit stream. Thus, each error is extended by a
factor of three. Multiple errors can be spaced the same as
the tap spacing, so they can be canceled in the descrambler.
In this case, two wrongs do make a right. Given all that, if a
single error is made the whole packet is discarded anyway,
so the error extension property has no effect on the packet
error rate. It should be taken into account if a forward error
correction scheme is contemplated.
Descrambling is self synchronizing and is done by a
polynomial division using a prescribed polynomial. A shift
register holds the last quotient and the output is the exclusiveor of the data and the sum of taps in the shift register.
ISL3873
SAMPLES
AT 2X CHIP
RATE
CORRELATION
PEAK
CORRELATION TIME
T0 + 1 SYMBOL CORRELATOR
T0
OUTPUT REPEATS
CORRELATOR OUTPUT IS THE RESULT OF CORRELATING
THE PSEUDO NOISE(PN) SEQUENCE WITH THE RECEIVED SIGNAL
EARLY
ON-TIME
LATE
T0 + 2 SYMBOLS
FIGURE 18. CORRELATION PROCESS
Data Demodulation in the CCK Modes
In this mode, the demodulator uses Complementary Code
Keying (CCK) modulation for the two highest data rates. It is
slaved to the low rate processor which it depends on for
acquisition of initial timing and phase tracking information.
The low rate section acquires the signal, locks up symbol
and carrier tracking loops, and determines the data rate to
be used for the MPDU data.
The demodulator for the CCK modes takes over when the
preamble and header have been acquired and processed.
On the last bit of the header, the phase of the signal is
captured and used as a phase reference for the high rate
differential demodulator.
The signal from the A/D converters is carrier frequency and
phase corrected by a DESPIN stage. This removes the
frequency offset and aligns the I and Q Channels properly for
the correlators. The sample rate is decimated to 11MSPS for
the correlators after the DESPIN since the data is now
synchronous in time.
The demodulator knows the symbol timing, so the
correlation is batch processed over each symbol. The
correlation outputs from the correlator are compared to each
other in a biggest picker and the chosen one determines 6
bits of the symbol. The QPSK phase of the chosen one
determines two more bits for a total of 8 bits per symbol. Six
bits come from which of the 64 correlators had the largest
output and the last two are determined from the QPSK
differential demod of that output. In the 5.5Mbps mode, only
4 of the correlator outputs are monitored. This demodulates
2 bits for which of 4 correlators had the largest output and 2
more for the QPSK demodulation of that output for a total of
4 bits per symbol.
Equalizer Description
The ISL3873 employs a Decision Feedback Equalizer (DFE)
to improve performance in the presence of significant
multipath distortion. The DFE combats Inter Chip
Interference (ICI) and Inter Symbol Interference (ISI). The
equalizer is trained on the sample data collected during the
first part of the acquisition after the AGC has settled and the
27
antenna selected. The same data is used for CMF
calculations and equalizer training. Once the equalizer has
been set up, it is used to process the incoming symbols in a
decision feedback manner. After the Fast Walsh transform is
performed, the detected symbols are corrected for ICI before
the bigger picker where the symbol decision process is
performed. Once a symbol has been demodulated, the
calculated residual energy from that symbol is subtracted
from the incoming data for the next symbol. That corrects for
the ISI component. The DFE is not adapted during the
packet as the channel impulse response is not expected to
vary significantly during that brief time. Register CR10 bits 4
and 5 can disable these equalizers separately.
Tracking
Carrier tracking is performed on the de-rotated signal
samples from the complex multiplier in a four phase Costas
loop. This forms the error term that is integrated in the lead/lag
filter for the NCO, closing the loop. Tracking is only measured
when there is a chip transition. Note that this tracking is
dependent on a positive SNR in the chip rate bandwidth.
The symbol clock is tracked by a sample interpolator that
can adjust the sample timing forwards and backwards by 72
increments of 1/8th chip. This approach means that the
ISL3873 can only track an offset in timing for a finite interval
before the limits of the interpolator are reached. Thus,
continuous demodulation is not possible.
Locked Oscillator Tracking
Symbol tracking can be slaved to the carrier offset tracking
for improved performance as long as at both the transmitting
and the receiving radios, the bit clocks and carrier frequency
clocks are locked to common crystal oscillators. A bit carried
in the SERVICE field (bit 2) indicates whether or not the
transmitter has locked clocks. When the same bit is set at
the receiver (CR6 bit 2), the receiver knows it can track the
bit clock by counting down the carrier tracking offset. This is
much more accurate than tracking the bit clock directly.
CR33 bit 6 can enable or disable this capability.
ISL3873
VDDA (ANALOG)
GND (ANALOG)
VDD (DIGITAL)
GND (DIGITAL)
CCA to
MAC
(INTERNAL)
RX_IF_DET
RX_IF_AGC
6
6-BIT
A/D
6
CORRELATOR
BARKER
6-BIT
A/D
BIT
SYNC
8
PEAK
EXTRACT.
8
RXD TO MAC
EQUAL.
BIAS
ADDER
SYMBOL
DECISION
MUX
MUX
RECEIVE
STATE
MACHINE
ANTENNA
SWITCH
CONTROL
TIMING
GENERATOR
MCLK
RESET
RX_PE
MCLK
FIGURE 19. DSSS BASEBAND PROCESSOR, RECEIVE SECTION
28
MD_RDY TO MAC
LOOP
FILTER
TEST CONTROL
ANTSEL
CCK
CORREL
RXCLK TO MAC
DECISION FEEDBACK
EQUALIZER
COHERENT
TIMING
INTEGRATOR
ANTSEL
AND RECEIVE
SIGNALS TO MAC
RX_DATA
DESCRAMBLER
SYMBOL
TRACKING
NCO
INTERNAL TRANSMIT
DPSK
DEMOD
PREAMBLE/HEADER
CRC-16 DETECT
RXQ
CMF
TRAINING
CHANNEL
MATCHED FILTER
RXI
CLEAR CHANNEL
ASSESSMENT/
SIGNAL QUALITY
DIVERSITY
CONTROL
DOWN CONVERT
ANT SEL
INTERPOLATING
BUFFER
RX_RF_AGC
AGC
CONTROL
6-BIT
DAC
6-BIT
DAC
6-BIT
DAC
TXI
TXQ
ISL3873
This section indicates the typical performance measures for
a radio design. The performance data below should be used
as a guide. In general, the actual performance depends on
the application, interference environment, RF/IF
implementation and radio component selection.
the link do not have locked oscillators, then symbol tracking
is done by a conventional early-late chip tracking method.
Eb/N0
7
8
9
10
11
12
1.E-01
Overall Eb/N0 Versus BER Performance
1.E-02
BER 2.0
The PRISM chip set has been designed to be robust and
energy efficient in packet mode communications. The
demodulator uses coherent processing for data
demodulation. The figures below show the performance of
the baseband processor when used in conjunction with the
HFA3783 IF and the PRISM recommended IF filters. Off the
shelf test equipment are used for the RF processing. The
curves should be used as a guide to assess performance in
a complete implementation.
Factors for carrier phase noise, multipath, and other
degradations will need to be considered on an
implementation by implementation basis in order to predict
the overall performance of each individual system.
1.E-03
BER 1.0
The PRISM demodulator performs with an implementation
loss of less than 4dB from theoretical in a AWGN
environment with low phase noise local oscillators. For the
1 and 2Mbps modes, the observed errors occurred in
groups of 4 and 6 errors. This is because of the error
extension properties of differential decoding and
descrambling. For the 5.5 and 11Mbps modes, the errors
occur in symbols of 4 or 8 bits each and are further
extended by the descrambling. Therefore the error patterns
are less well defined.
Clock Offset Tracking Performance
The PRISM baseband processor is designed to accept data
clock offsets of up to ±25ppm for each end of the link (TX
and RX). This effects both the acquisition and the tracking
performance of the demodulator. The budget for clock offset
error is 0.75dB at ±50ppm. No appreciable degradation was
seen for operation in AWGN at ±50ppm. Symbol tracking is
accomplished by one of two methods. If both ends of the link
employ locked oscillators for their bit timing and carrier
frequency generation, symbol tracking is done by dividing
down the carrier frequency offset. If either one of the ends of
29
1.E-04
THY 1, 2
1.E-05
1.E-06
1.E-07
1.E-08
FIGURE 20. BER vs Eb/N0 PERFORMANCE FOR PSK MODES
1.E+00
5
6
7
8
Eb/N0
9
10
11
12
13
14
1.E-01
BER 11
1.E-02
1.E-03
BER
Figure 18 shows the curves for theoretical DBPSK/DQPSK
demodulation with coherent demodulation and
descrambling as well as the PRISM performance measured
for DBPSK and DQPSK. The theoretical performance for
DBPSK and DQPSK are the same as shown on the
diagram. Figure 21 shows the theoretical and actual
performance of the CCK modes. The losses in both figures
include RF and IF radio losses; they do not reflect the
ISL3873 losses alone. The ISL3873 baseband processing
losses from theoretical are, by themselves, a small
percentage of the overall loss.
1.E+00
BER
Demodulator Performance
1.E-04
1.E-05
THY 11
THY 5.5
BER 5.5
1.E-06
1.E-07
1.E-08
1.E-09
FIGURE 21. BER vs Eb/N0 PERFORMANCE FOR CCK MODES
Carrier Offset Frequency Performance
The correlators used for acquisition for all modes and for
demodulation in the 1 and 2Mbps modes are time invariant
matched filter correlators otherwise known as parallel
correlators. They use two samples per chip and are tapped
at every other shift register stage. Their performance with
carrier frequency offsets is determined by the phase roll rate
due to the offset. For an offset of +50ppm (combined for both
TX and RX) will cause the carrier to phase roll 22.5 degrees
over the length of the correlator. This causes a loss of
0.22dB in correlation magnitude which translates directly to
Eb/N0 performance loss. In the PRISM chip design, the
carrier phase locked loop is inactive during acquisition.
During tracking, the carrier tracking loop corrects for offset,
so that no degradation is noted. In the presence of high
multipath and high SNR, however, some degradation is
expected.
ISL3873
RSSI Performance
100
The RSSI value is reported on CR62 in hex and is linear with
signal level in dB. Figure 22 shows the RSSI curve
measured on a whole evaluation radio. This takes into
account the full gain adjust range of all radio parts. To get
signal level in dBm on a radio, simply subtract the RSSI
value in decimal from 100.
120
RSSI
100
90
80
70
60
50
40
PER
30
MEAN
20
STDDEV
10
0
-10
RSSI IN DE
80
-5
0
5
10
15
20
25
SNR IN THE SPREAD BANDWIDTH AT 1Mbps
FIGURE 23. SIGNAL QUALITY MEASURE AND PER vs SNR
60
40
ED Threshold
0
-100
-80
-60
-40
-20
0
SIGNAL LEVEL IN dBm
FIGURE 22. RSSI vs SIGNAL LEVEL
Signal Quality Estimate
A signal quality measure is available on CR51 for use by the
MAC. This measure is the SNR in the carrier tracking loop
and can be used to determine when the demodulator is
working near to the noise floor and likely to make errors.
Figure 23 shows the performance of the SQ measure versus
signal to noise level.
The performance of the ED threshold is shown in Figure 24.
Setting this threshold will effect CCA only. Using ED as part
of the CCA measure will allow deferral to large signals even
if they are not correlated to the desired spread signals.
ED can be read from CR61 bit 4. Using ED and RSSI can
assist the MAC in determining the presence of non
correlating signals such as frequency hoppers or microwave
ovens. For example, the MAC can elect to try to transmit
over microwave oven interference but not count the results in
rate shifting algorithms.
40
ED THRESHOLD VALUE IN DECIMAL
20
30
20
10
0
STARTS MISSING
MISSING
-10
0
10
20
30
SNR IN SPREAD BANDWIDTH
FIGURE 24. ED THRESHOLD vs SNR IN dB AT 1Mbps
30
40
ISL3873
Plastic Ball Grid Array Packages (BGA)
o
A1
CORNER
A
D
V192.14x14
192 BALL PLASTIC BALL GRID ARRAY PACKAGE
A1
CORNER I.D.
INCHES
E
B
TOP VIEW
0.15
M C A B
0.006
0.08
M C
0.003
b
A1
CORNER
D1
16 15 14 13 12 11 10 9 8 7 6 5 4 3 2 1
A1
CORNER I.D.
A
B
C
D
E
F
G
H E1
J
K
L
M
N
P
R
T
S
A
SYMBOL
MIN
MAX
MIN
MAX
NOTES
A
-
0.059
-
1.40
-
A1
0.012
0.016
0.31
0.41
-
A2
0.033
0.039
0.83
0.99
-
b
0.016
0.020
0.41
0.51
7
D/E
0.547
0.555
13.90
14.10
-
D1/E1
0.468
0.476
11.90
12.10
-
N
192
192
-
e
0.032 BSC
0.80 BSC
-
MD/ME
16 x 16
16 x 16
3
bbb
0.004
0.10
-
aaa
0.005
0.12
Rev. 1 1/01
NOTES:
1. Controlling dimension: MILLIMETER. Converted inch
dimensions are not necessarily exact.
2. Dimensioning and tolerancing conform to ASME Y14.5M-1994.
3. “MD” and “ME” are the maximum ball matrix size for the “D”
and “E” dimensions, respectively.
4. “N” is the maximum number of balls for the specific array size.
5. Primary datum C and seating plane are defined by the spherical crowns of the contact balls.
6. Dimension “A” includes standoff height “A1”, package body
thickness and lid or cap height “A2”.
7. Dimension “b” is measured at the maximum ball diameter,
parallel to the primary datum C.
e
S
A
BOTTOM VIEW
MILLIMETERS
ALL ROWS AND COLUMNS
A1
A2
bbb C
8. Pin “A1” is marked on the top and bottom sides adjacent to A1.
9. “S” is measured with respect to datum’s A and B and defines
the position of the solder balls nearest to package centerlines. When there is an even number of balls in the outer row
the value is “S” = e/2.
aaa C
C
A
SEATING PLANE
SIDE VIEW
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Intersil products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design and/or specifications at any time without notice.
Accordingly, the reader is cautioned to verify that data sheets are current before placing orders. Information furnished by Intersil is believed to be accurate and reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result from its use.
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