Sep 1999 LT1813: 100MHz, 750V/µs Amplifier Draws Only 3mA

DESIGN FEATURES
LT1813: 100MHz, 750V/µs Amplifier
Draws Only 3mA
by George Feliz
Introduction
The LT1813 is a 100MHz dual operational amplifier that has been
optimized for supply voltages under
12V. It features an easy-to-use voltage
feedback topology with high impedance inputs, yet it slews 750V/µs
with only 3mA supply current. DC
performance has not been neglected
—the device has a 1.5mV maximum
VOS and a 400nA maximum IOS.
Performance
A summary of important specifications of the LT1813, compared to its
higher voltage brethren, is shown in
Table 1. A key figure of merit is the
ratio of gain bandwidth to supply
current (GBW/ISUPPLY, expressed in
units of MHz/mA). The new process
employed by the LT1813 forsakes high
supply voltage operation for a 3×–4×
increase in MHz/mA compared to the
LT1361 and LT1364. Blazing speed
from such a modest amount of supply
current is extremely attractive for low
power applications. The LT1813 also
propagates the family traits of
matched, high input impedance inputs and low VOS, IB, IOS and input
noise. The improved common mode
input range of the LT1813 adds to its
utility in low supply voltage applica-
Table 1. Comparison of dual, high speed op amps (VS = ±5V, 25°C)
LT1813
LT1364
LT1361
100MHz
50MHz
37MHz
3.0mA
6.0mA
3.8mA
33.3MHz/mA
8.3MHz/mA
9.7MHz/mA
750V/µs
450V/µs
350V/µs
Input Common Mode Range
±4.0V
+3.4V, –3.2V
+3.4V, –3.2V
Output Swing
±4.0V
±4.1V
±4.0V
Output Current (VOUT = ±3V)
60mA
45mA
38mA
VOS (Max)
1.5mV
1.5mV
1.0mV
IB (Max)
4.0µA
2.0µA
1.0µA
IOS (Max)
400nA
350nA
250nA
A VOL (Min)
1.5V/mV
3.5V/mV
3V/mV
Input Noise Voltage
8nV/ Hz
9nV/ Hz
9nV/ Hz
Input Noise Curent
1pA/ Hz
1pA/ Hz
0.9pA/ Hz
CLOAD
1000pF
∞
∞
12.6V
36V
36V
Gain Bandwidth
Supply Current per Amplifier
GBW/I SUPPLY
Slew Rate
Max Supply Voltage (V+ to V –)
tions. Stability with capacitive loading is another distinctive and desirable
feature. Although the LT1813 is not
stable with unlimited capacitive loads,
it is stable with nearly two orders of
magnitude more capacitance than
competitors’ high speed amplifiers.
The small-signal transient response
100mV/DIV
100pF
500pF
1000pF
Circuit Design
200ns/DIV
Figure 1. LT1813 in a gain-of-one configuration, no RL; CL = 100pF, 500pF or 1000pF
Linear Technology Magazine • September 1999
in unity gain with CLOAD =100pF,
500pF and 1000pF is shown in
Figure␣ 1.
The LT1813 extends the frequency
response of applications such as active
filters, instrumentation amplifiers and
buffers. Figure 2 shows the LT1813
converting a single-ended signal to a
differential drive for the LTC1417
14-bit analog-to-digital converter
(ADC). Note that the top amplifier
provides unity voltage gain, but the
amplifier is configured in a noise-gain
of 2 to match the phase response of
the bottom amplifier, which has a
gain of –1. The filter in front of the
ADC reduces broadband noise. The
spurious free dynamic range (SFDR)
of this circuit is –79dB for a 425kHz,
2VP-P input.
A simplified schematic of the circuit
is shown in Figure 3. The circuit looks
similar to a current feedback amplifier, but both inputs are high
5
DESIGN FEATURES
impedance as in a traditional voltage
feedback amplifier. A complementary
cascade of emitter followers, Q1–Q4,
buffers the noninverting input and
drives one side of resistor R1. The
other side of the resistor is driven by
Q5–Q8, which form a buffer for the
inverting input. The input voltage
appears across the resistor, generating currents in Q3 and Q4 that are
mirrored by Q9–Q11 and Q13–Q15
into the high impedance node. Transistors Q17–Q24 form the output
stage. Bandwidth is set by R1, the
gm’s of Q3, Q4, Q7 and Q8 and the
compensation capacitor, CT.
The voltage drops of Q1–Q4 and
the diodes Q10 and Q14 set the input
common mode range of the amplifier.
The emitters of Q3 and Q4 follow the
noninverting input. As the input
approaches either supply rail, the
limiting voltage is determined by the
saturation of Q3 or Q4, which occurs
at approximately a VBE plus a VSAT
from the supply rail. Typically, the
input common mode range is 1V from
either supply rail, and is guaranteed
by the CMRR specification to be 1.5V
from either rail. This excellent input
range is achieved without compromising the output impedance of the
mirrors Q9–Q11 and Q13–Q15,
because Q25 and Q26 provide floating bias points for cascode devices Q9
and Q13. Lower bandwidth processes
cannot successfully use this tech-
1k
–
100Ω
1k
LT1813
+
IN
1k
1k
SERIAL
OUT
14 BITS
LTC1417
500pF
–
100Ω
+
LT1813
Figure 2. Single-ended to differential ADC buffer: 2VP-P input at 425kHz yields –79dB SFDR
nique and maintain high bandwidth,
due to phase shift in the mirror.
The current available to slew compensation capacitor CT is proportional
to the voltage that appears across R1.
This method of “slew boost” achieves
low distortion due to its inherent linearity with input step size. Large slew
currents can be generated without
increasing quiescent current. A low
value for R1 reduces the input noise
voltage to 8nV/√Hz and helps reduce
input offset voltage and drift. The
LT1813 is built with small-geometry,
multi-GHz transistors that produce
abundant bandwidth with meager
operating currents and allow for further reduction of idling supply current.
The output stage buffers the high
impedance node from the load by
providing current gain. The simplest
output stage would be two pairs of
complementary emitter followers,
which would provide a current gain of
BetaNPN × BetaPNP. Unfortunately, this
gain is insufficient for driving even
modest loads. Adding another emitterfollower or a Darlington configuration
reduces output swing and creates
instability with large capacitive loads.
The solution used on the LT1813
was to create a pair of composite
transistors formed by transistors
Q19–Q21 and Q22–Q24. The current
mirrors attached to the collectors of
emitter followers Q19 and Q22 provide additional current gain. The ratio
of transistor geometries Q20 to Q21
and Q23 to Q24 increase the current
gain by approximately fifteen. There
continued on page 15
V+
Q11
Q25
Q10
Q20
Q12
Q9
Q19
Q17
RC
Q7
–IN
Q5
C1
CC
Q3
OUT
R1
Q6
Q2
Q8
Q21
Q1
Q18
+IN
CT
Q4
Q22
C2
Q13
Q14
Q26
Q15
Q16
Q23
Q24
V–
Figure 3. LT1813 simplified schematic
6
Linear Technology Magazine • September 1999
DESIGN FEATURES
Single-Supply RGB
Video Amplifier
The LT1399 can be used with a single
supply voltage of 6V or more to drive
ground-referenced RGB video. As seen
in Figure 10, two 1N4148 diodes, D1
and D2, have been placed in series
with the output of the amplifier A1,
but within the feedback loop formed
by resistor R8. These diodes effectively level-shift A1’s output downward
by 2 diodes, allowing the circuit output to swing to ground.
Amplifier A1 is used in a positive
gain configuration. The feedback
resistor R8 is 324Ω. The gain resistor
is created from the parallel combination of R6 and R7, giving a
Thevenin-equivalent 80.4Ω connected
to 3.75V. This gives an AC gain of five
from the noninverting input of amplifier A1 to the cathode of D2. However,
the video input is also attenuated
before arriving at A1’s positive input.
Assuming a 75Ω source impedance
for the signal driving VIN, the Thevenin-equivalent signal arriving at A1’s
positive input is 3V + (0.4 • VIN), with
a source impedance of 714Ω. The
combination of these two inputs gives
an output at the cathode of D2 of 2 •
VIN with no additional DC offset. The
75Ω back termination resistor R9
halves the signal again such that
VOUT equals a buffered version of VIN.
R1
1k
Y
R2
1k
+
A1
1/3 LT1399
R-Y
–
R
R3
324Ω
R4
324Ω
R6
205Ω
+
R2
1k
R8
316Ω
R5
75Ω
A2
1/3 LT1399
–
Linear Technology has introduced the
LT1399 and LT1399HV triple 300MHz
current feedback amplifiers. Both of
these products are well suited for use
in component video applications. The
higher supply voltage rating of the
LT1399HV makes it an excellent
choice for LCD driver applications.
Both products feature 4.6mA of supply current per amplifier, 300MHz
–3dB bandwidth, an exceptional
0.1dB gain flatness of 150MHz, 800V/
µs slew rate and a shutdown pin for
each channel.
G
5V
R10
324Ω
R1
1000Ω
A1
1/3 LT1399
R2
1300Ω
+
A3
1/3 LT1399
–
ALL RESISTORS 1%
VS = ± 5V
R16
75Ω
VIN
–
R3
160Ω
C1
4.7µF
VS
6V TO 12V
R6
107Ω
+
B-Y
R13
1k
Conclusion
R11
75Ω
R9
845Ω
R12
1k
It is important to note that the
4.7µF capacitor C1 is required to
maintain the voltage drop across
diodes D1 and D2 when the circuit
output drops low enough that the
diodes might otherwise be reverse
biased. This means that this circuit
works fine for continuous video input,
but will require that C1 be charged
after a period of inactivity at the input.
+
D2
D1
1N4148 1N4148
R9
75Ω
VOUT
R8
324Ω
B
R14
324Ω
R4
75Ω
R5
2.32Ω
R7
324Ω
R15
324Ω
Figure 9. Buffered color-difference to RGB matrix
LT1813, continued from page 6
is no output swing penalty as the
swing is limited at the collectors of Q9
and Q13. The dynamics of the composites are not as benign as those of
emitter followers, so compensation is
required and is provided by C1 and
C2.
The stability with capacitive loads
is provided by the RC, CC network
between the output stage and the
Linear Technology Magazine • September 1999
Figure 10. Single-supply RGB video amplifier (one of three channels)
gain node. When the amplifier is driving a light or moderate load, the output
can follow the high impedance node
and the network is bootstrapped and
has no effect. When driving a heavy
load such as a capacitor or smallvalue resistor, the network is
incompletely bootstrapped and adds
to the compensation provided by CT.
The added capacitance provided by
CC slows down the amplifier and the
zero created by RC adds phase margin
to increase stability.
Conclusion
The combination of a high slew rate,
DC accuracy and a frugal 3mA-peramplifier supply current make the
LT1813 a compelling choice for low
voltage and low power, high speed
applications.
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