LINEAR TECHNOLOGY JUNE 2006 IN THIS ISSUE… COVER ARTICLE Reduce Charge Time for High Capacity Li-Ion Batteries with 2A Continuous Charging .............1 Tom Hack Issue Highlights ..................................2 Linear Technology in the News….........2 DESIGN FEATURES 650MHz Selectable-Gain Amplifier/ Differential ADC Driver Has Small Form but Many Functions ...................6 Cheng-Wei Pei Dual Step-Up Converter Drives White LEDs with 1000:1 PWM Dimming ......10 Keith Szolusha Hot Swap™ Controller Monitors and Reports Power Supply Status ............12 Josh Simonson Efficient Buck-Boost Converter Ideal for Power Saving Modes and Wide Input Voltage Ranges ..................................16 Kevin Ohlson Dual/Triple Power Supply Monitor for Undervoltage and Overvoltage on Positive and Negative Supplies .....19 Andrew Thomas High Speed Low Power RS485 Transceivers with Integrated Switchable Termination ....................27 Ray Schuler and Steven Tanghe VOLUME XVI NUMBER 2 Reduce Charge Time for High Capacity Li-Ion Batteries with 2A Continuous Charging Introduction The latest high capacity Li-Ion batteries meet the needs of power hungry portable devices, but they also increase the demands placed on battery chargers—demands that can be too much for a standard linear charger. For instance, a linear charger, operating at 1A charging current, charges a 1Ahr battery to 70% capacity within one hour, and fully charges it within three hours. Newer 2Ahr batteries need twice that current in order to be fully charged in the same amount of time. The problem is that a linear charger operating at 2A produces too much heat for continuous charging—it’s just too inefficient. The LTC4001 solves this problem by incorporating a high 1.5A VLDO™ Operates Down to 0.4V Output and Maintains 100mV Dropout .........................................................30 Bill Walter DESIGN IDEAS ....................................................33–46 (complete list on page 33) New Device Cameos ...........................46 Design Tools ......................................47 Sales Offices .....................................48 Figure 1. A typical LTC4001-based Li-Ion battery charger occupies minimal board real estate. by Tom Hack efficiency PWM charger to perform continuous 2A battery charging. It works with both standard and currentlimited wall adapters—where the latter lowers battery charger dissipation and operating temperature. Big Features; Small Footprint A full-featured battery charger based on the LTC4001 requires an area not much larger than a dime (Figure 1). Fully programmable timer and charge rate terminations are included. Automatic battery “topping off” is also included. Filtering prevents accidental recharge from occurring in noisy environments (such as found in GPRS cellular phones). The LTC4001 works readily with NTC thermistors for battery temperature sensing. Remote battery sensing is included. Soft-start is fully programmable. The LTC4001 also drives charge status LEDs and provides logic signals for microprocessor-based designs. The LTC4001 is tiny, fitting into a 4mm × 4mm package, but other factors also contribute to the charger’s small footprint. High operating frequency (1.5MHz) reduces the size of the inductors and capacitors. Input short circuit blocking is built in so no external diode is required. Current continued on page 3 L, LT, LTC, LTM, Burst Mode, OPTI-LOOP, Over-The-Top and PolyPhase are registered trademarks of Linear Technology Corporation. Adaptive Power, BodeCAD, C-Load, DirectSense, Easy Drive, FilterCAD, Hot Swap, LinearView, µModule, Micropower SwitcherCAD, Multimode Dimming, No Latency ΔΣ, No Latency Delta-Sigma, No RSENSE, Operational Filter, PanelProtect, PowerPath, PowerSOT, SmartStart, SoftSpan, Stage Shedding, SwitcherCAD, ThinSOT, True Color PWM, UltraFast and VLDO are trademarks of Linear Technology Corporation. Other product names may be trademarks of the companies that manufacture the products. L EDITOR’S PAGE Issue Highlights H igh capacity Li-Ion batteries meet the needs of power hungry portable devices, but they also increase the demands placed on battery chargers—demands that can be too much for a standard linear charger. The problem is that a linear chargers cannot efficiently produce enough current to quickly charge the battery. To solve this problem, the LTC4001 uses a high efficiency PWM charger, which makes continuous 2A battery charging practical. See our cover article for more about this breakthrough device. Featured Devices Below is a summary of the other devices featured in this issue. The LT6411 selectable-gain differential amplifier/ADC driver is a good fit for power-critical high-speed signal chain applications. It can produce gains of 1, –1, and 2 with no external components. The dual amplifiers inside the LT6411 allow for easy singleended-to-differential conversion for driving high-speed analog-to-digital converters (ADC). (Page 6) The LT3486 dual LED string driver has two 1.3A channels with high PWM dimming capability in a small 5mm × 3mm DFN package. Since both channels’ power switches are included in the IC, the circuit is simple and small. (Page 10) The LTC4215 combines a robust Hot Swap circuit with an I2C interface and data converter to allow power monitoring as well as hot-plug functionality and fault isolation. (Page 12) The LTC3532 is a 300mA buckboost converter, which incorporates automatic Burst Mode® operation, adjustable switching frequency, and integrated soft-start. The LTC3532 is ideal for miniature disk-drive applications or any application that requires high efficiency over a wide range of output currents and input voltages. (Page 16) The LTC2909 is a highly customizable power supply monitoring solution with adjustable input thresh2 olds, input polarity selection, a multi-mode reset timer, and an opendrain RST output. Adjustable input thresholds allow the user to set any trip threshold for the comparator, subject only to the accuracy limitations of the part, instead of having to pick from a factory-set limited collection of thresholds. (Page 19) The LTC2859 and LTC2861 combine a logic-selectable integrated termination resistor with a rugged 20Mbps RS485/RS422 transceiver, providing a single die impedancematched network solution in a tiny package. (Page 27) The LTC3026 is a 1.5A VLDO with input voltage capability down to 1.14V and a low adjustable output voltage from 0.4V to 2.6V. The part also has a very low dropout voltage of only 100mV while delivering up to 1.5A of output current, enabling it to optimize battery run time from single cell applications with a low VIN to VOUT differential. (Page 30) Design Ideas and Cameos Design Ideas start on page 33, including a better way to combine battery packs using an ideal diode—improving on standard diode solutions. L Linear Technology in the News… Linear’s LTM4600 µModule Racks Up Awards Linear Technology’s LTM4600 10A DC/DC µModule™ has been steadily collecting major product awards. The LTM4600 was recently awarded EDN magazine’s Innovation of the Year award in the Power Systems and Modules category, and named Product of the Year by both Electronic Products magazine and AnalogZone. To that, add an Innovation Award from China’s Electronic Engineering & Product World, and a Product Award from EDAW/ Nikkei Electronics in China. Linear Expands Design Capabilities Linear Technology has further expanded its analog design resources in two new locations. In April, Linear announced the opening of the company’s first European design center in Munich. Lothar Maier, CEO of Linear Technology, stated, “We are excited to open our first European design center in the key technology hub of Munich. By centrally locating our latest design center on the European continent, this puts us in an even better position to develop the right products for our major European customers. Germany represents an ideal location for our first European design center, with its focus on the automotive, industrial and communications markets.” And in May, the company opened a new design center in Dallas. With this announcement, Linear now has 12 design centers worldwide. Linear Announces Digital Power Alliance Linear Technology has announced an alliance with Primarion, Inc. to produce digital point-of-load (POL) products for the networking, computing and telecommunications infrastructure markets. Linear’s LTC7510 digital POL controller is the first in a family of PMBus-compliant digital DC/DC controllers. This family of digital power management and conversion products provides flexible, system-level control of sophisticated power management systems. These controller ICs ensure seamless power management in high availability systems, enabling real-time programming and monitoring of key parameters to optimize performance and maximize system uptime. L Linear Technology Magazine • June 2006 DESIGN FEATURES L A Bare Bones Charger LTC4001, continued from page 1 sensing is internal, so there is no need for an expensive milliohm-sized current sense resistor. Inside the LTC4001 The LTC4001 is the basis for a complete 2A Li-Ion battery charger (Figure 2). A 50mA linear charger provides cell conditioning while a synchronous buck charger provides constant-current/constant-voltage high rate charging (up to 2A). Protection and lockouts guard against a variety of events including: shorts at the battery and wall adapter inputs; improper programming of the charge current; open battery and/or overvoltage battery; defective battery; insufficient wall adapter voltage; chip over-temperature; battery over- or under- temperature. 1.5µH WALL ADAPTER 4.5V TO 5.5V + BATSENS BAT 10µF + 4.2V Li-Ion LTC4001 CHRG EN NTC SS GNDSENS PROG IDET TIMER 274Ω 0.1µF Figure 3. A bare bones battery charger 3 5 50mA SENSE CURRENT REVERSAL COMPARATOR + Q 1 2 PGND SW – RAMP SENSE FAULT DRIVER S PWM COMPARATOR SW 10µF MICROPROCESSOR INTERFACE PVIN CLK VINSENSE PVIN PGND 8 OSCILLATOR indicator lights, battery temperature monitoring, and a timer (which may be provided by a microprocessor). In place of a timer, charge terminates when charge current drops below onetenth the high rate charge current (an Figure 3 shows a bare bones 2A battery charger. With only five additional components, this charger offers a high efficiency, high power solution. This implementation leaves out status RD 9 BAT 16 VINSENSE BATSENS OVERCURRENT COMPARATOR SHUTDOWN COMPARATOR + – – + – SS SHUTDOWN LOW BATTERY OVERCURRENT PWM ON TRICKLE ON LOGIC FAULT FAULT TIMER TIMER NTC NTC COMPARATOR CHIP OVERTEMP COMPARATOR CHARGE CURRENT ERROR AMP – + – DISCHARGE SS PROG SHORTED 11 PROG ERROR AMP + 1.2V CHRG TFAULT PROG SHORT COMPARATOR FLOAT VOLTAGE ERROR AMP SOFT-START COPMPARATOR 1.1V + – 15 RECHARGE COMPARATOR LOW CURRENT VIN GOOD RECHARGE + – 10 EN CHRG + – – + + – 7 EN UNDERVOLTAGE COMPARATOR + – 6 IDET COMPARATOR + – 14 LOW-BATTERY COMPARATOR SS SS LOW 150mV OVERVOLTAGE CHIP OVER TEMP BATTERY OVERVOLTAGE COMPARATOR + – CONNECT VOLTAGE REFERENCE GND 17 IDET 13 1.2V PROG 12 GNDSENS 4 Figure 2. Simplified block diagram of the LTC4001 Li-Ion battery charger Linear Technology Magazine • June 2006 3 L DESIGN FEATURES The LED also indicates when the battery is nearly full charged. As the battery approaches the float voltage and charge current drops below the IDET threshold the LED is dimly lit. This is difficult to see, so a better approach uses two LEDs to indicate all charger states (Figure 5). VIN D1 CHRG R1 1k LTC4001 CHRG Interfacing with Microprocessors Figure 4. A simple status indicator VIN R1 27k Q1 2N3906 Q2 TP0610 D1 GRN C/X D2 AMB CHRG R2 1k R3 1k LTC4001 CHRG Q3 2N7002 Figure 5. Full featured status indication IDET threshold equal to 200mA in this case). Internal charge termination may be completely defeated by connecting the timer pin to the IDET pin instead of ground (allowing a microprocessor complete control of charge termination). Adding Status Lights The CHRG pin indicates a variety of charger states (Table 1). Adding a resistor and LED in series with this pin to VIN (Figure 4) indicates charger off (LED off), high rate charging or cell conditioning (LED on continuously at high brightness), and battery temperature out of range/NTC fault (LED blinking). The interface in Figure 6 can distinguish between all states available on the CHRG pin. To detect cell conditioning or high rate charging, force the digital output pin, OUT, high and measure the voltage on the CHRG pin. The N-channel mosfet pulls CHRG low even with a 2k pull-up resistor. Near end of charge, the NMOS turns off, and CHRG sinks only 30µA. The IN pin is pulled high by the 2k resistor connected to OUT. If OUT is placed into a high impedance state, the 30µA sink current from the CHRG pin pulls IN low. When charging stops, CHRG opens and OUT stays high, even with a 390k pull up resistor. If a battery temperature fault occurs during high rate charging, the CHRG pin blinks using a serrated pulse pattern. Nominal timing of this pattern is given in Figure 7. The extra edges provide rapid indication to a microprocessor and may be used to drive a microprocessor interrupt line for low processor overhead, but still provide for a visible fault indication when using LEDs. Operation with Conventional and Current Limited Wall Adapters Wall adapters with or without current limiting may be used with the LTC4001, but the lowest power dissipation battery charging occurs with a current limited wall adapter. To use this feature, program the LTC4001 above the wall adapter current limit. For example, if the wall adapter current limit is 2A, set the LTC4001 charge current slightly higher than 2A (allowing for tolerances). To understand operation with a current limited wall adapter, assume battery voltage VBAT is initially below VTRIKL, the trickle charge threshold (Figure 8). Battery charging begins at approximately 50mA, well below the wall adapter current limit so the voltage into the LTC4001 (VIN) is the wall adapter’s rated output voltage (VADAPTER). Battery voltage rises eventually reaching VTRIKL. The linear charger shuts off and the PWM (high rate) charger turns on using soft start. Battery charging current rises during the soft-start cycle causing a corresponding increase in wall adapter load current. When the wall adapter reaches current limit, the wall adapter output voltage collapses, and VIN VDD Battery Temperature Sensing By adding one resistor and one thermistor, battery temperature sensing may be included. The LTC4001 is designed for Vishay Dale’s “R-T Curve 2” therm- LTC4001 CHRG R1 390k R2 2k µPROCESSOR OUT IN Figure 6. A microprocessor interface Table 1. CHRG Behavior 4 istors, but any thermistor with an RCOLD-to-RHOT ratio of about 7 will also work. If battery sensing is not needed, the NTC pin is grounded. Charger State CHRG Behavior Not charging Open High rate Charging and IBAT>IDET Or cell conditioning NMOS turned on pulling pin low High Rate Charging and IBAT<IDET 30µA pull down current NTC temperature fault while charging at IBAT>IDET Blink 20µs 667ms Figure 7. CHRG temperature fault waveform Linear Technology Magazine • June 2006 DESIGN FEATURES L LINEAR CHARGING VADAPTER WALL ADAPTER IN CURRENT LIMIT So how does LTC4001 dissipation stack up against a 2A linear charger? Most of a linear charger’s dissipation occurs in the series pass element so the dissipation is approximately equal to the voltage drop in the pass element times the charge current. Worst case dissipation occurs at the lowest battery voltage where high rate charging occurs (to make a valid comparison to the LTC4001 this would be 2.85V). For a 5.0V input, this translates into a dissipation of 4.3W! Higher input voltage makes the situation even worse. PWM CHARGING VBAT + VDROP VIN ILIMIT IBAT ITRICKLE VTRIKL VFLOAT VBAT Figure 8. Idealized charging behavior Low Dissipation Trickle charging uses a linear charger but low charge current produces low power dissipation, typically 256mW (VIN = 5V, VBAT = 0). High rate charging uses a high efficiency buck switcher and total charger dissipation is approximately 1.2W at 2A (Figure 9). High rate charging with a current limited wall adapter produces even lower charger dissipation (537mW at VBAT = 4.2V with a 2A current limited wall adapter) because there is very little voltage drop for the battery charging path inside the LTC4001. 1.25 TOTAL APPLICATION CIRCUIT POWER DISSIPATION (W) the LTC4001 PWM charger duty cycle ramps up to 100% (the top-side PMOS switch in the LTC4001 buck regulator stays on continuously.) As the battery voltage approaches VFLOAT, the float voltage error amplifier commands the PWM charger to deliver less than ILIMIT. The wall adapter exits current limit and VIN jumps back up to VADAPTER. Battery charging current continues to drop as VBAT rises, dropping to zero at VFLOAT. Because the voltage drop in the LTC4001 is very low when charge current is highest, power dissipation is also very low. R12 27k Q2 TP0610 D1 GRN C/X D2 AMB CHRG R1 1k Q3 2N7002 R2 1k A full featured battery charger is shown in Figure 10. It includes a three hour timer, battery temperature monitoring, programmable charge and IDET currents, remote sensing, and status lights. A fault light has been included that indicates when a shorted battery is detected or when the battery is out of normal temperature range. VIN = 5V VBAT = 4V 1.00 Conclusion The LTC4001 sets a new standard for small, low parts count, full-featured, high efficiency Li-Ion battery chargers. Low power dissipation makes continuous 2A battery charging practical, cutting dissipation to approximately one fifth the dissipation of a straight linear charger. L 0.75 0.50 0.25 0 500 1500 1000 2000 IBAT (mA) Figure 9. High rate charger power dissipation VIN 2A CURRENT LIMITED WALL ADAPTER 4.5V TO 5.5V Q1 2N3906 A Charger with All the Bells and Whistles R8 10k L1 1.5µH 2A SW VINSENSE PVIN C1 10µF 10V SENSE BATSENS BAT 10µF 10V PGND + 2Ahr 4.2V Li-Ion LTC4001 CHRG GNDSENS EN GND NTC FAULT EXT NTC PROG R9 1.33k R10 10k AT 25°C D3 RED FAULT R3 1k R4 2A 549Ω C3 0.22µF R5 1A 1.10k TIMER SS IDET C4 0.1µF R7 0.1A 1.10k R6 0.2A 549Ω L1: VISHAY DALE IHLP-2525AH-01 R10: NTC VISHAY DALE NTHS0603N02N1002J Figure 10. A full featured battery charger Linear Technology Magazine • June 2006 5 L DESIGN FEATURES 650MHz Selectable-Gain Amplifier/Differential ADC Driver Has Small Form but Many Functions by Cheng-Wei Pei Introduction Operational amplifiers have always been an important part of an analog designer’s bag of tools. Even the most basic tools can be improved, and some recent advancements have increased the utility of the workhorse op amp. For instance, the advent of dual and quad op amp packages allowed engineers to produce a variety of applications from a single device. The new LT6411 selectable-gain differential amplifier/ADC driver makes a good thing better by adding internal gain and feedback resistors and an easy-to-use flow-through pin layout. The LT6411 can produce gains of 1, –1, and 2 with no external components. The dual amplifiers inside the LT6411 allow for easy singleended-to-differential conversion for driving high-speed analog-to-digital converters (ADC). The wide bandwidth (650MHz), low distortion (–77dBc harmonic distortion at 30MHz) and high slew rate (3300V/µs) preserve signal fidelity even at high frequencies, while the low supply current (16mA total) enables the LT6411 to be used in power-critical high-speed signal chain applications. Form factor is also not an issue—all of these features fit When blazing fast speed, low power, or the flexibility of selectable gains is necessary, a circuit designer need only reach into his bag of tools and pull out the LT6411. into the tiny 3mm × 3mm 16-pin QFN package. When blazing fast speed, low power, or the flexibility of selectable gains is necessary, a circuit designer need only reach into his bag of tools and pull out the LT6411. Internal Topology and Gain Selection The LT6411 contains two internal current-feedback amplifiers with matched feedback and gain resistors. The integrated 370Ω resistors take the guesswork out of selecting the optimal feedback resistor for good AC response. Another common source of frustration with current-feedback amplifiers is maintaining a tight PC board layout to prevent excessive capacitance at the inverting input node. This node is +V IN + IN – LT6411 + – – + Single-Supply Operation and Level-Shifting The LT6411 operates on a wide supply voltage range, from 4.5V–12.6V. Figures 1–3 show the part with dual supplies. However, the LT6411 performance remains excellent with a +V IN + OUT – –V +V LT6411 OUT + Figure 1. Standard non-inverting gain of 2 configuration, shown with split supplies. 6 internal to the LT6411, so the layout of the circuit board is nearly optimal from the get-go. Selecting between the basic gain configurations of the LT6411 is a simple matter of pin strapping. Figure 1 shows the standard non-inverting gain of 2 configuration, with differential input and output. Figure 2 shows the non-inverting gain of 1 configuration, and Figure 3 shows an inverting gain of 1 configuration. Note that in Figure 2, the non-inverting inputs are tied together with the gain resistor. In theory, the gain resistor pin could be left floating, but in practice, the parasitic pin and pad capacitances cause the gain to peak up to 6dB at high frequencies, which causes excessive ringing in the transient response. At 650MHz (the bandwidth of the LT6411), 2pF of parasitic capacitance has a reactance of just 122Ω. LT6411 + – OUT + – + OUT – IN – IN + –V Figure 2. Standard non-inverting gain of 1 configuration, shown with dual supplies. The gain resistors, shown here tied to the inputs, should not be left floating (see text). + – IN – – + OUT – OUT + –V Figure 3. Standard inverting gain of 1 configuration, shown with dual supplies. Linear Technology Magazine • June 2006 DESIGN FEATURES L single supply, especially important for many common applications such as high speed ADC driving, where only a single supply is available. If the input signals into the LT6411 are already DC level-shifted above ground so that the input and output common-mode ranges are met (the input operates to within 1V of the supplies, the output with a 1k load swings to within 1.3V of the supplies), no additional work needs to be done. However, if the input signal is AC-coupled or centered around ground, then level-shifting is necessary. This section presents some of the methods to level-shift the input and output voltages of the LT6411. Figure 4 shows a center-tapped transformer providing the DC voltages necessary for single-supply operation. The input signal (shown single-ended, but can also be differential by driving the other end of the primary) is provided through the transformer primary, and the transformer secondary presents a balanced signal to the LT6411. If the input signal is coming from a 50Ω signal source, the two 24.9Ω resistors provide the appropriate termination. Note that the DC voltage provided at the center tap of the transformer is VO(DC)/2, due to the fact that the LT6411 has a non-inverting gain of 2. Alternatively, the two gain resistor pins could be AC-coupled to ground through a capacitor, and the DC gain of the LT6411 would be unity. The total differential gain in this configuration is 2; if a differential gain of 1 is desired, simply tie the gain resistor pins to the corresponding non-inverting input pins. For a single-ended input, Figure 5 shows a simple AC-coupled method of providing the correct input and output DC levels for the LT6411. The input is AC-coupled through a large capacitor (typically 0.1µF or larger), and the total differential gain at the output is +2. The DC voltage is provided through the second non-inverting input. In the case of a differential inputs, Figure 6 shows a similar configuration, where the two inputs are AC-coupled to the LT6411, and the DC level is provided through two resistors. The choice of size for the resistance R is Linear Technology Magazine • June 2006 V+ LT6411 MIA-COM ETC1-IT 1:1 IN + – 24.9Ω • • OUT + VO,DC VOLTAGE GAIN = 2 24.9Ω – + OUT – VO,DC VO,DC 2 Figure 4. DC level-shifting through the center tap of a transformer for single-supply operation. The DC voltage source at the center tap should serve as a low-impedance AC ground. The two 24.9Ω resistors provide a 50Ω termination, if necessary. V+ IN CLARGE LT6411 + – fCO ≈ OUT + VO,DC 1 2π • 370Ω • CLARGE – + VO,DC OUT – VO,DC Figure 5. A simple AC-coupled method of providing DC level shifting. The DC voltage source, which does not need to have a low impedance, is provided at the non-inverting input of the second amplifier. The AC input impedance of this circuit is 370Ω. +V IN + LT6411 CLARGE + – fCO ≈ 1 2πCLARGE • R IN – OUT + VO,DC R – + CLARGE OUT – VO,DC R VO,DC 2 Figure 6. A DC voltage source and two resistors sets the DC input and output level of the LT6411. The two resistors must be large enough not to overload the inputs. a trade-off between loading the differential inputs (for smaller values of R) and increasing voltage offset and noise (for larger values of R, due to the input bias currents and current noise in the non-inverting inputs). Values of R up to 10k work well in practice. The DC and AC inputs both have a gain of 2 in this configuration. For a gain of 1, simply tie the gain resistor pins to the corresponding non-inverting input pins. Figures 4–6 all have a lower frequency limitation, defined by either the transformer’s magnetizing inductance or the size of the AC-coupling capacitor. What if voltage level-shifting and response down to zero Hertz 7 L DESIGN FEATURES V+ IN + VI,DC RB +V RB LT6411 + – VO,DC IN – VI,DC RA V+ – + RA VO,DC OUT + VO,DC GAIN DIFF = 2 • VO,DC VO,DC = OUT – RB RA + RB (V+ • RA ) + (VI,DC • RB) R A + RB VO,DC Figure 7. Two identical resistive dividers (RA and RB) shift the DC level of the input (and output) to within the limits of the LT6411. VO,DC must have a low impedance at the frequencies of interest, and must be capable of sourcing and sinking currents through the internal resistors. (DC) is necessary? Figure 7 shows one method of level-shifting and maintaining signal response down to DC. The RA-RB resistive dividers set the input common-mode voltage for the LT6411. Choose RA and RB so that the common-mode voltage at the input of the LT6411 is the same as VO(DC), the desired output common-mode voltage. The gain of the LT6411 with the resistive divider is 2 • RB/(RA + RB), and this circuit has a response to DC. If the input DC common-mode level (VI, DC) is greater than the VO,DC, then RB should be attached to ground instead of the positive supply. (that is, VDC in the Figure 11 must be appropriate for both the amplifier and the ADC), or that one of the techniques from the previous section must be used to level-shift the input. If dual supplies are used for the LT6411, then interfacing with a single-supply ADC might require AC-coupling at the output, as shown in Figure 9. The DC level of the ADC input is established by VDC and two 499Ω resistors. Figure 8’s circuit, though simple, has an important drawback: all of the wideband noise of the amplifier couples into the ADC input, and thus degrades the signal-to-noise ratio (SNR) of the signal. In most cases, the input signal is limited to some frequency band less than the DC–650MHz bandwidth of the LT6411; thus, any extra bandwidth beyond that introduces unnecessary noise. The simplest approach to fixing this problem aptpears in Figure 10. A single shunt capacitor creates an RC lowpass filter that limits the noise bandwidth of the amplifier output, improving the SNR. For an even sharper cutoff lowpass filter, Figure 11 shows a more involved approach. The inductors and capacitor create a second-order lowpass filter, with R1 ensuring that the frequency peaking is not excessive. R2’s primary function is to ensure that the ADC +V IN + LT6411 + – VDC 24.9Ω High Speed ADC Driving Modern high resolution, high speed ADCs typically feature differential inputs with capacitive sample-and-hold circuits. Driving these inputs requires an amplifier with high bandwidth, fast settling times, good transient response and good distortion performance at the frequencies of interest. The figures in this section show various configurations designed to get the maximum performance out of the LT6411. The LT6411 is optimized for driving 12-bit and 14-bit high-speed converters such as the 14-bit, 80Msps LTC2249. For many applications, the only additional recommended interfacing components would be small series resistors, to help isolate the LT6411 from the ADC’s capacitive input. Figure 8 shows the simplest configuration for the LT6411 driving the ADC. The single supply operation means that either the inputs must have an appropriate DC common-mode level 8 ADC IN – – + VDC 24.9Ω Figure 8. LT6411 shown interfacing to an ADC. A small series resistance is recommended to isolate the ADC’s capacitive input. +V IN + fCO ≈ LT6411 + – 0V CLARGE 1 2π • 499Ω • CLARGE 499Ω IN – 0V – + ADC CLARGE 499Ω –V VDC Figure 9. Level-shifting the LT6411 output to within the input common-mode range of the singlesupply ADC. The resistors used must be large enough not to excessively load the outputs of the LT6411. Some ADCs have a VCM output that can be used for VDC. Linear Technology Magazine • June 2006 DESIGN FEATURES L Conclusion The LT6411 selectable-gain amplifier/ADC driver is extremely flexible, featuring a multitude of possible configurations with a minimal number of external components. The LT6411 is a dual op amp, a differential ADC driver, and a selectable-gain amplifier all in one. In most basic dual op amp functions, all that is required is a few power supply bypass capacitors to get excellent AC performance to over 600MHz. In addition, the LT6411 comes in the tiny 3mm × 3mm 16-lead QFN package and consumes only 16mA total. The LT6411 also has a shutdown feature that reduces the power supply current to 700µA total. With this unique set of features, the LT6411 can provide myriad functions without breaking the budget of size or power. L for the latest information on LTC products, visit www.linear.com Linear Technology Magazine • June 2006 +V LT6411 + – R C – + ADC R fCUTOFF = (4πRC)–1 Figure 10. An RC lowpass filter is shown at the output of the LT6411. Reducing the bandwidth at the output also reduces the amount of wideband noise seen by the ADC. V+ R2 LT6411 + – L R1 C R2 – + ADC L R1 fCUTOFF > 2π 2LC –1 R1 + R2 ≤ 150Ω Figure 11. A second-order LC lowpass filter, offering a flatter passband and sharper stopband rolloff than an RC filter. The series resistor R1 controls peaking near the cutoff frequency, and the parallel resistor R2 ensures that the ADC sees a low source impedance at very high frequencies. V+ LT6411 + – R1 54.9Ω R1 54.9Ω 6 L 390nH R2 80.6Ω – + 9 R2 80.6Ω 3 C 15pF GAIN (dB) inputs do not see too high of a source impedance. Figure 12 shows some sample values, configured for a cutoff of around 50MHz and almost no gain peaking near the cutoff frequency. Simulating the filter is a good way to determine optimal component values, especially when taking into account the series resistance of the inductor and component tolerances. Some high frequency applications contain very narrow-band signals, where a bandpass filter would provide the best noise limiting, and thus the highest SNR. Figure 13 shows a simple RLC bandpass filter. The value of R determines the quality factor (Q) of the filter—the larger the resistor value, the more narrow-band the filter. This comes at the cost of more pass-band loss (depending on the parasitic components of the inductor and capacitor) and higher sensitivity to component tolerances and variations. As the bandpass filter gets narrower, a small shift in center frequency can significantly attenuate the desired output signal. LTC2249 0 –3 –6 L 390nH –9 –12 1 10 100 FREQUENCY (MHz) 1000 Figure 12. The circuit of Figure 11, configured for a cutoff frequency of around 50MHz. V+ LT6411 + – R L – + C ADC R fCENTER = (2π√LC)–1 Figure 13. For narrow-band applications, an LC bandpass filter does an excellent job limiting the noise at the ADC input while maintaining a low component count. The series resistor affects the width of the pass-band and the pass-band attenuation. 9 L DESIGN FEATURES Dual Step-Up Converter Drives White LEDs with 1000:1 PWM Dimming by Keith Szolusha Introduction Notebook computers, large-screen handheld PDAs, dashboard displays, and automotive and avionic in-cabin entertainment LCD panels are illuminated with strings of high power white LEDs. White LEDs are the preferred over other backlight technologies because they provide true white light at a high enough intensity for daylight viewing, and enough dimming capability for nighttime use. LEDs also offer relatively long life spans and a lack of hazardous materials. LED strings lining the edges of these LCD panels provide uniform brightness when driven with a constant current. The maximum switch voltage of an LED driver limits the number of LEDs that it can drive in series. It may seem that paralleling LEDs is a good way to increase the capacity of a driver IC, but parallel LEDs must be wellmatched in forward voltage; otherwise un-matched LED strings cause uneven currents and thus uneven brightness. LEDs can be specially sorted (binned) for matching characteristics, but this increases cost. VIN 4V TO 16V 10µH ZLLS400 A915AY-100M 10µH A915AY-100M COUT1 2.2µF 35V 100mA 8 TO 10 WHITE LEDs <34V SW1 CTRL1 BRIGHTNESS ADJUST SHDN OVP2 CTRL1 CTRL2 PWM1 100k COUT2 2.2µF 35V 2.8k CTRL2 BRIGHTNESS ADJUST 0.1µF REF FDN5630 PWM2 FB1 VC1 GND RT PWM1 100mA 8 TO 10 WHITE LEDs <34V SW2 VIN SHDN LT3486EFE FDN5630 RSENSE1 2Ω 1% Dual LED String Step-Up Driver ZLLS400 CIN 4.7µF 25V OVP1 FB2 VC2 63.4k PWM2 100k RSENSE2 2Ω 1% 2.8k PWM INPUT 4700pF 100Hz 1000:1 DIM RATIO PWM INPUT 100Hz 1000:1 DIM RATIO 4700pF Figure 1. LED driver uses 4V–16V input to drive two strings of eight-to-ten 100mA LEDs (less than 34V total in string) with 1000:1 PWM dimming 90 100 85 EFFICIENCY (%) 50 40 30 20 10 LT3486 2× 10× WHITE 100mA LEDs 0 8 9 10 11 12 13 14 15 16 17 18 INPUT VOLTAGE (V) 100 80 80 LED CURRENT 75 60 70 40 65 60 VIN = 12V 2× 8× LEDs 0 20 40 60 80 PWM DUTY CYCLE (%) 20 0 100 Figure 2. Efficiency of the circuit in Figure 1 and efficiency as a function of PWM duty cycle 10 LED CURRENT (mA) 60 EFFICIENCY (%) 80 70 120 EFFICIENCY 90 A better solution is a dual channel LED driver to drive two strings of LEDs. This saves the space and cost of duplicating components, such as the driver IC and input capacitors. Each string is driven with the same regulated constant current, thus providing uniform brightness. One IC that has these features is the LT3486 dual LED string driver, which has two 1.3A channels with high PWM dimming capability in a small 5mm × 3mm DFN package. Since both channels’ power switches are included in the IC, the circuit is simple and small. The LT3486 is a dual step-up LED driver. Each channel has an efficient, low side 1.3A npn power switch with low VCE(sat) of 300mV (at 750mA switch current). The IC is designed to drive a string of LEDs from a wide input voltage range. Each LED string total voltage can be as high as 38V in a typical application, but may be limited to 34V if the overvoltage protection (OVP) pin is used to protect the switch when the LED string is open. Figure 1 demonstrates the LT3486 as a dual LED string step-up converter driving a total of 16–20 white LEDs at 100mA from a 4V–16V input voltage range source. The total voltage of the LEDs cannot exceed 34V. The circuit is kept small and simple with the single ceramic input capacitor and two small ceramic output capacitors. With a high 800kHz switching frequency, the inductors and capacitors can be small in size while the efficiency of the circuit remains high, as shown in Figure 2. As PWM duty cycle is decreased from 100%, the circuit efficiency drops slightly, but remains high during the PWM on-time. Not only is the operating efficiency high, but the converter shutdown current Linear Technology Magazine • June 2006 DESIGN FEATURES L FEEDBACK VOLTAGE (mV) 250 VIN = 3.6V TA = 25°C ILED 200mA/DIV 200 IL 500mA/DIV 150 PWM 5V/DIV 100 VIN = 12V 0.2ms/DIV 8/8 LEDs PWM FREQ = 1kHz 50 Figure 4. PWM dimming waveforms 0 0 1 0.5 1.5 CONTROL VOLTAGE (V) function of CTRL pin voltage. The low 200mV FB pin (and current sense voltage) accuracy is typically 3% at full current with the CTRL pin pulled high (above 1.5V) but as the CTRL pin voltage is lowered to 150mV, the FB pin voltage is also reduced to about 40mV. Below this 5:1 dimming ratio, the LEDs are turned off as the CTRL pin voltage is pulled below 75mV. Another method of reducing the brightness of the LEDs is digital PWM dimming. The PWM MOSFET in series with the LEDs creates the waveform shown in Figure 4 when the string of LEDs is PWM’d at 100mA constant current. During PWM on-time, the current is a well-regulated 100mA. During PWM off-time, the current is zero. Because the current is either 100mA or zero, the LED color is preserved as if the LED were driven by a constant 100mA current. Dimming is simply a function of the average, instead of instantaneous, current. The advanced PWM function in the LT3486 2 Figure 3. FB pin voltage vs CTRL pin voltage consumption is less than 1µA (typically 100nA), merely sipping from the battery when the IC is off. 1000:1 PWM Dimming and 10:1 Brightness Control As shown in Figure 1, LED brightness can be controlled on the LT3486 with an analog voltage input to the CTRL pin or a digital PWM signal to the gates of the PWM dimming MOSFET and the PWM pin. Analog brightness control reduces the LED current from 100mA to a lower value by reducing the internal sense resistor voltage. Although this is a simple way to decrease the brightness of the LED, the accuracy of the LED current control is reduced and the chromaticity of the LED changes at lower currents. The graph in Figure 3 displays the LT3486 typical FB pin voltage dropping as a CIN 10µF 5V D1 D2 L1 10µH COUT1 2.2µF SW1 25mA L2 10µH VIN COUT2 2.2µF SW2 OVP1 OVP2 CTRL1 CTRL2 OFF ON SHDN REF PWM1 LT3486 1.25V REF PWM2 FB1 CREF 0.1µF FB2 RT VC1 2.8k 8.06Ω 25mA 4.7nF VC2 RT Doubler Delivers Greater than 34V to LED Strings GPS navigation and in-cabin entertainment displays are increasingly popular in mainstream consumer vehicles. The advantage of using two LED drivers each with 8-LED strings, instead of a single 16-LED string, is that the maximum switch voltage remains that of a single 8-LED string (less than 34V total string voltage at 100mA). Even so, LCD panel screen sizes are pushing beyond the standard 6" and 7", requiring more LEDs and string voltages above 34V. The circuit in Figure 7 uses a charge pump voltage doubler to drive two strings of LEDs to voltages as high continued on page 44 2.8k 4.7nF 8.06Ω CIN: 10V, X7R COUT1, COUT2: 35V, X5R D1, D2: ZETEX ZHCS400 L1, L2: TOKO D53LC TYPE A Figure 5. LED driver uses 5V input to drive two strings of eight 25mA LEDs (less than 34V total in either string) with 5:1 brightness control Linear Technology Magazine • June 2006 is particularly fast in returning the LED to its programmed LED current. Its short minimum dimming on-time (10µs on-time) allows a 1000:1 digital PWM dimming ratio with 100Hz PWM frequency—fast enough to avoid visible flicker. For instance, a combination of two LT3486s driving four LED strings (R-G-G-B) in a top-end display provides 1000:1 dimming while maintaining the true-color of the display even during very dim nighttime operation. When a PWM signal is used for brightness control, but less than a 5:1 dimming range is needed and the chromaticity of the LEDs is not especially important, the PWM signal can be fed into an RC filter such as the one in Figure 6. This turns the PWM input into an analog CTRL pin voltage controlling the LED current directly, eliminating the need for the PWM dimming MOSFETs. The 5V, 16-LED converter in Figure 5 can deliver up to a 5:1 analog dimming range at the CTRL pins with such a filter without the need for the two additional PWM dimming MOSFETs. In this case, the LT3486 PWM pins are tied high to the 1.25V REF pin. PWM 10kHz TYP LT3486 R1 10k C1 1µF CTRL1,2 Figure 6. Achieving 5:1 brightness control with a filtered PWM signal 11 L DESIGN FEATURES Hot Swap Controller Monitors and Reports Power Supply Status by Josh Simonson Z1 SA14A CONNECTOR 1 High availability systems are designed to achieve an ideal of zero down time. To approach this goal, the system needs to be able to operate during routine maintenance and upgrades, which often involves cards being inserted and removed from a live backplane. These systems must also be designed for failsafe operation by isolating faulty boards before they cause backplane disturbances. Hot swapping requires a power switch to initially isolate the board, and a controller to turn on the switch slowly to minimize backplane disturbances. Since the Hot Swap controller monitors card voltage and current, it is an obvious place to integrate higher level monitoring with a data converter. This provides detailed information about the health of the power path and the power consumption of downstream circuits. Such information RS 0.005Ω VIN 12V CONNECTOR 2 Introduction CF 0.1µF R1 34k 1% Q1 FDC653N R5 10Ω R2 1.02k 1% UV VDD SENSE+ SENSE– GATE SOURCE FB OV ON ADIN SDAI GPIO LTC4215UF SDAO EN SCL SS ALERT R3 3.4k 1% SDA SCL ALERT TIMER INTVCC ADR0 ADR1 ADR2 GND CTIMER 0.68µF GND C3 0.1µF R7 30.1k 1% R8 3.57k 1% + VOUT 12V CL 330µF R4 24k CSS 6.8nF NC BACKPLANE PLUG-IN CARD Figure 1. In a typical application the LTC4215 uses an external N-channel pass transistor to isolate the hot swapped board from the backplane when it is first inserted. After a debounce time the controller can begin to apply power to the board or wait for a turn-on command from a host processor. Power is ramped gradually to minimize any backplane disturbance. After the power-up process is complete, the LTC4215 continues to monitor for faults in the power path. can be used to monitor performance over time and identify boards that are drifting towards failure or marginal performance. The LTC4215 combines a robust Hot Swap circuit with an I2C interface and data converter to allow power monitoring as well as hot-plug functionality Table 1. A few of the LTC4215’s many features Feature Benefits Wide Input Voltage Range: Operates from inputs of 2.9V to 15V, with 24V absolute maximum ❏ Suitable for 3.3V, 5V and 12V systems 8-bit ADC: ADC monitors current, output voltage and external pin voltage and measures off-state current in the FET to determine FET failures ❏ Increases reliability ❏ Simplifies design because part functions on a semi-regulated supply ❏ Large overvoltage transient range eases design tolerances for transient protection ❏ Board power information provides an early warning of board failure ❏ Verify board is staying within its allotted power ❏ Allows integrity check of redundant supply paths ❏ Allows active power management to safely maximize power utilization within the chassis cooling constraints I2C/SMBus: Communicates as a read-write ❏ Improves integration with the host system. Interface allows the host to configure the part, determine which faults are present or have occurred, and read back ADC slave device using a 2-wire serial interface measurements Fast Short Circuit Response: Fast (<1µs) current limit response to shorts ❏ Protects connector from overcurrent Alerts Host after Faults: When configured (using I2C), faults activate an active pulldown on the ALERT pin ❏ Interrupting the host for immediate fault servicing limits system damage 12 ❏ Limits the disturbance to the input supply from a short circuit ❏ Reduces the bus traffic for polling Linear Technology Magazine • June 2006 DESIGN FEATURES L and fault isolation (see Table 1). In a typical application the LTC4215 uses an external N-channel pass transistor to isolate the hot swapped board from the backplane when it is first inserted (Figure 1). After a debounce time the controller can begin to apply power to the board or wait for a turnon command from a host processor. Power is ramped gradually to minimize any backplane disturbance. After the power-up process is complete, the LTC4215 continues to monitor for faults in the power path. The LTC4215 provides the means for quantitatively measuring the board current and voltages with an onboard ADC and multiplexer. It reports this information using the I2C serial communication bus when polled by a host processor. The device interrupts the host for specific fault conditions, if configured to do so. The LTC4215 works in applications from 12V (with transients to 24V) down to 3.3V where the operating input voltage could drop to 2.9V. Functionally, the LTC4215 is very similar to the LTC4260 (Linear Technology, Nov. 2004) operating in a lower voltage range. Table 2 compares the major features of the LTC4215 and LTC4260. Special attention should be paid to the power-up sequence because of the added soft start pin and some changes in the function of the TIMER pin relative to the LTC4260. Since both parts can be used for 12V systems, Table 2 may be used to select the part with optimal set of features for a specific 12V application. An N-channel pass transistor, Q1, controls the application of power to the board as in Figure 1. A series sense resistor, RS, allows the LTC4215 to measure the current in the power- Table 2. Comparison of the LTC4260 and LTC4215 Feature LTC4215 LTC4260 VDD Abs Max 24V 100V VDD Min 2.9V 8.5V Recommend TransZorb for 12V Yes No Current Limit/Circuit Breaker 25mV 50mV Circuit Breaker Precision 10% 10% Current Limit Precision (FB = 0) 35% 25% Current Limit vs Circuit Breaker VTH 75mV, 25mV Both 50mV Current Limit Foldback Only during startup Always Gate RC Network Optional Required Soft Start Yes, Required No Timer Pin Optional Required OC Timer 20µs External/Adjustable ∆VGATE at 12V (0µA) 6V 8.5V Built In Overvoltage Threshold 15.6V None ADC Source LSB 60mV 400mV ADC VSENSE LSB 151µV 300µV ADC ADIN LSB 4.85mV 10mV Internally Generated VCC 3.1V 5.5V Package 4mm × 5mm QFN 5mm × 5mm QFN Linear Technology Magazine • June 2006 path. Resistor R5 suppresses self oscillations in Q1. Resistors R1–R3 select the undervoltage (UV) and overvoltage (OV) thresholds. Capacitor CF allows these thresholds to be filtered as needed. R7 and R8 select the powergood threshold and set the foldback current limit level. Capacitor CSS sets a maximum slew rate to control the inrush current and CTIMER is used to set the startup time. C3 is used to bypass the internal core voltage. Typically, the pins on the connector are staggered so that bulk power is applied first with the longest pins, followed by communication lines on medium length pins, and last, Hot Swap control lines such as the supply for the UV, OV, or EN pins. The UV, OV and EN pins must be in the correct state for a programmable debounce period of 100ms before Q1 is allowed to turn on. At this point the ON pin turns the part on immediately if it is high, or holds the part off if it is low. When the ON pin is held low, Q1 is turned on through the I2C bus by writing to the ON bit in the control register. Measure Real-Time Board Power with Integrated ADC Monitoring the supply voltage and current in real-time is a useful way of tracking the health of the power path. New data can be compared with historical data for the same card to detect changes in power consumption that could indicate that the card is behaving abnormally. An abnormal card can be shut down and flagged for service, perhaps before a more severe fault or system malfunction occurs. The LTC4215 includes an 8-bit data converter that continuously monitors three voltages: the ADIN pin, the SOURCE pin and the current sense voltage between the SENSE+ (VDD) and SENSE– pins. The ADIN pin is an uncommitted ADC input which allows the user to monitor any available voltage. The ADIN pin is monitored with a 1.235V full scale. The ADIN pin is connected directly to a data converter input without any signal scaling. The SOURCE pin uses a 1:12.5 divider at the input which gives a 15.4V full 13 L DESIGN FEATURES scale. The SENSE voltage amplifier has a voltage gain of 32, which results in a 38.4mV full scale. The converter uses a sophisticated oversampling and offset cancellation method that preserves the full 8-bit dynamic range on the SENSE channel. If the data converter reads more than 1mV on the VDD-SENSE channel while the external switch is turned off, the LTC4215 generates a FET-SHORT fault to indicate that the switch may be damaged. The presence of this condition is indicated in STATUS register bit C5 and logged to FAULT register bit D5. The LTC4215 takes no action in this condition other than logging the fault and generating an alert if configured to do so. The results from each conversion are stored in three ADC registers (see Table 3) and updated 10 times a second. Setting the test mode control register bit halts the data converter so that the registers can be written to and read from for software testing. Versatile Inrush Current Control Once the inputs to the LTC4215 reach the correct values for the part to turn the external switch on and an internal 100ms debounce timer has expired, the LTC4215 turns on. The startup time is determined by the capacitor on the TIMER pin, or 100ms if the TIMER pin is tied to VCC. During this time the circuit breaker is disabled to prevent an overcurrent fault from occurring, the power-good signal from the GPIO pin is also disabled to prevent turning on a load before the current limit has reached the full value via the Soft Start and Foldback pins. The inrush current slew rate (dI/dt) is limited via the SS pin. The inrush current is also folded back from 25mV to 10mV via the FB pin. An optional RC network on the external MOSFET gate can be used to set the inrush current below the foldback level by setting the maximum slope of the output voltage. The various inrush current profiles obtainable by these three methods are detailed in Figures 2 thru 6, which show a 12V system with a 25mΩ sense resistor, 14 Table 3. LTC4215 registers Register Description CONTROL Register turns-on or turns-off the pass transistor and controls whether the part will auto-retry or latchoff after a fault. It also configures the behavior of the GPIO pin. ALERT Alert register enables which faults interrupt the host using the ALERT pin. At power-up the default is to not alert on faults. STATUS Status register provides pass transistor (on/off), EN (high/low) and GPIO (high/low) conditions. It also lists five fault present conditions. FAULT Fault register logs overcurrent, overvoltage, undervoltage, power-bad, FET short and EN changed state faults. SENSE ADC data for the VDD-SENSE voltage measurement. SOURCE ADC data for the SOURCE pin voltage measurement. ADIN ADC data for the ADIN pin voltage measurement. or 1A current limit, starting up into a 470µF capacitive load. At the end of the startup period the current limit circuit is checked. If the current limit is still regulating the current, the LTC4215 determines that the output failed to come up and generates an overcurrent fault. If the current limit circuit is not active then the current limit threshold is moved to 75mV, the power-good signal to the GPIO pin is enabled and the 25mV circuit breaker is armed. The SS pin sets the current slew rate limit at startup. It starts at ground, which corresponds to a negative voltage on the sense resistor and results in the MOSFET being turned off. A current into the soft-start capacitor produces a ramp that corresponds to increasing VDD-SENSE voltage. When the current limit circuit releases the gate (when the commanded VDDSENSE voltage becomes positive) the current from the SS pin is stopped to wait for the GATE pin to rise and start to turn on the MOSFET. Once the current limit circuit begins to regulate the VDD-SENSE voltage, the current from the SS pin is resumed and the ramp continues until it reaches the foldback level. It is important that the SS pin stop the ramp while the GATE pin slews because the ramp would otherwise continue and result in an uncontrolled step in current once the MOSFET threshold is reached. An uncontrolled step may violate inrush specifications and cause supply glitches on the backplane. If the soft-start ramp reaches the foldback level, the foldback circuit stops the ramp, as shown in Figure 5. The ramp is allowed to continue as the voltage at the FB pin rises and increases the foldback current limit, still limited in slope and limited in magnitude by foldback as well. If an RC network is placed on the GATE pin to manually set the inrush current to a value below the foldback level (Figure 4), the current limit circuit will leave regulation when it is unable to achieve the VDD-SENSE voltage commanded by the SS and FB pins. If the startup timer expires during this inrush an overcurrent fault is not generated because the current limit is not active. The power-good output for GPIO is allowed to relay the state of the FB pin, and the circuit breaker is armed. Either the output voltage finishes rising and a power good is asserted when the FB pin crosses it’s 1.235V threshold, or the current rises to the circuit breaker threshold and the part generates an overcurrent fault. In the event there is an overcurrent condition after startup, the current limit circuit limits the VDD-SENSE voltage to 75mV while the circuit breaker waits for a 20µs timeout before producing an overcurrent fault. After any overcurrent fault, the part waits Linear Technology Magazine • June 2006 DESIGN FEATURES L VGATE 5V/DIV VOUT 5V/DIV VGATE 5V/DIV VOUT 5V/DIV IINRUSH 500mA/DIV IINRUSH 200mA/DIV 2ms/DIV 5ms/DIV Figure 2. Inrush current is limited by foldback. This allows the fastest startup of the load, with the inrush only lasting 9ms with a peak current of 1A. Figure 3. Inrush current is limited by a 1µF SS capacitor. This provides the fastest startup in a system with demanding inrush current slew rate requirements. Inrush dI/dt is reduced to 12mA/ms. VGATE 5V/DIV VOUT 5V/DIV VGATE 5V/DIV VOUT 5V/DIV IINRUSH 100mA/DIV IINRUSH 500mA/DIV 10ms/DIV 2ms/DIV Figure 4. Inrush current is limited by a 0.1µF GATE capacitor. This minimizes the power in the switch, allowing the use of smaller components at the cost of speed. The inrush current is only 100mA, but startup takes 55ms. Figure 5. Inrush current is limited by a 68nF SS capacitor and FB. Soft start controls the inrush current slew rate while the current limit is modulated by foldback. This allows the fastest startup while also protecting the backplane from current surges. VGATE 5V/DIV VOUT 5V/DIV IINRUSH 100mA/DIV 10ms/DIV Figure 6. Inrush current with 1µF SS capacitor and 0.1µF GATE capacitor. Soft start limits the inrush current slew rate until the GATE capacitor limits the inrush current by limiting the dV/dt at the output. This minimizes the power in the switch while protecting the backplane from inrush current surges. for a cool-down period of 50 times the startup time before allowing the part to restart by any means, including auto-retry, I2C, or cycling the EN, UV or ON pins. Controlled Turn-Off When the LTC4215 is turned off by a fault or I2C transaction, the GATE pin is pulled down with a 1mA current source. Once the GATE pin is below the SOURCE pin, a diode from SOURCE to GATE turns on and the voltage at the SOURCE pin is discharged by the same 1mA current. If there is a short that causes the sense voltage to exceed 75mV, a 400mA pull-down from GATE to SOURCE removes the gate charge of the switch. Once the sense voltage falls to 75mV, the current limit regulates there for Linear Technology Magazine • June 2006 20µs before turning the gate off with the 1mA current source. If there is significant inductance between VDD (SENSE+) and bulk capacitance, across a connector for instance, it is possible that a short circuit at the output with a very fast rise time could cause the input voltage to collapse while the current through this inductance slews. In this case, after 2µs, the VDD undervoltage lockout circuit turns on and discharges the GATE pin with the 400mA pull-down to the SOURCE pin and quickly turns the switch off. Save Power with Precise 25mV Circuit Breaker For supplies with lower voltages and higher currents, a 50mV circuit breaker threshold may result in too much power dissipation in the sense resistor, or cut excessively into the input supply voltage tolerance of downstream circuits. To reduce this problem the LTC4215 has a precision circuit breaker at 25mV with a low 10% tolerance. This allows the use of smaller and less expensive sense resistors with lower power ratings. In systems where the circuit breaker has only 20% accuracy the designer must be able to safely provide 40% more power than the card actually consumes to ensure that the slot doesn’t suffer from heat and supply limitations on the high side or produce a fault in normal operation on the low side. The precise 10% accuracy of the LTC4215 cuts this guard-band in half and safely allows the use of 20% more continued on page 37 15 L DESIGN FEATURES Efficient Buck-Boost Converter Ideal for Power Saving Modes and Wide by Kevin Ohlson Input Voltage Ranges Introduction Portable handheld electronic tools, gadgets, and toys are approaching the multi-function equivalent of a Swiss army knife. Previously separate functions are now miniaturized and combined into single pocket sized package. Unless owned by a teenager or a Hollywood agent, most gadgets spend most of their time in a low power state, waiting to come to life to make a phone call, pulse a photoflash or spin music and video to a tiny hard disk drive. These devices are powered from a variety of power sources, which means at some point the input voltage is higher than or lower than the 3.3V or 3.6V voltage needed to power the internal electronics. The DC/DC converters in the latest portables must be able to step up and step down voltage, maintain very high efficiency during idle and standby modes, and respond quickly and efficiently during peak power demands. 3 Figure 1. Lithium ion battery to 2.5V–5.25V converter in 1.4cm2 Linear Technology offers a family of buck-boost converters capable of supplying from 200mA to 2A with excellent efficiency. The latest addition to this lineup is the LTC3532, a 300mA buck-boost converter, which incorporates automatic Burst Mode operation, adjustable switching frequency, and integrated soft-start. The LTC3532 is ideal for miniature disk- 4 SW1 SW2 SW D SW A VOUT 6 SW B GATE DRIVERS AND ANTICROSS CONDUCTION 2 SW C PHASE CONTROL ERROR AMP BURST – VIN Features + 7 drive applications or any application that requires high efficiency over a wide range of output currents and input voltages. 1.22V FB 9 + – 1A 10 VC Figure 2. The LTC3532 four switch buck-boost converter uses a single inductor and features peak current clamp and automatic Burst Mode control. 16 The input voltage range of a LTC3532based converter is 2.4V to 5.5V and its output range can be programmed from 2.4V to 5.25V, making it ideal for devices operating from multiple sources such as battery, USB, and wall adapters. The LTC3532 is available in either an MS10 package, which is pin compatible with the LTC3440 converter, or in an exposed pad 3mm × 3mm DFN. With these tiny packages, an entire converter can be squeezed into the smallest spaces, as shown in Figure 1. Using a fixed frequency fourswitch architecture and a patented control method, the converter needs only a single inductor to regulate a constant output voltage with input voltages greater than or less than the output. The four switch topology of the LTC3532 (see the output stage schematic in Figure 2) allows the regulator to smoothly transition from buck mode to buck-boost mode and boost mode by correctly phasing the four output switches (A, B, C, and D) in response to the error amp output voltage, VC. During buck mode, switch D is on while switches A and B act like a buck converter. At the other extreme, boost mode, switch A is always on while switches C and D implement a synchronous boost converter. When VIN and VOUT approach the same voltage, all four switches commutate with the on time for each pair controlled by the voltage at VC. The four switch architecture inherently provides output disconnect, which prevents current flow between VIN and VOUT in shutdown mode. Linear Technology Magazine • June 2006 DESIGN FEATURES L 100 EFFICIENCY 100 80 10 70 POWER LOSS 1 60 3V 3.6V 4.2V 50 V = 3.3V 40 OUT 0.1 1 Li-Ion to 3.3V Converter Ideal for Miniature Hard Disk Drives IL 500mA/DIV ILOAD 500mA/DIV 0.1 VOUT = 3.3V 0.01 1000 10 100 LOAD CURRENT (mA) SHDN/SS pin reaches 1V, the internal control voltage is clamped until the pin rises to 2V. VOUT 1V/DIV POWER LOSS (mW) EFFICIENCY (%) 90 1000 Figure 3. High efficiency is possible over a wide range of load currents using automatic Burst Mode control. The capabilities of the four switch architecture is exploited in the LTC3532 when in Burst Mode operation as well. An innovative (patent pending) Burst Mode control circuit optimizes the firing of the four switches depending on whether in buck mode, boost mode, or buck-boost mode. Optimal switching control and a low 35µA Burst Mode quiescent current means the converter increases the battery life of a system by keeping efficiency above 80% in Burst Mode operation at loads as low as 300µA. Measured efficiency over a load current ranging from 0.1mA to 500mA is shown in Figure 3. A resistor and filter capacitor connected to the burst pin set the level of load current at which the converter automatically switches between continuous and Burst Mode operation. Peak inductor current is limited two ways. The first method monitors current in switch A and sources a 10ms/DIV Figure 4. A soft clamp peak current control keeps the converter in continuous control mode when peak inductor current is reached. fraction of that current into the FB pin when the peak current exceeds 1.1A. This effectively lowers the VOUT set point providing a closed loop method of controlling the peak current. In higher load and transient situations a comparator opens switches A and B, thus providing a hard peak current limit of 1.3A. Figure 4 shows inductor current and VOUT responding to increasing load. As the inductor current reaches current limit, VOUT drops and the control loop stays in continuous operation. When VOUT is low either at start-up or recovering from a short circuit the current limit clamp level is reduced by half providing a foldback function. Switching frequency may be programmed with an external resistor to a frequency between 300kHz and 2MHz, which allows a trade off between component size and efficiency. Soft start is performed by controlling the slew on the SHDN/SS pin. Once the converter is enabled, when the A miniature, 1” or smaller disk drive in standby may draw 40mA. In idle or track seeking modes the drives current increases to 150mA and when reading or writing data the load might peak at 200mA to 300mA. Even during these transitions, the supply should be well regulated with very low ripple throughout the discharge cycle of the system battery. Compared with a step-down regulator in a 3.3V system a buckboost converter such as the LTC3532, which maintains regulation even as the battery drops below 3.3V, allows all the energy in the battery to be used. The converter keeps accurate regulation during load transitions when battery ESR may cause the input voltage to drop below VOUT. A Li-ion to 3.3V application, which uses a tiny multi-layer chip inductor, is shown in Figure 5. Capable of load steps up to 400mA with battery voltage as low as 3.0V, the converter delivers efficiency greater than 90% in continuous mode with loads between 30mA and 200mA as shown in Figure 6. Lower power burst mode efficiency is greater than 80% at sub 1mA loads. This circuit uses a soft start capacitor connected to SHDN/ SS to limit inrush current. Transient responses to load steps are shown in D2 4.7µH VOUT 3.3V 340k D1 VIN 2.5V TO 4.2V SW1 SW2 VIN VOUT SHDN/SS 412k BURST RT LTC3532 FB VC 12.1k 220pF 220pF GND 22µF × 4 4.7µF 4.7nF SD 0.1µF 249k 86.6k 1k 200k L1 = FDK M1PF2520D4R7 D1, D2 = MBRM110LT Figure 5. A high efficiency converter using a tiny multi-layer inductor ideal for miniature HDD applications Linear Technology Magazine • June 2006 17 L DESIGN FEATURES 200µs/DIV 100 95 EFFICIENCY (%) 90 the value of the resistor on the BURST pin lowers the current at which burst mode is entered. Figure 8 shows the relationship between the burst pin resistor and the output current value at which the transition between continuous and burst operation takes place. If desired, the operating mode may be forced by driving the BURST pin above or below the thresholds. VOUT 100mV/DIV VIN = 4.2V 85 VIN = 3V 80 VOUT 500mV/DIV 75 70 BURST 5V/DIV 65 SHDN/SS 5V/DIV 60 55 VOUT = 3.3V 50 0.1 1 10 100 LOAD CURRENT (mA) ILOAD 100mA/DIV 1000 Figure 6. Efficiency is over 85% for all power saving modes of a typical 1” hard disk drive Figure 7. VOUT response to load transitions is well controlled. 1.12V the converter switches to continuous mode. As the load decreases and the voltage on the burst pin drops below 0.88V, the converter switches back to Burst Mode operation. Raising 70 60 LOAD CURRENT (mA) Figure 7. As the load is increased from 10mA to 50mA the transition from burst mode to continuous operation occurs. Steps from 150mA to 300mA and back show VOUT transients have peak amplitude of only 20mV. An RC network on the BURST pin enables automatic Burst Mode operation to maintain high efficiency at light loads without external control. Burst operation is controlled by the voltage on the burst pin. During operation a small fraction of the output current passing through switch D is mirrored out of the BURST pin. The mirrored current produces a voltage across the burst pin resistor that is proportional to the average load current. Figure 7 shows the burst pin responding with a voltage proportional to the load current. When the load increases and forces the burst pin voltage above R1 499Ω Control Input Current for USB Applications 10ms/DIV 50 40 LEAVE BURST 30 20 VOUT = 3.3V VIN = 3.6V ENTER BURST 10 150 250 350 450 BURST RESISTOR (kΩ) 550 Figure 8. The load at which the converter transitions from Burst Mode to continuous mode is programmable with a single resistor. Many devices now are powered and recharged from USB ports which have the restriction of a maximum current draw of 500mA. A converter that typically supplies 500mA would not nominally exceed the USB current limits. However, tolerances of host regulators, USB bus powered hubs, and cable drops result in a rather poorly regulated USB voltage which may vary from 5.25V down to 4.35V or lower during a transient. For example, if a 5V to 3.6V converter circuit is responding to a peak load of 500mA and the USB voltage drops to 4.35V, at 80% efficiency the input current exceeds 500mA. Figure 9 shows an input current monitor, which controls VOUT to clamp the input current to 500mA. Current injected into the FB node changes the effective set point of the output voltage within the voltage mode control loop. In fact, the output voltage of a continued on page 37 – + Q1 2N3906 LT1677 VOUT 3.6V 3.3µH VIN R2 0.05Ω D2 MBRM110LT SW1 VIN SHDN/SS BURST RT C1 4.7µF 43.2k LTC3532 SW2 169k VOUT FB VC 12.1k C2 220pF 12.1k D1 1N914 + LT1677 – GND 22µF 86.6k R3 24k Figure 9. A few components may be used to limit input current for USB and other applications 18 Linear Technology Magazine • June 2006 DESIGN FEATURES L Dual/Triple Power Supply Monitor for Undervoltage and Overvoltage on Positive and Negative Supplies by Andrew Thomas Introduction An accurate power supply monitor can signal when a supply overvoltage or undervoltage condition threatens to cause system failures, allowing the system to deal with the situation gracefully. The LTC2909 is a highly customizable monitoring solution with adjustable input thresholds, input polarity selection, a multimode reset timer, and an open-drain RST output. Adjustable input thresholds allow the user to set any trip threshold for the comparator, subject only to the accuracy limitations of the part, instead of having to pick from a factory-set limited collection of thresholds. Each adjustable input can be configured in either polarity, allowing it to monitor negative or positive supply voltages for undervoltage or overvoltage. Polarity selection is controlled by simple connection of the SEL pin—no external components required. The multimode timer pin can be configured a number of ways to suit a large variety of applications, allowing full control over the reset timeout, elimination of the external timing capacitor, or removal of the timeout altogether. The RST pin is an opendrain output—it can be pulled up to an appropriate voltage for the device receiving the RST signal, independent of the supply for the LTC2909. The output can be wired-OR connected with other supervisors or other opendrain logic, allowing any of a number of conditions to issue a reset. Minimal Space Required Figure 1 shows how the LTC2909, with just a few components, can monitor a 24V supply for both undervoltage and overvoltage. Almost any two reLinear Technology Magazine • June 2006 24V ±5% 47k 4.12M 11.5k 10k VCC ADJ1 RST REF SEL ADJ2 82.5k Any Polarity, Undervoltage or Overvoltage 3.3V SYSTEM RESET TMR GND Figure 1. A 24V undervoltage and overvoltage monitor set conditions in any system can be monitored by appropriate connection of the LTC2909. The small size of the LTC2909 (available in 8-pin 3mm × 2mm DFN and TSOT-23 packages) keeps the monitoring solution small, and the high accuracy of the part keeps The two adjustable inputs can be configured in either polarity, allowing the LTC2909 to monitor negative or positive supply voltages for undervoltage or overvoltage. Polarity selection is controlled by simple connection of the SEL pin—no external components required. system uptime high without sacrificing reliability. The separate VCC pin of the LTC2909 incorporates a shunt regulator, which allows the part to be powered from any high availability supply, even a high voltage rail. Furthermore, the low quiescent current consumed by the LTC2909 makes it suitable for low power applications like batterypowered handheld devices. The most common application of a supply monitor is determining when a positive supply is below some critical threshold required for the proper operation of powered devices. Less common, but no more difficult for the LTC2909, are scenarios that require negative supply monitoring, or determining when the voltage exceeds some value beyond which functionality might be impaired or powered devices damaged. The connection of the SEL threestate input pin determines whether each of the ADJ input comparators is configured as positive-polarity (input must be above the threshold or RST is asserted low) or as negative-polarity (input must be below the threshold or RST is asserted low). Inputs that are configured as negative-polarity are useful for resetting when the monitored voltage is more positive (or less negative) than it should be. In other words, a negative-polarity input can monitor a positive supply for overvoltage (OV) or a negative supply for undervoltage (UV). Similarly, a positive-polarity input is useful for issuing a reset when the monitored voltage is more negative (or less positive) than it should be, so it may monitor a positive voltage for undervoltage or a negative voltage for overvoltage. Conventionally, the terms overvoltage and undervoltage refer to the absolute value of the monitored voltage, so a –5V supply at –4.3V is undervoltage. Connecting SEL to ground configures both adjustable inputs as negative polarity. In this mode, the part may be used as a dual negative undervoltage monitor, or a dual positive overvoltage monitor. If desired, it also functions as 19 L DESIGN FEATURES a single negative undervoltage monitor with a single positive overvoltage monitor. Connecting SEL to VCC configures both inputs as positive polarity, useful for dual positive undervoltage or dual negative overvoltage monitors, as well as a single positive undervoltage monitor with a single negative overvoltage monitor. Finally, leaving the SEL pin open configures ADJ1 as positive polarity, and ADJ2 as negative polarity. In this configuration, the part can monitor one positive and one negative supply both for undervoltage, or both for overvoltage. It can also function as a window (undervoltage and overvoltage) monitor for one positive or negative supply. These polarity selections and the corresponding applications are summarized in Table 1. Adjustable Inputs The LTC2909 inputs are fully adjustable for ultimate monitoring flexibility. Each ADJ pin connects directly to the high-impedance input of a comparator whose other input is tied to an internal 500mV (nominal) reference. Setting the threshold voltage is as simple as connecting a resistor divider from the supply so that the ADJ input is at 500mV when the monitored supply is at the desired threshold. By choosing the correct external resistors, the nominal trip point can be set to any desired value. The typical configuration of resistors for a positive supply is as shown in Figure 2. For a negative supply, some offset is needed to allow the resistor tap point to lie at 500mV. This offset is provided by the REF pin on the LTC2909, which provides a buffered 1V reference (with 1.5% accuracy over the operating temperature and supply voltage range). Thus, the typical divider connection for a negative supply is as shown in Figure 3. Note that positive supplies with nominal trip points below 500mV should be considered “negative” for monitoring purposes (since they require an upwards shift to reach 0.5V). Monitoring a single supply for UV and OV can be accomplished with three resistors, as shown in Figure 4 for a positive supply and Figure 5 for a negative supply. 20 Selection of resistor values is driven by two factors: nominal trip point and current consumption. In particular, the selection of R1 is driven by current consumption, and the ratio of the other resistors to R1 determines the trip point. If the monitored voltage is typically close to its nominal trip threshold, the voltage across R1 is approximately 0.5V, so the current consumed by the resistor divider is about 0.5V/R1. Supplies that operate substantially away from their threshold cause the current consumption to deviate from the estimate above by about the same percentage by which they deviate from the threshold. In most applications, the current consumption should be minimized. However, as the current is reduced, the impact of leakage at the tap point on the monitoring accuracy becomes more severe. The leakage current is drawn from the driving-point impedance at the ADJ input, so the fractional error is approximately: ILEAK • R1 • R2 (R1+ R2) VMON R2 ADJx + R1 – 0.5V + – Figure 2. Monitoring a positive supply VMON ADJ1 + R3 R2 – ADJ2 + R1 0.5V + – – Figure 4. Monitoring a positive supply for UV and OV 500mV or for UV/OV circuits: ILEAK • (R1+ R2) • R3 / (R1+ R2 + R3) 500mV and ILEAK • R1 • (R2 + R3) / (R1+ R2 + R3) 500mV As a rule of thumb, the current in the divider should be at least 100 times the expected leakage, including the 15nA maximum internal to the part and any external leakage sources. The rest of the resistor values are determined by the choice of trip point. Since the accuracy of the LTC2909 thresholds is guaranteed to 1.5% over the operating temperature and supply range, the trip points should usually be set 1.5% beyond the specified operating range of the monitored supply. For example, a 5V ±10% supply should have a 4.425V undervoltage trip point, not 4.5V. See the sidebar on threshold accuracy for an explanation. Given a desired trip point, and the value of R1 chosen as above, it is then possible to calculate the appropriate REF R1 ADJx + R2 – VMON 0.5V + – Figure 3. Monitoring a negative supply REF ADJ1 + R1 R2 – ADJ2 + R3 VMON 0.5V + – – Figure 5. Monitoring a negative supply for UV and OV Linear Technology Magazine • June 2006 DESIGN FEATURES L values of the rest of the resistors. When monitoring a positive supply for a single fault condition, the user should choose V − 500mV R2 = R1• TRIP 500mV Similarly, for a negative supply (or positive supply with trip voltage below 0.5 V), R2 = R1 • 500mV − VTRIP 500mV Note that if the desired trip voltage is below ground, the value VTRIP should be negative. The situation is slightly more complicated when only three resistors are used to monitor a single supply for UV and OV. For a positive supply with desired trip thresholds VTRIP(UV) and VTRIP(OV), the appropriate values are R2 = R1 • VTRIP(OV ) − VTRIP(UV ) VTRIP(UV ) and R3 = R1 • VTRIP(UV ) − 500mV VTRIP(OV ) • 500mV VTRIP(UV ) Table 1. SEL connection for various input polarities ADJ1 ADJ2 SEL Pin Positive polarity: Positive UV or Negative OV Positive polarity: Positive UV or Negative OV VCC Positive polarity: Positive UV or Negative OV Negative polarity: Negative UV or Positive OV Open Negative polarity: Negative UV or Positive OV Negative polarity: Negative UV or Positive OV GND Finally, for a negative supply with desired trip thresholds VTRIP(UV) and VTRIP(OV), the appropriate values are: R2 = R1 • VTRIP(UV ) − VTRIP(OV ) 1V − VTRIP(UV ) and R3 = R1 • 500mV − VTRIP(UV ) 1V − VTRIP(OV ) • 500mV 1V − VTRIP(UV ) Tables 2 and 3 show suggested values of resistors for monitoring a number of standard supply voltages for UV, OV or UV and OV. Table 2 gives values for nominal supply accuracy of 5% (6.5% trip points), and Table 3 gives values for 10% supplies (11.5% Table 2. Suggested resistor values for 5% monitoring Nominal Voltage R1 24 5% UV R2 R1 232k 10.2M 15 115k 12 5% OV 5% UV and OV R2 R3 R2 R1 102k 5.11M 82.5k 11.5k 4.12M 3.09M 200k 6.19M 76.8k 10.7k 2.37M 49.9k 1.07M 102k 2.49M 76.8k 10.7k 1.87M 9 115k 1.82M 78.7k 1.43M 162k 22.6k 2.94M 5 137k 1.15M 137k 1.33M 76.8k 10.7k 732k 3.3 221k 1.15M 340k 2.05M 76.8k 10.7k 453k 2.5 115k 422k 51.1k 221k 137k 19.1k 576k 1.8 63.4k 150k 115k 324k 82.5k 11.5k 221k 1.5 59.0k 107k 137k 301k 76.8k 10.7k 158k 1.2 127k 158k 102k 158k 187k 26.1k 267k 1.0 200k 174k 100k 113k 107k 15.0k 105k –5 133k 1.37M 118k 1.37M 174k 20.0K 2.00M –9 97.6k 1.74M 115k 2.32M 182k 22.6k 3.65M –12 107k 2.49M 40.2k 1.07M 40.2k 5.11k 1.07M –15 107k 3.09M 309k 10.2M 309k 40.2k 10.2M Linear Technology Magazine • June 2006 trip points). In the tables, the values of R1 have been chosen to minimize the threshold error using standard 1% resistor values, while maintaining the divider current consumption near 5µA. UVLO The LTC2909 features a third highaccuracy comparator on the VCC pin, which allows the part to function in some applications as a triple supply monitor. The polarity of the VCC comparator is fixed to be positive, so the comparator creates an accurate UVLO. The threshold of the UVLO is also fixed, and is set at 11.5% below the nominal threshold voltage specified in the part number. Versions are available for standard logic supplies: LTC2909-2.5 for 2.5V supplies (2.175V nominal threshold), LTC2909-3.3 for 3.3V supplies (2.921V nominal threshold), and LTC2909-5 for 5.0V supplies (4.425V nominal threshold). The LTC2909-2.5 is recommended for designs that do not want monitoring of the VCC pin. The UVLO then functions merely to ensure that RST is not allowed to go high while the VCC voltage is too low to guarantee proper accuracy of the ADJ input thresholds. The accuracy of the UVLO threshold is the same as the ADJ thresholds: ±1.5% guaranteed over the operating temperature range. Glitch Immunity A monitored supply generally has highfrequency components riding on its DC value. These may be caused by load transients acting on non-zero output impedance (whether due to supply line impedance or regulation bandwidth), output ripple of the supply, coupling 21 L DESIGN FEATURES 22 Table 3. Suggested resistor values for 10% monitoring Nominal Voltage R1 24 102k 4.22M 115k 6.04M 39.2k 10.2k 2.05M 15 200k 5.11M 200k 6.49M 41.2k 10.7k 1.33M 12 115k 2.32M 107k 2.74M 41.2k 10.7k 1.05M 9 113k 1.69M 140k 2.67M 73.2k 19.1k 1.37M 5 113k 887k 113k 1.15M 115k 30.1k 1.13M 3.3 221k 1.07M 294k 1.87M 226k 59.0k 1.37M 2.5 102k 348k 301k 1.37M 41.2k 10.7k 178k 1.8 137k 301k 86.6k 261k 63.4k 16.5k 174k 1.5 48.7k 80.6k 43.2k 102k 51.1k 13.3k 107k 1.2 137k 154k 63.4k 107k 80.6k 21.0k 115k 1.0 200k 154k 137k 169k 174k 45.3k 169k –5 115k 1.13M 200k 2.43M 115k 24.3k 1.37M –9 127k 2.15M 215k 4.53M 51.1k 11.8k 1.07M –12 115k 2.55M 41.2k 1.15M 130k 30.9k 3.57M –15 115k 3.16M 309k 10.7M 47.5k 11.5k 1.62M 10% UV R2 R1 10% OV R2 toggling at the reset output. Because the timeout is defeated in comparator mode, the LTC2909 is free to chatter in that mode, so a small amount of one-sided hysteresis is added to the comparator thresholds. See “Timeout Control” below for a description of the hysteresis behavior. The other concern that must be addressed is identifying which transients cause a problem for the devices on the supply bus. It can generally be assumed that those devices can continue to operate through short 700 600 MAXIMUM ALLOWABLE GLITCH DURATION (µs) from nearby high-frequency signals, or noise. Ideally, the supply monitor should decide whether the supply voltage transient threatens the functionality of any of the devices which are powered by that voltage rail, and issue a reset if (and only if) it does. Unfortunately, a real supervisor cannot use an omniscient algorithm to know what exactly is connected to the bus or how those devices respond to supply transients. Given this, a number of possible approaches exist, addressing some of the concerns related to supply transients. These techniques focus on eliminating two undesirable situations that result from using a simple comparator. One undesirable effect that must be prevented is rapid toggling of the reset output (“chattering”), caused by ripple, coupling, or noise on a supply voltage that is near the threshold. A common solution is to add hysteresis to the monitor threshold, which prevents chattering as long as the transient amplitude is less than the amount of hysteresis. Adding hysteresis effectively worsens the threshold accuracy, thereby unnecessarily reducing system uptime, or tightening the system requirements on supply voltage. For this reason, the LTC2909 uses other methods to prevent chattering, and does not have threshold hysteresis, unless the part is configured in comparator mode, where it would otherwise be more susceptible to chattering than usual (as explained below). The primary defense against chattering is the programmed reset timeout period. If at any time during the reset timeout the supplies become invalid, the timer is immediately zeroed, and starts timing again from the beginning of the period when the supplies become valid again. Thus, any time the supply voltage is close enough to the threshold that the amplitude of the supply transients take the supply into the invalid region, RST remains low as long as the time between transients is less than the reset timeout. That is to say, the reset timeout prevents transients with frequency greater than 1/tRST from causing undesired 500 400 300 RESET OCCURS ABOVE CURVE 200 100 0 1 10 0.1 100 GLITCH PERCENTAGE PAST THRESHOLD (%) Figure 6. Allowable glitch duration as a function of magnitude R1 10% UV and OV R2 R3 duration excursions outside the valid supply region, particularly because local decoupling capacitors help prevent such transients from appearing at the devices. If possible, the supervisor should not issue a reset during these conditions. Consider, for example, what happens when a system spins up a hard drive connected to a monitored supply bus. The bus voltage briefly dips, possibly falling outside the valid region, and then returns, approximately, to its previous value. This is normal, expected behavior, and a microprocessor that is also connected to that bus should function normally through the transient (otherwise there is no way the system can ever safely use the hard drive). The supply monitor should not issue a reset to the microcontroller during such a transient. To solve this problem, the LTC2909 has low-pass filtering on the comparator outputs, so that short duration glitches on the monitored supply are not passed through to the control logic. For most systems, the response of the system to a glitch depends on the Linear Technology Magazine • June 2006 DESIGN FEATURES L possible, from the corresponding supply. Negative-polarity applications may also oscillate when the RST is driving a large load, which causes a voltage difference between the ground of the 0.5V internal reference, and the ground of the monitored voltage. Several factors can help eliminate this source of oscillation. First and foremost, the current sunk by RST should be kept below 1mA if possible. Good grounding practice is also important. Input resistor dividers which connect to ground should have a Kelvin-sense trace directly to the GND pin, and the path from the monitored supply ground to the GND pin should be low impedance (preferably through a good ground plane). Timeout Control As described above, the LTC2909 has a reset timeout delay which helps reduce the sensitivity of the monitor to supply glitches. For convenience, this reset timeout can be controlled in three different ways. If a 200ms timeout is appropriate for the application (based on expected noise distributions and system timing specifications), no external components are needed to set the timeout—simply tie the TMR pin Why Is Threshold Accuracy Important? In monitored systems, there is some voltage level beyond which the proper function of the devices connected to a supply bus cannot be guaranteed. Ideally, that is the voltage at which the supervisor should issue a reset, since this guarantees the proper function of the system while permitting the maximum allowable variation in supply voltage. Thus, in the ideal case, the power supply tolerance is as loose as the devices on the bus will tolerate. Of course, any real supervisor has limited accuracy, which tightens the system constraints. Typically, monitor accuracy is specified as a percentage band around the nominal trip point in which the threshold is guaranteed to lie, such as ±1.5%. To prevent nuisance resets when the supply is operating normally, the supply tolerance and monitor accuracy bands should not overlap. As an example, a supply with a specified tolerance of ±5%, monitored by a 1.5% accurate monitor must have its nominal threshold set at 6.5% to prevent nuisance resets. With that accuracy band, the supervisor is not guaranteed to issues a reset until the supply has reached the other end of the monitor accuracy band, at 8%. Therefore, the devices attached to the supply must function properly to at least an 8% deviation in supply voltage. If this is not possible, a supply with tighter tolerance must be provided. For comparison, if the 1.5% accurate supply monitor is replaced by a less accurate 2.5% device, the power supply tolerance must be tightened to ±3% to ensure the same 8% operation band, thus complicating the power supply design. L Linear Technology Magazine • June 2006 10000 RESET TIMEOUT PERIOD, tRST (ms) energy contained in the glitch, rather than just the voltage amplitude of the glitch. The duration of the glitch also factors into that energy, so the probability of a failure increases as the duration of the glitch increases (e.g. a 20% glitch on the supply may only be tolerable for 100µs, whereas a 5% glitch is tolerable for 1ms). The filtering on the LTC2909 comparators reflects this tendency. Figure 6 shows a typical curve of the maximum glitch duration that does not result in the LTC2909 issuing a reset, versus the percentage amount the glitch goes into the invalid region. Some of these concerns can be exacerbated by circuit board layout, so it is also important that some care be taken in the layout near the LTC2909. In applications which use negative polarity comparators, capacitive coupling from the RST output to the negative-polarity input can cause the part to oscillate at approximately 1/tRST if the negative-polarity input is sufficiently close to threshold: the capacitive coupling creates AC negative feedback around the part. To prevent this oscillation, the RST line should be kept away from the relevant ADJ inputs, and, where 1000 100 10 1 0.1 1 10 100 TMR PIN CAPACITANCE, CTMR (nF) 1000 Figure 7. Reset timeout period as a function of capacitance to ground, and the LTC2909 uses an internal 200ms delay generator. For applications that require timeout periods other than 200ms, the delay can be set by connecting the TMR pin to a grounded capacitor, where the delay is set at approximately 9ms per nF of capacitance. To ensure timer accuracy, the timing capacitor should be a low leakage ceramic type. Leakage currents over 500nA substantially impair timer function. As an example, for a 50ms delay, the timer capacitor should be 50/9 = 5.6nF. Figure 7 shows the typical timeout period as a function of the capacitor on the TMR pin. Due to inherent capacitance on the TMR pin, the minimum attainable timeout period in external mode is about 400µs, with no external capacitor connected to the pin. The maximum timeout is limited to nine seconds (1µF capacitor) by startup concerns. Assuming that the timer capacitor is initially discharged during the power-up sequence, the LTC2909 initially sees that the TMR voltage is near ground, and thus operates in internal timeout mode. As soon as the part is powered, a 2µA current source begins pulling up on the TMR pin, charging the timer capacitor towards the ground sense threshold (approximately 250mV). If all three supply inputs (VCC and both ADJ inputs) become valid, and the 200ms internal timeout period completes before the TMR voltage reaches the ground sense threshold, RST goes high after a much shorter delay than was intended. If this startup behavior is not a problem in a given system, the 23 L DESIGN FEATURES maximum timeout is limited only by the availability of large capacitors with leakage currents below 500nA. Finally, there are some systems where the reset timeout delay is undesirable. For example, this may be the case in applications where the user is not using the LTC2909 RST pin as a system reset line. If the user ties the TMR pin to VCC, the LTC2909 is put into comparator mode. In comparator mode, the timeout delay is bypassed, and the comparator outputs are connected directly to the RST drive circuitry. Due to the glitch-rejecting low-pass filter in the comparators, there will still be some delay from the inputs to the RST output, based on the amount of overdrive on the input. As shown by Figure 6, the propagation delay for large overdrives is about 25µs. In comparator mode, because the reset timeout has been removed, the glitch and oscillation immunity of the part have been decreased. To prevent undesired “chattering” of the RST output when the input voltages are very close to threshold, a small amount of one-sided hysteresis is added to all three comparators. The hysteresis is “one-sided” in the sense that the validto-invalid transition is unaffected, but the invalid-to-valid threshold is moved about 0.7% into the valid region. Thus, for the ADJ inputs, the threshold voltages in comparator mode are a function of the SEL pin state. Nominal values are shown in Table 4. Shunt Regulator In most systems, it is possible to identify one supply as the one with highest availability—that is to say the supply which is most likely to be on, first to power up, last to shut down, and so on. There are a number of advantages to powering a supply supervisor from this highest-availability supply. First, the RST pull-down circuits are powered by the part supply. Thus, having the part supply come up first helps guarantee that RST never floats high due to insufficient pull-down strength. Conversely, powering the part from a high-availability supply helps maximize the uptime of the system because the LTC2909 will not release the RST output unless the part is properly powered. The problem in many systems is that the high-availability supply is also a relatively high-voltage supply. For example, the highest availability supply in an automotive system is the 12V (nominal) battery voltage, and in a telecom system it is likely to be a 48V supply. Most supply supervisors require an external voltage regulator to operate from these supplies, but the LTC2909 saves components by integrating a 6.5V shunt regulator into the VCC pin. All that is required is a series-dropping resistor between the high-voltage supply and the VCC pin. This scheme allows the LTC2909 to be powered from an arbitrarily high voltage, subject only to constraint by the power dissipation in the shunt resistor. Furthermore, the VCC pin can be used to power other low voltage parts, as long as their supply current (which should be less than 5mA) is factored into the selection of the resistor. The shunt regulation voltage is nominally 6.5V, and is guaranteed to lie between 6.0V and 6.9V across the entire operating temperature range and across a wide range of shunt current. Selection of the series resistor is driven by the shunt regulator bias current. The shunt regulator bias will be set by the amount of current flowing through the resistor (based on its value and the voltage drop across it), minus the supply current of the part, including any load drawn from the REF pin, and the load currents of any other devices that take advantage of the 6.5V supply at the VCC pin. The series resistor should be chosen to bias the shunt regulator somewhere between 50µA and 10mA, ideally around 1mA. These design constraints impose the following limits on the series resistor. The maximum load drawn from the reference, plus the maximum load drawn by other devices connected to the VCC pin, plus 150µA for the LTC2909 must be less than the minimum current through the resistor by at least 50µA: (IREF + IDEVICES )MAX + 150µA + 50µA ≤ Input ADJ1 ADJ2 24 SEL = GND SEL Open SEL = VCC Rising 500.0mV 503.5mV 503.5mV Falling 496.5mV 500.0mV 500.0mV Rising 500.0mV 500.0mV 503.5mV Falling 496.5mV 496.5mV 500.0mV R SERIES This ensures that the shunt regulator is biased with at least 50µA of current. On the other side, the minimum load on the reference, plus the minimum load drawn by other devices on VCC must be less than the maximum current through the resistor by at most 10mA: (IREF + IDEVICES )MIN + 10mA ≥ VSUPPLY(MAX ) − 6 V R SERIES This ensures that the regulator is never shunting more than 10mA of current. In summary, the series resistor is required to satisfy: VSUPPLY(MAX ) − 6 V (IREF + IDEVICES )MIN + 10mA ≤ Table 4. Nominal ADJ thresholds in comparator mode VSUPPLY(MIN) − 6.9 V ≤R VSUPPLY(MIN) − 6.9 V (IREF + IDEVICES )MAX + 200µA As an example, consider operation from an automobile battery. For purposes of this example, the operating range of the battery supply is approximately 10V to 60V, and we can suppose that the user’s loading of REF and external current use can each range Linear Technology Magazine • June 2006 DESIGN FEATURES L from 0µA to 100µA. The minimum value of R is then 54V/10mA = 5.4k, and the maximum is 3.1V/400µA = 7.75k. Given these constraints, a value of 6.8k is probably optimal. The above equation is actually overly restrictive. In cases where the supply voltage is very close to the shunt regulation voltage, it may be impossible to satisfy the above equation because the maximum allowable value is less than the minimum. In these cases, it may be assumed that the maximum allowable value is 1k instead of the value predicted by the formula above, as long as the VCC pin is not used to power other devices. There are scenarios where the shunt regulator cannot satisfy the needs for VCC (e.g. those with a very large possible supply range). These applications must use an external voltage regulator of some sort, which, of course, should have a regulation voltage below 6V. A final consideration is the power dissipation in the series resistor, which may be quite high for high voltage supplies. The series resistor must be rated to handle a power of at least ( VSUPPLY(MAX) − 6V) 2 A rough rule of thumb suitable for many applications (those that have fairly constant REF current draw, and have minimum supply voltages well above 6V) is that the resistor rating should be at least 0.1 Watt per 100 RP2A 1.43M RP2A2 169k 10k* MANUAL RESET PUSHBUTTON RP1 49.9k RN2 2.49M RN1 107k *OPTIONAL FOR ESD volts of maximum supply, multiplied by the ratio of maximum to minimum supply voltage. Returning to the automobile battery example from above, the power dissipated in the 6.8k resistor could be as large as 542/6800 = 0.43W (the rule of thumb would give 0.36W), so a 0.5W resistor is best. In reality, of course, the battery is unlikely to stay at 60V for long enough to heat up the resistor substantially. If we were to take a more reasonable DC maximum of 16V, the resistor only needs to handle about 15mW. ±12V UV Monitor with Manual Reset Figure 8 shows a LTC2909 configured as an undervoltage monitor for a system with ±12V supplies, and a 1.8V logic bus. The part is powered from the high-availability 12V supply RCC 27k 0.25W RP2B 1.91M SYSTEM CBYP 100nF 5V M2 RPU 10k VCC ADJ1 RST LTC2909-25 SEL ADJ2 RP1A 18.7k RP1B 13.7k RP1B2 681k M1 REF TMR GND M1, M2: FDG6301N OR SIMILAR IF LOADING OF RST WILL EXCEED 1nF, A 1nF BYPASS CAPACITOR ON M1’s DRAIN IS RECOMMENDED Figure 9. A 48V telecom UV/OV monitor with hysteresis Linear Technology Magazine • June 2006 1.8V RPU 10k VCC ADJ1 RST FAULT OUTPUT LTC2909-25 REF SEL ADJ2 TMR GND CTMR 2.2nF Figure 8. ±12V undervoltage monitor with pushbutton reset VUV(RISING): 43.3V VUV(FALLING): 38.7V VOV(RISING): 71.6V VOV(FALLING): 70.2V RP2 1.07M Applications R SERIES VIN 36V TO 72V 12V –12V CBYP 100nF RCC 10k through the series dropping resistor RCC. The floating condition of SEL sets the polarity for one positive and one negative UV. The reset timeout is set to 20ms nominal by CTMR, which allows faster recovery from faults. Finally, the pushbutton allows the user to drive ADJ1 to ground, manually forcing a reset condition. The release of the pushbutton is debounced by the LTC2909’s reset timeout. If ESD from people touching the pushbutton is a concern, a 10k resistor in series with the pushbutton limits the current flow into the LTC2909 to prevent damage. 48V Telecom UV/OV Monitor with Hysteresis Telecom supply specifications usually require some amount of hysteresis in the acceptable voltage range. Since the LTC2909 does not generally have hysteresis in its thresholds, the hysteresis must be externally added. Figure 9 shows the LTC2909 configured to monitor a 48V nominal supply bus for UV and OV. The NMOS devices lower the UV threshold (by reducing R2 for ADJ1) and raise the OV threshold (by reducing R1 for ADJ2) while the RST is high. This has the effect of widening the acceptable supply window once the supply becomes good. The resistors are chosen so that the window is 43.3V–70.2V when the supply is outside the window, and 38.7V–71.6V once the supply is good. Since the part is powered from the 48V bus, the series-dropping resistor is required to be a 0.25W device to handle the power dissipated when the bus is overvoltage. 25 L DESIGN FEATURES DC/DC D1: 1N5238B OR SIMILAR Q1, Q2: FFB2227 OR SIMILAR RS 0.01Ω VIN 12V DC/DC M1 IRLZ34 RG2 10Ω 2N6507 CG 10nF CBYP1 100nF PWRGD LT1641-2 GND TIMER SYSTEM CBYP3 100nF RCC 4.7k RP2A 2.49M RP2B 2.05M Q2 RL2 100k RG1 1k SENSE GATE 3.3V RL1 4.7k Q1 D1 VCC 2.5V RPU1 4.7k CIRCUIT BREAKER AND CROWBAR ADJ1 TMR REF SEL TMR ADJ2 GND RFB1 10k RP1A 102k 12V OV AND 3.3V OV DETECT RP1B 340k RP2E 1.15M VCC ADJ1 LTC2909-25 REF SEL RP2D 1.07M VCC RST LTC2909-25 RFB2 100k ON FB CT 680nF CBYP2 100nF VCC RST RP2C 221k ADJ2 GND ADJ1 RREF 10.7k RP1C 51.1k RP1D 49.9k NTC THERMISTOR NTHS-1206N01 R25 = 100k R = 10.7k AT 85°C 2.5V OV AND T > 85°C DETECT SEL RPU2 10k LTC2909-25 RST REF TMR ADJ2 RP1E 221k GND 12V, 3.3V and 2.5V UV DETECT Figure 10. Automotive supply system with overvoltage, overcurrent and overtemperature protection The recommended NMOS device is FDG6301N, which combines both NMOS devices in one SC70-6 package. Other devices may be used as long as the threshold voltage is guaranteed to be much less than 5V, and the drainsource breakdown is greater than 10V. Note that if the RST output is loaded with a large capacitance, the feedback through the gate-drain capacitance of M1 can cause the circuit to oscillate unless a bypass capacitor is placed on M1’s drain. Automotive Supply System Figure 10 shows three LTC2909s in a full-featured automotive supply system, providing overvoltage, overcurrent, and over -temperature protection in addition to an undervoltage system reset. The system uses an LT1641-2 Hot Swap controller as a controlled electronic circuit breaker. The IRLZ34 logic-NFET serves as the disconnect switch, and the 10mΩ sense resistor sets a current limit of 4.7A. After an over-current fault, the LT1641-2 reconnects after a delay of 160ms (set by CT). The two LTC2909s on the left are responsible for detecting overvoltage and over-temperature conditions. To guarantee that they function properly, they must be powered from the 12V input. The VCC pins are tied together, and the supply current flows through just one dropping resistor, so the voltage tends to regulate at whichever of the shunt regulation voltages is the lower of the two. When any of the supply voltages goes overvoltage, or the temperature sensor is heated above 85°C, the shared RST line is pulled low by one of the two LTC2909s. This takes the LT1641-2 ON input low, disconnecting the power switch. At the same time, current is pulled through Q2, turning on Q1, which triggers the 2N6507 SCR and thereby crowbars the 12V supply to the system, removing the overvoltage condition. After the fault condition disappears, the LTC2909s apply a 200ms timeout before reconnecting to the 12V input. The third LTC2909 serves to provide a master reset to the system when any of the three supplies are undervoltage, whether because insufficient input voltage is present, or because one of the protection faults has tripped. The third monitor function is provided by the UVLO. A Dale NTHS-1206N01 NTC thermistor with room temperature resistance of 100k is used to detect the temperature, and may be physically located wherever temperature monitoring is needed. The thermistor forms part of a resistor divider from the buffered reference output to ground. As long as the temperature is below 85°C, the thermistor resistance is greater than RREF, so ADJ1 is above its threshold, and RST is allowed to go high. If the temperature rises, the thermistor resistance decreases, pulling down on ADJ1, and causing a reset when its resistance is equal to or lower than RREF. Conclusion The LTC2909 is a true one-size-fits-all power supply monitor—a way to simplify design and parts stock. It provides a compact solution to monitoring any two supplies for almost any fault condition, where input polarity selection and a buffered reference output allow monitoring of OV conditions and negative supplies. Precision comparators, including a third input on the part’s VCC, increase system reliability. To simplify design further, no regulated voltage is required—a built-in shunt regulator on VCC allows operation from a high-voltage high-availability supply. An accurate model of the LTC2909 is included with SwitcherCAD (available at www.linear.com), as an aid to rapid development. L For more information on parts featured in this issue, go to http://www.linear.com 26 Linear Technology Magazine • June 2006 DESIGN FEATURES L High Speed Low Power RS485 Transceivers with Integrated Switchable Termination by Ray Schuler and Steven Tanghe Introduction The LTC2859 and LTC2861 combine a logic-selectable integrated termination resistor with a rugged 20Mbps RS485/RS422 transceiver, providing a single die impedance-matched network solution in a tiny package. The low power driver features logicselectable reduced-slew rate mode for operation below 250kbps with low EMI emissions. The 1/8-unit load receiver provides a failsafe output over the full RS485 common mode range for up to 256 nodes. Both receiver inputs and driver outputs feature robust ESD protection exceeding ±15kV. The half duplex LTC2859 is available in a 3mm × 3mm DFN, while the full duplex LTC2861 is available in both 4mm × 3mm DFN and 16-pin SSOP packages (see photo in Figure 1). Block diagrams for the LTC2859 and LTC2861 are shown in Figure 2. Figure 1. Photograph of the LTC2861 SSOP, LTC2861 4mm × 3mm DFN, and LTC2959 3mm × 3mm DFN packages the data. The termination usually consists of discrete resistors that have the RE DE A 15kV SLEEP/SHUTDOWN LOGIC AND DELAY LTC2859 RO 120Ω DE RO DI B 15kV Z 15kV DI DI DRIVER DRIVER Y 15kV Figure 2. Block diagrams of the LTC2859 and LTC2861. LTC2859 R SLO RECEIVER SLO LTC2859 120Ω 120Ω NODE 1 120Ω B 15kV LTC2859 D A 15kV SLEEP/SHUTDOWN LOGIC AND DELAY TE RECEIVER 120Ω DE RE TE RS485 transceivers typically communicate over twisted-pair cables with characteristic impedance ranging from 100Ω to 120Ω. Proper termination of the cable is important to minimize reflections that can otherwise corrupt RO RE TE LTC2861 LTC2859 SLO Switchable Termination R same resistance as the characteristic impedance of the cable, connected differentially across the cable at both ends. When using LTC2859/LTC2861 transceivers, however, no external resistors are necessary. These devices have an integrated 120Ω resistor on the receiver inputs that can be enabled logically to terminate the cable where needed. Figure 3 shows an example of a properly connected network using LTC2859 transceivers with integrated termination resistors enabled on the two end devices. Short connection to R D RO RE TE DE NODE 2 DI SLO 120Ω R D RO RE TE DE NODE 3 DI SLO D RO RE TE DE DI SLO NODE 4 Figure 3. A properly connected RS485 network using LTC2859 transceivers with selectable terminators. Linear Technology Magazine • June 2006 27 L DESIGN FEATURES 1V/DIV 1V/DIV TE1 = 5V TE2 = 0V TE3 = 0V TE4 = 5V 400ns/DIV 1V/DIV TE1 = 5V TE2 = 5V TE3 = 0V TE4 = 0V (a) 400ns/DIV TE1 = 5V TE2 = 0V TE3 = 5V TE4 = 0V (b) 400ns/DIV (c) Figure 4. Differential received signals at node 4 from network in Figure 3. properly terminated with termination at nodes 1 and 4 (a). Improperly terminated with termination at nodes 1 and 2 (b), and nodes 1 and 3 (c) 150 135 140 130 120 140 115 110 MAGNITUDE (Ω) RESISTANCE (Ω) 120 130 PHASE 80 60 –15 –20 20 100 0 20 40 60 80 TEMPERATURE (°C) 100 120 (a) 110 –10 –10 40 120 105 –5 100 –5 5 10 0 COMMON MODE VOLTAGE (V) 15 (b) PHASE (DEGREES) RESISTANCE (Ω) 125 95 –40 –20 0 MAGNITUDE 0 0.1 1 10 FREQUENCY (MHz) –25 100 (c) Figure 5. LTC2859 termination resistance vs temperature (a), common mode voltage (b), and frequency (c) the intermediate stages along the cable should be maintained as these stubs will produce unwanted reflections. To illustrate the importance of proper termination placement in an RS485 system, consider the network of Figure 3 where four LTC2859 transceivers are spaced equidistantly along three hundred feet of cat5e cable. Signals are driven from node 1 and received at node 4. The integrated termination resistors in the LTC2859 devices are switched in or out at various locations along the cable to illustrate the effect of termination placement on the received waveforms. No external resistors are used. With proper resistive termination applied at the ends of the line (nodes 1 and 4) the received waveform has clean transitions, as shown in Figure 4a. If the end resistive termination is moved from node 4 to nodes 2 or 3, the waveforms of figures 4b and 4c result, respectively. It is clear that placement of termination resistors 28 can have a large impact on the signal integrity. The termination resistor in the LTC2859 and LTC2861 is enabled by pulling the termination enable (TE) pin high. The resistor is disconnected when the termination enable is pulled to a logic low or the device is unpowered. Figures 5 (a), (b), and (c) show the resistance is maintained well over temperature, common mode voltage, and frequency. The inclusion of a selectable 120Ω resistive termination on the LTC2859 and LTC2861 is a significant advantage over other RS485 transceivers. When modifications or additions are made to an RS485 network, the required termination changes can be made by logical control of the termination enable pin on the desired transceiver. This can be done through the use of a simple jumper or through higher level system control where manual intervention is prohibitive. A valuable benefit is that every node in the network is capable of providing termination without the use of external resistors, making network re-configuration more manageable. Controller based configuration of a network with the aid of switchable termination can be extended beyond simple network additions and reductions to include fault protection as well. RS485 networks supporting bus lengths up to 4000 feet are at risk of breaks or disconnects in the cabling that can interrupt service. Figure 6 shows a dual-master controlled ring network making use of the LTC2859 logic-controlled termination to protect against such open cable faults. When a break in a network is detected, through loss of response from one or more nodes, the master controller enters a low data rate mode. Even though the break imposes a severe impedance mismatch, low data rate communication is still possible up to the point of the break (as a rule of thumb, communication without termination is possible if the two-way Linear Technology Magazine • June 2006 DESIGN FEATURES L RO RE TE DE DI RO RE TE SLO D R DE RO RE TE SLO DI D R D 120Ω LTC2859 LTC2859 LTC2859 SLO DI R 120Ω 120Ω DE LTC2859 RO R RE TE DE DI D 120Ω SLO LTC2859 LTC2859 R RO RE TE R D DE LTC2859 120Ω 120Ω DI RO RE TE SLO RO 120Ω R D DE DI SLO RO RE TE D DE DI SLO XOR DI Figure 6. Line break-tolerant RS485 topology cable propagation delay is less than 10% of the bit time). Each sequentially addressed transceiver on the network is polled by the masters to determine where the break has occurred. If the network is constructed of LTC2859 and LTC2861 transceivers with selectable termination, the masters can instruct the two nodes on either side of the line break to enable termination resistance. The two master controllers may now access each node of the bus at high data rates until physical repairs have been made. An XOR of the RO pin from each transceiver on the master controller can optionally Linear Technology Magazine • June 2006 reduce the I/O pin count to the micro controller. Driver The LTC2859/LTC2861 drivers can deliver RS485/RS422 signals up to 20Mbps. Figure 7 shows waveforms of the part operating at the maximum data rate. The LTC2859/LTC2861 also feature a reduced-slew rate mode (SLO mode), which is entered by setting the SLO pin to a logic low level. SLO mode increases the driver transition time to reduce high frequency EMI emissions from equipment and cables. In this mode the driver data rate is limited to about 250kbps. Slew limiting also mitigates the adverse affects of improper line termination and long stubs. continued on page 32 DI Z Y 2V/DIV Y–Z 20ns/DIV Figure 7. Driver outputs toggling at the maximum data rate of 20Mbps 29 L DESIGN FEATURES 1.5A VLDO Operates Down to 0.4V Output and Maintains 100mV Dropout by Bill Walter Introduction L1 10µH SW 5V BOOST BST CONVERTER 4.7µF IN VIN = 1.5V 0.4V 4.7µF OFF ON + – VOUT = 1.2V, 1.5A OUT 8.06k ADJ SHDN LTC3026 100k GND COUT 10µF 4.02k PG L1: MURATA LQH2MCN100K02 Figure 1. Block diagram of the LTC3026 and typical application circuit with a low VIN to VOUT differential. The low VIN capability is important as many emerging handheld applications are using a 1.5V main rail, and require output voltages from 1.25V to 0.5V to drive low voltage microprocessor and microcontroller cores. Additionally, the LTC3026 offers low input quiescent current (<1mA) at 1.5V input and less than 1µA in shutdown, maximizing run time in battery-powered applications. Internal protection circuitry includes current limiting, thermal limiting, and reverse-current 70 60 RIPPLE REJECTION (dB) As the power supply voltage levels for digital ICs continue to fall, supply noise and efficiency become increasing concerns. Although many efficient switching DC/DC converters are available, switching noise at the output can be unacceptable at low supply levels, or the size can be prohibitive where multiple supply voltages are required. One commonly used solution is to follow a switching DC/DC converter with a low dropout (LDO) regulator to provide an efficient low noise supply. Nevertheless, LDOs can introduce other problems when used as a low voltage post regulator, such as requiring large and expensive tantalum capacitors, dropping out at a voltage of 300mV or more, requiring supply voltages greater than 1.8V, no ability to produce output voltages below 1.2V, poor supply rejection, poor load regulation, etc. The LTC3026 is a 1.5A VLDO with input voltage capability down to 1.14V and a low adjustable output voltage from 0.4V to 2.6V. The part also has a very low dropout voltage of only 100mV while delivering up to 1.5A of output current, enabling it to optimize battery run time from single cell applications 50 40 30 VBST = 5V VIN = 1.5V VOUT =1.2V IOUT = 800mA COUT = 10µF 20 10 0 100 1k 10k 100k FREQUENCY (Hz) 1M 10M Figure 2. Ripple rejection of the LTC3026 VIN = 2.5V TO ADDITIONAL REGULATORS 10µH BST SW IN BST SW 4.7µF LTC3026 VOUT1 1.8V, 1.5A OUT 1µF LTC3026 IN VOUT2 1.5V, 1.5A OUT 14k SHDN ADJ 100k 4.7µF GND 11k COUT1 10µF 4.02k PG LTC3026 WITH BOOST ENABLED FANOUT: 3-LTC3026 FOR VIN <1.4V 5-LTC3025 FOR VIN >1.4V SHDN 100k 1µF PG1 COUT2 10µF ADJ GND 4.02k PG2 PG BOOTSTRAPPED LTC3026 (BOOST DISABLED) Figure 3. Multiple LTC3026s operating from single boost converter. 30 Linear Technology Magazine • June 2006 DESIGN FEATURES L 150 DROPOUT (mV) protection. The LTC3026 regulator is available in a low profile 10-lead DFN (3mm × 3mm × 0.75mm) and 10-pin MSOP packages with exposed pad, offering a very compact and thermally efficient solution. Low Voltage Operation and Excellent Ripple Rejection To allow operation at low input voltages, the LTC3026 includes a boost converter that provides the necessary headroom for the internal LDO circuitry as shown in Figure 1. This feature offers the flexibility of lower input voltage capability if an external 5V rail is not available in the system. Figure 2 shows the input ripple rejection for the application circuit of Figure 1. From Figure 2 we see that the ripple rejection is greater than 30dB all the way up to 3MHz. The exceptional ripple rejection makes the LTC3026 an excellent choice for post DC/DC switching supply regulation. 100 1.2V 1.5V 2.0V 2.6V 50 0 0 1.0 0.5 1.5 IOUT (A) Figure 4. Dropout voltage vs output current for various output voltages enabled. Figure 3 shows an application with the boost converter of one LTC3026 driving the boost pin of a second LTC3026 with boost converter disabled. In this application up to five LTC3026 LDOs can be bootstrapped from one LTC3026 with its boost converter enabled. Figure 4 shows that the dropout voltage of the LTC3026 is very low and well controlled over a wide range of output voltages. Multiple VLDO Outputs from Single Boost efficiency dropping the 1.8V input down to 1.5V. In this application the LTC3026 boost converter is disabled and the Boost pin is driven by the external 5V supply. The fast transient response of the LTC3026’s output stage overcomes the traditional tradeoffs between dropout voltage, quiescent current and load transient response inherent in most LDO regulator architectures. Figure 6 shows that the LTC3026 only undershoots about 10mV for a full load transient step (0A to 1.5A), and overshoots about 30mV for a full load transient step (1.5A to 0mA). Additionally, the LTC3026 is designed to detect an overshoot condition and automatically loads the output to bring it back into regulation, as can 1.5A IOUT 0mA OUT AC 20mV/DIV Excellent Transient Response The boost converter of the LTC3026 and Efficiency Post DC/DC can be disabled by connecting the Buck Converter SW pin to ground (GND). Disabling the boost converter allows the Boost pin to be driven by an external 5V supply or from the Boost pin of a second LTC3026 with boost converter Figure 5 shows a typical application for the LTC3026 as post regulator for a buck DC/DC switching converter. The LTC1773 an efficient 1.8V output while the LTC3026 provides over 80% VOUT = 1.5V COUT = 10µF VIN = 1.7V VB = 5V 100µs/DIV Figure 6. Output load transient response 4.5V ≤ VIN ≤ 5.5V 33pF 30k 200pF 1 ITH 0.1µF 2 3 4 CIN 47µF 10V SW 10 LTC1773 5 RUN/SS SENSE– SYNC/FCB VIN VFB TG GND BG RSENSE 0.04Ω 9 8 L1 2.5µH 7 1µF SW BST LTC3026 IN OUT SHDN ADJ GND PG 11k 6 Si9942DY 1µF CBUCK 47µF 10V 100k 1% 80.6k 1% VBUCK 1.8V 2A 100k 4.02k VOUT 1.5V 1.5A COUT 10µF PG CIN, CBUCK: TAIYO YUDEN LMK550BJ476MM L1: CDRH5D28 RSENSE: IRC LR1206-01-R040-F Figure 5. Efficient low noise 1.5V output from 1.8V DC/DC buck converter Linear Technology Magazine • June 2006 31 L DESIGN FEATURES be seen in Figure 6. Once regulation is achieved the part disables the output load thus keeping supply current low. The transient scope photo also shows that the LTC3026 has excellent load regulation. The LTC3026 is designed to be stable with a wide range of ceramic output capacitors as small as 10µF. Conclusion The LTC3026 is an excellent choice for low voltage applications where efficient, low noise supplies are required. The exceptional ripple rejection and very low dropout of the LTC3026 makes it especially well suited as a post regulator for switching supplies. L for the latest information on LTC products, visit www.linear.com LTC2859, continued from page 29 Figure 8 shows single-ended and differential driver outputs in normal and SLO mode with corresponding frequency spectrums operating at 250kbps. SLO mode significantly reduces the high frequency harmonics. The LTC2859 and LTC2861 drivers feature current limiting that protects them from faults such as shorting the outputs to the power supply or ground. Short circuit current is limited to below the ±250mA RS485 standard, with typical clamp currents of ±150mA. If fault voltages are greater than approximately ±10V, currents are reduced further to limit power dissipation. The LTC2859 and LTC2861 also feature thermal shutdown protection, disabling the part if a fault condition causes it to overheat. Figure 9 shows the driver output I-V characteristic when driven by a curve tracer. Overcurrent protection engages on the positive and negative sweeps limiting the driver output current. X, Z Y–Z 10dB/DIV Y–Z 1.25MHz/DIV FREQUENCY SPECTRUM OF SAME SIGNAL NORMAL MODE DRIVER OUTPUT AT 125kHz INTO 100Ω RESISTOR X, Z Y–Z 10dB/DIV Y–Z SLO MODE DRIVER OUTPUT AT 125kHz INTO 100Ω RESISTOR 1.25MHz/DIV FREQUENCY SPECTRUM OF SAME SIGNAL Figure 8. Time and frequency domain waveforms of the LTC2859 driver output in normal and reduced-EMI SLO mode Receiver The LTC2859 features a low power receiver using just 540µA of current (typical). The LTC2859/LTC2861 failsafe feature guarantees the receiver output to be a logic HIGH state when the inputs are shorted, open, or terminated, but not driven for more than about 3µs. The delay prevents signal zero crossings from being interpreted as a shorted input and causing RO to go inadvertently high. This failsafe feature is guaranteed to work for inputs spanning the entire common mode range of –7V to +12V. The receiver output is internally driven high (to VCC) or low (to ground) with no external pull-up needed. The RO pin of the disabled receiver becomes high impedance with leakage 32 50mA/DIV 50mA/DIV DE = 5V DI = 0V 5V/DIV (a) DE = 5V DI = 5V 5V/DIV (b) Figure 9. Curve traced I-V characteristic of the LTC2861 driver output showing current limiting. (a) Pin is driven low by the LTC2861 driver. (b) Pin is driven high by the LTC2861 driver of less than ±1µA for voltages within the supply range. Conclusion Improperly terminated RS485 cabling can severely distort signals leading to losses in data integrity. Correcting network termination without logic- selectable termination often requires physical inspection of expansive networks. The inclusion of selectable termination resistance on the rugged LTC2859 and LTC2861 provide complete solutions to RS485 networking with next-generation remote network tuning capability. L Linear Technology Magazine • June 2006 DESIGN IDEAS L A Complete 500mA Linear Charger and 300mA Synchronous Buck Converter in a Tiny 3mm × 3mm by Ashish Kirtania DFN Package Introduction The LTC4080 is a full-featured, singlecell 4.2V Li-Ion battery charger with an integrated synchronous buck DC/DC converter designed primarily for handheld applications. Its tiny 3mm × 3mm DFN package and low external component count provide space-savings in today’s crowded circuit boards. The high operating frequency (2.25MHz) of the switching regulator minimizes overall solution footprint further by allowing the use of tiny, low profile inductors and ceramic capacitors. To extend battery life, the buck regulator offers high efficiency burst mode operation in which the DESIGN IDEAS A Complete 500mA Linear Charger and 300mA Synchronous Buck Converter in a Tiny 3mm × 3mm DFN Package ......33 Ashish Kirtania Triple Amps Handle High-Res Workstation Video from Single Supplies ............................................35 Jon Munson 500mA Output Current Low Noise Dual Mode Charge Pump ..................36 Yang Wen R3 510Ω D1 VCC (3.75V to 5.5V) VCC R4, 510Ω ACPR D2 CIN 4.7µF LTC4080 SW EN_BUCK FB MODE GND Julian Zhu New Dual Input USB/AC Linear Li-Ion Battery Chargers .....................45 Alfonso Centuori Linear Technology Magazine • June 2006 CPL 10pF PROG R1 715k R2 806k 4.2V Li-Ion BATTERY VOUT (1.5V/300mA) COUT 4.7µF Figure 1. Full featured Li-Ion charger with thermal management and efficient buck regulator in a compact, single IC solution regulator typically consumes only 20µA at no load. ing the PROG pin voltage (VPROG)and applying the following equation: Battery Charger Features ICHRG = The LTC4080 battery charger uses a unique constant-current, constant-voltage, constant-temperature algorithm with programmable charge current up to 500mA and a final float voltage of 4.2V±0.5%. The maximum charge current is programmed using a single external resistor (RPROG) from the PROG pin to ground. The charge current (ICHRG) out of the BAT pin can be determined at any time by monitor100 80 EFFICIENCY (%) 60 40 20 0 0.01 1000 EFFICIENCY (Burst) EFFICIENCY (PWM) VPROG • 400 RPROG In typical operation, the charge cycle begins in constant current mode. When the battery approaches the final float voltage of 4.2V, the charge current starts to decrease as the battery charger switches to constant-voltage mode. When the charge current drops to 10% of the full-scale charge current, commonly referred to as the C/10 point, the open-drain charge status pin, CHRG, assumes a high impedance state. An internal thermal regulator reduces the programmed charge current 100 POWER LOSS 10 (PWM) POWER LOSS (Burst) 0 POWER LOSS (mW) Step-Up/Step-Down Charge Pump DC/DC Converter Provides up to 150mA in a Tiny 2mm × 2mm DFN Package .........43 L1, 1OµH* + *COILCRAFT LPO1704-103M Mitchell Lee Joseph Duncan CBAT 4.7µF RPROG 806Ω Ideal Diodes Combine Battery Stacks (Minimize Heat and Voltage Loss) ......40 High Efficiency, Low Input Voltage, Synchronous Buck Controller Drives up to 15A Load Current .....................41 CHRG EN_CHRG Tiny Buck-Boost Converter for Low Current Applications..................38 Eddy Wells 500mA BAT VBAT = 3.8V 0.1 VOUT = 1.5V L = 10µH C = 4.7µF 0.01 0.1 1 10 100 1000 LOAD CURRENT (mA) Figure 2. Efficiency of the buck regulator in Figure 1 VOUT 20mV/DIV AC COUPLED ILOAD 250mA/DIV I=0 50µs/DIV Figure 3. Transient response of the buck regulator to a 0.5mA–200mA load step 33 L DESIGN IDEAS if the die temperature attempts to rise above a preset value of approximately 115°C. This feature not only protects the LTC4080 and external components from excessive temperature, it can also reduce the charge time by allowing the user to set a higher maximum charge current—essentially taking into account typical, instead of worst case, ambient temperatures for a given application. An internal safety timer sets the maximum time for a charge cycle, typically 4.5 hours. When this time elapses, the charge cycle terminates and the CHRG pin assumes a high impedance state even if C/10 has not been reached yet. A new charge cycle of 2.25hr automatically starts if the battery voltage falls below the recharge threshold (typically 4.1V). Trickle Charge and Defective Battery Detection At the beginning of a charge cycle, if the battery voltage is below 2.9V, the charger goes into trickle charge mode, reducing the charge current to 10% of the programmed value. If the low voltage condition persists for one quarter of the charge cycle (1.125hr), the battery is assumed to be defective, the charge cycle terminates and the CHRG output blinks at a frequency of 2Hz with a 75% duty cycle. If, for any reason, the battery voltage rises above 2.9V, the charge cycle restarts. Undervoltage Lockout An internal undervoltage lockout circuit monitors the input voltage and keeps the battery charger in shutdown until the input rises above 3.6V and approximately 80mV above the battery voltage. The undervoltage condition is indicated by a high-impedance state of the open-drain status output pin ACPR. Undervoltage Charge Current Limiting The battery charger in the LTC4080 includes undervoltage charge current limiting that prevents full charge current until the input supply voltage 34 reaches approximately 300mV above the battery voltage. This feature is particularly useful if the LTC4080 is powered from a supply with long leads or any relatively high output impedance. Buck Converter Features The buck converter in LTC4080 is powered from the BAT pin and has a programmable output voltage (0.8V to VBAT) providing a maximum load current of 300mA. It has two modes of operation, constant frequency mode and Burst Mode operation, selectable via the MODE pin. In constant frequency mode, also referred to as PWM mode, the switching regulator uses current mode control scheme with internal compensation and provides efficiencies up to 91% with very low ripple. The operating frequency of the switching regulator is set at 2.25MHz to minimize possible interference with the AM band. The switching regulator and the battery charger can run simultaneously or independently of each other. Burst Mode Operation Burst Mode operation offers higher efficiency at light loads at the cost of higher ripple at the output voltage. In this mode, the inductor current swings between a maximum value (IPEAK) and a minimum value (IZERO) irrespective of the load as long as the FB pin voltage (VFB) is less than the reference voltage of 0.8V. Once VFB exceeds 0.8V, the control logic turns off both switches along with most of the circuitry and the regulator draws only about 20µA from the battery. When the output voltage drops about 2% from its nominal value, the switching regulator wakes up and the inductor current starts ramping again. To minimize the output voltage ripple, the regulator is limited to a maximum load current of 55mA in Burst Mode operation. Short-Circuit Protection In the event of a short circuit at the output or during start-up, the shallow negative slope (~VOUT/L) of the induc- tor current may prevent the inductor from discharging enough to avoid a cumulative runaway situation over a number of switching cycles. Even the hard current limit on the main PMOS switch is no guarantee against inductor current runaway because of current sense blanking. The switching regulator in the LTC4080 prevents inductor current runaway by imposing a current limit on the synchronous NMOS switch. If the inductor current through the NMOS switch at the end of a discharge cycle is not below this limit, the regulator skips the next inductor charging cycle. Buck Undervoltage Lockout To prevent unreliable operation, when VBAT is less than 2.7V, an undervoltage lockout circuit prevents the switching regulator from turning on. However, if the regulator is already running and the battery voltage is dropping, the undervoltage comparator does not turn it off until VBAT becomes less than 2.5V. Global Thermal Shutdown The LTC4080 includes a global thermal shutdown which turns off the entire part (both battery charger and switching regulator) if the die temperature exceeds 160°C. The part resumes normal operation once the temperature drops approximately 14°C. Conclusion The LTC4080, with its complete Li-Ion battery charger and a moderately high current buck converter in a small 3mm × 3mm package, offers a very compact solution with minimum number of external components. Thermal regulation of the battery charger and the high efficiency of the converter reduce charge times and simplify thermal management. L for the latest information on LTC products, visit www.linear.com Linear Technology Magazine • June 2006 DESIGN IDEAS L Triple Amps Handle High-Res Workstation Video from Single Supplies Introduction Cutting edge, high resolution workstation displays demand cutting edge bandwidth and slew rate specs of their video amplifiers. Displays supporting 1920 × 1200 pixels, for example, must handle over 200Mpixels/s, less than five nanoseconds per pixel! System supply voltages have also been dropping in order to accommodate the new, faster digital processor technologies. Historically the fast amplifiers used for these formats required a total power supply of at least 6V, particularly in cable driver applications. Enter the LT6557 and LT6558, ultra-fast triple video amplifiers with internally fixed gains of two and unity, respectively. These devices have been specifically engineered to operate on single supply voltages down to 3.3V and yet maintain high bandwidth. Now it is practical to use a low voltage digital supply to directly power the analog video circuitry within high resolution products. EN 22µF IN R GND IN R GND R 500Ω 22µF IN G IN G GND G 500Ω 22µF IN B IN B GND B 500Ω LT6557 + – + – + – V+ V+ R OUT G 500Ω V+ G OUT B 500Ω The LT6557 and LT6558 are bipolar voltage-feedback topology parts that are designed for exceptionally high slew-rate and large output swing capabilities for their operating voltage. A blazing slew rate of 2200V/µs is responsible for assuring that 400MHz of bandwidth is available, regardless of signal amplitude. These parts also include a single-resistor-programmable biasing feature that eliminates having to place resistor networks in the signal path to establish the correct DC operating point in single-supply operation. In the typical application for the LT6557 as a cable-driver or the LT6558 as an input buffer, as shown in Figure 1 and Figure 2, the only external components in the signal path are the coupling capacitors and termination resistors, simplifying layout and preventing frequency response anomalies. The output voltage of these amplifiers can swing to within 800mV of either rail, thus there is 1.7V of swing EN BCV OUT R 500Ω Features for Performance and Ease-of-Use V+ B 5V 75Ω 220µF 5V 75Ω Figure 1. An LT6557 single-supply RGB cable driver Linear Technology Magazine • June 2006 75Ω 75Ω 22µF IN G 75Ω 75Ω 22µF 220µF 5V 22µF IN R 220µF 5V 75Ω 412Ω IN B 75Ω 75Ω GND IN R GND R IN G GND G IN B GND B by Jon Munson available on 3.3V, and 3.4V of swing on 5.0V. This means that there is plenty of swing available for RGB or HD video waveforms, plus allowance for offset variations due to AC-coupled picture content, for the unity gain LT6558 to operate on 3.3V or the LT6557 with gain of two to operate on 5V. Figure 3 shows the time response of the LT6558 to 700mVP-P pulses 6ns wide while operating from 3.3V (as in Figure 2). Note that for high fidelity waveform capture, a coupling circuit like that in Figure 1 is used, with a blocking capacitor and the signal measurement taken after a 75Ω double-termination. The same circuit operating at 5V exhibits even less overshoot. The typical current consumption is about 22mA per amplifier, and an enable feature is provided to permit a less than 1mA total current draw when the part is not in use. Both parts are available in a leaded SSOP-16 package or the leadless 5mm × 3mm DFN-16. The DFN model includes a bottom-side ground pad for enhanced thermal performance. continued on page 42 LT6558 + – BCV V+ OUT R V+ R + – OUT G V+ G + – OUT B V+ B 3.3V 205Ω 75Ω 3.3V 75Ω TO VIDEO ADC INPUTS 3.3V 75Ω 3.3V Figure 2. An LT6558 single-supply RGB buffer/ADC driver 35 L DESIGN IDEAS 500mA Output Current Low Noise by Yang Wen Dual Mode Charge Pump Introduction Charge pump (inductorless) DC/DC converters are popular in spaceconstrained applications with low to moderate load current (10mA–500mA) requirements. The devices in the LTC3203 family are low noise, high efficiency regulating charge pumps that can supply up to 500mA of output current from a single 2.7V to 5.5V supply. The LTC3203-1 and LTC3203B-1 produce a selectable fixed 4.5V or 5V output. The LTC3203B produces an adjustable output voltage. The LTC3203-1 features automatic Burst Mode operation at light load to achieve low supply current whereas the LTC3203B and LTC3203B-1 operate at constant frequency to minimize both input and output noise. High switching frequency (1MHz) makes it possible to use only four tiny low cost ceramic capacitors and two resistors for operation. The device also has two user selectable conversion modes for optimizing the efficiency of the charge pump. Additional features include low shutdown current (<1µA), soft-start at power-on and short circuit protection. The LTC3203 family is available in a 10-lead thermally enhanced DFN package, making it possible to build a complete converter in less than 0.04in2. A typical application circuit is shown in Figure 1. Low Noise Operation The constant frequency architecture achieves regulation by sensing the output voltage and regulating the amount of charge transferred per cycle. This method of regulation provides much lower input and output voltage ripple than that of burst mode regulated switched capacitor charge pumps. The LTC3203B and LTC3203B-1 make filtering input and output noise less demanding than burst mode switched capacitor charge pumps where switching frequencies depend on load current and can range over sev36 OUTPUT PROGRAMMING ON/OFF SHDN VSEL LTC3203-1 VIN 2.2µF 2.2µF R1* 100k VIN VOUT C1+ C2+ C1– C2– MODE GND IOUT(MAX) VSEL VOUT 300mA 300mA 500mA 500mA LOW 4.5V HIGH 5V LOW 4.5V HIGH 5V 10µF VOUT 500mA 2.2µF *R1 316k 357k 357k 402k Figure 1. Typical application VIN 20mV/DIV AC-COUPLED VIN 20mV/DIV AC-COUPLED VOUT 20mV/DIV AC-COUPLED VOUT 20mV/DIV AC-COUPLED VIN = 3.6V CIN = 2.2µF COUT = 10µF IOUT = 300mA 2x MODE 500ns/DIV VIN = 4V CIN = 2.2µF COUT = 10µF IOUT = 300mA 1.5x MODE 500ns/DIV Figure 3. Input and output noise in 2× mode Figure 2. Input and output noise in 1.5× mode eral orders of magnitude. The charge pump operates on two phases, where a break-before-make circuit prevents switch cross-conduction. The higher frequency noise due to the non-overlap “notches” is easily filtered by a small input capacitor and PCB parasitic inductance. Figures 2 and 3 shows the low input and output ripple with a 300mA load. The device is powered from a 3.6V input and produces a regulated 4.5V output. The input voltage source has 0.1Ω impedance. stacked on top of VIN and connected to the output. The two flying capacitors operate out of phase to minimize both input and output ripple. Alternatively, in 1.5× mode, it uses a split-capacitor technique rather than doubling. The flying capacitors are charged in series during the first clock phase, and stacked in parallel on top of VIN on the second clock phase. With this technique, the input current is reduced from more than twice the load current to just over 1.5 times the load current, resulting in approximately 25% less input current than what would be required for operating in 2× mode charge pump to drive the same load. Therefore, the efficiency at higher VIN is increased to approximately 90% with VIN at somewhere between 3V and 4V. Figure 4 shows the conversion efficiency at 300mA load current for 4.5V VOUT and 5V VOUT, respectively. Dual Mode Conversion The LTC3203 family offers both 1.5× and 2× boost modes—selected by the mode pin. In the 2× mode, the chip works as a dual-phase regulated voltage doubler. The flying capacitors are charged on alternate clock phases from VIN. While one capacitor is being charged from VIN, the other is Linear Technology Magazine • June 2006 DESIGN IDEAS L LTC4215, continued from page 15 power than a slot with a 20% accurate circuit breaker. Detect Insertion Events via the ENABLE Pin The EN pin can be used to sense the insertion of a board when the LTC4215 is used in backplane resident application. A short pin on the connector pulls EN to ground once the other, longer pins have already been connected. Once the EN pin crosses its falling 1.107V threshold the LTC4215 turns on the external switch after a 100ms debounce delay. Because a falling edge on the EN pin corresponds to the LTC3532, continued from page 18 DC/DC converter may be dynamically programmed by sourcing or sinking 90 80 VOUT = 5V 70 VOUT = 4.5V 60 Conclusion 50 40 30 20 10 0 2.5 3 3.5 4 VIN (V) 4.5 5 5.5 Figure 4. Efficiency vs VIN at 300mA load current the VIN threshold at which the charge pump will switch from 1.5× mode to 2× mode as VIN falls and vice versa. at which point the external switch is turned off with a 1mA current. current at the FB node. Referring to Figure 9, the equation for the input current clamp level is: simplifies the design of Lithium-Ion or multi-cell powered handheld electronics. With a highly efficient automatic Burst Mode operation, the converter maximizes battery life in portable devices with widely varying load requirements. Soft start, programmable switching frequency and external compensation make the LTC3532 suitable to a wide variety of applications. Two package options, an MS10 leaded package and a 3mm × 3mm DFN, plus the ability to operate efficiently at high frequency, enable the designer to minimize board area and component height. L 1.22V R1 • R3 R2 Figure 10 shows VOUT dropping when input current reaches 500mA as the load increases. In USB applications where the input voltage is nominally 5V, a Schottky diode is used to limit peak voltages on the SW1 pin. IIN 500mA/DIV IOUT 500mA/DIV VIN = 4.3V VOUT = 3.6V 10ms/DIV Figure 10. As load increases, the input current is clamped to 500mA using the circuit of Figure 9 Linear Technology Magazine • June 2006 With low operating current, low external parts count and robust protection features, the LTC3203 family is well suited for low power step-up/ step-down DC/DC conversion. The shutdown, dual mode conversion, selectable output voltage and low noise operation features provide additional value and functionality. The simple and versatile LTC3203 family is ideal for moderate power DC/DC conversion applications. L insertion of a new board, the LTC4215 clears the fault register (except for the EN Changed State bit) so that a previously recorded fault does not prevent the new board from starting up. Whenever the EN pin rises or falls, the EN Changed State bit in the FAULT register is set to indicate that a board has either been inserted or removed. A STATUS register bit contains the complement of the state of the EN pin to indicate if a board is present. When the board is unplugged, the short EN pin is the first to disconnect. The EN pin pulls up with an internal 10µA current source until the voltage reaches the rising 1.235V threshold, ICLAMP = VOUT 1V/DIV The 10% hysteresis on the MODE pin prevents the chip from hunting between the two modes. 100 EFFICIENCY (%) The conversion mode should be chosen based on considerations of efficiency, available output current and VOUT ripple. With a given VIN, the 1.5× mode gives a higher efficiency at lower available output current. The 2× mode gives a higher available output current at lower efficiency. Moreover, the output voltage ripple in the 2× mode is lower due to the out-of-phase operation of the two flying capacitors. Typically, at low VIN, the 2× mode should be selected, and at higher VIN, the 1.5× mode should be selected. The MODE pin has a precision comparator. By connecting a resistive divider from VIN to the MODE input pin, the user can accurately program Conclusion Linear Technology’s new LTC3532 synchronous buck-boost converter Conclusion The LTC4215 is a smart power gateway for hot swappable circuits. It provides fault isolation, closely monitors the health of the power path and provides an unprecedented level of control over the inrush current profile. It logs faults, provides real-time status information, and can interrupt the host if necessary. Meanwhile an internal 8-bit ADC continuously monitors board current and voltages. These features make the LTC4215 an ideal power gateway for high availability systems. L 37 L DESIGN IDEAS Tiny Buck-Boost Converter for Low Current Applications by Eddy Wells Introduction One common challenge for many battery powered portable applications is creating a regulated output voltage above or below the input source. Traditional buck-boost approaches, such as a dual inductor SEPIC converter or cascaded regulators, are unacceptable in most portable devices because of their large solution size and low efficiency. Smaller footprint, integrated charge-pump solutions can switch between buck and boost operation, but charge-pumps achieve good efficiency at only a few operating voltages, while efficiency dips below 50% at others. Another compact and simple approach forgoes a portion of the battery capacity and uses a buck (step-down) only solution, but the advantages are hard to justify when much of the battery capacity is not used, as with certain Li-Ion chemistries and a 3.3V output, or with two alkaline cells and a 3.0V or 2.5V output. two small ceramic capacitors, a miniature inductor, and the ThinSOT IC. All versions of the part are available in a thermally enhanced 3mm × 3mm DFN packages. A photo of the LTC3531 demo-board is shown in Figure 1. Generating a Clean 5V from a Noisy USB Cable Figure 1. Compact 3.3V buck-boost application The LTC3531 is a single inductor 200mA buck-boost converter that generates a regulated output voltage from a wide input voltage between 1.8V and 5.5V while maintaining high efficiency. It is an excellent fit for low power applications where a tiny total solution size is required. The LTC3531 is available with fixed outputs [3.0V or 3.3V] or with an adjustable output that can be set between 2.0V and 5.0V. The fixed output versions require only IL 200mA/DIV BOOST BUCK-BOOST BOOST VOUT 0.5V/DIV VIN 0.5V/DIV 5.5V 5V 4.5V 20µs/DIV Generating a clean 5V output from a USB cable or wall adapter can be a challenge when the combination of source impedance and load transients cause noise and voltage droops. USB cable voltage can vary between 5.25V and 4.35V, while the maximum allowed decoupling capacitance on the input is 10µF. The trace labeled “VIN” in Figure 2 shows what can occur with a 4.7µF input capacitor and a 100mA load step from a powered device. The LTC3531 produces a clean 5V output (VOUT) with less than 100mV of peakto-peak ripple using a 22µF VOUT capacitor. VIN and VOUT are DC aligned at 500mV per division in Figure 2, showing significant improvement in noise and voltage droop. Inductor current is also shown with operation in both boost (VIN < VOUT) and buckboost (VIN ≅ VOUT) modes. A complete schematic of the USB to 5V application is shown in Figure 3 along with efficiency and power loss curves versus load current. The Figure 2. Noisy USB cable input to clean 5V output 4.5 95 SW2 VIN VOUT LTC3531 4.7µF R2 1M 22µF FB SHDN ON OFF GND VOUT 5V 200mA R1 324k EFFICIENCY (%) SW1 80 POWER LOSS 75 COKE 1 GRAPHITE/ Li-POLYMER 3.5 3 2.5 70 2 65 0.1 Figure 3. USB to 5V application 38 10 85 POWER LOSS (mW) USB 4.35V TO 5.25V 4 EFFICIENCY VOLTAGE (V) 90 10µH 100 1 10 100 LOAD CURRENT (mA) 0.1 1000 0 20 40 60 USED CAPACITY (%) 80 100 Figure 4. Typical 1C lithium-ion/polymer capacity curves Linear Technology Magazine • June 2006 DESIGN IDEAS L Li-Ion + – SW1 SW2 VIN VOUT LTC3531-3.3 2.2µF VOUT 3.3V 150mA 10µF SHDN GND 350 95 300 90 EFFICIENCY (%) VIN 2.7V TO 5V 100 BOOST MODE 85 80 MAXIMUM IOUT (mA) 10µH BUCK MODE 4SW MODE 75 70 3.3VOUT AT 100mA 60 1.5 2 2.5 3 3.5 4 4.5 INPUT VOLTAGE (V) 5 Maximizing Li-Ion Capacity for 3.3V When compared to a straight buck converter, the LTC3531 allows lower input voltage operation when providing a 3.3V output from a Li-Ion input source. Typical capacity curves for 100 + – LTC3531-3 2.2µF VOUT 3V 80mA 47µF SHDN GND 80 10 2.5VIN 3.2VIN 75 1.8VIN 70 65 ON OFF 60 0.1 1 POWER LOSS AT 3.2VIN 1 10 100 LOAD CURRENT (mA) Figure 7. Two AA or two AAA to 3.0V application 0.1 1000 POWER LOSS (mW) 2 x AA ALKALINE + – SW2 VOUT EFFICIENCY (%) 85 SW1 100 2 2.5 3 3.5 4 4.5 5 5.5 Figure 6. Output current capability vs input voltage (VOUT = 3.3V) coke and graphite anode Li-Ion batteries are shown in Figure 4. Coke types have a lower cut-off voltage at 2.5V, where graphite types have a flatter discharge curve and a 3.0V cut-off. Solid lithium polymer batteries have discharge curves similar to graphite. The equivalent series resistance (ESR) of the Li-Ion battery causes additional voltage drops at the terminal at higher load currents. To make matters worse, the battery protector circuit adds additional series resistance and the effects of ESR lower system efficiency as the battery is discharged. 90 VIN 150 VIN (V) 10µH VIN 1.8V TO 3.2V 200 0 1.5 5.5 Figure 5. Lithium Ion to 3.3V schematic and 100mA efficiency curve. LTC3531 operates in Burst Mode operation, with just 20µA of quiescent current, providing high efficiency over several decades of load current. All four switches have an RDS(ON) of about 0.5Ω when operating at 5V, providing >90% efficiency at higher load currents. 250 50 65 ON OFF L = 10µH VOUT = 3.3V To guarantee a 3.3V output, a buck only design may need to use a cut-off voltage of 3.5V or 3.6V. This translates to a capacity loss of approximately 45% for the coke cell and 20% for the graphite—both significant reductions in run time. Furthermore, while graphite or polymer cells are more popular because of their flat discharge curve, new chemistries with greater capacity per volume are on the horizon with expected discharge curves resembling the coke anode. The wide input voltage range of the LTC3531 allows a regulated 3.3V to be produced from all Li-Ion chemistries, two or three alkaline cells, or a 5V source such as USB. The LTC3531 automatically transitions between buck, 4-switch (buck-boost), and boost modes based on the voltage difference between VIN and VOUT. Figure 5 shows a 3.3V application circuit, along with efficiency vs input voltage for a 100mA load. Maximum load current capability vs input voltage (VOUT = 3.3V) is shown in Figure 6. As expected, efficiency and load current capability are reduced with input voltage. 3.0V Flash Memory Application from Two Alkaline Cells IL 200mA/DIV VOUT 50mV/DIV ILOAD 100mA/DIV VIN = 2.5V 20µs/DIV VOUT = 3V COUT = 47µF LOAD STEP = 10mA TO 100mA Figure 8. Transient response of the circuit in Figure 7 Linear Technology Magazine • June 2006 Inexpensive MP3 players and other relatively low capacity, low cost portable devices often replace a hard disk drive (HDD) with flash memory and Li-Ion batteries with disposable alkaline—a good fit for the LTC3531. A complete schematic for a two cell alkaline to 3V flash memory supply is shown in Figure 7. Efficiency is better continued on page 40 39 L DESIGN IDEAS Ideal Diodes Combine Battery Stacks (Minimize Heat and Voltage Loss) by Mitchell Lee Introduction Modifications Combining multiple battery stacks to serve a common load is an easy task for a diode. Each stack delivers whatever current it can muster to the load, but back-feeding from a fresh battery to one mostly discharged is precluded by the presence of the blocking diode. If you’re concerned with heat dissipation in the diode and voltage drop at end of discharge, diodes may leave you pining for a better solution. Operation over a range of 10V to 36V is practical, and lower voltage operation is feasible by converting to a logic level MOSFET. Because the forward regulation point is a function of the battery voltage, the 10M–10kΩ divider should be adjusted to keep the drop across the 10kΩ resistor in the range of 10mV to 50mV. In Figure 1 the forward drop exceeds 26.4mV when the product of the load current and MOSFET RDS(ON) thus dictate. For the 55mΩ IRF540, this point is reached at load currents of 500mA. Given some finite RDS(ON) there is a practical limit for the load current in any MOSFET, where RDS(ON) • ILOAD ceases to provide any advantage over a diode. In the case of the IRF540 this point arrives in the 5A-to-10A range. For higher current applications, substitute a lower RDS(ON) MOSFET. L Active Diode V+ than 80% with fresh alkaline cells and better than 70% with depleted batteries. Note that overall efficiency is lower in this application (relative to the Li-Ion application) because of the lower drive voltage for the switches—switch RDS(ON) increases with decreasing drive voltage. The adjustable version of the part can be used to power lower voltage flash memories (i.e. 2.5V) with similar performance results. Peak current requirements for flash memory are typically lower than a HDD, since there is no disk spin-up, 10k + 26.4mV – 1M 1M IRF540 LT1494 BAT54W + LTC3531, continued from page 39 10M 26.4V 16 ALKALINE CELLS – The circuit in Figure 1 implements an active, ideal diode using a low resistance MOSFET and a micropower op amp. The MOSFET functionally replaces the diode. Placing it in the negative lead of the battery stack permits use of an N-channel device in a simple arrangement, driven on by an op amp if there is a slight forward voltage and off if the voltage reverses. The forward voltage drop is regulated at 26.4mV, giving freedom from oscillations and preventing reverse current flow. Static current drain for the entire circuit is less than 4µA. 18V 100nF –VOUT Figure 1. Battery pack ideal diode but load transients still occur when memory is accessed. The LTC3531’s response to a 10mA–100mA load step is shown in Figure 8. Burst Mode® operation provides a rapid transient response since there is no compensation loop to slew. Peak-to-peak voltage ripple plus load step is under 50mV with a 47µF output capacitor. The output voltage ripple stays fairly constant over input voltage. Half the output capacitance (22µF) results in approximately twice the load step plus voltage ripple (100mV). Summary The LTC3531 provides a simple, compact buck-boost solution for lower current, portable applications. A complete solution, 1mm in height, can fit in a 35mm2 footprint. The part maintains high efficiency over a wide range of input voltages and load currents, extending battery run time, while providing the flexibility to address many designs such as 2-cell alkaline, USB, and present day or emerging Li-Ion chemistries. L For more information on parts featured in this issue, go to http://www.linear.com 40 Linear Technology Magazine • June 2006 DESIGN IDEAS L High Efficiency, Low Input Voltage, Synchronous Buck Controller Drives by Joseph Duncan up to 15A Load Current Introduction IPRG The LTC3822 is a synchronous stepdown DC/DC converter that drives external N-channel power MOSFETs to maximize average current drive for the lowest cost. Its No RSENSE constant frequency architecture minimizes the number of external components, and a programmable frequency of up to 750kHz allows the use of small surface-mount inductors and capacitors. This DC/DC controller is optimized for 3.3VIN and Lithium-Ion applications allowing VOUT as low as 0.6V while maintaining 1% precision. The all N-channel MOSFET drive simplifies component selection as well as drastically increasing the current capabilities of a typical circuit. Even with 3.3V gate drive, the LTC3822 is capable of controlling more than 15A load current while maintaining high efficiency. RUN ITH LTC3822 SW GND 59k GND VOUT 1.8V 500mV/DIV VIN = 4.2V RLOAD = 1Ω Figure 2. Sample footprint for application circuit in Figure 1 ramps the output voltage from 0V to its final value in 800µs (Figure 3). This is done without the need for an external capacitor. The LTC3822 incorporates No RSENSE technology to sense the inductor current from the drain to source voltage (VDS) of the top-side CMDSH-3 Si4866 0.22µH 0.22µF 47µF 2x VIN 2.75V TO 4.5V VOUT 1.8V 100µF 15A 2x CMDSH-3 BG 118k Figure 4. High current application delivering 1.8V at 15A. Linear Technology Magazine • June 2006 BG 118k VFB 59k VOUT 1.8V 100µF 8A Figure 1. Typical application delivering 1.8V at 8A. Si4866 680pF 0.47µH VFB BOOST 5.1k 0.22µF 680pF TG FREQ SW BOOST 5.1k Figure 1 shows a 1.8V, 8A application that operates over input voltages between 2.75V and 4.5V, perfect for 3.3V or Li-Ion inputs. This application occupies much less space than would be expected for its current capabilities, as shown in Figure 2. During startup, the inter nal soft-start circuitry smoothly VIN LTC3822 VIN 2.75V TO 4.5V FDS6898A TG FREQ ITH CMDSH-3 47µF RUN Compact, 1.8V, 8A Application IPRG VIN 200µs/DIV Figure 3. Internal soft-start ramps the output voltage smoothly without requiring an external capacitor. power MOSFET. The maximum load current that the controller is capable of driving is determined by the RDS(ON) of this MOSFET. Since the LTC3822 incorporates all N-channel MOSFET drive, lower RDS(ON) (and cheaper) devices are available for the top-side MOSFET, when compared to traditional complementary MOSFET drive. Increasing the Current to 20A Figures 4 and 5 show two ways to raise the current capability of the regulator by lowering the RDS(ON) of the MOSFETs. In Figure 4, MOSFETs with a much lower RDS(ON) than those of Figure 1 are used. Because they are in individual SO-8 packages, their thermal capabilities are also higher. This application is designed for a 15A continuous current load. Figure 5 in41 L DESIGN IDEAS 5V SECONDARY SUPPLY CMDSH-3 IPRG VIN 47µF RUN ITH LTC3822 SW 0.22µF 0.47µH IL 2A/DIV VOUT 1.8V 100µF 10A BOOST 5.1k VOUT 100mV/DIV FDS6898A TG FREQ VIN 2.75V TO 4.5V ILOAD 2A/DIV 680pF GND BG VFB 59k VIN = 3.3V VOUT = 1.8V ILOAD = 1A TO 3A 118k 40µs/DIV Figure 6. Transient performance of the converter in Figure 1 Figure 5. High efficiency application deriving gate drive voltage from a secondary 5V supply. stead utilizes a secondary 5V supply to provide a higher gate drive voltage to the MOSFETs. Higher gate drive voltages lower RDS(ON) while simultaneously allowing the use of cheaper logic-level MOSFETs. The maximum load current can also be tailored using the current limit programming pin, IPRG. This three-state pin sets the peak current sense voltage across the top-side MOSFET. Combining all three high current approaches (utilizing low RDS(ON) MOSFETs, powering the gate drive from a secondary 5V supply, and setting current limit to its highest value) enables applications in excess of 20A. The LTC3822 incorporates OPTILOOP® compensation to enable the user to choose optimal component values to compensate the loop over a wide range of operating conditions with the minimum number of output capacitors. Figure 6 shows the tran- Conclusion LT6557, continued from page 35 matically tracks downward with the supply if below 4V. The selection of input bias point may depend on the application, but the values shown for the programming resistors in Figures 1 and 2 are representative of most designs. grammable biasing, these devices offer minimal parts-count AC-coupled amplifier solutions for very high-resolution applications. The LT6557, with its gain of two, is designed for RGB output ports such as in video routers and KVM switch products. The LT6558, with unity gain, is designed as an RGB input port buffer and/or ADC driver, such as in computer or home-theater display products. L The LT6557 and LT6558 are designed specifically with single-supply ACcoupled operation in mind. Each input includes an internal currentcontrolled bias voltage source like that shown in Figure 3. A single external resistor RBCV programs the input bias voltages as shown in Figure 4 for the LT6557. The LT6558 RBCV function is similar to Figure 4, but is optimized for producing higher biasing levels to account for the lower gain and auto- Conclusion The LT6557 and LT6558 triple video amplifiers are optimized specifically for operation on low voltage single supplies. With preset gain and pro- V+ VBCV = 48mV (TYPICAL) VSUPPLY = 3.3V VS = 5V 1.5 1.0 0.5 9.1k 2ns/DIV Figure 3. Fast pulse response of LT6558 on 3.3V single supply 42 VBCV RSET 2.5k IN V • 9.1k VBIAS(IN) = BCV RSET 2.5 2.0 I= OUTPUT 100mV/DIV The LTC3822 delivers currents as high as 20A for single-output applications using a minimum number of components in a tiny complete solution footprint. L INPUT VOLTAGE (V) Automatic Biasing Feature OPTI-LOOP Compensation sient response for the circuit in Figure 1 with a load step of 1A to 3A. The output overshoots by approximately 100mV on a 1.8V output and then settles in about 50µs. Figure 4. Simplified schematic of LT6557 input biasing circuit (LT6558 similar) 0 200 300 400 500 RBCV (Ω) 600 700 Figure 5. Relationship of LT6557 input bias voltage to programming resistor RBCV Linear Technology Magazine • June 2006 DESIGN IDEAS L Step-Up/Step-Down Charge Pump DC/DC Converter Provides up to 150mA in a Tiny 2mm × 2mm DFN Package by Julian Zhu Introduction A wide variety of handheld and portable applications are powered by Li-Ion batteries or AA cells. The wide input voltage range of a single Li-Ion battery (2.7V–4.2V) or 2 AA cells (1.8V–3.0V) requires a DC/DC converter that can step-up or step-down the input voltage to provide a fixed output voltage such as 3.3V or 2.5V. The new LTC3240 step-up/step-down DC/DC converter is ideally suited for such applications and can provide up to 150mA in a tiny 6-lead 2mm × 2mm DFN package. For input voltages greater than the regulated output voltage the LTC3240 operates as a low dropout regulator. When the input voltage decreases to within about 150mV of the regulated output voltage, the LTC3240 automatically switches to step-up Figure 1. LTC3240-3.3 step-up/stepdown converter capable of delivering current up to 150mA mode. In step-up mode, the LTC3240 operates as a constant frequency (1.2MHz) voltage doubling charge pump. The LTC3240 requires only three tiny external ceramic capacitors for an ultra small application footprint as shown in Figure 1. 1µF Li-Ion OR 3-CELL NiMH C+ VOUT C– VIN 2.7V TO 4.5V 1µF LTC3240-3.3 4.7µF GND OFF ON 3.3V IOUT = 150mA SHDN Figure 2. The regulated 3.3V from battery voltage 100 IOUT = 30mA 80 3.40 70 3.35 3.30 3.25 40 10 0 2.2 2.7 3.2 3.7 4.2 4.7 INPUT VOLTAGE (V) 5.2 5.7 Figure 3. Output voltage vs input voltage (full range) Linear Technology Magazine • June 2006 VOUT 20mV/DIV DOUBLER TO LDO MODE (VIN RISING) 30 20 A typical application circuit for LTC3240-3.3 is shown in Figure 2. The input can be a single Li-Ion battery or three AA cells. Figure 3 shows the output voltage variation for the entire input voltage range at a load current of 30mA. A new or recharged battery starts out at its highest terminal voltage. As the battery discharges, its terminal voltage continues to drop until the next recharge. The LTC3240 optimizes the output efficiency by continuing to operate in the step-down (LDO mode) for most of the battery life. When the battery voltage gets low enough it automatically switches into chargepump mode to squeeze out maximum energy from the battery before its next ILOAD = 1mA 50 3.20 Application for Li-Ion or Three AA Battery Input to 3.3V Out ILOAD = 40mA 60 3.15 3.10 1.7 LDO TO DOUBLER MODE (VIN FALLING) 90 3.45 EFFICIENCY (%) OUTPUT VOLTAGE (V) 3.50 The LTC3240 features low no load operating current (65µA typical) and ultra low shutdown current (<1µA). Built-in soft-start circuitry prevents excessive inrush current during start-up. The thermal-shutdown and current-limit circuitry allow the parts to survive a continuous output short-circuit. 1.8 2.3 2.8 3.3 3.8 4.3 4.8 SUPPLY VOLTAGE (V) 5.3 5.8 Figure 4. Efficiency vs input voltage (LTC3240-3.3) VIN = 2.5V ILOAD = 150mA 2µs/DIV Figure 5. Output voltage ripple (LTC3240-3.3) 43 L DESIGN IDEAS LT3486, continued from page 11 as 70V while both providing both overvoltage protection and remaining below the 42V maximum switch voltage. The charge pump Schottky diodes and capacitors double the effective output voltage for a given duty cycle while the LT3486 LED driver continues to regulate the 100mA constant LED current. The LEDs in Figure 7 have higher forward voltage than those in Figure 1 at 100mA, ripple voltage (<1% for 3.3V output) at 150mA (See Figure 5). To extend battery life at light loads, in charge pump mode, the part operates in high efficiency Burst Mode operation. In this case, the LTC3240 delivers a minimum amount of charge for a few cycles, and then enters a low current state until the output drops low enough to require another burst of charge. Conclusion resulting in a total string voltage as high as 40V. If more LEDs are needed, the string voltage can be stacked up to 70V before hitting the overvoltage protection level, but the peak switch current limit cannot be exceeded. As the string voltage and LED current goes up, the minimum input voltage also rises. Figure 8 shows the typical peak switch current limit dropping as duty cycle increases. In addition to the peak inductor current, the voltage doubler also adds additional charge pump capacitor current. ZLLS400 ZLLS400 COUT1B 2.2µF ZLLS400 25V PVIN 8V TO 18V CPV(IN) 4.7µF 25V 2.2µF 25V ZLLS400 10µH A915AY-100M COUT1A 2.2µF 25V 100mA 10 WHITE LEDs >34V CTRL1 BRIGHTNESS ADJUST SHDN VIN OVP2 CTRL1 CTRL2 FB1 VC1 GND RT PWM1 2.8k PWM INPUT 4700pF 100Hz 5V 1000:1 DIM RATIO ZLLS400 COUT2A 2.2µF 35V REF 0.1µF CTRL2 BRIGHTNESS ADJUST FB2 63.4k The LT3486 is a dual 1.3A LED string driver with 1000:1 PWM dimming capability. Its 3% LED current accuracy, low sense voltage, low shutdown current, overvoltage protection and wide input voltage range make it ideal for high power LCD panels in a variety of applications including automotive displays and notebook computers. The simple 5:1 analog dimming ratio and more precise 1000:1 PWM dimming ratio provide the displays with enough brightness control for daylight and nighttime use while retaining their color characteristics across brightness levels. L 1800 FDN5630 PWM2 VC2 Conclusion 100mA 10 WHITE LEDs >34V SW2 OVP1 PWM1 100k 2.2µF 25V CV(IN) 4.7µF 25V SHDN LT3486EFE FDN5630 RSENSE1 2Ω 1% 10µH A915AY-100M VIN 5V SW1 COUT2B 2.2µF 25V ZLLS400 The LTC3240 step-up/step-down charge pump DC/DC converter provides fixed regulated output voltage with currents up to 150mA from a wide input voltage range in a small 6-lead 2mm × 2mm DFN package. It is ideally suited for efficient DC/DC conversion in space-constrained applications such as battery-powered handheld electronics. L PEAK SWITCH CURRENT LIMIT (mA) recharge cycle. Figure 4 shows the efficiency of LTC3240-3.3 as a function of input voltage. In step-up (charge pump) mode, the LTC3240 uses a unique architecture to optimize the charge transferred to the output in each clock cycle, thus minimizing the output ripple. The part only needs a 4.7µF, 0603 size ceramic capacitor to obtain a 32mV maximum PWM2 100k RSENSE2 2Ω 1% 2.8k 4700pF PWM INPUT 100Hz 5V 1000:1 DIM RATIO Figure 7. LED driver uses 8V–18V input to drive two strings of ten 100mA LEDs (40V max per string) with 1000:1 PWM dimming VIN = 3.6V 1700 1600 1500 1400 1300 1200 20 30 40 50 60 70 80 DUTY CYCLE (%) 90 100 Figure 8. Typical peak switch current limit drops as duty cycle increases above 50% For more information on parts featured in this issue, go to http://www.linear.com 44 Linear Technology Magazine • June 2006 DESIGN IDEAS L New Dual Input USB/AC Linear by Alfonso Centuori Li-Ion Battery Chargers Introduction Digital Cameras, PDAs, mobile phones and MP3 players all use batteries that are commonly charged via either a wall adapter or through the USB. The LTC4076 and LTC4077 lithium-ion battery chargers are specifically designed to detect power at the inputs and automatically select the appropriate source for charging. Using a constant current/constant voltage algorithm, the chargers can be programmed to deliver up to 950mA of charge current with a final float voltage accuracy of ±0.6%. The LTC4076 and LTC4077 include an internal Pchannel power MOSFET and thermal regulation circuitry with no blocking diode or external sense resistor required. Thus, the basic dual-source charger requires only three external components. The LTC4076 offers a programmable current based termination scheme. The CHRG open-drain status pin can be programmed to indicate the battery charge state according to the needs of the application. The PWR open-drain status pin indicates that enough voltage is present at one of the inputs to charge a battery. With power applied on both inputs, LTC4076 and LTC4077 can be put into shutdown mode, reducing the DCIN supply current to 20µA, the USBIN supply current to 10µA and the battery drain current to less than 2µA. Internal thermal feedback regulates the charge current to maintain IBAT = 475mA IBAT = 100mA WALL ADAPTER USB POWER IBAT = 475mA IBAT = 95mA WALL ADAPTER USB POWER IDC 1.24k 1% 1k 1.24k 1% 1k 1µF IDC 2.1k 1% + CHRG ITERM GND 1-CELL Li-Ion BATTERY 1k 1% Figure 1. An LTC4076 USB/wall adapter Li-Ion charger a constant die temperature during high power operation or high ambient temperature conditions. USB Compatibility The HPWR pins of the LTC4076 and LTC4077 provide an easy method to choose between two different USB power modes: high power (usually ≤ 500mA) and low power (usually ≤ 100mA). With the LTC4076, a logic high on the HPWR pin sets the charge current to 100% of the current programmed by the IUSB pin resistor, while a logic low on the HPWR pin sets the charge current to 20% of the current programmed by the IUSB pin resistor. With the LTC4077, a logic high on the HPWR pin sets the charge current to the value programmed by the IUSB pin resistor, while a logic low on the HPWR pin sets the current to the value programmed by the IUSBL pin resistor. 800mA (WALL) 475mA (USB) 100mA (USB) BAT 1k CHRG IUSBL GND A weak pull down on the HPWR pin sets the part to default to the low power state. The HPWR pin provides a simple control for managing charge current as shown in Figure 1 and Figure 2. The presence of a wall adapter takes priority over the USB input and the IDC 1.24kΩ resistor sets the total maximum charge current to 800mA. When USB power is present, and a wall adapter is not, the IUSB 2.1kΩ resistor sets the charge current to 476mA with HPWR in its high state, and 95mA with HPWR in its low state for the LTC4076 (or 100mA with IUSBL 2kΩ resistor for the LTC4077). Programmability The LTC4076 and LTC4077 provide a great deal of design flexibility including programmable charge current and programmable current termination. The charge currents are programmed using a resistor from the IDC, IUSB and IUSBL (LTC4077 only) pins to ground as indicated in the following equations: ICHG = 1000 V RIDC (Wall Adapter Present) ICHG = 1000 V RIUSB (USB HPWR = High) ICHG = 200 V RIUSB (USBHPWR = Low, LTC4076 Only) ICHG = 200 V RIUSBL (USBHPWR = Low, LTC4077 Only) 1k + PWR 2.1k 1% 1µF 1µF IUSB BAT USBIN PWR USBIN 1µF DCIN IUSB HPWR LTC4077 DCIN 800mA (WALL) 475mA (USB) 95mA (USB) HPWR LTC4076 1-CELL Li-Ion BATTERY 2k 1% Figure 2. An LTC4077 USB/wall adapter Li-Ion charger Linear Technology Magazine • June 2006 45 L NEW DEVICE CAMEOS LTC4076 OR LTC4077 WALL ADAPTER USB POWER BAT DCIN 1µF USBIN 1µF 500mA + IUSB RISET 2k 1% IDC GND Figure 3. LTC4076 or LTC4077 dual input charger circuit. The wall adapter charge current and the USB charge current (HPWR = High) are both programmed to be 500mA with just one resistor. Both the LTC4076 and LTC4077 terminate the charge cycle based on the battery current. For the LTC4076 the current threshold is programmable and for the LTC4077 the current threshold is fixed, as described below. For LTC4076, the programmable current detection threshold, ITERM, is set by connecting a resistor, RITERM, from ITERM to ground. The following formula programs the termination current: ITERM = 100 V RITERM For the LTC4077 the termination current is fixed at 10% of the programmed charge current as set by IDC or IUSB (HPWR = High). When HPWR in its low state, the termination current is 50% of the current programmed by RIUSBL. The condition of the CHRG pin indicates the charge state. A strong pull-down on the CHRG pin indicates that the battery is charging. When the current termination threshold is reached the CHRG pin assumes a high impedance state. Avoiding Unnecessary Charge Cycles LTC4076 and LTC4077 are designed to avoid unnecessary charge cycles to extend the life of Li-Ion batteries. When power is first applied or when exiting shutdown, the LTC4076 and LTC4077 check the voltage on the BAT pin to determine its initial state. If the BAT pin voltage is below the recharge threshold of 4.1V (which corresponds to approximately 80%–90% battery capacity), LTC4076 and LTC4077 enter charge mode and begin a charge cycle. If the BAT pin is above 4.1V, the battery is nearly full and the charger does not initiate a charge cycle and instead enters standby mode. When in standby mode, the chargers continuously monitor the BAT pin voltage. When the BAT pin voltage drops below 4.1V, the charge cycle is automatically restarted. This feature eliminates the need for periodic charge cycle initiations, ensures that the battery is always fully charged, and prolongs battery life by reducing the number of unnecessary charge cycles. Conclusion LTC4076 and LTC4077 are complete Linear Li-Ion battery chargers compatible with portable USB applications. They are designed to accommodate charging from both a wall adapter and a USB input. The versatility, low quiescent current, simplicity, high level of integration and small size of the LTC4076 and LTC4077 provide an ideal choice for many portable USB applications. LTC4076 and LTC4077 are available in a small 10-lead low profile 3mm × 3mm DFN package. L New Device Cameos New Timing and High Voltage Output Enhance Push Button On/Off Controller The LTC2954 is an upgrade to the LTC2950 family of push button controllers. Two features have been added: new push button controlled interrupt timing, and a high voltage open drain enable output capable of 33V operation. The LTC2954 now provides interrupt and power down modes to allow more flexible and reliable system shut down. Once in the powered on state, momentary presses on the push button provide interrupts to system logic. This can be useful for system housekeeping and power down under 46 firmware control. Should the system require a forced power down, a long duration press and hold of the push button automatically releases the enable pin, thus turning system power off. The timing of this latter power down mode is adjustable with external capacitance on the power down timing pin. The new high voltage enable output of the LTC2954-2 is ideally suited to drive the gate of a high voltage power PFET. This allows a user to connect/ disconnect input power from its load simply by toggling the de-bounced push button input. The LTC2954 operates over a wide 2.7V to 26V input voltage range while consuming only 6µA of supply current and is available in 8-Pin 3mm × 2mm DFN and ThinSOT™ Packages. Two versions of the part are available to accommodate either positive or negative enable polarities. L For further information on any of the devices mentioned in this issue of Linear Technology, use the reader service card or call the LTC literature service number: 1-800-4-LINEAR Ask for the pertinent data sheets and Application Notes. Linear Technology Magazine • June 2006 DESIGN TOOLS L DESIGN TOOLS Product Information Applications Handbooks Linear Technology offers high-performance analog products across a broad product range. Current product information and design tools are available at www.linear.com. Our CD-ROM product selector tool, which is updated quarterly, and our most recent databook series can be obtained from your local Linear Sales office (see the back of this magazine) or requested from www.linear.com. Linear Applications Handbook, Volume I — Almost a thousand pages of application ideas covered in depth by 40 Application Notes and 33 Design Notes. This catalog covers a broad range of real world linear circuitry. In addition to detailed, systems-oriented circuits, this handbook contains broad tutorial content together with liberal use of schematics and scope photography. A special feature in this edition includes a 22-page section on SPICE macromodels. www.linear.com Product information and application solutions are available at www.linear.com through powerful search tools, which yield weighted results from our data sheets, application notes, design notes, Linear Technology magazine issues and other LTC publications. The LTC website simplifies the product selection process by providing convenient search methods, complete application solutions and design simulation programs for power, filter, op amp and data converter applications. Search methods include a text search for a particular part number, keyword or phrase, or a powerful parametric search engine. After selecting a desired product category, engineers can specify and sort by key parameters and specifications that satisfy their design requirements. Visit www.linear.com/mylinear to register and access your MyLinear home page. Here you can store your favorite LTC products, categories, product tables, contact information, preferences and more. Purchase Products Online Credit Card Purchases—Purchase online direct from Linear Technology at www.linear.com using a credit card. Create a personalized account to check order history, shipment information and reorder products. Linear Express Distribution — Get the parts you need. Fast. Most devices are stocked for immediate delivery. Credit terms and low minimum orders make it easy to get you up and running. Place and track orders online. Apply today at www.linear.com or call (866) 546-3271. CD-ROM The June 2006 CD-ROM contains product data sheets, application notes and Design Notes. Use your browser to view product categories and select products from parametric tables or simply choose products and documents from part number, application note or design note indexes. Linear Applications Handbook, Volume II — Continues the stream of real world linear circuitry initiated by Volume I. Similar in scope to Volume I, this book covers Application Notes 40 through 54 and Design Notes 33 through 69. References and articles from non-LTC publications that we have found useful are also included. Linear Applications Handbook, Volume III — This 976-page handbook includes Application Notes 55 through 69 and Design Notes 70 through 144. Subjects include switching regulators, measurement and control circuits, filters, video designs, interface, data converters, power products, battery chargers and CCFL inverters. An extensive subject index references circuits in Linear data sheets, design notes, application notes and Linear Technology magazines. Brochures Power Management for Portable Products — The solutions in this product selection guide solve real-life problems for cell phones, digital cameras, PDAs and other portable devices, maximizing battery run time and saving space. Circuits are shown for Li-Ion battery chargers, battery managers, USB support, system power regulation, display drivers, white LED drivers, photoflash chargers, DC/DC converters and RF PA power supply and control. Automotive Electronic Solutions — This selection guide features high performance, high reliability solutions for a wide range of functions commonly used in today’s automobiles, including telematics, infotainment systems, body electronics, engine management, safety systems and GPS navigation systems. Industrial Signal Chain — This product selection guide highlights analog-to-digital converters, digital-to-analog converters, amplifiers, comparators, filters, voltage references, RMS-to-DC converters and silicon oscillators designed for demanding industrial applications. These precise, flexible and rugged devices feature parameters fully guaranteed over the –40°C to 85°C temperature range. Wireless & RF Solutions — This brochure presents high performance RF solutions for use in various transceiver architectures employed in 2G, 2.5G and 3G cellular basestations, wireless point-to-point radios, WiMAX, broadband wireless access, satellite receivers, GPS receivers, cable and VOD infrastructure equipment, RFID readers, wireless handheld transceivers and software defined radios. Software SwitcherCAD™ III/LTC SPICE — LTC SwitcherCAD III is a fully functional SPICE simulator with enhancements and models to ease the simulation of switching regulators. This SPICE is a high performance circuit simulator and integrated waveform viewer, and also includes schematic capture. Our enhancements to SPICE result in much faster simulation of switching regulators than is possible with normal SPICE simulators. SwitcherCAD III includes SPICE, macromodels for 80% of LTC’s switching regulators and over 200 op amp models. It also includes models of resistors, transistors and MOSFETs. With this SPICE simulator, most switching regulator waveforms can be viewed in a few minutes on a high performance PC. Circuits using op amps and transistors can also be easily simulated. Download at www.linear.com FilterCAD™ 3.0 — FilterCAD 3.0 is a computer aided design program for creating filters with Linear Technology’s filter ICs. FilterCAD is designed to help users without special expertise in filter design to design good filters with a minimum of effort. It can also help experienced filter designers achieve better results by playing “what if” with the configuration and values of various components and observing the results. With FCAD, you can design lowpass, highpass, bandpass or notch filters with a variety of responses, including Butterworth, Bessel, Chebychev, elliptic and minimum Q elliptic, plus custom responses. Download at www.linear.com SPICE Macromodel Library — This library includes LTC op amp SPICE macromodels. The models can be used with any version of SPICE for analog circuit simulations. These models run on SwitcherCAD III/LTC SPICE. Noise Program — This PC program allows the user to calculate circuit noise using LTC op amps, determine the best LTC op amp for a low noise application, display the noise data for LTC op amps, calculate resistor noise and calculate noise using specs for any op amp. Battery Charger Solutions — This guide identifies optimum charging solutions for single-cell batteries, multi-cell batteries and battery packs, regardless of chemistry. Linear offers a broad range of charger solutions, including linear chargers, linear chargers with regulators, pulse chargers, switchmode monolithic chargers, switchmode controller chargers, and switchmode smart battery chargers. 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