V16N2 - JUNE

LINEAR TECHNOLOGY
JUNE 2006
IN THIS ISSUE…
COVER ARTICLE
Reduce Charge Time
for High Capacity Li-Ion Batteries
with 2A Continuous Charging .............1
Tom Hack
Issue Highlights ..................................2
Linear Technology in the News….........2
DESIGN FEATURES
650MHz Selectable-Gain Amplifier/
Differential ADC Driver Has Small
Form but Many Functions ...................6
Cheng-Wei Pei
Dual Step-Up Converter Drives White
LEDs with 1000:1 PWM Dimming ......10
Keith Szolusha
Hot Swap™ Controller Monitors and
Reports Power Supply Status ............12
Josh Simonson
Efficient Buck-Boost Converter Ideal
for Power Saving Modes and Wide Input
Voltage Ranges ..................................16
Kevin Ohlson
Dual/Triple Power Supply Monitor
for Undervoltage and Overvoltage
on Positive and Negative Supplies .....19
Andrew Thomas
High Speed Low Power RS485
Transceivers with Integrated
Switchable Termination ....................27
Ray Schuler and Steven Tanghe
VOLUME XVI NUMBER 2
Reduce Charge Time
for High Capacity
Li-Ion Batteries with 2A
Continuous Charging
Introduction
The latest high capacity Li-Ion batteries meet the needs of power hungry
portable devices, but they also increase the demands placed on battery
chargers—demands that can be too
much for a standard linear charger.
For instance, a linear charger, operating at 1A charging current, charges a
1Ahr battery to 70% capacity within
one hour, and fully charges it within
three hours. Newer 2Ahr batteries need
twice that current in order to be fully
charged in the same amount of time.
The problem is that a linear charger
operating at 2A produces too much
heat for continuous charging—it’s just
too inefficient. The LTC4001 solves
this problem by incorporating a high
1.5A VLDO™ Operates Down to 0.4V
Output and Maintains 100mV Dropout
.........................................................30
Bill Walter
DESIGN IDEAS
....................................................33–46
(complete list on page 33)
New Device Cameos ...........................46
Design Tools ......................................47
Sales Offices .....................................48
Figure 1. A typical LTC4001-based
Li-Ion battery charger occupies
minimal board real estate.
by Tom Hack
efficiency PWM charger to perform
continuous 2A battery charging. It
works with both standard and currentlimited wall adapters—where the latter
lowers battery charger dissipation and
operating temperature.
Big Features; Small Footprint
A full-featured battery charger based
on the LTC4001 requires an area not
much larger than a dime (Figure 1).
Fully programmable timer and charge
rate terminations are included. Automatic battery “topping off” is also
included. Filtering prevents accidental
recharge from occurring in noisy environments (such as found in GPRS
cellular phones). The LTC4001 works
readily with NTC thermistors for battery temperature sensing. Remote
battery sensing is included. Soft-start
is fully programmable. The LTC4001
also drives charge status LEDs and
provides logic signals for microprocessor-based designs.
The LTC4001 is tiny, fitting into a
4mm × 4mm package, but other factors also contribute to the charger’s
small footprint. High operating frequency (1.5MHz) reduces the size of
the inductors and capacitors. Input
short circuit blocking is built in so
no external diode is required. Current
continued on page 3
L, LT, LTC, LTM, Burst Mode, OPTI-LOOP, Over-The-Top and PolyPhase are registered trademarks of Linear Technology
Corporation. Adaptive Power, BodeCAD, C-Load, DirectSense, Easy Drive, FilterCAD, Hot Swap, LinearView, µModule,
Micropower SwitcherCAD, Multimode Dimming, No Latency ΔΣ, No Latency Delta-Sigma, No RSENSE, Operational Filter,
PanelProtect, PowerPath, PowerSOT, SmartStart, SoftSpan, Stage Shedding, SwitcherCAD, ThinSOT, True Color PWM,
UltraFast and VLDO are trademarks of Linear Technology Corporation. Other product names may be trademarks of the
companies that manufacture the products.
L EDITOR’S PAGE
Issue Highlights
H
igh capacity Li-Ion batteries
meet the needs of power hungry
portable devices, but they also
increase the demands placed on battery chargers—demands that can be
too much for a standard linear charger.
The problem is that a linear chargers
cannot efficiently produce enough current to quickly charge the battery. To
solve this problem, the LTC4001 uses
a high efficiency PWM charger, which
makes continuous 2A battery charging
practical. See our cover article for more
about this breakthrough device.
Featured Devices
Below is a summary of the other devices featured in this issue.
The LT6411 selectable-gain differential amplifier/ADC driver is a good
fit for power-critical high-speed signal
chain applications. It can produce
gains of 1, –1, and 2 with no external
components. The dual amplifiers inside
the LT6411 allow for easy singleended-to-differential conversion for
driving high-speed analog-to-digital
converters (ADC). (Page 6)
The LT3486 dual LED string driver
has two 1.3A channels with high PWM
dimming capability in a small 5mm ×
3mm DFN package. Since both channels’ power switches are included in
the IC, the circuit is simple and small.
(Page 10)
The LTC4215 combines a robust
Hot Swap circuit with an I2C interface and data converter to allow
power monitoring as well as hot-plug
functionality and fault isolation.
(Page 12)
The LTC3532 is a 300mA buckboost converter, which incorporates
automatic Burst Mode® operation,
adjustable switching frequency, and
integrated soft-start. The LTC3532 is
ideal for miniature disk-drive applications or any application that requires
high efficiency over a wide range of
output currents and input voltages.
(Page 16)
The LTC2909 is a highly customizable power supply monitoring
solution with adjustable input thresh2
olds, input polarity selection, a
multi-mode reset timer, and an opendrain RST output. Adjustable input
thresholds allow the user to set any
trip threshold for the comparator,
subject only to the accuracy limitations
of the part, instead of having to pick
from a factory-set limited collection of
thresholds. (Page 19)
The LTC2859 and LTC2861 combine a logic-selectable integrated
termination resistor with a rugged
20Mbps RS485/RS422 transceiver,
providing a single die impedancematched network solution in a tiny
package. (Page 27)
The LTC3026 is a 1.5A VLDO with
input voltage capability down to
1.14V and a low adjustable output
voltage from 0.4V to 2.6V. The part
also has a very low dropout voltage
of only 100mV while delivering up to
1.5A of output current, enabling it to
optimize battery run time from single
cell applications with a low VIN to VOUT
differential. (Page 30)
Design Ideas and Cameos
Design Ideas start on page 33, including a better way to combine battery
packs using an ideal diode—improving
on standard diode solutions. L
Linear Technology in the News…
Linear’s LTM4600 µModule
Racks Up Awards
Linear Technology’s LTM4600 10A
DC/DC µModule™ has been steadily
collecting major product awards.
The LTM4600 was recently awarded
EDN magazine’s Innovation of the
Year award in the Power Systems
and Modules category, and named
Product of the Year by both Electronic
Products magazine and AnalogZone.
To that, add an Innovation Award
from China’s Electronic Engineering & Product World, and a Product
Award from EDAW/ Nikkei Electronics in China.
Linear Expands
Design Capabilities
Linear Technology has further expanded its analog design resources
in two new locations. In April, Linear announced the opening of the
company’s first European design
center in Munich.
Lothar Maier, CEO of Linear
Technology, stated, “We are excited
to open our first European design
center in the key technology hub of
Munich. By centrally locating our
latest design center on the European
continent, this puts us in an even
better position to develop the right
products for our major European
customers. Germany represents an
ideal location for our first European
design center, with its focus on the
automotive, industrial and communications markets.”
And in May, the company opened
a new design center in Dallas. With
this announcement, Linear now has
12 design centers worldwide.
Linear Announces
Digital Power Alliance
Linear Technology has announced
an alliance with Primarion, Inc. to
produce digital point-of-load (POL)
products for the networking, computing and telecommunications
infrastructure markets. Linear’s
LTC7510 digital POL controller is the
first in a family of PMBus-compliant digital DC/DC controllers. This
family of digital power management
and conversion products provides
flexible, system-level control of
sophisticated power management
systems. These controller ICs ensure
seamless power management in
high availability systems, enabling
real-time programming and monitoring of key parameters to optimize
performance and maximize system
uptime. L
Linear Technology Magazine • June 2006
DESIGN FEATURES L
A Bare Bones Charger
LTC4001, continued from page 1
sensing is internal, so there is no need
for an expensive milliohm-sized current sense resistor.
Inside the LTC4001
The LTC4001 is the basis for a
complete 2A Li-Ion battery charger
(Figure 2). A 50mA linear charger
provides cell conditioning while a
synchronous buck charger provides
constant-current/constant-voltage
high rate charging (up to 2A). Protection and lockouts guard against a
variety of events including: shorts at
the battery and wall adapter inputs;
improper programming of the charge
current; open battery and/or overvoltage battery; defective battery;
insufficient wall adapter voltage; chip
over-temperature; battery over- or
under- temperature.
1.5µH
WALL ADAPTER
4.5V TO 5.5V
+
BATSENS
BAT
10µF
+
4.2V
Li-Ion
LTC4001
CHRG
EN
NTC
SS GNDSENS
PROG IDET TIMER
274Ω
0.1µF
Figure 3. A bare bones battery charger
3
5
50mA
SENSE
CURRENT
REVERSAL
COMPARATOR
+
Q
1
2
PGND SW
–
RAMP
SENSE
FAULT
DRIVER
S
PWM
COMPARATOR
SW
10µF
MICROPROCESSOR
INTERFACE
PVIN
CLK
VINSENSE
PVIN
PGND
8
OSCILLATOR
indicator lights, battery temperature
monitoring, and a timer (which may
be provided by a microprocessor). In
place of a timer, charge terminates
when charge current drops below onetenth the high rate charge current (an
Figure 3 shows a bare bones 2A battery charger. With only five additional
components, this charger offers a
high efficiency, high power solution.
This implementation leaves out status
RD
9
BAT
16
VINSENSE
BATSENS
OVERCURRENT
COMPARATOR
SHUTDOWN
COMPARATOR
+ –
– +
–
SS
SHUTDOWN
LOW BATTERY
OVERCURRENT
PWM ON
TRICKLE ON
LOGIC
FAULT
FAULT
TIMER
TIMER
NTC
NTC
COMPARATOR
CHIP
OVERTEMP
COMPARATOR
CHARGE CURRENT
ERROR AMP
– +
–
DISCHARGE SS
PROG SHORTED
11
PROG
ERROR
AMP
+
1.2V
CHRG
TFAULT
PROG SHORT
COMPARATOR
FLOAT VOLTAGE
ERROR AMP
SOFT-START
COPMPARATOR
1.1V
+
–
15
RECHARGE
COMPARATOR
LOW CURRENT
VIN GOOD
RECHARGE
+
–
10
EN
CHRG
+
–
– +
+
–
7
EN
UNDERVOLTAGE
COMPARATOR
+
–
6
IDET
COMPARATOR
+
–
14
LOW-BATTERY
COMPARATOR
SS
SS LOW
150mV
OVERVOLTAGE
CHIP OVER TEMP
BATTERY
OVERVOLTAGE
COMPARATOR
+
–
CONNECT
VOLTAGE
REFERENCE
GND
17
IDET
13
1.2V
PROG
12
GNDSENS
4
Figure 2. Simplified block diagram of the LTC4001 Li-Ion battery charger
Linear Technology Magazine • June 2006
3
L DESIGN FEATURES
The LED also indicates when the
battery is nearly full charged. As the
battery approaches the float voltage
and charge current drops below the
IDET threshold the LED is dimly lit. This
is difficult to see, so a better approach
uses two LEDs to indicate all charger
states (Figure 5).
VIN
D1
CHRG
R1
1k
LTC4001
CHRG
Interfacing with
Microprocessors
Figure 4. A simple status indicator
VIN
R1
27k
Q1
2N3906
Q2
TP0610
D1
GRN
C/X
D2
AMB
CHRG
R2
1k
R3
1k
LTC4001
CHRG
Q3
2N7002
Figure 5. Full featured status indication
IDET threshold equal to 200mA in this
case). Internal charge termination may
be completely defeated by connecting
the timer pin to the IDET pin instead
of ground (allowing a microprocessor
complete control of charge termination).
Adding Status Lights
The CHRG pin indicates a variety of
charger states (Table 1). Adding a
resistor and LED in series with this
pin to VIN (Figure 4) indicates charger
off (LED off), high rate charging or
cell conditioning (LED on continuously at high brightness), and battery
temperature out of range/NTC fault
(LED blinking).
The interface in Figure 6 can distinguish between all states available on
the CHRG pin. To detect cell conditioning or high rate charging, force
the digital output pin, OUT, high and
measure the voltage on the CHRG pin.
The N-channel mosfet pulls CHRG low
even with a 2k pull-up resistor. Near
end of charge, the NMOS turns off,
and CHRG sinks only 30µA. The IN
pin is pulled high by the 2k resistor
connected to OUT. If OUT is placed
into a high impedance state, the 30µA
sink current from the CHRG pin pulls
IN low. When charging stops, CHRG
opens and OUT stays high, even with
a 390k pull up resistor.
If a battery temperature fault occurs during high rate charging, the
CHRG pin blinks using a serrated
pulse pattern. Nominal timing of this
pattern is given in Figure 7. The extra
edges provide rapid indication to a
microprocessor and may be used to
drive a microprocessor interrupt line
for low processor overhead, but still
provide for a visible fault indication
when using LEDs.
Operation with Conventional
and Current Limited
Wall Adapters
Wall adapters with or without current limiting may be used with the
LTC4001, but the lowest power dissipation battery charging occurs with
a current limited wall adapter. To use
this feature, program the LTC4001
above the wall adapter current limit.
For example, if the wall adapter current
limit is 2A, set the LTC4001 charge
current slightly higher than 2A (allowing for tolerances).
To understand operation with a
current limited wall adapter, assume
battery voltage VBAT is initially below
VTRIKL, the trickle charge threshold
(Figure 8). Battery charging begins
at approximately 50mA, well below
the wall adapter current limit so
the voltage into the LTC4001 (VIN)
is the wall adapter’s rated output
voltage (VADAPTER). Battery voltage
rises eventually reaching VTRIKL. The
linear charger shuts off and the PWM
(high rate) charger turns on using
soft start. Battery charging current
rises during the soft-start cycle causing a corresponding increase in wall
adapter load current. When the wall
adapter reaches current limit, the wall
adapter output voltage collapses, and
VIN
VDD
Battery Temperature Sensing
By adding one resistor and one thermistor, battery temperature sensing may
be included. The LTC4001 is designed
for Vishay Dale’s “R-T Curve 2” therm-
LTC4001
CHRG
R1
390k R2
2k
µPROCESSOR
OUT
IN
Figure 6. A microprocessor interface
Table 1. CHRG Behavior
4
istors, but any thermistor with an
RCOLD-to-RHOT ratio of about 7 will also
work. If battery sensing is not needed,
the NTC pin is grounded.
Charger State
CHRG Behavior
Not charging
Open
High rate Charging and IBAT>IDET
Or cell conditioning
NMOS turned on pulling pin low
High Rate Charging and IBAT<IDET
30µA pull down current
NTC temperature fault while charging at
IBAT>IDET
Blink
20µs
667ms
Figure 7. CHRG temperature fault waveform
Linear Technology Magazine • June 2006
DESIGN FEATURES L
LINEAR CHARGING
VADAPTER
WALL ADAPTER IN CURRENT LIMIT
So how does LTC4001 dissipation
stack up against a 2A linear charger?
Most of a linear charger’s dissipation
occurs in the series pass element so
the dissipation is approximately equal
to the voltage drop in the pass element
times the charge current. Worst case
dissipation occurs at the lowest battery voltage where high rate charging
occurs (to make a valid comparison to
the LTC4001 this would be 2.85V). For
a 5.0V input, this translates into a dissipation of 4.3W! Higher input voltage
makes the situation even worse.
PWM
CHARGING
VBAT + VDROP
VIN
ILIMIT
IBAT
ITRICKLE
VTRIKL
VFLOAT
VBAT
Figure 8. Idealized charging behavior
Low Dissipation
Trickle charging uses a linear charger
but low charge current produces low
power dissipation, typically 256mW
(VIN = 5V, VBAT = 0). High rate charging
uses a high efficiency buck switcher
and total charger dissipation is approximately 1.2W at 2A (Figure 9).
High rate charging with a current
limited wall adapter produces even
lower charger dissipation (537mW at
VBAT = 4.2V with a 2A current limited
wall adapter) because there is very little
voltage drop for the battery charging
path inside the LTC4001.
1.25
TOTAL APPLICATION CIRCUIT POWER
DISSIPATION (W)
the LTC4001 PWM charger duty cycle
ramps up to 100% (the top-side PMOS
switch in the LTC4001 buck regulator
stays on continuously.) As the battery
voltage approaches VFLOAT, the float
voltage error amplifier commands the
PWM charger to deliver less than ILIMIT.
The wall adapter exits current limit
and VIN jumps back up to VADAPTER.
Battery charging current continues
to drop as VBAT rises, dropping to zero
at VFLOAT.
Because the voltage drop in the
LTC4001 is very low when charge
current is highest, power dissipation
is also very low.
R12
27k
Q2
TP0610
D1
GRN
C/X
D2
AMB
CHRG
R1
1k
Q3
2N7002
R2
1k
A full featured battery charger is shown
in Figure 10. It includes a three hour
timer, battery temperature monitoring, programmable charge and IDET
currents, remote sensing, and status
lights. A fault light has been included
that indicates when a shorted battery
is detected or when the battery is out
of normal temperature range.
VIN = 5V
VBAT = 4V
1.00
Conclusion
The LTC4001 sets a new standard for
small, low parts count, full-featured,
high efficiency Li-Ion battery chargers.
Low power dissipation makes continuous 2A battery charging practical,
cutting dissipation to approximately
one fifth the dissipation of a straight
linear charger. L
0.75
0.50
0.25
0
500
1500
1000
2000
IBAT (mA)
Figure 9. High rate charger power dissipation
VIN
2A
CURRENT LIMITED
WALL ADAPTER
4.5V TO 5.5V
Q1
2N3906
A Charger with All
the Bells and Whistles
R8
10k
L1
1.5µH 2A
SW
VINSENSE
PVIN
C1
10µF
10V
SENSE
BATSENS
BAT
10µF
10V
PGND
+
2Ahr
4.2V
Li-Ion
LTC4001
CHRG
GNDSENS
EN
GND
NTC
FAULT
EXT
NTC
PROG
R9
1.33k
R10
10k
AT 25°C
D3
RED
FAULT
R3
1k
R4
2A
549Ω
C3
0.22µF
R5
1A
1.10k
TIMER SS
IDET
C4
0.1µF
R7
0.1A
1.10k
R6
0.2A
549Ω
L1: VISHAY DALE IHLP-2525AH-01
R10: NTC VISHAY DALE NTHS0603N02N1002J
Figure 10. A full featured battery charger
Linear Technology Magazine • June 2006
5
L DESIGN FEATURES
650MHz Selectable-Gain
Amplifier/Differential ADC Driver
Has Small Form but Many Functions
by Cheng-Wei Pei
Introduction
Operational amplifiers have always
been an important part of an analog
designer’s bag of tools. Even the most
basic tools can be improved, and some
recent advancements have increased
the utility of the workhorse op amp. For
instance, the advent of dual and quad
op amp packages allowed engineers
to produce a variety of applications
from a single device. The new LT6411
selectable-gain differential amplifier/ADC driver makes a good thing
better by adding internal gain and
feedback resistors and an easy-to-use
flow-through pin layout.
The LT6411 can produce gains of
1, –1, and 2 with no external components. The dual amplifiers inside
the LT6411 allow for easy singleended-to-differential conversion for
driving high-speed analog-to-digital
converters (ADC). The wide bandwidth
(650MHz), low distortion (–77dBc
harmonic distortion at 30MHz) and
high slew rate (3300V/µs) preserve
signal fidelity even at high frequencies,
while the low supply current (16mA
total) enables the LT6411 to be used
in power-critical high-speed signal
chain applications. Form factor is also
not an issue—all of these features fit
When blazing fast speed,
low power, or the flexibility
of selectable gains is
necessary, a circuit designer
need only reach into his bag
of tools and pull out
the LT6411.
into the tiny 3mm × 3mm 16-pin QFN
package. When blazing fast speed, low
power, or the flexibility of selectable
gains is necessary, a circuit designer
need only reach into his bag of tools
and pull out the LT6411.
Internal Topology and
Gain Selection
The LT6411 contains two internal
current-feedback amplifiers with
matched feedback and gain resistors.
The integrated 370Ω resistors take the
guesswork out of selecting the optimal
feedback resistor for good AC response.
Another common source of frustration
with current-feedback amplifiers is
maintaining a tight PC board layout
to prevent excessive capacitance at
the inverting input node. This node is
+V
IN +
IN –
LT6411
+
–
–
+
Single-Supply Operation
and Level-Shifting
The LT6411 operates on a wide supply voltage range, from 4.5V–12.6V.
Figures 1–3 show the part with dual
supplies. However, the LT6411 performance remains excellent with a
+V
IN +
OUT –
–V
+V
LT6411
OUT +
Figure 1. Standard non-inverting gain of 2
configuration, shown with split supplies.
6
internal to the LT6411, so the layout
of the circuit board is nearly optimal
from the get-go.
Selecting between the basic gain
configurations of the LT6411 is a simple matter of pin strapping. Figure 1
shows the standard non-inverting gain
of 2 configuration, with differential
input and output. Figure 2 shows the
non-inverting gain of 1 configuration,
and Figure 3 shows an inverting gain of
1 configuration. Note that in Figure 2,
the non-inverting inputs are tied together with the gain resistor. In theory,
the gain resistor pin could be left floating, but in practice, the parasitic pin
and pad capacitances cause the gain
to peak up to 6dB at high frequencies, which causes excessive ringing
in the transient response. At 650MHz
(the bandwidth of the LT6411), 2pF of
parasitic capacitance has a reactance
of just 122Ω.
LT6411
+
–
OUT +
–
+
OUT –
IN –
IN +
–V
Figure 2. Standard non-inverting gain of 1
configuration, shown with dual supplies. The
gain resistors, shown here tied to the inputs,
should not be left floating (see text).
+
–
IN –
–
+
OUT –
OUT +
–V
Figure 3. Standard inverting gain of 1
configuration, shown with dual supplies.
Linear Technology Magazine • June 2006
DESIGN FEATURES L
single supply, especially important for
many common applications such as
high speed ADC driving, where only a
single supply is available. If the input
signals into the LT6411 are already
DC level-shifted above ground so that
the input and output common-mode
ranges are met (the input operates to
within 1V of the supplies, the output
with a 1k load swings to within 1.3V
of the supplies), no additional work
needs to be done. However, if the input signal is AC-coupled or centered
around ground, then level-shifting is
necessary. This section presents some
of the methods to level-shift the input
and output voltages of the LT6411.
Figure 4 shows a center-tapped
transformer providing the DC voltages
necessary for single-supply operation.
The input signal (shown single-ended,
but can also be differential by driving
the other end of the primary) is provided
through the transformer primary, and
the transformer secondary presents a
balanced signal to the LT6411. If the
input signal is coming from a 50Ω
signal source, the two 24.9Ω resistors
provide the appropriate termination.
Note that the DC voltage provided
at the center tap of the transformer
is VO(DC)/2, due to the fact that the
LT6411 has a non-inverting gain of
2. Alternatively, the two gain resistor
pins could be AC-coupled to ground
through a capacitor, and the DC gain
of the LT6411 would be unity. The
total differential gain in this configuration is 2; if a differential gain of 1
is desired, simply tie the gain resistor
pins to the corresponding non-inverting input pins.
For a single-ended input, Figure 5
shows a simple AC-coupled method of
providing the correct input and output
DC levels for the LT6411. The input is
AC-coupled through a large capacitor
(typically 0.1µF or larger), and the total
differential gain at the output is +2.
The DC voltage is provided through
the second non-inverting input.
In the case of a differential inputs,
Figure 6 shows a similar configuration,
where the two inputs are AC-coupled
to the LT6411, and the DC level is
provided through two resistors. The
choice of size for the resistance R is
Linear Technology Magazine • June 2006
V+
LT6411
MIA-COM
ETC1-IT
1:1
IN
+
–
24.9Ω
• •
OUT +
VO,DC
VOLTAGE GAIN = 2
24.9Ω
–
+
OUT –
VO,DC
VO,DC
2
Figure 4. DC level-shifting through the center tap of a transformer for single-supply operation.
The DC voltage source at the center tap should serve as a low-impedance AC ground. The two
24.9Ω resistors provide a 50Ω termination, if necessary.
V+
IN
CLARGE
LT6411
+
–
fCO ≈
OUT +
VO,DC
1
2π • 370Ω • CLARGE
–
+
VO,DC
OUT –
VO,DC
Figure 5. A simple AC-coupled method of providing DC level shifting. The DC voltage source,
which does not need to have a low impedance, is provided at the non-inverting input of the
second amplifier. The AC input impedance of this circuit is 370Ω.
+V
IN +
LT6411
CLARGE
+
–
fCO ≈
1
2πCLARGE • R
IN –
OUT +
VO,DC
R
–
+
CLARGE
OUT –
VO,DC
R
VO,DC
2
Figure 6. A DC voltage source and two resistors sets the DC input and output level of the LT6411.
The two resistors must be large enough not to overload the inputs.
a trade-off between loading the differential inputs (for smaller values of
R) and increasing voltage offset and
noise (for larger values of R, due to the
input bias currents and current noise
in the non-inverting inputs). Values of
R up to 10k work well in practice. The
DC and AC inputs both have a gain
of 2 in this configuration. For a gain
of 1, simply tie the gain resistor pins
to the corresponding non-inverting
input pins.
Figures 4–6 all have a lower frequency limitation, defined by either
the transformer’s magnetizing inductance or the size of the AC-coupling
capacitor. What if voltage level-shifting and response down to zero Hertz
7
L DESIGN FEATURES
V+
IN +
VI,DC
RB
+V
RB
LT6411
+
–
VO,DC
IN –
VI,DC
RA
V+
–
+
RA
VO,DC
OUT +
VO,DC
GAIN DIFF = 2 •
VO,DC
VO,DC =
OUT –
RB
RA + RB
(V+ • RA ) + (VI,DC • RB)
R A + RB
VO,DC
Figure 7. Two identical resistive dividers (RA and RB) shift the DC level of the input (and output)
to within the limits of the LT6411. VO,DC must have a low impedance at the frequencies of
interest, and must be capable of sourcing and sinking currents through the internal resistors.
(DC) is necessary? Figure 7 shows one
method of level-shifting and maintaining signal response down to DC.
The RA-RB resistive dividers set the
input common-mode voltage for the
LT6411. Choose RA and RB so that the
common-mode voltage at the input
of the LT6411 is the same as VO(DC),
the desired output common-mode
voltage. The gain of the LT6411 with
the resistive divider is 2 • RB/(RA +
RB), and this circuit has a response
to DC. If the input DC common-mode
level (VI, DC) is greater than the VO,DC,
then RB should be attached to ground
instead of the positive supply.
(that is, VDC in the Figure 11 must be
appropriate for both the amplifier and
the ADC), or that one of the techniques
from the previous section must be used
to level-shift the input.
If dual supplies are used for the
LT6411, then interfacing with a
single-supply ADC might require
AC-coupling at the output, as shown
in Figure 9. The DC level of the ADC
input is established by VDC and two
499Ω resistors.
Figure 8’s circuit, though simple,
has an important drawback: all of the
wideband noise of the amplifier couples
into the ADC input, and thus degrades
the signal-to-noise ratio (SNR) of the
signal. In most cases, the input signal
is limited to some frequency band less
than the DC–650MHz bandwidth of the
LT6411; thus, any extra bandwidth
beyond that introduces unnecessary
noise. The simplest approach to fixing
this problem aptpears in Figure 10. A
single shunt capacitor creates an RC
lowpass filter that limits the noise
bandwidth of the amplifier output,
improving the SNR.
For an even sharper cutoff lowpass
filter, Figure 11 shows a more involved
approach. The inductors and capacitor
create a second-order lowpass filter,
with R1 ensuring that the frequency
peaking is not excessive. R2’s primary
function is to ensure that the ADC
+V
IN +
LT6411
+
–
VDC
24.9Ω
High Speed ADC Driving
Modern high resolution, high speed
ADCs typically feature differential inputs with capacitive sample-and-hold
circuits. Driving these inputs requires
an amplifier with high bandwidth, fast
settling times, good transient response
and good distortion performance at the
frequencies of interest. The figures in
this section show various configurations designed to get the maximum
performance out of the LT6411.
The LT6411 is optimized for driving
12-bit and 14-bit high-speed converters such as the 14-bit, 80Msps
LTC2249. For many applications,
the only additional recommended
interfacing components would be
small series resistors, to help isolate
the LT6411 from the ADC’s capacitive
input. Figure 8 shows the simplest
configuration for the LT6411 driving
the ADC. The single supply operation
means that either the inputs must have
an appropriate DC common-mode level
8
ADC
IN –
–
+
VDC
24.9Ω
Figure 8. LT6411 shown interfacing to an ADC. A small series resistance is recommended to
isolate the ADC’s capacitive input.
+V
IN +
fCO ≈
LT6411
+
–
0V
CLARGE
1
2π • 499Ω • CLARGE
499Ω
IN –
0V
–
+
ADC
CLARGE
499Ω
–V
VDC
Figure 9. Level-shifting the LT6411 output to within the input common-mode range of the singlesupply ADC. The resistors used must be large enough not to excessively load the outputs of the
LT6411. Some ADCs have a VCM output that can be used for VDC.
Linear Technology Magazine • June 2006
DESIGN FEATURES L
Conclusion
The LT6411 selectable-gain amplifier/ADC driver is extremely flexible,
featuring a multitude of possible configurations with a minimal number of
external components. The LT6411 is a
dual op amp, a differential ADC driver,
and a selectable-gain amplifier all in
one. In most basic dual op amp functions, all that is required is a few power
supply bypass capacitors to get excellent AC performance to over 600MHz.
In addition, the LT6411 comes in the
tiny 3mm × 3mm 16-lead QFN package
and consumes only 16mA total. The
LT6411 also has a shutdown feature
that reduces the power supply current
to 700µA total. With this unique set
of features, the LT6411 can provide
myriad functions without breaking
the budget of size or power. L
for
the latest information
on LTC products,
visit
www.linear.com
Linear Technology Magazine • June 2006
+V
LT6411
+
–
R
C
–
+
ADC
R
fCUTOFF = (4πRC)–1
Figure 10. An RC lowpass filter is shown at the output of the LT6411. Reducing the bandwidth at
the output also reduces the amount of wideband noise seen by the ADC.
V+
R2
LT6411
+
–
L
R1
C
R2
–
+
ADC
L
R1
fCUTOFF > 2π 2LC
–1
R1 + R2 ≤ 150Ω
Figure 11. A second-order LC lowpass filter, offering a flatter passband and sharper stopband
rolloff than an RC filter. The series resistor R1 controls peaking near the cutoff frequency,
and the parallel resistor R2 ensures that the ADC sees a low source impedance at very high
frequencies.
V+
LT6411
+
–
R1
54.9Ω
R1
54.9Ω
6
L
390nH
R2
80.6Ω
–
+
9
R2
80.6Ω
3
C
15pF
GAIN (dB)
inputs do not see too high of a source
impedance. Figure 12 shows some
sample values, configured for a cutoff
of around 50MHz and almost no gain
peaking near the cutoff frequency.
Simulating the filter is a good way to
determine optimal component values,
especially when taking into account
the series resistance of the inductor
and component tolerances.
Some high frequency applications
contain very narrow-band signals,
where a bandpass filter would provide the best noise limiting, and thus
the highest SNR. Figure 13 shows a
simple RLC bandpass filter. The value
of R determines the quality factor (Q)
of the filter—the larger the resistor
value, the more narrow-band the
filter. This comes at the cost of more
pass-band loss (depending on the
parasitic components of the inductor
and capacitor) and higher sensitivity to
component tolerances and variations.
As the bandpass filter gets narrower,
a small shift in center frequency can
significantly attenuate the desired
output signal.
LTC2249
0
–3
–6
L
390nH
–9
–12
1
10
100
FREQUENCY (MHz)
1000
Figure 12. The circuit of Figure 11, configured for a cutoff frequency of around 50MHz.
V+
LT6411
+
–
R
L
–
+
C
ADC
R
fCENTER = (2π√LC)–1
Figure 13. For narrow-band applications, an LC bandpass filter does an excellent job limiting the
noise at the ADC input while maintaining a low component count. The series resistor affects the
width of the pass-band and the pass-band attenuation.
9
L DESIGN FEATURES
Dual Step-Up Converter Drives White
LEDs with 1000:1 PWM Dimming
by Keith Szolusha
Introduction
Notebook computers, large-screen
handheld PDAs, dashboard displays,
and automotive and avionic in-cabin
entertainment LCD panels are illuminated with strings of high power white
LEDs. White LEDs are the preferred
over other backlight technologies because they provide true white light at
a high enough intensity for daylight
viewing, and enough dimming capability for nighttime use. LEDs also offer
relatively long life spans and a lack
of hazardous materials. LED strings
lining the edges of these LCD panels
provide uniform brightness when
driven with a constant current.
The maximum switch voltage of an
LED driver limits the number of LEDs
that it can drive in series. It may seem
that paralleling LEDs is a good way
to increase the capacity of a driver
IC, but parallel LEDs must be wellmatched in forward voltage; otherwise
un-matched LED strings cause uneven
currents and thus uneven brightness.
LEDs can be specially sorted (binned)
for matching characteristics, but this
increases cost.
VIN
4V TO 16V
10µH
ZLLS400 A915AY-100M
10µH
A915AY-100M
COUT1
2.2µF
35V
100mA
8 TO 10 WHITE
LEDs <34V
SW1
CTRL1
BRIGHTNESS
ADJUST
SHDN
OVP2
CTRL1
CTRL2
PWM1
100k
COUT2
2.2µF
35V
2.8k
CTRL2
BRIGHTNESS
ADJUST
0.1µF
REF
FDN5630
PWM2
FB1
VC1 GND RT
PWM1
100mA
8 TO 10 WHITE
LEDs <34V
SW2
VIN
SHDN LT3486EFE
FDN5630
RSENSE1
2Ω
1%
Dual LED String
Step-Up Driver
ZLLS400
CIN
4.7µF
25V
OVP1
FB2
VC2
63.4k
PWM2
100k
RSENSE2
2Ω
1%
2.8k
PWM INPUT 4700pF
100Hz
1000:1 DIM RATIO
PWM INPUT
100Hz
1000:1 DIM RATIO
4700pF
Figure 1. LED driver uses 4V–16V input to drive two strings of eight-to-ten
100mA LEDs (less than 34V total in string) with 1000:1 PWM dimming
90
100
85
EFFICIENCY (%)
50
40
30
20
10 LT3486
2× 10× WHITE 100mA LEDs
0
8 9 10 11 12 13 14 15 16 17 18
INPUT VOLTAGE (V)
100
80
80
LED CURRENT
75
60
70
40
65
60
VIN = 12V
2× 8× LEDs
0
20
40
60
80
PWM DUTY CYCLE (%)
20
0
100
Figure 2. Efficiency of the circuit in Figure 1 and efficiency as a function of PWM duty cycle
10
LED CURRENT (mA)
60
EFFICIENCY (%)
80
70
120
EFFICIENCY
90
A better solution is a dual channel LED driver to drive two strings of
LEDs. This saves the space and cost of
duplicating components, such as the
driver IC and input capacitors. Each
string is driven with the same regulated constant current, thus providing
uniform brightness.
One IC that has these features is
the LT3486 dual LED string driver,
which has two 1.3A channels with
high PWM dimming capability in a
small 5mm × 3mm DFN package.
Since both channels’ power switches
are included in the IC, the circuit is
simple and small.
The LT3486 is a dual step-up LED
driver. Each channel has an efficient,
low side 1.3A npn power switch with
low VCE(sat) of 300mV (at 750mA
switch current). The IC is designed
to drive a string of LEDs from a wide
input voltage range. Each LED string
total voltage can be as high as 38V in a
typical application, but may be limited
to 34V if the overvoltage protection
(OVP) pin is used to protect the switch
when the LED string is open.
Figure 1 demonstrates the LT3486
as a dual LED string step-up converter
driving a total of 16–20 white LEDs at
100mA from a 4V–16V input voltage
range source. The total voltage of the
LEDs cannot exceed 34V. The circuit
is kept small and simple with the
single ceramic input capacitor and two
small ceramic output capacitors. With
a high 800kHz switching frequency,
the inductors and capacitors can be
small in size while the efficiency of
the circuit remains high, as shown
in Figure 2. As PWM duty cycle is
decreased from 100%, the circuit efficiency drops slightly, but remains
high during the PWM on-time. Not
only is the operating efficiency high,
but the converter shutdown current
Linear Technology Magazine • June 2006
DESIGN FEATURES L
FEEDBACK VOLTAGE (mV)
250
VIN = 3.6V
TA = 25°C
ILED
200mA/DIV
200
IL
500mA/DIV
150
PWM
5V/DIV
100
VIN = 12V
0.2ms/DIV
8/8 LEDs
PWM FREQ = 1kHz
50
Figure 4. PWM dimming waveforms
0
0
1
0.5
1.5
CONTROL VOLTAGE (V)
function of CTRL pin voltage. The low
200mV FB pin (and current sense
voltage) accuracy is typically 3% at
full current with the CTRL pin pulled
high (above 1.5V) but as the CTRL pin
voltage is lowered to 150mV, the FB
pin voltage is also reduced to about
40mV. Below this 5:1 dimming ratio,
the LEDs are turned off as the CTRL
pin voltage is pulled below 75mV.
Another method of reducing the
brightness of the LEDs is digital PWM
dimming. The PWM MOSFET in series
with the LEDs creates the waveform
shown in Figure 4 when the string of
LEDs is PWM’d at 100mA constant
current. During PWM on-time, the
current is a well-regulated 100mA.
During PWM off-time, the current is
zero. Because the current is either
100mA or zero, the LED color is preserved as if the LED were driven by
a constant 100mA current. Dimming
is simply a function of the average,
instead of instantaneous, current. The
advanced PWM function in the LT3486
2
Figure 3. FB pin voltage vs CTRL pin voltage
consumption is less than 1µA (typically 100nA), merely sipping from the
battery when the IC is off.
1000:1 PWM Dimming and
10:1 Brightness Control
As shown in Figure 1, LED brightness
can be controlled on the LT3486 with
an analog voltage input to the CTRL
pin or a digital PWM signal to the
gates of the PWM dimming MOSFET
and the PWM pin. Analog brightness
control reduces the LED current from
100mA to a lower value by reducing
the internal sense resistor voltage.
Although this is a simple way to decrease the brightness of the LED, the
accuracy of the LED current control
is reduced and the chromaticity of the
LED changes at lower currents. The
graph in Figure 3 displays the LT3486
typical FB pin voltage dropping as a
CIN
10µF
5V
D1
D2
L1
10µH
COUT1
2.2µF
SW1
25mA
L2
10µH
VIN
COUT2
2.2µF
SW2
OVP1
OVP2
CTRL1
CTRL2
OFF ON
SHDN
REF
PWM1
LT3486
1.25V
REF
PWM2
FB1
CREF
0.1µF
FB2
RT
VC1
2.8k
8.06Ω
25mA
4.7nF
VC2
RT
Doubler Delivers Greater
than 34V to LED Strings
GPS navigation and in-cabin entertainment displays are increasingly
popular in mainstream consumer
vehicles. The advantage of using two
LED drivers each with 8-LED strings,
instead of a single 16-LED string, is
that the maximum switch voltage
remains that of a single 8-LED string
(less than 34V total string voltage at
100mA). Even so, LCD panel screen
sizes are pushing beyond the standard
6" and 7", requiring more LEDs and
string voltages above 34V.
The circuit in Figure 7 uses a charge
pump voltage doubler to drive two
strings of LEDs to voltages as high
continued on page 44
2.8k
4.7nF
8.06Ω
CIN: 10V, X7R
COUT1, COUT2: 35V, X5R
D1, D2: ZETEX ZHCS400
L1, L2: TOKO D53LC TYPE A
Figure 5. LED driver uses 5V input to drive two strings of eight 25mA LEDs
(less than 34V total in either string) with 5:1 brightness control
Linear Technology Magazine • June 2006
is particularly fast in returning the
LED to its programmed LED current.
Its short minimum dimming on-time
(10µs on-time) allows a 1000:1 digital
PWM dimming ratio with 100Hz PWM
frequency—fast enough to avoid visible
flicker. For instance, a combination of
two LT3486s driving four LED strings
(R-G-G-B) in a top-end display provides
1000:1 dimming while maintaining the
true-color of the display even during
very dim nighttime operation.
When a PWM signal is used for
brightness control, but less than a
5:1 dimming range is needed and the
chromaticity of the LEDs is not especially important, the PWM signal can
be fed into an RC filter such as the
one in Figure 6. This turns the PWM
input into an analog CTRL pin voltage
controlling the LED current directly,
eliminating the need for the PWM
dimming MOSFETs. The 5V, 16-LED
converter in Figure 5 can deliver up
to a 5:1 analog dimming range at the
CTRL pins with such a filter without
the need for the two additional PWM
dimming MOSFETs. In this case, the
LT3486 PWM pins are tied high to the
1.25V REF pin.
PWM
10kHz TYP
LT3486
R1
10k
C1
1µF
CTRL1,2
Figure 6. Achieving 5:1 brightness
control with a filtered PWM signal
11
L DESIGN FEATURES
Hot Swap Controller Monitors and
Reports Power Supply Status by Josh Simonson
Z1
SA14A
CONNECTOR 1
High availability systems are designed
to achieve an ideal of zero down time.
To approach this goal, the system
needs to be able to operate during routine maintenance and upgrades, which
often involves cards being inserted and
removed from a live backplane. These
systems must also be designed for
failsafe operation by isolating faulty
boards before they cause backplane
disturbances.
Hot swapping requires a power
switch to initially isolate the board,
and a controller to turn on the switch
slowly to minimize backplane disturbances. Since the Hot Swap controller
monitors card voltage and current, it
is an obvious place to integrate higher
level monitoring with a data converter.
This provides detailed information
about the health of the power path
and the power consumption of downstream circuits. Such information
RS
0.005Ω
VIN
12V
CONNECTOR 2
Introduction
CF
0.1µF
R1
34k
1%
Q1
FDC653N
R5
10Ω
R2
1.02k
1%
UV VDD SENSE+ SENSE– GATE SOURCE
FB
OV
ON
ADIN
SDAI
GPIO
LTC4215UF
SDAO
EN
SCL
SS
ALERT
R3
3.4k
1%
SDA
SCL
ALERT
TIMER INTVCC ADR0 ADR1 ADR2 GND
CTIMER
0.68µF
GND
C3
0.1µF
R7
30.1k
1%
R8
3.57k
1%
+
VOUT
12V
CL
330µF
R4
24k
CSS
6.8nF
NC
BACKPLANE PLUG-IN
CARD
Figure 1. In a typical application the LTC4215 uses an external N-channel pass transistor to
isolate the hot swapped board from the backplane when it is first inserted. After a debounce time
the controller can begin to apply power to the board or wait for a turn-on command from a host
processor. Power is ramped gradually to minimize any backplane disturbance. After the power-up
process is complete, the LTC4215 continues to monitor for faults in the power path.
can be used to monitor performance
over time and identify boards that are
drifting towards failure or marginal
performance.
The LTC4215 combines a robust Hot
Swap circuit with an I2C interface and
data converter to allow power monitoring as well as hot-plug functionality
Table 1. A few of the LTC4215’s many features
Feature
Benefits
Wide Input Voltage Range: Operates from
inputs of 2.9V to 15V, with 24V absolute
maximum
❏ Suitable for 3.3V, 5V and 12V systems
8-bit ADC: ADC monitors current, output
voltage and external pin voltage and
measures off-state current in the FET to
determine FET failures
❏ Increases reliability
❏ Simplifies design because part functions on a semi-regulated supply
❏ Large overvoltage transient range eases design tolerances for transient protection
❏ Board power information provides an early warning of board failure
❏ Verify board is staying within its allotted power
❏ Allows integrity check of redundant supply paths
❏ Allows active power management to safely maximize power utilization within the
chassis cooling constraints
I2C/SMBus: Communicates as a read-write ❏ Improves integration with the host system. Interface allows the host to configure
the part, determine which faults are present or have occurred, and read back ADC
slave device using a 2-wire serial interface
measurements
Fast Short Circuit Response: Fast (<1µs)
current limit response to shorts
❏ Protects connector from overcurrent
Alerts Host after Faults: When configured
(using I2C), faults activate an active pulldown on the ALERT pin
❏ Interrupting the host for immediate fault servicing limits system damage
12
❏ Limits the disturbance to the input supply from a short circuit
❏ Reduces the bus traffic for polling
Linear Technology Magazine • June 2006
DESIGN FEATURES L
and fault isolation (see Table 1). In a
typical application the LTC4215 uses
an external N-channel pass transistor to isolate the hot swapped board
from the backplane when it is first
inserted (Figure 1). After a debounce
time the controller can begin to apply
power to the board or wait for a turnon command from a host processor.
Power is ramped gradually to minimize
any backplane disturbance. After the
power-up process is complete, the
LTC4215 continues to monitor for
faults in the power path.
The LTC4215 provides the means
for quantitatively measuring the board
current and voltages with an onboard
ADC and multiplexer. It reports this
information using the I2C serial communication bus when polled by a host
processor. The device interrupts the
host for specific fault conditions, if
configured to do so.
The LTC4215 works in applications
from 12V (with transients to 24V) down
to 3.3V where the operating input
voltage could drop to 2.9V. Functionally, the LTC4215 is very similar to
the LTC4260 (Linear Technology, Nov.
2004) operating in a lower voltage
range. Table 2 compares the major
features of the LTC4215 and LTC4260.
Special attention should be paid to the
power-up sequence because of the
added soft start pin and some changes
in the function of the TIMER pin relative to the LTC4260. Since both parts
can be used for 12V systems, Table
2 may be used to select the part with
optimal set of features for a specific
12V application.
An N-channel pass transistor, Q1,
controls the application of power to the
board as in Figure 1. A series sense
resistor, RS, allows the LTC4215 to
measure the current in the power-
Table 2. Comparison of the LTC4260 and LTC4215
Feature
LTC4215
LTC4260
VDD Abs Max
24V
100V
VDD Min
2.9V
8.5V
Recommend TransZorb for 12V
Yes
No
Current Limit/Circuit Breaker
25mV
50mV
Circuit Breaker Precision
10%
10%
Current Limit Precision (FB = 0)
35%
25%
Current Limit vs Circuit Breaker
VTH
75mV, 25mV
Both 50mV
Current Limit Foldback
Only during startup
Always
Gate RC Network
Optional
Required
Soft Start
Yes, Required
No
Timer Pin
Optional
Required
OC Timer
20µs
External/Adjustable
∆VGATE at 12V (0µA)
6V
8.5V
Built In Overvoltage Threshold
15.6V
None
ADC Source LSB
60mV
400mV
ADC VSENSE LSB
151µV
300µV
ADC ADIN LSB
4.85mV
10mV
Internally Generated VCC
3.1V
5.5V
Package
4mm × 5mm QFN
5mm × 5mm QFN
Linear Technology Magazine • June 2006
path. Resistor R5 suppresses self
oscillations in Q1. Resistors R1–R3
select the undervoltage (UV) and
overvoltage (OV) thresholds. Capacitor
CF allows these thresholds to be filtered
as needed. R7 and R8 select the powergood threshold and set the foldback
current limit level. Capacitor CSS sets
a maximum slew rate to control the
inrush current and CTIMER is used to
set the startup time. C3 is used to
bypass the internal core voltage.
Typically, the pins on the connector are staggered so that bulk power
is applied first with the longest pins,
followed by communication lines on
medium length pins, and last, Hot
Swap control lines such as the supply
for the UV, OV, or EN pins. The UV,
OV and EN pins must be in the correct
state for a programmable debounce
period of 100ms before Q1 is allowed to
turn on. At this point the ON pin turns
the part on immediately if it is high,
or holds the part off if it is low. When
the ON pin is held low, Q1 is turned
on through the I2C bus by writing to
the ON bit in the control register.
Measure Real-Time Board
Power with Integrated ADC
Monitoring the supply voltage and
current in real-time is a useful way
of tracking the health of the power
path. New data can be compared with
historical data for the same card to
detect changes in power consumption
that could indicate that the card is
behaving abnormally. An abnormal
card can be shut down and flagged for
service, perhaps before a more severe
fault or system malfunction occurs.
The LTC4215 includes an 8-bit data
converter that continuously monitors three voltages: the ADIN pin, the
SOURCE pin and the current sense
voltage between the SENSE+ (VDD)
and SENSE– pins. The ADIN pin is
an uncommitted ADC input which
allows the user to monitor any available voltage.
The ADIN pin is monitored with
a 1.235V full scale. The ADIN pin is
connected directly to a data converter
input without any signal scaling. The
SOURCE pin uses a 1:12.5 divider
at the input which gives a 15.4V full
13
L DESIGN FEATURES
scale. The SENSE voltage amplifier
has a voltage gain of 32, which results
in a 38.4mV full scale. The converter
uses a sophisticated oversampling
and offset cancellation method that
preserves the full 8-bit dynamic range
on the SENSE channel.
If the data converter reads more
than 1mV on the VDD-SENSE channel
while the external switch is turned off,
the LTC4215 generates a FET-SHORT
fault to indicate that the switch may be
damaged. The presence of this condition is indicated in STATUS register
bit C5 and logged to FAULT register
bit D5. The LTC4215 takes no action
in this condition other than logging
the fault and generating an alert if
configured to do so.
The results from each conversion
are stored in three ADC registers
(see Table 3) and updated 10 times a
second. Setting the test mode control
register bit halts the data converter so
that the registers can be written to and
read from for software testing.
Versatile Inrush
Current Control
Once the inputs to the LTC4215 reach
the correct values for the part to turn
the external switch on and an internal
100ms debounce timer has expired,
the LTC4215 turns on. The startup
time is determined by the capacitor on
the TIMER pin, or 100ms if the TIMER
pin is tied to VCC. During this time the
circuit breaker is disabled to prevent
an overcurrent fault from occurring,
the power-good signal from the GPIO
pin is also disabled to prevent turning
on a load before the current limit has
reached the full value via the Soft Start
and Foldback pins. The inrush current
slew rate (dI/dt) is limited via the SS
pin. The inrush current is also folded
back from 25mV to 10mV via the FB
pin. An optional RC network on the
external MOSFET gate can be used
to set the inrush current below the
foldback level by setting the maximum
slope of the output voltage. The various
inrush current profiles obtainable by
these three methods are detailed in
Figures 2 thru 6, which show a 12V
system with a 25mΩ sense resistor,
14
Table 3. LTC4215 registers
Register
Description
CONTROL
Register turns-on or turns-off the pass transistor and controls whether
the part will auto-retry or latchoff after a fault. It also configures the
behavior of the GPIO pin.
ALERT
Alert register enables which faults interrupt the host using the ALERT
pin. At power-up the default is to not alert on faults.
STATUS
Status register provides pass transistor (on/off), EN (high/low) and GPIO
(high/low) conditions. It also lists five fault present conditions.
FAULT
Fault register logs overcurrent, overvoltage, undervoltage, power-bad,
FET short and EN changed state faults.
SENSE
ADC data for the VDD-SENSE voltage measurement.
SOURCE
ADC data for the SOURCE pin voltage measurement.
ADIN
ADC data for the ADIN pin voltage measurement.
or 1A current limit, starting up into a
470µF capacitive load.
At the end of the startup period
the current limit circuit is checked.
If the current limit is still regulating
the current, the LTC4215 determines
that the output failed to come up and
generates an overcurrent fault. If the
current limit circuit is not active then
the current limit threshold is moved
to 75mV, the power-good signal to the
GPIO pin is enabled and the 25mV
circuit breaker is armed.
The SS pin sets the current slew rate
limit at startup. It starts at ground,
which corresponds to a negative voltage on the sense resistor and results
in the MOSFET being turned off. A
current into the soft-start capacitor
produces a ramp that corresponds to
increasing VDD-SENSE voltage. When
the current limit circuit releases the
gate (when the commanded VDDSENSE voltage becomes positive) the
current from the SS pin is stopped
to wait for the GATE pin to rise and
start to turn on the MOSFET. Once the
current limit circuit begins to regulate
the VDD-SENSE voltage, the current
from the SS pin is resumed and the
ramp continues until it reaches the
foldback level. It is important that the
SS pin stop the ramp while the GATE
pin slews because the ramp would
otherwise continue and result in an
uncontrolled step in current once
the MOSFET threshold is reached.
An uncontrolled step may violate inrush specifications and cause supply
glitches on the backplane.
If the soft-start ramp reaches the
foldback level, the foldback circuit
stops the ramp, as shown in Figure 5.
The ramp is allowed to continue as
the voltage at the FB pin rises and
increases the foldback current limit,
still limited in slope and limited in
magnitude by foldback as well.
If an RC network is placed on the
GATE pin to manually set the inrush
current to a value below the foldback
level (Figure 4), the current limit circuit
will leave regulation when it is unable
to achieve the VDD-SENSE voltage
commanded by the SS and FB pins.
If the startup timer expires during
this inrush an overcurrent fault is not
generated because the current limit
is not active. The power-good output
for GPIO is allowed to relay the state
of the FB pin, and the circuit breaker
is armed. Either the output voltage
finishes rising and a power good is
asserted when the FB pin crosses it’s
1.235V threshold, or the current rises
to the circuit breaker threshold and the
part generates an overcurrent fault.
In the event there is an overcurrent
condition after startup, the current
limit circuit limits the VDD-SENSE
voltage to 75mV while the circuit
breaker waits for a 20µs timeout before
producing an overcurrent fault. After
any overcurrent fault, the part waits
Linear Technology Magazine • June 2006
DESIGN FEATURES L
VGATE
5V/DIV
VOUT
5V/DIV
VGATE
5V/DIV
VOUT
5V/DIV
IINRUSH
500mA/DIV
IINRUSH
200mA/DIV
2ms/DIV
5ms/DIV
Figure 2. Inrush current is limited by foldback. This allows the fastest
startup of the load, with the inrush only lasting 9ms with a peak
current of 1A.
Figure 3. Inrush current is limited by a 1µF SS capacitor. This
provides the fastest startup in a system with demanding inrush
current slew rate requirements. Inrush dI/dt is reduced to 12mA/ms.
VGATE
5V/DIV
VOUT
5V/DIV
VGATE
5V/DIV
VOUT
5V/DIV
IINRUSH
100mA/DIV
IINRUSH
500mA/DIV
10ms/DIV
2ms/DIV
Figure 4. Inrush current is limited by a 0.1µF GATE capacitor. This
minimizes the power in the switch, allowing the use of smaller
components at the cost of speed. The inrush current is only 100mA,
but startup takes 55ms.
Figure 5. Inrush current is limited by a 68nF SS capacitor and FB.
Soft start controls the inrush current slew rate while the current limit
is modulated by foldback. This allows the fastest startup while also
protecting the backplane from current surges.
VGATE
5V/DIV
VOUT
5V/DIV
IINRUSH
100mA/DIV
10ms/DIV
Figure 6. Inrush current with 1µF SS capacitor and 0.1µF GATE capacitor. Soft start limits the inrush current slew rate until the GATE capacitor
limits the inrush current by limiting the dV/dt at the output. This minimizes the power in the switch while protecting the backplane from inrush
current surges.
for a cool-down period of 50 times the
startup time before allowing the part
to restart by any means, including
auto-retry, I2C, or cycling the EN, UV
or ON pins.
Controlled Turn-Off
When the LTC4215 is turned off by
a fault or I2C transaction, the GATE
pin is pulled down with a 1mA current
source. Once the GATE pin is below the
SOURCE pin, a diode from SOURCE
to GATE turns on and the voltage at
the SOURCE pin is discharged by the
same 1mA current.
If there is a short that causes the
sense voltage to exceed 75mV, a 400mA
pull-down from GATE to SOURCE
removes the gate charge of the switch.
Once the sense voltage falls to 75mV,
the current limit regulates there for
Linear Technology Magazine • June 2006
20µs before turning the gate off with
the 1mA current source.
If there is significant inductance
between VDD (SENSE+) and bulk
capacitance, across a connector for
instance, it is possible that a short
circuit at the output with a very fast
rise time could cause the input voltage
to collapse while the current through
this inductance slews. In this case,
after 2µs, the VDD undervoltage lockout
circuit turns on and discharges the
GATE pin with the 400mA pull-down
to the SOURCE pin and quickly turns
the switch off.
Save Power with Precise
25mV Circuit Breaker
For supplies with lower voltages
and higher currents, a 50mV circuit
breaker threshold may result in too
much power dissipation in the sense
resistor, or cut excessively into the
input supply voltage tolerance of
downstream circuits. To reduce this
problem the LTC4215 has a precision
circuit breaker at 25mV with a low
10% tolerance. This allows the use
of smaller and less expensive sense
resistors with lower power ratings.
In systems where the circuit breaker
has only 20% accuracy the designer
must be able to safely provide 40%
more power than the card actually consumes to ensure that the slot
doesn’t suffer from heat and supply
limitations on the high side or produce
a fault in normal operation on the low
side. The precise 10% accuracy of the
LTC4215 cuts this guard-band in half
and safely allows the use of 20% more
continued on page 37
15
L DESIGN FEATURES
Efficient Buck-Boost Converter Ideal
for Power Saving Modes and Wide
by Kevin Ohlson
Input Voltage Ranges
Introduction
Portable handheld electronic tools,
gadgets, and toys are approaching
the multi-function equivalent of a
Swiss army knife. Previously separate
functions are now miniaturized and
combined into single pocket sized
package. Unless owned by a teenager
or a Hollywood agent, most gadgets
spend most of their time in a low power
state, waiting to come to life to make
a phone call, pulse a photoflash or
spin music and video to a tiny hard
disk drive. These devices are powered
from a variety of power sources, which
means at some point the input voltage is higher than or lower than the
3.3V or 3.6V voltage needed to power
the internal electronics. The DC/DC
converters in the latest portables must
be able to step up and step down
voltage, maintain very high efficiency
during idle and standby modes, and
respond quickly and efficiently during
peak power demands.
3
Figure 1. Lithium ion battery to
2.5V–5.25V converter in 1.4cm2
Linear Technology offers a family
of buck-boost converters capable of
supplying from 200mA to 2A with
excellent efficiency. The latest addition to this lineup is the LTC3532, a
300mA buck-boost converter, which
incorporates automatic Burst Mode
operation, adjustable switching frequency, and integrated soft-start. The
LTC3532 is ideal for miniature disk-
4
SW1
SW2
SW D
SW A
VOUT
6
SW B
GATE
DRIVERS
AND
ANTICROSS
CONDUCTION
2
SW C
PHASE
CONTROL
ERROR
AMP
BURST
–
VIN
Features
+
7
drive applications or any application
that requires high efficiency over a
wide range of output currents and
input voltages.
1.22V
FB
9
+
–
1A
10
VC
Figure 2. The LTC3532 four switch buck-boost converter uses a single inductor
and features peak current clamp and automatic Burst Mode control.
16
The input voltage range of a LTC3532based converter is 2.4V to 5.5V and
its output range can be programmed
from 2.4V to 5.25V, making it ideal
for devices operating from multiple
sources such as battery, USB, and
wall adapters. The LTC3532 is available in either an MS10 package, which
is pin compatible with the LTC3440
converter, or in an exposed pad 3mm
× 3mm DFN. With these tiny packages,
an entire converter can be squeezed
into the smallest spaces, as shown
in Figure 1.
Using a fixed frequency fourswitch architecture and a patented
control method, the converter needs
only a single inductor to regulate a
constant output voltage with input
voltages greater than or less than the
output. The four switch topology of
the LTC3532 (see the output stage
schematic in Figure 2) allows the
regulator to smoothly transition from
buck mode to buck-boost mode and
boost mode by correctly phasing the
four output switches (A, B, C, and D)
in response to the error amp output
voltage, VC. During buck mode, switch
D is on while switches A and B act like
a buck converter. At the other extreme,
boost mode, switch A is always on
while switches C and D implement a
synchronous boost converter. When
VIN and VOUT approach the same voltage, all four switches commutate with
the on time for each pair controlled
by the voltage at VC. The four switch
architecture inherently provides
output disconnect, which prevents
current flow between VIN and VOUT in
shutdown mode.
Linear Technology Magazine • June 2006
DESIGN FEATURES L
100
EFFICIENCY
100
80
10
70
POWER LOSS
1
60
3V
3.6V
4.2V
50
V
= 3.3V
40 OUT
0.1
1
Li-Ion to 3.3V Converter
Ideal for Miniature
Hard Disk Drives
IL
500mA/DIV
ILOAD
500mA/DIV
0.1
VOUT = 3.3V
0.01
1000
10
100
LOAD CURRENT (mA)
SHDN/SS pin reaches 1V, the internal
control voltage is clamped until the
pin rises to 2V.
VOUT
1V/DIV
POWER LOSS (mW)
EFFICIENCY (%)
90
1000
Figure 3. High efficiency is possible over a
wide range of load currents using automatic
Burst Mode control.
The capabilities of the four switch architecture is exploited in the LTC3532
when in Burst Mode operation as well.
An innovative (patent pending) Burst
Mode control circuit optimizes the firing of the four switches depending on
whether in buck mode, boost mode, or
buck-boost mode. Optimal switching
control and a low 35µA Burst Mode
quiescent current means the converter
increases the battery life of a system
by keeping efficiency above 80% in
Burst Mode operation at loads as low
as 300µA. Measured efficiency over a
load current ranging from 0.1mA to
500mA is shown in Figure 3. A resistor
and filter capacitor connected to the
burst pin set the level of load current
at which the converter automatically
switches between continuous and
Burst Mode operation.
Peak inductor current is limited
two ways. The first method monitors
current in switch A and sources a
10ms/DIV
Figure 4. A soft clamp peak current control
keeps the converter in continuous control
mode when peak inductor current is reached.
fraction of that current into the FB pin
when the peak current exceeds 1.1A.
This effectively lowers the VOUT set
point providing a closed loop method
of controlling the peak current. In
higher load and transient situations
a comparator opens switches A and
B, thus providing a hard peak current
limit of 1.3A. Figure 4 shows inductor
current and VOUT responding to increasing load. As the inductor current
reaches current limit, VOUT drops and
the control loop stays in continuous
operation. When VOUT is low either at
start-up or recovering from a short
circuit the current limit clamp level is
reduced by half providing a foldback
function.
Switching frequency may be programmed with an external resistor
to a frequency between 300kHz and
2MHz, which allows a trade off between component size and efficiency.
Soft start is performed by controlling
the slew on the SHDN/SS pin. Once
the converter is enabled, when the
A miniature, 1” or smaller disk drive
in standby may draw 40mA. In idle or
track seeking modes the drives current
increases to 150mA and when reading
or writing data the load might peak at
200mA to 300mA. Even during these
transitions, the supply should be well
regulated with very low ripple throughout the discharge cycle of the system
battery. Compared with a step-down
regulator in a 3.3V system a buckboost converter such as the LTC3532,
which maintains regulation even as the
battery drops below 3.3V, allows all the
energy in the battery to be used. The
converter keeps accurate regulation
during load transitions when battery
ESR may cause the input voltage to
drop below VOUT. A Li-ion to 3.3V application, which uses a tiny multi-layer
chip inductor, is shown in Figure 5.
Capable of load steps up to 400mA
with battery voltage as low as 3.0V,
the converter delivers efficiency greater
than 90% in continuous mode with
loads between 30mA and 200mA as
shown in Figure 6. Lower power burst
mode efficiency is greater than 80% at
sub 1mA loads. This circuit uses a soft
start capacitor connected to SHDN/
SS to limit inrush current. Transient
responses to load steps are shown in
D2
4.7µH
VOUT
3.3V
340k
D1
VIN
2.5V TO
4.2V
SW1
SW2
VIN
VOUT
SHDN/SS
412k
BURST
RT
LTC3532
FB
VC
12.1k
220pF
220pF
GND
22µF × 4
4.7µF
4.7nF SD
0.1µF
249k
86.6k
1k
200k
L1 = FDK M1PF2520D4R7
D1, D2 = MBRM110LT
Figure 5. A high efficiency converter using a tiny multi-layer inductor ideal for miniature HDD applications
Linear Technology Magazine • June 2006
17
L DESIGN FEATURES
200µs/DIV
100
95
EFFICIENCY (%)
90
the value of the resistor on the BURST
pin lowers the current at which burst
mode is entered. Figure 8 shows the
relationship between the burst pin resistor and the output current value at
which the transition between continuous and burst operation takes place.
If desired, the operating mode may be
forced by driving the BURST pin above
or below the thresholds.
VOUT
100mV/DIV
VIN = 4.2V
85
VIN = 3V
80
VOUT
500mV/DIV
75
70
BURST
5V/DIV
65
SHDN/SS
5V/DIV
60
55
VOUT = 3.3V
50
0.1
1
10
100
LOAD CURRENT (mA)
ILOAD
100mA/DIV
1000
Figure 6. Efficiency is over 85% for all power
saving modes of a typical 1” hard disk drive
Figure 7. VOUT response to load
transitions is well controlled.
1.12V the converter switches to continuous mode. As the load decreases
and the voltage on the burst pin drops
below 0.88V, the converter switches
back to Burst Mode operation. Raising
70
60
LOAD CURRENT (mA)
Figure 7. As the load is increased from
10mA to 50mA the transition from
burst mode to continuous operation
occurs. Steps from 150mA to 300mA
and back show VOUT transients have
peak amplitude of only 20mV.
An RC network on the BURST pin
enables automatic Burst Mode operation to maintain high efficiency at light
loads without external control. Burst
operation is controlled by the voltage
on the burst pin. During operation a
small fraction of the output current
passing through switch D is mirrored
out of the BURST pin. The mirrored
current produces a voltage across the
burst pin resistor that is proportional
to the average load current. Figure 7
shows the burst pin responding with
a voltage proportional to the load
current. When the load increases and
forces the burst pin voltage above
R1
499Ω
Control Input Current
for USB Applications
10ms/DIV
50
40
LEAVE BURST
30
20
VOUT = 3.3V
VIN = 3.6V
ENTER BURST
10
150
250
350
450
BURST RESISTOR (kΩ)
550
Figure 8. The load at which the converter
transitions from Burst Mode to continuous
mode is programmable with a single resistor.
Many devices now are powered and
recharged from USB ports which
have the restriction of a maximum
current draw of 500mA. A converter
that typically supplies 500mA would
not nominally exceed the USB current
limits. However, tolerances of host
regulators, USB bus powered hubs,
and cable drops result in a rather
poorly regulated USB voltage which
may vary from 5.25V down to 4.35V or
lower during a transient. For example,
if a 5V to 3.6V converter circuit is responding to a peak load of 500mA and
the USB voltage drops to 4.35V, at 80%
efficiency the input current exceeds
500mA.
Figure 9 shows an input current
monitor, which controls VOUT to clamp
the input current to 500mA. Current
injected into the FB node changes the
effective set point of the output voltage within the voltage mode control
loop. In fact, the output voltage of a
continued on page 37
–
+
Q1
2N3906
LT1677
VOUT
3.6V
3.3µH
VIN
R2
0.05Ω
D2
MBRM110LT
SW1
VIN
SHDN/SS
BURST
RT
C1
4.7µF
43.2k
LTC3532
SW2
169k
VOUT
FB
VC
12.1k
C2
220pF
12.1k
D1
1N914
+
LT1677
–
GND
22µF
86.6k
R3
24k
Figure 9. A few components may be used to limit input current for USB and other applications
18
Linear Technology Magazine • June 2006
DESIGN FEATURES L
Dual/Triple Power Supply Monitor for
Undervoltage and Overvoltage on
Positive and Negative Supplies
by Andrew Thomas
Introduction
An accurate power supply monitor
can signal when a supply overvoltage
or undervoltage condition threatens
to cause system failures, allowing
the system to deal with the situation
gracefully.
The LTC2909 is a highly customizable monitoring solution with
adjustable input thresholds, input
polarity selection, a multimode reset
timer, and an open-drain RST output.
Adjustable input thresholds allow the
user to set any trip threshold for the
comparator, subject only to the accuracy limitations of the part, instead
of having to pick from a factory-set
limited collection of thresholds.
Each adjustable input can be
configured in either polarity, allowing it to monitor negative or positive
supply voltages for undervoltage or
overvoltage. Polarity selection is controlled by simple connection of the
SEL pin—no external components
required.
The multimode timer pin can be
configured a number of ways to suit
a large variety of applications, allowing full control over the reset timeout,
elimination of the external timing
capacitor, or removal of the timeout
altogether. The RST pin is an opendrain output—it can be pulled up to
an appropriate voltage for the device
receiving the RST signal, independent
of the supply for the LTC2909. The
output can be wired-OR connected
with other supervisors or other opendrain logic, allowing any of a number
of conditions to issue a reset.
Minimal Space Required
Figure 1 shows how the LTC2909, with
just a few components, can monitor
a 24V supply for both undervoltage
and overvoltage. Almost any two reLinear Technology Magazine • June 2006
24V ±5%
47k
4.12M
11.5k
10k
VCC
ADJ1
RST
REF
SEL
ADJ2
82.5k
Any Polarity,
Undervoltage or Overvoltage
3.3V
SYSTEM
RESET
TMR
GND
Figure 1. A 24V undervoltage
and overvoltage monitor
set conditions in any system can be
monitored by appropriate connection
of the LTC2909. The small size of the
LTC2909 (available in 8-pin 3mm ×
2mm DFN and TSOT-23 packages)
keeps the monitoring solution small,
and the high accuracy of the part keeps
The two adjustable inputs
can be configured in
either polarity, allowing
the LTC2909 to monitor
negative or positive supply
voltages for undervoltage
or overvoltage. Polarity
selection is controlled by
simple connection of the SEL
pin—no external
components required.
system uptime high without sacrificing
reliability. The separate VCC pin of the
LTC2909 incorporates a shunt regulator, which allows the part to be powered
from any high availability supply, even
a high voltage rail. Furthermore, the
low quiescent current consumed by
the LTC2909 makes it suitable for
low power applications like batterypowered handheld devices.
The most common application of a
supply monitor is determining when
a positive supply is below some critical threshold required for the proper
operation of powered devices. Less
common, but no more difficult for the
LTC2909, are scenarios that require
negative supply monitoring, or determining when the voltage exceeds
some value beyond which functionality
might be impaired or powered devices
damaged.
The connection of the SEL threestate input pin determines whether
each of the ADJ input comparators is
configured as positive-polarity (input
must be above the threshold or RST
is asserted low) or as negative-polarity
(input must be below the threshold or
RST is asserted low). Inputs that are
configured as negative-polarity are
useful for resetting when the monitored
voltage is more positive (or less negative) than it should be. In other words,
a negative-polarity input can monitor
a positive supply for overvoltage (OV)
or a negative supply for undervoltage
(UV). Similarly, a positive-polarity input is useful for issuing a reset when
the monitored voltage is more negative
(or less positive) than it should be, so
it may monitor a positive voltage for
undervoltage or a negative voltage for
overvoltage. Conventionally, the terms
overvoltage and undervoltage refer to
the absolute value of the monitored
voltage, so a –5V supply at –4.3V is
undervoltage.
Connecting SEL to ground configures both adjustable inputs as negative
polarity. In this mode, the part may be
used as a dual negative undervoltage
monitor, or a dual positive overvoltage
monitor. If desired, it also functions as
19
L DESIGN FEATURES
a single negative undervoltage monitor
with a single positive overvoltage monitor. Connecting SEL to VCC configures
both inputs as positive polarity, useful
for dual positive undervoltage or dual
negative overvoltage monitors, as well
as a single positive undervoltage monitor with a single negative overvoltage
monitor. Finally, leaving the SEL pin
open configures ADJ1 as positive polarity, and ADJ2 as negative polarity.
In this configuration, the part can
monitor one positive and one negative
supply both for undervoltage, or both
for overvoltage. It can also function as a
window (undervoltage and overvoltage)
monitor for one positive or negative
supply. These polarity selections and
the corresponding applications are
summarized in Table 1.
Adjustable Inputs
The LTC2909 inputs are fully adjustable for ultimate monitoring flexibility.
Each ADJ pin connects directly to the
high-impedance input of a comparator
whose other input is tied to an internal
500mV (nominal) reference. Setting
the threshold voltage is as simple as
connecting a resistor divider from the
supply so that the ADJ input is at
500mV when the monitored supply
is at the desired threshold. By choosing the correct external resistors, the
nominal trip point can be set to any
desired value.
The typical configuration of resistors for a positive supply is as shown in
Figure 2. For a negative supply, some
offset is needed to allow the resistor
tap point to lie at 500mV. This offset
is provided by the REF pin on the
LTC2909, which provides a buffered
1V reference (with 1.5% accuracy over
the operating temperature and supply
voltage range). Thus, the typical divider
connection for a negative supply is as
shown in Figure 3. Note that positive
supplies with nominal trip points
below 500mV should be considered
“negative” for monitoring purposes
(since they require an upwards shift
to reach 0.5V). Monitoring a single
supply for UV and OV can be accomplished with three resistors, as shown
in Figure 4 for a positive supply and
Figure 5 for a negative supply.
20
Selection of resistor values is driven
by two factors: nominal trip point and
current consumption. In particular,
the selection of R1 is driven by current consumption, and the ratio of
the other resistors to R1 determines
the trip point. If the monitored voltage is typically close to its nominal
trip threshold, the voltage across R1
is approximately 0.5V, so the current
consumed by the resistor divider is
about 0.5V/R1. Supplies that operate
substantially away from their threshold cause the current consumption
to deviate from the estimate above by
about the same percentage by which
they deviate from the threshold.
In most applications, the current
consumption should be minimized.
However, as the current is reduced,
the impact of leakage at the tap point
on the monitoring accuracy becomes
more severe. The leakage current is
drawn from the driving-point impedance at the ADJ input, so the fractional
error is approximately:
ILEAK • R1 • R2 (R1+ R2)
VMON
R2
ADJx
+
R1
–
0.5V
+
–
Figure 2. Monitoring a positive supply
VMON
ADJ1
+
R3
R2
–
ADJ2
+
R1
0.5V
+
–
–
Figure 4. Monitoring a positive
supply for UV and OV
500mV
or for UV/OV circuits:
ILEAK • (R1+ R2) • R3 / (R1+ R2 + R3)
500mV
and
ILEAK • R1 • (R2 + R3) / (R1+ R2 + R3)
500mV
As a rule of thumb, the current in
the divider should be at least 100 times
the expected leakage, including the
15nA maximum internal to the part
and any external leakage sources.
The rest of the resistor values are
determined by the choice of trip point.
Since the accuracy of the LTC2909
thresholds is guaranteed to 1.5% over
the operating temperature and supply
range, the trip points should usually
be set 1.5% beyond the specified operating range of the monitored supply.
For example, a 5V ±10% supply should
have a 4.425V undervoltage trip point,
not 4.5V. See the sidebar on threshold
accuracy for an explanation.
Given a desired trip point, and the
value of R1 chosen as above, it is then
possible to calculate the appropriate
REF
R1
ADJx
+
R2
–
VMON
0.5V
+
–
Figure 3. Monitoring a negative supply
REF
ADJ1
+
R1
R2
–
ADJ2
+
R3
VMON
0.5V
+
–
–
Figure 5. Monitoring a negative
supply for UV and OV
Linear Technology Magazine • June 2006
DESIGN FEATURES L
values of the rest of the resistors.
When monitoring a positive supply
for a single fault condition, the user
should choose
V
− 500mV
R2 = R1• TRIP
500mV
Similarly, for a negative supply
(or positive supply with trip voltage
below 0.5 V),
R2 = R1 •
500mV − VTRIP
500mV
Note that if the desired trip voltage
is below ground, the value VTRIP should
be negative. The situation is slightly
more complicated when only three
resistors are used to monitor a single
supply for UV and OV. For a positive
supply with desired trip thresholds
VTRIP(UV) and VTRIP(OV), the appropriate
values are
R2 = R1 •
VTRIP(OV ) − VTRIP(UV )
VTRIP(UV )
and
R3 = R1 •
VTRIP(UV ) − 500mV VTRIP(OV )
•
500mV
VTRIP(UV )
Table 1. SEL connection for various input polarities
ADJ1
ADJ2
SEL Pin
Positive polarity:
Positive UV or Negative OV
Positive polarity:
Positive UV or Negative OV
VCC
Positive polarity:
Positive UV or Negative OV
Negative polarity:
Negative UV or Positive OV
Open
Negative polarity:
Negative UV or Positive OV
Negative polarity:
Negative UV or Positive OV
GND
Finally, for a negative supply with
desired trip thresholds VTRIP(UV) and
VTRIP(OV), the appropriate values are:
R2 = R1 •
VTRIP(UV ) − VTRIP(OV )
1V − VTRIP(UV )
and
R3 = R1 •
500mV − VTRIP(UV ) 1V − VTRIP(OV )
•
500mV
1V − VTRIP(UV )
Tables 2 and 3 show suggested
values of resistors for monitoring a
number of standard supply voltages
for UV, OV or UV and OV. Table 2 gives
values for nominal supply accuracy
of 5% (6.5% trip points), and Table 3
gives values for 10% supplies (11.5%
Table 2. Suggested resistor values for 5% monitoring
Nominal
Voltage
R1
24
5% UV
R2
R1
232k
10.2M
15
115k
12
5% OV
5% UV and OV
R2
R3
R2
R1
102k
5.11M
82.5k
11.5k
4.12M
3.09M
200k
6.19M
76.8k
10.7k
2.37M
49.9k
1.07M
102k
2.49M
76.8k
10.7k
1.87M
9
115k
1.82M
78.7k
1.43M
162k
22.6k
2.94M
5
137k
1.15M
137k
1.33M
76.8k
10.7k
732k
3.3
221k
1.15M
340k
2.05M
76.8k
10.7k
453k
2.5
115k
422k
51.1k
221k
137k
19.1k
576k
1.8
63.4k
150k
115k
324k
82.5k
11.5k
221k
1.5
59.0k
107k
137k
301k
76.8k
10.7k
158k
1.2
127k
158k
102k
158k
187k
26.1k
267k
1.0
200k
174k
100k
113k
107k
15.0k
105k
–5
133k
1.37M
118k
1.37M
174k
20.0K
2.00M
–9
97.6k
1.74M
115k
2.32M
182k
22.6k
3.65M
–12
107k
2.49M
40.2k
1.07M
40.2k
5.11k
1.07M
–15
107k
3.09M
309k
10.2M
309k
40.2k
10.2M
Linear Technology Magazine • June 2006
trip points). In the tables, the values
of R1 have been chosen to minimize
the threshold error using standard
1% resistor values, while maintaining the divider current consumption
near 5µA.
UVLO
The LTC2909 features a third highaccuracy comparator on the VCC pin,
which allows the part to function in
some applications as a triple supply monitor. The polarity of the VCC
comparator is fixed to be positive, so
the comparator creates an accurate
UVLO. The threshold of the UVLO is
also fixed, and is set at 11.5% below
the nominal threshold voltage specified in the part number. Versions are
available for standard logic supplies:
LTC2909-2.5 for 2.5V supplies (2.175V
nominal threshold), LTC2909-3.3
for 3.3V supplies (2.921V nominal
threshold), and LTC2909-5 for 5.0V
supplies (4.425V nominal threshold).
The LTC2909-2.5 is recommended for
designs that do not want monitoring
of the VCC pin. The UVLO then functions merely to ensure that RST is
not allowed to go high while the VCC
voltage is too low to guarantee proper
accuracy of the ADJ input thresholds.
The accuracy of the UVLO threshold
is the same as the ADJ thresholds:
±1.5% guaranteed over the operating
temperature range.
Glitch Immunity
A monitored supply generally has highfrequency components riding on its DC
value. These may be caused by load
transients acting on non-zero output
impedance (whether due to supply line
impedance or regulation bandwidth),
output ripple of the supply, coupling
21
L DESIGN FEATURES
22
Table 3. Suggested resistor values for 10% monitoring
Nominal
Voltage
R1
24
102k
4.22M
115k
6.04M
39.2k
10.2k
2.05M
15
200k
5.11M
200k
6.49M
41.2k
10.7k
1.33M
12
115k
2.32M
107k
2.74M
41.2k
10.7k
1.05M
9
113k
1.69M
140k
2.67M
73.2k
19.1k
1.37M
5
113k
887k
113k
1.15M
115k
30.1k
1.13M
3.3
221k
1.07M
294k
1.87M
226k
59.0k
1.37M
2.5
102k
348k
301k
1.37M
41.2k
10.7k
178k
1.8
137k
301k
86.6k
261k
63.4k
16.5k
174k
1.5
48.7k
80.6k
43.2k
102k
51.1k
13.3k
107k
1.2
137k
154k
63.4k
107k
80.6k
21.0k
115k
1.0
200k
154k
137k
169k
174k
45.3k
169k
–5
115k
1.13M
200k
2.43M
115k
24.3k
1.37M
–9
127k
2.15M
215k
4.53M
51.1k
11.8k
1.07M
–12
115k
2.55M
41.2k
1.15M
130k
30.9k
3.57M
–15
115k
3.16M
309k
10.7M
47.5k
11.5k
1.62M
10% UV
R2
R1
10% OV
R2
toggling at the reset output. Because
the timeout is defeated in comparator
mode, the LTC2909 is free to chatter
in that mode, so a small amount of
one-sided hysteresis is added to the
comparator thresholds. See “Timeout
Control” below for a description of the
hysteresis behavior.
The other concern that must be
addressed is identifying which transients cause a problem for the devices
on the supply bus. It can generally
be assumed that those devices can
continue to operate through short
700
600
MAXIMUM ALLOWABLE
GLITCH DURATION (µs)
from nearby high-frequency signals,
or noise. Ideally, the supply monitor
should decide whether the supply
voltage transient threatens the functionality of any of the devices which
are powered by that voltage rail, and
issue a reset if (and only if) it does. Unfortunately, a real supervisor cannot
use an omniscient algorithm to know
what exactly is connected to the bus or
how those devices respond to supply
transients. Given this, a number of
possible approaches exist, addressing some of the concerns related to
supply transients. These techniques
focus on eliminating two undesirable
situations that result from using a
simple comparator.
One undesirable effect that must
be prevented is rapid toggling of the
reset output (“chattering”), caused by
ripple, coupling, or noise on a supply
voltage that is near the threshold.
A common solution is to add hysteresis to the monitor threshold,
which prevents chattering as long as
the transient amplitude is less than
the amount of hysteresis. Adding
hysteresis effectively worsens the
threshold accuracy, thereby unnecessarily reducing system uptime, or
tightening the system requirements
on supply voltage. For this reason,
the LTC2909 uses other methods to
prevent chattering, and does not have
threshold hysteresis, unless the part
is configured in comparator mode,
where it would otherwise be more
susceptible to chattering than usual
(as explained below).
The primary defense against chattering is the programmed reset timeout
period. If at any time during the reset
timeout the supplies become invalid,
the timer is immediately zeroed, and
starts timing again from the beginning of the period when the supplies
become valid again. Thus, any time
the supply voltage is close enough to
the threshold that the amplitude of the
supply transients take the supply into
the invalid region, RST remains low as
long as the time between transients
is less than the reset timeout. That
is to say, the reset timeout prevents
transients with frequency greater
than 1/tRST from causing undesired
500
400
300
RESET OCCURS
ABOVE CURVE
200
100
0
1
10
0.1
100
GLITCH PERCENTAGE PAST THRESHOLD (%)
Figure 6. Allowable glitch duration
as a function of magnitude
R1
10% UV and OV
R2
R3
duration excursions outside the valid
supply region, particularly because local decoupling capacitors help prevent
such transients from appearing at the
devices. If possible, the supervisor
should not issue a reset during these
conditions.
Consider, for example, what happens when a system spins up a hard
drive connected to a monitored supply
bus. The bus voltage briefly dips, possibly falling outside the valid region,
and then returns, approximately, to
its previous value. This is normal,
expected behavior, and a microprocessor that is also connected to that bus
should function normally through the
transient (otherwise there is no way the
system can ever safely use the hard
drive). The supply monitor should not
issue a reset to the microcontroller
during such a transient.
To solve this problem, the LTC2909
has low-pass filtering on the comparator outputs, so that short duration
glitches on the monitored supply are
not passed through to the control logic.
For most systems, the response of
the system to a glitch depends on the
Linear Technology Magazine • June 2006
DESIGN FEATURES L
possible, from the corresponding supply. Negative-polarity applications may
also oscillate when the RST is driving a
large load, which causes a voltage difference between the ground of the 0.5V
internal reference, and the ground of
the monitored voltage. Several factors
can help eliminate this source of oscillation. First and foremost, the current
sunk by RST should be kept below 1mA
if possible. Good grounding practice is
also important. Input resistor dividers
which connect to ground should have a
Kelvin-sense trace directly to the GND
pin, and the path from the monitored
supply ground to the GND pin should
be low impedance (preferably through
a good ground plane).
Timeout Control
As described above, the LTC2909 has
a reset timeout delay which helps
reduce the sensitivity of the monitor
to supply glitches. For convenience,
this reset timeout can be controlled in
three different ways. If a 200ms timeout is appropriate for the application
(based on expected noise distributions
and system timing specifications), no
external components are needed to set
the timeout—simply tie the TMR pin
Why Is Threshold Accuracy Important?
In monitored systems, there is some voltage level beyond which the proper
function of the devices connected to a supply bus cannot be guaranteed.
Ideally, that is the voltage at which the supervisor should issue a reset,
since this guarantees the proper function of the system while permitting the
maximum allowable variation in supply voltage. Thus, in the ideal case, the
power supply tolerance is as loose as the devices on the bus will tolerate.
Of course, any real supervisor has limited accuracy, which tightens the
system constraints. Typically, monitor accuracy is specified as a percentage
band around the nominal trip point in which the threshold is guaranteed
to lie, such as ±1.5%. To prevent nuisance resets when the supply is operating normally, the supply tolerance and monitor accuracy bands should
not overlap.
As an example, a supply with a specified tolerance of ±5%, monitored
by a 1.5% accurate monitor must have its nominal threshold set at 6.5%
to prevent nuisance resets. With that accuracy band, the supervisor is not
guaranteed to issues a reset until the supply has reached the other end of
the monitor accuracy band, at 8%. Therefore, the devices attached to the
supply must function properly to at least an 8% deviation in supply voltage.
If this is not possible, a supply with tighter tolerance must be provided.
For comparison, if the 1.5% accurate supply monitor is replaced by a less
accurate 2.5% device, the power supply tolerance must be tightened to
±3% to ensure the same 8% operation band, thus complicating the power
supply design. L
Linear Technology Magazine • June 2006
10000
RESET TIMEOUT PERIOD, tRST (ms)
energy contained in the glitch, rather
than just the voltage amplitude of
the glitch. The duration of the glitch
also factors into that energy, so the
probability of a failure increases as
the duration of the glitch increases
(e.g. a 20% glitch on the supply may
only be tolerable for 100µs, whereas
a 5% glitch is tolerable for 1ms). The
filtering on the LTC2909 comparators
reflects this tendency. Figure 6 shows
a typical curve of the maximum glitch
duration that does not result in the
LTC2909 issuing a reset, versus the
percentage amount the glitch goes into
the invalid region.
Some of these concerns can be exacerbated by circuit board layout, so it is
also important that some care be taken
in the layout near the LTC2909. In applications which use negative polarity
comparators, capacitive coupling from
the RST output to the negative-polarity
input can cause the part to oscillate
at approximately 1/tRST if the negative-polarity input is sufficiently close
to threshold: the capacitive coupling
creates AC negative feedback around
the part. To prevent this oscillation,
the RST line should be kept away from
the relevant ADJ inputs, and, where
1000
100
10
1
0.1
1
10
100
TMR PIN CAPACITANCE, CTMR (nF)
1000
Figure 7. Reset timeout period
as a function of capacitance
to ground, and the LTC2909 uses an
internal 200ms delay generator.
For applications that require timeout periods other than 200ms, the
delay can be set by connecting the TMR
pin to a grounded capacitor, where the
delay is set at approximately 9ms per
nF of capacitance. To ensure timer accuracy, the timing capacitor should be
a low leakage ceramic type. Leakage
currents over 500nA substantially
impair timer function. As an example,
for a 50ms delay, the timer capacitor
should be 50/9 = 5.6nF.
Figure 7 shows the typical timeout
period as a function of the capacitor on
the TMR pin. Due to inherent capacitance on the TMR pin, the minimum
attainable timeout period in external
mode is about 400µs, with no external
capacitor connected to the pin. The
maximum timeout is limited to nine
seconds (1µF capacitor) by startup
concerns. Assuming that the timer
capacitor is initially discharged during
the power-up sequence, the LTC2909
initially sees that the TMR voltage is
near ground, and thus operates in
internal timeout mode. As soon as
the part is powered, a 2µA current
source begins pulling up on the TMR
pin, charging the timer capacitor
towards the ground sense threshold
(approximately 250mV). If all three
supply inputs (VCC and both ADJ
inputs) become valid, and the 200ms
internal timeout period completes
before the TMR voltage reaches the
ground sense threshold, RST goes high
after a much shorter delay than was
intended. If this startup behavior is
not a problem in a given system, the
23
L DESIGN FEATURES
maximum timeout is limited only by
the availability of large capacitors with
leakage currents below 500nA.
Finally, there are some systems
where the reset timeout delay is undesirable. For example, this may be the
case in applications where the user is
not using the LTC2909 RST pin as a
system reset line. If the user ties the
TMR pin to VCC, the LTC2909 is put
into comparator mode. In comparator
mode, the timeout delay is bypassed,
and the comparator outputs are
connected directly to the RST drive
circuitry. Due to the glitch-rejecting
low-pass filter in the comparators,
there will still be some delay from the
inputs to the RST output, based on the
amount of overdrive on the input. As
shown by Figure 6, the propagation
delay for large overdrives is about
25µs.
In comparator mode, because the
reset timeout has been removed, the
glitch and oscillation immunity of the
part have been decreased. To prevent
undesired “chattering” of the RST output when the input voltages are very
close to threshold, a small amount of
one-sided hysteresis is added to all
three comparators. The hysteresis is
“one-sided” in the sense that the validto-invalid transition is unaffected, but
the invalid-to-valid threshold is moved
about 0.7% into the valid region. Thus,
for the ADJ inputs, the threshold
voltages in comparator mode are a
function of the SEL pin state. Nominal
values are shown in Table 4.
Shunt Regulator
In most systems, it is possible to identify one supply as the one with highest
availability—that is to say the supply
which is most likely to be on, first to
power up, last to shut down, and so
on. There are a number of advantages
to powering a supply supervisor from
this highest-availability supply. First,
the RST pull-down circuits are powered by the part supply. Thus, having
the part supply come up first helps
guarantee that RST never floats high
due to insufficient pull-down strength.
Conversely, powering the part from a
high-availability supply helps maximize the uptime of the system because
the LTC2909 will not release the RST
output unless the part is properly
powered.
The problem in many systems is
that the high-availability supply is
also a relatively high-voltage supply.
For example, the highest availability
supply in an automotive system is the
12V (nominal) battery voltage, and in
a telecom system it is likely to be a
48V supply. Most supply supervisors
require an external voltage regulator
to operate from these supplies, but
the LTC2909 saves components by
integrating a 6.5V shunt regulator into
the VCC pin. All that is required is a
series-dropping resistor between the
high-voltage supply and the VCC pin.
This scheme allows the LTC2909 to be
powered from an arbitrarily high voltage, subject only to constraint by the
power dissipation in the shunt resistor.
Furthermore, the VCC pin can be used
to power other low voltage parts, as
long as their supply current (which
should be less than 5mA) is factored
into the selection of the resistor.
The shunt regulation voltage is
nominally 6.5V, and is guaranteed to
lie between 6.0V and 6.9V across the
entire operating temperature range
and across a wide range of shunt current. Selection of the series resistor
is driven by the shunt regulator bias
current. The shunt regulator bias
will be set by the amount of current
flowing through the resistor (based on
its value and the voltage drop across
it), minus the supply current of the
part, including any load drawn from
the REF pin, and the load currents of
any other devices that take advantage
of the 6.5V supply at the VCC pin. The
series resistor should be chosen to
bias the shunt regulator somewhere
between 50µA and 10mA, ideally
around 1mA.
These design constraints impose the
following limits on the series resistor.
The maximum load drawn from the
reference, plus the maximum load
drawn by other devices connected
to the VCC pin, plus 150µA for the
LTC2909 must be less than the minimum current through the resistor by
at least 50µA:
(IREF + IDEVICES )MAX + 150µA + 50µA
≤
Input
ADJ1
ADJ2
24
SEL = GND
SEL Open
SEL = VCC
Rising
500.0mV
503.5mV
503.5mV
Falling
496.5mV
500.0mV
500.0mV
Rising
500.0mV
500.0mV
503.5mV
Falling
496.5mV
496.5mV
500.0mV
R SERIES
This ensures that the shunt regulator is biased with at least 50µA of
current. On the other side, the minimum load on the reference, plus the
minimum load drawn by other devices
on VCC must be less than the maximum current through the resistor by
at most 10mA:
(IREF + IDEVICES )MIN + 10mA
≥
VSUPPLY(MAX ) − 6 V
R SERIES
This ensures that the regulator is
never shunting more than 10mA of
current. In summary, the series resistor is required to satisfy:
VSUPPLY(MAX ) − 6 V
(IREF + IDEVICES )MIN + 10mA
≤
Table 4. Nominal ADJ thresholds in comparator mode
VSUPPLY(MIN) − 6.9 V
≤R
VSUPPLY(MIN) − 6.9 V
(IREF + IDEVICES )MAX + 200µA
As an example, consider operation
from an automobile battery. For purposes of this example, the operating
range of the battery supply is approximately 10V to 60V, and we can suppose
that the user’s loading of REF and
external current use can each range
Linear Technology Magazine • June 2006
DESIGN FEATURES L
from 0µA to 100µA. The minimum
value of R is then 54V/10mA = 5.4k,
and the maximum is 3.1V/400µA =
7.75k. Given these constraints, a value
of 6.8k is probably optimal.
The above equation is actually
overly restrictive. In cases where the
supply voltage is very close to the
shunt regulation voltage, it may be
impossible to satisfy the above equation because the maximum allowable
value is less than the minimum. In
these cases, it may be assumed that the
maximum allowable value is 1k instead
of the value predicted by the formula
above, as long as the VCC pin is not
used to power other devices. There are
scenarios where the shunt regulator
cannot satisfy the needs for VCC (e.g.
those with a very large possible supply
range). These applications must use
an external voltage regulator of some
sort, which, of course, should have a
regulation voltage below 6V.
A final consideration is the power
dissipation in the series resistor, which
may be quite high for high voltage supplies. The series resistor must be rated
to handle a power of at least
( VSUPPLY(MAX) − 6V)
2
A rough rule of thumb suitable for
many applications (those that have
fairly constant REF current draw, and
have minimum supply voltages well
above 6V) is that the resistor rating
should be at least 0.1 Watt per 100
RP2A
1.43M
RP2A2
169k
10k*
MANUAL
RESET
PUSHBUTTON
RP1
49.9k
RN2
2.49M
RN1
107k
*OPTIONAL FOR ESD
volts of maximum supply, multiplied
by the ratio of maximum to minimum
supply voltage.
Returning to the automobile battery example from above, the power
dissipated in the 6.8k resistor could
be as large as 542/6800 = 0.43W (the
rule of thumb would give 0.36W), so
a 0.5W resistor is best. In reality, of
course, the battery is unlikely to stay
at 60V for long enough to heat up
the resistor substantially. If we were
to take a more reasonable DC maximum of 16V, the resistor only needs
to handle about 15mW.
±12V UV Monitor
with Manual Reset
Figure 8 shows a LTC2909 configured as an undervoltage monitor for
a system with ±12V supplies, and a
1.8V logic bus. The part is powered
from the high-availability 12V supply
RCC
27k
0.25W
RP2B
1.91M
SYSTEM
CBYP
100nF
5V
M2
RPU
10k
VCC
ADJ1
RST
LTC2909-25
SEL
ADJ2
RP1A
18.7k
RP1B
13.7k
RP1B2
681k
M1
REF
TMR
GND
M1, M2: FDG6301N OR SIMILAR
IF LOADING OF RST WILL EXCEED 1nF,
A 1nF BYPASS CAPACITOR ON M1’s
DRAIN IS RECOMMENDED
Figure 9. A 48V telecom UV/OV monitor with hysteresis
Linear Technology Magazine • June 2006
1.8V
RPU
10k
VCC
ADJ1
RST
FAULT
OUTPUT
LTC2909-25
REF
SEL
ADJ2
TMR
GND
CTMR
2.2nF
Figure 8. ±12V undervoltage monitor with pushbutton reset
VUV(RISING): 43.3V
VUV(FALLING): 38.7V
VOV(RISING): 71.6V
VOV(FALLING): 70.2V
RP2
1.07M
Applications
R SERIES
VIN
36V TO 72V
12V
–12V
CBYP
100nF
RCC
10k
through the series dropping resistor
RCC. The floating condition of SEL
sets the polarity for one positive and
one negative UV. The reset timeout is
set to 20ms nominal by CTMR, which
allows faster recovery from faults. Finally, the pushbutton allows the user
to drive ADJ1 to ground, manually
forcing a reset condition. The release
of the pushbutton is debounced by
the LTC2909’s reset timeout. If ESD
from people touching the pushbutton
is a concern, a 10k resistor in series
with the pushbutton limits the current flow into the LTC2909 to prevent
damage.
48V Telecom UV/OV
Monitor with Hysteresis
Telecom supply specifications usually
require some amount of hysteresis in
the acceptable voltage range. Since the
LTC2909 does not generally have hysteresis in its thresholds, the hysteresis
must be externally added. Figure 9
shows the LTC2909 configured to
monitor a 48V nominal supply bus for
UV and OV. The NMOS devices lower
the UV threshold (by reducing R2 for
ADJ1) and raise the OV threshold (by
reducing R1 for ADJ2) while the RST
is high. This has the effect of widening
the acceptable supply window once
the supply becomes good. The resistors are chosen so that the window
is 43.3V–70.2V when the supply is
outside the window, and 38.7V–71.6V
once the supply is good. Since the
part is powered from the 48V bus, the
series-dropping resistor is required
to be a 0.25W device to handle the
power dissipated when the bus is
overvoltage.
25
L DESIGN FEATURES
DC/DC
D1: 1N5238B OR SIMILAR
Q1, Q2: FFB2227 OR SIMILAR
RS
0.01Ω
VIN
12V
DC/DC
M1
IRLZ34
RG2
10Ω
2N6507
CG
10nF
CBYP1
100nF
PWRGD LT1641-2
GND
TIMER
SYSTEM
CBYP3
100nF
RCC
4.7k
RP2A
2.49M
RP2B
2.05M
Q2
RL2
100k
RG1
1k
SENSE GATE
3.3V
RL1 4.7k
Q1
D1
VCC
2.5V
RPU1
4.7k
CIRCUIT BREAKER AND CROWBAR
ADJ1
TMR
REF
SEL
TMR
ADJ2
GND
RFB1
10k
RP1A
102k
12V OV AND 3.3V OV DETECT
RP1B
340k
RP2E
1.15M
VCC
ADJ1
LTC2909-25
REF
SEL
RP2D
1.07M
VCC
RST
LTC2909-25
RFB2
100k
ON
FB
CT
680nF
CBYP2
100nF
VCC
RST
RP2C
221k
ADJ2
GND
ADJ1
RREF
10.7k
RP1C
51.1k
RP1D
49.9k
NTC THERMISTOR
NTHS-1206N01
R25 = 100k
R = 10.7k AT 85°C
2.5V OV AND T > 85°C DETECT
SEL
RPU2
10k
LTC2909-25
RST
REF
TMR
ADJ2
RP1E
221k
GND
12V, 3.3V and 2.5V UV DETECT
Figure 10. Automotive supply system with overvoltage, overcurrent and overtemperature protection
The recommended NMOS device
is FDG6301N, which combines both
NMOS devices in one SC70-6 package.
Other devices may be used as long as
the threshold voltage is guaranteed to
be much less than 5V, and the drainsource breakdown is greater than 10V.
Note that if the RST output is loaded
with a large capacitance, the feedback
through the gate-drain capacitance of
M1 can cause the circuit to oscillate
unless a bypass capacitor is placed
on M1’s drain.
Automotive Supply System
Figure 10 shows three LTC2909s
in a full-featured automotive supply system, providing overvoltage,
overcurrent, and over -temperature protection in addition to an
undervoltage system reset. The system
uses an LT1641-2 Hot Swap controller as a controlled electronic circuit
breaker. The IRLZ34 logic-NFET serves
as the disconnect switch, and the
10mΩ sense resistor sets a current
limit of 4.7A. After an over-current
fault, the LT1641-2 reconnects after
a delay of 160ms (set by CT).
The two LTC2909s on the left are
responsible for detecting overvoltage
and over-temperature conditions. To
guarantee that they function properly,
they must be powered from the 12V
input. The VCC pins are tied together,
and the supply current flows through
just one dropping resistor, so the voltage tends to regulate at whichever of
the shunt regulation voltages is the
lower of the two.
When any of the supply voltages
goes overvoltage, or the temperature
sensor is heated above 85°C, the
shared RST line is pulled low by one
of the two LTC2909s. This takes the
LT1641-2 ON input low, disconnecting
the power switch. At the same time,
current is pulled through Q2, turning
on Q1, which triggers the 2N6507
SCR and thereby crowbars the 12V
supply to the system, removing the
overvoltage condition. After the fault
condition disappears, the LTC2909s
apply a 200ms timeout before reconnecting to the 12V input.
The third LTC2909 serves to provide
a master reset to the system when any
of the three supplies are undervoltage,
whether because insufficient input
voltage is present, or because one of
the protection faults has tripped. The
third monitor function is provided by
the UVLO.
A Dale NTHS-1206N01 NTC thermistor with room temperature resistance
of 100k is used to detect the temperature, and may be physically located
wherever temperature monitoring is
needed. The thermistor forms part of
a resistor divider from the buffered
reference output to ground. As long
as the temperature is below 85°C, the
thermistor resistance is greater than
RREF, so ADJ1 is above its threshold,
and RST is allowed to go high. If the
temperature rises, the thermistor resistance decreases, pulling down on
ADJ1, and causing a reset when its
resistance is equal to or lower than
RREF.
Conclusion
The LTC2909 is a true one-size-fits-all
power supply monitor—a way to simplify design and parts stock. It provides
a compact solution to monitoring any
two supplies for almost any fault condition, where input polarity selection
and a buffered reference output allow
monitoring of OV conditions and negative supplies. Precision comparators,
including a third input on the part’s
VCC, increase system reliability. To
simplify design further, no regulated
voltage is required—a built-in shunt
regulator on VCC allows operation from
a high-voltage high-availability supply.
An accurate model of the LTC2909 is
included with SwitcherCAD (available
at www.linear.com), as an aid to rapid
development. L
For more information on parts featured in this issue,
go to http://www.linear.com
26
Linear Technology Magazine • June 2006
DESIGN FEATURES L
High Speed Low Power RS485
Transceivers with Integrated
Switchable Termination
by Ray Schuler and Steven Tanghe
Introduction
The LTC2859 and LTC2861 combine
a logic-selectable integrated termination resistor with a rugged 20Mbps
RS485/RS422 transceiver, providing a single die impedance-matched
network solution in a tiny package.
The low power driver features logicselectable reduced-slew rate mode for
operation below 250kbps with low EMI
emissions. The 1/8-unit load receiver
provides a failsafe output over the full
RS485 common mode range for up to
256 nodes. Both receiver inputs and
driver outputs feature robust ESD
protection exceeding ±15kV. The half
duplex LTC2859 is available in a 3mm
× 3mm DFN, while the full duplex
LTC2861 is available in both 4mm ×
3mm DFN and 16-pin SSOP packages
(see photo in Figure 1). Block diagrams
for the LTC2859 and LTC2861 are
shown in Figure 2.
Figure 1. Photograph of the LTC2861 SSOP,
LTC2861 4mm × 3mm DFN, and LTC2959
3mm × 3mm DFN packages
the data. The termination usually consists of discrete resistors that have the
RE
DE
A
15kV
SLEEP/SHUTDOWN
LOGIC AND DELAY
LTC2859
RO
120Ω
DE
RO
DI
B
15kV
Z
15kV
DI
DI
DRIVER
DRIVER
Y
15kV
Figure 2. Block diagrams of the LTC2859 and LTC2861.
LTC2859
R
SLO
RECEIVER
SLO
LTC2859
120Ω
120Ω
NODE 1
120Ω
B
15kV
LTC2859
D
A
15kV
SLEEP/SHUTDOWN
LOGIC AND DELAY
TE
RECEIVER
120Ω
DE
RE
TE
RS485 transceivers typically communicate over twisted-pair cables with
characteristic impedance ranging from
100Ω to 120Ω. Proper termination of
the cable is important to minimize
reflections that can otherwise corrupt
RO RE TE
LTC2861
LTC2859
SLO
Switchable Termination
R
same resistance as the characteristic
impedance of the cable, connected
differentially across the cable at both
ends. When using LTC2859/LTC2861
transceivers, however, no external
resistors are necessary. These devices
have an integrated 120Ω resistor on
the receiver inputs that can be enabled
logically to terminate the cable where
needed. Figure 3 shows an example
of a properly connected network using
LTC2859 transceivers with integrated
termination resistors enabled on the
two end devices. Short connection to
R
D
RO RE TE
DE
NODE 2
DI
SLO
120Ω
R
D
RO RE TE
DE
NODE 3
DI
SLO
D
RO RE TE
DE
DI
SLO
NODE 4
Figure 3. A properly connected RS485 network using LTC2859 transceivers with selectable terminators.
Linear Technology Magazine • June 2006
27
L DESIGN FEATURES
1V/DIV
1V/DIV
TE1 = 5V
TE2 = 0V
TE3 = 0V
TE4 = 5V
400ns/DIV
1V/DIV
TE1 = 5V
TE2 = 5V
TE3 = 0V
TE4 = 0V
(a)
400ns/DIV
TE1 = 5V
TE2 = 0V
TE3 = 5V
TE4 = 0V
(b)
400ns/DIV
(c)
Figure 4. Differential received signals at node 4 from network in Figure 3. properly terminated with termination
at nodes 1 and 4 (a). Improperly terminated with termination at nodes 1 and 2 (b), and nodes 1 and 3 (c)
150
135
140
130
120
140
115
110
MAGNITUDE (Ω)
RESISTANCE (Ω)
120
130
PHASE
80
60
–15
–20
20
100
0
20 40 60 80
TEMPERATURE (°C)
100 120
(a)
110
–10
–10
40
120
105
–5
100
–5
5
10
0
COMMON MODE VOLTAGE (V)
15
(b)
PHASE (DEGREES)
RESISTANCE (Ω)
125
95
–40 –20
0
MAGNITUDE
0
0.1
1
10
FREQUENCY (MHz)
–25
100
(c)
Figure 5. LTC2859 termination resistance vs temperature (a), common mode voltage (b), and frequency (c)
the intermediate stages along the cable
should be maintained as these stubs
will produce unwanted reflections.
To illustrate the importance of proper termination placement in an RS485
system, consider the network of Figure
3 where four LTC2859 transceivers
are spaced equidistantly along three
hundred feet of cat5e cable. Signals
are driven from node 1 and received
at node 4. The integrated termination
resistors in the LTC2859 devices are
switched in or out at various locations along the cable to illustrate the
effect of termination placement on
the received waveforms. No external
resistors are used.
With proper resistive termination
applied at the ends of the line (nodes
1 and 4) the received waveform has
clean transitions, as shown in Figure
4a. If the end resistive termination is
moved from node 4 to nodes 2 or 3,
the waveforms of figures 4b and 4c
result, respectively. It is clear that
placement of termination resistors
28
can have a large impact on the signal
integrity.
The termination resistor in the
LTC2859 and LTC2861 is enabled by
pulling the termination enable (TE)
pin high. The resistor is disconnected
when the termination enable is pulled
to a logic low or the device is unpowered. Figures 5 (a), (b), and (c) show
the resistance is maintained well over
temperature, common mode voltage,
and frequency.
The inclusion of a selectable 120Ω
resistive termination on the LTC2859
and LTC2861 is a significant advantage
over other RS485 transceivers. When
modifications or additions are made to
an RS485 network, the required termination changes can be made by logical
control of the termination enable pin
on the desired transceiver. This can
be done through the use of a simple
jumper or through higher level system
control where manual intervention is
prohibitive. A valuable benefit is that
every node in the network is capable of
providing termination without the use
of external resistors, making network
re-configuration more manageable.
Controller based configuration of
a network with the aid of switchable
termination can be extended beyond
simple network additions and reductions to include fault protection as
well. RS485 networks supporting bus
lengths up to 4000 feet are at risk of
breaks or disconnects in the cabling
that can interrupt service. Figure 6
shows a dual-master controlled ring
network making use of the LTC2859
logic-controlled termination to protect
against such open cable faults.
When a break in a network is detected, through loss of response from
one or more nodes, the master controller enters a low data rate mode. Even
though the break imposes a severe
impedance mismatch, low data rate
communication is still possible up
to the point of the break (as a rule
of thumb, communication without
termination is possible if the two-way
Linear Technology Magazine • June 2006
DESIGN FEATURES L
RO RE TE
DE
DI
RO RE TE
SLO
D
R
DE
RO RE TE
SLO
DI
D
R
D
120Ω
LTC2859
LTC2859
LTC2859
SLO
DI
R
120Ω
120Ω
DE
LTC2859
RO
R
RE
TE
DE
DI
D
120Ω
SLO
LTC2859
LTC2859
R
RO RE TE
R
D
DE
LTC2859
120Ω
120Ω
DI
RO RE TE
SLO
RO
120Ω
R
D
DE
DI
SLO
RO RE TE
D
DE
DI
SLO
XOR
DI
Figure 6. Line break-tolerant RS485 topology
cable propagation delay is less than
10% of the bit time). Each sequentially
addressed transceiver on the network
is polled by the masters to determine
where the break has occurred. If the
network is constructed of LTC2859
and LTC2861 transceivers with selectable termination, the masters can
instruct the two nodes on either side
of the line break to enable termination
resistance. The two master controllers may now access each node of the
bus at high data rates until physical
repairs have been made. An XOR of
the RO pin from each transceiver on
the master controller can optionally
Linear Technology Magazine • June 2006
reduce the I/O pin count to the micro
controller.
Driver
The LTC2859/LTC2861 drivers can
deliver RS485/RS422 signals up to
20Mbps. Figure 7 shows waveforms
of the part operating at the maximum
data rate. The LTC2859/LTC2861 also
feature a reduced-slew rate mode (SLO
mode), which is entered by setting the
SLO pin to a logic low level.
SLO mode increases the driver transition time to reduce high frequency
EMI emissions from equipment and
cables. In this mode the driver data
rate is limited to about 250kbps. Slew
limiting also mitigates the adverse
affects of improper line termination
and long stubs.
continued on page 32
DI
Z
Y
2V/DIV
Y–Z
20ns/DIV
Figure 7. Driver outputs toggling at
the maximum data rate of 20Mbps
29
L DESIGN FEATURES
1.5A VLDO Operates Down to 0.4V
Output and Maintains 100mV Dropout
by Bill Walter
Introduction
L1
10µH
SW
5V BOOST BST
CONVERTER
4.7µF
IN
VIN = 1.5V
0.4V
4.7µF
OFF ON
+
–
VOUT = 1.2V,
1.5A
OUT
8.06k
ADJ
SHDN
LTC3026
100k
GND
COUT
10µF
4.02k
PG
L1: MURATA LQH2MCN100K02
Figure 1. Block diagram of the LTC3026 and typical application circuit
with a low VIN to VOUT differential. The
low VIN capability is important as many
emerging handheld applications are
using a 1.5V main rail, and require
output voltages from 1.25V to 0.5V
to drive low voltage microprocessor
and microcontroller cores. Additionally, the LTC3026 offers low input
quiescent current (<1mA) at 1.5V
input and less than 1µA in shutdown,
maximizing run time in battery-powered applications. Internal protection
circuitry includes current limiting,
thermal limiting, and reverse-current
70
60
RIPPLE REJECTION (dB)
As the power supply voltage levels
for digital ICs continue to fall, supply
noise and efficiency become increasing concerns. Although many efficient
switching DC/DC converters are available, switching noise at the output
can be unacceptable at low supply
levels, or the size can be prohibitive
where multiple supply voltages are
required. One commonly used solution is to follow a switching DC/DC
converter with a low dropout (LDO)
regulator to provide an efficient low
noise supply. Nevertheless, LDOs
can introduce other problems when
used as a low voltage post regulator,
such as requiring large and expensive
tantalum capacitors, dropping out at
a voltage of 300mV or more, requiring
supply voltages greater than 1.8V, no
ability to produce output voltages below 1.2V, poor supply rejection, poor
load regulation, etc.
The LTC3026 is a 1.5A VLDO with
input voltage capability down to 1.14V
and a low adjustable output voltage
from 0.4V to 2.6V. The part also has a
very low dropout voltage of only 100mV
while delivering up to 1.5A of output
current, enabling it to optimize battery
run time from single cell applications
50
40
30
VBST = 5V
VIN = 1.5V
VOUT =1.2V
IOUT = 800mA
COUT = 10µF
20
10
0
100
1k
10k
100k
FREQUENCY (Hz)
1M
10M
Figure 2. Ripple rejection of the LTC3026
VIN = 2.5V
TO ADDITIONAL
REGULATORS
10µH
BST
SW
IN
BST
SW
4.7µF
LTC3026
VOUT1
1.8V, 1.5A
OUT
1µF
LTC3026
IN
VOUT2
1.5V, 1.5A
OUT
14k
SHDN
ADJ
100k
4.7µF
GND
11k
COUT1
10µF
4.02k
PG
LTC3026 WITH BOOST ENABLED FANOUT:
3-LTC3026 FOR VIN <1.4V
5-LTC3025 FOR VIN >1.4V
SHDN
100k
1µF
PG1
COUT2
10µF
ADJ
GND
4.02k
PG2
PG
BOOTSTRAPPED LTC3026
(BOOST DISABLED)
Figure 3. Multiple LTC3026s operating from single boost converter.
30
Linear Technology Magazine • June 2006
DESIGN FEATURES L
150
DROPOUT (mV)
protection. The LTC3026 regulator is
available in a low profile 10-lead DFN
(3mm × 3mm × 0.75mm) and 10-pin
MSOP packages with exposed pad,
offering a very compact and thermally
efficient solution.
Low Voltage Operation and
Excellent Ripple Rejection
To allow operation at low input voltages, the LTC3026 includes a boost
converter that provides the necessary headroom for the internal LDO
circuitry as shown in Figure 1. This
feature offers the flexibility of lower
input voltage capability if an external
5V rail is not available in the system.
Figure 2 shows the input ripple rejection for the application circuit of
Figure 1. From Figure 2 we see that the
ripple rejection is greater than 30dB all
the way up to 3MHz. The exceptional
ripple rejection makes the LTC3026
an excellent choice for post DC/DC
switching supply regulation.
100
1.2V
1.5V
2.0V
2.6V
50
0
0
1.0
0.5
1.5
IOUT (A)
Figure 4. Dropout voltage vs output
current for various output voltages
enabled. Figure 3 shows an application with the boost converter of one
LTC3026 driving the boost pin of a
second LTC3026 with boost converter
disabled. In this application up to five
LTC3026 LDOs can be bootstrapped
from one LTC3026 with its boost converter enabled. Figure 4 shows that
the dropout voltage of the LTC3026
is very low and well controlled over a
wide range of output voltages.
Multiple VLDO Outputs
from Single Boost
efficiency dropping the 1.8V input
down to 1.5V. In this application the
LTC3026 boost converter is disabled
and the Boost pin is driven by the
external 5V supply.
The fast transient response of the
LTC3026’s output stage overcomes
the traditional tradeoffs between
dropout voltage, quiescent current
and load transient response inherent
in most LDO regulator architectures.
Figure 6 shows that the LTC3026
only undershoots about 10mV for a
full load transient step (0A to 1.5A),
and overshoots about 30mV for a full
load transient step (1.5A to 0mA).
Additionally, the LTC3026 is designed
to detect an overshoot condition and
automatically loads the output to
bring it back into regulation, as can
1.5A
IOUT
0mA
OUT
AC 20mV/DIV
Excellent Transient Response
The boost converter of the LTC3026 and Efficiency Post DC/DC
can be disabled by connecting the Buck Converter
SW pin to ground (GND). Disabling
the boost converter allows the Boost
pin to be driven by an external 5V
supply or from the Boost pin of a
second LTC3026 with boost converter
Figure 5 shows a typical application
for the LTC3026 as post regulator for
a buck DC/DC switching converter.
The LTC1773 an efficient 1.8V output
while the LTC3026 provides over 80%
VOUT = 1.5V
COUT = 10µF
VIN = 1.7V
VB = 5V
100µs/DIV
Figure 6. Output load transient response
4.5V ≤ VIN ≤ 5.5V
33pF
30k
200pF
1
ITH
0.1µF
2
3
4
CIN
47µF
10V
SW
10
LTC1773
5
RUN/SS
SENSE–
SYNC/FCB
VIN
VFB
TG
GND
BG
RSENSE
0.04Ω
9
8
L1
2.5µH
7
1µF
SW
BST
LTC3026
IN
OUT
SHDN
ADJ
GND
PG
11k
6
Si9942DY
1µF
CBUCK
47µF
10V
100k
1%
80.6k
1%
VBUCK
1.8V
2A
100k
4.02k
VOUT
1.5V
1.5A
COUT
10µF
PG
CIN, CBUCK: TAIYO YUDEN LMK550BJ476MM
L1: CDRH5D28
RSENSE: IRC LR1206-01-R040-F
Figure 5. Efficient low noise 1.5V output from 1.8V DC/DC buck converter
Linear Technology Magazine • June 2006
31
L DESIGN FEATURES
be seen in Figure 6. Once regulation is
achieved the part disables the output
load thus keeping supply current low.
The transient scope photo also shows
that the LTC3026 has excellent load
regulation. The LTC3026 is designed to
be stable with a wide range of ceramic
output capacitors as small as 10µF.
Conclusion
The LTC3026 is an excellent choice
for low voltage applications where efficient, low noise supplies are required.
The exceptional ripple rejection and
very low dropout of the LTC3026
makes it especially well suited as a post
regulator for switching supplies. L
for
the latest information
on LTC products,
visit
www.linear.com
LTC2859, continued from page 29
Figure 8 shows single-ended and
differential driver outputs in normal
and SLO mode with corresponding
frequency spectrums operating at
250kbps. SLO mode significantly
reduces the high frequency harmonics.
The LTC2859 and LTC2861 drivers feature current limiting that
protects them from faults such as
shorting the outputs to the power
supply or ground. Short circuit current is limited to below the ±250mA
RS485 standard, with typical clamp
currents of ±150mA. If fault voltages
are greater than approximately ±10V,
currents are reduced further to limit
power dissipation. The LTC2859 and
LTC2861 also feature thermal shutdown protection, disabling the part if
a fault condition causes it to overheat.
Figure 9 shows the driver output I-V
characteristic when driven by a curve
tracer. Overcurrent protection engages
on the positive and negative sweeps
limiting the driver output current.
X, Z
Y–Z
10dB/DIV
Y–Z
1.25MHz/DIV
FREQUENCY SPECTRUM OF SAME SIGNAL
NORMAL MODE DRIVER OUTPUT
AT 125kHz INTO 100Ω RESISTOR
X, Z
Y–Z
10dB/DIV
Y–Z
SLO MODE DRIVER OUTPUT AT
125kHz INTO 100Ω RESISTOR
1.25MHz/DIV
FREQUENCY SPECTRUM OF SAME SIGNAL
Figure 8. Time and frequency domain waveforms of the LTC2859
driver output in normal and reduced-EMI SLO mode
Receiver
The LTC2859 features a low power
receiver using just 540µA of current
(typical). The LTC2859/LTC2861 failsafe feature guarantees the receiver
output to be a logic HIGH state when
the inputs are shorted, open, or terminated, but not driven for more than
about 3µs. The delay prevents signal
zero crossings from being interpreted
as a shorted input and causing RO to
go inadvertently high. This failsafe feature is guaranteed to work for inputs
spanning the entire common mode
range of –7V to +12V.
The receiver output is internally
driven high (to VCC) or low (to ground)
with no external pull-up needed. The
RO pin of the disabled receiver becomes high impedance with leakage
32
50mA/DIV
50mA/DIV
DE = 5V
DI = 0V
5V/DIV
(a)
DE = 5V
DI = 5V
5V/DIV
(b)
Figure 9. Curve traced I-V characteristic of the LTC2861 driver output showing current limiting.
(a) Pin is driven low by the LTC2861 driver. (b) Pin is driven high by the LTC2861 driver
of less than ±1µA for voltages within
the supply range.
Conclusion
Improperly terminated RS485 cabling
can severely distort signals leading
to losses in data integrity. Correcting
network termination without logic-
selectable termination often requires
physical inspection of expansive
networks. The inclusion of selectable
termination resistance on the rugged
LTC2859 and LTC2861 provide complete solutions to RS485 networking
with next-generation remote network
tuning capability. L
Linear Technology Magazine • June 2006
DESIGN IDEAS L
A Complete 500mA Linear Charger
and 300mA Synchronous Buck
Converter in a Tiny 3mm × 3mm
by Ashish Kirtania
DFN Package
Introduction
The LTC4080 is a full-featured, singlecell 4.2V Li-Ion battery charger with
an integrated synchronous buck
DC/DC converter designed primarily
for handheld applications. Its tiny
3mm × 3mm DFN package and low
external component count provide
space-savings in today’s crowded
circuit boards. The high operating
frequency (2.25MHz) of the switching
regulator minimizes overall solution
footprint further by allowing the use of
tiny, low profile inductors and ceramic
capacitors. To extend battery life, the
buck regulator offers high efficiency
burst mode operation in which the
DESIGN IDEAS
A Complete 500mA Linear Charger and
300mA Synchronous Buck Converter in
a Tiny 3mm × 3mm DFN Package ......33
Ashish Kirtania
Triple Amps Handle High-Res
Workstation Video from Single
Supplies ............................................35
Jon Munson
500mA Output Current Low Noise
Dual Mode Charge Pump ..................36
Yang Wen
R3
510Ω
D1
VCC
(3.75V
to 5.5V)
VCC
R4, 510Ω
ACPR
D2
CIN
4.7µF
LTC4080
SW
EN_BUCK
FB
MODE
GND
Julian Zhu
New Dual Input USB/AC Linear
Li-Ion Battery Chargers .....................45
Alfonso Centuori
Linear Technology Magazine • June 2006
CPL
10pF
PROG
R1
715k
R2
806k
4.2V
Li-Ion
BATTERY
VOUT
(1.5V/300mA)
COUT
4.7µF
Figure 1. Full featured Li-Ion charger with thermal management
and efficient buck regulator in a compact, single IC solution
regulator typically consumes only
20µA at no load.
ing the PROG pin voltage (VPROG)and
applying the following equation:
Battery Charger Features
ICHRG =
The LTC4080 battery charger uses
a unique constant-current, constant-voltage, constant-temperature
algorithm with programmable charge
current up to 500mA and a final float
voltage of 4.2V±0.5%. The maximum
charge current is programmed using
a single external resistor (RPROG) from
the PROG pin to ground. The charge
current (ICHRG) out of the BAT pin can
be determined at any time by monitor100
80
EFFICIENCY (%)
60
40
20
0
0.01
1000
EFFICIENCY
(Burst)
EFFICIENCY
(PWM)
VPROG • 400
RPROG
In typical operation, the charge
cycle begins in constant current mode.
When the battery approaches the final
float voltage of 4.2V, the charge current starts to decrease as the battery
charger switches to constant-voltage
mode. When the charge current drops
to 10% of the full-scale charge current,
commonly referred to as the C/10
point, the open-drain charge status
pin, CHRG, assumes a high impedance state.
An internal thermal regulator reduces the programmed charge current
100
POWER
LOSS
10
(PWM)
POWER LOSS
(Burst)
0
POWER LOSS (mW)
Step-Up/Step-Down Charge Pump DC/DC
Converter Provides up to 150mA in a
Tiny 2mm × 2mm DFN Package .........43
L1, 1OµH*
+
*COILCRAFT LPO1704-103M
Mitchell Lee
Joseph Duncan
CBAT
4.7µF
RPROG
806Ω
Ideal Diodes Combine Battery Stacks
(Minimize Heat and Voltage Loss) ......40
High Efficiency, Low Input Voltage,
Synchronous Buck Controller Drives
up to 15A Load Current .....................41
CHRG
EN_CHRG
Tiny Buck-Boost Converter for
Low Current Applications..................38
Eddy Wells
500mA
BAT
VBAT = 3.8V
0.1
VOUT = 1.5V
L = 10µH
C = 4.7µF
0.01
0.1
1
10
100
1000
LOAD CURRENT (mA)
Figure 2. Efficiency of the
buck regulator in Figure 1
VOUT
20mV/DIV
AC COUPLED
ILOAD
250mA/DIV
I=0
50µs/DIV
Figure 3. Transient response of the buck
regulator to a 0.5mA–200mA load step
33
L DESIGN IDEAS
if the die temperature attempts to rise
above a preset value of approximately
115°C. This feature not only protects
the LTC4080 and external components
from excessive temperature, it can
also reduce the charge time by allowing the user to set a higher maximum
charge current—essentially taking
into account typical, instead of worst
case, ambient temperatures for a given
application.
An internal safety timer sets the
maximum time for a charge cycle,
typically 4.5 hours. When this time
elapses, the charge cycle terminates
and the CHRG pin assumes a high
impedance state even if C/10 has not
been reached yet. A new charge cycle
of 2.25hr automatically starts if the
battery voltage falls below the recharge
threshold (typically 4.1V).
Trickle Charge and Defective
Battery Detection
At the beginning of a charge cycle,
if the battery voltage is below 2.9V,
the charger goes into trickle charge
mode, reducing the charge current to
10% of the programmed value. If the
low voltage condition persists for one
quarter of the charge cycle (1.125hr),
the battery is assumed to be defective,
the charge cycle terminates and the
CHRG output blinks at a frequency of
2Hz with a 75% duty cycle. If, for any
reason, the battery voltage rises above
2.9V, the charge cycle restarts.
Undervoltage Lockout
An internal undervoltage lockout circuit monitors the input voltage and
keeps the battery charger in shutdown
until the input rises above 3.6V and
approximately 80mV above the battery
voltage. The undervoltage condition is
indicated by a high-impedance state
of the open-drain status output pin
ACPR.
Undervoltage Charge
Current Limiting
The battery charger in the LTC4080
includes undervoltage charge current
limiting that prevents full charge current until the input supply voltage
34
reaches approximately 300mV above
the battery voltage. This feature is
particularly useful if the LTC4080
is powered from a supply with long
leads or any relatively high output
impedance.
Buck Converter Features
The buck converter in LTC4080 is
powered from the BAT pin and has a
programmable output voltage (0.8V
to VBAT) providing a maximum load
current of 300mA. It has two modes
of operation, constant frequency mode
and Burst Mode operation, selectable
via the MODE pin. In constant frequency mode, also referred to as PWM
mode, the switching regulator uses
current mode control scheme with
internal compensation and provides
efficiencies up to 91% with very low
ripple. The operating frequency of the
switching regulator is set at 2.25MHz
to minimize possible interference with
the AM band. The switching regulator and the battery charger can run
simultaneously or independently of
each other.
Burst Mode Operation
Burst Mode operation offers higher
efficiency at light loads at the cost of
higher ripple at the output voltage. In
this mode, the inductor current swings
between a maximum value (IPEAK) and
a minimum value (IZERO) irrespective of
the load as long as the FB pin voltage
(VFB) is less than the reference voltage
of 0.8V. Once VFB exceeds 0.8V, the
control logic turns off both switches
along with most of the circuitry and
the regulator draws only about 20µA
from the battery. When the output voltage drops about 2% from its nominal
value, the switching regulator wakes
up and the inductor current starts
ramping again. To minimize the output
voltage ripple, the regulator is limited
to a maximum load current of 55mA
in Burst Mode operation.
Short-Circuit Protection
In the event of a short circuit at the
output or during start-up, the shallow
negative slope (~VOUT/L) of the induc-
tor current may prevent the inductor
from discharging enough to avoid a
cumulative runaway situation over
a number of switching cycles. Even
the hard current limit on the main
PMOS switch is no guarantee against
inductor current runaway because of
current sense blanking. The switching
regulator in the LTC4080 prevents
inductor current runaway by imposing
a current limit on the synchronous
NMOS switch. If the inductor current
through the NMOS switch at the end
of a discharge cycle is not below this
limit, the regulator skips the next
inductor charging cycle.
Buck Undervoltage Lockout
To prevent unreliable operation, when
VBAT is less than 2.7V, an undervoltage
lockout circuit prevents the switching
regulator from turning on. However, if
the regulator is already running and
the battery voltage is dropping, the
undervoltage comparator does not
turn it off until VBAT becomes less
than 2.5V.
Global Thermal Shutdown
The LTC4080 includes a global thermal
shutdown which turns off the entire
part (both battery charger and switching regulator) if the die temperature
exceeds 160°C. The part resumes normal operation once the temperature
drops approximately 14°C.
Conclusion
The LTC4080, with its complete Li-Ion
battery charger and a moderately high
current buck converter in a small 3mm
× 3mm package, offers a very compact
solution with minimum number of
external components. Thermal regulation of the battery charger and the
high efficiency of the converter reduce
charge times and simplify thermal
management. L
for
the latest information
on LTC products,
visit
www.linear.com
Linear Technology Magazine • June 2006
DESIGN IDEAS L
Triple Amps Handle High-Res
Workstation Video from
Single Supplies
Introduction
Cutting edge, high resolution workstation displays demand cutting edge
bandwidth and slew rate specs of their
video amplifiers. Displays supporting
1920 × 1200 pixels, for example, must
handle over 200Mpixels/s, less than
five nanoseconds per pixel! System
supply voltages have also been dropping in order to accommodate the new,
faster digital processor technologies.
Historically the fast amplifiers used for
these formats required a total power
supply of at least 6V, particularly in
cable driver applications. Enter the
LT6557 and LT6558, ultra-fast triple
video amplifiers with internally fixed
gains of two and unity, respectively.
These devices have been specifically
engineered to operate on single supply voltages down to 3.3V and yet
maintain high bandwidth. Now it is
practical to use a low voltage digital
supply to directly power the analog
video circuitry within high resolution
products.
EN
22µF
IN R
GND
IN R
GND R 500Ω
22µF
IN G
IN G
GND G 500Ω
22µF
IN B
IN B
GND B 500Ω
LT6557
+
–
+
–
+
–
V+
V+ R
OUT G
500Ω
V+ G
OUT B
500Ω
The LT6557 and LT6558 are bipolar
voltage-feedback topology parts that
are designed for exceptionally high
slew-rate and large output swing capabilities for their operating voltage.
A blazing slew rate of 2200V/µs is
responsible for assuring that 400MHz
of bandwidth is available, regardless
of signal amplitude. These parts also
include a single-resistor-programmable biasing feature that eliminates
having to place resistor networks in
the signal path to establish the correct
DC operating point in single-supply
operation. In the typical application
for the LT6557 as a cable-driver or
the LT6558 as an input buffer, as
shown in Figure 1 and Figure 2, the
only external components in the signal
path are the coupling capacitors and
termination resistors, simplifying layout and preventing frequency response
anomalies.
The output voltage of these amplifiers can swing to within 800mV of
either rail, thus there is 1.7V of swing
EN
BCV
OUT R
500Ω
Features for Performance
and Ease-of-Use
V+ B
5V
75Ω
220µF
5V
75Ω
Figure 1. An LT6557 single-supply RGB cable driver
Linear Technology Magazine • June 2006
75Ω
75Ω
22µF
IN G
75Ω
75Ω
22µF
220µF
5V
22µF
IN R
220µF
5V
75Ω
412Ω
IN B
75Ω
75Ω
GND
IN R
GND R
IN G
GND G
IN B
GND B
by Jon Munson
available on 3.3V, and 3.4V of swing on
5.0V. This means that there is plenty of
swing available for RGB or HD video
waveforms, plus allowance for offset
variations due to AC-coupled picture
content, for the unity gain LT6558 to
operate on 3.3V or the LT6557 with
gain of two to operate on 5V. Figure 3
shows the time response of the LT6558
to 700mVP-P pulses 6ns wide while
operating from 3.3V (as in Figure 2).
Note that for high fidelity waveform
capture, a coupling circuit like that
in Figure 1 is used, with a blocking
capacitor and the signal measurement
taken after a 75Ω double-termination.
The same circuit operating at 5V exhibits even less overshoot. The typical
current consumption is about 22mA
per amplifier, and an enable feature
is provided to permit a less than 1mA
total current draw when the part is
not in use. Both parts are available
in a leaded SSOP-16 package or the
leadless 5mm × 3mm DFN-16. The
DFN model includes a bottom-side
ground pad for enhanced thermal
performance.
continued on page 42
LT6558
+
–
BCV
V+
OUT R
V+ R
+
–
OUT G
V+ G
+
–
OUT B
V+ B
3.3V
205Ω
75Ω
3.3V
75Ω
TO VIDEO
ADC INPUTS
3.3V
75Ω
3.3V
Figure 2. An LT6558 single-supply RGB buffer/ADC driver
35
L DESIGN IDEAS
500mA Output Current Low Noise
by Yang Wen
Dual Mode Charge Pump
Introduction
Charge pump (inductorless) DC/DC
converters are popular in spaceconstrained applications with low to
moderate load current (10mA–500mA)
requirements. The devices in the
LTC3203 family are low noise, high
efficiency regulating charge pumps
that can supply up to 500mA of
output current from a single 2.7V
to 5.5V supply. The LTC3203-1 and
LTC3203B-1 produce a selectable fixed
4.5V or 5V output. The LTC3203B
produces an adjustable output voltage.
The LTC3203-1 features automatic
Burst Mode operation at light load to
achieve low supply current whereas
the LTC3203B and LTC3203B-1 operate at constant frequency to minimize
both input and output noise. High
switching frequency (1MHz) makes it
possible to use only four tiny low cost
ceramic capacitors and two resistors
for operation. The device also has two
user selectable conversion modes for
optimizing the efficiency of the charge
pump. Additional features include low
shutdown current (<1µA), soft-start
at power-on and short circuit protection. The LTC3203 family is available
in a 10-lead thermally enhanced DFN
package, making it possible to build
a complete converter in less than
0.04in2. A typical application circuit
is shown in Figure 1.
Low Noise Operation
The constant frequency architecture
achieves regulation by sensing the output voltage and regulating the amount
of charge transferred per cycle. This
method of regulation provides much
lower input and output voltage ripple
than that of burst mode regulated
switched capacitor charge pumps.
The LTC3203B and LTC3203B-1
make filtering input and output noise
less demanding than burst mode
switched capacitor charge pumps
where switching frequencies depend
on load current and can range over sev36
OUTPUT PROGRAMMING
ON/OFF
SHDN VSEL
LTC3203-1
VIN
2.2µF
2.2µF
R1*
100k
VIN
VOUT
C1+
C2+
C1–
C2–
MODE
GND
IOUT(MAX) VSEL VOUT
300mA
300mA
500mA
500mA
LOW 4.5V
HIGH 5V
LOW 4.5V
HIGH 5V
10µF
VOUT
500mA
2.2µF
*R1
316k
357k
357k
402k
Figure 1. Typical application
VIN
20mV/DIV
AC-COUPLED
VIN
20mV/DIV
AC-COUPLED
VOUT
20mV/DIV
AC-COUPLED
VOUT
20mV/DIV
AC-COUPLED
VIN = 3.6V
CIN = 2.2µF
COUT = 10µF
IOUT = 300mA
2x MODE
500ns/DIV
VIN = 4V
CIN = 2.2µF
COUT = 10µF
IOUT = 300mA
1.5x MODE
500ns/DIV
Figure 3. Input and output noise in 2× mode
Figure 2. Input and output noise in 1.5× mode
eral orders of magnitude. The charge
pump operates on two phases, where
a break-before-make circuit prevents
switch cross-conduction. The higher
frequency noise due to the non-overlap
“notches” is easily filtered by a small
input capacitor and PCB parasitic
inductance. Figures 2 and 3 shows
the low input and output ripple with
a 300mA load. The device is powered
from a 3.6V input and produces a
regulated 4.5V output. The input voltage source has 0.1Ω impedance.
stacked on top of VIN and connected to
the output. The two flying capacitors
operate out of phase to minimize both
input and output ripple. Alternatively,
in 1.5× mode, it uses a split-capacitor technique rather than doubling.
The flying capacitors are charged in
series during the first clock phase,
and stacked in parallel on top of VIN
on the second clock phase. With this
technique, the input current is reduced
from more than twice the load current
to just over 1.5 times the load current, resulting in approximately 25%
less input current than what would
be required for operating in 2× mode
charge pump to drive the same load.
Therefore, the efficiency at higher VIN
is increased to approximately 90%
with VIN at somewhere between 3V and
4V. Figure 4 shows the conversion efficiency at 300mA load current for 4.5V
VOUT and 5V VOUT, respectively.
Dual Mode Conversion
The LTC3203 family offers both 1.5×
and 2× boost modes—selected by the
mode pin. In the 2× mode, the chip
works as a dual-phase regulated
voltage doubler. The flying capacitors are charged on alternate clock
phases from VIN. While one capacitor
is being charged from VIN, the other is
Linear Technology Magazine • June 2006
DESIGN IDEAS L
LTC4215, continued from page 15
power than a slot with a 20% accurate
circuit breaker.
Detect Insertion Events
via the ENABLE Pin
The EN pin can be used to sense the
insertion of a board when the LTC4215
is used in backplane resident application. A short pin on the connector pulls
EN to ground once the other, longer
pins have already been connected.
Once the EN pin crosses its falling
1.107V threshold the LTC4215 turns
on the external switch after a 100ms
debounce delay. Because a falling
edge on the EN pin corresponds to the
LTC3532, continued from page 18
DC/DC converter may be dynamically
programmed by sourcing or sinking
90
80
VOUT = 5V
70
VOUT = 4.5V
60
Conclusion
50
40
30
20
10
0
2.5
3
3.5
4
VIN (V)
4.5
5
5.5
Figure 4. Efficiency vs VIN
at 300mA load current
the VIN threshold at which the charge
pump will switch from 1.5× mode to
2× mode as VIN falls and vice versa.
at which point the external switch is
turned off with a 1mA current.
current at the FB node. Referring to
Figure 9, the equation for the input
current clamp level is:
simplifies the design of Lithium-Ion or
multi-cell powered handheld electronics. With a highly efficient automatic
Burst Mode operation, the converter
maximizes battery life in portable
devices with widely varying load requirements. Soft start, programmable
switching frequency and external
compensation make the LTC3532
suitable to a wide variety of applications. Two package options, an MS10
leaded package and a 3mm × 3mm
DFN, plus the ability to operate efficiently at high frequency, enable the
designer to minimize board area and
component height. L
1.22V R1
•
R3
R2
Figure 10 shows VOUT dropping
when input current reaches 500mA as
the load increases. In USB applications
where the input voltage is nominally
5V, a Schottky diode is used to limit
peak voltages on the SW1 pin.
IIN
500mA/DIV
IOUT
500mA/DIV
VIN = 4.3V
VOUT = 3.6V
10ms/DIV
Figure 10. As load increases, the input
current is clamped to 500mA using the
circuit of Figure 9
Linear Technology Magazine • June 2006
With low operating current, low
external parts count and robust protection features, the LTC3203 family
is well suited for low power step-up/
step-down DC/DC conversion. The
shutdown, dual mode conversion,
selectable output voltage and low noise
operation features provide additional
value and functionality. The simple
and versatile LTC3203 family is ideal
for moderate power DC/DC conversion
applications. L
insertion of a new board, the LTC4215
clears the fault register (except for
the EN Changed State bit) so that a
previously recorded fault does not
prevent the new board from starting
up. Whenever the EN pin rises or falls,
the EN Changed State bit in the FAULT
register is set to indicate that a board
has either been inserted or removed.
A STATUS register bit contains the
complement of the state of the EN
pin to indicate if a board is present.
When the board is unplugged, the
short EN pin is the first to disconnect.
The EN pin pulls up with an internal
10µA current source until the voltage
reaches the rising 1.235V threshold,
ICLAMP =
VOUT
1V/DIV
The 10% hysteresis on the MODE
pin prevents the chip from hunting
between the two modes.
100
EFFICIENCY (%)
The conversion mode should be
chosen based on considerations of
efficiency, available output current
and VOUT ripple. With a given VIN, the
1.5× mode gives a higher efficiency at
lower available output current. The 2×
mode gives a higher available output
current at lower efficiency. Moreover,
the output voltage ripple in the 2×
mode is lower due to the out-of-phase
operation of the two flying capacitors.
Typically, at low VIN, the 2× mode
should be selected, and at higher VIN,
the 1.5× mode should be selected.
The MODE pin has a precision
comparator. By connecting a resistive
divider from VIN to the MODE input
pin, the user can accurately program
Conclusion
Linear Technology’s new LTC3532
synchronous buck-boost converter
Conclusion
The LTC4215 is a smart power gateway
for hot swappable circuits. It provides
fault isolation, closely monitors the
health of the power path and provides
an unprecedented level of control over
the inrush current profile. It logs faults,
provides real-time status information,
and can interrupt the host if necessary. Meanwhile an internal 8-bit ADC
continuously monitors board current
and voltages. These features make the
LTC4215 an ideal power gateway for
high availability systems. L
37
L DESIGN IDEAS
Tiny Buck-Boost Converter for
Low Current Applications
by Eddy Wells
Introduction
One common challenge for many battery powered portable applications is
creating a regulated output voltage
above or below the input source. Traditional buck-boost approaches, such
as a dual inductor SEPIC converter or
cascaded regulators, are unacceptable
in most portable devices because of
their large solution size and low efficiency. Smaller footprint, integrated
charge-pump solutions can switch
between buck and boost operation, but
charge-pumps achieve good efficiency
at only a few operating voltages, while
efficiency dips below 50% at others.
Another compact and simple approach
forgoes a portion of the battery capacity and uses a buck (step-down) only
solution, but the advantages are hard
to justify when much of the battery
capacity is not used, as with certain
Li-Ion chemistries and a 3.3V output,
or with two alkaline cells and a 3.0V
or 2.5V output.
two small ceramic capacitors, a miniature inductor, and the ThinSOT IC.
All versions of the part are available
in a thermally enhanced 3mm × 3mm
DFN packages. A photo of the LTC3531
demo-board is shown in Figure 1.
Generating a Clean 5V
from a Noisy USB Cable
Figure 1. Compact 3.3V buck-boost application
The LTC3531 is a single inductor
200mA buck-boost converter that
generates a regulated output voltage
from a wide input voltage between
1.8V and 5.5V while maintaining high
efficiency. It is an excellent fit for low
power applications where a tiny total
solution size is required. The LTC3531
is available with fixed outputs [3.0V
or 3.3V] or with an adjustable output
that can be set between 2.0V and 5.0V.
The fixed output versions require only
IL
200mA/DIV
BOOST
BUCK-BOOST
BOOST
VOUT
0.5V/DIV
VIN
0.5V/DIV
5.5V
5V
4.5V
20µs/DIV
Generating a clean 5V output from
a USB cable or wall adapter can be
a challenge when the combination of
source impedance and load transients
cause noise and voltage droops. USB
cable voltage can vary between 5.25V
and 4.35V, while the maximum allowed decoupling capacitance on the
input is 10µF. The trace labeled “VIN”
in Figure 2 shows what can occur with
a 4.7µF input capacitor and a 100mA
load step from a powered device. The
LTC3531 produces a clean 5V output
(VOUT) with less than 100mV of peakto-peak ripple using a 22µF VOUT
capacitor. VIN and VOUT are DC aligned
at 500mV per division in Figure 2,
showing significant improvement in
noise and voltage droop. Inductor
current is also shown with operation
in both boost (VIN < VOUT) and buckboost (VIN ≅ VOUT) modes.
A complete schematic of the USB
to 5V application is shown in Figure 3 along with efficiency and power
loss curves versus load current. The
Figure 2. Noisy USB cable input to clean 5V output
4.5
95
SW2
VIN
VOUT
LTC3531
4.7µF
R2
1M
22µF
FB
SHDN
ON OFF
GND
VOUT
5V
200mA
R1
324k
EFFICIENCY (%)
SW1
80
POWER LOSS
75
COKE
1
GRAPHITE/
Li-POLYMER
3.5
3
2.5
70
2
65
0.1
Figure 3. USB to 5V application
38
10
85
POWER LOSS (mW)
USB
4.35V TO
5.25V
4
EFFICIENCY
VOLTAGE (V)
90
10µH
100
1
10
100
LOAD CURRENT (mA)
0.1
1000
0
20
40
60
USED CAPACITY (%)
80
100
Figure 4. Typical 1C lithium-ion/polymer
capacity curves
Linear Technology Magazine • June 2006
DESIGN IDEAS L
Li-Ion
+
–
SW1
SW2
VIN
VOUT
LTC3531-3.3
2.2µF
VOUT
3.3V
150mA
10µF
SHDN
GND
350
95
300
90
EFFICIENCY (%)
VIN
2.7V TO
5V
100
BOOST
MODE
85
80
MAXIMUM IOUT (mA)
10µH
BUCK
MODE
4SW
MODE
75
70
3.3VOUT AT 100mA
60
1.5
2
2.5
3 3.5 4 4.5
INPUT VOLTAGE (V)
5
Maximizing Li-Ion
Capacity for 3.3V
When compared to a straight buck
converter, the LTC3531 allows lower
input voltage operation when providing a 3.3V output from a Li-Ion input
source. Typical capacity curves for
100
+
–
LTC3531-3
2.2µF
VOUT
3V
80mA
47µF
SHDN
GND
80
10
2.5VIN
3.2VIN
75
1.8VIN
70
65
ON OFF
60
0.1
1
POWER LOSS AT 3.2VIN
1
10
100
LOAD CURRENT (mA)
Figure 7. Two AA or two AAA to 3.0V application
0.1
1000
POWER LOSS (mW)
2 x AA
ALKALINE
+
–
SW2
VOUT
EFFICIENCY (%)
85
SW1
100
2
2.5
3
3.5
4
4.5
5
5.5
Figure 6. Output current capability
vs input voltage (VOUT = 3.3V)
coke and graphite anode Li-Ion batteries are shown in Figure 4. Coke types
have a lower cut-off voltage at 2.5V,
where graphite types have a flatter
discharge curve and a 3.0V cut-off.
Solid lithium polymer batteries have
discharge curves similar to graphite.
The equivalent series resistance
(ESR) of the Li-Ion battery causes additional voltage drops at the terminal at
higher load currents. To make matters
worse, the battery protector circuit
adds additional series resistance and
the effects of ESR lower system efficiency as the battery is discharged.
90
VIN
150
VIN (V)
10µH
VIN
1.8V TO
3.2V
200
0
1.5
5.5
Figure 5. Lithium Ion to 3.3V schematic and 100mA efficiency curve.
LTC3531 operates in Burst Mode operation, with just 20µA of quiescent
current, providing high efficiency over
several decades of load current. All four
switches have an RDS(ON) of about 0.5Ω
when operating at 5V, providing >90%
efficiency at higher load currents.
250
50
65
ON OFF
L = 10µH
VOUT = 3.3V
To guarantee a 3.3V output, a buck
only design may need to use a cut-off
voltage of 3.5V or 3.6V. This translates
to a capacity loss of approximately
45% for the coke cell and 20% for the
graphite—both significant reductions
in run time. Furthermore, while graphite or polymer cells are more popular
because of their flat discharge curve,
new chemistries with greater capacity
per volume are on the horizon with
expected discharge curves resembling
the coke anode.
The wide input voltage range of the
LTC3531 allows a regulated 3.3V to be
produced from all Li-Ion chemistries,
two or three alkaline cells, or a 5V
source such as USB. The LTC3531
automatically transitions between
buck, 4-switch (buck-boost), and boost
modes based on the voltage difference
between VIN and VOUT. Figure 5 shows
a 3.3V application circuit, along with
efficiency vs input voltage for a 100mA
load. Maximum load current capability
vs input voltage (VOUT = 3.3V) is shown
in Figure 6. As expected, efficiency and
load current capability are reduced
with input voltage.
3.0V Flash Memory
Application from Two
Alkaline Cells
IL
200mA/DIV
VOUT
50mV/DIV
ILOAD
100mA/DIV
VIN = 2.5V
20µs/DIV
VOUT = 3V
COUT = 47µF
LOAD STEP = 10mA TO 100mA
Figure 8. Transient response of the circuit in Figure 7
Linear Technology Magazine • June 2006
Inexpensive MP3 players and other
relatively low capacity, low cost portable devices often replace a hard
disk drive (HDD) with flash memory
and Li-Ion batteries with disposable
alkaline—a good fit for the LTC3531.
A complete schematic for a two cell
alkaline to 3V flash memory supply is
shown in Figure 7. Efficiency is better
continued on page 40
39
L DESIGN IDEAS
Ideal Diodes Combine Battery Stacks
(Minimize Heat and Voltage Loss)
by Mitchell Lee
Introduction
Modifications
Combining multiple battery stacks to
serve a common load is an easy task for
a diode. Each stack delivers whatever
current it can muster to the load, but
back-feeding from a fresh battery to
one mostly discharged is precluded by
the presence of the blocking diode. If
you’re concerned with heat dissipation
in the diode and voltage drop at end
of discharge, diodes may leave you
pining for a better solution.
Operation over a range of 10V to 36V
is practical, and lower voltage operation is feasible by converting to a logic
level MOSFET. Because the forward
regulation point is a function of the
battery voltage, the 10M–10kΩ divider
should be adjusted to keep the drop
across the 10kΩ resistor in the range
of 10mV to 50mV.
In Figure 1 the forward drop exceeds
26.4mV when the product of the load
current and MOSFET RDS(ON) thus
dictate. For the 55mΩ IRF540, this
point is reached at load currents of
500mA.
Given some finite RDS(ON) there is
a practical limit for the load current
in any MOSFET, where RDS(ON) • ILOAD
ceases to provide any advantage over
a diode. In the case of the IRF540 this
point arrives in the 5A-to-10A range.
For higher current applications, substitute a lower RDS(ON) MOSFET. L
Active Diode
V+
than 80% with fresh alkaline cells and
better than 70% with depleted batteries. Note that overall efficiency is
lower in this application (relative to the
Li-Ion application) because of the lower
drive voltage for the switches—switch
RDS(ON) increases with decreasing drive
voltage. The adjustable version of the
part can be used to power lower voltage
flash memories (i.e. 2.5V) with similar
performance results.
Peak current requirements for flash
memory are typically lower than a
HDD, since there is no disk spin-up,
10k
+
26.4mV
–
1M
1M
IRF540
LT1494
BAT54W
+
LTC3531, continued from page 39
10M
26.4V
16 ALKALINE CELLS
–
The circuit in Figure 1 implements
an active, ideal diode using a low resistance MOSFET and a micropower
op amp. The MOSFET functionally
replaces the diode. Placing it in the
negative lead of the battery stack permits use of an N-channel device in a
simple arrangement, driven on by an
op amp if there is a slight forward voltage and off if the voltage reverses.
The forward voltage drop is regulated at 26.4mV, giving freedom from
oscillations and preventing reverse
current flow. Static current drain for
the entire circuit is less than 4µA.
18V
100nF
–VOUT
Figure 1. Battery pack ideal diode
but load transients still occur when
memory is accessed. The LTC3531’s
response to a 10mA–100mA load step
is shown in Figure 8. Burst Mode®
operation provides a rapid transient
response since there is no compensation loop to slew. Peak-to-peak
voltage ripple plus load step is under
50mV with a 47µF output capacitor.
The output voltage ripple stays fairly
constant over input voltage. Half the
output capacitance (22µF) results in
approximately twice the load step plus
voltage ripple (100mV).
Summary
The LTC3531 provides a simple,
compact buck-boost solution for
lower current, portable applications.
A complete solution, 1mm in height,
can fit in a 35mm2 footprint. The
part maintains high efficiency over
a wide range of input voltages and
load currents, extending battery run
time, while providing the flexibility to
address many designs such as 2-cell
alkaline, USB, and present day or
emerging Li-Ion chemistries. L
For more information on parts featured in this issue,
go to http://www.linear.com
40
Linear Technology Magazine • June 2006
DESIGN IDEAS L
High Efficiency, Low Input Voltage,
Synchronous Buck Controller Drives
by Joseph Duncan
up to 15A Load Current
Introduction
IPRG
The LTC3822 is a synchronous stepdown DC/DC converter that drives
external N-channel power MOSFETs
to maximize average current drive for
the lowest cost. Its No RSENSE constant
frequency architecture minimizes the
number of external components, and
a programmable frequency of up to
750kHz allows the use of small surface-mount inductors and capacitors.
This DC/DC controller is optimized
for 3.3VIN and Lithium-Ion applications allowing VOUT as low as 0.6V
while maintaining 1% precision. The
all N-channel MOSFET drive simplifies component selection as well as
drastically increasing the current
capabilities of a typical circuit. Even
with 3.3V gate drive, the LTC3822 is
capable of controlling more than 15A
load current while maintaining high
efficiency.
RUN
ITH
LTC3822
SW
GND
59k
GND
VOUT
1.8V
500mV/DIV
VIN = 4.2V
RLOAD = 1Ω
Figure 2. Sample footprint for
application circuit in Figure 1
ramps the output voltage from 0V to its
final value in 800µs (Figure 3). This is
done without the need for an external
capacitor. The LTC3822 incorporates
No RSENSE technology to sense the
inductor current from the drain to
source voltage (VDS) of the top-side
CMDSH-3
Si4866
0.22µH
0.22µF
47µF
2x
VIN
2.75V TO 4.5V
VOUT
1.8V
100µF 15A
2x
CMDSH-3
BG
118k
Figure 4. High current application delivering 1.8V at 15A.
Linear Technology Magazine • June 2006
BG
118k
VFB
59k
VOUT
1.8V
100µF 8A
Figure 1. Typical application delivering 1.8V at 8A.
Si4866
680pF
0.47µH
VFB
BOOST
5.1k
0.22µF
680pF
TG
FREQ
SW
BOOST
5.1k
Figure 1 shows a 1.8V, 8A application that operates over input voltages
between 2.75V and 4.5V, perfect for
3.3V or Li-Ion inputs. This application
occupies much less space than would
be expected for its current capabilities,
as shown in Figure 2.
During startup, the inter nal soft-start circuitry smoothly
VIN
LTC3822
VIN
2.75V TO 4.5V
FDS6898A
TG
FREQ
ITH
CMDSH-3
47µF
RUN
Compact, 1.8V,
8A Application
IPRG
VIN
200µs/DIV
Figure 3. Internal soft-start ramps the output
voltage smoothly without requiring an external
capacitor.
power MOSFET. The maximum load
current that the controller is capable
of driving is determined by the RDS(ON)
of this MOSFET. Since the LTC3822
incorporates all N-channel MOSFET
drive, lower RDS(ON) (and cheaper)
devices are available for the top-side
MOSFET, when compared to traditional complementary MOSFET drive.
Increasing the Current to 20A
Figures 4 and 5 show two ways to
raise the current capability of the
regulator by lowering the RDS(ON) of
the MOSFETs. In Figure 4, MOSFETs
with a much lower RDS(ON) than those
of Figure 1 are used. Because they
are in individual SO-8 packages, their
thermal capabilities are also higher.
This application is designed for a 15A
continuous current load. Figure 5 in41
L DESIGN IDEAS
5V SECONDARY
SUPPLY
CMDSH-3
IPRG
VIN
47µF
RUN
ITH
LTC3822
SW
0.22µF
0.47µH
IL
2A/DIV
VOUT
1.8V
100µF 10A
BOOST
5.1k
VOUT
100mV/DIV
FDS6898A
TG
FREQ
VIN
2.75V TO 4.5V
ILOAD
2A/DIV
680pF
GND
BG
VFB
59k
VIN = 3.3V
VOUT = 1.8V
ILOAD = 1A TO 3A
118k
40µs/DIV
Figure 6. Transient performance
of the converter in Figure 1
Figure 5. High efficiency application deriving gate drive voltage from a secondary 5V supply.
stead utilizes a secondary 5V supply
to provide a higher gate drive voltage
to the MOSFETs. Higher gate drive
voltages lower RDS(ON) while simultaneously allowing the use of cheaper
logic-level MOSFETs. The maximum
load current can also be tailored using the current limit programming
pin, IPRG. This three-state pin sets
the peak current sense voltage across
the top-side MOSFET. Combining all
three high current approaches (utilizing low RDS(ON) MOSFETs, powering
the gate drive from a secondary 5V
supply, and setting current limit to its
highest value) enables applications in
excess of 20A.
The LTC3822 incorporates OPTILOOP® compensation to enable the
user to choose optimal component
values to compensate the loop over
a wide range of operating conditions
with the minimum number of output
capacitors. Figure 6 shows the tran-
Conclusion
LT6557, continued from page 35
matically tracks downward with the
supply if below 4V. The selection of
input bias point may depend on the
application, but the values shown for
the programming resistors in Figures
1 and 2 are representative of most
designs.
grammable biasing, these devices
offer minimal parts-count AC-coupled
amplifier solutions for very high-resolution applications. The LT6557, with
its gain of two, is designed for RGB
output ports such as in video routers and KVM switch products. The
LT6558, with unity gain, is designed
as an RGB input port buffer and/or
ADC driver, such as in computer or
home-theater display products. L
The LT6557 and LT6558 are designed
specifically with single-supply ACcoupled operation in mind. Each
input includes an internal currentcontrolled bias voltage source like that
shown in Figure 3. A single external
resistor RBCV programs the input bias
voltages as shown in Figure 4 for the
LT6557. The LT6558 RBCV function is
similar to Figure 4, but is optimized
for producing higher biasing levels to
account for the lower gain and auto-
Conclusion
The LT6557 and LT6558 triple video
amplifiers are optimized specifically
for operation on low voltage single
supplies. With preset gain and pro-
V+
VBCV = 48mV (TYPICAL)
VSUPPLY = 3.3V
VS = 5V
1.5
1.0
0.5
9.1k
2ns/DIV
Figure 3. Fast pulse response of
LT6558 on 3.3V single supply
42
VBCV
RSET
2.5k
IN
V
• 9.1k
VBIAS(IN) = BCV
RSET
2.5
2.0
I=
OUTPUT
100mV/DIV
The LTC3822 delivers currents as high
as 20A for single-output applications
using a minimum number of components in a tiny complete solution
footprint. L
INPUT VOLTAGE (V)
Automatic Biasing Feature
OPTI-LOOP Compensation
sient response for the circuit in Figure
1 with a load step of 1A to 3A. The
output overshoots by approximately
100mV on a 1.8V output and then
settles in about 50µs.
Figure 4. Simplified schematic of LT6557
input biasing circuit (LT6558 similar)
0
200
300
400
500
RBCV (Ω)
600
700
Figure 5. Relationship of LT6557 input bias
voltage to programming resistor RBCV
Linear Technology Magazine • June 2006
DESIGN IDEAS L
Step-Up/Step-Down Charge Pump
DC/DC Converter Provides up to
150mA in a Tiny 2mm × 2mm
DFN Package
by Julian Zhu
Introduction
A wide variety of handheld and portable
applications are powered by Li-Ion
batteries or AA cells. The wide input
voltage range of a single Li-Ion battery
(2.7V–4.2V) or 2 AA cells (1.8V–3.0V)
requires a DC/DC converter that can
step-up or step-down the input voltage
to provide a fixed output voltage such
as 3.3V or 2.5V. The new LTC3240
step-up/step-down DC/DC converter
is ideally suited for such applications
and can provide up to 150mA in a tiny
6-lead 2mm × 2mm DFN package.
For input voltages greater than
the regulated output voltage the
LTC3240 operates as a low dropout
regulator. When the input voltage
decreases to within about 150mV of the
regulated output voltage, the LTC3240
automatically switches to step-up
Figure 1. LTC3240-3.3 step-up/stepdown converter capable of delivering
current up to 150mA
mode. In step-up mode, the LTC3240
operates as a constant frequency
(1.2MHz) voltage doubling charge
pump. The LTC3240 requires only
three tiny external ceramic capacitors
for an ultra small application footprint
as shown in Figure 1.
1µF
Li-Ion OR
3-CELL NiMH
C+
VOUT
C–
VIN
2.7V TO 4.5V
1µF
LTC3240-3.3
4.7µF
GND
OFF ON
3.3V
IOUT = 150mA
SHDN
Figure 2. The regulated 3.3V from battery voltage
100
IOUT = 30mA
80
3.40
70
3.35
3.30
3.25
40
10
0
2.2
2.7
3.2 3.7 4.2 4.7
INPUT VOLTAGE (V)
5.2
5.7
Figure 3. Output voltage vs input voltage
(full range)
Linear Technology Magazine • June 2006
VOUT
20mV/DIV
DOUBLER TO LDO
MODE (VIN RISING)
30
20
A typical application circuit for
LTC3240-3.3 is shown in Figure 2.
The input can be a single Li-Ion battery
or three AA cells. Figure 3 shows the
output voltage variation for the entire
input voltage range at a load current
of 30mA.
A new or recharged battery starts
out at its highest terminal voltage. As
the battery discharges, its terminal
voltage continues to drop until the
next recharge. The LTC3240 optimizes
the output efficiency by continuing to
operate in the step-down (LDO mode)
for most of the battery life. When the
battery voltage gets low enough it
automatically switches into chargepump mode to squeeze out maximum
energy from the battery before its next
ILOAD = 1mA
50
3.20
Application for Li-Ion or
Three AA Battery Input
to 3.3V Out
ILOAD = 40mA
60
3.15
3.10
1.7
LDO TO DOUBLER
MODE (VIN FALLING)
90
3.45
EFFICIENCY (%)
OUTPUT VOLTAGE (V)
3.50
The LTC3240 features low no load
operating current (65µA typical) and
ultra low shutdown current (<1µA).
Built-in soft-start circuitry prevents
excessive inrush current during
start-up. The thermal-shutdown
and current-limit circuitry allow the
parts to survive a continuous output
short-circuit.
1.8 2.3
2.8 3.3 3.8 4.3 4.8
SUPPLY VOLTAGE (V)
5.3
5.8
Figure 4. Efficiency vs input voltage
(LTC3240-3.3)
VIN = 2.5V
ILOAD = 150mA
2µs/DIV
Figure 5. Output voltage ripple (LTC3240-3.3)
43
L DESIGN IDEAS
LT3486, continued from page 11
as 70V while both providing both
overvoltage protection and remaining below the 42V maximum switch
voltage. The charge pump Schottky
diodes and capacitors double the
effective output voltage for a given
duty cycle while the LT3486 LED
driver continues to regulate the 100mA
constant LED current. The LEDs in
Figure 7 have higher forward voltage
than those in Figure 1 at 100mA,
ripple voltage (<1% for 3.3V output)
at 150mA (See Figure 5).
To extend battery life at light
loads, in charge pump mode, the part
operates in high efficiency Burst Mode
operation. In this case, the LTC3240
delivers a minimum amount of charge
for a few cycles, and then enters a low
current state until the output drops
low enough to require another burst
of charge.
Conclusion
resulting in a total string voltage as
high as 40V. If more LEDs are needed,
the string voltage can be stacked up
to 70V before hitting the overvoltage
protection level, but the peak switch
current limit cannot be exceeded. As
the string voltage and LED current
goes up, the minimum input voltage
also rises. Figure 8 shows the typical
peak switch current limit dropping as
duty cycle increases. In addition to
the peak inductor current, the voltage
doubler also adds additional charge
pump capacitor current.
ZLLS400
ZLLS400
COUT1B
2.2µF ZLLS400
25V
PVIN
8V TO 18V
CPV(IN)
4.7µF
25V
2.2µF
25V
ZLLS400
10µH
A915AY-100M
COUT1A
2.2µF
25V
100mA
10 WHITE
LEDs >34V
CTRL1
BRIGHTNESS
ADJUST
SHDN
VIN
OVP2
CTRL1
CTRL2
FB1
VC1 GND RT
PWM1
2.8k
PWM INPUT 4700pF
100Hz
5V
1000:1 DIM RATIO
ZLLS400
COUT2A
2.2µF
35V
REF
0.1µF
CTRL2
BRIGHTNESS
ADJUST
FB2
63.4k
The LT3486 is a dual 1.3A LED string
driver with 1000:1 PWM dimming capability. Its 3% LED current accuracy,
low sense voltage, low shutdown current, overvoltage protection and wide
input voltage range make it ideal for
high power LCD panels in a variety
of applications including automotive
displays and notebook computers. The
simple 5:1 analog dimming ratio and
more precise 1000:1 PWM dimming
ratio provide the displays with enough
brightness control for daylight and
nighttime use while retaining their
color characteristics across brightness
levels. L
1800
FDN5630
PWM2
VC2
Conclusion
100mA
10 WHITE
LEDs >34V
SW2
OVP1
PWM1
100k
2.2µF
25V
CV(IN)
4.7µF
25V
SHDN LT3486EFE
FDN5630
RSENSE1
2Ω
1%
10µH
A915AY-100M
VIN
5V
SW1
COUT2B
2.2µF
25V
ZLLS400
The LTC3240 step-up/step-down
charge pump DC/DC converter
provides fixed regulated output voltage
with currents up to 150mA from a wide
input voltage range in a small 6-lead
2mm × 2mm DFN package. It is ideally
suited for efficient DC/DC conversion
in space-constrained applications
such as battery-powered handheld
electronics. L
PEAK SWITCH CURRENT LIMIT (mA)
recharge cycle. Figure 4 shows the
efficiency of LTC3240-3.3 as a function
of input voltage.
In step-up (charge pump) mode, the
LTC3240 uses a unique architecture
to optimize the charge transferred to
the output in each clock cycle, thus
minimizing the output ripple. The part
only needs a 4.7µF, 0603 size ceramic
capacitor to obtain a 32mV maximum
PWM2
100k
RSENSE2
2Ω
1%
2.8k
4700pF
PWM INPUT
100Hz
5V
1000:1 DIM RATIO
Figure 7. LED driver uses 8V–18V input to drive two strings of ten
100mA LEDs (40V max per string) with 1000:1 PWM dimming
VIN = 3.6V
1700
1600
1500
1400
1300
1200
20
30
40
50 60 70 80
DUTY CYCLE (%)
90
100
Figure 8. Typical peak switch current limit
drops as duty cycle increases above 50%
For more information on parts featured in this issue,
go to http://www.linear.com
44
Linear Technology Magazine • June 2006
DESIGN IDEAS L
New Dual Input USB/AC Linear
by Alfonso Centuori
Li-Ion Battery Chargers
Introduction
Digital Cameras, PDAs, mobile phones
and MP3 players all use batteries that
are commonly charged via either a
wall adapter or through the USB. The
LTC4076 and LTC4077 lithium-ion
battery chargers are specifically designed to detect power at the inputs
and automatically select the appropriate source for charging. Using a
constant current/constant voltage
algorithm, the chargers can be programmed to deliver up to 950mA of
charge current with a final float voltage accuracy of ±0.6%. The LTC4076
and LTC4077 include an internal Pchannel power MOSFET and thermal
regulation circuitry with no blocking
diode or external sense resistor required. Thus, the basic dual-source
charger requires only three external
components.
The LTC4076 offers a programmable current based termination
scheme. The CHRG open-drain status
pin can be programmed to indicate
the battery charge state according to
the needs of the application. The PWR
open-drain status pin indicates that
enough voltage is present at one of the
inputs to charge a battery. With power
applied on both inputs, LTC4076 and
LTC4077 can be put into shutdown
mode, reducing the DCIN supply
current to 20µA, the USBIN supply
current to 10µA and the battery drain
current to less than 2µA.
Internal thermal feedback regulates the charge current to maintain
IBAT = 475mA
IBAT = 100mA
WALL
ADAPTER
USB
POWER
IBAT = 475mA
IBAT = 95mA
WALL
ADAPTER
USB
POWER
IDC
1.24k
1%
1k
1.24k
1%
1k
1µF
IDC
2.1k
1%
+
CHRG
ITERM
GND
1-CELL
Li-Ion
BATTERY
1k
1%
Figure 1. An LTC4076 USB/wall adapter Li-Ion charger
a constant die temperature during
high power operation or high ambient
temperature conditions.
USB Compatibility
The HPWR pins of the LTC4076 and
LTC4077 provide an easy method to
choose between two different USB
power modes: high power (usually
≤ 500mA) and low power (usually ≤
100mA).
With the LTC4076, a logic high on
the HPWR pin sets the charge current
to 100% of the current programmed by
the IUSB pin resistor, while a logic low
on the HPWR pin sets the charge current to 20% of the current programmed
by the IUSB pin resistor.
With the LTC4077, a logic high on
the HPWR pin sets the charge current to the value programmed by the
IUSB pin resistor, while a logic low
on the HPWR pin sets the current to
the value programmed by the IUSBL
pin resistor.
800mA (WALL)
475mA (USB)
100mA (USB)
BAT
1k
CHRG
IUSBL
GND
A weak pull down on the HPWR
pin sets the part to default to the low
power state. The HPWR pin provides
a simple control for managing charge
current as shown in Figure 1 and Figure 2. The presence of a wall adapter
takes priority over the USB input and
the IDC 1.24kΩ resistor sets the total
maximum charge current to 800mA.
When USB power is present, and a
wall adapter is not, the IUSB 2.1kΩ
resistor sets the charge current to
476mA with HPWR in its high state,
and 95mA with HPWR in its low state
for the LTC4076 (or 100mA with IUSBL
2kΩ resistor for the LTC4077).
Programmability
The LTC4076 and LTC4077 provide a
great deal of design flexibility including programmable charge current and
programmable current termination.
The charge currents are programmed
using a resistor from the IDC, IUSB
and IUSBL (LTC4077 only) pins to
ground as indicated in the following
equations:
ICHG =
1000 V
RIDC
(Wall Adapter Present)
ICHG =
1000 V
RIUSB
(USB HPWR = High)
ICHG =
200 V
RIUSB
(USBHPWR = Low,
LTC4076 Only)
ICHG =
200 V
RIUSBL
(USBHPWR = Low,
LTC4077 Only)
1k
+
PWR
2.1k
1%
1µF
1µF
IUSB
BAT
USBIN
PWR
USBIN
1µF
DCIN
IUSB
HPWR
LTC4077
DCIN
800mA (WALL)
475mA (USB)
95mA (USB)
HPWR
LTC4076
1-CELL
Li-Ion
BATTERY
2k
1%
Figure 2. An LTC4077 USB/wall adapter Li-Ion charger
Linear Technology Magazine • June 2006
45
L NEW DEVICE CAMEOS
LTC4076
OR
LTC4077
WALL
ADAPTER
USB
POWER
BAT
DCIN
1µF
USBIN
1µF
500mA
+
IUSB
RISET
2k
1%
IDC
GND
Figure 3. LTC4076 or LTC4077 dual input charger circuit. The wall adapter charge current and
the USB charge current (HPWR = High) are both programmed to be 500mA with just one resistor.
Both the LTC4076 and LTC4077
terminate the charge cycle based on
the battery current. For the LTC4076
the current threshold is programmable
and for the LTC4077 the current
threshold is fixed, as described below.
For LTC4076, the programmable
current detection threshold, ITERM, is
set by connecting a resistor, RITERM,
from ITERM to ground. The following
formula programs the termination
current:
ITERM =
100 V
RITERM
For the LTC4077 the termination
current is fixed at 10% of the programmed charge current as set by IDC
or IUSB (HPWR = High). When HPWR
in its low state, the termination current
is 50% of the current programmed
by RIUSBL.
The condition of the CHRG pin
indicates the charge state. A strong
pull-down on the CHRG pin indicates
that the battery is charging. When
the current termination threshold is
reached the CHRG pin assumes a high
impedance state.
Avoiding Unnecessary
Charge Cycles
LTC4076 and LTC4077 are designed
to avoid unnecessary charge cycles
to extend the life of Li-Ion batteries.
When power is first applied or when
exiting shutdown, the LTC4076 and
LTC4077 check the voltage on the BAT
pin to determine its initial state. If the
BAT pin voltage is below the recharge
threshold of 4.1V (which corresponds
to approximately 80%–90% battery
capacity), LTC4076 and LTC4077
enter charge mode and begin a charge
cycle. If the BAT pin is above 4.1V, the
battery is nearly full and the charger
does not initiate a charge cycle and
instead enters standby mode. When in
standby mode, the chargers continuously monitor the BAT pin voltage.
When the BAT pin voltage drops below
4.1V, the charge cycle is automatically
restarted. This feature eliminates the
need for periodic charge cycle initiations, ensures that the battery is
always fully charged, and prolongs
battery life by reducing the number
of unnecessary charge cycles.
Conclusion
LTC4076 and LTC4077 are complete
Linear Li-Ion battery chargers compatible with portable USB applications.
They are designed to accommodate
charging from both a wall adapter
and a USB input. The versatility, low
quiescent current, simplicity, high
level of integration and small size of
the LTC4076 and LTC4077 provide an
ideal choice for many portable USB applications. LTC4076 and LTC4077 are
available in a small 10-lead low profile
3mm × 3mm DFN package. L
New Device Cameos
New Timing and High Voltage
Output Enhance Push Button
On/Off Controller
The LTC2954 is an upgrade to the
LTC2950 family of push button
controllers. Two features have been
added: new push button controlled
interrupt timing, and a high voltage
open drain enable output capable of
33V operation.
The LTC2954 now provides interrupt and power down modes to allow
more flexible and reliable system
shut down. Once in the powered on
state, momentary presses on the push
button provide interrupts to system
logic. This can be useful for system
housekeeping and power down under
46
firmware control. Should the system
require a forced power down, a long
duration press and hold of the push
button automatically releases the enable pin, thus turning system power
off. The timing of this latter power
down mode is adjustable with external capacitance on the power down
timing pin.
The new high voltage enable output
of the LTC2954-2 is ideally suited to
drive the gate of a high voltage power
PFET. This allows a user to connect/
disconnect input power from its load
simply by toggling the de-bounced
push button input.
The LTC2954 operates over a wide
2.7V to 26V input voltage range while
consuming only 6µA of supply current
and is available in 8-Pin 3mm × 2mm
DFN and ThinSOT™ Packages. Two
versions of the part are available to
accommodate either positive or negative enable polarities. L
For further information on any
of the devices mentioned in this
issue of Linear Technology, use
the reader service card or call the
LTC literature service number:
1-800-4-LINEAR
Ask for the pertinent data sheets
and Application Notes.
Linear Technology Magazine • June 2006
DESIGN TOOLS L
DESIGN TOOLS
Product Information
Applications Handbooks
Linear Technology offers high-performance analog
products across a broad product range. Current
product information and design tools are available at
www.linear.com. Our CD-ROM product selector tool,
which is updated quarterly, and our most recent databook
series can be obtained from your local Linear Sales
office (see the back of this magazine) or requested from
www.linear.com.
Linear Applications Handbook, Volume I — Almost a
thousand pages of application ideas covered in depth by
40 Application Notes and 33 Design Notes. This catalog
covers a broad range of real world linear circuitry. In
addition to detailed, systems-oriented circuits, this
handbook contains broad tutorial content together with
liberal use of schematics and scope photography. A
special feature in this edition includes a 22-page section
on SPICE macromodels.
www.linear.com
Product information and application solutions are available at www.linear.com through powerful search tools,
which yield weighted results from our data sheets,
application notes, design notes, Linear Technology
magazine issues and other LTC publications. The LTC
website simplifies the product selection process by
providing convenient search methods, complete application solutions and design simulation programs for
power, filter, op amp and data converter applications.
Search methods include a text search for a particular part
number, keyword or phrase, or a powerful parametric
search engine. After selecting a desired product category,
engineers can specify and sort by key parameters and
specifications that satisfy their design requirements.
Visit www.linear.com/mylinear to register and access
your MyLinear home page. Here you can store your
favorite LTC products, categories, product tables, contact
information, preferences and more.
Purchase Products Online
Credit Card Purchases—Purchase online direct from
Linear Technology at www.linear.com using a credit card.
Create a personalized account to check order history,
shipment information and reorder products.
Linear Express Distribution — Get the parts you need.
Fast. Most devices are stocked for immediate delivery.
Credit terms and low minimum orders make it easy to get
you up and running. Place and track orders online. Apply
today at www.linear.com or call (866) 546-3271.
CD-ROM
The June 2006 CD-ROM contains product data sheets,
application notes and Design Notes. Use your browser
to view product categories and select products from
parametric tables or simply choose products and documents from part number, application note or design
note indexes.
Linear Applications Handbook, Volume II — Continues
the stream of real world linear circuitry initiated by Volume
I. Similar in scope to Volume I, this book covers Application Notes 40 through 54 and Design Notes 33 through
69. References and articles from non-LTC publications
that we have found useful are also included.
Linear Applications Handbook, Volume III —
This 976-page handbook includes Application Notes 55
through 69 and Design Notes 70 through 144. Subjects
include switching regulators, measurement and control
circuits, filters, video designs, interface, data converters,
power products, battery chargers and CCFL inverters.
An extensive subject index references circuits in Linear
data sheets, design notes, application notes and Linear
Technology magazines.
Brochures
Power Management for Portable Products — The
solutions in this product selection guide solve real-life
problems for cell phones, digital cameras, PDAs and
other portable devices, maximizing battery run time
and saving space. Circuits are shown for Li-Ion battery
chargers, battery managers, USB support, system
power regulation, display drivers, white LED drivers,
photoflash chargers, DC/DC converters and RF PA power
supply and control.
Automotive Electronic Solutions — This selection guide
features high performance, high reliability solutions for
a wide range of functions commonly used in today’s
automobiles, including telematics, infotainment systems,
body electronics, engine management, safety systems
and GPS navigation systems.
Industrial Signal Chain — This product selection guide
highlights analog-to-digital converters, digital-to-analog
converters, amplifiers, comparators, filters, voltage
references, RMS-to-DC converters and silicon oscillators designed for demanding industrial applications.
These precise, flexible and rugged devices feature
parameters fully guaranteed over the –40°C to 85°C
temperature range.
Wireless & RF Solutions — This brochure presents high
performance RF solutions for use in various transceiver
architectures employed in 2G, 2.5G and 3G cellular
basestations, wireless point-to-point radios, WiMAX,
broadband wireless access, satellite receivers, GPS
receivers, cable and VOD infrastructure equipment, RFID
readers, wireless handheld transceivers and software
defined radios.
Software
SwitcherCAD™ III/LTC SPICE — LTC SwitcherCAD III is
a fully functional SPICE simulator with enhancements
and models to ease the simulation of switching regulators. This SPICE is a high performance circuit simulator
and integrated waveform viewer, and also includes
schematic capture. Our enhancements to SPICE result
in much faster simulation of switching regulators than is
possible with normal SPICE simulators. SwitcherCAD III
includes SPICE, macromodels for 80% of LTC’s switching
regulators and over 200 op amp models. It also includes
models of resistors, transistors and MOSFETs. With this
SPICE simulator, most switching regulator waveforms
can be viewed in a few minutes on a high performance
PC. Circuits using op amps and transistors can also be
easily simulated. Download at www.linear.com
FilterCAD™ 3.0 — FilterCAD 3.0 is a computer aided design program for creating filters with Linear Technology’s
filter ICs. FilterCAD is designed to help users without
special expertise in filter design to design good filters
with a minimum of effort. It can also help experienced
filter designers achieve better results by playing “what if”
with the configuration and values of various components
and observing the results. With FCAD, you can design
lowpass, highpass, bandpass or notch filters with a
variety of responses, including Butterworth, Bessel,
Chebychev, elliptic and minimum Q elliptic, plus custom
responses. Download at www.linear.com
SPICE Macromodel Library — This library includes LTC
op amp SPICE macromodels. The models can be used
with any version of SPICE for analog circuit simulations.
These models run on SwitcherCAD III/LTC SPICE.
Noise Program — This PC program allows the user to
calculate circuit noise using LTC op amps, determine the
best LTC op amp for a low noise application, display the
noise data for LTC op amps, calculate resistor noise and
calculate noise using specs for any op amp.
Battery Charger Solutions — This guide identifies
optimum charging solutions for single-cell batteries,
multi-cell batteries and battery packs, regardless of
chemistry. Linear offers a broad range of charger solutions, including linear chargers, linear chargers with
regulators, pulse chargers, switchmode monolithic
chargers, switchmode controller chargers, and switchmode smart battery chargers.
Linear Technology Magazine • June 2006
47
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www.linear.com
Linear Technology Magazine • June 2006