DESIGN FEATURES SOT-23 Switching Regulators Deliver Low Noise Outputs in a Small Footprint by Steve Pietkiewicz Introduction L1 4.7µH 100 90 EFFICIENCY (%) As portable electronics designers continue to press for reduction in component sizes, Linear Technology introduces the LT1611 and LT1613 SOT-23 switching regulators. These current mode, constant frequency devices contain internal 36V switches capable of generating output power in the range of 400mW to 2W, in a 5lead SOT-23 package. The LT1613 has a standard positive feedback pin and is designed to regulate positive voltages. The LT1611 has a novel feedback scheme designed to directly regulate negative output voltages without the use of level-shifting circuitry. Boost, single-ended primary inductance converters (SEPIC) and inverting configurations are possible with the LT1613 and LT1611. The high voltage switch allows hard-todo, yet popular DC/DC converter functions like four cells to 5V, 5V to –5V, 5V to –15V or 5V to 15V to be easily realized. Both devices switch at a frequency of 1.4MHz, allowing the use of tiny inductors and capacitors. Many of the components specified for use with the LT1613 and LT1611 are 2mm or less in height, providing a low profile solution. The input voltage range is 1V to 10V, with 2mA quiescent current. In shutdown mode, the quiescent current drops to 0.5µA. The VIN C1 15µF SHDN VIN = 2.8V VIN = 1.5V 50 0 50 100 150 200 250 300 350 400 LOAD CURRENT (mA) TA01a Figure 2. Efficiency of Figure 1’s1613boost converter constant frequency switching produces low amplitude output ripple that is easy to filter, unlike the low frequency ripple typical of pulseskipping or PFM type converters. Internally compensated current mode control provides good transient response. LT1613 Boost Converter Provides a 5V Output Figure 1’s circuit details a boost converter that delivers 5V at 200mA from a 3.3V input. The input can range from 1.5V to 4.5V, making the circuit usable from a variety of input sources, such as a 2- or 3-cell battery, single Li-Ion cell or 3.3V supply. Efficiency, shown in Figure 2, reaches 88% from a 4.2V input. Start-up waveforms from a 3.3V input into a 47Ω load are LT1613 5V to 15V Boost Converter By changing the value of the resistive divider, a 15V supply can be generated in a similar manner to the 5V converter shown in Figure 1. Figure 4 depicts the converter. L1’s value has been changed to 10µH to provide the same di/dt slope with a higher input voltage. The converter delivers 15V at 60mA from a 5V input, at efficiencies up to 85%, as shown in the efficiency graph of Figure 5. LT1613 4-Cell to 5V SEPIC A 4-cell battery presents a unique challenge to the DC/DC converter designer. A fresh battery measures about 6.5V, above the 5V output, while at end of life the battery voltage will measure 3.5V, below the 5V output. Simple switching regulator topologies like boost or buck can only increase or decrease an input voltage, VOUT 5V/200mA R1 374k LT1613 + C2 15µF VOUT 1V/DIV FB GND L1: MURATA LQH3C4R7M24 OR SUMIDA CD43-4R7 C1, C2: AVX TAJA156M010R D1: MOTOROLA MBR0520 VIN = 3.5V 70 D1 SW SHDN 80 60 VIN 3.3V + VIN = 4.2V pictured in Figure 3; the converter reaches regulation in approximately 250µs. The device requires some bulk capacitance due to the internal compensation network used. A 10µF ceramic output capacitor can be used with the addition of a phase-lead capacitor paralleled with R1; this capacitor is typically in the 10pF– 100pF range. R2 121k (814) 237-1431 (847) 956-0666 (803) 946-0362 (800) 441-2447 IL1 500mA/DIV 1613 • TA01 Figure 1. This boost converter steps up a 1.5V to 4.2V input to 5V. It can deliver 250mA from a 3.3V input. Linear Technology Magazine • February 1999 SHDN 5V/DIV 100µs/DIV Figure 3. Boost converter start-up with 3.3V input into a 50Ω load 11 DESIGN FEATURES L1 10µH + VOUT 15V/50mA VIN C1 15µF SHDN R1 1.37M 1% + SW LT1613 SHDN 1nF L1: MURATA LQH3C100 C1: AVX TAJB226M016 C2: AVX TAJA475M025 D1: MOTOROLA MBR0520 EFFICIENCY (%) SW 1M SHDN (800) 441-2447 70 65 60 1613 • TA01 (800) 441-2447 Figure 6. This single-ended primary inductance converter (SEPIC) generates 5V from an input voltage above or below 5V. 200µs, with a maximum perturbation under 200mV. The double trace of VOUT under load in Figure 8 is actually switching ripple at 1.4MHz caused by the ESR of output capacitor C2. A better (lower ESR) output capacitor will decrease the output ripple. 85 80 VIN = 6.5V 75 70 VIN = 3.6V 65 VIN = 5V 60 55 50 0 10 20 30 40 50 60 70 80 90 100 LOAD CURRENT (mA) 1611 TA02 Figure 5. Efficiency of Figure 4’s circuit which will not do the trick in this situation. The solution is a SEPIC. A dual-winding inductor or two separate inductors are required to make this converter. Figure 6 details the circuit. A Sumida CLS62-150 15µH dual inductor is specified in the application, although two 15µH units can be used instead. Up to 125mA can be generated from a 3.6V input. Figure 7’s graph shows converter efficiency, which peaks at 77%. Transient response with a 5mA to 105mA load step is pictured in Figure 8. The converter settles to final value inside LT1611 5V to –5V Inverting Converter 50 0 A low noise –5V output can be generated using an inverting topology with the LT1611. This circuit, shown in Figure 9, bears some similarity to the SEPIC described above, but the output is in series with the second inductor. This results in a very low noise output. The circuit can deliver –5V at up to 150mA from a 5V input, or up to 100mA from a 3V input. Efficiency, described in Figure 10, peaks at 75%. Figure 11 illustrates the start-up waveforms. During startup, the switch-current increases to approximately 1A. At this current, the inductance of the Sumida unit decreases, resulting in the increased VOUT 100mV/DIV AC COUPLED 1611 TA02 ripple current noticeable in the switchcurrent trace of Figure 11. After the circuit has reached regulation, the ripple current decreases by about a factor of two. Switching waveforms with a 100mA load are shown in Figure 12. Output voltage ripple is caused by ripple current in the inductor multiplied by output capacitor ESR. Although the 20mVP-P ripple pictured in Figure 12 is low, significant improvement can be obtained by judicious component selection. Figure 13 details the same 5 to –5V C3 0.22µF L1A 22µH + VIN C1 22µF SHDN 25 50 75 100 125 150 175 200 225 250 LOAD CURRENT (mA) Figure 7. Efficiency of Figure 6’s SEPIC reaches 77%. VIN 5V L1B 22µH SW D1 LT1611 29.4k SHDN VOUT –5V/150mA NFB GND 10k + 105mA 5mA 200µs/DIV Figure 8. SEPIC transient response at 5V input with a 5mA to 105mA load step 12 C2 15µF (847) 956-0666 (803) 946-0362 55 ILOAD VOUT 5V/175mA + 324k L1: SUMIDA CLS62-150 15µH C1, C2: AVX TAJA156M016 C3: X7R CERAMIC D1: MOTOROLA MBR0520 1613 • TA01 VIN = 5V FB GND (814) 237-1431 (803) 946-0362 D1 L1B 15µH LT1613 R2 121k VIN = 6.5V VIN = 3.6V VIN C1 15µF SHDN 85 75 + C2 22µF Figure 4. This 4-cell to 15V boost converter can deliver 50mA from a 3V input. 80 VIN 4V–7V FB GND C3 0.22µF L1A 15µH D1 EFFICIENCY (%) VIN 3V–7V L1: SUMIDA CLS62-220 22µH C1, C2: AVX TAJB226010 C3: X7R CERAMIC D1: MOTOROLA MBR0520 C2 22µF (847) 956-0666 (803) 946-0362 1613 • TA01 (800) 441-2447 Figure 9. This inverting converter delivers –5V at 150mA from a 5V input. Linear Technology Magazine • February 1999 DESIGN FEATURES 85 80 VOUT 2V/DIV VIN = 5V EFFICIENCY (%) 75 70 VIN = 3V ISW 500mA/DIV 65 60 55 VSHDN 5V/DIV 50 0 25 50 75 100 LOAD CURRENT (mA) 125 150 200µs/DIV Figure 11. 5V to –5V inverting converter start-up into a 47Ω load 1611 TA02 Figure 10. 5V to –5V inverting converter efficiency reaches 76%. VOUT 200mV/DIV AC COUPLED converter function with better output capacitors. Now, output ripple measures just 4mV P-P. Additionally, transient response is improved by the addition of phase lead capacitor C5. Figure 14 depicts load transient response of a 25mA to 125mA load step. Maximum perturbation is under 30mV and the converter reaches final value in approximately 250µs. It is important to take notice of how Figures 9 and 13 are drawn. D1’s cathode is returned to the LT1611’s GND pin before both connect to the ground plane. This connection combines the current of the switch and diode, which conduct on alternate phases. The summation of both currents equals a current with no abrupt changes, minimizing di/dt induced voltages caused by the few nanohenries of inductance in the ground plane. This summed current is then depos- ISW 100mA/DIV VSW 10V/DIV 100ns/DIV Figure 12. Switching waveforms of inverting converter with 100mA load ited into the ground plane. If this technique is not followed, 100mV spikes can appear at the converter output (I speak from experience: my first several breadboards had this problem). Many systems, such as personal computers, have a 12V supply available. Although the LT1611 VIN pin 5V OR 12V (SEE TEXT) has a 10V maximum, the 36V switch allows a 12V supply to be used for the inductor while the LT1611’s VIN pin is still driven from 5V, as indicated in Figure 13. Significantly more output power can be obtained in this manner, as illustrated in the efficiency graph of Figure 15. continued on page 23 L1A 22µH C2 0.22µF VIN 5V D1 VIN C1 22µF SHDN SW LT1611 SHDN VOUT –5V/150mA 29.4k C5 2.2nF NFB GND 10k C3 4.7µF + + L1B 22µH VOUT 20mV/DIV AC COUPLED C4 68µF ILOAD L1: SUMIDA CLS62-220 22µH C1: AVX TAJB226010 C2: X7R CERAMIC C3: Y5V CERAMIC C4: SANYO POSCAP 10TPC68M D1: MOTOROLA MBR0520 (847) 956-0666 (803) 946-0362 125mA 25mA 1613 • TA01 (619) 661-6835 (800) 441-2447 Figure 13. Low noise inverting converter; component selection and feedforward capacitor C5 reduce noise to 4mVP-P. Linear Technology Magazine • February 1999 200µs/DIV Figure 14. Transient response of low noise inverting converter is under 30mV for a 25mA to 125mA load step. Steady-state output ripple is 4mVP-P. 13 DESIGN FEATURES causes distortion) by making the glitch impulse both ultralow and uniform with code. Op Amp Selection Considerations A significant advantage of the LTC1597 is the ability to choose the I-to-V output op amp to optimize system accuracy, speed, power and cost. Table 1 shows a sampling of op amps and their relevant specifications for this application. The LTC1597 is designed to minimize the sensitivity of INL and DNL to op amp offset; this sensitivity has been greatly reduced compared to that of competing multiplying DACs. Figure 10 summarizes the effects of op amp offset for both modes of operation. Note that the bipolar LSB size is twice its unipolar counterpart. As Figure 10 shows, op amp offset has a minimal effect on DAC linearity; it merely shifts the end points. LT1611/LT1613, continued from page 13 The amplifier’s input bias current, which flows through the feedback resistor, adds to the output offset voltage. The amplifier’s finite DC openloop gain also degrades accuracy. The DAC gain error is inversely proportional to the open-loop gain and feedback factor of the op amp. In unipolar mode at full-scale the feedback factor is 0.5; for a 0.2LSB of gain error (REF = 10V) at 16 bits, the openloop amplifier gain should be greater than 650,000. The op amp’s input voltage and current noise also limit DC accuracy. Noise effects accuracy similarly to voltage and current offsets and adds in an RMS fashion. As with any precision application, and with wide bandwidth amplifiers in particular, the noise bandwidth should be minimized with a filter on the output of the op amp to maximize resolution. VIN 80 + SHDN 70 Wherever system requirements demand true 16-bit accuracy over temperature, the LTC1597 provides the best solution. The LTC1597 has outstanding 1LSB linearity over temperature, ultralow glitch impulse, on-chip 4-quadrant resistors, low power consumption, asynchronous clear and a versatile parallel interface.Combined with the LT1468 op amp, the LTC1597 provides the best in its class, 1.7µs settling time to 0.0015%, while maintaining superb DC linearity specifications. SW D1 L1B 15µH LT1611 68.1k SHDN VOUT –10V/60mA NFB GND 10k 65 + EFFICIENCY (%) 75 C1 22µF Conclusion: C2 0.22µF L1A 15µH VIN 3.6V–7V 85 Referring to Table 1, the LT1001 provides excellent DC precision, low noise and low power dissipation. The LT1468 provides the optimum solution for applications requiring DC precision, low noise and fast 16-bit settling. C3 6.8µF 60 55 50 0 50 100 150 200 250 LOAD CURRENT (mA) 300 350 1611 TA02 Figure 15. 12V supply at L1A increases efficiency to 81% and output current to 350mA. 85 VIN = 6.5V EFFICIENCY (%) 75 VIN = 3.6V (847) 956-0666 (803) 946-0362 1613 • TA01 (800) 441-2447 Figure 16. 4-Cell to –10V inverting converter delivers 75mA from a 4V input. LT1611 4-Cell to –10V Inverting Converter A –10V low noise output can be generated in a similar manner as the –5V circuit described above. Figure 16’s circuit can deliver –10V at up to 60mA from a 3.6V input. Efficiency, graphed in Figure 17, reaches a high of 78%. 80 70 L1: SUMIDA CL562-150 C1: AVX TAJB226M010 C2: X7R CERAMIC C3: AVX TAJA685M016 D1: MOTOROLA MBR0520 VIN = 5V 65 60 Conclusion 55 The flexibility of individually controlled outputs in multiple-supply applications can make several LT1611/ LT1613 converters attractive compared to a multiple-output flyback 50 0 25 50 75 100 LOAD CURRENT (mA) 125 150 1611 TA02 Figure 17. 4-cell to –10V converter efficiency Linear Technology Magazine • February 1999 design with one large switching regulator and a custom transformer. Changing an output voltage on a multiple output flyback requires changing the transformer turns ratio, hardly a simple task. Conversely, individual control of each output, using the multiple LT1611/LT1613 approach, provides for complete control of each output voltage as well as supply sequencing. The LT1611 and LT1613 SOT-23 switchers provide small, low noise solutions to power generation needs in tight spaces. 23