DESIGN FEATURES New UltraFast Comparators: Rail-toRail Inputs and 2.4V Operation Allow Use on Low Supplies by Glen Brisebois Introduction The new LT1711 family of UltraFast comparators has fully differential railto-rail inputs and outputs and operates on supplies as low as 2.4V, allowing unfettered application on low voltages. The LT1711 (single) and LT1712 (dual) are specified at 4.5ns of propagation delay and 100MHz toggle frequency. The low power LT1713 (single) and LT1714 (dual) are specified at 7ns of propagation delay and 65MHz toggle frequency. All of these comparators are fully equipped to support multiple-supply applications, and have latch-enable pins and complementary outputs like the popular LT1016, LT1671 and LT1394. They are available in MSOP and SSOP packages, fully specified over commercial and industrial temperature ranges on 2.7V, 5V and ±5V supplies. rent sources and sinks feeding the NPN and PNP differential pairs formed by Q3–Q4 (protected by fast diodes D11–D12) and Q1–Q2 (protected by D1–D2). This approach makes the inputs truly fully differential and noninteracting, unlike approaches that resort to resistors and diode clamps. Even with the inputs at opposite rails, the input bias currents are still a simple function of the input transistor base currents and remain in the µA region. Both input stages feed the level shifting transistors Q5– Q6, and the remainder of the differential voltage gain circuit flows with a delightful symmetry towards the output. Note that the channels are identical, with polarity yet unassigned, and are therefore interchangeable in layout. The symmetry, broken only by the latch-enable circuit, is enhanced by the fact that all of the transistors are well matched, complementary 6GHz fT BJTs. Each output stage ends in two Bakerclamped common emitter transistors, Circuit Description Figure 1 shows a simplified schematic of the LT1711 through LT1714. The front end consists of eight cur- allowing full rail-to-rail output swing. All the comparators guarantee full 5V TTL output capability over temperature, even when supplied with only 3V. Output rise and fall times are fast, at 2ns for the LT1711 and LT1712 and 4ns for the LT1713 and LT1714. Jitter is among the lowest for any monolithic comparator, at 11psRMS for the LT1711 and LT1712 and 15psRMS for the LT1713 and LT1714. Some Applications Simultaneous Full-Duplex 75MBaud Interface with Only Two Wires The circuit of Figure 2 shows a simple, fully bidirectional, differential 2-wire interface that gives good results to 75MBaud, using the low power LT1714. Eye diagrams under conditions of unidirectional and bidirectional communication are shown in Figures 3 and 4. Although not as pristine as the unidirectional VCC I1 I2 I9 R1 D4 R2 D35 Q5 Q6 Q10 D15 GND VCC Q34 Q22 Q13 D38 D14 Q4 OUTB1 I5 Q3 D9 D10 D11 Q8 Q42 D18 I14 Q14 R3 I6 I7 R4 D39 Q41 Q15 I11 Q7 D12 GND D33 OUTA1 D37 D13 D32 D36 Q35 D30 VEE VCC VCC Q29 Q23 Q17 BIAS INB1 VCC Q30 Q28 Q16 Q12 D31 R9 Q19 Q9 VEE VEE VCC Q31 R8 Q11 Q2 VEE Q1 Q18 LE1 D29 I25 I18 D34 D28 INA1 R6 I13 VCC I10 I4 R5 I12 VEE VCC D3 I24 I23 I3 D2 D1 D27 D17 R11 Q43 R12 I16 I8 VEE GND Q40 I15 I21 I22 I20 Figure 1. LT1711–LT1714 simplified schematic Linear Technology Magazine • February 2001 21 DESIGN FEATURES 3V 4 + 14 2 2 3V 1/2 LT1714 RXD ALL DIODES = BAV99 – 13 16 1 – 3 R2a 2.55k 3V 15 + 11 R3a 124Ω 1/2 LT1714 TXD 8 – 6 10 12 9 R3b 124Ω R2b 2.55k R1b 499Ω RXD 16 13 3 R1a 499Ω R1c 499Ω R2c 2.55k R0a 140Ω R0b 140Ω R3c 124Ω 3V 5 5 7 14 1/2 LT1714 1 15 3V 4 + 3V SIX FEET TWISTED PAIR ZO 120Ω + 11 7 TXD 1/2 LT1714 R3d 124Ω R1d 499Ω 12 – 6 8 10 9 R2d 2.55k Figure 2. 75MBaud full-duplex interface on two wires per for mance of Figure 3, the performance under simultaneous bidirectional operation is still excellent. Because the LT1714 input voltage range extends 100mV beyond both supply rails, the circuit works with a full ±3V of ground potential difference. The circuit works well with the resistor values shown, but other sets of values can be used. The starting point is the characteristic impedance, ZO, of the twisted-pair cable. The input impedance of the resistive network should match the characteristic impedance and is given by: RIN = 2 • RO • (R1||(R2 + R3) (RO + 2 • (R1||(R2 + R3))) This comes out to 120Ω for the values shown. The Thevenin equivalent source voltage is given by: 5ns/DIV 5ns/DIV Figure 3. Performance of Figure 1’s circuit operated unidirectionally; eye is wide open (cursors show bit interval of 13.3ns or 75MBaud). Figure 4. Performance of Figure 1’s circuit operated simultaneous-bidirectionally; crosstalk appears as noise. Eye is slightly shut but performance is still excellent. C4 100pF R5 7.5k V • (R2 + R3 – R1) VTh = S • (R2 + R3 + R1) R6 162Ω 22 C3 100pF R7 15.8k VS RO (RO + 2 • (R1||(R2 + R3))) This amounts to an attenuation factor of 0.0978 with the values shown. (The actual voltage on the lines will be cut in half again due to the 120Ω ZO.) This attenuation factor is important because it is the key to deciding the ratio of the R2, R3 resistor divider in the receiver path. This divider allows the receiver to reject the large signal of the local transmitter and instead sense the attenuated signal of the remote transmitter. Note The author having already designed R2 + R3 to be 2.653kΩ (by allocating input impedance across RO, R1, and R2 + R3 to get the requisite 120Ω), R2 and R3 then become 2529Ω and 123.5Ω, respectively. The nearest 1% value for R2 is 2.55k and that for R3 is 124Ω. that in the above equations, R2 and R3 are not yet fully determined because they only appear as a sum. This allows the designer to now place an additional constraint on their values. The R2, R3 divider ratio should be set to one-half of the attenuation factor mentioned above or R3/R2 = 1/2 • 0.0976.1 2 – R9 2k 3 + C2 0.1µF R8 2k VS R1 1k 2 R2 1k R4 210Ω VS 6 SINE VS = 2.7V TO 6V 1 + 7 LT1713 3 LT1806 4 1MHz AT CUT VS 7 – 4 SQUARE 8 5 6 C1 0.1µF R3 1k Figure 5. LT1713 comparator configured as a series-resonant crystal oscillator; the LT1806 op amp is configured as a bandpass filter with a Q of 5 and fC of 1MHz. Linear Technology Magazine • February 2001 DESIGN FEATURES 1MHz Series-Resonant Crystal Oscillator with Square and Sinusoid Outputs Figure 5 shows a classic 1MHz seriesresonant crystal oscillator. At series resonance, the crystal is a low impedance and the positive feedback connection brings about oscillation at the series resonance frequency. The RC feedback to the – input ensures that the circuit does not find a stable DC operating point and refuse to oscillate. The comparator output is a 1MHz square wave (top trace of Figure 6), with jitter measured at 28psRMS on a 5V supply and 40 psRMS on a 3V supply. At pin 2 of the comparator, on the other side of the crystal, is a clean sine wave except for the presence of the small high frequency glitch (middle trace of Figure 6). This glitch is caused by the fast edge of the comparator output feeding back through crystal capacitance. Amplitude stability of the sine wave is maintained by the is the bottom trace of Figure 6. Distortion was measured at –70dBc and –55dBc on the second and third harmonics, respectively. A 3V/DIV B 1V/DIV Conclusion C 1V/DIV 200ns/DIV Figure 6. Oscillator waveforms with VS = 3V: Trace A = comparator output; Trace B = crystal feedback to pin 2 of the LT1713; Trace C = buffered, inverted and bandpass filtered output of LT1806 fact that the sine wave is basically a filtered version of the square wave. Hence, the usual amplitude-control loops associated with sinusoidal oscillators are not necessary.2 The sine wave is filtered and buffered by the fast, low noise LT1806 op amp. To remove the glitch, the LT1806 is configured as a bandpass filter with a Q of 5 and unity gain center frequency of 1MHz. The final sinusoidal output The fully differential rail-to-rail inputs of the new LT1711 family of fast comparators make them useful across a wide variety of applications. The high speed, low jitter performance of this family, coupled with their small package sizes and 2.4V operation, makes them attractive where PCB real estate is at a premium and bandwidth-topower ratios must be optimized. 1 Using the design value of R2 + R3 = 2.653k rather than the implementation value of 2.55k + 124Ω = 2.674k. 2 Amplitude will be a linear function of comparator output swing, which is supply dependent and therefore adjustable. The important difference here is that any added amplitude stabilization or control loop will not be faced with the classical task of avoiding regions of nonoscillation vs clipping. LT1618, continued from page 7 L1 10µH VIN 2.7V TO 5V D1 0.619Ω 80mA 90 85 9 10kHz TO 50kHz PWM BRIGHTNESS ADJUST 8 7 VIN SW SHDN ISP ISN R3 5.1k 4 VIN = 5V 80 3 2 R1 2M LT1618 IADJ FB VC GND 5 1 C2 1µF CC 0.1µF Linear Technology Magazine • February 2001 VIN = 2.8V 65 50 51Ω 51Ω 51Ω 51Ω 10 20 30 40 50 60 70 80 LED CURRENT (mA) Figure 10. High power white LED driver efficiency Figure 9. High power white LED driver For larger LCD displays where a greater amount of light output is needed, multiple strings of LEDs can be driven in parallel. When driving parallel strings, ballast resistors should be added to compensate for LED forward voltage variations. The amount of ballasting needed depends on the LEDs used and how well they 70 55 R2 121k (408) 573-4150 (408) 573-4150 (800) 282-9855 (847) 956-0666 High Power White LED Driver VIN = 3.3V 75 60 10 C3 0.1µF C1: TAIYO YUDEN JMK212BJ475 C2: TAIYO YUDEN TMK316BJ105 D1: ON SEMICONDUCTOR MBR0530 L1: SUMIDA CR43-100 EFFICIENCY (%) C1 4.7µF are matched. The circuit in Figure 9 is ideal for larger displays, providing constant current drive for twenty white LEDs from a single Li-Ion cell. Efficiency reaches a respectable 82%, as seen in Figure 10. Conclusion The constant-current/constant-voltage operation of the LT1618 makes the device an ideal choice for a variety of constant-current designs. The device provides accurate output current regulation or input current limiting, along with excellent output voltage regulation. With a wide input voltage range and the ability to produce outputs up to 35V, the LT1618 works well in many different applications. References 1. Kim, Dave. “Tiny Regulators Drive White LED Backlights.” Linear Technology Design Note 231 (May 2000). 23