LINEAR TECHNOLOGY FEBRUARY 2001 IN THIS ISSUE… COVER ARTICLE New 5-Lead SOT-23 Oscillator is Small, Very Stable and Easy to Use .................................................... 1 Andy Crofts Issue Highlights ........................... 2 VOLUME XI NUMBER 1 New 5-Lead SOT-23 Oscillator is Small, Very Stable and Easy to Use by Andy Crofts LTC® in the News .......................... 2 DESIGN FEATURES Current-Limited DC/DC Converter Simplifies USB Power Supplies ..... 6 Bryan Legates 2.3MHz Monolithic, Continuous Time, Differential Lowpass Filter Provides Solutions for Wide Band CDMA Applications ....................... 8 Nello Sevastopoulos and Mike Kultgen Very Low Cost Li-Ion Battery Charger Requires Little Area and Few Components ........................ 12 David Laude Synchronous Buck Controller Extends Battery Life and Fits in a Small Footprint ......................... 13 Peter Guan New No RSENSE™ Controllers Deliver Very Low Output Voltages .......... 16 Christopher B. Umminger New UltraFast™ Comparators: Rail-to-Rail Inputs and 2.4V Operation Allow Use on Low Supplies ..................................... 21 Glen Brisebois High Efficiency Synchronous PWM Controller Boosts 1V to 3.3V or 5V .................................................. 24 San-Hwa Chee DESIGN INFORMATION Rail-to-Rail 14-Bit Dual DAC in a Space Saving 16-Pin SSOP Package .................................................. 28 Introduction Enter the LTC1799 Generating a periodic waveform of arbitrary frequency is not always a trivial task. Low cost RC oscillators can be built using discrete components such as comparators, resistors and capacitors, or by using simple integrated circuits such as the industry-standard 555 timer in conjunction with several discrete components. These solutions are bulky and inaccurate, especially at frequencies above a few hundred kilohertz. Very accurate oscillators with a predetermined frequency may be realized using either crystals or ceramic resonators as stable frequency elements; crystal oscillators offer the highest perfor mance, although they are costly. These circuits are also bulky, sensitive to acceleration forces and tend to be less robust than RC oscillators. Generating various frequencies from a single crystal or ceramic oscillator requires additional circuitry that will add to the component list and consume PC board space. The LTC1799 offers an alternative that combines the frequency stability and accuracy of a ceramic resonator with the flexibility and ease of use of an RC oscillator, while requiring less space than either. The LTC1799 is the only oscillator IC that can accurately generate a square wave signal at any frequency from 5kHz to 20MHz without the use of a crystal, ceramic element or existing clock reference. A complete oscillator circuit requires only an LTC1799, a frequency-setting resistor (RSET) and a bypass capacitor, as illustrated in Figure 1. With a 0.1% resistor, the frequency accuracy is typically better than ±0.6%. The LTC1799’s internal master oscillator is a resistance to frequency converter with an output range of 500kHz to 20MHz. A programmable on-chip frequency divider divides the frequency by 1, 10 or 100, extending the frequency range to greater than three decades (5kHz to 20MHz). DESIGN IDEAS ............................................ 29–37 complete list on page 29 1 2 RSET 20k 0.1% 5MHz ±1.6%* (27°C) LTC1799 3V Hassan Malik continued on page 3 C1 0.1µF 3 VCC OUT 5 GND SET DIV 4 New Device Cameos ................... 38 Design Tools ............................... 39 Sales Offices .............................. 40 *INCLUDING ERROR CONTRIBUTION FROM RESISTOR Figure 1. A complete oscillator solution , LTC and LT are registered trademarks of Linear Technology Corporation. Adaptive Power, Burst Mode, C-Load, DirectSense, FilterCAD, Hot Swap, LinearView, Micropower SwitcherCAD, No Latency ∆Σ, No RSENSE, Operational Filter, OPTI-LOOP, Over-The-Top, PolyPhase, PowerSOT, SwitcherCAD and UltraFast are trademarks of Linear Technology Corporation. Other product names may be trademarks of the companies that manufacture the products. EDITOR’S PAGE Issue Highlights Happy New Year and welcome to our first issue of 2001, beginning the eleventh year of Linear Technology magazine. Our lead article in this issue introduces the LTC1799, a new oscillator IC in a SOT-23 package. The LTC1799 offers the frequency stability and accuracy of a ceramic resonator with the flexibility and ease of use of an RC oscillator, while requiring less space than either. It is the only oscillator IC that can accurately generate a square wave at any frequency from 5kHz to 20MHz without the use of a crystal, ceramic element or existing clock reference. A complete circuit requires only an LTC1799, a frequency-setting resistor and a bypass capacitor. With a 0.1% resistor, the frequency accuracy is typically better than ±0.5%. In the filter realm, we present the LTC1566-1, a new monolithic 7th order continuous time lowpass filter featuring differential input and output terminals. The LTC1566-1 operates from a single 5V supply and dual supplies of up to ±5V, comes in an SO-8 package and requires no external components other than power supply decoupling capacitors. The filter is designed to have a flat passband from DC to 2MHz and a steep transition band. The filter cutoff is set at 2.3MHz to accommodate differential filtering needs in wideband CDMA base stations. Also introduced in this issue is the new LT1711/12/13/14 family of UltraFast comparators, which has fully differential rail-to-rail inputs and outputs and operates on supplies as low as 2.4V, allowing unfettered application on low voltages. The LT1711 (single) and LT1712 (dual) are specified at 4.5ns of propagation delay and 100MHz toggle frequency. The low power LT1713 (single) and LT1714 (dual) are specified at 7ns of propagation delay and 65MHz toggle frequency. As usual, this issue introduces a variety of new power products. The 2 newly released LT1618 DC/DC converter provides USB devices with accurate input current control. Several requirements must be met by a device powered from the USB connector: input capacitors smaller than 10µF are required to minimize inrush currents at plug-in; upon power-up, the device must draw less than 100mA of current from the USB bus and can increase its input current to 500mA only when given permission by the USB controller. These requirements can be easily met using the LT1618. The part combines a traditional voltage feedback loop with a unique current feedback loop to operate as a constant-current, constant-voltage source. The LTC1734 is a precision, low cost, single-cell, linear Li-Ion battery charger with constant voltage and constant current control. The small quantity and low cost of external components results in a very low overall system cost and the part’s 6-pin SOT23 package allows for a compact design solution. Previous products usually required an external current sensing resistor and blocking diode whose functions are included in the LTC1734. The LTC1773 is a synchronous DC/DC controller that packs high output current capability and low operating quiescent current in a small MSOP-10 package. Its input voltage range is from 2.65V to 8.5V, ideal for 1- or 2-cell Li-Ion batteries as well as 3- to 6-cell NiCd and NiMH battery packs. A precise internal undervoltage lockout circuit prevents deep discharge of the battery below 2.5V. The LTC1773’s high operating frequency of 550kHz allows the use of small, surface mount components to provide a compact power supply solution. The LTC1778 and the LTC3711 are buck regulators that deliver the low output voltages and high efficiencies required by today’s portable supplies. The LTC1778 is a step-down controller that provides synchronous LTC in the News… On January 16, Linear Technology Corporation announced its financial results for the 2nd quarter of fiscal year 2001. Robert H. Swanson, Chairman & CEO, stated, “Once again we had a strong financial performance. We achieved record levels of sales and profits with sales increasing 11% and profits 12% sequentially from the September quarter. Our return on sales was a record 44.4%. As with other semiconductor companies, we have seen a slowdown in net bookings from the very robust levels experienced in previous quarters. Interestingly, gross bookings in this quarter still exceeded our net billings, however cancellations of some bookings from previous quarters caused our net bookings to be slightly less than our net billings. Nevertheless, our backlog continues to be strong and demand for product in the near term appears to be firm. Consequently, we currently estimate that we will grow sales and profits sequentially in the 8%–10% range for the March quarter. The Company reported sales of $258,450,000, and net income of $114,758,000 for the 2nd quarter. Net sales were up 59% over the same quarter last year. Diluted earnings per share were $0.34, an increase of 77% over the 2nd quarter last year. drive for two external N-channel MOSFET switches. Its true current mode architecture has an adjustable current limit, can be easily compensated, is stable with ceramic output capacitors and does not require a sense resistor. The LTC1778 operates on input voltages from 4V to 36V and output voltages from 0.8V up to 90% of VIN. Switching frequencies up to nearly 2MHz can be chosen, allowing wide latitude in trading off efficiency for component size. The LTC3711 is essentially the same as the LTC1778 but includes a 5-bit VID interface. continued on page 5 Linear Technology Magazine • February 2001 DESIGN FEATURES Divider Setting DIV (Pin 4) Connection LTC1799 5V Table 1. Frequency range vs divider setting 1 Frequency Range ÷1 GND > 500kHz* ÷10 Floating 50kHz to 1M Hz ÷100 V+ ≤ 100kHz 2 RT 100k THERMISTOR C1 0.1µF RT: YSI 44011 3 OUT VCC 5 OUT GND DIV SET 4 (800) 765-4974 *At frequencies above 10MHz (RSET <10k), the LTC1799 may suffer reduced accuracy on supplies less than 4V. Figure 4. Temperature-to-frequency converter LTC1799, continued from page 1 typically draws 1mA of supply current. Figure 1 shows a circuit that generates a precision 5MHz signal. LTC1799: Advantages in Precision, where N is the on-chip divider setting Resolution and Size fOSC = 10MHz • 10kΩ/(N • RSET) OUTPUT FREQUENCY ERROR (%) of 1, 10 or 100, depending on the state of the DIV pin. A proprietary feedback loop maintains this accurate relationship over all operating conditions, providing a temperature coefficient that is typically less than ±0.004%/°C. The LTC1799 operates over a 2.7V to 5.5V supply range, with a voltage coefficient of 0.05%/V. It 3.5 3.0 2.5 2.0 1.5 1.0 0.5 0 –0.5 –1.0 –1.5 –2.0 –2.5 –3.0 –3.5 TA = 27°C WORST-CASE HIGH TYPICAL HIGH WORST-CASE LOW 1k 10k TYPICAL LOW 100k 1M RSET (Ω) GUARANTEED LIMITS APPLY TO 5k TO 200k ONLY Figure 2. Accuracy of the output frequency equation NORMALIZED OUTPUT FREQUENCY DRIFT (%) 2.0 WORST-CASE HIGH 1.5 1.0 TYPICAL HIGH 0.5 0.0 –0.5 –1.0 –1.5 TYPICAL LOW RSET = 31.6k WORST-CASE LOW –2.0 –40 –30 –20 –10 0 10 20 30 40 50 60 70 80 90 TEMPERATURE (°C) Figure 3. Output frequency temperature drift Linear Technology Magazine • February 2001 With a frequency tolerance 0.5% typical and 1.5% worst-case, the performance of the LTC1799 is similar to that of ceramic resonators and vastly superior to oscillators that use discrete resistors and capacitors. Its stingy temperature and voltage coefficients (typically ±0.004%/°C and 0.05%/V, respectively) maintain accuracy over all operating conditions. Unlike oscillators using crystals, the LTC1799 has infinite frequency resolution; the output frequency can be set to any value in the 5kHz to 20MHz range. The programmed frequency is limited only by the choice of RSET. This feature allows the clock frequency to be changed late in a design cycle by changing the value of a resistor instead of stocking crystals in many different frequencies. The LTC1799’s SOT-23 package and low component count (one resistor, one capacitor) result in an efficient use of PCB space, requiring less space than any crystal, ceramic resonator or discrete oscillator solution. LTC1799 includes a programmable frequency divider. The DIV input pin may be connected to GND to pass the master oscillator output directly to the OUT pin. When the DIV pin is left floating, the LTC1799 divides the master oscillator frequency by 10 before driving OUT. Connect DIV to V+ to divide the master oscillator by 100 to generate frequencies below 100kHz. Table 1 suggests the proper DIV pin setting for the desired frequency. The frequency ranges overlap near 100kHz and 1MHz, allowing a choice of settings. Since the supply current increases with smaller values of RSET, the lower divider setting is usually preferred. Once the divider setting has been selected, calculate the proper resistor value using this simple equation: RSET = 10kΩ • 10MHz/(N • fOSC) Since the oscillator frequency, fOSC, is dependent on the resistor value, RSET, any error in the resistor will create error in fOSC. Performance Rivals Ceramic Resonators The LTC1799 obeys its frequency vs RSET equation within 1.5% at room temperature with any RSET from 10k 1400 MAX Frequency Set by Single Resistor and Ranged by an Internal Frequency Divider The heart of the LTC1799 is a master oscillator that performs a precise resistance-to-frequency conversion. RSET can be any value from 3.32k to 1M, generating master oscillator frequencies between 30MHz and 1kHz with guaranteed 1.5% accuracy for resistors between 5k and 200k. To extend its frequency range, the 1200 FREQUENCY (kHz) Selecting the proper resistor is straightforward because the LTC1799 follows a simple relationship between RSET and frequency: MIN 1000 800 600 TYP 400 200 0 –20 –10 0 10 20 30 40 50 60 70 80 90 TEMPERATURE (°C) Figure 5. Output frequency vs temperature for Figure 4’s circuit 3 DESIGN FEATURES U1 LTC1799 1 3V 2 RSET C1 0.1µF 3 VCC 5 OUT GND SET 4 DIV U3 LTC1067-50 3V C4 1µF U2 74HC4520 SW1 1 3V 2 16 10 C2 0.1µF 7 8 9 15 CLOCK A Q1A ENABLE A Q2A VDD Q3A ENABLE B Q4A RESET A Q1B VSS Q2B CLOCK B Q3B RESET B Q4B 3 ÷2 4 ÷4 5 ÷8 6 ÷16 11 ÷32 12 ÷64 1 C3 0.1µF 2 3 R61 10k 4 R51 5.11k 5 R31 51.1k R11 100k 6 7 R21 20k 8 13 ÷128 14 ÷256 V+ CLK NC AGND + V V – SA SB LPA LPB BPA BPB HPA/NA INV A HPB/NB INV B 16 15 14 R62 14k 13 R52 5.11k 12 11 R32 51.1k OUT 10 9 R22 20k RH1 249k VSQUARE RL1 51.1k Figure 6. 80Hz to 8kHz sine wave generator to 200k for a frequency range of 5kHz to 10MHz. With a 5V supply, this range is extended to resistors as low as 5k, for frequencies up to 20MHz. Figure 2 shows the frequency deviation from the equation over the range of possible values for RSET. Figure 3 shows the output frequency variation over the industrial temperature range. Applications Temperature-to-Frequency Converter In Figure 4, the frequency-setting resistor is replaced by a thermistor to create a temperature-to-frequency converter. The thermistor resistance is 100k at 25°C, 333k at 0°C and 16.3k at 70°C, a span that fits nicely in the LTC1799’s permitted range for RSET. With its low tempco and high linearity, the LTC1799 adds less than ±0.5°C of error to the output frequency. Figure 5 plots the typical and worst case output frequency vs temperature (error due to the thermistor is not shown). 80Hz to 8kHz Sine Wave Generator Figure 6 shows the LTC1799 providing both the clock source and the input to a switched capacitor filter to generate a low distortion sine wave output. The 74HC4520 counter divides the frequency by 64 before driving the filter with a square wave. An ideal square wave will have only odd harmonics. The LTC1067-50 filter building block is configured as a lowpass filter with a stopband notch at the third harmonic of the desired sine-wave frequency. The fifth and higher-order harmonics are attenuated by 60dB or greater. The resulting sine wave has less than 0.1% distortion. This design can generate any tone from 78Hz (the LTC1799 minimum output frequency of 5kHz/64) to 8kHz, limited by maximum clocking frequency of the LTC1067-50 at a 3V supply. Figure 7 shows a scope capture for a 1kHz tone (RSET = 158kΩ). Digital Frequency Control Figure 8 shows the details of an LTC1799 controlled by a 12-bit voltage output D/A converter. Since the LTC1799 is a resistance-to-frequency converter, the input voltage between VCC and SET must be measured and used to create a current. Therefore, the DAC and op amps create a digitally controlled resistor between VCC and SET. Figure 9 shows the measured output frequency vs input code. The linearity is excellent except at the endpoints; the low frequency accuracy VSQUARE OUT Figure 7. Scope capture for a 100kHz tone (RSET = 158k) 4 Linear Technology Magazine • February 2001 DESIGN FEATURES 3V C3 0.1µF U3 LTC1659 CLK DIN CS/LD 1 2 3 4 CLK VCC DIN VOUT CS/LD REF DOUT GND 8 7 3V 6 5 C2 0.1µF R5 10k 3 3V R6 10k R1 10k 10 R2 10k 9 R7 10k 2 4 + U2A 1/4 LT1491 – 1 5kHz TO 85kHz U1 LTC1799 11 + U2C 1/4 LT1491 1 3V 2 8 – R8 R9 20k 10k R4 10k R3 10k 7 RS 10k + U2B 1/4 LT1491 – C1 0.1µF 3 VCC OUT 5 OUT GND SET DIV 4 3V 5 6 Figure 8. Digitally controlled oscillator with 5kHz to 85kHz range 100 Conclusion The LTC1799 is a tiny, accurate, easyto-use oscillator that is programmed by a single resistor. With a typical frequency accuracy of better than 0.5% and low temperature and supply dependence, the LTC1799 provides performance that approaches that of crystal oscillators and ceramic resonators without sacrificing PCB space. Furthermore, the output frequency has unlimited resolution because it is resistor -programmable. With its resistance-to-frequency conversion architecture, the LTC1799 delivers an unprecedented combination of simplicity, stability, precision, frequency range and resolution in a tiny SOT-23 package. 75 fOUT (kHz) is limited by op amp offset and gain errors, while the highest frequency is limited by the op amp’s maximum output voltage. 50 25 0 0 1024 2048 DAC CODE 3072 4096 Figure 9. Input code vs output frequency for Figure 8’s circuit Issue Highlights, continued from page 2 The LTC1700 synchronous PWM controller boosts input voltages as low as 0.9V to 3.3V or 5V. It uses a constant frequency, current mode PWM architecture but does not require a current sense resistor; instead, it senses the VDS across the external N-channel MOSFET. This reduces component count and improves high load current efficiency. The LTC1700 offers high efficiency over the entire load current range. During continuous mode operation, the LTC1700 Linear Technology Magazine • February 2001 consumes 540µA; it drops to 180µA in Sleep mode. In shutdown, the quiescent current is just 10µA. Our Design Information section introduces the LTC1654, a 14-bit railto-rail voltage output dual DAC in the 16-pin SSOP package. This part offers a convenient solution for applications where density, resolution and power are critical. The LTC1654 is guaranteed to be 14-bit monotonic over temperature with a typical differential nonlinearity of only 0.3LSB. Our Design Ideas section features a number of novel circuits, including a white LED driver, a 48V Hot Swap™ circuit with reverse-battery protection, a collection of circuits using a dual DAC to adjust gain and phase in RF applications, a high current, multioutput PolyPhase™ supply for computer and networking applications and an ultralow noise 48V to 5V step-down converter. The issue concludes with a trio of New Device Cameos. 5 DESIGN FEATURES Current-Limited DC/DC Converter Simplifies USB Power Supplies by Bryan Legates Introduction currents when the device is plugged into the USB port; when first plugged in, the device must draw less than 100mA from the port and, for high power devices, the current drawn from the port can increase to 500mA only after it is given permission to do so by the USB controller. These requirements can be easily met using the Many portable Universal Serial Bus (USB) devices power themselves from the USB host or hub power supply when plugged into the USB port. Several requirements must be met to ensure the integrity of the bus: the USB specification dictates that the input capacitance of a device must be less than 10µF to minimize inrush L1 10µH C1 4.7µF 3 VOUT 12V 2 7 ISN SW R1 909k ISP LT1618 8 3.3V OFF ON 0V 90 D1 9 1 FB VIN C2 4.7µF R2 107k SHDN IADJ GND 20k 4 VC 5 13k 0 20 40 60 80 100 120 140 160 LOAD CURRENT (mA) 10nF Figure 2. USB to 12V boost efficiency (408) 573-4150 (602) 244-6600 (847) 956-0667 3 C3 0.47µF L1 10µH 0.1Ω C1 4.7µF VOUT 2V/DIV 8 9 20k D1 VOUT 5V 2 7 ISN SW L2 10µH ISP LT1618 5 VC C2 10µF Figure 4. USB to 5V SEPIC during start-up 10 2k 13k 1ms/DIV R2 107k SHDN IADJ GND IIN 50mA/DIV R1 316k 1 FB VIN 4 3.3V 100mA 500mA 0V 70 65 Figure 1. USB to 12V boost converter with selectable 100mA/500mA current limit 3.3V OFF ON 0V 75 10 2k C1: TAIYO YUDEN JMK212BJ475 C2: TAIYO YUDEN EMK316BJ475 D1: ON SEMICONDUCTOR MBR0520 L1: SUMIDA CR43-100 VIN 5V 80 60 3.3V 100mA 500mA 0V IIN 85 EFFICIENCY (%) 0.1Ω VIN 5V LT1618 DC/DC converter, which provides an accurate input current control ideal for USB applications. The LT1618 combines a traditional voltage feedback loop with a unique current feedback loop to operate as a constant-current, constant-voltage source. VOUT 2V/DIV 10nF 50mA/DIV C1: TAIYO YUDEN JMK212BJ475 C2: TAIYO YUDEN JMK316BJ106 C3: TAIYO YUDEN EMK212BJ474 D1: ON SEMICONDUCTOR MBR0520 L1: SUMIDA CR43-100 (408) 573-4150 (408) 573-4150 (408) 573-4150 (800) 282-9855 (847) 956-0666 Figure 3. USB to 5V SEPIC converter 6 1ms/DIV Figure 5. USB to 5V SEPIC start-up with shorted output Linear Technology Magazine • February 2001 DESIGN FEATURES 80 L1 10µH VIN 2.7V TO 5V D1 2.49Ω 20mA EFFICIENCY (%) 75 C1 4.7µF 9 70 10kHz TO 50kHz PWM BRIGHTNESS ADJUST 65 60 8 7 VIN SW SHDN ISP 50 100 150 200 250 LOAD CURRENT (mA) 300 350 R3 5.1k LT1618 4 IADJ GND USB to 12V Boost Converter Figure 1 shows a 5V to 12V boost converter ideal for USB applications. The converter has a selectable 100mA/500mA input current limit, allowing the device to be easily switched between the USB low and high power modes. Efficiency, shown 80 75 VIN = 4.2V EFFICIENCY (%) 70 VIN = 3.3V 65 VIN = 2.7V 60 55 50 45 40 0 5 10 15 20 LED CURRENT (mA) Figure 8. Li-Ion white LED driver efficiency Linear Technology Magazine • February 2001 R1 2M 1 FB VC 10 C2 1µF C3 0.1µF R2 100k CC 0.1µF Figure 6. USB to 5V SEPIC efficiency In addition to providing an accurate input current limit, the LT1618 can also be used to provide an accurately regulated output current for current-source applications. Driving white LEDs is one application for which the device is ideally suited. With an input voltage range of 1.6V to 18V, the LT1618 works from a variety of input sources. The 36V switch rating allows output voltages of up to 35V to be generated, easily driving up to eight white LEDs in series. The 1.4MHz switching frequency allows the use of low profile inductors and capacitors, which, along with the LT1618’s MSOP-10 package, helps to minimize board area. 2 ISN 5 0 3 C1: TAIYO YUDEN JMK212BJ475 C2: TAIYO YUDEN TMK316BJ105 D1: ON SEMICONDUCTOR MBR0530 L1: SUMIDA CLQ4D10-100 (408) 573-4150 (408) 573-4150 (800) 282-9855 (847) 956-0666 Figure 7. Li-Ion white LED driver in Figure 2, exceeds 85%. If the load demands more current than the converter can provide with the input current limited to 100mA (or 500mA), the output voltage will simply decrease and the LT1618 will operate in constant-current mode. For example, with an input current limit of 100mA, about 35mA can be provided to the 12V output. If the load increases to 50mA, the output voltage will reduce to approximately 8V to maintain a constant 100mA input current. USB to 5V SEPIC Converter with Short-Circuit Protection Unlike boost converters, SEPICs (single-ended primary inductance) converters) have an output that is DC-isolated from the input, so an input current limit not only helps soft start the output, but also provides excellent short-circuit protection. The 5V SEPIC converter shown in Figure 3 is ideal for applications that need the output voltage to go to zero during shutdown. The accurate input current limit ensures USB device compliance even under output fault conditions. Figure 4 shows the startup characteristic of the SEPIC converter with a 50mA load. By limiting the input current to 100mA, the output is effectively soft started, smoothly increasing and not overshooting its final 5V value. Figure 5 shows that the input current does not exceed 100mA even with the out- put shorted to ground (thus the flat output voltage waveform in the oscilloscope photo). Efficiency is shown in Figure 6. This converter also has a selectable input current limit of either 100mA or 500mA, making it ideal for high power USB applications. Li-Ion White LED Driver The circuit in Figure 7 is capable of driving six white LEDs from a single Li-Ion cell. LED brightness can be easily adjusted using a pulse width modulated (PWM) signal, as shown, or using a DC voltage to drive the IADJ pin directly, without the R3, C3 lowpass filter. If brightness control is not needed, simply connect the IADJ pin to ground. The typical output voltage is about 22V and the R1, R2 output divider sets the maximum output voltage to around 26V to protect the LT1618 if the LEDs are disconnected. The LT1618’s constant current loop regulates 50mV across the 2.49Ω sense resistor, setting the LED current to 20mA. Efficiency for this circuit, shown in Figure 8, exceeds 70%, which is significantly higher than the 30% to 50% efficiencies obtained when using a charge pump for LED drive. No current flows in the LEDs when the LT1618 is turned off. Their high forward voltages prevent them from turning on, ensuring a true low current shutdown with no excess battery leakage or light output. continued on page 23 7 DESIGN FEATURES 2.3MHz Monolithic, Continuous Time, Differential Lowpass Filter Provides Solutions for Wide Band CDMA by Nello Sevastopoulos and Mike Kultgen Applications Introducing the LTC1566-1: 2.3MHz Lowpass in SO-8 Setting the Input and Output and the 85dB attenuation floor Common Mode Levels The LTC1566-1 is a new monolithic 7th order continuous time lowpass filter featuring differential input and output terminals; it operates from a single 5V supply and dual supplies of up to ±5V and it is packaged in an 8-pin surface mount SO-8 package. The LTC1566-1 requires no external components other than power supply decoupling capacitors. It replaces bulky discrete designs built from differential amplifiers, op amps, precision resistors and capacitors. The filter is designed to have a flat passband from DC to 2MHz and a steep transition band. The –3dB cutoff frequency is 2.3MHz and the attenuation at 3.5MHz is in excess of 38dB. The filter gain gradually rolls off past 5MHz extends beyond 100MHz. This gain performance cannot be obtained with discrete components without trimming passive components. The filter cutoff is set at 2.3MHz to accommodate differential filtering needs in wide band CDMA base stations. Figure 1 shows the measured amplitude response and group delay. The LTC1566-1 is a fully integrated, continuous time, differential filter; its passband, its cutoff frequency and its transition band are fixed. Depending on demand, other filter cutoff frequencies as well as other lowpass filter responses up to 7th order can be provided. The passband gain is internally set to 4V/V (12dB); it can be lowered with three external resistors. R1* 1 GAIN 0 GAIN (dB) –10 –20 0.8 –30 0.6 –40 –50 0.4 GROUP DELAY –60 0.2 0.0 –70 –80 0.5 1.0 10 FREQUENCY (MHz) 50 Figure 1. LTC1566-1 gain and group delay vs frequency + – – + 2R2* R ×1 8 OUT + ×1 7 OUT – 6 V+ 5 VODC + A3 – – R1* 10 IN + A1 ×1 VIN– 20 GROUP DELAY (µs) VIN+ Figure 2 shows the block diagram of the LTC1566-1. The high input impedance of “floating amplifiers” A1 and A2 allows external resistors R1 IN – 2 + R 7th ORDER FILTER NETWORK WITH 12dB GAIN A2 ×1 – INPUT AMPLIFIERS WITH COMMON MODE TRANSLATION CIRCUIT + UNITY GAIN OUTPUT BUFFERS WITH DC REFERENCE ADJUSTMENT GND 3 V– 4 *NEEDED ONLY FOR PASSBAND GAIN < 12dB, R2 (IN+ – IN–) = (VIN+ – VIN–) • R1 + R2 Figure 2. LTC1566-1 block diagram 8 Linear Technology Magazine • February 2001 DESIGN FEATURES 0.1µF + – IN + 2 IN – OUT + 8 VOUT+ OUT – 7 VOUT– V+ 6 + – LTC1566-1 VIN+ + – 1 VIN– 3 10k 0.1µF 4 GND 5V 10k V– VODC 2 Figure 3. Single 5V supply operation, DC-coupled inputs – 30 VS = 5V VS = ±5V S/N THD, SNR (dB) – 40 – 50 – 70 – 80 – 90 0.5 5.1k 1.0 3.5 1.5 2.0 2.5 3.0 DIFFERENTIAL OUTPUT (VP-P) 4.0 Figure 5. Total harmonic distortion and signal-to-noise ratio vs differential output voltage for single 5V and ±5V supplies Linear Technology Magazine • February 2001 8 VOUT+ OUT – 7 VOUT– V+ 6 LTC1566-1 0.1µF 4 GND 5V 5.1k V– VODC AC COUPLED INPUT VIN (COMMON MODE) = VOUT (COMMON MODE) = 0.1µF 5 V+ 2 Figure 4. Single 5V supply operation, AC-coupled inputs allowing floating amplifiers A1 and A2 to operate at an input common mode voltage dictated by the differential signal source driving the filter. Another unique feature of the LTC1566-1 is the ability to introduce a differential offset voltage at the output of the filter. As seen in the block diagram, Figure 2, if a DC voltage is applied at pin 5 with respect to pin 3, it will be added to the differential voltage seen between pins 7 and 8. The DC output common mode voltage is therefore the arithmetic average of the DC voltages applied at pin 3 and pin 5. This output DC offset control can be used for sideband suppression of differential modulators, calibration of A/Ds or simple signal summation. Figure 3 shows a typical connection for single-supply operation where the differential output is DC biased at one-half the power supply voltage. The input can be DC or AC coupled (Figure 3, Figure 4). AC coupling should be used if the common mode input voltage is outside the input range of the filter, as illustrated in Figure 4. Dynamic Range – 60 IN – 3 5 SINGLE 5V SUPPLY: 1V ≤ VIN (COMMON MODE) ≤ 3V ±5V SUPPLY: –4V ≤ VIN COMMON MODE) ≤ 3V 2 OUT + 100k 0.1µF V + + VIN– VIN (COMMON MODE) = IN and 2R2 to be added to attenuate the differential input signal and to lower the effective passband gain of the circuit, if necessary. For example, if a gain of 2 (6dB) is desired, R1 = R2. The LTC1566-1 is also capable of providing common mode voltage level shifting; that is, it can process differential input signals and provide filtered output differential signals with different common mode voltage levels. This is quite desirable, as components along a differential signal path may be optimized for a specific DC common mode level. For instance, the common mode output of a differential demodulator can be different than the required common mode input of a differential A/D converter. The common mode translation is performed through unity-gain input buffers A1 and A2 and op amp A3 (Figure 2). Amplifier A3 forces the input amplifiers to operate with a common mode voltage dictated by the biasing of pin 3 (the ground pin) while VIN– IN + 100k VIN+ 0.1µF + – 1 The total output in-band noise (DC to 2MHz) is typically 230µVRMS. Figure 5 shows the output signal-to-noise ratio vs differential output voltage. With a 1VRMS output level (2.8VP-P differential) the filter features 73dB SINAD (S/N and THD). Note that the maximum dynamic range of the IC is independent of its power supply volt- age. With dual 5V supplies, however, the filter can accept differential signals with wider common mode levels. The out-of-band noise is almost negligible due to the steep roll-off of the filter transition band. Input referred, the noise at each input terminal of the filter (pins 1 and 2) is 41µVRMS or –138dBm/Hz. Intermodulation The coexistence of AMPS (American Mobile Phone System), CDMA (Code Division Multiple Access), and wide-band CDMA (WCDMA) cellular systems has increased the need for linearity in the transmitter and receiver circuits. In a CDMA or WCDMA transmitters, intermodulation of components in the spread-spectrum signal creates spectral regrowth and, consequently, adjacent channel interference. CDMA and WCDMA must operate in the presence of AMPS signals in the same channel (cochannel interference). Intermodulation between the AMPS signal and the CDMA/WCDMA signal desensitizes the receiver. Intermodulation is reduced by making the circuit as linear as possible. Both receiver and transmitter linearity can be characterized by measuring the intermodulation of two tones in the passband. When two tones of equal amplitude are simultaneously applied to a weakly nonlinear circuit, the output spectrum above the two fundamentals will include the second and third harmonics of the input sources, the sum 9 DESIGN FEATURES A Brief Overview of Filter Technologies. Switched capacitor filter technology allows the filter cutoff frequency to be tuned with an external or internal clock, the value of the cutoff frequency being a multiple of the clock frequency. This highly convenient feature is always mitigated by the “sampled data” nature of the filter that always requires a clock, a “digital” element in the middle of a pure analog function. If a clock is not available, the clock must be designed as a separate circuit. If a clock is available, it often has to be conditioned to provide the appropriate multiple of the required cutoff frequency (for example, 100 times the filter cutoff). Typically, the clock will be routed from the digital part of the system into the analog board space, creating board layout issues. Today, fully integrated switched capacitor lowpass filters are widely available. Most of them require an external clock because either the filter product by choice did not integrate the clock or the inaccuracy of the internal clock makes it useless. The cutoff frequency of most of these products is well below 100kHz.1 and the difference frequencies of the two input sources (IM2) and the sum and differences of twice one input source and the other (IM3). Furthermore, if the same two tones are applied to an LTC1566-1 lowpass filter, the filter selectivity will attenuate the out-of-band spurs. The 2nd order intermodulation products (IM2) and some 3rd order intermodulation RC active filter technology provides wide dynamic range and much higher cutoff frequencies than monolithic switched capacitor filters. RC active filters are mainly realized with discrete components (op amps, resistors and capacitors). With the availability of high speed op amps, cutoff frequencies of a few MHz or more can be obtained. The filter shape and cutoff frequency are determined by the appropriate choice of the discrete passive components required to build the filter. LTC’s newer RC active “monolithic” filters (the LTC1562, LTC1562-2 and LTC1563-X) integrate the op amps, the precision capacitors and some resistors to provide a compact filtering solution for cutoff frequencies up to 300kHz. The cutoff frequency, the filter type and the filter shape are programmed with external resistors. The internal capacitors are trimmed to better than 1% to provide more accurate filtering than their discrete counterparts. Furthermore, the newer FilterCAD 3.0 filter design software (see Linear Technology X:3, September 2000) allows the system designer to easily realize simple or complex filter functions using the above ICs. Fully integrating a discrete RC active filter implies integrating both the active components and all passive components, thus losing the tunability of the filter. However, it is worth mentioning that tunable monolithic continuous time filters have been realized for specific high speed applications with reduced dynamic range. The tuning elements are either transconductors or MOSFETs, replacing the traditional resistor of an RC active filter. For applications where the filter cutoff frequency is fixed, monolithic continuous time filters with preset cutoff frequencies can provide most of the advantages of the discrete RC active filters and eliminate the cumbersome, and sometimes hard to get, external passive components. Furthermore, if the signal path is differential, a discrete differential RC active design becomes complex and a fully integrated solution is desirable. 1 The new 10th order linear phase lowpass filter family, the LTC1569-X, is the only IC to solve the clock generation and clock routing problem by providing a precision internal clock that can be easily programmed via a single external resistor. × 450kHz – 2MHz = 1.1MHz). The IM2 products, 2MHz + 455kHz and 2MHz – 455kHz, are also shown and are weaker than the IM3s, as expected. The suppression of the IM2 products is due to the fully differential nature of the LTC1566-1, which tends to cancel them. Furthermore it can be proven that the IM3 products increase by approximately 3dB for each dB of products (IM3), however, may fall within the passband or in the vicinity of the band edge of the circuit and their presence can limit system performance. Figure 6 shows the actual test circuit with 455kHz and 2MHz tones simultaneously applied at the filter’s differential inputs. Figure 7 shows the measured IM3 products (2 × 2MHz – 455kHz = 3.55MHz, and 2 15V HP33120 0° π 1 0° Σ Σ VIN π 49.9Ω LC FILTER HP89410 LTC1566-1 49.9Ω LC FILTER MINI-CIRCUIT SPLITTERS V+ 2 2 3 4 IN+ OUT+ 8 IN– OUT– 7 GND V– 0.1µF 2.49k 6 V+ VODC 2.49k 3 7 + LT1363 4.99k 5V 5 2 – 4 6 HP89410 RIN = 1M 4.99k 0.1µF –15V Figure 6. Test circuit for intermodulation distortion 10 Linear Technology Magazine • February 2001 DESIGN FEATURES 0 OUTPUT VOLTAGE (dBm) VS = 5V INPUTS 450kHz, 2MHz 1.1MHz (IM3) –20 3.55MHz (IM3) –40 2.45MHz (IM2) –60 1.55MHz (IM2) –80 NOISE FLOOR –100 –25 –20 –15 –10 VIN (dBm) –5 0 Figure 7. 450kHz/2MHz intermodulation, VS = 5V input signal increase, so their presence in the passband must be minimized or eliminated. As shown in Figure 7, the excellent linearity of the LTC1566-1 provides an intermodulation ratio (IM) of better than 70dB for output levels of –3Bm or lower. The IM performance of the LTC1566-1 is better than some commercially available passive LC filter modules. Out-of-Band Attenuation The amplitude response of a filter is routinely tested with an input signal of varying frequency and constant amplitude; yet, a common requirement in communication systems is the ability to process an in-band signal in the presence of large out-of-band interference. The active filter should be designed to meet these stringent requirements. Figure 8 shows a test circuit that simulates the case where the LTC1566-1 receives a constantamplitude, in-band signal in the presence of strong out-of-band interference. Figure 9 shows the measured filter output. Three out-of-band tones (3MHz, 5MHz, 10MHz) are summed with a –2dBm (0.5VP-P) 2MHz in-band signal. Figure 9 should be compared with the gain response of the filter shown in Figure 1. As can be seen in Figure 9, the LTC1566-1 can attenuate a 12dBm (2.52V P-P ) 10MHz out-of-band signal by 50dB, while amplifying an in-band 0.5VP-P (–2dBm) 2MHz signal without gain error. The maximum allowable amplitude of the 10MHz out-of-band signal is 13dBm (2.82VP-P); a larger signal will warp the passband gain. This excellent dynamic performance is attributable to the internal architecture of the LTC1566-1, which provides band-limiting at early stages. Similar observations can be made for the 5MHz and 3MHz cases of Figure 9, although large 3MHz signals will warp the passband gain sooner; this is expected, because high ampli- 20 VS = 5V 10 2MHz 0 OUTPUT LEVEL (dBm) 20 –10 3MHz –20 –30 5MHz –40 –50 10MHz –60 –70 –25 –20 –15 –10 –5 0 5 INPUT LEVEL (dBm) 10 15 Figure 9. Out-of-band rejection, VS = 5V tude out-of-band 3MHz tones will also form in-band IM3 and IM2 products. To conclude, the LTC1566-1 attenuates out-of-band signals that are smaller than, equal to or larger than in-band signals. Conclusion The LTC1566-1 is a monolithic, selfcontained, fully differential lowpass filter with outstanding linearity; it can process a wide spectrum of input signals and, in addition to filtering, it can provide common mode DC level shifting. Authors can be contacted at (408) 432-1900 15V HP33120 3, 5, 10MHz 0° π Σ Σ 1 π 2 3 49.9Ω LC FILTER HP89410 0° MINI-CIRCUIT SPLITTERS 4 THE DIFFERENTIAL INPUT VOLTAGE IS A –2dBm, 2MHz SIGNAL SUMMED WITH A VARIABLE AMPLITUDE 3MHz, 5MHz OR 10MHz INTERFERER OUT+ IN– 7 OUT– V– 6 V+ VODC 2.49k 8 IN+ GND 0.1µF 2.49k LTC1566-1 49.9Ω 3 7 + LT1363 4.99k 5V 5 2 – 4 6 HP89410 RIN = 1M 4.99k 0.1µF V+ 2 –15V Figure 8. Test circuit for out-of-band rejection Linear Technology Magazine • February 2001 11 DESIGN FEATURES Very Low Cost Li-Ion Battery Charger Requires Little Area and Few Components by David Laude A Simple, Low Cost The LTC1734 is a precision, low cost, Li-Ion Charger Introduction single-cell, linear Li-Ion battery charger with constant voltage and constant current control. The small quantity and low cost of external components results in a very low overall system cost and the part’s 6-pin SOT23 package allows for a compact design solution. Previous products usually required an external current sensing resistor and blocking diode whose functions are included in the LTC1734. Other features include: ❏ 1% accurate float voltage options of 4.1V or 4.2V ❏ Programmable constant current range of 200mA to 700mA ❏ Charging current monitor output and manual shutdown for use with a microcontroller ❏ Automated shutdown with no battery drain after supply removal ❏ Undervoltage lockout ❏ Self protection for overcurrent and overtemperature Applications include such compact devices as cellular phones, digital cameras and handheld computers. The LTC1734 can also be used as a general purpose current source or for charging nickel-cadmium or nickelmetal-hydride batteries. A battery charger programmed for 300mA in the constant current mode with a charge current monitoring function is shown in Figure 1. The PNP transistor is needed to source the charging current and resistor R1 is used to program the maximum charging current. Note that no external current sense resistor or diode to block current is required. When the supply is opened or shorted to ground, the charger shuts down and no quiescent current flows from the battery to the charger. This feature extends battery life. Capacitor C2 can consist of up to 100µF of bypass caps, which would normally be distributed along the battery line. The supply voltage can range from 4.75V to 8V, but power dissipation of the PNP may become excessive near the higher end. The programming pin (PROG) accomplishes several functions. It is used to set the current in the constant current mode, monitor the charging current and manually shut down the charger. In the constant current mode, the LTC1734 maintains the PROG pin at 1.5V. The PROG pin voltage drops below 1.5V as the constant voltage mode is entered and charge current drops off. The charge current is always one thousand times the current through R1 and is therefore proportional to the PROG pin voltage. At 1.5V the current is the full 300mA, whereas at 0.15V the current is 300mA/10 or 30mA. If the grounded side of R1 is pulled above 2.15V or is allowed to float, the charger enters the manual shutdown mode and charging ceases. These features enable smarter charging by allowing a microcontroller to monitor the charging current and shut down the charger at the appropriate time. The ISENSE and BAT pins are used to monitor charge current and battery voltage, respectively; the DRIVE pin controls the PNP’s base. A Programmable Constant Current Source An example of a programmable current source is shown in Figure 2. To ensure that only the constant current mode is activated, the BAT pin is tied to ground to prevent the constant voltage control loop from engaging. The control inputs either float or are connected to ground. This can be achieved by driving them from the drains of NMOS FETs or from the collectors of NPNs. When both inputs are floating, manual shutdown is continued on page 15 LTC1734 1 3 VIN 5V 2 C1 1µF ISENSE VCC GND LTC1734 DRIVE BAT PROG 6 4 R1 5.1k Q1: ZETEX FMMT549 (631) 543-7100 CHARGE CURRENT MONITOR Figure 1. Simple, low cost charger programmed for 300mA output current 12 1 Q1 5 C2 10µF SINGLE Li-Ion CELL VIN 5V C1 1µF Q1: ZETEX FCX589 3 2 ISENSE VCC GND DRIVE BAT PROG 6 Q1 5 4 LOAD R1 3k (631) 543-7100 R2 7.5k CONTROL 1 CONTROL 2 Figure 2. Programmable current source with output current of 0mA, 200mA, 500mA or 700mA Linear Technology Magazine • February 2001 DESIGN FEATURES Synchronous Buck Controller Extends Battery Life and Fits in a Small Footprint by Peter Guan Introduction Operation Portable electronic devices continue to decrease in size and their supply voltages are also falling, but load current requirements are increasing as a result of higher processing speed and improved features. This trend places more constraints on today’s portable power supplies, but Linear Technology has the solution. The LTC1773 is a synchronous DC/DC controller that packs high output current capability and low operating quiescent current in a small MSOP-10 package. Its input voltage range is from 2.65V to 8.5V; this is ideal for 1- or 2-cell Li-Ion batteries as well as 3- to 6-cell NiCd and NiMH battery packs because it allows the batteries to operate near end of charge. A precise internal undervoltage lockout circuit prevents deep discharge of the battery below 2.5V. Popular features such as OPTILOOP™ compensation, soft start and Burst Mode™ operation are also included. Combined with its small MSOP package, the LTC1773’s high operating frequency of 550kHz allows the use of small, surface mount components to provide a compact power supply solution. Figure 1 shows a typical application of the LTC1773 in a 5V to 2.5V/3A step-down converter. Figure 2 shows its efficiency vs load current. The LTC1773 uses a constant frequency, current mode architecture to drive an external pair of complementary power MOSFETs. An internal oscillator sets Portable electronic devices continue to decrease in size and their supply voltages are also falling, but load current requirements are increasing as a result of higher processing speed and improved features. This trend places more constraints on today’s portable power supplies, but Linear Technology has the solution. the operating frequency of the device. The P-channel MOSFET turns on with every oscillator cycle and turns off when the internal current comparator trips, indicating that the inductor current has reached a level set by the ITH pin. An internal error amplifier, in turn, drives the ITH pin by monitoring the output voltage through an external resistive divider connected to the VFB pin. While the P-channel MOSFET is off, the synchronous N-channel MOSFET turns on until either the inductor current starts to reverse, as indicated by the SW pin going below ground, or until the beginning of the next cycle. Synchronous, Burst Mode and Forced Continuous Operation Three modes of operation can be selected through the SYNC/FCB pin. Tying it above 0.8V or leaving it floating will enable Burst Mode operation, which increases efficiency during light load conditions. During Burst Mode operation, the peak inductor current limit is clamped to about a third of the maximum value and the ITH pin is monitored to determine whether the device will go into a power-saving Sleep mode. The ITH level is inversely proportional to the output voltage error. When the inductor’s average current is higher than the load requirement, the output voltage rises while the ITH level drops. When ITH dips below 0.22V, the device goes into Sleep mode, turning off the external VIN 2.65V TO 8.5V 2 1 0.1µF 47pF 220pF CC VIN SYNC/FCB – RUN/SS SENSE ITH TG LTC1773 30k RC SW 4 VFB GND 5 BG CIN 68µF RSENSE 0.025Ω 9 L1 3µH 7 10 VOUT 2.5V 6 R1 80.6k Si9801DY R2 169k Figure 1. 5V to 2.5V/3A step-down converter Linear Technology Magazine • February 2001 90 VIN = 5V 85 VIN = 8V 80 75 70 65 + COUT 180µF L1: SUMIDA CDRH6D28-3R0 (847) 956-0667 VIN = 3.3V 95 + 8 EFFICIENCY (%) 3 100 60 55 1 10 100 1000 OUTPUT CURRENT (mA) 5000 Figure 2. Efficiency of Figure 1’s circuit with several input voltages 13 DESIGN FEATURES placed at the pin to program its rise time to ensure a soft start at the output by limiting the amount of charge current into the output capacitors. The RUN/SS pin also serves another function: if the pin is tied below 0.65V, the part goes into shutdown and consumes less than 10µA of input current. VOUT 100mV/DIV IL 2A/DIV VOUT 100mV/DIV IL 2A/DIV Fault Protection VIN = 5V 10mV/DIV VOUT = 2.5V 100mA TO 5A LOAD STEP Figure 3a. Load-step response, Burst Mode operation power MOSFETs and most of the internal circuitry; in this state, the LTC1773 consumes only 80µA of quiescent current. At this point, the load current is being supplied by the output capacitor. As the output droops, ITH will be driven higher. When ITH rises above 0.27V, the device resumes normal operation. For frequency-sensitive applications, Burst Mode operation can be inhibited by tying the SYNC/FCB pin to below 0.8V to force continuous operation, which will continually drive the external power MOSFETs synchronously regardless of the output load. The inductor current is allowed to reverse in this case. In addition to being a logic input threshold, the 0.8V threshold of the SYNC/FCB pin can also be used to regulate a secondary winding output by forcing continuous synchronous operation regardless of the primary output load. A logic-level clock signal connected to the SYNC/FCB pin synchronizes the operating frequency to an external source between 585kHz and 750kHz. Burst Mode operation is automatically disabled during synchronization to reduce noise. Instead, cycle skipping occurs under light load conditions because current reversal is not allowed. This boosts the low current efficiency while providing low output ripple. Run/Soft Start Upon power up, the RUN/SS pin is pulled high by an internal current source; an external capacitor can be 14 The LTC1773 incorporates protection features such as programmable current limit, input undervoltage lockout, output overvoltage protection and frequency foldback when the output falls out of regulation. One of the advantages of a current mode switching regulator is that current is regulated during every clock cycle, thus providing current overload protection on a pulse-by-pulse basis. Current limit is programmed through an external high-side sense resistor. The maximum sense voltage across this resistor is 100mV. But taking into account current ripple, input noise and sense resistor tolerance, 70mV should be used in choosing the proper sense resistor (RSENSE = 70mV/IOUT). To protect a battery power source from deep discharge near its end of charge, an internal undervoltage lockout circuit shuts down the device when VIN drops below 2.5V. This reduces the current consumption to about 2µA. A built-in 150mV hysteresis ensures reliable operation with noisy supplies. During transient overshoots and other more serious conditions that may cause the output to rise out of regulation (>7.5%), an internal overvoltage comparator will turn off the main MOSFET and turn on the synchronous MOSFET until the overvoltage condition is cleared. During this time, if the main MOSFET is defective or shorted to ground, current will flow directly from VIN to ground, blowing the system fuse and saving the other board components. In addition, if the output is shorted to ground, the frequency of the oscillator is reduced to about 55kHz, 1/10 of the nominal frequency. This fre- 10mV/DIV VIN = 5V VOUT = 2.5V 100mA TO 5A LOAD STEP Figure 3b. Load-step response, continuous mode operation quency foldback ensures that the inductor current has enough time to decay, thereby preventing runaway. The oscillator’s frequency will gradually increase back to 550kHz when the VFB pin rises above 0.4V. Dropout Operation During the discharging of a battery source, when the input supply voltage decreases toward the output voltage, the duty cycle increases toward the maximum on-time. The output voltage will then be determined by VIN minus the I • R voltage drops across the external P-channel MOSFET, the sense resistor and the inductor. OPTI-LOOP Compensation To meet stringent transient response requirements, other switching regulators may need to use many large and expensive output capacitors to reduce the output voltage droop during a load step. The LTC1773, with OPTI-LOOP compensation, requires fewer output capacitors and also allows the use of inexpensive ceramic capacitors. The ITH pin of the LTC1773 allows users to choose the proper component values to compensate the loop so that the transient response can be optimized with the minimum number of output capacitors. Line and Load Regulation The current mode architecture of the LTC1773 ensures excellent line and load regulation without cumbersome compensation and excessive output Linear Technology Magazine • February 2001 DESIGN FEATURES capacitance. Figures 3a and 3b show the response of the LTC1773 to a 100mA to 5A load step during Burst Mode and continuous mode operations, respectively. 2.7V ≤ VIN ≤ 6V 47pF 30k 1 220pF 2 0.1µF 3 1.8V/7A Application Figure 4 shows a step-down application from 3.3V to 1.8V at 7A. When operating below 5V, care should be taken to choose the proper sublogiclevel MOSFETs that have relatively low gate charge. For high current (>3A) applications, single P-channel and N-channel MOSFETs should be used instead of complementary MOSFETs in one package. A good figure of merit for MOSFETs is the RDS(ON) gatecharge product. The lower this value is, the higher the application’s efficiency will be. In addition to normal step-down applications, the LTC1773 can also be used in a zeta converter configuration that will do both step-down and step-up conversions, as shown in Figure 5. This application is ideal for battery-powered operation, in which a regulated 3.3V output is maintained during the entire discharge cycle of a Li-Ion battery from 4.7V to 2.5V. Conclusion The LTC1773 offers flexibility, high efficiency and many other popular features in a small MSOP-10 package. For low voltage portable systems that require small footprint and high efficiency, the LTC1773 is the ideal solution. 4 5 100pF LTC1773 ITH 10 SW RUN/SS SENSE SYNC/FCB 8 VIN VFB TG GND BG 99k 1% RSENSE 0.01Ω 9 – M1 7 6 M2 + 100pF 80.6k 1% D2* MBRS340T3 CIN: PANASONIC SPECIAL POLYMER COUT: KEMET T510687K004AS L1: TOKO TYPE D104C 919AS-1RON RSENSE: IRC LR2512-01-R010-J M1: FAIRCHILD FDS6375 M2: SILICONIX Si9804DY (714) 737-7334 (408) 986-0424 (847) 699-3430 (361) 992-7900 (408) 822-2126 (800) 554-5565 Linear Technology Magazine • February 2001 + CIN 150µF 6.3V 4.7µF 6.3V 0.1µF COUT 680µF 4V ×2 *NOTE: D2 IS OPTIONAL. IF REMOVED, EFFICIENCY DROPS BY 1% Figure 4. 3.3V to 1.8V/7A regulator 33pF 2.7V ≤ VIN ≤ 4.2V 30k 200pF 1 2 0.1µF VIN 3 4 5 LTC1773 ITH SW RUN/SS SENSE– SYNC/FCB VIN VFB TG GND BG 10 9 RSENSE 0.025Ω 8 M1 47µF L1 2µH 7 + + M2 80.6k 1% CIN: COUT: L1: RSENSE: M1: M2: 249k 1% SANYO POSCAP 6TPA150M AVX TPSD227M006R0100 COILTRONICS CTX2-4 IRC LR1206-01-R033-F SILICONIX Si9803DY SILICONIX Si9804DY VOUT 3.3V 1A 6 + CIN 150µF 6.3V COUT 220µF 6.3V L1 2µH (714) 373-7334 (207) 282-5111 (561) 752-5000 (361) 992-7900 (800) 554-5565 Figure 5. Single Li-Ion cell to 3.3V/1A synchronous zeta converter Conclusion LTC1734, continued from page 12 entered. Connecting Control 1 to ground causes 500mA of current to flow into the load, whereas Control 2 results in 200mA of current. When both control inputs are grounded the current is 700mA. A voltage DAC, VOUT 1.8V 7A L1 1µH connected to the PROG pin through a resistor, could also be used to control the current. Applications include charging nickel-cadmium or nickelmetal-hydride batteries, driving LEDs or biasing bridge circuits. Low cost, small footprint, reduced component count, precision and versatility make the LTC1734 an excellent solution for implementing compact and inexpensive battery chargers or constant current sources. 15 DESIGN FEATURES New No RSENSE Controllers Deliver Very Low Output Voltages by Christopher B. Umminger Valley Current Control Digital system voltages are dropping of very high efficiency DC/DC step- Enables tON(MIN) < 100ns Introduction ever lower, yet battery voltages are not. This forces DC/DC step-down converters in portable products to operate at lower duty cycles. Unfortunately, low duty cycle operation decreases efficiency due to both increased switching losses and the increased importance of I2R losses at low output voltages. Furthermore, conventional control architectures often have difficulty operating with very short switch on-times. The LTC1778 and LTC3711 with VID address these problems with a new architecture for buck regulators that delivers the low output voltages and high efficiencies that modern portable supplies require. The LTC1778 is a step-down controller that provides synchronous drive for two external N-channel MOSFET switches. It comes with a variety of features to ease the design ION down converters. The true current mode control architecture has an adjustable current limit, can be easily compensated, is stable with ceramic output capacitors and does not require a power-wasting sense resistor. An optional discontinuous mode of operation increases efficiency at light loads. The LTC1778 operates over a wide range of input voltages from 4V to 36V and output voltages from 0.8V up to 90% of VIN. Switching frequencies up to nearly 2MHz can be chosen, allowing wide latitude in trading off efficiency for component size. Fault protection features include a power-good output, current limit foldback, optional short-circuit shutdown timer and an overvoltage soft latch. The LTC3711 is essentially the same as the LTC1778 but includes a 5-bit VID interface. VON Power supplies for modern portable computers require that voltages as high as 24V from a battery pack or wall adapter be converted down to levels from 2.5V to as low as 0.8V. Such a large ratio of input to output voltage means that a buck regulator must operate with duty cycles down to 3%. At 300kHz operation, this implies a main switch on-time of only 110ns. Conventional current mode regulators have difficulty achieving on-times this short, forcing lower frequency operation and the use of larger components. To overcome this limitation, the LTC1778 family uses a valley current control architecture that is illustrated in Figure 1. Current is sensed by the voltage drop between the SW (or SENSE+) and PGND (or SENSE–) pins while the bottom switch, M2, is turned on. During this time the negative VIN TG TOP S Q M1 L1 VVON D= • CT IION R Q BG COUT ONE SHOT DELAY VOUT + M2 – ICMP –133mV TO 267mV + – + 20k –7µA TO 13µA VRNG PGND/SENSE– SW/SENSE+ × 0.8V 0.5 – 2 EA 0µA TO 10µA 1 240k + 1.7mS R2 VFB – 3.3µA ITH 0V TO 2.4V R1 CC Figure 1. LTC1778 main control loop 16 Linear Technology Magazine • February 2001 DESIGN FEATURES DROPOUT REGION 1.0 0.5 0 0 0.25 0.50 0.75 DUTY CYCLE (VOUT/VIN) 1.0 Figure 2. Maximum switching frequency vs duty cycle Flexible One-Shot Timer Keeps Frequency Constant voltage across inductor L1 causes the current flowing through it to decay. When it reaches the level set by the current-control threshold (ITH) voltage, the current comparator (ICMP) trips. This sets the latch, turning off the bottom switch and turning on the top (or main) switch, M1. After a controlled delay determined by a one-shot timer, the top switch turns off again and the cycle repeats. The currentcontrol threshold is set by an error amplifier (EA) that compares the divided output voltage with a 0.8V reference in order to keep the threshold at a level that matches the load current. This control loop has several advantages compared to peak-curCSS 0.1µF 1 R3 11k R4 39k RPG 100k 2 3 CC1 510pF 4 RC 20k 5 CC2 100pF 6 7 R1 14.0k R2 30.1k 8 Although the LTC1778 does not contain an internal oscillator, switching frequency is kept approximately constant through the use of a flexible one-shot timer that controls the top switch on-time. A current entering the ION pin (IION) charges an internal timing capacitor (CT) to the voltage applied at the VON pin (VVON) to determine the on-time: tON = CT • VVON/IION. For a buck regulator running at a constant frequency, the on-time is proportional to VOUT/VIN. By connecting a resistor (RON) from VIN to the ION pin and connecting VOUT to the VON pin (if available), the one-shot duration can be made proportional to VOUT and inversely proportional to VIN. The converter will then operate at an ap- LTC1778 RUN/SS BOOST PGOOD TG VRNG SW FCB ITH SGND 16 15 VIN = 5V 90 EFFICIENCY (%) 1.5 100 rent controllers that use an internal oscillator. Because only a one-shot timer determines the top switch on-time, it can be made very short for low duty cycle applications. Another advantage is that slope compensation is not required. Furthermore, response to a load step increase can be very fast since the loop does not have to wait for an oscillator pulse before the top switch is turned on and current begins increasing. 70 VOUT = 2.5V EXTVCC = 5V f = 250kHz 60 0.01 BG INTVCC ION VIN VFB EXTVCC 0.1 1 LOAD CURRENT (A) 10 Figure 4. Efficiency vs load current for Figure 3’s circuit proximately constant frequency equal to (RON • CT)–1. In most applications, the output voltage is not intended to change. Thus, some versions of the LTC1778 do not make the VON pin available and it defaults internally to 0.7V. By adjusting the value of RON, a wide range of operating frequencies can be selected. However, an important limit is set by the 500ns minimum off-time of the top switch. This is the minimum time required by the LTC1778 to turn on the bottom switch, sense the current and then shut it off. At a given switching frequency, it places a limit on the maximum duty cycle as illustrated in Figure 2. For example, at 200kHz operation, the LTC1778 can accommodate duty cycles up to 90%. Attempting to M1 14 VIN 5V TO 28V CIN 10µF 50V ×3 CB 0.22µF L1, 1.8µH + PGND VIN = 25V 80 DB CMDSH-3 13 M2 D1 COUT1-2 180µF 4V ×2 COUT3 22µF 6.3V X7R VOUT 2.5V 10A 12 11 + SWITCHING FREQUENCY (MHz) 2.0 CVCC 4.7µF 10 RF 1Ω 9 CF 0.1µF RON 1.40M CIN: UNITED CHEMICON THCR70E1H26ZT COUT1-2: CORNELL DUBILIER ESRE181E04B L1: SUMIDA CEP125-IR8MC-H M1: SILICONIX Si4884 M2: SILICONIX Si4874 D1: DIODES, INC. B340A (847) 696-2000 (508) 996-8561 (847) 956-0667 (800) 554-5565 (805) 446-4800 Figure 3. 2.5V/10A converter switches at 250kHz Linear Technology Magazine • February 2001 17 DESIGN FEATURES operate at duty cycles above this limit will cause the output voltage to drop out of regulation, down to a value that satisfies the duty cycle limit. Thus, the LTC1778 can be used in exceptionally high frequency buck converters, provided that the duty cycle is low enough. For example, a 10V to 2.5V converter can be run at frequencies as high as 1.5MHz. No RSENSE Operation Raises Efficiency at Low VOUT The LTC1778 offers true current mode control without the need for a sense resistor, an expensive component that is sometimes difficult to procure. The current comparator monitors the voltage drop between the SW and PGND pins, determining inductor current using the on-resistance of the bottom MOSFET. In addition to eliminating the sense resistor, this technique also simplifies the board layout and improves efficiency. The efficiency gain is especially noticeable in low output voltage applications where the resistor sense voltage is a significant fraction of the output voltage. For example, a 50mV sense voltage reduces efficiency by 5% in a 1V output converter. The LTC1778 allows the current sense range to be adjusted using the VRNG pin to accommodate a variety of MOSFET on-resistances. The power supply designer can easily trade off efficiency and cost in the choice of VOUT 50mV/DIV IL 5A/DIV MOSFET on-resistance. The voltage presented at the VRNG pin should be ten times the nominal sense voltage at maximum load current, for example, VRNG = 1V corresponds to a nominal sense voltage of 100mV. Connecting this pin to INTVCC or ground defaults the nominal sense voltage to 140mV or 70mV, respectively. Current is limited at 150% and –50% of the nominal level set by the VRNG pin. For those applications that require more accurate current measurement, the LTC3711 and some versions of the LTC1778 make available one or both of the current comparator inputs as separate SENSE+ and SENSE– pins. Connecting the inputs to a precise sense resistor placed in series with the source of the bottom MOSFET switch determines current more accurately. This is especially beneficial for applications that need a more accurate current limit or seek to actively position the output voltage as the load current varies. will be turned off and the bottom switch turned on until the output is pulled back below the power-good threshold. In an undervoltage condition, if the output falls by 25%, a short-circuit latch-off timer will be started. If the output has not recovered within this time, both switches will be shut off, stopping the converter. Undervoltage/short-circuit latch-off can be overridden. In this case, if the output voltage continues to fall below 50% of the regulation point, the current limit will be reduced, or folded back, to about one fourth of its maximum value. Popular Features from Other Controllers Remain Continuous synchronous operation at light loads reduces efficiency due to the large amount of current consumed by switching losses. Efficiency is improved by operating the converter in discontinuous mode. In this mode, the bottom switch is turned off at the instant that inductor current starts Output is Protected to reverse, even though the current from a Variety of Faults control threshold (ITH) is below that The LTC1778 comes with a number of level. The top switch, however, is not fault protection features. The output turned on until the ITH level rises back voltage is continuously monitored for to the point corresponding to zero out-of-range conditions. If it deviates inductor current. During the time both by more than ±7.5% from the regula- switches are off, the output current is tion point, an open drain power-good provided solely by the output capacioutput will pull low to indicate the tor and switching losses are avoided. out-of-regulation condition. In an The switching frequency becomes proovervoltage situation, the top switch portional to the load current in this mode of operation. The LTC1778 contains its own internal low dropout regulator that provides the 5V gate drive required for logic-level MOSFETs. However, it is also able to accept an external 5V to 7V supply if one is available. Connecting such a supply to the EXTVCC pin disables the internal regulator; all controller and gate drive power is LOAD STEP = 1A TO 10A VIN = 15V then drawn from the external supply. VOUT = 2.5V If the external drive comes from a high FCB = INTVCC efficiency source, overall efficiency can be improved. Furthermore, connecting the VIN and EXTVCC pins together to an external 5V supply allows the controller to convert low input volt20µs/DIV ages such as 3.3V and 2.5V. Figure 5. Transient response of Figure 3’s circuit 18 Linear Technology Magazine • February 2001 DESIGN FEATURES RPG 100k 2 3 CC1 470pF 4 RC 33k 5 CC2 100pF 6 7 R1 11.5k R2 24.9k 8 C2 2200pF LTC1778 RUN/SS BOOST PGOOD VRNG FCB ITH SGND TG SW 16 15 DB CMDSH-3 CB 0.22µF M1 14 BG INTVCC ION VIN VFB EXTVCC VOUT 2.5V 3A L1, 1µH + PGND VIN 9V TO 18V CIN 10µF 25V 13 COUT 120µF 4V M2 12 11 CVCC 4.7µF 10 RF 1Ω 9 CF 0.1µF RON 220k CIN: TAIYO YUDEN TMK432BJ106MM COUT: CORNELL DUBILIER ESRD121M04B L1: TOKO D63LCB M1, M2: 1/2 SILICONIX Si9802 LTC3711 Adds VID Interface for 0.9V – 2.0V Microprocessor Core Supplies (408) 573-4150 (508) 996-8561 (847) 699-3430 (800) 554-5565 Figure 6. 2.5V/3A converter switches at 1.4MHz Design Examples Figure 3 shows a typical application circuit using the LTC1778EGN. This 16-pin SSOP version of the part does not make all of the pin functions available. The VON input is internally set to 0.7V and the SENSE+ and SENSE– pins are cobonded with the SW and PGND pins, respectively. The circuit delivers a regulated 2.5V output at up to 10A from input voltages between 5V and 28V. The power MOSFETs from Siliconix are optimized for low duty cycle applications. The 1.4MΩ RON sets the 250kHz switching frequency. This switching frequency yields good efficiency with reasonable component sizes. Figure 4 shows that the efficiency of this circuit ranges from 90% to 95%, depending upon output current and input voltage. At light loads, below about 2A, the circuit enters discontinuous mode to keep the efficiency high. The response to a 1A to 10A load step is shown in Figure 5. Note the discontinuous mode operation with the 1A load and the rapid increase in inductor current after the load step. Figure 6 shows a very high switching frequency buck regulator that allows the use of small power components. This circuit delivers a 2.5V output at up to 3A while switching at Linear Technology Magazine • February 2001 Unlike many other current mode controllers, the LTC1778 can also be used in applications with a high output voltage, nearly up to the full input voltage. Figure 7 illustrates this with a 12V output circuit that can deliver up to 5A. This circuit uses the LTC1778EGN-1, which replaces the PGOOD pin with the VON pin. Tying this pin high sets the internal VON level to 2.4V, reducing the required value of the RON resistor for 300kHz operation. This circuit has excellent efficiency, reaching 97% at 5A with a 24V VIN. 1.4MHz. The minimum off-time constraint limits the duty cycle in this circuit to below 30%, as illustrated in Figure 2. Thus, the minimum permissible VIN to avoid dropout is 9V. A pair of low-gate-charge MOSFETs in a single SO-8 package was chosen to minimize the significant switching losses at this high frequency. Efficiency runs about 80% to 85% with a 12V input. CSS 0.1µF 1 2 3 CC1 2.2nF 4 RC 20k 5 CC2 100pF 6 7 R1 10k R2 140k 8 C2 2200pF LTC1778-1 RUN/SS VON VRNG FCB ITH SGND BOOST TG SW 16 15 Many low voltage microprocessors now require digital control of the output voltage and active voltage positioning to improve load transient response. The LTC3711 specifically addresses these needs. It uses the LTC1778 control architecture for handling the low duty cycles while adding a 5-bit VID interface. The VID code selects an output voltage in the range of 0.9V to 2.0V, compatible with Intel mobile Pentium® processors. The LTC1778 and LTC3711 both include a trimmed error amplifier Pentium is a registered trademark of Intel Corp. DB CMDSH-3 CB 0.22µF 14 CIN 22µF 50V M1 L1, 10µH + PGND BG INTVCC ION VIN VFB EXTVCC 13 M2 D1 VIN 14V TO 28V VOUT 12V 5A COUT 220µF 16V 12 11 + 1 + CSS 0.1µF CVCC 4.7µF 10 RF 1Ω 9 CF 0.1µF RON 1.6M CIN: UNITED CHEMICON THCR70E1H226ZT COUT: SANYO 16SV220M L1: SUMIDA CDRH127-100 M1, M2: FAIRCHILD FDS7760A D1: DIODES, INC. B340A (847) 696-2000 (619) 661-6835 (847) 956-0667 (408) 822-2126 (805) 446-4800 Figure 7. 12V/5A converter switches at 300kHz 19 DESIGN FEATURES 1 CSS 0.1µF 2 3 RRNG1 RRNG2 4.99k 45.3k RPG 100k 4 5 VID2 VID1 RUN/SS VID0 BOOST VON TG PGOOD SW VRNG 24 23 DB CMDSH-3 22 CB 0.33µF 21 M1 M2 ×2 7 RVP1 12.4k CC1 180pF 8 9 CFB 100pF 10 11 RON 330k 12 FCB ITH SENSE+ PGND SGND ION BG INTVCC VFB VOSENSE VIN EXTVCC VID3 VID4 D1 + 19 VIN 7V TO 24V VOUT 1.5V 15A COUT 270µF 2V ×3 RSENSE 0.003Ω 18 17 16 + 6 L1 1µH 20 LTC3711 RVP2 40.2k CIN 22µF 50V ×3 CVCC 4.7µF RF 1Ω 15 14 CF 0.1µF 13 CIN: UNITED CHEMICON THCR70E1H26ZT (847) 696-2000 COUT: CORNELL DUBILIER ESRE271M02B (508) 996-8561 L1: SUMIDA CEP125-IR0MC-H (847) 956-0667 M1: INTERNATIONAL RECTIFIER IRF7811A (310) 332-3331 D1: MICROSEMI UPS840 (617) 926-0404 Figure 8. 1.5V/15A CPU core voltage regulator with active voltage positioning Conclusion transconductance that is constant over temperature. This feature allows more aggressive compensation of the control loop for faster transient response as well as enabling accurate active voltage positioning. Active voltage positioning lowers the output voltage in a controlled manner as the load current increases. This is useful in microprocessor power supplies where large load current transients are the main cause of output voltage error. An example of a VID controlled LTC3711 application with active voltage positioning is shown in Figure 8. To facilitate the voltage positioning, the SENSE+ pin is used with a current sense resistor at the source of M2. The voltage positioning gain is accurately set using resistors RVP1 and RVP2 along with the trimmed transconductance of the error amplifier. This circuit positions the output voltage about 65mV above a 1.5V nominal output at no load, drooping to 65mV below the nominal output at full load. Voltage positioning allows the number of output capacitors to be reduced from five to three and still maintain a ±100mV specification on the output voltage. The LTC1778/LTC3711 step-down DC/DC controllers are designed for power supplies operating over a wide input and output range. The valley current control architecture enables very low voltage outputs to be obtained from high input voltage sources such as battery packs and wall adapters. Eliminating the sense resistor improves efficiency and saves both board space and component cost. The LTC1778 and LTC3711 are excellent choices for delivering the low output voltages and high efficiencies required by modern portable power supplies. for the latest information on LTC products, visit www.linear-tech.com 20 Linear Technology Magazine • February 2001 DESIGN FEATURES New UltraFast Comparators: Rail-toRail Inputs and 2.4V Operation Allow Use on Low Supplies by Glen Brisebois Introduction The new LT1711 family of UltraFast comparators has fully differential railto-rail inputs and outputs and operates on supplies as low as 2.4V, allowing unfettered application on low voltages. The LT1711 (single) and LT1712 (dual) are specified at 4.5ns of propagation delay and 100MHz toggle frequency. The low power LT1713 (single) and LT1714 (dual) are specified at 7ns of propagation delay and 65MHz toggle frequency. All of these comparators are fully equipped to support multiple-supply applications, and have latch-enable pins and complementary outputs like the popular LT1016, LT1671 and LT1394. They are available in MSOP and SSOP packages, fully specified over commercial and industrial temperature ranges on 2.7V, 5V and ±5V supplies. rent sources and sinks feeding the NPN and PNP differential pairs formed by Q3–Q4 (protected by fast diodes D11–D12) and Q1–Q2 (protected by D1–D2). This approach makes the inputs truly fully differential and noninteracting, unlike approaches that resort to resistors and diode clamps. Even with the inputs at opposite rails, the input bias currents are still a simple function of the input transistor base currents and remain in the µA region. Both input stages feed the level shifting transistors Q5– Q6, and the remainder of the differential voltage gain circuit flows with a delightful symmetry towards the output. Note that the channels are identical, with polarity yet unassigned, and are therefore interchangeable in layout. The symmetry, broken only by the latch-enable circuit, is enhanced by the fact that all of the transistors are well matched, complementary 6GHz fT BJTs. Each output stage ends in two Bakerclamped common emitter transistors, Circuit Description Figure 1 shows a simplified schematic of the LT1711 through LT1714. The front end consists of eight cur- allowing full rail-to-rail output swing. All the comparators guarantee full 5V TTL output capability over temperature, even when supplied with only 3V. Output rise and fall times are fast, at 2ns for the LT1711 and LT1712 and 4ns for the LT1713 and LT1714. Jitter is among the lowest for any monolithic comparator, at 11psRMS for the LT1711 and LT1712 and 15psRMS for the LT1713 and LT1714. Some Applications Simultaneous Full-Duplex 75MBaud Interface with Only Two Wires The circuit of Figure 2 shows a simple, fully bidirectional, differential 2-wire interface that gives good results to 75MBaud, using the low power LT1714. Eye diagrams under conditions of unidirectional and bidirectional communication are shown in Figures 3 and 4. Although not as pristine as the unidirectional VCC I1 I2 I9 R1 D4 R2 D35 Q5 Q6 Q10 D15 GND VCC Q34 Q22 Q13 D38 D14 Q4 OUTB1 I5 Q3 D9 D10 D11 Q8 Q42 D18 I14 Q14 R3 I6 I7 R4 D39 Q41 Q15 I11 Q7 D12 GND D33 OUTA1 D37 D13 D32 D36 Q35 D30 VEE VCC VCC Q29 Q23 Q17 BIAS INB1 VCC Q30 Q28 Q16 Q12 D31 R9 Q19 Q9 VEE VEE VCC Q31 R8 Q11 Q2 VEE Q1 Q18 LE1 D29 I25 I18 D34 D28 INA1 R6 I13 VCC I10 I4 R5 I12 VEE VCC D3 I24 I23 I3 D2 D1 D27 D17 R11 Q43 R12 I16 I8 VEE GND Q40 I15 I21 I22 I20 Figure 1. LT1711–LT1714 simplified schematic Linear Technology Magazine • February 2001 21 DESIGN FEATURES 3V 4 + 14 2 2 3V 1/2 LT1714 RXD ALL DIODES = BAV99 – 13 16 1 – 3 R2a 2.55k 3V 15 + 11 R3a 124Ω 1/2 LT1714 TXD 8 – 6 10 12 9 R3b 124Ω R2b 2.55k R1b 499Ω RXD 16 13 3 R1a 499Ω R1c 499Ω R2c 2.55k R0a 140Ω R0b 140Ω R3c 124Ω 3V 5 5 7 14 1/2 LT1714 1 15 3V 4 + 3V SIX FEET TWISTED PAIR ZO 120Ω + 11 7 TXD 1/2 LT1714 R3d 124Ω R1d 499Ω 12 – 6 8 10 9 R2d 2.55k Figure 2. 75MBaud full-duplex interface on two wires per for mance of Figure 3, the performance under simultaneous bidirectional operation is still excellent. Because the LT1714 input voltage range extends 100mV beyond both supply rails, the circuit works with a full ±3V of ground potential difference. The circuit works well with the resistor values shown, but other sets of values can be used. The starting point is the characteristic impedance, ZO, of the twisted-pair cable. The input impedance of the resistive network should match the characteristic impedance and is given by: RIN = 2 • RO • (R1||(R2 + R3) (RO + 2 • (R1||(R2 + R3))) This comes out to 120Ω for the values shown. The Thevenin equivalent source voltage is given by: 5ns/DIV 5ns/DIV Figure 3. Performance of Figure 1’s circuit operated unidirectionally; eye is wide open (cursors show bit interval of 13.3ns or 75MBaud). Figure 4. Performance of Figure 1’s circuit operated simultaneous-bidirectionally; crosstalk appears as noise. Eye is slightly shut but performance is still excellent. C4 100pF R5 7.5k V • (R2 + R3 – R1) VTh = S • (R2 + R3 + R1) R6 162Ω 22 C3 100pF R7 15.8k VS RO (RO + 2 • (R1||(R2 + R3))) This amounts to an attenuation factor of 0.0978 with the values shown. (The actual voltage on the lines will be cut in half again due to the 120Ω ZO.) This attenuation factor is important because it is the key to deciding the ratio of the R2, R3 resistor divider in the receiver path. This divider allows the receiver to reject the large signal of the local transmitter and instead sense the attenuated signal of the remote transmitter. Note The author having already designed R2 + R3 to be 2.653kΩ (by allocating input impedance across RO, R1, and R2 + R3 to get the requisite 120Ω), R2 and R3 then become 2529Ω and 123.5Ω, respectively. The nearest 1% value for R2 is 2.55k and that for R3 is 124Ω. that in the above equations, R2 and R3 are not yet fully determined because they only appear as a sum. This allows the designer to now place an additional constraint on their values. The R2, R3 divider ratio should be set to one-half of the attenuation factor mentioned above or R3/R2 = 1/2 • 0.0976.1 2 – R9 2k 3 + C2 0.1µF R8 2k VS R1 1k 2 R2 1k R4 210Ω VS 6 SINE VS = 2.7V TO 6V 1 + 7 LT1713 3 LT1806 4 1MHz AT CUT VS 7 – 4 SQUARE 8 5 6 C1 0.1µF R3 1k Figure 5. LT1713 comparator configured as a series-resonant crystal oscillator; the LT1806 op amp is configured as a bandpass filter with a Q of 5 and fC of 1MHz. Linear Technology Magazine • February 2001 DESIGN FEATURES 1MHz Series-Resonant Crystal Oscillator with Square and Sinusoid Outputs Figure 5 shows a classic 1MHz seriesresonant crystal oscillator. At series resonance, the crystal is a low impedance and the positive feedback connection brings about oscillation at the series resonance frequency. The RC feedback to the – input ensures that the circuit does not find a stable DC operating point and refuse to oscillate. The comparator output is a 1MHz square wave (top trace of Figure 6), with jitter measured at 28psRMS on a 5V supply and 40 psRMS on a 3V supply. At pin 2 of the comparator, on the other side of the crystal, is a clean sine wave except for the presence of the small high frequency glitch (middle trace of Figure 6). This glitch is caused by the fast edge of the comparator output feeding back through crystal capacitance. Amplitude stability of the sine wave is maintained by the is the bottom trace of Figure 6. Distortion was measured at –70dBc and –55dBc on the second and third harmonics, respectively. A 3V/DIV B 1V/DIV Conclusion C 1V/DIV 200ns/DIV Figure 6. Oscillator waveforms with VS = 3V: Trace A = comparator output; Trace B = crystal feedback to pin 2 of the LT1713; Trace C = buffered, inverted and bandpass filtered output of LT1806 fact that the sine wave is basically a filtered version of the square wave. Hence, the usual amplitude-control loops associated with sinusoidal oscillators are not necessary.2 The sine wave is filtered and buffered by the fast, low noise LT1806 op amp. To remove the glitch, the LT1806 is configured as a bandpass filter with a Q of 5 and unity gain center frequency of 1MHz. The final sinusoidal output The fully differential rail-to-rail inputs of the new LT1711 family of fast comparators make them useful across a wide variety of applications. The high speed, low jitter performance of this family, coupled with their small package sizes and 2.4V operation, makes them attractive where PCB real estate is at a premium and bandwidth-topower ratios must be optimized. 1 Using the design value of R2 + R3 = 2.653k rather than the implementation value of 2.55k + 124Ω = 2.674k. 2 Amplitude will be a linear function of comparator output swing, which is supply dependent and therefore adjustable. The important difference here is that any added amplitude stabilization or control loop will not be faced with the classical task of avoiding regions of nonoscillation vs clipping. LT1618, continued from page 7 L1 10µH VIN 2.7V TO 5V D1 0.619Ω 80mA 90 85 9 10kHz TO 50kHz PWM BRIGHTNESS ADJUST 8 7 VIN SW SHDN ISP ISN R3 5.1k 4 VIN = 5V 80 3 2 R1 2M LT1618 IADJ FB VC GND 5 1 C2 1µF CC 0.1µF Linear Technology Magazine • February 2001 VIN = 2.8V 65 50 51Ω 51Ω 51Ω 51Ω 10 20 30 40 50 60 70 80 LED CURRENT (mA) Figure 10. High power white LED driver efficiency Figure 9. High power white LED driver For larger LCD displays where a greater amount of light output is needed, multiple strings of LEDs can be driven in parallel. When driving parallel strings, ballast resistors should be added to compensate for LED forward voltage variations. The amount of ballasting needed depends on the LEDs used and how well they 70 55 R2 121k (408) 573-4150 (408) 573-4150 (800) 282-9855 (847) 956-0666 High Power White LED Driver VIN = 3.3V 75 60 10 C3 0.1µF C1: TAIYO YUDEN JMK212BJ475 C2: TAIYO YUDEN TMK316BJ105 D1: ON SEMICONDUCTOR MBR0530 L1: SUMIDA CR43-100 EFFICIENCY (%) C1 4.7µF are matched. The circuit in Figure 9 is ideal for larger displays, providing constant current drive for twenty white LEDs from a single Li-Ion cell. Efficiency reaches a respectable 82%, as seen in Figure 10. Conclusion The constant-current/constant-voltage operation of the LT1618 makes the device an ideal choice for a variety of constant-current designs. The device provides accurate output current regulation or input current limiting, along with excellent output voltage regulation. With a wide input voltage range and the ability to produce outputs up to 35V, the LT1618 works well in many different applications. References 1. Kim, Dave. “Tiny Regulators Drive White LED Backlights.” Linear Technology Design Note 231 (May 2000). 23 DESIGN FEATURES High Efficiency Synchronous PWM Controller Boosts 1V to 3.3V or 5V by San-Hwa Chee Introduction CPU power supplies continue to fall toward the 1V level, although other circuits still require the traditional 3.3V or 5V rails. Since the LTC1700 is capable of operating at an input voltage as low as 0.9V, it can boost the latest CPU power supply voltages to provide the missing 3.3V or 5V rail. The LTC1700 uses a constant frequency, current mode PWM architecture but does not require a current sense resistor; instead, it senses the VDS across the external N-channel MOSFET. This reduces component count and improves high load current efficiency. Efficiency is further increased at high load currents through the use of a synchronous P-channel MOSFET. With Burst Mode operation selected, efficiency at low load currents is enhanced, thereby providing high efficiency over the C1 22µF L1 1.8µH 2 R3 220Ω C3 220pF C5 220pF C4 0.1µF 4 1 3 5 R2 100k ITH LTC1700 RUN/SS SGND VFB SYNC/MODE SW BG PGND TG VOUT C2 68µF 6.3V VIN 3.3V TO 4.2V M2 10 8 + C6 10µF M1 + 9 6 7 R1 316k C1: TAIYO YUDEN LMK432BJ226MM CERAMIC C2: AVX TAJB686K006R C6: TAIYO YUDEN JMK316BJ106ML CERAMIC C7: SANYO POSCAP 6TPB330M L1: TOKO 919AS–1R8N (D104C TYPE) M1: SILICONIX Si9804 M2: SILICONIX Si9803 (408) 573-4150 (207) 282-5111 (619) 661-6835 (800) 554-5565 VOUT 5V/2A C7 330µF 6V entire load current range. The LTC1700 operates at 530kHz but can be externally synchronized to frequencies between 400kHz and 750kHz. During continuous mode operation, the LTC1700 consumes 540µA; it drops to 180µA in Sleep mode. In shutdown, the quiescent current is just 10µA. The LTC1700 is available in the 10-lead MSOP package. 3.3V to 4.2V Input, 5V/2A Output Regulator Figure 1 shows a 10W output application circuit. Since the LTC1700 is operating at 530kHz, a small valued inductor is sufficient for this circuit. The input capacitors consist of a small 22µF ceramic capacitor in parallel with a B-case size tantalum capacitor. The ceramic capacitor provides a low overall ESR while the tantalum provides the bulk capacitance. In applications where the input is connected very close to a low impedance supply, the input tantalum capacitor may not be needed. A Sanyo POSCAP capacitor is used for the output capacitor because of its high ripple Figure 1. 3.3 to 4.2V input, 5V/2A output converter OUTPUT VOLTAGE RIPPLE (AC COUPLED) 100mV/DIV 100 EFFICIENCY (%) 90 VIN = 4.2V 80 VIN = 3.3V 70 INDUCTOR CURRENT 2A/DIV 60 50 0.001 1.000 0.100 0.010 LOAD CURRENT (A) Figure 2. Efficiency of Figure 1’s circuit 24 VIN = 3.7V LOAD STEP = 100mA TO 1.75A 0.2ms/DIV Figure 3. Load-step response of Figure 1’s circuit Linear Technology Magazine • February 2001 DESIGN FEATURES VOUT 1.205V REFERENCE 7 VIN + C1 PARASITIC DIODE OF MOSFET C2 L1 + – SC VOUT SW = 1 WHEN VOUT = <2.3V 10 + C6 C7 SHDN TO TG START-UP OSCILLATOR S QB L1 R M1 Q + ICMP 60mV 1Ω – LTC1700 Figure 4. Start-up components of the LTC1700 Start-Up and MOSFET Selection When the voltage at the VOUT pin is below 2.3V, the LTC1700 operates in the start-up mode. In this mode, only the start-up circuitry in the LTC1700 is active and both of the external MOSFETs are turned off. Figure 4 shows the components that control start-up. In this mode, the current limit is set at 60mA and the internal MOSFET is used to bring the output voltage up. The start-up oscillator, C1 10µF L1 4.6µH 220pF 2.2k 1 2 270pF 470pF 4 3 R2 100k R1 95.3k 5 SGND LTC1700 SW BG ITH PGND RUN/SS TG VFB VOUT SYNC/MODE M1 9 6 7 150pF C1: TAIYO YUDEN JMK316BJ106ML CERAMIC C2: AVX TAJB686K006R C3: TAIYO YUDEN JMK325BJ226M CERAMIC C4: SANYO POSCAP 6TPA150M L1: SUMIDA CEP1234R6 M1: SILICONIX Si6466 M2: FAIRCHILD FDS6375 C2 68µF 6.3V VIN 3.3V M2 10 8 + C3 22µF + which is different from the main oscillator, runs at 210kHz at a duty cycle of 50%. Due to the low current limit, the output should not be heavily loaded during the start-up phase, as this will cause the output to “hang.” Once the output rises above 2.3V, the rest of the internal circuitry of the LTC1700 comes alive and the external MOSFETs begin switching. The start-up circuitry will then be shut down. In some applications, the input voltage is high enough that start-up mode is not needed, resulting in the VOUT 5V/3A C4 150µF 6.3V ×3 100 90 EFFICIENCY (%) current rating. Once again, a ceramic capacitor is used in parallel with the POSCAP for reduced ESR and high frequency decoupling. Figure 2 shows the efficiency curves for input voltages of 3.3V and 4.2V. Note that the maximum efficiency reaches 95% at a load current of 2A. A load step from 100mA to 1.75A was applied and its response is shown in Figure 3. 80 70 (408) 573-4150 (207) 282-5111 60 (619) 661-6835 (847) 956-0667 (800) 554-5565 (408) 822-2126 50 0.001 Figure 5. 3.3V input to 5V/3A output regulator Linear Technology Magazine • February 2001 1.000 0.100 0.010 LOAD CURRENT (A) Figure 6. Efficiency of Figure 5’s circuit 25 DESIGN FEATURES OUTPUT VOLTAGE RIPPLE (AC COUPLED) 100mV/DIV INDUCTOR CURRENT 2A/DIV VIN = 3.3V LOAD STEP = 300mA TO 2.6A 0.2ms/DIV Figure 7. Load-step response of Figure 5’s circuit circuit being able to power up at full load current. Figure 1 shows an example of this. The required input voltage to power up with full load current is: VIN > 2.3 + VF where VF is the forward voltage of the parasitic diode across the external P-channel MOSFET, which is dependent on the load current. For a load current less than 3A, a VF of 0.75V can be used. Since the switchover from the internal MOSFET to the external MOSFETs occurs at VOUT = 2.3V, the MOSFETs selected should have a Figure 5 shows a 3.3V input to 5V output circuit that can provide a maximum of 3A output current. Like the circuit in Figure 1, this circuit will bypass the start-up mode and therefore is capable of starting up at full load. Figure 6 shows that its efficiency reaches 88% at load currents of 2A to 3A. Figure 7 shows the load step response. 1 2 200pF 0.1µF 4 3 100k The circuit shown in Figure 8 is capable of supplying 4A of load current. To obtain this output current capability, two IRF7811A N-channel MOSFETs are paralleled to obtain the required peak inductor current. Two Si9803DY are used for the synchronous P-channel MOSFET because of the amount of RMS current through these devices. The Si9803DYs are mounted on an area of copper adequate to effectively remove the maximum amount of heat. Due to the 3.3V Input, 5V/3A Output Regulator C1 22µF L1 0.68µH 220pF 2.2k 3.3V Input, 5V/4A Output Regulator threshold of 2.5V or lower. This will guarantee a smooth transition out of the start-up mode. 5 SGND LTC1700 SW BG ITH RUN/SS PGND TG VFB SYNC/MODE VOUT 9 6 C2 68µF 6.3V VIN 3.3V M2 10 8 + M1 C5 4.7µF C3 22µF + VOUT 5V/4A C4 150µF 6.3V ×2 7 316k C1: TAIYO YUDEN JMK316BJ226ML CERAMIC C2: AVX TAJB686K006R C3: TAIYO YUDEN JMK325BJ226M CERAMIC C4: PANASONIC EEUEOJ151R L1: SUMIDA CDEP134-0R6NC-H M1: SILICONIX Si9803 ×2 M2: INTERNATIONAL RECTIFIER IR7811 ×2 (408) 573-4150 (207) 282-5111 (714) 373-7334 (847) 956-0667 (800) 554-5565 (408) 822-2126 Figure 8. 3.3V input, 5V/4A output regulator 26 Linear Technology Magazine • February 2001 DESIGN FEATURES L1 2.2µH 470pF 33k 1 2 300pF 470pF 4 3 30k 5 SGND LTC1700 SW BG ITH PGND RUN/SS TG VFB SYNC/MODE VOUT C2 68µF 6.3V C3 22µF M1 9 6 100 90 M2 10 8 + + EFFICIENCY (%) C1 10µF VIN 2.5V VOUT 3.3V/1.8A C4 220µF 6.3V Burst Mode OPERATION ENABLED 80 70 Burst Mode OPERATION DISABLED 60 7 50 53.6k 40 0.001 C1: TAIYO YUDEN JMK316BJ106ML CERAMIC C2: AVX TAJB686K006R C3: TAIYO YUDEN JMK325BJ226M CERAMIC C4: KEMET T520D227M00AS L1: MURATA LQN6C M1: SILICONIX Si9802 M2: SILICONIX Si9803 0.100 0.010 LOAD CURRENT (A) 1.000 (408) 573-4150 (207) 282-5111 (408) 986-0424 (814) 237-1431 (800) 554-5565 Figure 10. Efficiency of Figure 9’s circuit Figure 9. 2.5V input, 3.3V/1.8A output regulator 2.5V Input, 3.3V/1.8A Output Regulator high RMS ripple current going into the output capacitors, two Panasonic SP capacitors are required. The maximum efficiency of 92% occurs at load currents between 2A to 3A. This circuit has no problem starting up into a load that exhibits a resistive characteristic. Figure 9 shows a circuit that takes an input of 2.5V and steps it up to 3.3V. Both MOSFETs are selected with a guaranteed threshold voltage of 3V. Its efficiency and load step response are shown in Figures 10 and 11, respectively. Due to its low input voltage, this circuit cannot start up into a heavy load. Conclusion Through the use of VDS sensing and a synchronous topology, the LTC1700 provides high efficiency at high load currents. Selectable Burst Mode operation allows high efficiency to be obtained at low load currents. With its low operating voltage, the LTC1700 can easily be used to step up low voltages to the traditional 3.3V or 5V supply rails. OUTPUT VOLTAGE RIPPLE (AC COUPLED) 100mV/DIV INDUCTOR CURRENT 2A/DIV VIN = 2.5V LOAD STEP = 100mA TO 1.6A 100µs/DIV Figure 11. Load-step response of Figure 9’s circuit Linear Technology Magazine • February 2001 27 DESIGN INFORMATION Rail-to-Rail 14-Bit Dual DAC in a Space Saving 16-Pin SSOP Package by Hassan Malik Linear Technology introduces the LTC1654, a 14-bit rail-to-rail voltage output dual DAC in a space saving 16-pin SSOP package. This part offers a convenient solution for applications where density, resolution and power are critical parameters. The LTC1654 is guaranteed to be 14-bit monotonic over temperature with a typical differential nonlinearity of only 0.3LSB. The supply voltage range is 2.7V to 5.5V. The LTC1654 is software programmable for two different speed/power modes of operation: a FAST mode with 3.5µs settling time and 750µA supply current and a SLOW mode with 8µs settling time and 450µA supply current. Either of the two DACs can be independently set to the FAST or the SLOW mode of operation. The output amplifiers swing to within 450mV of either supply rail when sourcing or sinking 5mA and are capable of driving over 300pF of load capacitance. The output noise voltage density at 1kHz is 540nV/√Hz in SLOW mode and 320nV/√Hz in FAST mode. The LTC1654 has separate REFHI and REFLO pins for each DAC and two different gain modes. A gain of one is set by connecting the X1 /X1/2 pin to REFLO and a gain of one-half is set by connecting this pin to VOUT. The two different gain modes allow the user to set different output spans. The REFHI inputs have an operating range from ground to VCC and the REFLO inputs have an operating range from ground to VCC – 1.5V. A block diagram of the part is shown in Figure 1. The LTC1654 allows each of the DACs to be individually shut down, in which state they consume less than 4µA/DAC. The REFHI input goes into a high impedance state when the DAC is in shutdown. The respective speed states are retained in shutdown as long as the supply voltage is maintained above the minimum value of 2.7V. When the supply voltage is first applied, both DACs are active and in SLOW mode, with all zeros loaded in the input shift register and DAC latches. The LTC1654 has a double-buffered 3-wire serial interface consisting of clock, data and chip select pins. This interface is SPI/QSPI and continued on page 33 MICROWIRE is a trademark of National Semiconductor Corp. 2.7V TO 5.5V 16 5 CS/LD 3 SCK 4 SDI LTC1654 VCC REFHI B 14 VOUT B 15 CONTROL LOGIC µP INPUT LATCH DAC REGISTER + DAC B – 32-BIT SHIFT REGISTER INPUT LATCH VCC 7 SDO 2 CLR DAC REGISTER + DAC A X1/X1/2 B 1 REFHI A 10 VOUT A 9 X1/X1/2 A 8 – POWER-ON RESET DGND 6 REFLO B REFLO A 13 11 AGND 12 Figure 1. LTC1654 block diagram 28 Linear Technology Magazine • February 2001 DESIGN IDEAS Charge Pump Powers White LEDs by Steven Martin Introduction Most of today’s portable equipment uses a liquid crystal display to convey information to its user. Until recently, those displays have been monochrome and have used a low voltage light emitting diode (LED) such as red or green for backlighting. With the advent of color liquid crystal displays, a white backlight source is needed. Recently, a revolution in LED technology has lead to the long-coveted white LED. However, white LEDs, which are actually blue LEDs with a special phosphor in the lens, operate at a considerably higher voltage than red or green LEDs. White LEDs can require anywhere from 3.5V to 3.9V to operate at 15mA. In battery powered systems, it’s impractical to drive the LEDs directly from the battery and still control the LED current. To properly control the LED current a somewhat larger voltage is needed, where the excess is used for control. DESIGN IDEAS Charge Pump Powers White LEDs ................................................... 29 Steven Martin 48V Hot Swap Circuit Blocks Reverse Battery Voltage ........................... 30 Mitchell Lee Using the LTC1662 3µ A DAC to Cure RF Implementation Ills ................ 31 Derek Redmayne LTC1628-SYNC Minimizes Input Capacitors in a Multioutput, High Current Power Supply ................. 34 Figure 1. LTC3200 evaluation circuit Figure 1 shows the LTC3200-5 constant frequency voltage doubler used to drive five white LEDs. Figure 2 is the schematic diagram of this circuit. The LTC3200-5 produces a regulated 5V output from an input as low as 3V. Switching at 2MHz, the constant frequency operation of the charge pump is ideal for low noise environments such as cellular telephones or internet communication devices. Since it produces a regulated 5V output, the additional voltage dropped across the resistors controls the LED current. The resistors also provide ballasting to ensure that the LEDs Wei Chen continued on page 37 1µF Reduce EMI with Ultralow Noise 48V to 5V, 10W DC/DC Converter ........ 36 4 Rick Brewster 5 3V TO 4.4V Li-Ion BATTERY 1µF C– VIN 6 C+ 1 VOUT UP TO 5 LEDS 1µF LTC3200-5 3 ON OFF Authors can be contacted at (408) 432-1900 run at similar currents despite moderate differences in forward voltage. Figure 3 shows how the adjustable LTC3200 can be used to control the LED current directly by controlling the voltage on the ballast resistors. The LTC3200 regulates the anodes of the LEDs until the FB pin comes to balance at 1.268V. The feedback LED’s current is precisely controlled and the remaining LEDs are moderately well controlled by virtue of their similarity and the 1.268V ballast voltage. Since the current is more precisely controlled, up to six LEDs can be powered by the adjustable LTC3200. SHDN (APPLY PWM WAVEFORM FOR ADJUSTABLE BRIGHTNESS CONTROL) GND VSHDN 100Ω 100Ω 100Ω 100Ω 100Ω 2 t Figure 2. Li-Ion battery-powered 5V white or blue LED driver Linear Technology Magazine • February 2001 29 DESIGN IDEAS 48V Hot Swap Circuit Blocks Reverse Battery Voltage Hot Swap controllers guard against inrush current and short circuits, but reverse battery installation is another matter. In central office applications, OR-ing diodes block reversed input voltages. In systems with a single power source OR-ing diodes become unnecessary; although a single diode could be retained for reverse conditions, its forward drop loss is a significant penalty. The circuit shown in Figure 1 eliminates the loss associated with a blocking diode. The LT1641 and Q1 handle familiar hot swap chores of undervoltage lockout, inrush control, short circuit protection and system reset, while Q2 and Q3 handle reverse input situations. Under positive input conditions, Q2’s body diode is forward biased and power reaches the LT1641 and Q1, allowing them to function in the normal manner. When the LT1641 is commanded to turn on, GATE (pin 6) and R7 slowly charge C1, thereby limiting the inrush current to CLOAD. The LT1641 drives both Q1 and Q2 fully into enhancement mode, minimizing losses and eliminating the drop in Q2’s body diode. Q3 is included as part of the circuitry that blocks reverse inputs, yet it must “get out of the way” when positive inputs are present. With a positive input, Q3’s emitter is pulled up, dragging along its base via D2. Since the base is slightly negative with respect to the emitter, Q3 is off. A small current flows through resistor R9 to ground, but this is of no consequence. If the LT1641 is in the off state, GATE pulls low. Current flows from the forward-biased collector -base junction of Q3 to the LT1641’s GATE pin, but is limited by R7. When the LT1641 turns on, GATE pulls up above the input supply and D3 * 36V–72V INPUT LONG RSNUB 100Ω D5 R7 1M C1, 10nF/160V R9 100k SHORT D4 INTERNATIONAL RECTIFIER IRF530 (310) 322-3331 ON SEMICONDUCTOR MPSA 42 (602) 244-6600 1N5245, 15V 1N4148 DIODES INC. SMAT70A (805) 446-4800 BAV21 BODY DIODE R1 35.7k 1% R6 1k 8 7 6 VCC SENSE GATE PWRGD 1 ON 3 LT1641 R2 1.24k 1% C3 100nF LONG CLOAD 100µF, TYP D1 Q3 D2 Q1, Q2: Q3: D1: D2: D3: D4–D5: * OUTPUT Q1 R5 10Ω R8 10Ω CSNUB 10nF the collector-base junction of Q3 becomes reverse-biased. Q3 is off, so no collector current flows and the LT1641’s GATE output is not loaded by Q3. If a reverse input polarity is applied, Q3 goes to work and ensures that Q2 is held off. Negative inputs pull the emitter of Q3 below ground. Q3 turns on, biased by R9, and the collector effectively shorts Q2’s gate and source. With Q2 in the off state, no current can flow into RS, the LT1641 or Q1. The circuit can take up to –100V in this condition. Of course, some leakage current does flow in Q2; this is absorbed by D3. This circuit can handle instantaneous steps from zero to ±75V. As shown, short circuit protection limits the output to 2.5A. Q1, Q2, and RS can be scaled to handle loads in excess of 1kW. * RS 0.02Ω Q2 by Mitchell Lee FB GND TIMER 4 5 RESET R3 35.7k 1% 2 R4 1.21k 1% C2 1µF 10V RTN Figure 1. Reverse-battery protected LT1641 48V Hot Swap circuit 30 Linear Technology Magazine • February 2001 DESIGN IDEAS Using the LTC1662 3µA DAC to Cure RF Implementation Ills by Derek Redmayne Introduction The small form factor and almost nonexistent power consumption of the LTC1662 dual 10-bit DAC make it attractive as a “tweak” to a design that has already gone through initial qualifications and requires some additional improvements. The small size may allow the system designer to add it to a design without altering those areas that have already passed UL or FCC certification. In other words, the part is so small that it can be added to a design without ripping up and rerouting the entire circuit. The power consumption of the LTC1662 is so low that even very restricted power budgets in isolated or solar powered applications remain intact, or even unchanged. The examples shown demonstrate only a few of many possible uses. Minimizing Carrier Feedthrough in IQ Modulation In Figure 1, the DAC adjusts bias voltage in an I and Q modulator to minimize carrier feedthrough. The output of the modulator is ideally one frequency only, the sum of the carrier and the modulation. This frequency is the upper sideband, where the carrier and lower sideband are suppressed. The two channels of the DAC allow I and Q to be adjusted independently. In addition to being suitable for sealed, encapsulated, immersed or otherwise physically inaccessible locations, the LTC1662 does not have the shock and vibration sensitivity of potentiometers and is less expensive than a good potentiometer. If the I and Q components are added (the Q being a 90-degree-shifted version of both the carrier and the modulation), the lower sideband is suppressed and a single sideband (SSB) results. A single sideband is used in many RF applications, as it Linear Technology Magazine • February 2001 allows all the transmitted energy to be contained within the one sideband. Without the carrier present, at low modulation depths, minimal energy is required. Single sideband also occupies less spectrum for a given bandwidth. This technique can also be used to synthesize frequencies. The balanced modulator/demodulator will produce carrier feedthrough if a DC offset is present at the modulation terminal. The sum and difference of the carrier and the DC In Figure 2a, the DAC adjusts gain and phase by use of a varactor diode. Because it is not a multiplying DAC, it is not readily adaptable for direct DC or low frequency gain control, at least not high precision gain control over wide dynamic ranges. It can, however, be used to subtly adjust gain in cases where an excitation current needs to be adjusted; in the case of high frequencies, it can be used in conjunction with varactor diodes, PIN diodes or modulator/ I INPUT OPTIONAL LTC1662 CS/LD SCK DIN 5V 1 8 2 7 3 6 4 5 3µA DUAL DAC 20k 2.74k 1% R1 100k R2 2.74k 1% 0.1µF 1µF 5V 2.74k 1% 2.74k 1% 5V 20k 0.1µF OPTIONAL offset is fC, which is carrier feedthrough. One section of the DAC provides an adjustment range for the I section, via an attenuator that is just adequate to cover the worst-case offset of the modulator/demodulator. The other half of the DAC is used in the same manner for the Q section. Gain/Phase “Tweaks” Made Easy The setting for these DACs can be determined in manufacturing, stored and simply initialized each time power is applied. In a more complex design, the carrier feedthrough could be evaluated in the field and adjusted in real time or as temperature changes. LO 90° I + Q MODULATOR RF 0° 5V 100k 2.74k 1% 2.74k 1% 2.74k 1% 2.74k 1% Q INPUT Figure 1. Using the LTC1662 to trim for minimum carrier feedthrough 31 DESIGN IDEAS 5V 5V 70MHz ATTENUATION CONTROL PHASE CONTROL 1k 1k 10k 10k 1/2 LTC1662 1/2 LTC1662 0.1µF 10k 0.01µF 10k 0.1µF 20Ω 70MHz IN 49.9Ω 0.01µF OUT 47pF ZC830* 10pF 49.9Ω 20pF 49.9Ω ZC830* *ZETEX (516) 543-7100 Figure 2a. 70MHz attenuation control 6dB DIRECTIONAL COUPLER A 70MHz RF IN B 5V 70MHz RF OUT C 49.9Ω 6 1µF 4 SAMPLED 70MHz 8 10k 1/2 LTC1662 20pF 10k 0.1µF DELAY NTE610* 2 *NTE (973) 748-5089 Figure 2b. Using a varactor to tweak delay through a directional coupler demodulators to adjust IF or RF gain. It can also be used to tweak VCOs, phase-locked loops, biasing schemes for power devices, avalanche photodiodes or optical tuning or modulating devices that rely on electrostatic fields. The circuit in Figure 2a produces a range of phase adjustment of approximately 2 degrees and a range of amplitude adjustment of 3.5dB at 70MHz. The two adjustments are somewhat interactive. Depending on physical implementation, this circuit produces nominally zero degrees of phase shift at 70MHz and nominal insertion loss of 12dB. Figures 2b–2d show a few additional examples of how a micropower DAC can be used to produce subtle changes in gain or phase in RF circuitry. This type of tweak is often required in RF circuitry; if it is required at inaccessible points in a device or when installed in inaccessible locations, the ability to implement these tweaks remotely, with no space or power penalty, is valuable. The circuit in Figure 2b shows how one may adjust the phase of a signal from the isolated port of a 6dB directional coupler over a range of approximately 2.5 degrees at 70MHz. This is of the order that may be required to compensate for phase variation in the coupler or delay variation in transmission lines. The photos taken from a network analyzer (Figures 3a and 3b) show the maximum phase adjustment range vs frequency, as well as the variation in the attenuation factor over the range of phase adjustment. The examples shown in Figures 2c and 2d show examples of varactors used to reduce even-order harmonic distortion by using cancellation in two diodes. Figure 2d produces greater delay than Figure 2c and the circuit may be extended to produce yet more delay. In all of these circuits, the performance is very much dependent on physical implementation. 10V 10k 10k 1/2 LTC1662 10k 1/2 LTC1662 100pF 1µF ZC830* 10k 1000pF 0.1µF 10k 1000pF 49.9Ω 0.01µF IN 49.9Ω OUT 49.9Ω 10k ZC830* *ZETEX ZC830* 0.1µF IN OUT 20pF 49.9Ω 49.9Ω ZC830* 49.9Ω 10k *ZETEX (516) 543-7100 Figure 2c. Low distortion phase control 32 20pF 0.1µF 0.01µF (516) 543-7100 Figure 2d. Greater phase control Linear Technology Magazine • February 2001 DESIGN IDEAS In all cases, the adjustment range of trimmer circuits should be minimized, either by adjustment of the DAC output signal or by padding the varactor with series and parallel capacitance. This will reduce the phase noise contribution from the DAC and offset variation effects, as well as to minimize the harmonic distortion introduced by the varactor. The wideband noise of the LTC1662, as with any micropower DAC, is a potential pitfall in these applications, as it may degrade phase noise performance; however, the Figure 3a. Range of phase shift vs frequency for Figure 2b’s circuit LTC1662 has the virtue of very low 1/f noise. Bandwidth limiting the output to less than 1Hz will produce the benefit of low noise. If your application is not sensitive to low frequency phase noise, the output filters may not be necessary. Figure 3b. Maximum gain variation vs frequency for Figure 2b’s circuit 2.0 2.0 INL (LSB) 1.0 0.5 0 –0.5 –1.0 –1.5 –2.0 0 8192 16383 INPUT CODE Figure 2. LTC1654 typical integral nonlinearity (INL) LTC1654, continued from page 28 MICROWIRE™ compatible. The maximum clock rate is 33MHz. Double buffering allows individual load and update capability for each DAC. There are three different methods for loading data into the serial interface: a 24-bit word without daisy chaining, a 32-bit word without daisy chaining and a 32-bit word with daisy chaining. The 24-bit word loading method requires eight bits for control and address followed by sixteen bits of data. The last two LSBs in the 16-bit Linear Technology Magazine • February 2001 data segment are “don’t cares.” The 32-bit word loading method without daisy chaining requires eight “don’t cares” followed by eight bits for control and address and sixteen bits of data. The 32-bit word loading method with daisy chaining is the same as above except that the DOUT pin is used. When the REFHI pins are connected to VCC and the LTC1654 is configured for a gain of one, the voltage outputs swing from ground to VCC. The typical differential and integral nonlinearities are shown in Figures 2 and 3, respectively. VOUT is as follows for the two different gain configurations: Gain of one (X1 /X1/2 pin connected to REFLO): VOUT = (VREFHI – VREFLO) • (SDI/16384) + VREFLO where SDI is the decimal representation of the digital data input. Gain of one-half (X 1 /X 1/2 pin connected to VOUT ): 1.5 1.0 DNL (LSB) 1.5 0.5 0 –0.5 –1.0 –1.5 –2.0 0 8192 16383 INPUT CODE Figure 3. LTC1654 typical differential nonlinearity (DNL) VOUT = (1/2)(VREFHI – VREFLO) • (SDI/16384) + VREFLO In any rail-to-rail DAC, the output swing is limited to voltages within the supply range. If the DAC offset is negative, the output for the lowest codes limits at 0V. Similarly, limiting can occur near full-scale when the REFHI pin is tied to VCC. This can be avoided by ensuring that VREFHI is less than VCC by at least 15mV or by using the gain of one-half configuration. 33 DESIGN IDEAS LTC1628-SYNC Minimizes Input Capacitors in a Multioutput, High Current Power Supply C2 1000pF INTVCC1 by Wei Chen 10Ω 10Ω 1µF 10Ω 1µF LTC1629 0.1µF 1 10k 0.01µF RUN/SS 2 SENSE1+ 3 – 2.7k 6 49.9k 7 10k 8 680pF 9 100pF 56pF R14 6.98k 10Ω C20 1000pF 10Ω VIN PLLIN BG1 PHASMD EXTVCC ITH INTVCC PGND VDIFFOUT BG2 28 L1 1.0µH Q1 26 25 R9 0.002Ω 0.47µF D1 24 23 Q2–Q3 22 21 20 18 14 SENSE2+ TG2 16 11 VOS– SW2 12 VOS+ D2 + C11 1µF C12 10µF 6.3V 19 SENSE2– BOOST2 AMPMD CLK1 27 13 VO1SENSE– VO1SENSE+ EAIN SGND 10 R13 8.06k SW1 PLLFLTR BOOST1 4 12.1k TG1 SENSE1 5 47pF CLKOUT 0.47µF 1µF L2 1.0µH Q4 17 R15 0.002Ω VOUT1+ 1.5V/60A MAX 15 Q5– Q6 D3 + 10Ω C22–C27 270µF, 2V ×7 10Ω VOUT1– 10Ω C35 1000pF 10Ω 10Ω 1µF 16V VIN+ 12V LTC1628-SYNC 1 47pF SENSE1+ TG1 3 SENSE1– SW1 5 10k CLK1 6 7 0.01µF 56pF 8 9 0.1µF 10 1000pF 10k 11 12 100pF 13 R28 8.06k 14 C66 47pF C59 1000pF R32 17.4k 470pF PGOOD 2 4 1nF RUN/SS1 VOSENSE1 BOOST1 PLLFLTR PLLIN FCB ITH1 SGND 3.3VOUT ITH2 VIN BG1 EXTVCC INTVCC PGND BG2 BOOST2 VOSENSE2 TG SENSE2– SW2 SENSE2+ RUN/SS2 10Ω 28 1µF PGOOD R22 100k 27 L3 1.0µH Q7 26 R23 0.002Ω CIN 270µF 16V ×3 VIN– 25 0.47µF D4 24 Q8– Q9 23 22 + C51 1µF Q1–Q12: FAIRCHILD FDS7764A L1–L3: SUMIDA CEP125-1R0-MC-H L4: SUMIDA CEP125-1R5-MC-H D1, D3, D4, D6: ON SEMICONDUCTOR MBRS340T3 (602) 244-6600 D2, D5: ON SEMICONDUCTOR BAT54A D5 21 C52 10µF 6.3V 20 19 0.47µF 18 16 1µF L4 1.5µH Q10 17 15 C60 0.1µF R26 0.003Ω + Q11– Q12 D6 VOUT2+ 2.5V/15A C62–C63 180µF, 4V ×2 VOUT2– 10Ω 10Ω VO2SENSE+ Figure 1. LTC1628-Sync/LTC1629 2-output PolyPhase supply 34 Linear Technology Magazine • February 2001 DESIGN IDEAS current seen by the input bus is further reduced. In addition, the differential amp within the LTC1629 enables true remote sensing to ensure accurate voltage regulation at the CPU supply pins. Table 1 compares the input ripple current requirements of the multiphase design and a conventional single-phase design. The multiphase technique reduces input capacitance by almost 60%. The complete schematic diagram and efficiency measurements are shown in Figures 1 and 2, respectively. With a 12V input and 250kHz switching frequency, greater than 80% efficiency is maintained for both outputs over most of the load range. Table 1. Comparison of input ripple current and input capacitors for single-phase and PolyPhase configurations Worst-Case Input Number of Input Caps Ripple Current (ARMS) (OSCON 16SP270M) at 65° C) Phases Single-Phase PolyPhase (LTC1629 + LTC1628-SYNC) 23.4 7 10.2 3 Introduction In broadband networking and high speed computing applications, multiple output, high current, low voltage power supplies are needed to power FPGAs, flash memories, DSPs and microprocessors. One such example calls for a maximum current of 60A to power the CPU and up to 15A to power the memory. A custom DC/DC module is usually expensive and the external circuitry for synchronization further increases the cost of individual supplies. This design idea presents a low cost, high efficiency, dual-output power supply design using LTC’s latest PolyPhase products, the LTC1628100 VIN = 12V VOUT = 1.5V fS = 200kHz EFFICIENCY (%) 90 80 70 60 50 0 10 20 30 40 LOAD CURRENT (A) 50 60 Figure 2. Efficiency vs load current for Figure 1’s circuit SYNC and LTC1629. The input is 12V and the outputs are 1.5V at 60A max for the CPU and 2.5V at 15A max for the memory. The design uses the LTC1629 and one channel of the LTC1628-SYNC to configure the 3-phase supply for the CPU power and the remaining channel of LTC1628-SYNC for the memory supply. With only twelve SO-8 MOSFETs and two SSOP-28 controllers, the complete power supply occupies a footprint of only 3" × 3". Efficiencies of 85% and 89% are achieved for outputs of 1.6V/ 60A and 2.5V/15A, respectively. Other Applications For applications with more than two outputs, several LTC1628-SYNCs can be teamed with the LTC1629 for greater than 2-phase operation. Figure 3 shows an example using the LTC1629 and LTC1628-SYNC in a 3-output, 4-phase application. Because four synchronous buck circuits are interleaved 90 degree out of phase, the net input ripple current is greatly reduced. Design Details The newly released LTC1628-SYNC is a dual-output, PolyPhase, current mode controller. Unlike other versions of the LTC1628, it has a PLLIN pin that enables external synchronization. In conjunction with the LTC1629, it can be used to implement a true 3-phase circuit for CPU power while the second output of the LTC1628-SYNC is used to generate the memory power supply. Because the channel used for the memory power switches at 300 degrees with respect to the other three channels used for CPU power, the net ripple Conclusion The synchronization capability of the LTC1628-SYNC helps minimize input capacitance and avoid beat frequencies. Teamed with LTC1629, it can effectively provide a 3-phase solution for multiple output applications and minimize the size and cost of the complete power supply. IIN VIN = 12V U1 OPEN I1 PHASMD TG1 CLKOUT TG2 BUCK: 2.5V/15A IIN BUCK: 2.5V/15A I2 VOUT1 2.5V/30A I1 LTC1629 U2 I3 BUCK 1.5V/15A TG1 PLLIN TG2 BUCK 1.8V/15A VOUT2 1.5V/15A VOUT3 1.8V/15A I2 I3 I4 LTC1628-SYNC I4 Figure 3. Block diagram of LTC1628-Sync/LTC1629 3-input, 4-phase application Linear Technology Magazine • February 2001 35 DESIGN IDEAS Reduce EMI with Ultralow Noise 48V to 5V, 10W DC/DC Converter by Rick Brewster Introduction Increasingly, designers are using ultralow noise controllers to avoid EMI problems. Lower operating voltages and more sensitive measurements have created the need for quieter supplies. Extra filtering components and shielding are usually required, as is a careful board layout. Ultralow noise switching regulator controllers such as the LT1683 reduce or eliminate the need for extra filtering. The LT1683 controller uses external MOSFETs to create ultralow noise DC/DC converters. Control of the switch voltage and switch current slew rates reduces switcher noise. The LT1683’s use of external switches allows for greater flexibility in the selection of voltage and current ratings of the supply. + 39µF 63V 2µs/DIV Figure 2. Voltage on one of the MOSFET drains and on the sense resistor Circuit Details T1 FZT649 10µF 20V 2N3904 1N4148 CS 50mV/DIV Figure 1 shows the schematic of an ultralow noise 48V to 5V converter using a push-pull forward converter topology. The output broadband noise is a very low 200µV (bandwidth = 48V 510Ω 0.5W 51k DRAIN 20V/DIV + OPTIONAL D3 D1 8.3 + 11V 68µF 20V 2.7M 14 5 130k 6 1.3nF 7 16.9k 8 25k 3.6k 16 25k 3.6k 15 1.5k 12 SHDN GATE A SYNC CT CAP B LT1683 RT GATE B RVSL CS D2 1 10pF 200V 5pF 19 30pF M1 M2 4 PGND VC SS GND 11 FB NFB 20 7.5k 9 30pF 10 10nF MIDCOM 31244 SILICONIX Si9422 ON SEMICONDUCTOR MBRS340 ON SEMICONDUCTOR MBR0530 (605) 886-4385 (800) 554-5565 (602) 244-6600 Figure 1. Ultralow noise 48V to 5V DC/DC converter 36 A 5V/2A 22µF 2×100µF POSCAP 10pF 200V 5pF 2 18 + 0.1Ω RCSL 13 T1: M1, M2: D1, D2: D3: CAP A V5 0.22µF 22nF 150µF OS-CON 17 3 VIN GCL B 22µF 2.49k + 100MHz) at 2A output (10W). The LT1683 contains all the control circuitry for the converter: oscillator, error amp, gate drivers and protection circuitry. The low noise is achieved by controlling the voltage slew rate of the MOSFET drain and the current slew rate of the MOSFET current. The capacitor divider network from the drain to Cap A or Cap B yields an effective 0.33pF capacitor that provides the voltage slew rate feedback information. The current slew feedback occurs internal to the LT1683 by means of the 100mΩ sense resistor. The resistors on the RVSL and RCSL pins allow the user to optimize the slew rates. The trade-off is between noise and converter efficiency. During design, monitor the output supply noise while slowing down the slew rates via the slew control resistors. Adjust the slew rate until the noise requirement is satisfied. In general, the efficiency loss is only a few percent. Figure 2 shows the voltage on the drain of one of the MOSFETs and the voltage across the sense resistor. Linear Technology Magazine • February 2001 DESIGN IDEAS A 200µV/DIV voltage. The CS pin provides the feedback for pulse-by-pulse current control and slew control. A large signal on CS, indicative of a fault, also shuts the MOSFETs off. Converter efficiency is improved by use of a bootstrap winding that powers the part when the converter is up and running. Efficiency at the low noise setting is approximately 77%. 200µVP-P B 20mV/DIV Conclusion The LT1683 provides a unique way to produce an efficient, ultralow noise supply. Novel control circuitry quiets the switcher, allowing a new supply solution for sensitive electronic systems. The use of external MOSFET switches allows the voltage and current ratings of the supply to be tailored to the application. 5µs/DIV Figure 3. 5V output noise (bandwidth = 100MHz) with undervoltage lockout, ensuring that the input is up and running before the converter is allowed to start. In addition, the GCL pin prevents excessive gate voltage on the MOSFET and protects against the MOSFETs turning on without sufficient gate Because of the voltage slew control, clamps or snubbers on the MOSFET drains are not required and switch ringing is greatly reduced. Figure 3 shows the noise at the outputs. The output noise is a very low 200µVP-P. The SHDN pin provides the supply LTC3200, continued from page 29 1µF 1 2 3V TO 4.4V Li-Ion BATTERY 1µF C+ 3 C– VOUT VIN 1µF LTC3200 FB ON OFF 6 (APPLY PWM WAVEFORM FOR ADJUSTABLE BRIGHTNESS CONTROL) SGND SHDN VSHDN UP TO 6 LEDS 8 PGND 7 5 82Ω 82Ω 82Ω 82Ω 82Ω 82Ω 4 t Figure 3. White or blue LED driver with LED current control Conclusion In either constant voltage or current controlled applications of the LTC3200, the LED brightness can be controlled by applying a PWM signal (approximately 100Hz) to the SHDN pin. Varying the pulse width from 4% to 100% gives the LEDs a linear appearance of brightness control from full-on to full-off. Linear Technology Magazine • February 2001 In the tiny 6-pin SOT or 8-pin MSOP packages, the LTC3200 family of charge pumps provides a simple solution for powering white LEDs. Its small size, low external parts count and low noise, constant frequency operation is ideally suited for both communications and other portable products. http://www.linear-tech.com/ezone/zone.html Articles, Design Ideas, Tips from the Lab… 37 NEW DEVICE CAMEOS New Device Cameos LTC1701B: Tiny Step-Down Regulator Switches at 1MHz For low to medium power applications that must fit in small spaces, the LTC1701B provides a DC/DC converter that consumes less than 0.3in2 of PC board space. The part’s 1MHz switching frequency allows the use of tiny, low cost capacitors and inductors, which, along with the tiny SOT-23 package, results in a very compact solution. The LTC1701B operates continuously down to very low load currents to provide low output ripple at the expense of light load efficiency, while the LTC1701 incorporates automatic power saving Burst Mode operation to reduce gate charge losses, providing better efficiency at light loads. The LTC1701/LTC1701B operate from an input supply from 2.5V to 5.5V; the output voltage is adjustable from 1.25V to 5V. A built-in 0.28Ω P-channel MOSFET switch allows up to 500mA of output current at high efficiencies (up to 94%). In dropout, the internal switch can be turned on continuously, maximizing the usable battery life. Its combination of a high switching frequency, low ripple and an onboard P-channel MOSFET in a tiny SOT-23 package makes the LTC1701B ideal for low noise, space-critical portable applications. LTC Releases Popular High Speed Amplifiers and Comparators in the SpaceSaving SOT-23 Package Ideal for saving board space, the new SOT -23 versions of Linear Technology’s high speed amplifiers and comparators feature the same performance as their SO counterparts but require much less board area. The 400MHz LT1395S5/LT1395S6 current feedback amplifiers provide the bandwidth and output current required for cable driver and video applications. A gain flatness of 0.1dB to 100MHz ensures signal fidelity and 38 the 80mA output current is sufficient for cable drivers. The parts draw only 4.6mA and operate on all supplies from a single 4V to ±6V. A shutdown feature is included in the LT1395S6. The 100MHz LT1812S5/LT1812S6 voltage feedback amplifiers have a maximum offset voltage of 1.5mV and a maximum input offset current of 400nA. Drawing only 3mA supply current, the devices have a slew rate of 750V/µs and can drive a 100Ω load to ±3.5V with ±5V supplies. These easy-to-use devices are stable with load capacitances as high as 1000pF. The LT1812S6 includes a power-saving shutdown feature. Operating from supplies as low as 2.5V, the 325MHz LT1806S6 and the 180MHz LT1809S6 amplifiers provide the distortion and noise performance required by low voltage signal conditioning systems. Rail-to-rail inputs and outputs allow the entire supply range to be used and the high output current capability, 60mA typical on a 3V supply, is ideal for cable driver applications. The LT1806S6 is optimized for noise and DC performance, featuring a low voltage noise of 3.5nV/√Hz and a maximum offset voltage of 700µV. The LT1809S6 is optimized for slew rate and distortion, featuring a slew rate of 350V/µs and low harmonic distortion of –90dBc at fC = 5MHz (VS = 5V, VO = 2VP-P). Both parts are fully specified for 3V, 5V and ±5V operation. A shutdown function is included. The LT1719S6 4.5ns comparator makes fast comparisons painlessly on 3V and 5V supplies while consumFor further information on any of the devices mentioned in this issue of Linear Technology, use the reader service card or call the LTC literature service number: 1-800-4-LINEAR Ask for the pertinent data sheets and Application Notes. ing only 4.2mA. Internal hysteresis ensures clean transitions even on slow-moving input signals. The input common mode range extends to 100mV below ground and the outputs are rail-to-rail. The propagation time is 4.5ns with 20mV overdrive and 7ns with 5mV overdrive. A shutdown pin reduces the supply current to 80µA max. LT1494 and LT1672/ LT1673/LT1674 Ultralow Power Rail-to-Rail Op Amps The LT1494 and LT1672/73/74 are the newest members of the industry’s lowest power, precision, rail-to-rail op amp family. All devices have excellent amplifier specifications: 375µV max input offset voltage, 100pA max input offset current and 0.4µV/°C typical drift while operating from a maximum of only 1.5µA supply current for the LT1494 and 2µA/amplifier for the LT1672–74. A minimum openloop gain (AVOL) of 100V/mV ensures that gain errors are small. Both the supply rejection and the common mode rejection ratios are greater than 90dB. All devices exhibit little change in characteristics over the wide supply range of 2.2V to ±15V. They feature reverse battery protection (–18V min) and Over-The-Top™ operation (the ability to operate with the inputs above the positive supply). The LT1494 is a single rail-to-rail amplifier with a gain bandwidth product (GBW) of 3kHz and a slew rate (SR) of 0.4V/ms. The LT1672/73/74 are single, dual and quad decompensated versions of the LT1494. They are stable with a gain of five and are four times faster than the LT1494 (GBW = 12kHz, SR = 1.6V/ms), with an increase in supply current of only 500nA/amp. The LT1494 and LT1672 single come in 8-lead MSOP, SO and PDIP packages. The LT1673 dual comes in 8-lead SO and PDIP packages and the LT1674 quad comes in 14-lead SO and PDIP packages. Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. Linear Technology Magazine • February 2001 DESIGN TOOLS DESIGN TOOLS Technical Books 1990 Linear Databook, Vol I —This 1440 page collection of data sheets covers op amps, voltage regulators, references, comparators, filters, PWMs, data conversion and interface products (bipolar and CMOS), in both commercial and military grades. The catalog features well over 300 devices. $10.00 1992 Linear Databook, Vol II — This 1248 page supplement to the 1990 Linear Databook is a collection of all products introduced in 1991 and 1992. The catalog contains full data sheets for over 140 devices. The 1992 Linear Databook, Vol II is a companion to the 1990 Linear Databook, which should not be discarded. $10.00 1994 Linear Databook, Vol III —This 1826 page supplement to the 1990 and 1992 Linear Databooks is a collection of all products introduced since 1992. A total of 152 product data sheets are included with updated selection guides. The 1994 Linear Databook Vol III is a companion to the 1990 and 1992 Linear Databooks, which should not be discarded. $10.00 1995 Linear Databook, Vol IV —This 1152 page supplement to the 1990, 1992 and 1994 Linear Databooks is a collection of all products introduced since 1994. A total of 80 product data sheets are included with updated selection guides. The 1995 Linear Databook Vol IV is a companion to the 1990, 1992 and 1994 Linear Databooks, which should not be discarded. $10.00 1996 Linear Databook, Vol V —This 1152 page supplement to the 1990, 1992, 1994 and 1995 Linear Databooks is a collection of all products introduced since 1995. A total of 65 product data sheets are included with updated selection guides. The 1996 Linear Databook Vol V is a companion to the 1990, 1992, 1994 and 1995 Linear Databooks, which should not be discarded. $10.00 1997 Linear Databook, Vol VI —This 1360 page supplement to the 1990, 1992, 1994, 1995 and 1996 Linear Databooks is a collection of all products introduced since 1996. A total of 79 product data sheets are included with updated selection guides. The 1997 Linear Databook Vol VI is a companion to the 1990, 1992, 1994, 1995 and 1996 Linear Databooks, which should not be discarded. $10.00 1999 Linear Data Book, Vol VII — This 1968 page supplement to the 1990, 1992, 1994, 1995, 1996 and 1997 Linear Databooks is a collection of all product data sheets introduced since 1997. A total of 120 product data sheets are included, with updated selection guides. The 1999 Linear Databooks is a companion to the previous Linear Databooks, which should not be discarded. $10.00 1990 Linear Applications Handbook, Volume I — 928 pages full of application ideas covered in depth by 40 Application Notes and 33 Design Notes. This catalog covers a broad range of “real world” linear circuitry. In addition to detailed, systems-oriented circuits, this handbook contains broad tutorial content together with liberal use of schematics and scope photography. A special feature in this edition includes a 22-page section on SPICE macromodels. $20.00 1993 Linear Applications Handbook, Volume II — Continues the stream of “real world” linear circuitry initiated by the 1990 Handbook. Similar in scope to the 1990 edition, the new book covers Application Notes 40 Linear Technology Magazine • February 2001 through 54 and Design Notes 33 through 69. References and articles from non-LTC publications that we have found useful are also included. $20.00 1997 Linear Applications Handbook, Volume III — This 976 page handbook maintains the practical outlook and tutorial nature of previous efforts, while broadening topic selection. This new book includes Application Notes 55 through 69 and Design Notes 70 through 144. Subjects include switching regulators, measurement and control circuits, filters, video designs, interface, data converters, power products, battery chargers and CCFL inverters. An extensive subject index references circuits in LTC data sheets, design notes, application notes and Linear Technology magazines. $20.00 1998 Data Converter Handbook — This impressive 1360 page handbook includes all of the data sheets, application notes and design notes for Linear Technology’s family of high performance data converter products. Products include A/D converters (ADCs), D/A converters (DACs) and multiplexers—including the fastest monolithic 16-bit ADC, the 3Msps, 12-bit ADC with the best dynamic performance and the first dual 12-bit DAC in an SO-8 package. Also included are selection guides for references, op amps and filters and a glossary of data converter terms. $10.00 Interface Product Handbook — This 424 page handbook features LTC’s complete line of line driver and receiver products for RS232, RS485, RS423, RS422, V.35 and AppleTalk® applications. Linear’s particular expertise in this area involves low power consumption, high numbers of drivers and receivers in one package, mixed RS232 and RS485 devices, 10kV ESD protection of RS232 devices and surface mount packages. Available at no charge Power Management Solutions Brochure — This 96 page collection of circuits contains real-life solutions for common power supply design problems. There are over 70 circuits, including descriptions, graphs and performance specifications. Topics covered include battery chargers, desktop PC power supplies, notebook PC power supplies, portable electronics power supplies, distributed power supplies, telecommunications and isolated power supplies, off-line power supplies and power management circuits. Selection guides are provided for each section and a variety of helpful design tools are also listed for quick reference. Available at no charge. Data Conversion Solutions Brochure␣ —␣ This 64 page collection of data conversion circuits, products and selection guides serves as excellent reference for the data acquisition system designer. Over 60 products are showcased, solving problems in low power, small size and high performance data conversion applications— with performance graphs and specifications. Topics covered include ADCs, DACs, voltage references and analog multiplexers. A complete glossary defines data conversion specifications; a list of selected application and design notes is also included. Available at no charge Telecommunications Solutions Brochure —This 76 page collection of application circuits and selection guides covers a wide variety of products targeted for telecommunications. Circuits solve real life problems for central office switching, cellular phones, high speed modems, base station, plus special sections covering –48V and Hot SwapTM applications. Many applications highlight new products such as Hot Swap controllers, power products, high speed amplifiers, A/D converters, interface transceivers and filters. Includes a telecommunications glossary, serial interface standards, protocol information and a complete list of key application notes and design notes. Available at no charge. Applications on Disk FilterCAD™ 3.0 CD-ROM — This CD-ROM contains the latest release of FilterCAD, version 3.0. FilterCAD is a powerful filter design tool that supports all of Linear Technology’s high performance active RC and switched capacitor filters. You can run FilterCAD directly from the CD or install it on your computer’s hard disk for much faster operation. FilterCAD requires a PC running Windows® 95 or later. The CD-ROM also contains a filter selection guide that lists all of Linear Technology’s filter products, along with links to their data sheets. Adobe® Acrobat Reader, version 3.0 or later, is required to use the selection guide. Available at no charge. Noise Disk — This IBM-PC (or compatible) program allows the user to calculate circuit noise using LTC op amps, determine the best LTC op amp for a low noise application, display the noise data for LTC op amps, calculate resistor noise and calculate noise using specs for any op amp. Available at no charge SPICE Macromodel Disk — This IBM-PC (or compatible) high density diskette contains the library of LTC op amp SPICE macromodels. The models can be used with any version of SPICE for general analog circuit simulations. The diskette also contains working circuit examples using the models and a demonstration copy of PSPICE™ by MicroSim. Available at no charge SwitcherCAD™ — The SwitcherCAD program is a powerful PC software tool that aids in the design and optimization of switching regulators. The program can cut days off the design cycle by selecting topologies, calculating operating points and specifying component values and manufacturer’s part numbers. 144 page manual included. $20.00 SwitcherCAD supports the following parts: LT1070 series: LT1070, LT1071, LT1072, LT1074 and LT1076. LT1082. LT1170 series: LT1170, LT1171, LT1172 and LT1176. It also supports: LT1268, LT1269 and LT1507. LT1270 series: LT1270 and LT1271. LT1371 series: LT1371, LT1372, LT1373, LT1375, LT1376 and LT1377. Micropower SwitcherCAD™ — The MicropowerSCAD program is a powerful tool for designing DC/DC converters based on Linear Technology’s micropower switching regulator ICs. Given basic design parameters, MicropowerSCAD selects a circuit topology and offers you a selection of appropriate Linear Technology switching regulator ICs. MicropowerSCAD also performs circuit simulations to select the other components which surround the DC/DC converter. In the case of a battery supply, MicropowerSCAD can perform a battery life simulation. 44 page manual included. $20.00 MicropowerSCAD supports the following LTC micropower DC/DC converters: LT1073, LT1107, LT1108, LT1109, LT1109A, LT1110, LT1111, LT1173, LTC1174, LT1300, LT1301 and LT1303. continued on page 40 39 DESIGN TOOLS, continued from page 39 CD-ROM Catalog LinearView — LinearView™ CD-ROM version 4.0 is Linear Technology’s latest interactive CD-ROM. It allows you to instantly access thousands of pages of product and applications information, covering Linear Technology’s complete line of high performance analog products, with easy-to-use search tools. The LinearView CD-ROM includes the complete product specifications from Linear Technology’s Databook library (Volumes I–VII) and the complete Applications Handbook collection (Volumes I–III). Our extensive collection of Design Notes and the complete collection of Linear Technology magazine are also included. A powerful search engine built into the LinearView CDROM enables you to select parts by various criteria, such as device parameters, keywords or part numbers. All product categories are represented: data conversion, references, amplifiers, power products, filters and interface circuits. 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Phone: +886 (2) 2521-7575 FAX: +886 (2) 2562-2285 Haselburger Damm 4 D-59387 Ascheberg Germany Phone: +49 (2593) 9516-0 FAX: +49 (2593) 951679 Zettachring 12 D-70567 Stuttgart Germany Phone: +49 (711) 1329890 FAX: +49 (711) 7285055 JAPAN Linear Technology KK 5F NAO Bldg. 1-14 Shin-Ogawa-cho Shinjuku-ku Tokyo, 162 Japan Phone: +81 (3) 3267-7891 FAX: +81 (3) 3267-8510 UNITED KINGDOM Linear Technology (UK) Ltd. The Coliseum, Riverside Way Camberley, Surrey GU15 3YL United Kingdom Phone: +44 (1276) 677676 FAX: +44 (1276) 64851 6F Tokyo Seimei Honmachi Bldg. 1-6-13 Awaza, Nishi-ku Osaka-shi, 550-0011, Japan Phone: +81 (6) 6533-5880 FAX: +81 (6) 6533-5885 Linear Technology Magazine • February 2001