A8514 Datasheet

A8514
Wide Input Voltage Range, High Efficiency
Fault Tolerant LED Driver
Features and Benefits
Description
• AEC-Q100 Qualified
• Wide input voltage range of 5 to 40 V for start/stop, cold
crank and load dump requirements
• Fully integrated LED current sinks and boost converter
with 60 V DMOS
• Sync function to synchronize boost converter switching
frequency up to 2.3 MHz, allowing operation above the
AM band
• Excellent input voltage transient response
• Single resistor primary OVP minimizes VOUT leakage
• Internal secondary OVP for redundant protection
• LED current of 80 mA per channel
• Drives up to 12 series LEDs in 4 parallel strings
• 0.7% to 0.8% LED to LED matching accuracy
• PWM and analog dimming inputs
• 5000:1 PWM dimming at 200 Hz
• Provides driver for external PMOS input disconnect switch
• Extensive protection against:
▫ Shorted boost switch or inductor
▫ Shorted FSET or ISET resistor
▫ Shorted output
▫ Open or shorted LED pin
▫ Open boost Schottky
▫ Overtemperature (OTP)
The A8514 is a multi-output white LED driver for small-size
LCD backlighting. It integrates a current-mode boost converter
with internal power switch and four current sinks. The boost
converter can drive up to 48 LEDs, 12 LEDs per string, at
80 mA. The LED sinks can be paralleled together to achieve
even higher LED currents, up to 320 mA. The A8514 can
operate with a single power supply, from 5 to 40 V, which
allows the part to withstand load dump conditions encountered
in automotive systems.
The A8514 can drive an external P-FET to disconnect the input
supply from the system in the event of a fault. The A8514
provides protection against output short and overvoltage,
open or shorted diode, open or shorted LED pin, shorted
boost switch or inductor, shorted FSET or ISET resistor, and
IC overtemperature. A dual level cycle-by-cycle current limit
function provides soft start and protects the internal current
switch against high current overloads.
The A8514 has a synchronization pin that allows PWM
switching frequencies to be synchronized in the range of
580 kHz to 2.3 MHz. The high switching frequency allows
the A8514 to operate above the AM radio band.
Package: 20-pin TSSOP with exposed
thermal pad (suffix LP)
Continued on the next page…
Applications:
LCD backlighting or LED lighting for:
▫ Automotive infotainment
▫ Automotive cluster
▫ Automotive center stack
Not to scale
Typical Application Circuit
VIN
8 to 16 V
CIN
4.7 μF
50 V
RSC
0.033 Ω
RADJ
249 Ω
Optional
RC
20 Ω
CC
22 nF
Q1
VC
CVDD
0.1 μF
100 kΩ
D1
2 A / 60 V
L1
10 μH
GATE
VSENSE
VIN
VDD
OVP
COUT
4.7 μF
50 V
A8514
PAD
RFSET
10 kΩ
ROVP
137 kΩ
SW
FAULT
PWM/EN
APWM
ISET
RISET
8.25 kΩ
VOUT
FSET/SYNC
AGND
LED1
LED2
LED3
LED4
COMP
PGND
CP
120 pF
RZ
150 Ω
CZ
0.47 μF
Figure 1. Application with VIN to ground short protection, using optional P-MOSFET for input disconnect protection
A8514-DS, Rev. 5
Wide Input Voltage Range, High Efficiency
Fault Tolerant LED Driver
A8514
Description (continued)
The A8514 is provided in a 20-pin TSSOP package (suffix LP) with
an exposed pad for enhanced thermal dissipation. It is lead (Pb) free,
with 100% matte tin lead frame plating.
Selection Guide
Part Number
Packing*
A8514KLPTR-T
4000 pieces per 13-in. reel
*Contact Allegro™ for additional packing options
Absolute Maximum Ratings*
Characteristic
Symbol
Rating
Unit
LEDx Pins
–0.3 to 55
V
OVP Pin
–0.3 to 60
V
VSENSE and GATE pins should not exceed VIN
by more than 0.4 V
–0.3 to 40
V
Continuous
–0.6 to 62
V
VIN, VSENSE, GATE Pins
SW Pin
Notes
–1.0
V
¯Ā¯Ū¯L̄¯T̄
¯ Pin
F̄
t < 50 ns
-0.3 to 40
V
ISET, FSET, APWM, COMP Pins
–0.3 to 5.5
V
–0.3 to 7
V
All Other Pins
Operating Ambient Temperature
TA
–40 to 125
ºC
Maximum Junction Temperature
TJ(max)
150
ºC
Tstg
–55 to 150
ºC
Storage Temperature
Range K
*Stresses beyond those listed in this table may cause permanent damage to the device. The Absolute Maximum ratings are
stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in the Electrical
Characteristics table is not implied. Exposure to Absolute Maximum-rated conditions for extended periods may affect device reliability.
Table of Contents
Specifications
Thermal Characteristics
Pin-out Diagram and Terminal List
Characteristic Performance
Functional Description
Enabling the IC
Powering up: LED pin short-to-ground check
Soft start function
Frequency selection
Sync
LED current setting and LED dimming
PWM dimming
APWM pin
Analog dimming
LED short detect
Overvoltage protection
Boost switch overcurrent protection
2
3
3
8
11
11
11
13
13
14
15
15
16
18
18
19
21
Input overcurrent protection and disconnect switch
Setting the current sense resistor
Input UVLO
VDD
Shutdown
Fault protection during operation
Application Information
22
23
23
23
23
24
26
Design Example for Boost Configuration
Design Example for SEPIC Configuration
Package Outline Drawing
26
30
34
Appendix A. Feedback Loop Calculations
A-1
Power Stage Transfer Function
Output to Control Transfer Function
Stabilizing the Closed Loop System
Measuring the Feedback Loop Gain and Phase Margin
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A-1
A-2
A-4
A-6
2
Wide Input Voltage Range, High Efficiency
Fault Tolerant LED Driver
A8514
GATE 1
20 SW
VSENSE 2
Pin-out Diagram
19 OVP
VIN 3
18 PGND
FAULT 4
17 PGND
COMP 5
16 PGND
APWM 6
15 VDD
PWM/EN 7
14 LED1
FSET/SYNC 8
13 LED2
ISET 9
12 LED3
AGND 10
11 LED4
Terminal List Table
Number
Name
1
GATE
Function
2
VSENSE
3
VIN
4
¯Ā¯Ū¯L̄¯T̄
¯
F̄
Indicates a fault condition. Connect a 100 kΩ resistor between this pin and the required logic
level voltage. The pin is an open drain type configuration that will be pulled low when a fault
occurs.
5
COMP
Output of the error amplifier and compensation node. Connect a series RZ-CZ network from
this pin to ground for control loop compensation.
6
APWM
Analog trimming option for dimming. Applying a digital PWM signal to this pin adjusts the
internal ISET current.
7
PWM/EN
PWM dimming pin, used to control the LED intensity by using pulse width modulation. Also
used to enable the A8514.
8
FSET/SYNC
Frequency/synchronization pin. A resistor RFSET from this pin to ground sets the switching
frequency. This pin can also be used to synchronize two or more A8514s in the system. The
maximum synchronization frequency is 2.3 MHz.
Output gate driver pin for external P-channel FET control.
Connect this pin to the negative sense side of the current sense resistor RSC. The threshold
voltage is measured as VIN – VSENSE . There is also a fixed current sink to allow for trip
threshold adjustment.
Input power to the A8514 as well as the positive input used for current sense resistor.
9
ISET
10
AGND
Connect the RISET resistor between this pin and ground to set the 100% LED current.
11,12,13,14
LEDx
Connect the cathodes of the LED strings to these pins.
15
VDD
Output of internal LDO; connect a 0.1 μF decoupling capacitor between this pin and ground.
16,17.18
PGND
19
OVP
Overvoltage Condition (OVP) sense; connect the ROVP resistor from VOUT to this pin to
adjust the overvoltage protection.
20
SW
The drain of the internal DMOS switch of the boost converter.
–
PAD
Exposed pad of the package providing enhanced thermal dissipation. This pad must be
connected to the ground plane(s) of the PCB with at least 8 vias, directly in the pad.
LED signal ground.
Power ground for internal DMOS device.
Thermal Characteristics may require derating at maximum conditions, see application information
Characteristic
Symbol
Test Conditions*
Value
Unit
On 2-layer PCB, 3 in.
40.0
ºC/W
On 4-layer PCB based on JEDEC standard
(estimated)
29.0
ºC/W
2
Package Thermal Resistance
RθJA
*Additional thermal information available on the Allegro website
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115 Northeast Cutoff
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Wide Input Voltage Range, High Efficiency
Fault Tolerant LED Driver
A8514
Functional Block Diagram
VDD
SW
Internal VCC
Regulator
UVLO
VIN
VREF
1.235 V
Ref
Internal VCC
AGND
∑
FSET/SYNC
Fault
+
–
Oscillator
Diode
Open
Driver
Circuit
+ Sense
COMP
–
+
Current
Sense
ISS
–
Internal
Soft Start
+
PGND
–
VSENSE
Thermal
Shutdown
Input Current
Sense Amplifier
IADJ
Fault
+
PMOS
Driver
PWM/EN
OVP
Sense
GOFF
Fault
Enable
–
GATE
OVP
VREF
Open/Short
LED Detect
PWM
100 kΩ
VREF
ISS
LED1
LED
Driver
APWM
LED2
LED3
ISET
Internal VCC
ISET
LED4
AGND
Fault
FAULT
PAD
PGND
AGND
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Wide Input Voltage Range, High Efficiency
Fault Tolerant LED Driver
A8514
ELECTRICAL CHARACTERISTICS1,2 Valid at VIN = 16 V, TA = 25°C, indicates specifications guaranteed by design and
characterization over the full operating temperature range with TA = TJ = –40°C to 125°C; unless otherwise noted
Characteristics
Symbol
Test Conditions
Min.
Typ.
Max.
Unit
5
–
40
V
Input Voltage Specifications
Operating Input Voltage Range3
VIN
UVLO Start Threshold
VUVLOrise
VIN rising
−
–
4.35
V
UVLO Stop Threshold
VUVLOfall
VIN falling
−
–
3.90
V
UVLO Hysteresis2
VUVLOHYS
300
450
600
mV
Input Currents
IQ
PWM/EN = VIH ; SW = 2 MHz, no load
−
5.5
10
mA
IQSLEEP
VIN = 16 V, VPWMEN = VFSETSYNC = 0 V
−
2
10.0
μA
Input Quiescent Current
Input Sleep Supply Current
Input Logic Levels (PWM/EN and APWM)
Input Logic Level-Low
VIL
VIN throughout operating input voltage range
–
–
400
mV
Input Logic Level-High
VIH
VIN throughout operating input voltage range
1.5
–
–
V
PWM/EN Pin Open Drain
Pull-down Resistor
RPWMEN
PWM/EN = 5 V
60
100
140
kΩ
APWM Pull-down Resistor
RAPWM
PWM/EN = VIH
60
100
140
kΩ
fAPWM
VIH = 2 V, VIL = 0 V
20
−
1000
kHz
44
48
52
dB
750
990
1220
μA/V
−
–350
−
μA
APWM
APWM Frequency2
Error Amplifier
Open Loop Voltage Gain
Transconductance
AVOL
gm
ΔICOMP = ±10 μA
Source Current
IEA(SRC)
VCOMP = 1.5 V
Sink Current
IEA(SINK)
VCOMP = 1.5 V
−
350
−
μA
COMP Pin Pull-down Resistance
RCOMP
¯ĀŪ¯L̄¯T̄
¯ =0
F̄
−
2000
−
Ω
VOVP(th)
OVP connected to VOUT
7.7
8.1
8.5
V
188
199
210
μA
−
0.1
1
μA
53
55
58
V
ISW = 0.750 A, VIN = 16 V
75
300
600
mΩ
VSW = 16 V, PWM/EN = VIL
−
0.1
1
μA
3.0
3.5
4.2
A
−
7.00
−
A
Overvoltage Protection
Overvoltage Threshold
OVP Sense Current
IOVPH
OVP Leakage Current
IOVPLKG
Secondary Overvoltage Protection
VOVP(sec)
ROVP = 40.2 kΩ, VIN = 16 V, PWM/EN = VIL
Boost Switch
Switch On-Resistance
RSW
Switch Leakage Current
ISWLKG
Switch Current Limit
ISW(LIM)
Secondary Switch Current Limit2
ISW(LIM2)
Higher than ISW(LIM)(max) for all conditions,
device latches when detected
Soft Start Boost Current Limit
ISWSS(LIM)
Initial soft start current for boost switch
−
700
−
mA
Minimum Switch On-Time
tSWONTIME
60
85
111
ns
Minimum Switch Off-Time
tSWOFFTIME
30
47
68
ns
Continued on the next page…
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115 Northeast Cutoff
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Wide Input Voltage Range, High Efficiency
Fault Tolerant LED Driver
A8514
ELECTRICAL CHARACTERISTICS1,2 (continued) Valid at VIN = 16 V, TA = 25°C, indicates specifications guaranteed by design
and characterization over the full operating temperature range with TA = TJ = –40°C to 125°C; unless otherwise noted
Characteristics
Symbol
Test Conditions
Min.
Typ.
Max.
Unit
1.8
2
2.2
MHz
Oscillator Frequency
RFSET = 10 kΩ
Oscillator Frequency
fSW
RFSET = 20 kΩ
0.9
1
1.1
MHz
RFSET = 35.6 kΩ
520
580
640
kHz
FSET/SYNC Pin Voltage
VFSET
−
1.00
−
V
FSET Frequency Range
fFSET
580
−
2500
kHz
fSWSYNC
580
−
2300
kHz
Synchronization Input
Minimum Off-Time
tPWSYNCOFF
150
−
−
ns
Synchronization Input
Minimum On-Time
tPWSYNCON
150
−
−
ns
RFSET = 10 kΩ
Synchronization
Synchronized PWM Frequency
SYNC Input Logic Voltage
VSYNC(H)
FSET/SYNC pin, high level
−
−
0.4
V
VSYNC(L)
FSET/SYNC pin, low level
2.0
−
−
V
LED Current Sinks
LEDx Accuracy
ErrLED
ISET = 120 μA
−
−
3
%
LEDx Matching
ΔLEDx
ISET = 120 μA
−
−
3
%
LEDx Regulation Voltage
VLED
VLED1=VLED2=VLED3 =VLED4, ISET = 120 μA
600
700
800
mV
ISET to ILEDx Current Gain
AISET
ISET = 120 μA
633
653
672
A/A
ISET Pin Voltage
VISET
0.988
1.003
1.018
V
Allowable ISET Current
ISET
20
−
120
μA
4.6
5.1
5.6
V
VLED Short Detect
VLEDSC
While LED sinks are in regulation, sensed
from LEDx pin to ground
Soft Start LEDx Current
ILEDSS
Current through each enabled LEDx pin
during soft start
−
2.0
–
mA
Maximum PWM Dimming
Until Off-Time2
tPWML
Measured while PWM/EN = low, during
dimming control and internal references
are powered-on (exceeding tPWML results in
shutdown)
−
32,750
−
fSW
cycles
Minimum PWM On-Time
tPWMH
First cycle when powering-up device
−
0.75
2
μs
−
0.5
1
μs
−
360
500
ns
PWM High to LED-On Delay
tdPWM(on)
Time between PWM enable and LED current
reaching 90% of maximum
PWM Low to LED-Off Delay
tdPWM(off)
Time between PWM enable going low and
LED current reaching 10% of maximum
Continued on the next page…
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Wide Input Voltage Range, High Efficiency
Fault Tolerant LED Driver
A8514
ELECTRICAL CHARACTERISTICS1,2 (continued) Valid at VIN = 16 V, TA = 25°C, indicates specifications guaranteed by design
and characterization over the full operating temperature range with TA = TJ = –40°C to 125°C; unless otherwise noted
Characteristics
Symbol
Test Conditions
Min.
Typ.
Max.
Unit
−
−104
−
μA
GATE Pin
GATE Pin Sink Current
IGSINK
VGS = VIN
Gate Fault Shutdown Greater than
2X Current2
tGFAULT2
−
−
3
μs
Gate Fault Shutdown Greater than
1–2X Current
tGFAULT1
−
10,000
−
fSW
cycles
−
–6.7
−
V
18.8
20.3
21.8
μA
Gate Voltage
VGS
Gate to source voltage measured when gate
is on
VSENSE Pin
VSENSE Pin Sink Current
IADJ
VSENSE Trip Point
VSENSEtrip1
Measured between VIN and VSENSE,
RADJ = 0 Ω
94
104
114
mV
VSENSE 2X Trip2
VSENSEtrip2
2X VSENSEtrip , instantaneous shutdown,
RADJ = 0 Ω
−
180
−
mV
VFAULT
IFAULT = 1 mA
−
−
0.5
V
IFAULTLKG
VFAULT = 5 V
−
−
1
μA
−
165
−
ºC
−
20
−
ºC
¯Ā¯Ū¯L̄
¯T̄
¯ Pin
F̄
¯ĀŪ¯L̄¯T̄
¯ Pull-Down Voltage
F̄
¯ĀŪ¯L̄¯T̄
¯ Pin Leakage Current
F̄
Thermal Protection (TSD)
Thermal Shutdown Threshold2
TSD
Thermal Shutdown Hysteresis2
TSDHYS
Temperature rising
1For
input and output current specifications, negative current is defined as coming out of the node or pin (sourcing); positive current is defined as
going into the node or pin (sinking).
2Ensured by design and characterization, not production tested.
3Minimum V = 5 V is only required at startup. After startup is completed, the IC is able to function down to V = 4 V.
IN
IN
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Wide Input Voltage Range, High Efficiency
Fault Tolerant LED Driver
A8514
Characteristic Performance
TA = TJ
10
9
8
7
6
5
4
3
2
1
0
VIN UVLO Start Threshold Voltage
versus Ambient Temperature
VUVLOrise (V)
IQSLEEP (μA)
VIN Input Sleep Mode Current
versus Ambient Temperature
-50 -40 -30 -20 -10 0
10 20 30 40 50 60 70 80 90 100 110 120 130
4.40
4.35
4.30
4.25
4.20
4.15
4.10
4.05
4.00
-50 -40 -30 -20 -10 0
Temperature (°C)
Temperature (°C)
VIN UVLO Stop Threshold Voltage
versus Ambient Temperature
VUVLOfall (V)
fSW (MHz)
Switching Frequency
versus Ambient Temperature
2.20
2.15
2.10
2.05
2.00
1.95
1.90
1.85
1.80
-50 -40 -30 -20 -10 0
10 20 30 40 50 60 70 80 90 100 110 120 130
3.70
3.69
3.68
3.67
3.66
3.65
3.64
3.63
3.62
3.61
3.60
-50 -40 -30 -20 -10 0
Temperature (°C)
OVP Pin Overvoltage Threshold
versus Ambient Temperature
8.4
8.3
VOVP(th) (V)
IOVPH (μA)
10 20 30 40 50 60 70 80 90 100 110 120 130
Temperature (°C)
OVP Pin Sense Current
versus Ambient Temperature
210
208
206
204
202
200
198
196
194
192
190
10 20 30 40 50 60 70 80 90 100 110 120 130
8.2
8.1
8.0
7.9
7.8
7.7
-50 -40 -30 -20 -10 0
10 20 30 40 50 60 70 80 90 100 110 120 130
Temperature (°C)
7.6
-50 -40 -30 -20 -10 0
10 20 30 40 50 60 70 80 90 100 110 120 130
Temperature (°C)
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Wide Input Voltage Range, High Efficiency
Fault Tolerant LED Driver
A8514
VSENSE Pin Sink Current
versus Ambient Temperature
Input Disconnect Switch Gate to Source Voltage
-6.3
versus Ambient Temperature
20.8
20.7
20.6
-6.5
IADJ (μA)
VGS (V)
-6.4
-6.6
-6.7
20.5
20.4
20.3
20.2
-6.8
20.1
20.0
-6.9
-50 -40 -30 -20 -10 0
-50 -40 -30 -20 -10 0
10 20 30 40 50 60 70 80 90 100 110 120 130
Temperature (°C)
Temperature (°C)
LED Current versus Ambient Temperature
ISET to LED Current Gain versus Ambient Temperature
AISET (A/A)
82
81
80
79
78
77
665
660
655
650
645
76
75
640
10 20 30 40 50 60 70 80 90 100 110 120 130
-50 -40 -30 -20 -10 0
Temperature (°C)
3
10 20 30 40 50 60 70 80 90 100 110 120 130
Temperature (°C)
LED to LED Matching Accuracy
versus Ambient Temperature
2
ΔLEDx (%)
ILED (mA)
ISET = 120 μA
670
83
-50 -40 -30 -20 -10 0
10 20 30 40 50 60 70 80 90 100 110 120 130
1
0
-1
-2
-3
-50 -40 -30 -20 -10 0
10 20 30 40 50 60 70 80 90 100 110 120 130
Temperature (°C)
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Wide Input Voltage Range, High Efficiency
Fault Tolerant LED Driver
A8514
Eﬃciency (%)
100
Eﬃciency for Various 4-String Configura ons
ILED = 70 mA, LED Vf ≈ 3.2 V
95
90
6 series LEDs each string
7 series LEDs each string
8 series LEDs each string
85
80
75
5
7
9
11
13
15
17
Input Voltage, VIN (V)
Eﬃciency for Various 4-String Configura ons
ILED = 80 mA, LED Vf ≈ 3.2 V
Eﬃciency (%)
100
95
90
6 series LEDs each string
85
7 series LEDs each string
8 series LEDs each string
80
75
5
7
9
11
13
15
17
Input Voltage, VIN (V)
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Wide Input Voltage Range, High Efficiency
Fault Tolerant LED Driver
A8514
Functional Description
The A8514 incorporates a current-mode boost controller with
internal DMOS switch, and four LED current sinks. It can be
used to drive four LED strings of up to 12 white LEDs in series,
with current up to 80 mA per string. For optimal efficiency,
the output of the boost stage is adaptively adjusted to the minimum voltage required to power all of the LED strings. This is
expressed by the following equation:
VOUT = max ( VLED1 ,..., VLED4 ) + VREG
(1)
where
VLEDx is the voltage drop across LED strings 1 through 4, and
VREG is the regulation voltage of the LED current sinks (typically 0.7 V at the maximum LED current).
Enabling the IC
The IC turns on when a logic high signal is applied on the
PWM/EN pin with a minimum duration of tPWMH for the first
clock cycle, and the input voltage present on the VIN pin is
greater than the 4.35 V necessary to clear the UVLO (VUVLOrise )
threshold. The power-up sequence is shown in figure 2. Before
the LEDs are enabled, the A8514 driver goes through a system
check to determine if there are any possible fault conditions that
might prevent the system from functioning correctly. Also, if the
FSET/SYNC pin is pulled low, the IC will not power-up. More
information on the FSET/SYNC pin can be found in the Sync
section of this datasheet.
Powering up: LED pin short-to-ground check
The VIN pin has a UVLO function that prevents the A8514
from powering-up until the UVLO threshold is reached. After
the VIN pin goes above UVLO, and a high signal is present on
the PWM/EN pin, the IC proceeds to power-up. As shown in
figure 3, at this point the A8514 enables the disconnect switch
and checks if any LEDx pins are shorted to ground and/or are not
used.
The LED detect phase starts when the GATE voltage of the
disconnect switch is equal to VIN – 4.5 V. After the voltage
threshold on the LEDx pins exceeds 120 mV, a delay of between
3000 and 4000 clock cycles is used to determine the status of the
pins. Thus, the LED detection duration varies with the switching
frequency, as shown in the following table:
Switching Frequency
(MHz)
Detection Time
(ms)
2
1.5 to 2
1
3 to 4
0.800
3.75 to 5
0.600
5 to 6.7
The LED pin detection voltage thresholds are as follows:
LED Pin Voltage
LED Pin Status
Action
<70 mV
Short-to-ground
Power-up is halted
150 mV
Not used
LED removed from operation
325 mV
LED pin in use
None
GATE = VIN – 4.5 V
VDD
GATE
C1
FSET/SYNC
C1
LEDx
LED detection period
C2
C2
ISET
C3
ISET
PWM/EN
C4
C3
C4
t
Figure 2. Power-up diagram; shows VDD (ch1, 2 V/div.), FSET/SYNC (ch2,
1 V/div.), ISET (ch3, 1 V/div.), and PWM/EN (ch4, 2 V/div.) pins,
time = 200 μs/div.
PWM/EN
t
Figure 3. Power-up diagram; shows the relationship of an LEDx pin with
respect to the gate voltage of the disconnect switch (if used) during the
LED detect phase, as well as the duration of the LED detect phase for a
switching frequency of 2 MHz; shows GATE (ch1, 5 V/div.), LED (ch2,
500 mV/div.), ISET (ch3, 1 V/div.), and PWM/EN (ch4, 5 V/div.) pins,
time = 500 μs/div.
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Wide Input Voltage Range, High Efficiency
Fault Tolerant LED Driver
A8514
LED1
LED1
LED detection period
C1
LED detection period
C1
C2
LED2
LED2
C2
C3
ISET
ISET
C3
C4
PWM/EN
PWM/EN
C4
t
t
4A. An LED detect occurring when both LED pins are selected to be used;
shows LED1 (ch1, 500 mV/div.), LED2 (ch2, 500 mV/div.), ISET (ch3,
1 V/div.), and PWM/EN (ch4, 5 V/div.) pins, time = 500 μs/div.
4B. Example with LED2 pin not being used; the detect voltage is
about 150 mV; shows LED1 (ch1, 500 mV/div.), LED2 (ch2, 500 mV/div.),
ISET (ch3, 1 V/div.), and PWM/EN (ch4, 5 V/div.) pins, time = 500 μs/div.
Short removed
Pin shorted
LED1
C1
LED2
C2
ISET
C3
C4
PWM/EN
t
4C. Example with one LED shorted to ground. The IC will not proceed with
power-up until the shorted LED pin is released, at which point the LED is
checked to see if it is being used; shows LED1 (ch1, 500 mV/div.), LED2 (ch2,
500 mV/div.), ISET (ch3, 1 V/div.), and PWM/EN (ch4, 5 V/div.) pins,
time = 1 ms/div.
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Wide Input Voltage Range, High Efficiency
Fault Tolerant LED Driver
A8514
All unused pins should be connected with a 2.37 kΩ resistor to
ground, as shown in figure 5. The unused pin, with the pull-down
resistor, will be taken out of regulation at this point and will not
contribute to the boost regulation loop.
If a LEDx pin is shorted to ground the A8514 will not proceed
with soft start until the short is removed from the LEDx pin. This
prevents the A8514 from powering-up and putting an uncontrolled amount of current through the LEDs.
Soft start function
During soft start the LEDx pins are set to sink (ILEDSS) and the
boost switch current is reduced to the ISWSS(LIM) level to limit
the inrush current generated by charging the output capacitors.
When the converter senses that there is enough voltage on the
LEDx pins the converter proceeds to increase the LED current to
the preset regulation current and the boost switch current limit is
switched to the ISW(LIM) level to allow the A8514 to deliver the
necessary output power to the LEDs. This is shown in figure 6.
In case during operation a fault occurs that will increase the
switching frequency, the FSET/SYNC pin is clamped to a
maximum switching frequency of no more than 3.5 MHz. If the
FSET/SYNC pin is shorted to GND the part will shut down. For
more details see the Fault Mode table later in this datasheet.
Inrush current caused by
enabling the disconnect
switch (when used)
Operation during
ISWSS(lim)
C1
IOUT
C2
IIN
Normal operation
ISW(lim)
C3
VOUT
C4
PWM/EN
Frequency selection
The switching frequency on the boost regulator is set by the resistor connected to the FSET/SYNC pin. The switching frequency
can be can be anywhere from 580 kHz to 2.3 MHz. Figure 7
shows the typical switching frequencies for various resistor
values, with the relationship between RFSET and typical switching
frequency given as:
GND
A8514
LED1
LED2
LED3
LED4
GND
2.1
(2)
where RFSET is in in kilohms, fSW is in megahertz, k = 20.9 and
RINT (internal resistance of FSET pin) = 0.6 kΩ.
A8514
Figure 6. Startup diagram showing the input current, output voltage, and
output current; shows IOUT (ch1, 200 mA/div.), IIN (ch2, 1 A/div.), VOUT
(ch3, 20 V/div.), and PWM/EN (ch4, 5 V/div.), time = 1 ms/div.
LED1
LED2
LED3
LED4
2.37 kΩ
Figure 5. Channel select setup: (left) using only LED1, LED2, and LED3,
and (right) using all four channels.
Switching Frequency, fSW (MHz)
fSW = k / (RFSET + RINT ), or
RFSET = k / fSW – RINT
t
1.9
1.7
1.5
1.3
1.1
0.9
0.7
0.5
10 12 14 16 18 20 22 24 26 28 30 32 34 36
Resistance for RFSET (kΩ)
Figure 7. Typical Switching Frequency versus value of RFSET resistor
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Wide Input Voltage Range, High Efficiency
Fault Tolerant LED Driver
A8514
Sync
The A8514 can also be synchronized using an external clock
on the FSET/SYNC pin. Figure 8 shows the correspondence of
a sync signal and the FSET/SYNC pin, and figure 9 shows the
result when a sync signal is detected: the LED current does not
show any variation while the frequency changeover occurs. At
power-up if the FSET/SYNC pin is held low, the IC will not
power-up. Only when the FSET/SYNC pin is tri-stated to allow
the pin to rise, to about 1 V, or when a synchronization clock is
detected, will the A8514 try to power-up.
The basic requirement of the sync signal is 150 ns minimum ontime and 150 ns minimum off time, as indicated by the specifications for tPWSYNCON and tPWSYNCOFF . Figure 10 shows the timing
for a synchronization clock into the A8514 at 2.2 MHz. Thus any
pulse with a duty cycle of 33% to 66% at 2.2 MHz can be used to
synchronize the IC.
VOUT
C1
IOUT
C2
C3
FSET/SYNC
SW node
C4
t
Figure 8. Diagram showing a synchronized FSET/SYNC pin and switch
node; shows VOUT (ch1, 20 V/div.), IOUT (ch2, 200 mA/div.), FSET/SYNC
(ch3, 2 V/div.), and SW node (ch4, 20 V/div.), time = 2 μs/div.
The SYNC pulse duty cycle ranges for selected switching frequencies are:
VOUT
SYNC Pulse Frequency
(MHz)
Duty Cycle Range
(%)
C1
2.2
33 to 66
C2
2
30 to 70
1
15 to 85
0.800
12 to 88
0.600
9 to 91
If during operation a sync clock is lost, the IC will revert to the
preset switching frequency that is set by the resistor RFSET. During this period the IC will stop switching for a maximum period
of about 7 μs to allow the sync detection circuitry to switch over
to the externally preset switching frequency.
If the clock is held low for more than 7 μs, the A8514 will shut
down. In this shutdown mode the IC will stop switching, the
input disconnect switch is open, and the LEDs will stop sinking
current. To shutdown the IC into low power mode, the user must
disable the IC using the PWM pin, by keeping the pin low for a
period of 32,750 clock cycles. If the FSET/SYNC pin is released
at any time after 7 μs, the A8514 will proceed to soft start.
IOUT
C3
FSET/SYNC
2 MHz operation
1 MHz operation
SW node
C4
t
Figure 9. Transition of the SW waveform when the SYNC pulse is
detected. The A8514 switching at 2 MHz, applied SYNC pulse at 1 MHz;
shows VOUT (ch1, 20 V/div.), IOUT (ch2, 200 mA/div.), FSET/SYNC (ch3,
2 V/div.), and SW node (ch4, 20 V/div.), time = 5 μs/div.
t PWSYNCON
154 ns
150 ns
150 ns
t PWSYNCOFF
T = 454 ns
Figure 10. SYNC pulse on and off time requirements.
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Wide Input Voltage Range, High Efficiency
Fault Tolerant LED Driver
A8514
VOUT
LED current setting and LED dimming
The maximum LED current can be up to 80 mA per channel,
and is set through the ISET pin. To set the ILED current, connect
a resistor, RISET, between this pin and ground, according to the
following formula:
RISET = (1.003 × 653) / ILED
(3)
where ILED is in A and RISET is in Ω. This sets the maximum current through the LEDs, referred to as the 100% current. Standard
RISET values, at gain equals 653, are as follows:
Standard Closest RISET
Resistor Value
(kΩ)
COMP
C2
C1
C3
PWM
ILED
C4
t
LED current per LED, ILED
(mA)
8.25
80
10.2
65
16.5
40
22.1
30
PWM dimming
The LED current can be reduced from the 100% current level
by PWM dimming using the PWM/EN pin. When the PWM/EN
pin is pulled high, the A8514 turns on and all enabled LEDs sink
100% current. When PWM/EN is pulled low, the boost converter
and LED sinks are turned off. The compensation (COMP) pin is
floated, and critical internal circuits are kept active. The typical
PWM dimming frequencies fall between 200 Hz and 1 kHz. Figures 11A to 11D provide examples of PWM switching behavior.
Figure 11B. Typical PWM diagram showing VOUT, ILED, and COMP pin as
well as the PWM signal. PWM dimming frequency is 500 Hz at 1% duty
cycle ; shows VOUT (ch1, 10 V/div.), COMP (ch2, 2 V/div.), PWM (ch3,
5 V/div.), and ILED (ch4, 50 mA/div.), time = 500 μs/div.
PWM
C1
ILED
C2
t
Figure 11C. Delay from rising edge of PWM signal to LED current; shows
PWM (ch1, 2 V/div.), and ILED (ch2, 50 mA/div.), time = 200 ns/div.
PWM
VOUT
COMP
C1
C2
C1
C3
PWM
ILED
C2
C4
ILED
t
Figure 11A. Typical PWM diagram showing VOUT, ILED, and COMP pin as
well as the PWM signal. PWM dimming frequency is 500 Hz at 50% duty
cycle; shows VOUT (ch1, 10 V/div.), COMP (ch2, 2 V/div.), PWM (ch3,
5 V/div.), and ILED (ch4, 50 mA/div.), time = 500 μs/div.
t
Figure 11D. Delay from falling edge of PWM signal to LED current turn off;
shows PWM (ch1, 2 V/div.), and ILED (ch2, 50 mA/div.), time = 200 ns/div.
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Wide Input Voltage Range, High Efficiency
Fault Tolerant LED Driver
A8514
Another important feature of the A8514 is the PWM signal to
LED current delay. This delay is typically less than 500 ns, which
allows greater accuracy at low PWM dimming duty cycles, as
shown in figure 12.
APWM pin
The APWM pin is used in conjunction with the ISET pin (see figure 13). This is a digital signal pin that internally adjusts the ISET
current. When this pin is not used it should be tied to ground.
The typical input signal frequency is between 20 kHz and 1 MHz.
The duty cycle of this signal is inversely proportional to the percentage of current that is delivered to the LEDs (figure 14).
To use this pin for a trim function, the user should set the maximum output current to a value higher than the required current by
at least 5%. The LED ISET current is then trimmed down to the
appropriate value. Another consideration that also is important
is the limitation of the user APWM signal duty cycle. In some
cases it might be preferable to set the maximum ISET current to be
25% to 50% higher, thus allowing the APWM signal to have duty
cycles that are between 25% and 50%.
APWM
10
ErrLED (%)
8
ISET
Worst-case
6
Typical
RISET
4
A8514
ISET
Current
Mirror
Current
Adjust
PWM
2
LED
Driver
0
0.1
1
10
100
PWM Duty Cycle, D (%)
Figure 13. Simplified block diagram of the APWM and ISET circuit.
12
80
70
60
50
40
30
20
10
0
10
%ErrLED
IOUT (mA)
Figure 12. Percentage Error of the LED current versus PWM duty cycle
(at 200 Hz PWM frequency).
IOUT = 80 mA
IOUT = 65 mA
8
6
IOUT = 65 mA
4
IOUT = 80 mA
2
0
0
20
40
60
80
100
APWM Duty Cycle (%)
Figure 14. Output current versus duty cycle; 200 kHz APWM signal.
0
20
40
60
80
100
APWM Duty Cycle (%)
Figure 15. Percentage Error of the LED current versus PWM duty cycle;
200 kHz APWM signal.
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Wide Input Voltage Range, High Efficiency
Fault Tolerant LED Driver
A8514
As an example, a system that delivers a full LED current of
80 mA per LED would deliver 60 mA of current per LED when
an APWM signal is applied with a duty cycle of 25% (figures
16 and 17).
Although the order in which APWM and the PWM signal are
enabled does not matter, when enabling the A8514 into low current output while PWM and APWM dimming, the APWM signal
should be enable before or at the same time as the PWM signal.
This sequence will prevent the light output intensity from changing during power up of the IC.
Figure 18 shows the sequencing of the APWM and PWM signal
during power-up to prevent inadvertent light intensity changes.
The full intensity light output with no APWM or PWM dimming
is 80 mA per channel.
ILED
ILED
C1
C1
C2
C2
APWM
APWM
PWM/EN
PWM/EN
C3
C3
t
t
Figure 16. Diagram showing the transition of LED current from 60 mA
to 80 mA, when a 25% duty cycle signal is removed from the APWM pin.
PWM = 1; shows ILED (ch1, 50 mA/div.), APWM (ch2, 10 V/div.), and
PWM/EN (ch3, 5 V/div.), time = 500 μs/div.
Figure 17. Diagram showing the transition of LED current from 80 mA
to 60 mA, when a 25% duty cycle signal is applied to the APWM pin;
PWM = 1; shows ILED (ch1, 50 mA/div.), APWM (ch2, 10 V/div.), and
PWM/EN (ch3, 5 V/div.), time = 500 μs/div.
APWM
C1
ILED
C1
IOUT
C2
PWM/EN
C2
C3
APWM
VOUT
C3
PWM/EN
C4
t
Figure 18. Diagram showing power-up sequencing LED current of 5 mA
per channel with a 10% duty cycle PWM signal and a 95% duty cycle
APWM signal; shows APWM (ch1, 5 V/div.), ILED (ch2, 50 mA/div.),
PWM/EN (ch3, 5 V/div.), and VOUT (ch4, 10 V/div.), time = 500 μs/div.
t
Figure 19. Transition of output current level when a 50% duty cycle signal
is applied to the APWM pin, in conjunction with a 50% duty cycle PWM
dimming being applied to the PWM pin; shows IOUT (ch1, 50 mA/div.),
APWM (ch2, 10 V/div.), and PWM/EN (ch3, 5 V/div.), time = 500 μs/div.
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Wide Input Voltage Range, High Efficiency
Fault Tolerant LED Driver
A8514
Although the APWM dimming function has a wide frequency
range, if this function is used strictly as an analog dimming
function it is recommended to use frequency ranges between
50 and 500 kHz for best accuracy. The frequency range must be
considered only if the user is not using this function as a closed
loop trim function. Another limitation is that the propagation
delay between this APWM signal and IOUT takes several milliseconds to change the actual LED current. This effect is shown in
figures 16, 17, and 19.
Analog dimming
The A8514 can also be dimmed by using an external DAC or
another voltage source applied either directly to the ground side
of the RISET resistor or through an external resistor to the ISET
pin (see figure 19). The limit of this type of dimming depends on
the range of the ISET pin. In the case of the A8514 the limit is
20 to 125 μA.
• For a single resistor (panel A of figure 20), the ISET current is
controlled by the following formula:
VISET – VDAC
ISET =
(4)
R
ISET
where VISET is the ISET pin voltage and VDAC is the DAC output
voltage.
When the DAC voltage is 0 V the LED current will be at its
DAC
R ISET
VDAC
maximum. To keep the internal gain amplifier stable, the user
should not decrease the current through the RISET resistor to less
than 20 μA
• For a dual-resistor configuration (panel B of figure 20), the ISET
current is controlled by the following formula:
ISET =
VISET
VDAC – VISET
–
RISET
R1
(5)
The advantage of this circuit is that the DAC voltage can be
higher or lower, thus adjusting the LED current to a higher or
lower value of the preset LED current set by the RISET resistor:
▫ VDAC = 1.003 V; the output is strictly controlled by RISET
▫ VDAC > 1.003 V; the LED current is reduced
▫ VDAC < 1.003 V; the LED current is increased
LED short detect
Both LEDx pins are capable of handling the maximum VOUT
that the converter can deliver, thus providing protection from the
LEDx pin to VOUT in the event of a connector short.
An LEDx pin that has a voltage exceeding VLEDSC will be
removed from operation (see figure 21). This is to prevent the IC
from dissipating too much power by having a large voltage present on an LEDx pin.
A8514
IOUT
ISET
GND
GND
C1
(A)
LED1
DAC
R1
A8514
VDAC
GND
C2
ISET
R ISET
GND
(B)
Figure 20. Simplified diagrams of voltage control of ILED: typical
applications using a DAC to control ILED using a single resistor (upper),
and dual resistors (lower).
PWM/EN
C3
t
Figure 21. Example of the disabling of an LED string when the LED pin
voltage is increased above 4.6 V; shows IOUT (ch1, 200 mA/div.), LED1
(ch2, 5 V/div.), and PWM/EN (ch3, 5 V/div.), time = 10 μs/div.
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Wide Input Voltage Range, High Efficiency
Fault Tolerant LED Driver
A8514
While the IC is being PWM-dimmed, the IC rechecks the disabled LED every time the PWM signal goes high, to prevent false
tripping of an LED short event. This also allows some self-correction if an intermittent LED pin short to VOUT is present.
Overvoltage protection
The A8514 has overvoltage protection (OVP) and open Schottky
diode (D1 in figure 1) protection. The OVP protection has a
default level of 8.1 V and can be increased up to 53 V by connecting resistor ROVP between the OVP pin and VOUT . When
the current into the OVP pin exceeds 199 μA (typical), the OVP
comparator goes low and the boost stops switching.
The following equation can be used to determine the resistance
for setting the OVP level:
where:
ROVP = ( VOUTovp – VOVP(th) ) / IOVPH
VOVP(th) is the pin OVP trip point found in the Electrical Characteristics table, and
IOVPH is the current into the OVP pin.
There are several possibilities for why an OVP condition would
be encountered during operation, the two most common being: a
disconnected output, and an open LED string. Examples of these
are provided in figures 22 and 23.
Figure 22 illustrates when the output of the A8514 is disconnected from load during normal operation. The output voltage
instantly increases up to OVP voltage level and then the boost
stops switching to prevent damage to the IC. If the output is
drained off, eventually the boost might start switching for a short
duration until the OVP threshold is hit again.
Figure 23 displays a typical OVP event caused by an open LED
string. After the OVP condition is detected, the boost stops
VOUTovp is the target overvoltage level,
Output disconnect
event detected
(6)
ROVP is the value of the external resistor, in Ω,
VOUT
LED string open
condition detected
VOUT
SW node
C2
C2
C1
C3
C1
C3
SW node
PWM
PWM
IOUT
IOUT
C4
C4
t
Figure 22. OVP protection in an output disconnect event; shows VOUT
(ch1, 10 V/div.), SW node (ch2, 50 V/div.), PWM (ch3, 5 V/div.), and
IOUT (ch4, 200 mA/div.), time = 1 ms/div.
t
Figure 23. OVP protection in an open LED string event; shows VOUT
(ch1, 10 V/div.), SW node (ch2, 50 V/div.), PWM (ch3, 5 V/div.), and
IOUT (ch4, 200 mA/div.), time = 500 μs/div.
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Wide Input Voltage Range, High Efficiency
Fault Tolerant LED Driver
A8514
switching, and the open LED string is removed from operation.
Afterwards VOUT is allowed to fall, and eventually the boost will
resume switching and the A8514 will resume normal operation.
A8514 also has built-in secondary overvoltage protection to
protect the internal switch in the event of an open diode condition. Open Schottky diode detection is implemented by detecting
overvoltage on the SW pin of the device. If voltage on the SW
pin exceeds the device safe operating voltage rating, the A8514
disables and remains latched. To clear this fault, the IC must be
shut down either by using the PWM/EN signal or by going below
Open diode
condition detected
the UVLO threshold on the VIN pin. Figure 24 illustrates this.
As soon as the switch node voltage (SW) exceeds 60 V, the IC
shuts down. Due to small delays in the detection circuit, as well
as there being no load present, the switch node voltage will rise
above the trip point voltage.
Figure 25 illustrates when the A8514 is being enabled during an
open diode condition. The IC goes through all of its initial LED
detection and then tries to enable the boost, at which point the
open diode is detected.
PWM
C1
SW node
C2
VOUT
IOUT
C3
C4
t
Figure 24. OVP protection in an open Schottky diode event, while the IC is
in normal operation; shows PWM (ch1, 5 V/div.), SW node (ch2, 50 V/div.),
VOUT (ch3, 20 V/div.), and IOUT (ch4, 200 mA/div.), time = 1 μs/div.
Open diode
condition detected
PWM
C1
SW node
C2
VOUT
C3
IOUT
C4
t
Figure 25. OVP protection when the IC is enabled during an open diode
condition; shows PWM (ch1, 5 V/div.), SW node (ch2, 50 V/div.), VOUT
(ch3, 10 V/div.), and IOUT (ch4, 200 mA/div.), time = 500 μs/div.
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Wide Input Voltage Range, High Efficiency
Fault Tolerant LED Driver
A8514
Boost switch overcurrent protection
The boost switch is protected with cycle-by-cycle current limiting
set at a minimum of 3.0 A. There is also a secondary current limit
that is sensed on the boost switch. When detected this current
limit immediately shuts down the A8514. The level of this cur-
C1
rent limit is set above the cycle-by-cycle current limit to protect
the switch from destructive currents when the boost inductor is
shorted. Various boost switch overcurrent conditions are shown in
figures 26 through 28.
C1
SW node
SW node
C2
IL
IL
VOUT
VOUT
C2
PWM/EN
C3
C4
PWM/EN
C3
C4
t
t
Figure 26. Normal operation of the switch node (SW); inductor current
(IL) and output voltage (VOUT) for 9 series LEDs in each of four strings
configuration; shows SW node (ch1, 20 V/div.), inductor current IL (ch2,
1 A/div.), VOUT (ch3, 10 V/div.), and PWM/EN (ch4, 5 V/div.),
time = 2 μs/div.
Figure 27. Cycle-by-cycle current limiting; inductor current (yellow trace, IL),
note reduction in output voltage as compared to normal operation with the
same configuration (figure 26); shows SW node (ch1, 20 V/div.), inductor
current IL (ch2, 1 A/div.), VOUT (ch3, 10 V/div.), and PWM/EN (ch4, 5 V/div.),
time = 2 μs/div.
PWM/EN
C1
FAULT
C2
SW node
C3
IL
C4
t
Figure 28. Secondary boost switch current limit; when this limit is hit, the
A8514 immediately shuts down; shows PWM (ch1, 5 V/div.), VOUT (ch2,
5 V/div.), SW node (ch3, 50 V/div.), and inductor current IL (ch4, 2 A/div.),
time = 100 ns/div.
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Wide Input Voltage Range, High Efficiency
Fault Tolerant LED Driver
A8514
Input overcurrent protection and disconnect switch
The primary function of the input disconnect switch is to protect
the system and the device from catastrophic input currents during
a fault condition. The external circuit implementing the disconnect is shown in figure 29. If the input disconnect switch is not
used, the VSENSE pin must be tied to VIN and the GATE pin
must be left open.
C1
FAULT
IIN
C2
(2) Disconnect switch
goes into a linear mode
When selecting the external PMOS, check for the following
parameters:
• Drain-source breakdown voltage V(BR)DSS > –40 V
• Gate threshold voltage (make sure it is fully conducting at
VGS = –4 V, and cut-off at –1 V)
• RDS(on): Make sure the on-resistance is rated at VGS = –4.5 V or
similar, not at –10 V; derate it for higher temperature
The input disconnect switch has two modes of operation:
• 1X mode When the input current is between one and two times
the preset current limit value, the disconnect switch enters a constant-current mode for a maximum duration of 10,000 cycles or
5 ms at 2 MHz. During this time, the Fault flag is set immediately
and the disconnect switch goes into a linear mode of operation,
in which the input current will be limited to a value approximate
to the 1X current trip point level (figure 30). If the fault corrects
itself before the expiration of the timer, the Fault flag will be
removed and normal operation will resume.
The user can also during this time decide whether to shut down
the A8514. To immediately shut down the device, pull the FSET/
SYNC pin low for more than 7 μs. After the FSET/SYNC pin has
been low for a period longer than 7 μs, the IC will stop switching,
the input disconnect switch will open, and the LEDx pins will
stop sinking current. The A8514 can be powered-down into low
power mode. To do so, disable the IC by keeping the PWM/EN
pin low for a period of 32,750 clock cycles. To keep the disconnect switch stable while the disconnect switch is in 1X mode, use
a 22 nF capacitor for CC and a 20 Ω resistor for RC.
VIN
RSC
(3) IIN limited to 3 A
(1) Initial fault
detected
GATE
(4) After 12.5 ms,
disconnect switch
shuts down
C3
C4
PWM/EN
t
Figure 30. Showing typical wave forms for a 3-A, 1X current limit under a
¯ĀŪ¯L̄¯T̄
¯ (ch1, 5 V/div.), IIN (ch2, 2 A/
fault condition; shows fSW = 800 kHz, F̄
div.), GATE (ch3, 5 V/div.), and PWM/EN (ch4, 5 V/div.), time = 5 ms/div.
FAULT
Fault flag set at
1X trip point
C1
C2
A8514 shuts down at
2X trip point
GATE
IIN
C3
C4
PWM/EN
t
Figure 31. 2X mode, secondary overcurrent fault condition. IIN is the input
current through the switch. The Fault flag is set at the 1X current limit, and
when the 2X current limit is reached the A8514 disables the gate of the
¯ĀŪ¯L̄¯T̄
¯ (ch1, 5 V/div.), GATE (ch2,
disconnect switch (GATE); shows F̄
10 V/div.), IIN (ch3, 2 A/div.), and PWM/EN (ch4, 5 V/div.), time = 5 μs/div.
VIN
Q1
To L1
RC
To L1
CC
RADJ
GATE
VSENSE
VIN
GATE
A8514
VSENSE
VIN
A8514
Figure 29. Typical circuit (left) with the input disconnect feature implemented, and (right) without the input disconnect feature.
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A8514
Wide Input Voltage Range, High Efficiency
Fault Tolerant LED Driver
• 2X current limit If the input current level goes above 2X of the
preset current limit threshold, the A8514 will shut down in less
than 3 μs regardless of user input (figure 31). This is a latched
condition. The Fault flag is also set to indicate a fault. This
feature is meant to prevent catastrophic failure in the system due
to inductor short to ground, switch pin short to ground, or output
short to ground.
Setting the current sense resistor
The typical threshold for the current sense circuit is 104 mV,
when RADJ is 0 Ω. This voltage can be trimmed by the RADJ
resistor. The typical 1X trip point should be set at about 3 A,
which coincides with the cycle-by-cycle current limit minimum
threshold.
For example, given 3 A of input current, and the calculated maximum value of the sense resistor, RSC = 0.033 Ω.
VIN
RADJ = (VSENSETRIP – VADJ ) / IADJ
(7)
VDD
C3
PWM/EN
C4
t
The RSC chosen is 0.03 Ω, a standard.
Also:
IOUT
C1
C2
Figure 32. Shutdown showing a falling input voltage (VIN); shows VIN
(ch1, 2 V/div.), IOUT (ch2, 200 mA/div.), VDD (ch3, 5 V/div.), and PWM/EN
(ch4, 2 V/div.), time = 5 ms/div.
The trip point voltage is calculated as:
VADJ = 3.0 A × 0.03 Ω = 0.090 V
RADJ = (0.104 – 0.09 V) / (20.3 μA) = 731 Ω
GATE
Input UVLO
When VIN and VSENSE rise above the VUVLOrise threshold, the
A8514 is enabled. A8514 is disabled when VIN falls below the
VUVLOfall threshold for more than 50 μs. This small delay is used
to avoid shutting down because of momentary glitches in the
input power supply. When VIN falls below 4.35 V, the IC will
shut down (see figure 32).
VDD
The VDD pin provides regulated bias supply for internal circuits.
Connect the capacitor CVDD with a value of 0.1 μF or greater to
this pin. The internal LDO can deliver no more than 2 mA of current with a typical VDD of about 3.5 V, enabling this pin to serve
¯T̄
¯ pin.
as the pull-up voltage for the F̄¯Ā¯Ū¯L̄
Shutdown
If the PWM/EN pin is pulled low for more than tPWML (32,750
clock cycles), the device enters shutdown mode and clears all
C1
IOUT
C2
VDD
C3
PWM/EN
C4
t
Figure 33. Shutdown using the enable function, showing the 16 ms
delay between the PWM/EN signal and when the VDD and GATE of
the disconnect switch turns off; shows GATE (ch1, 10 V/div.), IOUT (ch2,
200 mA/div.), VDD (ch3, 5 V/div.), and PWM/EN (ch4, 2 V/div.), time = 5
ms/div.
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23
Wide Input Voltage Range, High Efficiency
Fault Tolerant LED Driver
A8514
internal fault registers. As an example, at a 2 MHz clock frequency, it will take approximately 16.3 ms to shut down the IC
into the low power mode (figure 33). When the A8514 is shut
down, the IC will disable all current sources and wait until the
PWM/EN signal goes high to re-enable the IC. If faster shut
down is required, the FSET/SYNC pin can be used.
The possible fault conditions that the device can detect are: Open
LED pin, LED pin shorted to ground, shorted inductor, VOUT
short to ground, SW pin shorted to ground, ISET pin shorted to
ground, and input disconnect switch source shorted to ground.
Fault protection during operation
The A8514 constantly monitors the state of the system to determine if any fault conditions occur during normal operation. The
response to a triggered fault condition is summarized in the Fault
Mode table.
• Some of the protection features might not be active during
startup, to prevent false triggering of fault conditions.
Note the following:
• Some of these faults will not be protected if the input disconnect
switch is not being used. An example of this is VOUT short to
ground.
Fault Mode Table
Fault Name
Type
Active
Fault
Flag
Set
Primary switch
overcurrent protection
(cycle-by-cycle
current limit)
Auto-restart
Always
No
This fault condition is triggered by the cycle-bycycle current limit, ISW(LIM).
Secondary switch
current limit
Input disconnect
current limit
Secondary OVP
Latched
Latched
Latched
Always
Always
Always
Boost
Disconnect
switch
Sink
driver
Off for
a single
cycle
On
On
Yes
When the current through the boost switch exceeds
secondary current SW limit (ISW(LIM2)) the device
immediately shuts down the disconnect switch,
LED drivers, and boost. The Fault flag is set. To reenable the device, the PWM/EN pin must be pulled
low for 32,750 clock cycles.
Off
Off
Off
Yes
The device is immediately shut off if the voltage
across the input sense resistor is 2X the preset
current value. The Fault flag is set. If the input
current limit is between 1X and 2X, the Fault flag
is set but the IC will continue to operate normally
for tGFAULT1 or until it is shut down. To re-enable
the device the PWM/EN pin must be pulled low for
32,750 clock cycles.
Off
Off
Off
Yes
Secondary overvoltage protection is used for open
diode detection. When diode D1 opens, the SW pin
voltage will increase until VOVP(SEC) is reached. This
fault latches the IC. The input disconnect switch is
disabled as well as the LED drivers, and the Fault
flag is set. To re-enable the part the PWM pin must
be pulled low for 32,750 clock cycles.
Off
Off
Off
Description
Continued on the next page…
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Wide Input Voltage Range, High Efficiency
Fault Tolerant LED Driver
A8514
Fault Mode Table (continued)
Fault Name
LED Pin Short
Protection
LED Pin open
ISET Short Protection
FSET/SYNC Short
Protection
Overvoltage
Protection
Type
Auto-restart
Auto-restart
Auto-restart
Auto-restart
Auto-restart
Active
Startup
Normal
Operation
Always
Always
Always
Fault
Flag
Set
Description
Boost
Disconnect
Switch
Sink
driver
Off
On
Off
No
This fault prevents the device from starting-up if
either of the LEDx pins are shorted. The device
stops soft-start from starting while either of the
LEDx pins are determined to be shorted. After the
short is removed, soft-start is allowed to start.
No
When an LEDx pin is open the device will determine
which LED pin is open by increasing the output
voltage until OVP is reached. Any LED string not
in regulation will be turned off. The device will then
go back to normal operation by reducing the output
voltage to the appropriate voltage level.
On
On
Off for
open
pins.
On
for all
others.
No
This fault occurs when the ISET current goes above
150% of the maximum current. The boost will stop
switching, the disconnect switch will turn off, and
the IC will disable the LED sinks until the fault is
removed. When the fault is removed the IC will try
to to regulate to the preset LED current.
Off
On
Off
Yes
Fault occurs when the FSET/SYNC current goes
above 150% of maximum current, about 180 μA.
The boost will stop switching, the disconnect switch
will turn off, and the IC will disable the LED sinks
until the fault is removed. When the fault is removed
the IC will try to restart with soft-start.
Off
Off
Off
No
Fault occurs when OVP pin exceeds VOVP(th)
threshold. The A8514 will immediately stop
switching to try to reduce the output voltage. If the
output voltage decreases then the A8514 will restart
switching to regulate the output voltage.
Stop
during
OVP
event.
On
On
On
On
Off for
shorted
pins.
On
for all
others.
LED Short Protection
Auto-restart
Always
No
Fault occurs when the LED pin voltage exceeds
VLEDSC. When the LED short protection is detected
the LED string that is above the threshold will be
removed from operation.
Overtemperature
Protection
Auto-restart
Always
No
Fault occurs when the die temperature exceeds the
overtemperature threshold, 165°C.
Off
Off
Off
VIN UVLO
Auto-restart
Always
No
Fault occurs when VIN drops below VUVLO , 3.90 V
maximum. This fault resets all latched faults.
Off
Off
Off
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Wide Input Voltage Range, High Efficiency
Fault Tolerant LED Driver
A8514
Application Information
Design Example for Boost Configuration
This section provides a method for selecting component values
when designing an application using the A8514. The resulting
design is diagrammed in figure 34.
Then the OVP resistor is:
Assumptions: For the purposes of this example, the following are
given as the application requirements:
where both IOVPH and VOVP(th) are taken from the Electrical
Characteristics table.
• VBAT: 10 to 14 V
Chose a value of resistor that is higher value than the calculated
ROVP . In this case a value of 137 kΩ was selected. Below is the
actual value of the minimum OVP trip level with the selected
resistor:
• Quantity of LED channels, #CHANNELS : 4
• Quantity of series LEDs per channel, #SERIESLEDS : 10
• LED current per channel, ILED : 60 mA
ROVP = (VOUT(OVP) – VOVP(th) ) / IOVPH
(9)
= (34.7 (V) – 8.1 (V)) / 199 (μA) = 133.67 kΩ
VOUT(OVP) = 137 (kΩ) × 199 (μA) + 8.1 (V) = 35.36 V
• Vf at 60 mA: 3.2 V
Step 3b At this point a quick check must be done to see if the
• fSW : 2 MHz
conversion ratio is acceptable for the selected frequency.
• TA(max): 65°C
Dmaxofboost = 1 – tSWOFFTIME × fSW
• PWM dimming frequency: 200 Hz, 1% duty cycle
(10)
= 1 – 68 (ns) × 2.0 (MHz) = 86.4%
Procedure: The procedure consists of selecting the appropriate
configuration and then the individual component values, in an
ordered sequence.
where the minimum off-time (tSWOFFTIME) is found in the Electrical Characteristics table.
Step 1 Connect LEDs to pins LED1 and LED2.
The Theoretical Maximum VOUT is then calculated as:
Step 2 Determining the LED current setting resistor RISET:
RISET = (VISET × AISET) / ILED
(7)
VOUT(max) =
= (1.003 (V) × 653) / 60 mA = 10.92 kΩ
=
Choose a 11.00 kΩ resistor.
Step 3 Determining the OVP resistor. The OVP resistor is
connected between the OVP pin and the output voltage of the
converter.
on the LED requirements. The regulation voltage, VLED , of the
A8514 is 700 mV. A constant term, 2 V, is added to give margin
to the design due to noise and output voltage ripple.
= 10 × 3.2 V+ 0.7 V + 2 V
= 34.7 V
– Vd
(11)
10 (V)
– 0.4 (V) = 73.13 V
1 – 0.864
where Vd is the diode forward voltage.
Step 3a The first step is determining the maximum voltage based
VOUT(OVP) = #SERIESLEDS × Vf + VLED + 2 (V)
VIN(min)
1 – Dmaxofboost
(8)
The Theoretical Maximum VOUT value must be greater than the
value VOUT(OVP) . If this is not the case, the switching frequency
of the boost converter must be reduced to meet the maximum
duty cycle requirements.
Step 4 Selecting the inductor. The inductor must be chosen such
that it can handle the necessary input current. In most applications, due to stringent EMI requirements, the system must operate
in continuous conduction mode throughout the whole input voltage range.
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Wide Input Voltage Range, High Efficiency
Fault Tolerant LED Driver
A8514
Step 4a Determining the duty cycle, calculated as follows:
D(max) = 1 –
= 1–
Step 4d Double-check to make sure the ½ current ripple is less
than IIN(min):
VIN(min)
(12)
VOUT(OVP) + Vd
Step 4e This step is used to verify that there is sufficient slope
Step 4b Determining the maximum and minimum input current
to the system. The minimum input current will dictate the inductor value. The maximum current rating will dictate the current
rating of the inductor. First, the maximum input current, given:
= 4
ILED
(13)
0.060 (A) = 0.240 A
=
VOUT(OVP) IOUT
VIN(min)
compensation for the inductor chosen. The slope compensation
value is determined by the following formula:
3.6 fSW
Slope Compensation =
= 3.6 A /μs
(19)
2 10 6
Next insert the inductor value used in the design:
ΔILused =
then:
IIN(max) =
(14)
H
35.36 (V) 240 (mA)
= 0.94 A
10 (V)
0.90
=
Required Slope (min) =
(15)
35.36 (V) 240 (mA)
= 0.67 A
14 (V) 0.90
inductor operates in continuous conduction mode, the value of
the inductor must be set such that the ½ inductor ripple current is
not greater than the average minimum input current. A first past
assumes Iripple to be 40% of the maximum inductor current:
(16)
= 0.94 × 0.40 = 0.376 A
then:
=
VIN(min)
ΔIL fSW
(17)
D(max)
10 (V)
0.376 (A) 2 (MHz)
0.72 = 9.57 μH
ΔILused 1 10 –6
1
(1 – D(max))
(21)
=
0.36 (A) 1 10 –6
1
(1 – 0.72)
2.0 (MHz)
= 2.57 A/μs
Note: The slope compensation value is in A/μs, and 1×10 –6 is a
constant multiplier.
Step 4f Determining the inductor current rating. The inductor
Step 4c Determining the inductor value. To ensure that the
L=
10 (V) 0.72
= 0.36 A
10 (μH) 2.0 (MHz)
If the minimum required slope is greater than the calculated slope
compensation, the inductor value must be increased.
A good approximation of efficiency, η , can be taken from the
efficiency curves located in the datasheet. A value of 90% is a
good starting approximation.
ΔIL = IIN(max) × Iripple
(20)
fSW
Next, calculate minimum input current, as follows:
=
VIN(min) D(max)
Lused fSW
Calculate the minimum required slope:
where η is efficiency.
VOUT(OVP) IOUT
IIN(min) =
VIN(max) H
(18)
0.67 A > 0.19 A
A good inductor value to use would be 10 μH.
10 (V)
= 72.04%
35.36 (V) + 0.4 (V)
IOUT = #CHANNELS
IIN(min) > 1/2 ΔIL
current rating must be greater than the IIN(max) value plus the
ripple current ΔIL, calculated as follows:
L(min) = IIN(max) + 1/2 ΔILused
(22)
= 0.94 (A) + 0.36 (A) / 2 = 1.12 A
Step 5 Determining the resistor value for a particular switching
frequency. Use the RFSET values shown in figure 7. For example,
a 10 kΩ resistor will result in a 2 MHz switching frequency.
Step 6 Choosing the proper switching diode. The switching diode
must be chosen for three characteristics when it is used in LED
lighting circuitry. The most obvious two are: current rating of the
diode and reverse voltage rating.
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Wide Input Voltage Range, High Efficiency
Fault Tolerant LED Driver
A8514
The reverse voltage rating should be such that during operation
condition, the voltage rating of the device is larger than the maximum output voltage. In this case it is VOUT(OVP).
The peak current through the diode is calculated as:
Idp = IIN(max) + 1/2 ΔILused
(23)
must be chosen such that they can provide filtering for both the
boost converter and for the PWM dimming function. The biggest
factors that contribute to the size of the output capacitor are:
PWM dimming frequency and PWM duty cycle. Another major
contributor is leakage current, ILK . This current is the combination of the OVP leakage current as well as the reverse current of
the switching diode. In this design the PWM dimming frequency
is 200 Hz and the minimum duty cycle is 1%. Typically, the voltage variation on the output, VCOUT , during PWM dimming must
be less than 250 mV, so that no audible hum can be heard. The
capacitance can be calculated as follows:
= 200 μA
0.36 (A)
0.72 +
0.94 (A) × 12
1 – 0.72
=0.240 (A)
(25)
= 0.39 A
The output capacitor must have a current rating of at least
390 mA. The capacitor selected in this design was a 4.7 μF 50 V
capacitor with a 3 A current rating.
Step 8 Selecting input capacitor. The input capacitor must be
Step 7 Choosing the output capacitors. The output capacitors
1 – D(min)
fPWM(dimming) V
COUT
∆ILused
IIN(max) × 12
1 – D(max)
D(max) +
ICOUTrms = IOUT
= 0.94 (A) + 0.36 (A) / 2 = 1.12 A
The third major component in deciding the switching diode is the
reverse current, IR , characteristic of the diode. This characteristic
is especially important when PWM dimming is implemented.
During PWM off-time the boost converter is not switching. This
results in a slow bleeding off of the output voltage, due to leakage
currents. IR can be a large contributor, especially at high temperatures. On the diode that was selected in this design, the current
varies between 1 and 100 μA.
COUT = ILK
The rms current through the capacitor is given by:
selected such that it provides a good filtering of the input voltage
waveform. To reduce the switching frequency noise, a good rule
of thumb is to set the input voltage high frequency ripple ΔVIN to
be 1% of the minimum input voltage. The minimum input capacitor requirements are as follows:
CIN =
=
∆ILused
8
8
(26)
fSW
∆VIN
0.36 (A)
2 (MHz) 0.1 (V)
= 0.23 μF
(24)
1 – 0.01
= 3.96 μF
200 Hz 0.250 V
A capacitor larger than 3.96 μF should be selected due to degradation of capacitance at high voltages on the capacitor. A ceramic
4.7 μF 50 V capacitor is a good choice to fulfill this requirement.
Corresponding capacitors include:
The rms current through the capacitor is given by:
CINrms = ∆ILused / 12 = 0.104 A
(27)
A good ceramic input capacitor with ratings of 2.2 μF 50 V or
4.7 μF 50 V will suffice for this application. Corresponding
capacitors include:
Vendor
Value
Part number
Vendor
Value
Part number
Murata
4.7 μF 50 V
GRM32ER71H475KA88L
Murata
4.7 μF 50 V
GRM32ER71H475KA88L
Murata
2.2 μF 50 V
GRM31CR71H225KA88L
Murata
2.2 μF 50 V
GRM31CR71H225KA88L
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Wide Input Voltage Range, High Efficiency
Fault Tolerant LED Driver
A8514
The required electrolytic capacitor will be:
( ∆I ∆T ) 106
(30)
CIN =
= 59.4 μF
∆VIN
If this issue is important to the customer, it is recommended to
use a 50 V/ 68 μF, low ESR value electrolytic capacitor.
Selecting the electrolytic input capacitor according to dimming
transient
During a PWM dimming transient, unless an adequate input
capacitor is used, the input voltage drops significantly.
The input capacitor ripple current, ΔI , during the dimming transient is the same as the input maximum DC current, and can be
calculated by:
∆I = IINDC(max) =
VOUT IOUT
VIN(min) η
Step 9 Choosing the input disconnect switch components. Set
the input disconnect 1X current limit to 3 A by choosing a sense
resistor. The calculated maximum value of the sense resistor is:
(28)
RSC(max) = VSENSEtrip / 3.0 (A)
= 0.104 (V) / 3.0 (A) = 0.035 Ω
The RSC chosen is 0.033 Ω, a standard.
where VIN(min) is the minimum input voltage, 10 V.
Consider the case where: η = 0.88, VIN(min) = 10 V,
VOUT = 33.1 V, IOUT = 0.24 A, then:
The trip point voltage must be:
VADJ = 3.0 (A) × 0.033 (Ω) = 0.099 (V)
RADJ = (VSENSEtrip – VADJ ) / IADJ (typ)
(32)
RADJ = (0.104 (V) – 0.099 (V)) / 20.3 (μA) = 246.31 Ω
A value of 249 Ω was chosen for this design.
ΔI = IINDC(max)
= (33.1 × 0.24) / (10 × 0.88) = 0.9 A
Allowing VIN to drop by ΔVIN = 0.5 V, and considering the feedback loop bandwidth, or cross over frequency, to be fC = 30 kHz,
the input drop will last ΔT, calculated by :
ΔT = 1 / fC = 33 × 10–6 (second)
VIN
10 to 14 V
CIN
4.7 μF
RSC
0.033 Ω
RADJ
249 Ω
RC
20 Ω
D1
2 A / 60 V
L1
10 μH
GATE
VSENSE
VIN
VDD
CVDD
0.1 μF
OVP
COUT2
2.2 μF
COUT1
4.7 μF
A8514
PAD
RFSET
10 kΩ
VOUT
ROVP
137 kΩ
SW
FAULT
PWM/EN
APWM
ISET
RISET
11 kΩ
calculate RZ, CZ, and CP. Using L1 = 10 μH, COUT = (4.7 μF +
2.2 μF ), and fC = 30 kHz, the calculation results for RZ , CZ , and
CP are: RZ = 499 Ω, CZ = 100 nF, and CP = 320 pF.
CC
22 nF
VC
100 kΩ
Step 10 See appendix A for a detailed description of how to
(29)
Q1
(31)
FSET/SYNC
AGND
LED1
LED2
10 LEDs each string
LED3
LED4
COMP
PGND
CP
320 pF
RZ
499 Ω
CZ
100 nF
Figure 34. The schematic diagram showing calculated values from the design example above.
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Wide Input Voltage Range, High Efficiency
Fault Tolerant LED Driver
A8514
Design Example for SEPIC Configuration
This section provides a method for selecting component values
when designing an application using the A8514 in SEPIC (Single-Ended Primary-Inductor Converter) circuit. SEPIC topology
has the advantage that it can generate a positive output voltage
either higher or lower than the input voltage. The resulting design
is diagrammed in figure 35.
Assumptions: For the purposes of this example, the following are
given as the application requirements:
• VBAT: 6 to 14 V ( VIN(min): 5 V and VIN(max): 16 V )
• Quantity of LED channels, #CHANNELS : 4
A8514 is 700 mV. A constant term, 2 V, is added to give margin
to the design due to noise and output voltage ripple.
VOUT(OVP) = #SERIESLEDS × Vf + VLED + 2 (V)
(34)
= 4 × 3.3 (V) + 0.7 (V) + 2 (V) = 15.9 V
Then the OVP resistor is:
ROVP = (VOUT(OVP) – VOVP(th) ) / IOVPH
(35)
= (15.9 (V) – 8.1 (V)) / 0.199 (mA) = 39.196 kΩ
where both IOVPH and VOVP(th) are taken from the Electrical
Characteristics table.
In this case a value of 39.2 kΩ was selected. Below is the actual
value of the minimum OVP trip level with the selected resistor:
• Quantity of series LEDs per channel, #SERIESLEDS : 4
• LED current per channel, ILED : 60 mA
VOUT(OVP) = 39.2 (kΩ) × 0.199 (mA) + 8.1 (V) = 15.9 V
• LED Vf at 60 mA: ≈ 3.3 V
Step 3b At this point a quick check must be done to determine if
• fSW : 2 MHz
the conversion ratio is acceptable for the selected frequency.
• TA(max): 65°C
Dmax = 1 – tSWOFFTIME × fSW
• PWM dimming frequency: 200 Hz, 1% duty cycle
= 1 – 68 (ns) × 2 (MHz) = 86.4%
Procedure: The procedure consists of selecting the appropriate
configuration and then the individual component values, in an
ordered sequence.
where the minimum off-time (tSWOFFTIME) is found in the Electrical Characteristics table.
The Theoretical Maximum VOUT is then calculated as:
Step 1 Connecting LEDs to LEDx pins. If only some of the LED
channels are needed, the unused LEDx pins should be pulled to
ground using a 1.5 kΩ resistor.
Step 2 Determining the LED current setting resistor RISET:
RISET = (VISET × AISET) / ILED
(36)
VOUT(max) = VIN(min)
= 5 (V)
(33)
= (1.003 (V) × 653) / 0.60 (A) = 10.92 kΩ
Choose an 11.00 kΩ 1% resistor.
Dmax
1 – Dmax
– Vd
(37)
0.86
– 0.4 (V) = 30.3 V
1 – 0.86
where Vd is the diode forward voltage.
connected between the OVP pin and the output voltage of the
converter.
The Theoretical Maximum VOUT value must be greater than
the value VOUT(OVP) . If this is not the case, it may be necessary
to reduce the frequency to allow the boost to convert the voltage ratios.
Step 3a The first step is determining the maximum voltage based
Step 4 Selecting the inductor. The inductor must be chosen such
on the LED requirements. The regulation voltage, VLED , of the
that it can handle the necessary input current. In most applica-
Step 3 Determining the OVP resistor. The OVP resistor is
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Wide Input Voltage Range, High Efficiency
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A8514
tions, due to stringent EMI requirements, the system must operate
in continuous conduction mode throughout the whole input voltage range.
then:
L=
Step 4a Determining the duty cycle, calculated as follows:
D(max) =
=
VOUT(OVP) + Vd
VIN(min) + VOUT(OVP) + Vd
=
(38)
to the system. The minimum input current will dictate the inductor value. The maximum current rating will dictate the current
rating of the inductor. First, the maximum input current, given:
= 4
5 (V)
0.254 (A) 2 (MHz)
0.765 = 7.53 μH
than IIN(min):
Step 4b Determining the maximum and minimum input current
ILED
(43)
D(max)
Step 4d Double-check to make sure the ½ current ripple is less
15.9 (V) + 0.4 (V)
= 76.5%
5 (V) + 15.9 (V) + 0.4 (V)
IOUT = #CHANNELS
VIN(min)
ΔIL fSW
(39)
IIN(min) > 1/2 ΔIL
(44)
0.265 A > 0.127 A
A good inductor value to use would be 10 μH.
Step 4e Next insert the inductor value used in the design to deter-
mine the actual inductor ripple current:
0.060 (A) = 0.240 A
then:
IIN(max) =
=
VOUT(OVP) IOUT
VIN(min)
(40)
H
=
15.9 (V) 0.24 (A)
= 0.848 A
5 (V) 0.90
(45)
0.765
5 (V)
= 0.191 A
10 (μH) 2.0 (MHz)
current rating must be greater than the IIN(max) value plus half of
Next, calculate minimum input current, as follows:
VOUT(OVP) IOUT
IIN(min) =
VIN(max) H
15.9 (V)
16 (V)
VIN(min) D(max)
Lused fSW
Step 4f Determining the inductor current rating. The inductor
where η is efficiency.
=
ΔILused =
the ripple current ΔIL, calculated as follows:
(41)
0.24 (A)
= 0.265 A
0.90
Step 5 Determining the resistor value for a particular switching
tor operates in continuous conduction mode, the value of the
inductor must be set such that the ½ inductor ripple current is not
greater than the average minimum input current. As a first pass
assume Iripple to be 30% of the maximum inductor current:
= 0.848 × 0.30 = 0.254 A
(46)
= 0.848 (A) + 0.096 (A) = 0.944 A
Step 4c Determining the inductor value. To ensure that the induc-
ΔIL = IIN(max) × Iripple
L(min) = IIN(max) + 1/2 ΔILused
(42)
frequency. Use the RFSET values shown in figure 7. For example,
a 10 kΩ resistor will result in a 2 MHz switching frequency.
Step 6 Choosing the proper switching diode. The switching diode
must be chosen for three characteristics when it is used in LED
lighting circuitry. The most obvious two are: current rating of the
diode and reverse voltage rating.
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Wide Input Voltage Range, High Efficiency
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A8514
The reverse breakdown voltage rating for the output diode in a
SEPIC circuit should be:
VBD > VOUT(OVP)(max) + VIN(max)
(47)
> 15.9 (V) + 16 (V) = 31.9 V
because the maximum output voltage in this case is VOUT(OVP).
The peak current through the diode is calculated as:
Idp = IIN(max) + 1/2 ΔILused
A capacitor larger than 3.96 μF should be selected due to degradation of capacitance at high voltages on the capacitor. Select a
4.7 μF capacitor for this application.
The rms current through the capacitor is given by:
ICOUTrms = IOUT
(48)
= 0.848 (A) + 0.096 (A) = 0.944 A
The third major component in deciding the switching diode is the
reverse current, IR , characteristic of the diode. This characteristic
is especially important when PWM dimming is implemented.
During PWM off-time the boost converter is not switching. This
results in a slow bleeding off of the output voltage, due to leakage
currents. IR can be a large contributor, especially at high temperatures. On the diode that was selected in this design, the current
varies between 1 and 100 μA. It is often advantageous to pick a
diode with a much higher breakdown voltage, just to reduce the
reverse current. Therefore for this example, pick a diode rated for
a VBD of 60 V, instead of just 40 V.
= 0.240 (A)
COUT = ILK
1 – D(min)
fPWM(dimming) V
COUT
= 200 (μA)
(49)
(50)
0.765
= 0.433 A
1 – 0.765
The output capacitor must have a ripple current rating of at least
500 mA. The capacitor selected for this design is a 4.7 μF 50 V
capacitor with a 1.5 A current rating.
Step 8 Selecting input capacitor. The input capacitor must be
selected such that it provides a good filtering of the input voltage
waveform. A estimation rule is to set the input voltage ripple,
ΔVIN , to be 1% of the minimum input voltage. The minimum
input capacitor requirements are as follows:
CIN =
Step 7 Choosing the output capacitors. The output capacitors
must be chosen such that they can provide filtering for both the
boost converter and for the PWM dimming function. The biggest
factors that contribute to the size of the output capacitor are:
PWM dimming frequency and PWM duty cycle. Another major
contributor is leakage current, ILK . This current is the combination of the OVP leakage current as well as the reverse current of
the switching diode. In this design the PWM dimming frequency
is 200 Hz and the minimum duty cycle is 1%. Typically, the voltage variation on the output, VCOUT , during PWM dimming must
be less than 250 mV, so that no audible hum can be heard. The
capacitance can be calculated as follows:
D(max)
1 – D(max)
=
∆ILused
8
8
fSW
∆VIN
0.191 (A)
= 0.24 μF
2 (MHz) 0.05 (V)
The rms current through the capacitor is given by:
∆ILused
CINrms =
12
0.191 (A)
= 0.055 A
=
12
(51)
(52)
A good ceramic input capacitor with a rating of 2.2 μF 25 V will
suffice for this application.
1 – 0.01
= 3.96 μF
200 (Hz) 0.250 (V)
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Wide Input Voltage Range, High Efficiency
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A8514
Step 9 Selecting coupling capacitor CSW. The minimum capaci-
The rms current requirement of the coupling capacitor is given
by:
tance of CSW is related to the maximum voltage ripple allowed
across it:
CSW =
IOUT
1 – D(max)
D(max)
ICSWrms = IIN(max)
DMAX
fSW
(53)
∆VSW
0.24 (A) 0.765
=
= 0.92 μF
0.1 (V) 2 (MHz)
= 0.848 (A)
(54)
1 – 0.765
= 0.47 A
0.765
The voltage rating of the coupling capacitor must be greater than
VIN(max), or 16 V in this case. A ceramic capacitor rated for
2.2 μF 25 V will suffice for this application.
L2
10 μH
VIN
9 to 16 V
CIN
2.2 μF
RSC
0.033 Ω
RADJ
249 Ω
Q1
RC
20 Ω
CC
22 nF
VC
CVDD
0.1 μF
100 kΩ
CSW
2.2 μF
L1
10 μH
GATE
VSENSE
VIN
VDD
RISET
11 kΩ
RFSET
10 kΩ
A8514
OVP
PAD
LED1
FSET/SYNC
AGND
VOUT
ROVP
39.2 kΩ
SW
FAULT
PWM/EN
APWM
ISET
D1
2 A / 60 V
COUT
4.7 μF
LED2
LED3
LED4
COMP
PGND
CP
120 pF
RZ 150 Ω
CZ 0.47 μF
Figure 35. Typical application showing SEPIC configuration, with accurate input current sense, and VSENSE
to ground protection.
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Wide Input Voltage Range, High Efficiency
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A8514
Package LP, 20-Pin TSSOP with Exposed Thermal Pad
0.45
6.50±0.10
8º
0º
20
0.65
20
0.20
0.09
1.70
C
3.00
4.40±0.10
6.40±0.20
3.00
6.10
0.60 ±0.15
A
1
1.00 REF
2
4.12
0.25 BSC
20X
SEATING
PLANE
0.10 C
0.30
0.19
1.20 MAX
0.65 BSC
0.15
0.00
C
SEATING PLANE
GAUGE PLANE
1
2
4.12
B
PCB Layout Reference View
For Reference Only; not for tooling use (reference MO-153 ACT)
Dimensions in millimeters
Dimensions exclusive of mold flash, gate burrs, and dambar protrusions
Exact case and lead configuration at supplier discretion within limits shown
A Terminal #1 mark area
B Reference land pattern layout (reference IPC7351
SOP65P640X110-21M);
All pads a minimum of 0.20 mm from all adjacent pads; adjust as
necessary to meet application process requirements and PCB layout
tolerances; when mounting on a multilayer PCB, thermal vias at the
exposed thermal pad land can improve thermal dissipation (reference
EIA/JEDEC Standard JESD51-5)
C Exposed thermal pad (bottom surface); dimensions may vary with device
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Wide Input Voltage Range, High Efficiency
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A8514
Revision History
Revision
Revision Date
Rev. 5
May 20, 2013
Description of Revision
Update application information, add appendix A
Copyright ©2011-2013, Allegro MicroSystems, LLC
Allegro MicroSystems, LLC reserves the right to make, from time to time, such departures from the detail specifications as may be required to
permit improvements in the performance, reliability, or manufacturability of its products. Before placing an order, the user is cautioned to verify that
the information being relied upon is current.
Allegro’s products are not to be used in life support devices or systems, if a failure of an Allegro product can reasonably be expected to cause the
failure of that life support device or system, or to affect the safety or effectiveness of that device or system.
The information included herein is believed to be accurate and reliable. However, Allegro MicroSystems, LLC assumes no responsibility for its
use; nor for any infringement of patents or other rights of third parties which may result from its use.
For the latest version of this document, visit our website:
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Wide Input Voltage Range, High Efficiency
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A8514
Appendix A. Feed Back Loop Components Calculation
for Peak Current Control Boost Converter Used in LED Drivers Applications
This appendix provides an examination of the factors involved
in calculating the transfer function of a peak current controlled
boost converter, an output to control transfer function, and recommendations for stabilizing the feedback loop closed system.
An example of a complete small signal model of a peak-currentmode boost converter is shown in figure A-2. The A8514 is an
example of a boost converter that drives 4 LED strings with
10 LEDs in each string.
Power Stage Transfer Function
Using a frequency-based model, the transfer function (control to
output) of boost power stage peak-current control is given by the
following equation:
1+
TP(f )= AP ×
1+
2× ×f×j
2× ×f×j
× 1–
ωZ
ωRHP
2× ×f×j
2 × × f × j (2 × × f × j)2
–
× 1+
Q D × ωS
ωP
ωS2
(A-1)
AP is the DC gain,
ωZ is the angular frequency of the output capacitor
ESR zero, fZ ,
ωRHP is the angular frequency of the right-half plane
zero, fRHP ,
ωP is the angular frequency of the output load pole, fP ,
QD is the inductor peak current sampling double pole quality or
damping factor, and
AP , DC gain
The DC gain is defined as follows:
AP =
RS × REQ
1– D(nom)
×
RI
RS + RD + REQ
(A-3)
where
• D is the PWM duty cycle, calculated as:
where
D(nom) = (VOUT – VIN(nom)) / VOUT
(A-4)
VOUT = NL × Vf + VREG + VD
(A-5)
and
NL is the quantity of LEDs per string,
Vf is the nominal forward voltage drop for each
LED diode,
VREG is the current sink regulated voltage for each
LED string, and
VD is the Schottky diode forward voltage drop.
• RI is the current sense resistor, which is connected in series
with the boost power switch,
• RS is the LED sink pin sense resistor, which is usually located
inside the IC and can be calculated from the following
equation:
RS = VREG / ILED
(A-6)
where ILED is the current through one LED string,
ωS is the double-pole angular frequency oscillation .
Figure A-1 shows the plot of the power stage logarithmic transfer
function as gain, GP(f) , versus frequency. with GP(f) given by:
The next sections define the components of TP(f).
(A-2)
0
Gain, GP(f)
GP(f) = 20 × log( |TP(f)| )
¾–100
10
100
1¾×103
1¾× 104
1¾×105
1¾×106
1¾× 107
1¾×108
Frequency (Hz)
Figure A-1. Plot of power stage transfer function versus frequency
A8514-APPXA
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Wide Input Voltage Range, High Efficiency
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A8514
• REQ is the output nominal operating resistance, which is given
by the following equation:
REQ = VOUT / ILEDT
(A-7)
where ILEDT is the total output current through all LED
strings:
ILEDT = NS × ILED
(A-8)
and NS is the total quantity of LED strings, and
1. Apply a voltage, V1 , from one end of an LED string to the
other end.
2. Measure the current, I1 , through the string.
3. Repeat for another voltage, V2 , and measure the current, I2 .
4. Calculate the result as:
RD = (V2 – V1) / (I2 – I1)
(A-9)
QD , inductor peak current sampling double pole quality
1
× [0.5 – D(nom) + ( 1 – D(nom)) × IFSC]
(A-10)
where
IFSC is the implemented factor of inductor slope compensation,
and is given by:
IFSC = ( ISC / CSC ) × FSC
(A-11)
and
ISC is the IC implemented slope compensation in A/μs, which
can be taken from the IC datasheet, and
CSC is the calculated slope compensation also in A/μs,
given by:
CSC =
∆I × FSC × 10–6
(1/ fSW)× (1– D(max))
(A-12)
and
ΔI = (VIN(min) × D(max)) / L1 × fSW , and
(A-13)
FSC is the Ridley’s factor slope compensation, given by:
FSC = 1 – 0.18 / D(max)
(A-15)
ωRHP , angular frequency of the right-half plane
zero, fRHP
ωRHP = REQ / (1 – D(max))2 × L1 )
(A-16)
where
• RD is the total dynamic resistance of an LED string, which
can be taken from the datasheets of the LEDs being used, or
measured in the lab, as follows:
QD =
ωZ , angular frequency of the output capacitor
ESR zero, fZ
ωZ = 1 / (ESR × COUT )
(A-14)
D(max) = (VOUT – VIN(min)) / VOUT
(A-17)
ωP , angular frequency of the output load pole, fP
ωP =
RS + RD + REQ
(RS + RD + ESR) × REQ × COUT
(A-18)
ωS , angular frequency oscillation of the double pole that
occurs at half of the switching frequency, fSW
ωS = π × fSW
(A-19)
Output to Control Transfer Function
When using peak current mode control for a DC-to-DC converter,
a type II PI error amplifier compensation circuit is sufficient to
stabilize the converter. For controlling the current sink voltage
and as a result controlling the output, the A8514 IC uses a high
bandwidth transconductance amplifier, shown as A1 in figure A-2.
A transconductance amplifier is actually a voltage-controlled
current source. It converts any error voltage at its input pins to a
current flowing out of its output pin at VC. The transconductance
gain of the error amplifier, g , is defined as:
g = IAMP / Verror
(A-20)
In figure A-2, RAMP represents the output impedance of the
transconductance amplifier (A1). RAMP usually has a high value
and it is neglected in the calculation of the error amplifier transfer
function.
RZ, CZ , and CP represent the external Type II compensation network. From an AC point of view, the non-inverting pin of A1 is
connected to a DC reference voltage, VREG , which is a virtual
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A-2
Wide Input Voltage Range, High Efficiency
Fault Tolerant LED Driver
A8514
AC ground. Therefore, the transfer function of the compensation
circuit is derived as follows:
TEA(f ) =
=
(A-21)
VC(f)
VOUT(f)
tion to the pole at the origin. The pole at the origin is defined
by CP and RAMP . The zero is defined by RZ and CZ . The zero
frequency location is selected to compensate or cancel the power
train load pole. It is defined by:
fZEA = 1 / (2 × π × RZ × CZ )
–1 × IAMP × ZC (f)
RS + RD
RS
VERROR ×
(A-26)
100
(A-22)
TEA(f ) = –1 ×
RS × g
RS+RD
(A-23)
× ZC (f )
where
ZC(f ) =
1
1
RZ + 2 × × f × j× C × 2 × × f × j× C
Z
P
RZ +
1
1
+
2 × × f × j× CZ
2 × × f × j× CP
V IN
PWM
and
Driver
Mid-Band Gain
(A-24)
¾ 50
10
100
1¾×10 3
1¾×10 4
1¾×10 5
1¾× 10 6
1¾×10 7
1¾×108
Frequency (Hz)
(A-25)
The transfer function has a single pair of pole and zero in addi-
L1
50
0
Figure A-3 shows the logarithmic transfer function for the output
to control compensation circuit, with gain, GEA(f). given by:
GEA(f) = 20 × log( |TEA(f)| )
Gain, GP(f)
applying equation A-20:
Figure A-3. Plot of error amplifier stage transfer function versus frequency
VOUT
D1
COUT
Current Sense
Amplifier
D1
A2
Q1
ESR
RI
D2
D10
Adder
Slope
Compensation
Transconductance
(g) Amplifier
Vc
A1
Rz
Cp
Ram p
Vreg
Rs
Cz
Figure A-2. Small signal model of a peak-current-mode boost converter;
the four strings of the A8514 are represented by one string in this example
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The error amplifier pole frequency is selected to compensate for or cancel the power train ESR zero. This is the case if
the frequency of the ESR zero is small or below the switching frequency. Otherwise, it is selected to be at half switching
frequency. This pole frequency determines the end of mid-band
gain of the error amplifier transfer function, so it ensures that the
closed loop system cross-over frequency is below half switching frequency, which is important for stability issues. The pole
frequency is defined by:
fPEA =
1c. To compensate for the difference from the error amplifier gain
at fZEA and the actual mid-band gain, subtract an additional 3 dB:
–GP(fC) –3 dB
1d. Convert the calculated gain to a linear gain:
10
CZ × CP
CZ + CP
(A-27)
– GP(fC) –3
20
1e. Calculate RZ:
1
2 × × RZ ×
(A-29)
10
RZ =
g×
– GP(fC) –3
20
RS
RS+RD
(A-29)
(A-30)
Stabilizing the Closed Loop System
2. Select a value for CZ.
In this section, calculations are provided for selecting optimal
RZ , CZ , and CP . The closed loop system will be stable if the total
system transfer function rolls off while crossing over at a phase
margin of approximately 90° or –20 dB per decade. It is recommended that the phase margin does not fall below 45°. For higher
stability, the cross over frequency should be much less than
the right half plane zero and smaller than half of the switching
frequency.
2a. Calculate the frequency for the error-amplifier compensation
zero, fZEA . This zero should cancel the dominant low frequency
pole of power train. Therefore, fZEA should be close to fP . Usually it is selected to be 1/5 to 1/10 of fC:
To achieve that, first fix the mid-band gain of the error amplifier
transfer function. Make it equal in value to the power train gain
at the cross over frequency, but negative so the total closed loop
gain will be 0 dB. Then position the compensation pole and zero.
Here are step-by-step procedures on how to calculate the compensation network components:
1. Calculate RZ such that the negative mid-band gain of the error
amplifier will be equal to the power train gain at the required
system bandwidth or cross over frequency.
1a. Calculate the cross over frequency to be much less than
the RHP zero and lower than the half-switching frequency. A
20 to 30 kHz cross over frequency is appropriate for LED applications, calculated as follows:
fC = 0.015 × fSW
(A-28)
1b. Calculate, or preferably measure, the power train gain at fC ,
which is GP(fC ), then multiply it by –1.
fZEA = fC / 10
(A-31)
2b. Cz can be calculated by applying equation A-26:
CZ = 1 / (2 × π × RZ × fZEA )
(A-32)
3. Select a value for CP .
3a. Select a frequency for the error-amplifier compensation
pole, fPEA . This pole determines the error-amplifier end of the
mid-band region. It is selected to cancel the power train ESR
zero. However, if ceramic capacitors are used at the output, the
ESR zero will be at very high frequency. In this case, the fPEA is
selected to be at half of the switching frequency to ensure that
fC is at lower than half the switching frequency and as a result a
higher phase margin can be achieved. fPEA is given by:
fPEA = 0.5 × fSW
(A-33)
3b. CP can be calculated by applying equation A-27:
CP =
CZ
2 × × RZ × CZ × fPEA – 1
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(A-34)
A-4
Wide Input Voltage Range, High Efficiency
Fault Tolerant LED Driver
A8514
The closed-loop system transfer function is given by:
TS(f ) = TP(f ) × TEA(f )
(A-35)
The closed-loop system logarithmic transfer function gain is
given by:
GS(f ) = 20 × log(|TS(f )|)
(A-36)
Figure A-4 shows the closed loop logarithmic transfer function as
gain versus frequency. As shown in figure A-4, if the above methods are implemented the transfer function rolls off while crossing
over with around a –20 dB per decade, which results in around a
90° phase margin.
Finally, it is recommended to measure the gain and phase margin
of the whole system closed loop. If necessary, the compensation
components values could be tweaked to obtain the required cross
over frequency and phase margin.
Gain, GS(f)
100
0
¾ 100
10
100
1¾×103
1¾×104
1¾×105
1¾×106
1¾×107
1¾×108
Frequency (Hz)
Figure A-4. Plot of the whole system closed loop transfer function gain versus
frequency, with a cross over frequency, fC , of 30 kHz
Allegro MicroSystems, LLC
115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
A-5
Wide Input Voltage Range, High Efficiency
Fault Tolerant LED Driver
A8514
control loop at very low frequency.
Measuring the Feedback Loop Gain and Phase Margin
It is always necessary to measure the feedback loop gain and
phase margin of a power converter to make sure the converter
runs stably and responds quickly to line or load transients. In
addition, to calculate the feedback-loop component values, it
is necessary first to calculate or preferably to measure only the
power-stage transfer function at the required cross over frequency. Below, one method for measuring the power-stage and
the closed-loop whole system transfer functions is presented.
2. On the PCB cut the trace between VOUT and the LED strings.
3. Connect a 10 Ω resistor from VOUT to the LED strings.
4. Connect the sweeping signal, VS, leads from the spectrum
analyzer line (red) to VOUT and the neutral (black) to the LED
string, across the 10 Ω resistor.
5. Hook the voltage probe V2 (red) to VOUT (B1) and the ground
lead to PCB GND.
Power Stage Transfer Function Measurement
The power stage or control to output transfer function can be measured using any gain/phase analyzer. Figure A-5 shows a block
diagram for the whole closed-loop system. To measure the powerstage transfer function, implement the following steps:
1. First, temporarily, use a large value capacitor for CZ , say
4.7 μF, and a small value resistor for RZ , say 100 Ω, to roll-off the
6. Hook the voltage probe V1 (blue) to VC, so the gain would be
GP(f) = B1 / A2.
7. Run the sweep.
8. When the sweep is completed, to read the power stage gain
GP(fC) at the selected frequency, fC , place the analyzer screen
cursor at that frequency.
PWM
Q1
B1
VOUT
Driver
A2
Vc
A1
U1
–
COUT
Vs
AC Sweeping Signal
LED Strings
Cz
–
+
+
Rz
Cp
+
–
R1
10 ohm
Vreg
I-string
Figure A-5. Simplified block diagram for the closed-loop whole system to show how to measure
the gain of the power stage or closed-loop system gain and phase margin
Allegro MicroSystems, LLC
115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
A-6
A8514
Wide Input Voltage Range, High Efficiency
Fault Tolerant LED Driver
Whole Closed-Loop System Transfer Function Gain and
Phase Margin Measurement
The closed-loop whole system transfer function gain and phase
margin can be measured using the following steps:
1. Change RZ , CZ , and CP to be the same as the calculated values.
2. Follow same steps 2 through 5, shown above.
3. Hook the voltage probe V1 (blue) to A1, so the gain would be
GS(f) = B1 / A1.
4. Run the sweep.
5. When the sweep is completed, to read the phase margin at the
cross over frequency, fC, place the analyzer screen cursor at fC.
6. To read the gain margin, place the analyzer screen cursor where
the phase margin is zero.
The whole system closed loop is considered stable if the phase
margin is larger than 45°. It is also recommended to have the
gain margin as large as possible. Larger than around –7 dB
is sufficient.
Allegro MicroSystems, LLC
115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
A-7

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