ISL6532A
®
Data Sheet
May 5, 2008
ACPI Regulator/Controller for Dual
Channel DDR Memory Systems
FN9099.5
Features
The ISL6532A provides a complete ACPI compliant power
solution for up to 4 DIMM dual channel DDR/DDR2 Memory
systems. Included are both a synchronous buck controller
and integrated LDO to supply VDDQ with high current during
S0/S1 states and standby current during S3 state. During
S0/S1 state, a fully integrated sink-source regulator
generates an accurate (VDDQ/2) high current VTT voltage
without the need for a negative supply. A buffered version of
the VDDQ/2 reference is provided as VREF. An LDO
controller is also integrated for AGP core voltage regulation.
• Generates 3 Regulated Voltages
- Synchronous Buck PWM Controller with Standby LDO
- 3A Integrated Sink/Source Linear Regulator with
Accurate VDDQ/2 Divider Reference.
- Glitch-free Transitions During State Changes
- LDO Regulator for 1.5V Video and Core voltage
• Acpi Compliant Sleep State Control
• Integrated VREF Buffer
• PWM Controller Drives Low Cost N-Channel MOSFETs
The switching PWM controller drives two N-Channel
MOSFETs in a synchronous-rectified buck converter
topology. The synchronous buck converter uses voltagemode control with fast transient response. Both the switching
regulator and standby LDO provide a maximum static
regulation tolerance of ±2% over line, load, and temperature
ranges. The output is user-adjustable by means of external
resistors down to 0.8V.
• 250kHz Constant Frequency Operation
Switching memory core output between the PWM regulator
and the standby LDO during state transitions is
accomplished smoothly via the internal ACPI control
circuitry. The NCH signal provides synchronized switching of
a backfeed blocking switch during the transitions eliminating
the need to route 5V Dual to the memory supply.
• Simple Single-Loop Voltage-Mode PWM Control Design
An integrated soft-start feature brings all outputs into
regulation in a controlled manner when returning to S0/S1
state from any sleep state. During S0 the PGOOD signal
indicates VTT is within spec and operational.
Each output is monitored for under and overvoltage events.
The switching regulator has overcurrent protection. Thermal
shutdown is integrated.
Ordering Information
PART
NUMBER
PART
MARKING
ISL6532ACR*, **
ISL 6532ACR
ISL6532AIRZ*
(Note)
• 5V or 3.3V Down Conversion
• Fully-Adjustable Outputs with Wide Voltage Range: Down
to 0.8V supports DDR and DDR2 Specifications
• Fast PWM Converter Transient Response
• Under and Overvoltage Monitoring on All Outputs
• OCP on the Switching Regulator
• Integrated Thermal Shutdown Protection
• QFN Package Option
- QFN Compliant to JEDEC PUB95 MO-220 QFN - Quad
Flat No Leads - Product Outline
- QFN Near Chip Scale Package Footprint; Improves
PCB Efficiency, Thinner in Profile
• Pb-free Available (RoHS Compliant)
Applications
TEMP.
RANGE
(°C)
PACKAGE
PKG.
DWG. #
0 to +70 28 Ld 6x6 QFN L28.6x6
ISL6532ACRZ*, ** ISL6532 ACRZ 0 to +70 28 Ld 6x6 QFN L28.6x6
(Note)
• Tight Output Voltage Regulation
- All Outputs: ±2% Over-Temperature
(Pb-free)
ISL6532 AIRZ -40 to +85 28 Ld 6x6 QFN L28.6x6
(Pb-free)
• Single and Dual Channel DDR Memory Power Systems in
ACPI compliant PCs
• Graphics Cards - GPU and Memory Supplies
• ASIC Power Supplies
• Embedded Processor and I/O Supplies
• DSP Supplies
*Add “-T” suffix for tape and reel.
**Add “-TK” suffix for tape and reel. Please refer to TB347 for details on
reel specifications
NOTE: These Intersil Pb-free plastic packaged products employ special
Pb-free material sets; molding compounds/die attach materials and 100%
matte tin plate PLUS ANNEAL - e3 termination finish, which is RoHS
compliant and compatible with both SnPb and Pb-free soldering
operations. Intersil Pb-free products are MSL classified at Pb-free peak
reflow temperatures that meet or exceed the Pb-free requirements of
IPC/JEDEC J STD-020.
1
CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures.
1-888-INTERSIL or 1-888-468-3774 | Intersil (and design) is a registered trademark of Intersil Americas Inc.
Copyright © Intersil Americas Inc. 2002-2004, 2007. All Rights Reserved
All other trademarks mentioned are the property of their respective owners.
ISL6532A
Pinout
GNDP
LGATE
UGATE
P12V
S5#
S3#
NCH
ISL6532A
(28 LD QFN)
TOP VIEW
28
27
26
25
24
23
22
GNDP
1
21
PGOOD
5VSBY
2
20
PHASE
GNDQ
3
19
DRIVE2
18
FB2
GND
29
6
16
COMP
VDDQ
7
15
FB
8
9
10
11
12
13
14
VREF_IN
VTT
VREF_OUT
GNDA
OCSET
17
P5VSBY
5
VTTSNS
VTT
VDDQ
4
VDDQ
GNDQ
2
FN9099.5
May 5, 2008
Block Diagram
P5VSBY
S3#
VDDQ S3
REGULATOR
S5#
5VSBY
VOLTAGE
REFERENCE
0.800V
+
-
0.680V (-15%)
VDDQ(3)
5V
0.920V (+15%)
12VCC
3
POR
VTTSNS
+
VTT
- REG
+
VTT(2)
S3
GNDQ
S0
DISABLE
+
FB2
UV/OV3
NCH
UV/OV
SLEEP,
SOFT-START,
PGOOD,
AND FAULT
LOGIC
PWM ENABLE
S0/S3
12V
POR
SOFT-START
+
-
RU
VREF_IN
OSCILLATOR
{
RL
DRIVE2
650Ω OUTPUT
IMPEDANCE
ISL6532A
{
EA2
+
P12V
PWM
PWM
LOGIC
UGATE
250kHz
+
-
GNDA
+
COMP
EA1
UV/OV1
PHASE
+
OC
COMP
20μA
+
-
LGATE
UV/OV2
VREF_OUT
PGOOD
FB
COMP
OCSET
GNDP
FN9099.5
May 5, 2008
ISL6532A
Simplified Power System Diagram
12V
5VSBY
5V
ISL6532A
NCH
SLP_S3
SLEEP
STATE
LOGIC
SLP_S5
Q1
VDDQ
PWM
CONTROLLER
+
Q2
5VSBY/3V3SBY
STANDBY
LDO
VDDQ
VREF
LINEAR
CONTROLLER
Q3
VTT
REGULATOR
VAGP
VTT
+
+
Typical Application - 5V or 3.3V Input
5VSBY
+12V
+3.3V
+5V OR +3.3V
P12V
P5VSBY
5VSBY
CBP
RNCH
PGOOD
VDDQ
S3#
SLP_S3
SLP_S5
VREF
NCH
Q4
S5#
VREF_OUT
+
OCSET
ROCSET
VREF_IN
UGATE
+
Q1
PHASE
VTT
VTT
VDDQ
LOUT
2.5V
+
ISL6532A
LGATE
VTT
VDDQ
VDDQ
VDDQ
+
CVTT_OUT
VDDQ
CIN
Q2
CVDDQ_OUT
GNDQ
GNDQ
VTTSNS
DRIVE2
Q3
FB
COMP
VAGP
1.5V
FB2
+
GNDP
GNDA
COUT2
4
FN9099.5
May 5, 2008
ISL6532A
Typical Application - Input From 5V Dual
5VSBY
+12V
5V DUAL
P12V
P5VSBY
CBP
5VSBY
+3.3V
PGOOD
VDDQ
S3#
SLP_S3
VREF
NCH
S5#
SLP_S5
VREF_OUT
+
OCSET
UGATE
Q1
PHASE
VTT
VTT
VDDQ
LOUT
2.5V
+
ISL6532A
LGATE
VTT
+
CIN
ROCSET
VREF_IN
VDDQ
VDDQ
VDDQ
CVTT_OUT
VDDQ
Q2
CVDDQ_OUT
GNDQ
GNDQ
VTTSNS
DRIVE2
Q3
FB
VAGP
COMP
1.5V
FB2
+
GNDP
GNDA
COUT2
5
FN9099.5
May 5, 2008
ISL6532A
Absolute Maximum Ratings
Thermal Information
5VSBY, P5VSBY . . . . . . . . . . . . . . . . . . . . . . . . . GND - 0.3V to +7V
P12V . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . GND - 0.3V to +14V
UGATE, LGATE, NCH . . . . . . . . . . . . . . GND - 0.3V to P12V + 0.3V
All other Pins . . . . . . . . . . . . . . . . . . . . GND - 0.3V to 5VCC + 0.3V
ESD Classification . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . LEVEL 1
Thermal Resistance (Typical, Notes 1, 2) θJA (°C/W) θJC (°C/W)
QFN Package . . . . . . . . . . . . . . . . . . .
32
5
Maximum Junction Temperature (Plastic Package) . . . . . . +150°C
Maximum Storage Temperature Range . . . . . . . . . -65°C to +150°C
Pb-free reflow profile . . . . . . . . . . . . . . . . . . . . . . . . . .see link below
http://www.intersil.com/pbfree/Pb-FreeReflow.asp
Recommended Operating Conditions
Supply Voltage on 5VSBY . . . . . . . . . . . . . . . . . . . . . . . . +5V ±10%
Supply Voltage on P12V . . . . . . . . . . . . . . . . . . . . . . . . +12V ±10%
Supply Voltage onP5VSBY . . . . . . . . . . . . . . . . . . . . . . . +5V ±10%
Commercial Ambient Temperature Range. . . . . . . . . . 0°C to +70°C
Industrial Ambient Temperature Range . . . . . . . . . . -40°C to +85°C
Junction Temperature Range. . . . . . . . . . . . . . . . . -40°C to +125°C
CAUTION: Do not operate at or near the maximum ratings listed for extended periods of time. Exposure to such conditions may adversely impact product reliability and
result in failures not covered by warranty.
NOTES:
1. θJA is measured in free air with the component mounted on a high effective thermal conductivity test board with “direct attach” features. See
Tech Brief TB379.
2. For θJC, the “case temp” location is the center of the exposed metal pad on the package underside.
3. Limits established by characterization and are not production tested.
Electrical Specifications
Recommended Operating Conditions, Industrial Temperature Range, Unless Otherwise Noted. Refer to Block
and Simplified Power System Diagrams and Typical Application Schematics
PARAMETER
SYMBOL
TEST CONDITIONS
MIN
TYP
MAX
UNITS
ICC_S0
S3# and S5# HIGH, UGATE/LGATE Open
3.00
5.25
7.25
mA
ICC_S3
S3# LOW, S5# HIGH, UGATE/LGATE
Open
3.50
-
5
mA
ICC_S5
S5# LOW, S3# Don’t Care, UGATE/LGATE
Open
0.3
-
0.925
mA
Rising 5VSBY POR Threshold
4.00
-
4.35
V
Falling 5VSBY POR Threshold
3.55
-
3.95
V
Rising P12V POR Threshold
10.0
-
10.6
V
Falling P12V POR Threshold
8.80
-
9.75
V
220
250
280
kHz
5VSBY SUPPLY CURRENT
Nominal Supply Current
POWER-ON RESET
OSCILLATOR AND SOFT-START
PWM Frequency
fOSC
PWM Frequency
fOSC
200
240
280
kHz
Ramp Amplitude
ΔVOSC
-
1.5
-
V
Error Amp Reset Time
tRESET
Mechanical Off/S5 to S0
6.5
-
10
ms
tSS
Mechanical Off/S5 to S0
6.5
-
10
ms
0.784
0.800
0.816
V
0.780
0.800
0.820
V
Note 3
-
80
-
dB
GBWP
Note 3
15
-
-
MHz
SR
Note 3
-
6
-
V/μs
VDDQ Soft-Start Interval
Commercial Temperature Range
REFERENCE VOLTAGE
Reference Voltage
VREF
Reference Voltage
VREF
Commercial Temperature Range
PWM CONTROLLER ERROR AMPLIFIER
DC Gain
Gain-Bandwidth Product
Slew Rate
6
FN9099.5
May 5, 2008
ISL6532A
Electrical Specifications
Recommended Operating Conditions, Industrial Temperature Range, Unless Otherwise Noted. Refer to Block
and Simplified Power System Diagrams and Typical Application Schematics (Continued)
PARAMETER
SYMBOL
TEST CONDITIONS
MIN
TYP
MAX
UNITS
STATE LOGIC
S3# Transition Level
VS3
-
1.5
-
V
S5# Transition Level
VS5
-
1.5
-
V
UGATE and LGATE Source
IGATE
-
-0.8
-
A
UGATE and LGATE Sink
IGATE
-
0.8
-
A
-
-
6
mA
9.0
9.5
10.0
V
P5VSBY = 5.0V
-
-
650
mA
P5VSBY = 3.3V
-
-
550
mA
PWM CONTROLLER GATE DRIVERS
NCH BACKFEED CONTROL
NCH Current Sink
INCH
NCH Trip Level
VNCH
NCH = 0.8V
VDDQ STANDBY LDO
Output Drive Current
VTT REGULATOR
Upper Divider Impedance
RU
-
2.5
-
kΩ
Lower Divider Impedance
RL
-
2.5
-
kΩ
IVREF_OUT
-
-
2
mA
Periodic load applied with 30% duty cycle
and 10ms period using ISL6532AEVAL1
evaluation board (see Application Note
AN1056)
-3
-
3
A
Note 3
-
80
-
dB
GBWP
Note 3
9
-
-
MHz
SR
Note 3
-
6
-
V/μs
DRIVE2 High Output Voltage
10.0
10.2
-
V
DRIVE2 Low Output Voltage
-
0.16
0.40
V
DRIVE2 High Output Source Current
-.5
-1.4
-
mA
DRIVE2 Low Output Sink Current
.85
1.3
-
mA
VREF_OUT Buffer Source Current
Maximum VTT Load Current
IVTT_MAX
LINEAR REGULATOR
DC GAIN
Gain Bandwidth Product
Slew Rate
PGOOD
PGOOD Rising Threshold
VVTTSNS/VVDDQ S0
-
57.5
-
%
PGOOD Falling Threshold
VVTTSNS/VVDDQ S0
-
45.0
-
%
15
20
22.5
μA
PROTECTION
OCSET Current Source
IOCSET
VDDQ OV Level
VFB/VREF
S0
-
115
-
%
VDDQ UV Level
VFB/VREF
S0
-
85
-
%
Linear Regulator OV Level
VFB2/VREF
S0
-
115
-
%
Linear Regulator UV Level
VFB2/VREF
S0
-
85
-
%
Note 3
-
140
-
°C
Thermal Shutdown Limit
TSD
7
FN9099.5
May 5, 2008
ISL6532A
Functional Pin Description
5VSBY (Pin 2)
5VSBY is the bias supply of the ISL6532A. It is typically
connected to the 5V standby rail of an ATX power supply.
During S4/S5 sleep states the ISL6532A enters a reduced
power mode and draws less than 1mA (ICC_S5) from the
5VSBY supply. The supply to 5VSBY should be locally
bypassed using a 0.1μF capacitor.
P12V (Pin 25)
P12V provides the gate drive to the switching MOSFETs of
the PWM power stage. The VTT regulation circuit and the
Linear Driver are also powered by P12V. P12V is not
required except during S0/S1/S2 operation. P12V is typically
connected to the +12V rail of an ATX power supply.
5VSBY (Pin 11)
The FB pin is also monitored for under and overvoltage
events.
PHASE (Pin 20)
Connect this pin to the upper MOSFET’s source. This pin is
used to monitor the voltage drop across the upper MOSFET
for overcurrent protection.
OCSET (Pin 12)
Connect a resistor (ROCSET) from this pin to the drain of the
upper MOSFET, ROCSET, an internal 20μA current source
(IOCSET), and the upper MOSFET ON-resistance (rDS(ON)).
Set the converter overcurrent (OC) trip point according to
Equation 1:
I OCSET xR OCSET
I PEAK = -----------------------------------------------r DS ( ON )
(EQ. 1)
An overcurrent trip cycles the soft-start function.
This pin provides the VDDQ output power during S3 sleep
state. The regulator is capable of providing standby VDDQ
power from either the 5VSBY or 3.3VSBY rail. It is
recommended that the 5VSBY rail be used as the output
current handling capability of the standby LDO is higher than
with the 3.3VSBY rail.
GND, GNDA, GNDP, GNDQ (Pins 1, 3, 4, 17, 29)
The GND terminals of the ISL6532A provide the return path
for the VTT LDO, standby LDO and switching MOSFET gate
drivers. High ground currents are conducted directly through
the exposed paddle of the QFN package which must be
electrically connected to the ground plane through a path as
low in inductance as possible. GNDA is the Analog ground
pin, GNDQ is the return for the VTT regulator and GNDP is
the return for the upper and lower gate drives.
UGATE (Pin 26)
UGATE drives the upper (control) FET of the VDDQ
synchronous buck switching regulator. UGATE is driven
between GND and P12V.
LGATE (Pin 27)
LGATE drives the lower (synchronous) FET of the VDDQ
synchronous buck switching regulator. LGATE is driven
between GND and P12V.
FB (Pin 15) and COMP (Pin 16)
The VDDQ switching regulator employs a single voltage
control loop. FB is the negative input to the voltage loop error
amplifier. The positive input of the error amplifier is
connected to a precision 0.8V reference and the output of
the error amplifier is connected to the COMP pin. The VDDQ
output voltage is set by an external resistor divider
connected to FB. With a properly selected divider, VDDQ can
be set to any voltage between the power rail (reduced by
converter losses) and the 0.8V reference. Loop
compensation is achieved by connecting an AC network
across COMP and FB.
8
VDDQ (Pins 7, 8, 9)
The VDDQ pins should be connected externally together to
the regulated VDDQ output. During S0/S1 states, the VDDQ
pins serve as inputs to the VTT regulator and to the VTT
Reference precision divider. During S3 state, the VDDQ pins
serve as an output from the integrated standby LDO.
VTT (Pins 5, 6)
The VTT pins should be connected externally together.
During S0/S1 states, the VTT pins serve as the outputs of
the VTT linear regulator. During S3 state, the VTT regulator is
disabled.
VTTSNS (Pin 10)
VTTSNS is used as the feedback for control of the VTT linear
regulator. Connect this pin to the VTT output at the physical
point of desired regulation.
VREF_OUT (Pin 13)
VREF_OUT is a buffered version of VTT and also acts as the
reference voltage for the VTT linear regulator. It is
recommended that a minimum capacitance of 0.1μF is
connected between VDDQ and VREF_OUT and also
between VREF_OUT and ground for proper operation.
VREF_IN (Pin 14)
A capacitor, CSS, connected between VREF_IN and ground
is required. This capacitor and the parallel combination of
the Upper and Lower Divider Impedance (RU||RL), sets the
time constant for the start up ramp when transitioning from
S3 to S0/S1/S2.
The minimum value for CSS can be found using
Equation 2:
C VTTOUT ⋅ V DDQ
C SS > -----------------------------------------------10 ⋅ 2A ⋅ R U || R L
(EQ. 2)
FN9099.5
May 5, 2008
ISL6532A
The calculated capacitance, CSS, will charge the output
capacitor bank on the VTT rail in a controlled manner without
reaching the current limit of the VTT LDO.
ACPI compliance is realized through the SLP_S3 and
SLP_S5 sleep signals and through monitoring of the 12V
ATX bus.
NCH (Pin 22)
Initialization
NCH is an open-drain output that controls the MOSFET
blocking backfeed from VDDQ to the input rail during sleep
states. A 2kΩ or larger resistor is to be tied between the 12V
rail and the NCH pin. Until the voltage on the NCH pin
reaches the NCH trip level, the PWM is disabled.
The ISL6532A automatically initializes upon receipt of input
power. Special sequencing of the input supplies is not
necessary. The Power-On Reset (POR) function continually
monitors the input bias supply voltages. The POR monitors
the bias voltage at the 5VSBY and P12V pins. The POR
function initiates soft-start operation after the bias supply
voltages exceed their POR thresholds.
If NCH is not actively utilized, it still must be tied to the 12V
rail through a resistor. For systems using 5V dual as the
input to the switching regulator, a time constant, in the form
of a capacitor, can be added to the NCH pad to delay start of
the PWM switcher until the 5V dual has switched from
5VSBY to 5VATX.
PGOOD (Pin 21)
Power Good is an open-drain logic output that changes to a
logic low if any of the three regulators are out of regulation in
S0/S1/S2 state. PGOOD will always be low in any state
other than S0/S1/S2.
SLP_S5# (Pin 24)
ACPI State Transitions
COLD START (S4/S5 TO S0 TRANSITION)
At the onset of a mechanical start, the ISL6532A receives it’s
bias voltage from the 5V Standby bus (5VSBY). As soon as
the SLP_S3 and SLP_S5 have transitioned HIGH, the
ISL6532A starts an internal counter. Following a cold start or
any subsequent S4/S5 state, state transitions are ignored
until the system enters S0/S1. None of the regulators will
begin the soft-start procedure until the 5V Standby bus has
exceeded POR, the 12V bus has exceeded POR and VNCH
has exceeded the trip level.
This pin accepts the SLP_S5# sleep state signal.
SLP_S3# (Pin 23)
This pin accepts the SLP_S3# sleep state signal.
FB2 (Pin 18)
Connect the output of the external linear regulator to this pin
through a properly sized resistor divider. The voltage at this
pin is regulated to 0.8V. This pin is monitored for under and
overvoltage events.
DRIVE2 (Pin 19)
Connect this pin to the gate terminal of an external
N-Channel MOSFET transistor. This pin provides the gate
voltage for the linear regulator pass transistor. It also
provides a means of compensating the error amplifier for
applications requiring the transient response of the linear
regulator to be optimized.
Functional Description
Overview
The ISL6532A provides complete control, drive, protection
and ACPI compliance for a regulator powering DDR memory
systems. It is primarily designed for computer applications
powered from an ATX power supply. A 250kHz Synchronous
Buck Regulator with a precision 0.8V reference provides the
proper Core voltage to the system memory of the computer.
An internal LDO regulator with the ability to both sink and
source current and an externally available buffered
reference that tracks the VDDQ output by 50% provides the
VTT termination voltage. The ISL6532A also features an
LDO regulator for 1.5V AGP Video and Core voltage.
9
Once all of these conditions are met, the PWM error
amplifier will first be reset by internally shorting the COMP
pin to the FB pin. This reset lasts for 2048 clock cycles,
which is typically 8.2ms (one clock cycle = 1/fOSC). The
digital soft-start sequence will then begin.
The PWM error amplifier reference input is clamped to a
level proportional to the soft-start voltage. As the soft-start
voltage slews up, the PWM comparator generates PHASE
pulses of increasing width that charge the output
capacitor(s). The internal VTT LDO will also soft-start
through the reference that tracks the output of the PWM
regulator. The reference for the AGP LDO controller will rise
relative to the soft-start reference. The soft-start lasts for
2048 clock cycles, which is typically 8.2ms. This method
provides a rapid and controlled output voltage rise.
Figure 1 shows the soft-start sequence for a typical cold
start. Due to the soft-start capacitance, CSS, on the
VREF_IN pin, the S5 to S0 transition profile of the VTT rail
will have a more rounded features at the start and end of the
soft-start whereas the VDDQ profile has distinct starting and
ending points to the ramp up.
By directly monitoring 12VATX and the SLP_S3 and SLP_S5
signals the ISL6532A can achieve PGOOD status
significantly faster than other devices that depend on
Latched_Backfeed_Cut for timing.
ACTIVE TO SLEEP (S0 TO S3 TRANSITION)
When SLP_S3 goes LOW with SLP_S5 still HIGH, the
ISL6532A will disable the VTT linear regulator and the AGP
LDO controller. The VDDQ standby regulator will be enabled
FN9099.5
May 5, 2008
ISL6532A
should be noted that the soft-start profile of the VTT LDO
output will vary according to the value of the capacitor on the
VREF_IN pin.
S3
S5
12VATX 2V/DIV
S3
5VSBY
1V/DIV
VDDQ
500mV/DIV
S5
12VATX 2V/DIV
VAGP
500mV/DIV
VAGP
VTT
500mV/DIV
500mV/DIV
VDDQ
500mV/DIV
VTT_FLOAT
PGOOD
5V/DIV
2048 CLOCK
CYCLES
12V POR
VTT
500mV/DIV
2048 CLOCK
CYCLES
SOFT-START ENDS
SOFT-START
INITIATES PGOOD COMPARATOR
ENABLED
FIGURE 1. TYPICAL COLD START
and the VDDQ switching regulator will be disabled. NCH is
pulled low to disable the backfeed blocking MOSFET.
PGOOD will also transition LOW. When VTT is disabled, the
internal reference for the VTT regulator is internally shorted
to the VTT rail. This allows the VTT rail to float. When
floating, the voltage on the VTT rail will depend on the
leakage characteristics of the memory and MCH I/O pins. It
is important to note that the VTT rail may not bleed down to 0V.
The VDDQ rail will be supported in the S3 state through the
standby VDDQ LDO. When S3 transitions LOW, the Standby
regulator is immediately enabled. The switching regulator is
disabled synchronous to the switching waveform. The shut
off time will range between 4µs and 8µs. The standby LDO is
capable of supporting up to 650mA of load with P5VSBY tied
to the 5V Standby Rail. The standby LDO may receive input
from either the 3.3V Standby rail or the 5V Standby rail
through the P5VSBY pin. It is recommended that the 5V
Standby rail be used as the current delivery capability of the
LDO is greater.
SLEEP TO ACTIVE (S3 TO S0 TRANSITION)
When SLP_S3 transitions from LOW to HIGH with SLP_S5
held HIGH and after the 12V rail exceeds POR, the
ISL6532A will enable the VDDQ switching regulator, disable
the VDDQ standby regulator, enable the VTT LDO and force
the NCH pin to a high impedance state turning on the
blocking MOSFET. The AGP LDO goes through a 2048
clock cycle soft-start. The internal short between the VTT
reference and the VTT rail is released. Upon release of the
short, the capacitor on VREF_IN is then charged up through
the internal resistor divider network. The VTT output will
follow this capacitor charge up, and acting as the S3 to S0
transition soft-start for the VTT rail. The PGOOD comparator
is enabled only after 2048 clock cycles, or typically 8.2ms,
have passed following the S3 transition to a HIGH state.
Figure 2 illustrates a typical state transition from S3 to S0. It
10
PGOOD
5V/DIV
2048 CLOCK
CYCLES
12V POR
PGOOD COMPARATOR
ENABLED
FIGURE 2. TYPICAL S3 to S0 STATE TRANSITION
ACTIVE TO SHUTDOWN (S0 TO S5 TRANSITION)
When the system transitions from active (S0) state to
shutdown (S4/S5) state, the ISL6532A IC disables all
regulators and forces the PGOOD pin and the NCH pin
LOW.
VDDQ Overcurrent Protection (S0 State)
The overcurrent function protects the switching converter
from a shorted output by using the upper MOSFET ONresistance, rDS(ON), to monitor the current. This method
enhances the converter’s efficiency and reduces cost by
eliminating a current sensing resistor.
The overcurrent function cycles the soft-start function in a
hiccup mode to provide fault protection. A resistor (ROCSET)
programs the overcurrent trip level (see Typical Application
Diagrams on page 4 and page 5). An internal 20μA (typical)
current sink develops a voltage across ROCSET that is
referenced to the converter input voltage. When the voltage
across the upper MOSFET (also referenced to the converter
input voltage) exceeds the voltage across ROCSET, the overcurrent function initiates a soft-start sequence. The initiation
of soft-start will affect all regulators. The VTT regulator is
directly affected as it receives it’s reference from VDDQ. The
AGP LDO will also be soft-started, and as such, the AGP
LDO voltage will be disabled while the VDDQ regulator is
disabled.
Figure 3 illustrates the protection feature responding to an
overcurrent event. At time T0, an overcurrent condition is
sensed across the upper MOSFET. As a result, the regulator
is quickly shutdown and the internal soft-start function begins
producing soft-start ramps. The delay interval seen by the
output is equivalent to three soft-start cycles. The fourth
FN9099.5
May 5, 2008
ISL6532A
internal soft-start cycle initiates a normal soft-start ramp of
the output, at time T1. The output is brought back into
regulation by time T2 as long as the overcurrent event has
cleared.
VDDQ
A small ceramic capacitor should be placed in parallel with
ROCSET to smooth the voltage across ROCSET in the
presence of switching noise on the input voltage.
Overvoltage and Undervoltage Protection
All three regulators are protected from faults through internal
Overvoltage and Undervoltage detection circuitry. If the any
rail falls below 85% of the targeted voltage, then an
undervoltage event is tripped. An undervoltage will disable
all three regulators for a period of 3 soft-start cycles, after
which a normal soft-start is initiated. If the output is still under
85% of target, the regulators will continue to be disabled and
soft-started in a hiccup mode until the fault is cleared. This
protection feature works much the same as the VDDQ PWM
overcurrent protection works. See Figure 3.
VAGP
VTT
500mV/DIV
If the any rail exceeds 115% of the targeted voltage, then all
three outputs are immediately disabled. The ISL6532A will
not re-enable the outputs until either the bias voltage is
toggled in order to initiate a POR or the S5 signal is forced
LOW and then back to HIGH.
INTERNAL SOFT-START FUNCTION
DELAY INTERVAL
Thermal Protection (S0/S3 State)
T1
T0
T2
TIME
FIGURE 3. VDDQ OVERCURRENT PROTECTION AND
VTT/VAGP LDO UNDER VOLTAGE PROTECTION
RESPONSES
Had the cause of the overcurrent still been present after the
delay interval, the overcurrent condition would be sensed
and the regulator would be shut down again for another
delay interval of three soft-start cycles. The resulting hiccup
mode style of protection would continue to repeat indefinitely.
The overcurrent function will trip at a peak inductor current
(IPEAK) determined by:
I OCSET x R OCSET
I PEAK = ---------------------------------------------------r DS ( ON )
(EQ. 3)
where IOCSET is the internal OCSET current source (20μA
typical). The OC trip point varies mainly due to the MOSFET
rDS(ON) variations. To avoid overcurrent tripping in the
normal operating load range, find the ROCSET resistor from
Equation 3 with:
1. The maximum rDS(ON) at the highest junction
temperature.
2. The minimum IOCSET from the specification table.
3. Determine IPEAK for:
( ΔI )
I PEAK > I OUT ( MAX ) + ---------- ,where ΔI is
2
the output inductor ripple current.
If the ISL6532A IC junction temperature reaches a nominal
temperature of +140°C, all regulators will be disabled. The
ISL6532A will not re-enable the outputs until the junction
temperature drops below +110°C and either the bias voltage
is toggled in order to initiate a POR or the SLP_S5 signal is
forced LOW and then back to HIGH.
Shoot-Through Protection
A shoot-through condition occurs when both the upper and
lower MOSFETs are turned on simultaneously, effectively
shorting the input voltage to ground. To protect from a shootthrough condition, the ISL6532A incorporates specialized
circuitry, which insures that complementary MOSFETs are
not ON simultaneously.
The adaptive shoot-through protection utilized by the VDDQ
regulator looks at the lower gate drive pin, LGATE, and the
upper gate drive pin, UGATE, to determine whether a
MOSFET is ON or OFF. If the voltage from UGATE or from
LGATE to GND is less than 0.8V, then the respective
MOSFET is defined as being OFF and the other MOSFET is
allowed to turned ON. This method allows the VDDQ
regulator to both source and sink current.
Since the voltage of the MOSFET gates are being measured
to determine the state of the MOSFET, the designer is
encouraged to consider the repercussions of introducing
external components between the gate drivers and their
respective MOSFET gates before actually implementing
such measures. Doing so may interfere with the shootthrough protection.
For an equation for the ripple current, see the section under
component guidelines titled “Output Inductor Selection” on
page 14.
11
FN9099.5
May 5, 2008
ISL6532A
Application Guidelines
12VATX
CBP
P12V
GNDP
Layout Considerations
Layout is very important in high frequency switching
converter design. With power devices switching efficiently at
250kHz, the resulting current transitions from one device to
another cause voltage spikes across the interconnecting
impedances and parasitic circuit elements. These voltage
spikes can degrade efficiency, radiate noise into the circuit,
and lead to device overvoltage stress. Careful component
layout and printed circuit board design minimizes these
voltage spikes.
VIN_DDR
ISL6532A
NCH
5VSBY
P5VSBY
5VSBY
GNDP
CIN
CBP
Q1 LOUT
UGATE
VDDQ
As an example, consider the turn-off transition of the control
MOSFET. Prior to turn-off, the MOSFET is carrying the full
load current. During turn-off, current stops flowing in the
MOSFET and is picked up by the lower MOSFET. Any
parasitic inductance in the switched current path generates a
large voltage spike during the switching interval. Careful
component selection, tight layout of the critical components,
and short, wide traces minimizes the magnitude of voltage
spikes.
LGATE
COMP
C1
R2
R1
FB
C3 R3
R4
VDDQ(3)
12
VTT(2)
VTT
COUT2
Q3
DRIVE2
GND PAD
VIN_AGP
R5
VAGP
FB2
R6
COUT3
LOAD
The switching components should be placed close to the
ISL6532A first. Minimize the length of the connections between
the input capacitors, CIN, and the power switches by placing
them nearby. Position both the ceramic and bulk input
capacitors as close to the upper MOSFET drain as possible.
VDDQ
LOAD
In order to dissipate heat generated by the internal VTT
LDO, the ground pad, pin 29, should be connected to the
internal ground plane through at least four vias. This allows
the heat to move away from the IC and also ties the pad to
the ground plane through a low impedance path.
COUT1
Q2
C2
There are two sets of critical components in the ISL6532A
switching converter. The switching components are the most
critical because they switch large amounts of energy, and
therefore tend to generate large amounts of noise. Next are
the small signal components which connect to sensitive
nodes or supply critical bypass current and signal coupling.
A multi-layer printed circuit board is recommended. Figure 4
shows the connections of the critical components in the
converter. Note that capacitors CIN and COUT could each
represent numerous physical capacitors. Dedicate one solid
layer, usually a middle layer of the PC board, for a ground
plane and make all critical component ground connections
with vias to this layer. Dedicate another solid layer as a
power plane and break this plane into smaller islands of
common voltage levels. Keep the metal runs from the
PHASE terminals to the output inductor short. The power
plane should support the input power and output power
nodes. Use copper filled polygons on the top and bottom
circuit layers for the phase nodes. Use the remaining printed
circuit layers for small signal wiring. The wiring traces from
the GATE pins to the MOSFET gates should be kept short
and wide enough to easily handle the 1A of drive current.
LOAD
PHASE
KEY
ISLAND ON POWER PLANE LAYER
ISLAND ON CIRCUIT PLANE LAYER
VIA CONNECTION TO GROUND PLANE
FIGURE 4. PRINTED CIRCUIT BOARD POWER PLANES
AND ISLANDS
Position the output inductor and output capacitors between the
upper and lower MOSFETs and the load.
The critical small signal components include any bypass
capacitors, feedback components, and compensation
components. Place the PWM converter compensation
components close to the FB and COMP pins. The feedback
resistors should be located as close as possible to the FB pin
with vias tied straight to the ground plane as required.
Feedback Compensation - PWM Buck Converter
Figure 5 highlights the voltage-mode control loop for a
synchronous-rectified buck converter. The output voltage
(VOUT) is regulated to the Reference voltage level. The error
amplifier output (VE/A) is compared with the oscillator (OSC)
triangular wave to provide a pulse-width modulated (PWM)
wave with an amplitude of VIN at the PHASE node.
The PWM wave is smoothed by the output filter (LO and CO).
FN9099.5
May 5, 2008
ISL6532A
OSC
PWM
COMPARATOR
6. Check Gain against Error Amplifier’s Open-Loop Gain.
LO
-
ΔVOSC
5. Place 2ND Pole at Half the Switching Frequency.
VIN
DRIVER
DRIVER
+
VDDQ
PHASE
7. Estimate Phase Margin - Repeat if Necessary.
Compensation Break Frequency Equations
CO
ESR
(PARASITIC)
ZFB
VE/A
ZIN
-
+
ERROR
AMP
1
f P1 = --------------------------------------------------------⎛ C 1 x C 2⎞
2π x R 2 x ⎜ ----------------------⎟
⎝ C1 + C2⎠
1
f Z2 = ------------------------------------------------------2π x ( R 1 + R 3 ) x C 3
1
f P2 = -----------------------------------2π x R 3 x C 3
REFERENCE
(EQ. 5)
Figure 6 shows an asymptotic plot of the DC-DC converter’s
gain vs frequency. The actual Modulator Gain has a high
gain peak due to the high Q factor of the output filter and is
not shown in Figure 6. Using the above guidelines should
give a Compensation Gain similar to the curve plotted. The
open loop error amplifier gain bounds the compensation
gain. Check the compensation gain at fP2 with the
capabilities of the error amplifier. The Closed Loop Gain is
constructed on the graph of Figure 6 by adding the
Modulator Gain (in dB) to the Compensation Gain (in dB).
This is equivalent to multiplying the modulator transfer
function to the compensation transfer function and plotting
the gain.
DETAILED COMPENSATION COMPONENTS
ZFB
C1
C2
VDDQ
ZIN
C3
R2
R3
R1
COMP
FB
+
R4
ISL6532A
REFERENCE
R ⎞
⎛
V DDQ = 0.8 × ⎜ 1 + ------1-⎟
R 4⎠
⎝
FIGURE 5. VOLTAGE-MODE BUCK CONVERTER
COMPENSATION DESIGN AND OUTPUT
VOLTAGE SELECTION
The modulator transfer function is the small-signal transfer
function of VOUT/VE/A . This function is dominated by a DC
Gain and the output filter (LO and CO), with a double pole
break frequency at FLC and a zero at FESR . The DC Gain of
the modulator is simply the input voltage (VIN) divided by the
peak-to-peak oscillator voltage ΔVOSC .
The compensation gain uses external impedance networks
ZFB and ZIN to provide a stable, high bandwidth (BW) overall
loop. A stable control loop has a gain crossing with
-20dB/decade slope and a phase margin greater than 45 °.
Include worst case component variations when determining
phase margin.
100
1
F ESR = -------------------------------------------2π x ESR x C O
(EQ. 4)
The compensation network consists of the error amplifier
(internal to the ISL6532A) and the impedance networks ZIN
and ZFB. The goal of the compensation network is to provide
a closed loop transfer function with the highest 0dB crossing
frequency (f0dB) and adequate phase margin. Phase margin
is the difference between the closed loop phase at f0dB and
180°. The following equations relate the compensation
network’s poles, zeros and gain to the components (R1 , R2 ,
R3 , C1 , C2 , and C3) in Figure 5. Use these guidelines for
locating the poles and zeros of the compensation network:
1. Pick Gain (R2/R1) for desired converter bandwidth.
2. Place 1ST Zero Below Filter’s Double Pole (~75% FLC).
fZ1 fZ2
fP1
fP2
80
OPEN LOOP
ERROR AMP GAIN
60
GAIN (dB)
Modulator Break Frequency Equations
1
F LC = ------------------------------------------2π x L O x C O
1
f Z1 = -----------------------------------2π x R 2 x C 2
40
20
20LOG
(R2/R1)
20LOG
(VIN/ΔVOSC)
0
COMPENSATION
GAIN
MODULATOR
GAIN
-20
CLOSED LOOP
GAIN
-40
fLC
-60
10
100
1k
fESR
10k
100k
1M
10M
FREQUENCY (Hz)
FIGURE 6. ASYMPTOTIC BODE PLOT OF CONVERTER GAIN
Feedback Compensation - AGP LDO Controller
Figure 7 shows the AGP LDO power and control stage. This
LDO, which uses a MOSFET as the linear pass element,
requires feedback compensation to insure stability of the
system. The LDO requires compensation because of the
output impedance of the error amplifier.
3. Place 2ND Zero at Filter’s Double Pole.
4. Place 1ST Pole at the ESR Zero.
13
FN9099.5
May 5, 2008
ISL6532A
Component Selection Guidelines
ISL6532A
VDDQ
Output Capacitor Selection - PWM Buck Converter
0.8V
REFERENCE
650Ω
+
-
OUTPUT
IMPEDANCE
DRIVE2
C25
VAGP
R10
FB2
R8
R9
ESR
R ⎞
⎛
V AGP = 0.8 × ⎜ 1 + ------8-⎟
R 9⎠
⎝
COUT
RLOAD
+
FIGURE 7. COMPENSATION AND OUTPUT VOLTAGE
SELECTION OF THE LINEAR
To properly compensate the LDO system, a 100kΩ 1%
resistor and a 680pF X5R ceramic capacitor, represented as
R10 and C25 in Figure 7, are used. This compensation will
insure a stable system with any MOSFET given the following
conditions:
τ = C OUT ⋅ ESR > 10μs
R FB = R 8 = 249Ω
(EQ. 6)
Maximum bandwidth will be realized at full load while
minimum bandwidth will be realized at no load. Bandwidth at
no load will be maximized as τ becomes closer to 10μs.
Output Voltage Selection
The output voltage of the VDDQ PWM converter can be
programmed to any level between VIN and the internal
reference, 0.8V. An external resistor divider is used to scale
the output voltage relative to the reference voltage and feed
it back to the inverting input of the error amplifier, see
Figure 5. However, since the value of R1 affects the values of
the rest of the compensation components, it is advisable to
keep its value less than 5kΩ. Depending on the value chosen
for R1, R4 can be calculated based on the Equation 7:
R1 × 0.8V
R 4 = -----------------------------------V DDQ - 0.8V
(EQ. 7)
If the output voltage desired is 0.8V, simply route VDDQ back
to the FB pin through R1, but do not populate R4.
The output voltage for the internal VTT linear regulator is set
internal to the ISL6532A to track the VDDQ voltage by 50%.
There is no need for external programming resistors.
As with the VDDQ PWM regulator, the AGP linear regulator
output voltage is set by means of an external resistor divider
as shown in Figure 7. For stability concerns described
earlier, the recommended value of the feedback resistor, R8,
is 249Ω. The voltage programming resistor, R9 can be
calculated based on the Equation 8:
R 8 × 0.8V
R 9 = ---------------------------------V AGP - 0.8V
(EQ. 8)
14
An output capacitor is required to filter the inductor current
and supply the load transient current. The filtering
requirements are a function of the switching frequency and
the ripple current. The load transient requirements are a
function of the slew rate (di/dt) and the magnitude of the
transient load current. These requirements are generally met
with a mix of capacitors and careful layout.
DDR memory systems are capable of producing transient
load rates above 1A/ns. High frequency capacitors initially
supply the transient and slow the current load rate seen by
the bulk capacitors. The bulk filter capacitor values are
generally determined by the ESR (Effective Series
Resistance) and voltage rating requirements rather than
actual capacitance requirements.
High frequency decoupling capacitors should be placed as
close to the power pins of the load as physically possible. Be
careful not to add inductance in the circuit board wiring that
could cancel the usefulness of these low inductance
components. Consult with the manufacturer of the load on
specific decoupling requirements.
Use only specialized low-ESR capacitors intended for
switching-regulator applications for the bulk capacitors. The
bulk capacitor’s ESR will determine the output ripple voltage
and the initial voltage drop after a high slew-rate transient.
An aluminum electrolytic capacitor’s ESR value is related to
the case size with lower ESR available in larger case sizes.
However, the Equivalent Series Inductance (ESL) of these
capacitors increases with case size and can reduce the
usefulness of the capacitor to high slew-rate transient
loading. Unfortunately, ESL is not a specified parameter.
Work with your capacitor supplier and measure the
capacitor’s impedance with frequency to select a suitable
component. In most cases, multiple electrolytic capacitors of
small case size perform better than a single large case
capacitor.
Output Capacitor Selection - LDO Regulators
The output capacitors used in LDO regulators are used to
provide dynamic load current. The amount of capacitance
and type of capacitor should be chosen with this criteria in
mind.
Output Inductor Selection
The output inductor is selected to meet the output voltage
ripple requirements and minimize the converter’s response
time to the load transient. The inductor value determines the
converter’s ripple current and the ripple voltage is a function
of the ripple current. The ripple voltage and current are
approximated by the following equations:
ΔI =
VIN - VOUT
Fs x L
x
VOUT
VIN
ΔVOUT = ΔI x ESR
(EQ. 9)
FN9099.5
May 5, 2008
ISL6532A
Increasing the value of inductance reduces the ripple current
and voltage. However, the large inductance values reduce
the converter’s response time to a load transient.
One of the parameters limiting the converter’s response to a
load transient is the time required to change the inductor
current. Given a sufficiently fast control loop design, the
ISL6532A will provide either 0% or 100% duty cycle in
response to a load transient. The response time is the time
required to slew the inductor current from an initial current
value to the transient current level. During this interval the
difference between the inductor current and the transient
current level must be supplied by the output capacitor.
Minimizing the response time can minimize the output
capacitance required.
The response time to a transient is different for the
application of load and the removal of load. The following
equations give the approximate response time interval for
application and removal of a transient load:
tRISE =
L x ITRAN
VIN - VOUT
tFALL =
L x ITRAN
VOUT
(EQ. 10)
where: ITRAN is the transient load current step, tRISE is the
response time to the application of load, and tFALL is the
response time to the removal of load. The worst case
response time can be either at the application or removal of
load. Be sure to check both of these equations at the
minimum and maximum output levels for the worst case
response time.
Input Capacitor Selection - PWM Buck Converter
Use a mix of input bypass capacitors to control the voltage
overshoot across the MOSFETs. Use small ceramic
capacitors for high frequency decoupling and bulk capacitors
to supply the current needed each time the upper MOSFET
turns on. Place the small ceramic capacitors physically close
to the MOSFETs and between the drain of upper MOSFET
and the source of lower MOSFET.
The important parameters for the bulk input capacitance are
the voltage rating and the RMS current rating. For reliable
operation, select bulk capacitors with voltage and current
ratings above the maximum input voltage and largest RMS
current required by the circuit. Their voltage rating should be
at least 1.25 times greater than the maximum input voltage,
while a voltage rating of 1.5 times is a conservative
guideline. For most cases, the RMS current rating
requirement for the input capacitor of a buck regulator is
approximately 1/2 the DC load current.
The maximum RMS current required by the regulator may be
closely approximated through Equation 11:
I RMS
MAX
=
V IN - V OUT V OUT 2
V OUT ⎛
2
1
-------------- × I OUT
+ ------ × ⎛ ----------------------------- × --------------⎞ ⎞
⎝
V IN
V IN ⎠ ⎠
12 ⎝ L × f s
MAX
For a through hole design, several electrolytic capacitors
may be needed. For surface mount designs, solid tantalum
capacitors can be used, but caution must be exercised with
regard to the capacitor surge current rating. These
capacitors must be capable of handling the surge-current at
power-up. Some capacitor series available from reputable
manufacturers are surge current tested.
MOSFET Selection - PWM Buck Converter
The ISL6532A requires 2 N-Channel power MOSFETs for
switching power and a third MOSFET to block backfeed from
VDDQ to the Input in S3 Mode. These should be selected
based upon rDS(ON) , gate supply requirements, and thermal
management requirements.
In high-current applications, the MOSFET power dissipation,
package selection and heatsink are the dominant design
factors. The power dissipation includes two loss
components; conduction loss and switching loss. The
conduction losses are the largest component of power
dissipation for both the upper and the lower MOSFETs.
These losses are distributed between the two MOSFETs
according to duty factor. The switching losses seen when
sourcing current will be different from the switching losses
seen when sinking current. When sourcing current, the
upper MOSFET realizes most of the switching losses. The
lower switch realizes most of the switching losses when the
converter is sinking current (see the following equations).
These equations assume linear voltage-current transitions
and do not adequately model power loss due the reverserecovery of the upper and lower MOSFET’s body diode. The
gate-charge losses are dissipated in part by the ISL6532A
and do not significantly heat the MOSFETs. However, large
gate-charge increases the switching interval, tSW which
increases the MOSFET switching losses. Ensure that both
MOSFETs are within their maximum junction temperature at
high ambient temperature by calculating the temperature
rise according to package thermal-resistance specifications.
A separate heatsink may be necessary depending upon
MOSFET power, package type, ambient temperature and air
flow.
Approximate Losses while Sourcing current
2
1
P UPPER = Io × r DS ( ON ) × D + --- ⋅ Io × V IN × t SW × f s
2
PLOWER = Io2 x rDS(ON) x (1 - D)
Approximate Losses while Sinking current
PUPPER = Io2 x rDS(ON) x D
2
1
P LOWER = Io × r DS ( ON ) × ( 1 - D ) + --- ⋅ Io × V IN × t SW × f s
2
Where: D is the duty cycle = VOUT / VIN ,
tSW is the combined switch ON and OFF time, and
fs is the switching frequency.
(EQ. 12)
(EQ. 11)
15
FN9099.5
May 5, 2008
ISL6532A
MOSFET Selection - AGP LDO
ISL6532A Application Circuit
The main criteria for selection of the linear regulator pass
transistor is package selection for efficient removal of heat.
Select a package and heatsink that maintains the junction
temperature below the rating with a maximum expected
ambient temperature.
Figure 8 shows an application circuit utilizing the ISL6532A.
Detailed information on the circuit, including a complete Billof-Materials and circuit board description, can be found in
Application Note AN1056.
The power dissipated in the linear regulator is:
P LINEAR ≅ I O × ( V IN - V OUT )
(EQ. 13)
where IO is the maximum output current and VOUT is the
nominal output voltage of the linear regulator.
VCC5
5VSBY
VCC12
+3.3V
C17,18
1μF
C26
0.1μF
VREF
S5#
SLP_S3
S3#
5VSBY
PGOOD
SLP_S5
P12V
PGOOD
VDDQ
VTT
1.25V
C20+
220μF
VDDQ
Q4
C22
R7
1000pF
C4,5
1μF
8.87kΩ
C1-3
2200μF
UGATE
2.5V 15AMAX
L2
2.1μH
VTT
VTT
C21
220μF
VDDQ
Q1,3
PHASE
ISL6532A
+
+
VREF_IN
C19
0.47μF
VDDQ
L1
2.1μH
NCH
OCSET
VREF_OUT
C27
0.1μF
Q5
C16
1μF
P5VSBY
R2
10.0kΩ
R1
4.99kΩ
LGATE
Q2,4
VDDQ
VDDQ
VDDQ
VTTSNS
C6-8
1800μF
C9-12
22μF
R4
1.74kΩ
GNDQ
GNDQ
DRIVE2
+
100kΩ
R10
+
C24
1μF
C23
220μF
R8
249Ω
R9
287Ω
GNDA
FB2
GNDP
C25 680pF
1.5V
GNDP
FB
VAGP
COMP
C15
1000pF
C14
6.8nF
R3
19.1kΩ
C13
56nF
R5
22.6Ω
R6
825Ω
FIGURE 8. DDR SDRAM AND AGP VOLTAGE REGULATOR USING THE ISL6532A
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ISL6532A
Quad Flat No-Lead Plastic Package (QFN)
Micro Lead Frame Plastic Package (MLFP)
L28.6x6
28 LEAD QUAD FLAT NO-LEAD PLASTIC PACKAGE
(COMPLIANT TO JEDEC MO-220VJJC ISSUE C)
MILLIMETERS
SYMBOL
MIN
NOMINAL
MAX
NOTES
A
0.80
0.90
1.00
-
A1
-
-
0.05
-
A2
-
-
1.00
A3
b
0.23
D
0.28
9
0.35
5, 8
6.00 BSC
D1
D2
9
0.20 REF
-
5.75 BSC
3.95
4.10
9
4.25
7, 8
E
6.00 BSC
-
E1
5.75 BSC
9
E2
3.95
e
4.10
4.25
7, 8
0.65 BSC
-
k
0.25
-
-
-
L
0.35
0.60
0.75
8
L1
-
-
0.15
10
N
28
2
Nd
7
3
Ne
7
3
P
-
-
0.60
9
θ
-
-
12
9
Rev. 1 10/02
NOTES:
1. Dimensioning and tolerancing conform to ASME Y14.5-1994.
2. N is the number of terminals.
3. Nd and Ne refer to the number of terminals on each D and E.
4. All dimensions are in millimeters. Angles are in degrees.
5. Dimension b applies to the metallized terminal and is measured
between 0.15mm and 0.30mm from the terminal tip.
6. The configuration of the pin #1 identifier is optional, but must be
located within the zone indicated. The pin #1 identifier may be
either a mold or mark feature.
7. Dimensions D2 and E2 are for the exposed pads which provide
improved electrical and thermal performance.
8. Nominal dimensions are provided to assist with PCB Land Pattern
Design efforts, see Intersil Technical Brief TB389.
9. Features and dimensions A2, A3, D1, E1, P & θ are present when
Anvil singulation method is used and not present for saw
singulation.
10. Depending on the method of lead termination at the edge of the
package, a maximum 0.15mm pull back (L1) maybe present. L
minus L1 to be equal to or greater than 0.3mm.
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