LM2595 1.0 A, Step-Down Switching Regulator The LM2595 regulator is monolithic integrated circuit ideally suited for easy and convenient design of a step−down switching regulator (buck converter). It is capable of driving a 1.0 A load with excellent line and load regulation. This device is available in adjustable output version and it is internally compensated to minimize the number of external components to simplify the power supply design. Since LM2595 converter is a switch−mode power supply, its efficiency is significantly higher in comparison with popular three−terminal linear regulators, especially with higher input voltages. The LM2595 operates at a switching frequency of 150 kHz thus allowing smaller sized filter components than what would be needed with lower frequency switching regulators. Available in a standard 5−lead TO−220 package with several different lead bend options, and D2PAK surface mount package. The other features include a guaranteed $4% tolerance on output voltage within specified input voltages and output load conditions, and $15% on the oscillator frequency. External shutdown is included, featuring 50 mA (typical) standby current. Self protection features include switch cycle−by−cycle current limit for the output switch, as well as thermal shutdown for complete protection under fault conditions. http://onsemi.com 5 Heatsink surface connected to Pin 3 TO−220 T SUFFIX CASE 314D 1 5 Pin Features • • • • • • • • • • Adjustable Output Voltage Range 1.23 V − 37 V Guaranteed 1.0 A Output Load Current Wide Input Voltage Range up to 40 V 150 kHz Fixed Frequency Internal Oscillator TTL Shutdown Capability Low Power Standby Mode, typ 50 mA Thermal Shutdown and Current Limit Protection Internal Loop Compensation Moisture Sensitivity Level (MSL) Equals 1 Pb−Free Packages are Available 1 5 February, 2009 − Rev. 2 Output Vin Ground Feedback ON/OFF D2PAK D2T SUFFIX CASE 936A ORDERING INFORMATION Simple High−Efficiency Step−Down (Buck) Regulator Efficient Pre−Regulator for Linear Regulators On−Card Switching Regulators Positive to Negative Converter (Buck−Boost) Negative Step−Up Converters Power Supply for Battery Chargers © Semiconductor Components Industries, LLC, 2009 1. 2. 3. 4. 5. Heatsink surface (shown as terminal 6 in case outline drawing) is connected to Pin 3 Applications • • • • • • TO−220 TV SUFFIX CASE 314B 1 See detailed ordering and shipping information in the package dimensions section on page 23 of this data sheet. DEVICE MARKING INFORMATION See general marking information in the device marking section on page 23 of this data sheet. 1 Publication Order Number: LM2595/D LM2595 12 V Unregulated DC Input R1=1K Feedback +Vin 4 LM2595 L1 68 mH Cff R2=3.0K Output 2 5V@1A Regulated Output 1 Cin 220 mF/ 50 V 3 GND 5 ON/OFF Cout 220 mF D1 1N5822 Figure 1. Typical Application Figure 2. Representative Block Diagram MAXIMUM RATINGS Rating Symbol Value Unit Maximum Supply Voltage Vin 45 V ON/OFF Pin Input Voltage ON/OFF −0.3 V ≤ V ≤ +Vin V Output −1.0 V PD Internally Limited W Thermal Resistance, Junction−to−Ambient RqJA 65 °C/W Thermal Resistance, Junction−to−Case RqJC 5.0 °C/W PD Internally Limited W Thermal Resistance, Junction−to−Ambient RqJA 70 °C/W Thermal Resistance, Junction−to−Case RqJC 5.0 °C/W Tstg −65 to +150 °C Minimum ESD Rating (Human Body Model: C = 100 pF, R = 1.5 kW) − 2.0 kV Lead Temperature (Soldering, 10 seconds) − 260 °C Maximum Junction Temperature TJ 150 °C Output Voltage to Ground (Steady−State) Power Dissipation Case 314B and 314D (TO−220, 5−Lead) Case 936A (D2PAK) Storage Temperature Range Stresses exceeding Maximum Ratings may damage the device. Maximum Ratings are stress ratings only. Functional operation above the Recommended Operating Conditions is not implied. Extended exposure to stresses above the Recommended Operating Conditions may affect device reliability. http://onsemi.com 2 LM2595 PIN FUNCTION DESCRIPTION Pin Symbol Description (Refer to Figure 1) 1 Output This is the emitter of the internal switch. The saturation voltage Vsat of this output switch is typically 1.0 V. It should be kept in mind that the PCB area connected to this pin should be kept to a minimum in order to minimize coupling to sensitive circuitry. 2 Vin This pin is the positive input supply for the LM2595 step−down switching regulator. In order to minimize voltage transients and to supply the switching currents needed by the regulator, a suitable input bypass capacitor must be present (Cin in Figure 1). 3 GND 4 Feedback This pin is the direct input of the error amplifier and the resistor network R2, R1 is connected externally to allow programming of the output voltage. 5 ON/OFF It allows the switching regulator circuit to be shut down using logic level signals, thus dropping the total input supply current to approximately 50 mA. The threshold voltage is typically 1.6 V. Applying a voltage above this value (up to +Vin) shuts the regulator off. If the voltage applied to this pin is lower than 1.6 V or if this pin is left open, the regulator will be in the “on” condition. Circuit ground pin. See the information about the printed circuit board layout. OPERATING RATINGS (Operating Ratings indicate conditions for which the device is intended to be functional, but do not guarantee specific performance limits. For guaranteed specifications and test conditions, see the Electrical Characteristics.) Rating Symbol Value Unit Operating Junction Temperature Range TJ −40 to +125 °C Supply Voltage Vin 4.5 to 40 V http://onsemi.com 3 LM2595 SYSTEM PARAMETERS ELECTRICAL CHARACTERISTICS Specifications with standard type face are for TJ = 25°C, and those with boldface type apply over full Operating Temperature Range −40°C to +125°C Symbol Characteristics Min Typ Max Unit LM2595 (Note 1, Test Circuit Figure 16) Feedback Voltage (Vin = 12 V, ILoad = 0.2 A, Vout = 5.0 V, ) VFB_nom Feedback Voltage (8.0 V ≤ Vin ≤ 40 V, 0.2 A ≤ ILoad ≤ 1.0 A, Vout = 5.0 V) VFB 1.193 1.18 h − Symbol Min Efficiency (Vin = 12 V, ILoad = 1.0 A, Vout = 5.0 V) Characteristics 1.23 Feedback Bias Current (Vout = 5.0 V) Ib 1.267 1.28 V 81 − % Typ Max Unit 25 100 200 nA 150 165 180 kHz 1.2 1.3 V Oscillator Frequency (Note 2) fosc Saturation Voltage (Iout = 1.0 A, Notes 3 and 4) Vsat 1.0 Max Duty Cycle “ON” (Note 4) DC 95 Current Limit (Peak Current, Notes 2 and 3) ICL Output Leakage Current (Notes 5 and 6) Output = 0 V Output = −1.0 V IL Quiescent Current (Note 5) Standby Quiescent Current (ON/OFF Pin = 5.0 V (“OFF”)) (Note 6) 135 120 1.2 1.15 V % 2.1 2.4 2.6 A 0.5 13 2.0 30 IQ 5.0 10 mA Istby 50 200 250 mA mA ON/OFF PIN LOGIC INPUT 1.6 Threshold Voltage Vout = 0 V (Regulator OFF) VIH Vout = Nominal Output Voltage (Regulator ON) VIL V 2.2 2.4 V 1.0 0.8 V ON/OFF Pin Input Current ON/OFF Pin = 5.0 V (Regulator OFF) IIH − 15 30 mA ON/OFF Pin = 0 V (regulator ON) IIL − 0.01 5.0 mA 1. External components such as the catch diode, inductor, input and output capacitors can affect switching regulator system performance. When the LM2595 is used as shown in the Figure 16 test circuit, system performance will be as shown in system parameters section. 2. The oscillator frequency reduces to approximately 30 kHz in the event of an output short or an overload which causes the regulated output voltage to drop approximately 40% from the nominal output voltage. This self protection feature lowers the average dissipation of the IC by lowering the minimum duty cycle from 5% down to approximately 2%. 3. No diode, inductor or capacitor connected to output (Pin 1) sourcing the current. 4. Feedback (Pin 4) removed from output and connected to 0 V. 5. Feedback (Pin 4) removed from output and connected to +12 V to force the output transistor “off”. 6. Vin = 40 V. http://onsemi.com 4 LM2595 TYPICAL PERFORMANCE CHARACTERISTICS (Circuit of Figure 16) 0.8 0.6 Vout, OUTPUT VOLTAGE CHANGE (%) Vout , OUTPUT VOLTAGE CHANGE (%) 1.0 Vin = 20 V ILoad = 200 mA Normalized at TJ = 25°C 0.4 0.2 0 -0.2 -0.4 -0.6 -0.8 -1.0 −50 −25 0 25 50 75 100 125 0.6 Vout = 5 V 0.4 0.2 0 −0.2 −0.4 −0.6 0 5.0 10 15 20 25 30 Figure 4. Line Regulation 35 40 3.0 SWITCHING CURRENT LIMIT (A) ILoad = 1 A 0.5 ILoad = 200 mA 0 L = 68 mH R_ind = 30 mW −25 0 25 60 75 100 2.0 1.0 10 30 50 70 90 110 TJ, JUNCTION TEMPERATURE (°C) TJ, JUNCTION TEMPERATURE (°C) Figure 5. Dropout Voltage Figure 6. Current Limit 160 Vout = 5 V Measured at GND Pin TJ = 25°C 11 10 9 ILoad = 1.0 A 8 7 ILoad = 200 mA 6 5 5 Vin = 12 V 0.0 −50 −30 −10 125 I stby , STANDBY QUIESCENT CURRENT (μA) INPUT - OUTPUT DIFFERENTIAL (V) 0.8 Figure 3. Normalized Output Voltage 12 I Q, QUIESCENT CURRENT (mA) 1.0 Vin, INPUT VOLTAGE (V) 1.0 4 0 ILoad = 200 mA TJ = 25°C 1.2 TJ, JUNCTION TEMPERATURE (°C) 1.5 −0.5 −50 1.4 10 15 20 25 30 35 40 140 VON/OFF = 5.0 V 120 100 80 Vin = 40 V 60 40 Vin = 12 V 20 0 −50 −25 0 25 60 75 100 Vin, INPUT VOLTAGE (V) TJ, JUNCTION TEMPERATURE (°C) Figure 7. Quiescent Current Figure 8. Standby Quiescent Current http://onsemi.com 5 130 125 LM2595 TYPICAL PERFORMANCE CHARACTERISTICS (Circuit of Figure 16) 1.0 1.2 NORMALIZED FREQUENCY (%) Vsat , SATURATION VOLTAGE (V) 1.3 1.1 1.0 0.9 −40°C 0.8 25°C 0.7 125°C 0.6 0.5 0 0.2 0.4 0.6 −3.0 −4.0 −5.0 −6.0 −7.0 0.8 −9.0 −50 1.0 0 25 50 75 100 TJ, JUNCTION TEMPERATURE (°C) Figure 9. Switch Saturation Voltage Figure 10. Switching Frequency 100 4.5 80 Ib , FEEDBACK PIN CURRENT (nA) 5.0 4.0 3.5 3.0 2.5 2.0 Vout ' 1.23 V ILoad = 200 mA 1.5 1.0 0.5 0 -50 −25 SWITCH CURRENT (A) -25 0 25 50 75 100 40 20 0 -20 -40 -60 -80 -100 -50 125 -25 0 25 12 V, 1 A 85 5 V, 1 A 80 75 3.3 V, 1 A 0 5 10 75 100 Figure 12. Feedback Pin Current 95 70 50 TJ, JUNCTION TEMPERATURE (°C) TJ, JUNCTION TEMPERATURE (°C) 90 15 20 25 30 VIN, INPUT VOLTAGE (V) Figure 13. Efficiency http://onsemi.com 6 125 60 Figure 11. Minimum Supply Operating Voltage EFFICIENCY (%) V in, INPUT VOLTAGE (V) −2.0 −8.0 0.4 0.3 0.0 −1.0 35 40 45 125 LM2595 TYPICAL PERFORMANCE CHARACTERISTICS (Circuit of Figure 16) A 10 V 100 mV Output 0 Voltage Change - 100 mV 0 1.2 A B 0.6 A 0 0.5 A 1.2 A C D Load Current 0.6 A 0.1 A 0 0 2 ms/div 100 ms/div Figure 14. Switching Waveforms Figure 15. Load Transient Response Vout = 5 V A: Output Pin Voltage, 10 V/div B: Switch Current, 0.6 A/div C: Inductor Current, 0.6 A/div, AC−Coupled D: Output Ripple Voltage, 50 mV/div, AC−Coupled Horizontal Time Base: 2.0 ms/div Adjustable Output Voltage Versions Feedback Vin LM2595 2 L1 68 mH Output 3 8.5 V - 40 V Unregulated DC Input 4 GND 5 Vout 5.0 V/1.0 A 1 ON/OFF CFF Cin 100 mF D1 1N5822 Cout 220 mF R2 Load R1 V out + V ǒ R2 + R1 ǒ1.0 ) R2 Ǔ R1 ref Ǔ V out 1.0 V ref Where Vref = 1.23 V, R1 between 1.0 k and 5.0 k Figure 16. Typical Test Circuit http://onsemi.com 7 LM2595 PCB LAYOUT GUIDELINES On the other hand, the PCB area connected to the Pin 1 As in any switching regulator, the layout of the printed (emitter of the internal switch) of the LM2595 should be circuit board is very important. Rapidly switching currents kept to a minimum in order to minimize coupling to sensitive associated with wiring inductance, stray capacitance and circuitry. parasitic inductance of the printed circuit board traces can Another sensitive part of the circuit is the feedback. It is generate voltage transients which can generate important to keep the sensitive feedback wiring short. To electromagnetic interferences (EMI) and affect the desired assure this, physically locate the programming resistors near operation. As indicated in the Figure 16, to minimize to the regulator, when using the adjustable version of the inductance and ground loops, the length of the leads LM2595 regulator. indicated by heavy lines should be kept as short as possible. For best results, single−point grounding (as indicated) or ground plane construction should be used. DESIGN PROCEDURE Buck Converter Basics This period ends when the power switch is once again turned on. Regulation of the converter is accomplished by varying the duty cycle of the power switch. It is possible to describe the duty cycle as follows: The LM2595 is a “Buck” or Step−Down Converter which is the most elementary forward−mode converter. Its basic schematic can be seen in Figure 17. The operation of this regulator topology has two distinct time periods. The first one occurs when the series switch is on, the input voltage is connected to the input of the inductor. The output of the inductor is the output voltage, and the rectifier (or catch diode) is reverse biased. During this period, since there is a constant voltage source connected across the inductor, the inductor current begins to linearly ramp upwards, as described by the following equation: I L(on) + t d + on , where T is the period of switching. T For the buck converter with ideal components, the duty cycle can also be described as: V d + out V in Figure 18 shows the buck converter, idealized waveforms of the catch diode voltage and the inductor current. ǒV IN * VOUTǓton Von(SW) L Power Switch Diode Voltage During this “on” period, energy is stored within the core material in the form of magnetic flux. If the inductor is properly designed, there is sufficient energy stored to carry the requirements of the load during the “off” period. L Vin Cout D Power Switch Off VD(FWD) Power Switch On Power Switch On RLoad Time Figure 17. Basic Buck Converter Inductor Current Ipk The next period is the “off” period of the power switch. When the power switch turns off, the voltage across the inductor reverses its polarity and is clamped at one diode voltage drop below ground by the catch diode. The current now flows through the catch diode thus maintaining the load current loop. This removes the stored energy from the inductor. The inductor current during this time is: I L(off) + Power Switch Off ILoad(AV) Imin Diode Power Switch Diode Power Switch Time Figure 18. Buck Converter Idealized Waveforms ǒV OUT * VDǓtoff L http://onsemi.com 8 LM2595 PROCEDURE (ADJUSTABLE OUTPUT VERSION: LM2595) Procedure Example Given Parameters: Vout = Regulated Output Voltage Vin(max) = Maximum DC Input Voltage ILoad(max) = Maximum Load Current Given Parameters: Vout = 5.0 V Vin(max) = 12 V ILoad(max) = 1.0 A 1. Programming Output Voltage To select the right programming resistor R1 and R2 value (see Figure 1) use the following formula: 1. Programming Output Voltage (selecting R1 and R2) Select R1 and R2: ǒ Ǔ R2 1.0 ) V out + V ref R1 ǒ V out + 1.23 1.0 ) ǒ where Vref = 1.23 V R2 + R1 Resistor R1 can be between 1.0 k and 5.0 kW. (For best temperature coefficient and stability with time, use 1% metal film resistors). ǒV R2 + R1 V out ref * 1.0 V out V ref Ǔ R2 R1 Select R1 = 1.0 kW Ǔ ǒ * 1.0 + Ǔ 5V * 1.0 1.23 V R2 = 3.07 kW, choose a 3.0k metal film resistor. Ǔ 2. Input Capacitor Selection (Cin) To prevent large voltage transients from appearing at the input and for stable operation of the converter, an aluminium or tantalum electrolytic bypass capacitor is needed between the input pin +Vin and ground pin GND This capacitor should be located close to the IC using short leads. This capacitor should have a low ESR (Equivalent Series Resistance) value. 2. Input Capacitor Selection (Cin) A 220 mF, 50 V aluminium electrolytic capacitor located near the input and ground pin provides sufficient bypassing. For additional information see input capacitor section in the “Application Information” section of this data sheet. 3. Catch Diode Selection (D1) A. Since the diode maximum peak current exceeds the regulator maximum load current the catch diode current rating must be at least 1.2 times greater than the maximum load current. For a robust design, the diode should have a current rating equal to the maximum current limit of the LM2595 to be able to withstand a continuous output short. B. The reverse voltage rating of the diode should be at least 1.25 times the maximum input voltage. 3. Catch Diode Selection (D1) A. For this example, a 1.0 A (for a robust design 3.0 A diode is recommended) current rating is adequate. B. For Vin = 12 V use a 20 V 1N5817 (1N5820) Schottky diode or any suggested fast recovery diode in the Table 2. http://onsemi.com 9 LM2595 PROCEDURE (ADJUSTABLE OUTPUT VERSION: LM2595) (CONTINUED) Procedure Example 4. Inductor Selection (L1) A. Calculate E x T [V x ms] constant: 4. Inductor Selection (L1) A. Use the following formula to calculate the inductor Volt x microsecond [V x ms] constant: E ǒ T+ V IN *V OUT *V Ǔ SAT V V IN OUT *V )V SAT D )V 1000 D 150 kHz ǒV +I ) T + ǒ12 * 5 * 1.0Ǔ E T + ǒ6Ǔ msǓ B. Match the calculated E x T value with the corresponding number on the vertical axis of the Inductor Value Selection Guide shown in Figure 19. This E x T constant is a measure of the energy handling capability of an inductor and is dependent upon the type of core, the core area, the number of turns, and the duty cycle. C. Next step is to identify the inductance region intersected by the E x T value and the maximum load current value on the horizontal axis shown in Figure 19. D. Select an appropriate inductor from Table 3. The inductor chosen must be rated for a switching frequency of 150 kHz and for a current rating of 1.15 x ILoad. The inductor current rating can also be determined by calculating the inductor peak current: I E 5.5 11.5 5 ) 0.5 1000 12 * 1 ) 0.5 150 kHz 6.7ǒV ǒV msǓ msǓ B. E x T = 19.2 [V x ms] C. ILoad(max) = 1.0 A Inductance Region = L30 D. Proper inductor value = 68 mH Choose the inductor from Table 3. ǒVin * VoutǓ ton p(max) Load(max) 2L where ton is the “on” time of the power switch and V t on + out x 1.0 V f osc in 5. Output Capacitor Selection (Cout) A. Since the LM2595 is a forward−mode switching regulator with voltage mode control, its open loop has 2−pole−1−zero frequency characteristic. The loop stability is determined by the output capacitor (capacitance, ESR) and inductance values. 5. Output Capacitor Selection (Cout) A. In this example, it is recommended to use a Nichicon PM capacitor: 220 mF/25 V For stable operation use recommended values of the output capacitors in Table 1. Low ESR electrolytic capacitors between 180 mF and 1000 mF provide best results. B. The capacitors voltage rating should be at least 1.5 times greater than the output voltage, and often much higher voltage rating is needed to satisfy low ESR requirement 6. Feedforward Capacitor (CFF) It provides additional loop stability mainly for higher input voltages. For Cff selection use Table 1. The compensation capacitor between 0.6 nF and 15 nF is wired in parallel with the output voltage setting resistor R2, The capacitor type can be ceramic, plastic, etc.. 6. Feedforward Capacitor (CFF) In this example, it is recommended to use a feedforward capacitor 4.7 nF. http://onsemi.com 10 LM2595 LM2595 Series Buck Regulator Design Procedures (continued) Table 1. RECOMMENDED VALUES OF THE OUTPUT CAPACITOR AND FEEDFORWARD CAPACITOR (Iload = 1.0 A) Nichicon Pm Capacitors Vin (V) Capacity/ESR/Voltage Range (mF/mW/V) 1000/60/10 1000/60/10 1000/60/10 470/120/10 220/110/25 180/290/25 180/290/25 82/190/35 82/190/35 35 1000/60/10 1000/60/10 1000/60/10 220/110/25 180/140/25 120/200/25 120/200/25 82/190/35 82/190/35 26 1000/60/10 470/120/10 220/110/25 220/110/25 180/140/25 120/200/25 120/200/25 82/190/35 20 1000/60/10 470/120/10 220/110/25 220/110/25 180/140/25 120/200/25 120/200/25 18 1000/60/10 470/120/10 220/110/25 220/110/25 180/140/25 120/200/25 120/200/25 12 470/120/10 470/120/10 220/110/25 220/110/25 180/140/25 10 470/120/10 470/120/10 220/110/25 220/110/25 Vout 2 3 4 6 9 12 15 24 28 Cff (nF) 10 4.7 4.7 4.7 1.5 1.5 1 0.6 0.6 E*T(V*us) 40 MAXIMUM LOAD CURRENT (A) Figure 19. Inductor Value Selection Guides (For Continuous Mode Operation) http://onsemi.com 11 LM2595 Table 2. DIODE SELECTION 1A Diodes Surface Mount VR Schottky 20V SK12 Ultra Fast Recovery 3A Diodes Through Hole Schottky Surface Mount Ultra Fast Recovery Schottky Ultra Fast Recovery 1N5817 Through Hole Schottky Ultra Fast Recovery 1N5820 SR102 SK32 SR302 MBR320 30 V SK13 MBRS130 40 V SK14 MBRS140 50 V or More All of these diodes are rated to at least 50 V MURS120 10BF10 1N5818 SR103 11DQ03 All of these diodes are rated to at least 50 V. MUR120 SK33 All of these diodes are rated to at least 50 V. MURS320 30WF10 1N5821 MBR330 31DQ03 1N5822 1N5819 SK34 10BQ040 SR104 MBRS340 MBR340 10MQ040 11DQ04 30WQ04 31DQ04 MBRS160 SR105 SK35 SR305 10BQ050 MBR150 MBR360 MBR350 10MQ060 11DQ05 30WQ05 31DQ05 http://onsemi.com 12 SR304 All of these diodes are rated to at least 50 V. MUR320 30WF10 LM2595 Table 3. INDUCTOR MANUFACTURERS PART NUMBERS Renco Pulse Engineering Coilcraft Inductance (mH) Current (A) Through Hole Surface Mount Through Hole Surface Mount Through Hole L4 68 0.32 RL−1284−68−43 RL1500−68 PE−53804 PE−53804−S − DO1608−68 L5 47 0.37 RL−1284−47−43 RL1500−47 PE−53805 PE−53805−S − DO1608−473 L6 33 0.44 RL−1284−33−43 RL1500−33 PE−53806 PE−53806−S − DO1608−333 L9 220 0.32 RL−5470−3 RL1500−220 PE−53809 PE−53809−S − DO3308−224 L10 150 0.39 RL−5470−4 RL1500−150 PE−53810 PE−53810−S − DO3308−154 Surface Mount L11 100 0.48 RL−5470−5 RL1500−100 PE−53811 PE−53811−S − DO3308−104 L12 68 0.58 RL−5470−6 RL1500−68 PE−53812 PE−53812−S − DO3308−683 L13 47 0.70 RL−5470−7 RL1500−47 PE−53813 PE−53813−S − DO3308−473 L14 33 0.83 RL−1284−33−43 RL1500−33 PE−53814 PE−53814−S − DO3308−333 L15 22 0.99 RL−1284−22−43 RL1500−22 PE−53815 PE−53815−S − DO3308−223 L16 15 1.24 RL−1284−15−43 RL1500−15 PE−53816 PE−53816−S − DO3308−153 L17 330 0.42 RL−5471−1 RL1500−330 PE−53817 PE−53817−S − DO3316−334 L18 220 0.55 RL−5471−2 RL1500−220 PE−53818 PE−53818−S − DO3316−224 L19 150 0.66 RL−5471−3 RL1500−150 PE−53819 PE−53819−S − DO3316−154 L20 100 0.82 RL−5471−4 RL1500−100 PE−53820 PE−53820−S − DO3316−104 L21 68 0.99 RL−5471−5 RL1500−68 PE−53821 PE−53821−S − DO3316−683 L22 47 1.17 RL−5471−6 − PE−53822 PE−53822−S − DO3316−473 L23 33 1.40 RL−5471−7 − PE−53823 PE−53823−S − DO3316−333 L24 22 1.70 RL−1283−22−43 − PE−53824 PE−53824−S RFB0810−220L DO3316−223 L26 330 0.80 RL−5471−1 − PE−53826 PE−53826−S RFB0810−331L DO3340P−334ML L27 220 1.00 RL−5471−2 − PE−53827 PE−53827−S RFB0810−221L DO3340P−224ML L28 150 1.20 RL−5471−3 − PE−53828 PE−53828−S RFB0810−151L DO3340P−154ML L29 100 1.47 RL−5471−4 − PE−53829 PE−53829−S RFB0810−101L DO3340P−104ML L30 68 1.78 RL−5471−5 − PE−53830 PE−53830−S RFB0810−680L DO3340P−683ML L35 47 2.15 RL−5473−1 − PE−53935 PE−53935−S RFB0810−470L DO3340P−473ML http://onsemi.com 13 LM2595 APPLICATION INFORMATION EXTERNAL COMPONENTS regulator loop stability. The ESR of the output capacitor and the peak−to−peak value of the inductor ripple current are the main factors contributing to the output ripple voltage value. Standard aluminium electrolytics could be adequate for some applications but for quality design, low ESR types are recommended. An aluminium electrolytic capacitor’s ESR value is related to many factors such as the capacitance value, the voltage rating, the physical size and the type of construction. In most cases, the higher voltage electrolytic capacitors have lower ESR value. Often capacitors with much higher voltage ratings may be needed to provide low ESR values that, are required for low output ripple voltage. Input Capacitor (Cin) The Input Capacitor Should Have a Low ESR For stable operation of the switch mode converter a low ESR (Equivalent Series Resistance) aluminium or solid tantalum bypass capacitor is needed between the input pin and the ground pin, to prevent large voltage transients from appearing at the input. It must be located near the regulator and use short leads. With most electrolytic capacitors, the capacitance value decreases and the ESR increases with lower temperatures. For reliable operation in temperatures below −25°C larger values of the input capacitor may be needed. Also paralleling a ceramic or solid tantalum capacitor will increase the regulator stability at cold temperatures. Feedfoward Capacitor (Adjustable Output Voltage Version) This capacitor adds lead compensation to the feedback loop and increases the phase margin for better loop stability. For CFF selection, see the design procedure section. RMS Current Rating of Cin The important parameter of the input capacitor is the RMS current rating. Capacitors that are physically large and have large surface area will typically have higher RMS current ratings. For a given capacitor value, a higher voltage electrolytic capacitor will be physically larger than a lower voltage capacitor, and thus be able to dissipate more heat to the surrounding air, and therefore will have a higher RMS current rating. The consequence of operating an electrolytic capacitor beyond the RMS current rating is a shortened operating life. In order to assure maximum capacitor operating lifetime, the capacitor’s RMS ripple current rating should be: The Output Capacitor Requires an ESR Value That Has an Upper and Lower Limit As mentioned above, a low ESR value is needed for low output ripple voltage, typically 1% to 2% of the output voltage. But if the selected capacitor’s ESR is extremely low (below 0.05 W), there is a possibility of an unstable feedback loop, resulting in oscillation at the output. This situation can occur when a tantalum capacitor, that can have a very low ESR, is used as the only output capacitor. At Low Temperatures, Put in Parallel Aluminium Electrolytic Capacitors with Tantalum Capacitors Irms > 1.2 x d x ILoad Electrolytic capacitors are not recommended for temperatures below −25°C. The ESR rises dramatically at cold temperatures and typically rises 3 times at −25°C and as much as 10 times at −40°C. Solid tantalum capacitors have much better ESR spec at cold temperatures and are recommended for temperatures below −25°C. They can be also used in parallel with aluminium electrolytics. The value of the tantalum capacitor should be about 10% or 20% of the total capacitance. The output capacitor should have at least 50% higher RMS ripple current rating at 150 kHz than the peak−to−peak inductor ripple current. where d is the duty cycle, for a buck regulator V t d + on + out T V in |V out| t on and d + + for a buck*boost regulator. T |V out| ) V in Output Capacitor (Cout) For low output ripple voltage and good stability, low ESR output capacitors are recommended. An output capacitor has two main functions: it filters the output and provides http://onsemi.com 14 LM2595 Catch Diode Locate the Catch Diode Close to the LM2595 The LM2595 is a step−down buck converter; it requires a fast diode to provide a return path for the inductor current when the switch turns off. This diode must be located close to the LM2595 using short leads and short printed circuit traces to avoid EMI problems. Use a Schottky or a Soft Switching Ultra−Fast Recovery Diode Since the rectifier diodes are very significant sources of losses within switching power supplies, choosing the rectifier that best fits into the converter design is an important process. Schottky diodes provide the best performance because of their fast switching speed and low forward voltage drop. They provide the best efficiency especially in low output voltage applications (5.0 V and lower). Another choice could be Fast−Recovery, or Ultra−Fast Recovery diodes. It has to be noted, that some types of these diodes with an abrupt turnoff characteristic may cause instability or EMI troubles. A fast−recovery diode with soft recovery characteristics can better fulfill some quality, low noise design requirements. Table 2 provides a list of suitable diodes for the LM2595 regulator. Standard 50/60 Hz rectifier diodes, such as the 1N4001 series or 1N5400 series are NOT suitable. VERTRICAL RESOLUTION 0.4 A/DIV ripple voltage. On the other hand it does require larger inductor values to keep the inductor current flowing continuously, especially at low output load currents and/or high input voltages. To simplify the inductor selection process, an inductor selection guide for the LM2595 regulator was added to this data sheet (Figure 19). This guide assumes that the regulator is operating in the continuous mode, and selects an inductor that will allow a peak−to−peak inductor ripple current to be a certain percentage of the maximum design load current. This percentage is allowed to change as different design load currents are selected. For light loads (less than approximately 300 mA) it may be desirable to operate the regulator in the discontinuous mode, because the inductor value and size can be kept relatively low. Consequently, the percentage of inductor peak−to−peak current increases. This discontinuous mode of operation is perfectly acceptable for this type of switching converter. Any buck regulator will be forced to enter discontinuous mode if the load current is light enough. 0.4 A Inductor Current Waveform 0 A Inductor 0.8 A Power Switch Current Waveform 0 A The magnetic components are the cornerstone of all switching power supply designs. The style of the core and the winding technique used in the magnetic component’s design has a great influence on the reliability of the overall power supply. Using an improper or poorly designed inductor can cause high voltage spikes generated by the rate of transitions in current within the switching power supply, and the possibility of core saturation can arise during an abnormal operational mode. Voltage spikes can cause the semiconductors to enter avalanche breakdown and the part can instantly fail if enough energy is applied. It can also cause significant RFI (Radio Frequency Interference) and EMI (Electro−Magnetic Interference) problems. HORIZONTAL TIME BASE: 2.0 ms/DIV Figure 20. Continuous Mode Switching Current Waveforms Selecting the Right Inductor Style Some important considerations when selecting a core type are core material, cost, the output power of the power supply, the physical volume the inductor must fit within, and the amount of EMI (Electro−Magnetic Interference) shielding that the core must provide. The inductor selection guide covers different styles of inductors, such as pot core, E−core, toroid and bobbin core, as well as different core materials such as ferrites and powdered iron from different manufacturers. For high quality design regulators the toroid core seems to be the best choice. Since the magnetic flux is contained within the core, it generates less EMI, reducing noise problems in sensitive circuits. The least expensive is the bobbin core type, which consists of wire wound on a ferrite rod core. This type of inductor generates more EMI due to the fact that its core is open, and the magnetic flux is not contained within the core. When multiple switching regulators are located on the same printed circuit board, open core magnetics can cause Continuous and Discontinuous Mode of Operation The LM2595 step−down converter can operate in both the continuous and the discontinuous modes of operation. The regulator works in the continuous mode when loads are relatively heavy, the current flows through the inductor continuously and never falls to zero. Under light load conditions, the circuit will be forced to the discontinuous mode when inductor current falls to zero for certain period of time (see Figure 20 and Figure 21). Each mode has distinctively different operating characteristics, which can affect the regulator performance and requirements. In many cases the preferred mode of operation is the continuous mode. It offers greater output power, lower peak currents in the switch, inductor and diode, and can have a lower output http://onsemi.com 15 LM2595 interference between two or more of the regulator circuits, especially at high currents due to mutual coupling. A toroid, pot core or E−core (closed magnetic structure) should be used in such applications. inductor and/or the LM2595. Different inductor types have different saturation characteristics, and this should be kept in mind when selecting an inductor. Exceeding an inductor’s maximum current rating may cause the inductor to overheat because of the copper wire losses, or the core may saturate. Core saturation occurs when the flux density is too high and consequently the cross sectional area of the core can no longer support additional lines of magnetic flux. This causes the permeability of the core to drop, the inductance value decreases rapidly and the inductor begins to look mainly resistive. It has only the DC resistance of the winding. This can cause the switch current to rise very rapidly and force the LM2595 internal switch into cycle−by−cycle current limit, thus reducing the DC output load current. This can also result in overheating of the VERTICAL RESOLUTION 25 mA/DIV Do Not Operate an Inductor Beyond its Maximum Rated Current 0.05 A Inductor Current Waveform 0A 0.05 A Power Switch Current Waveform 0A HORIZONTAL TIME BASE: 2.0 ms/DIV Figure 21. Discontinuous Mode Switching Current Waveforms GENERAL RECOMMENDATIONS Output Voltage Ripple and Transients Source of the Output Ripple Minimizing the Output Ripple In order to minimize the output ripple voltage it is possible to enlarge the inductance value of the inductor L1 and/or to use a larger value output capacitor. There is also another way to smooth the output by means of an additional LC filter (3 mH, 100 mF), that can be added to the output (see Figure 31) to further reduce the amount of output ripple and transients. With such a filter it is possible to reduce the output ripple voltage transients 10 times or more. Figure 22 shows the difference between filtered and unfiltered output waveforms of the regulator shown in Figure 31. The lower waveform is from the normal unfiltered output of the converter, while the upper waveform shows the output ripple voltage filtered by an additional LC filter. Since the LM2595 is a switch mode power supply regulator, its output voltage, if left unfiltered, will contain a sawtooth ripple voltage at the switching frequency. The output ripple voltage value ranges from 0.5% to 3% of the output voltage. It is caused mainly by the inductor sawtooth ripple current multiplied by the ESR of the output capacitor. Short Voltage Spikes and How to Reduce Them The regulator output voltage may also contain short voltage spikes at the peaks of the sawtooth waveform (see Figure 22). These voltage spikes are present because of the fast switching action of the output switch, and the parasitic inductance of the output filter capacitor. There are some other important factors such as wiring inductance, stray capacitance, as well as the scope probe used to evaluate these transients, all these contribute to the amplitude of these spikes. To minimize these voltage spikes, low inductance capacitors should be used, and their lead lengths must be kept short. The importance of quality printed circuit board layout design should also be highlighted. The Surface Mount Package D2PAK and its Heatsinking The other type of package, the surface mount D2PAK, is designed to be soldered to the copper on the PC board. The copper and the board are the heatsink for this package and the other heat producing components, such as the catch diode and inductor. The PC board copper area that the package is soldered to should be at least 0.4 in2 (or 100 mm2) and ideally should have 2 or more square inches (1300 mm2) of 0.0028 inch copper. Additional increasing of copper area beyond approximately 3.0 in2 (2000 mm2) will not improve heat dissipation significantly. If further thermal improvements are needed, double sided or multilayer PC boards with large copper areas should be considered. Voltage spikes caused by switching action of the output switch and the parasitic inductance of the output capacitor VERTRICAL RESOLUTION 20 mV/DIV Filtered Output Voltage Unfiltered Output Voltage Thermal Analysis and Design The following procedure must be performed to determine the operating junction temperature. First determine: 1. PD(max) maximum regulator power dissipation in the application. 2. TA(max) maximum ambient temperature in the application. HORIZONTAL TIME BASE: 5.0 ms/DIV Figure 22. Output Ripple Voltage Waveforms http://onsemi.com 16 LM2595 Packages Not on a Heatsink (Free−Standing) 3. TJ(max) maximum allowed junction temperature (125°C for the LM2595). For a conservative design, the maximum junction temperature should not exceed 110°C to assure safe operation. For every additional +10°C temperature rise that the junction must withstand, the estimated operating lifetime of the component is halved. 4. RqJC package thermal resistance junction−case. package thermal resistance junction−ambient. 5. RqJA (Refer to Maximum Ratings on page 2 of this data sheet or RqJC and RqJA values). For a free−standing application when no heatsink is used, the junction temperature can be determined by the following expression: TJ = (RqJA) (PD) + TA Where (RqJA) (PD) represents the junction temperature rise caused by the dissipated power and TA is the maximum ambient temperature. Packages on a Heatsink If the actual operating junction temperature is greater than the selected safe operating junction temperature determined in step 3, than a heatsink is required. The junction temperature will be calculated as follows: The following formula is to calculate the approximate total power dissipated by the LM2595: TJ = PD (RqJA + RqCS + RqSA) + TA PD = (Vin x IQ) + d x ILoad x Vsat Where RqJC is the thermal resistance junction−case, RqCS is the thermal resistance case−heatsink, RqSA is the thermal resistance heatsink−ambient. If the actual operating temperature is greater than the selected safe operating junction temperature, then a larger heatsink is required. where d is the duty cycle and for buck converter V t d + on + O , V in T IQ (quiescent current) and Vsat can be found in the LM2595 data sheet, Vin is minimum input voltage applied, VO is the regulator output voltage, ILoad is the load current. The dynamic switching losses during turn−on and turn−off can be neglected if proper type catch diode is used. The junction temperature can be determined by the following expression: Some Aspects That can Influence Thermal Design It should be noted that the package thermal resistance and the junction temperature rise numbers are all approximate, and there are many factors that will affect these numbers, such as PC board size, shape, thickness, physical position, location, board temperature, as well as whether the surrounding air is moving or still. Other factors are trace width, total printed circuit copper area, copper thickness, single− or double−sided, multilayer board, the amount of solder on the board or even color of the traces. The size, quantity and spacing of other components on the board can also influence its effectiveness to dissipate the heat. TJ = (RqJA) (PD) + TA where (RqJA)(PD) represents the junction temperature rise caused by the dissipated power and TA is the maximum ambient temperature. 12 to 25 V Unregulated DC Input Cin 100 mF/50 V R4 Feedback +Vin L1 100 mH LM2595 CFF ON/OFF GND D1 1N5819 R3 Cout 220 mF Figure 23. Inverting Buck−Boost Develops −12 V http://onsemi.com 17 −12 V @ 0.7 A Regulated Output LM2595 ADDITIONAL APPLICATIONS Using a delayed startup arrangement, the input capacitor can charge up to a higher voltage before the switch−mode regulator begins to operate. The high input current needed for startup is now partially supplied by the input capacitor Cin. It has been already mentioned above, that in some situations, the delayed startup or the undervoltage lockout features could be very useful. A delayed startup circuit applied to a buck−boost converter is shown in Figure 28. Figure 30 in the “Undervoltage Lockout” section describes an undervoltage lockout feature for the same converter topology. Inverting Regulator An inverting buck−boost regulator using the LM2595 is shown in Figure 23. This circuit converts a positive input voltage to a negative output voltage with a common ground by bootstrapping the regulators ground to the negative output voltage. By grounding the feedback pin, the regulator senses the inverted output voltage and regulates it. In this example the LM2595 is used to generate a −12 V output. The maximum input voltage in this case cannot exceed +28 V because the maximum voltage appearing across the regulator is the absolute sum of the input and output voltages and this must be limited to a maximum of 40 V. This circuit configuration is able to deliver approximately 0.25 A to the output when the input voltage is 12 V or higher. At lighter loads the minimum input voltage required drops to approximately 4.7 V, because the buck−boost regulator topology can produce an output voltage that, in its absolute value, is either greater or less than the input voltage. Since the switch currents in this buck−boost configuration are higher than in the standard buck converter topology, the available output current is lower. This type of buck−boost inverting regulator can also require a larger amount of startup input current, even for light loads. This may overload an input power source with a current limit less than 1.0 A. Such an amount of input startup current is needed for at least 2.0 ms or more. The actual time depends on the output voltage and size of the output capacitor. Because of the relatively high startup currents required by this inverting regulator topology, the use of a delayed startup or an undervoltage lockout circuit is recommended. The inverting regulator operates in a different manner than the buck converter and so a different design procedure has to be used to select the inductor L1 or the output capacitor Cout. The output capacitor values must be larger than what is normally required for buck converter designs. Low input voltages or high output currents require a large value output capacitor (in the range of thousands of mF). The recommended range of inductor values for the inverting converter design is between 68 mH and 220 mH. To select an inductor with an appropriate current rating, the inductor peak current has to be calculated. The following formula is used to obtain the peak inductor current: I (V ) |V |) O ) V in x t on [ Load in 2L 1 V in |V | O where t on + x 1.0 , and f osc + 52 kHz. V ) |V | f osc in O I peak Under normal continuous inductor current operating conditions, the worst case occurs when Vin is minimal. R4 Feedback 12 to 40 V Unregulated DC Input Cin 100 mF/50 V Design Recommendations: +Vin C1 0.1 mF L1 100 mH LM2595 CFF ON/OFF GND D1 1N5819 R2 47k R3 Cout 220 mF −12 V @ 0.25 A Regulated Output Figure 24. Inverting Buck−Boost Develops with Delayed Startup http://onsemi.com 18 LM2595 +V +Vin +Vin Shutdown Input 5.0 V 0 Cin R1 100 mF 47 k 0 LM2595 7 Off On R2 5.6 k 5 ON/OFF 6 +Vin GN D +Vin 7 LM2595 Cin 100 mF On Off Shutdown Input R3 470 Q1 2N3906 R2 47 k 5 ON/OFF 6 -Vout R1 12 k MOC8101 NOTE: This picture does not show the complete circuit. GN D -Vout NOTE: This picture does not show the complete circuit. Figure 25. Inverting Buck−Boost Regulator Shutdown Circuit Using an Optocoupler Figure 26. Inverting Buck−Boost Regulator Shutdown Circuit Using a PNP Transistor With the inverting configuration, the use of the ON/OFF pin requires some level shifting techniques. This is caused by the fact, that the ground pin of the converter IC is no longer at ground. Now, the ON/OFF pin threshold voltage (1.3 V approximately) has to be related to the negative output voltage level. There are many different possible shut down methods, two of them are shown in Figures 25 and 26. Negative Boost Regulator This example is a variation of the buck−boost topology and it is called negative boost regulator. This regulator experiences relatively high switch current, especially at low input voltages. The internal switch current limiting results in lower output load current capability. The circuit in Figure 27 shows the negative boost configuration. The input voltage in this application ranges from −5.0 V to −12 V and provides a regulated −12 V output. If the input voltage is greater than −12 V, the output will rise above −12 V accordingly, but will not damage the regulator. R4 Cout 470 mF Feedback +Vin Cin 100 mF/ 50 V −12 V Unregulated DC Input LM2595 ON/OFF GND D1 1N5822 R3 −12 V @ 0.25 A Regulated Output L1 100 mH Figure 27. Negative Boost Regulator Design Recommendations: values for the negative boost regulator is the same as for inverting converter design. Another important point is that these negative boost converters cannot provide current limiting load protection in the event of a short in the output so some other means, such as a fuse, may be necessary to provide the load protection. The same design rules as for the previous inverting buck−boost converter can be applied. The output capacitor Cout must be chosen larger than would be required for a what standard buck converter. Low input voltages or high output currents require a large value output capacitor (in the range of thousands of mF). The recommended range of inductor http://onsemi.com 19 LM2595 Delayed Startup There are some applications, like the inverting regulator already mentioned above, which require a higher amount of startup current. In such cases, if the input power source is limited, this delayed startup feature becomes very useful. To provide a time delay between the time when the input voltage is applied and the time when the output voltage comes up, the circuit in Figure 28 can be used. As the input voltage is applied, the capacitor C1 charges up, and the voltage across the resistor R2 falls down. When the voltage on the ON/OFF pin falls below the threshold value 1.3 V, the regulator starts up. Resistor R1 is included to limit the maximum voltage applied to the ON/OFF pin. It reduces the power supply noise sensitivity, and also limits the capacitor C1 discharge current, but its use is not mandatory. When a high 50 Hz or 60 Hz (100 Hz or 120 Hz respectively) ripple voltage exists, a long delay time can cause some problems by coupling the ripple into the ON/OFF pin, the regulator could be switched periodically on and off with the line (or double) frequency. +Vin R1 47 k 5 ON/OFF 6 R2 10 k R1 10 k ǒ Vth ≈ 13 V Figure 29. Undervoltage Lockout Circuit for Buck Converter +Vin LM2595 2 R2 15 k GN D Cin 100 mF 5 R3 47 k Z1 1N5242B R2 47 k Q1 2N3904 ON/OFF 3 GND Vth ≈ 13 V R1 15 k Vout NOTE: This picture does not show the complete circuit. Figure 30. Undervoltage Lockout Circuit for Buck−Boost Converter Some applications require the regulator to remain off until the input voltage reaches a certain threshold level. Figure 29 shows an undervoltage lockout circuit applied to a buck regulator. A version of this circuit for buck−boost converter is shown in Figure 30. Resistor R3 pulls the ON/OFF pin high and keeps the regulator off until the input voltage reaches a predetermined threshold level with respect to the ground Pin 3, which is determined by the following expression: Z1 GND NOTE: This picture does not show the complete circuit. Undervoltage Lockout [V ON/OFF 3 Q1 2N3904 Figure 28. Delayed Startup Circuitry th Cin 100 mF 5 R3 47 k Z1 1N5242B NOTE: This picture does not show the complete circuit. V LM2595 2 LM2595 7 Cin 100 mF +Vin +Vin +Vin C1 0.1 mF +Vin Adjustable Output, Low−Ripple Power Supply A 1.0 A output current capability power supply that features an adjustable output voltage is shown in Figure 31. This regulator delivers 1.0 A into 1.2 V to 35 V output. The input voltage ranges from roughly 3.0 V to 40 V. In order to achieve a 10 or more times reduction of output ripple, an additional L−C filter is included in this circuit. Ǔ (Q1) ) 1.0 ) R2 V R1 BE http://onsemi.com 20 LM2595 40 V Max Unregulated DC Input Feedback 4 +Vin LM2595 2 Cin 100 mF Output 3 GND 5 L1 100 mH L2 3 mH 1 ON/OFF CFF 2 to 35 V @ 1.0 A R2 50 k Cout 220 mF D1 1N5822 R1 1.21 k C1 100 mF Optional Output Ripple Filter Figure 31. 2 to 35 V Adjustable 1.0 A Power Supply with Low Output Ripple http://onsemi.com 21 Output Voltage LM2595 THE LM2595 STEP−DOWN VOLTAGE REGULATOR WITH 5.0 V @ 1.0 A OUTPUT POWER CAPABILITY. TYPICAL APPLICATION WITH THROUGH−HOLE PC BOARD LAYOUT 4 Unregulated DC Input +Vin +Vin = 10 V to 40 V Feedback L1 68 mH LM2595 2 Output 3 GND 5 Regulated Output Filtered 1 ON/OFF C1 100 mF /50 V R2 3.0 k D1 1N5819 ON/OFF C2 470 mF /25 V − − − − − − − 100 mF, 50 V, Aluminium Electrolytic 470 mF, 25 V, Aluminium Electrolytic 1.0 A, 40 V, Schottky Rectifier, 1N5819 100 mH, DO3340P, Coilcraft 1.0 kW, 0.25 W 3.0 kW, 0.25 W See Table 1 Vout2 = 5.0 V @ 1.0 A R1 1.0 k V C1 C2 D1 L1 R1 R2 Cff CFF ǒ Ǔ R2 out + V ref ) 1.0 ) R1 Vref = 1.23 V R1 is between 1.0 k and 5.0 k Figure 32. Schematic Diagram of the 5.0 V @ 1.0 A Step−Down Converter Using the LM2595−ADJ NOTE: Not to scale. NOTE: Not to scale. Figure 33. Printed Circuit Board Layout With Component Figure 34. Printed Circuit Board Layout Copper Side References • • • • National Semiconductor LM2595 Data Sheet and Application Note National Semiconductor LM2595 Data Sheet and Application Note Marty Brown “Practical Switching Power Supply Design”, Academic Press, Inc., San Diego 1990 Ray Ridley “High Frequency Magnetics Design”, Ridley Engineering, Inc. 1995 http://onsemi.com 22 LM2595 ORDERING INFORMATION Package Shipping† TO−220 (Pb−Free) 50 Units / Rail LM2595TVADJG TO−220 (F) (Pb−Free) 50 Units / Rail LM2595DSADJG D2PAK (Pb−Free) 50 Units / Rail LM2595DSADJR4G D2PAK (Pb−Free) 800 / Tape & Reel Device LM2595TADJG †For information on tape and reel specifications, including part orientation and tape sizes, please refer to our Tape and Reel Packaging Specifications Brochure, BRD8011/D. MARKING DIAGRAMS TO−220 TV SUFFIX CASE 314B TO−220 T SUFFIX CASE 314D LM 2595T−ADJ AWLYWWG LM 2595T−ADJ AWLYWWG D2PAK DS SUFFIX CASE 936A LM 2595−ADJ AWLYWWG 1 1 5 1 A WL Y WW G 5 = Assembly Location = Wafer Lot = Year = Work Week = Pb−Free Package http://onsemi.com 23 5 LM2595 PACKAGE DIMENSIONS TO−220 TV SUFFIX CASE 314B−05 ISSUE L C B −P− Q OPTIONAL CHAMFER E A U K NOTES: 1. DIMENSIONING AND TOLERANCING PER ANSI Y14.5M, 1982. 2. CONTROLLING DIMENSION: INCH. 3. DIMENSION D DOES NOT INCLUDE INTERCONNECT BAR (DAMBAR) PROTRUSION. DIMENSION D INCLUDING PROTRUSION SHALL NOT EXCEED 0.043 (1.092) MAXIMUM. L S W F 5X 0.24 (0.610) D 0.10 (0.254) M T P DIM A B C D E F G H J K L N Q S U V W J 5X G V M H T N M −T− SEATING PLANE INCHES MIN MAX 0.572 0.613 0.390 0.415 0.170 0.180 0.025 0.038 0.048 0.055 0.850 0.935 0.067 BSC 0.166 BSC 0.015 0.025 0.900 1.100 0.320 0.365 0.320 BSC 0.140 0.153 --0.620 0.468 0.505 --0.735 0.090 0.110 MILLIMETERS MIN MAX 14.529 15.570 9.906 10.541 4.318 4.572 0.635 0.965 1.219 1.397 21.590 23.749 1.702 BSC 4.216 BSC 0.381 0.635 22.860 27.940 8.128 9.271 8.128 BSC 3.556 3.886 --- 15.748 11.888 12.827 --- 18.669 2.286 2.794 TO−220 T SUFFIX CASE 314D−04 ISSUE F −T− B −Q− B1 DETAIL A-A A U C L J H G 5 PL 0.356 (0.014) M T Q NOTES: 1. DIMENSIONING AND TOLERANCING PER ANSI Y14.5M, 1982. 2. CONTROLLING DIMENSION: INCH. 3. DIMENSION D DOES NOT INCLUDE INTERCONNECT BAR (DAMBAR) PROTRUSION. DIMENSION D INCLUDING PROTRUSION SHALL NOT EXCEED 10.92 (0.043) MAXIMUM. DIM A B B1 C D E G H J K L Q U 1234 5 K D E SEATING PLANE M B B1 DETAIL A−A http://onsemi.com 24 INCHES MIN MAX 0.572 0.613 0.390 0.415 0.375 0.415 0.170 0.180 0.025 0.038 0.048 0.055 0.067 BSC 0.087 0.112 0.015 0.025 0.977 1.045 0.320 0.365 0.140 0.153 0.105 0.117 MILLIMETERS MIN MAX 14.529 15.570 9.906 10.541 9.525 10.541 4.318 4.572 0.635 0.965 1.219 1.397 1.702 BSC 2.210 2.845 0.381 0.635 24.810 26.543 8.128 9.271 3.556 3.886 2.667 2.972 LM2595 PACKAGE DIMENSIONS D2PAK D2T SUFFIX CASE 936A−02 ISSUE C −T− OPTIONAL CHAMFER A E U S K B NOTES: 1. DIMENSIONING AND TOLERANCING PER ANSI Y14.5M, 1982. 2. CONTROLLING DIMENSION: INCH. 3. TAB CONTOUR OPTIONAL WITHIN DIMENSIONS A AND K. 4. DIMENSIONS U AND V ESTABLISH A MINIMUM MOUNTING SURFACE FOR TERMINAL 6. 5. DIMENSIONS A AND B DO NOT INCLUDE MOLD FLASH OR GATE PROTRUSIONS. MOLD FLASH AND GATE PROTRUSIONS NOT TO EXCEED 0.025 (0.635) MAXIMUM. TERMINAL 6 V H 1 2 3 4 5 M D 0.010 (0.254) M T L G INCHES MIN MAX 0.386 0.403 0.356 0.368 0.170 0.180 0.026 0.036 0.045 0.055 0.067 BSC 0.539 0.579 0.050 REF 0.000 0.010 0.088 0.102 0.018 0.026 0.058 0.078 5 _ REF 0.116 REF 0.200 MIN 0.250 MIN DIM A B C D E G H K L M N P R S U V P N R C SOLDERING FOOTPRINT* 8.38 0.33 MILLIMETERS MIN MAX 9.804 10.236 9.042 9.347 4.318 4.572 0.660 0.914 1.143 1.397 1.702 BSC 13.691 14.707 1.270 REF 0.000 0.254 2.235 2.591 0.457 0.660 1.473 1.981 5 _ REF 2.946 REF 5.080 MIN 6.350 MIN 1.702 0.067 10.66 0.42 16.02 0.63 3.05 0.12 SCALE 3:1 1.016 0.04 mm Ǔ ǒinches *For additional information on our Pb−Free strategy and soldering details, please download the ON Semiconductor Soldering and Mounting Techniques Reference Manual, SOLDERRM/D. ON Semiconductor and are registered trademarks of Semiconductor Components Industries, LLC (SCILLC). SCILLC reserves the right to make changes without further notice to any products herein. SCILLC makes no warranty, representation or guarantee regarding the suitability of its products for any particular purpose, nor does SCILLC assume any liability arising out of the application or use of any product or circuit, and specifically disclaims any and all liability, including without limitation special, consequential or incidental damages. “Typical” parameters which may be provided in SCILLC data sheets and/or specifications can and do vary in different applications and actual performance may vary over time. All operating parameters, including “Typicals” must be validated for each customer application by customer’s technical experts. SCILLC does not convey any license under its patent rights nor the rights of others. SCILLC products are not designed, intended, or authorized for use as components in systems intended for surgical implant into the body, or other applications intended to support or sustain life, or for any other application in which the failure of the SCILLC product could create a situation where personal injury or death may occur. Should Buyer purchase or use SCILLC products for any such unintended or unauthorized application, Buyer shall indemnify and hold SCILLC and its officers, employees, subsidiaries, affiliates, and distributors harmless against all claims, costs, damages, and expenses, and reasonable attorney fees arising out of, directly or indirectly, any claim of personal injury or death associated with such unintended or unauthorized use, even if such claim alleges that SCILLC was negligent regarding the design or manufacture of the part. SCILLC is an Equal Opportunity/Affirmative Action Employer. This literature is subject to all applicable copyright laws and is not for resale in any manner. PUBLICATION ORDERING INFORMATION LITERATURE FULFILLMENT: Literature Distribution Center for ON Semiconductor P.O. Box 5163, Denver, Colorado 80217 USA Phone: 303−675−2175 or 800−344−3860 Toll Free USA/Canada Fax: 303−675−2176 or 800−344−3867 Toll Free USA/Canada Email: [email protected] N. American Technical Support: 800−282−9855 Toll Free USA/Canada Europe, Middle East and Africa Technical Support: Phone: 421 33 790 2910 Japan Customer Focus Center Phone: 81−3−5773−3850 http://onsemi.com 25 ON Semiconductor Website: www.onsemi.com Order Literature: http://www.onsemi.com/orderlit For additional information, please contact your local Sales Representative LM2595/D