AND8246/D A 160 W CRT TV Power Supply using NCP1337 Prepared by: Nicolas Cyr ON Semiconductor http://onsemi.com APPLICATION NOTE Introduction A 160 W TV Power Supply Design Valley switching converters, also known as quasi−resonant (QR) converters, allow designing flyback Switch−Mode Power Supplies (SMPS) with reduced Electro−Magnetic Interference (EMI) signature and improved efficiency. Thanks to the low level of generated noise, valley switching SMPS converters are therefore very well suited to applications dealing with RF and video signals, such as TVs. ON Semiconductor NCP1337 is a powerful valley switching controller, which eases the design of an EMI−friendly TV power supply with only a few surrounding components. Moreover, very low standby power (less than 1 W) can be achieved without any noise. Power Supply Specification • • • • • • • 160 W +135 V, 1 A max (135 W) regulated +20 V, 800 mA max (16 W) +12 V, 500 mA max (6 W) +8 V, 500 mA max (4 W) Standby output : +5 V, 100 mA derived from +8 V through a regulator Protections Standby Power • Automatic Valley Switching • Current−Mode • Soft Ripple Mode with Minimum Switching Frequency • • • Universal input 90 Vac to 265 Vac Outputs Main Features of the Controller • Input Voltage Output Power Short−circuit, over−power, over−voltage and brown−out below 1 W for Noise−Free Standby Auto−Recovery Short−Circuit Protection Independent of Auxiliary Voltage Over Voltage Protection Brown−Out Protection 2 Externally Triggerable Fault Comparators (Auto−Recovery or Permanent Latch) Internal 5 ms Soft−Start 500 mA Peak Current Source/Sink Capability 130 kHz Max Frequency Internal Leading Edge Blanking Internal Temperature Shutdown Direct Optocoupler Connection Dynamic Self−Supply © Semiconductor Components Industries, LLC, 2005 November, 2005 − Rev. 1 1 Publication Order Number: AND8246/D AND8246/D Schematic C19 R17 D13 135V 12 C20 D5 R2 R1 C8 R10 D141 T1 C4 6 D111 R3 Rbo1 Rhyst C2 Rbo2 C7 L1 8 D6 20V Out20V R35 R11 12V C141 Out12V 17 4 D11 D7 C5 R34 Out20V C131 Rbo M1 2 R31 D1 D14 18 C9 0V R33 11 DZ3 C26 Out12V 16 IC3 8V C13 1 8 2 7 C14 Out8V Out8V 15 D12 IC4 D16 Reg 5V OUT IN 3 6 4 5 14 C11 C10 C15 C3 IC1 5Vstby ADJUST C18 C16 C17 DZ2 13 X1 R4 C12 C1 R8 R19 SW1 R18 R5 F1 C25 R6 Rs2 Rs1 IC3x C21 D10 R12 mains R7 C23 http://onsemi.com 2 IC2 P1 R13 Q1 R21 R16 AND8246/D Design Steps At this time we don’t know the value of LLEAK, but we can choose a value of 3% of the primary inductance (i.e. 10 mH), which would not be too far from the final value. Considering 330 pF on the drain, at 375 V input voltage and 160 W of output power, which gives IP = 4.2 A, we obtain: VOVLEAK 730 V. But we only have 100 V available before reaching the MOSFET breakdown voltage. So we will need to add a clamp to limit the spike at turn−off. Please refer to application note AN1679 (available at www.onsemi.com) to calculate this clamp. You can also use a SPICE simulator to test the right values for the components. We chose to use an RCD clamp, using a 1N4937 diode, a 47 kW resistor and a 10 nF capacitor: it is an aggressive design (the maximum drain voltage will be very close to the maximum voltage allowable for the MOSFET), but it gives enough protection without degrading the efficiency too much. 1. Reflected Voltage Let us first start the design by selecting the amount of secondary voltage we want to reflect on the primary side, which will give us the primary to secondary turn ratio of the transformer. If we decide that we want to use a rather cheap and common 600 V MOSFET, we will select the turn ratio by: VINmax N · (VOUT VF) 600 V VINmax is 375 V and (VOUT + VF) is about 135.5 V. If we decide to keep a 100 V safety margin, it gives N < 0.92. We will choose a turn ratio of N = 0.91, which will give a reflected voltage of 123 V. 2. Peak Current Knowing the turn ratio, we can now calculate the peak primary current needed to supply the 75 W of output power. If we neglect the delay TW between the zero of the current and the valley of the drain voltage, we can calculate IPmax by: 5. Brown−Out Protection We want the power supply to turn on at 90 Vac, and turn off at 70 Vac. Start−up level is directly given by the resistor divider connected between high input voltage and BO pin, knowing that the threshold of the internal comparator is 500 mV. 90 Vac means 127 Vdc, so the ratio of the divider must be 254. Once the controller has started, an internal 10 mA current source is activated and flows out of BO pin, creating hysteresis. 70 Vac means 99 Vdc, so we want a 28 V hysteresis, corresponding to 22% of the start−up level. The corresponding threshold for the comparator is 390 mV, so the 10 mA current must create an offset of 110 mV across the equivalent resistance of the resistor divider. Those 2 conditions lead to 2 equations: VINmin N · (VOUT VF) IPmax 2 · POUT · h · N · VINmin · (VOUT VF) VINmin is 110 V and η is 85%. Plugging the other values gives us a maximum peak current of IPmax = 6.5 A. NCP1337 max current sense setpoint is 500 mV, so we should put a sense resistor RS = 0.5 V / 6.5 A = 0.077 W. We will use two standard 0.15 W resistors in parallel, that will allow IPmax = 6.67 A. 3. Primary Inductance To calculate the primary inductance LP, we need to decide the switching frequency range in which we allow the controller to operate. There are two constraints: at low line, maximum power, the switching frequency should be above the audible range (higher than 20 kHz). At high line, 50% nominal power, the switching period should be higher than 7.5 ms, to prevent the controller from jumping between valleys (because these discrete jumps between 2 valleys can generate noise in the transformer as well). If we still neglect TW, LP is then given by: 1 LP 2 · FSWmin · POUTmax RBOhigh RBOlow RBOlow and RBOhigh · RBOlow RBOhigh RBOlow 2 VINminN · (VOUTVF) h · N · VINmin · (VOUTVF) · 10−5 0.11 Solving these equations gives RBOhigh = 2.8 MW and RBOlow = 11 kW. But in reality there will be a non−negligible ripple on the DC input voltage, and the hysteresis should be increased in order to obtain the desired turn−on and turn−off levels. Final value for RBOlow is 15k (RBO2 in schematic), and 3.9 MW for RBOhigh (split in RBO = 2.7 MW and RBO1 = 1.2 MW to sustain the high voltage). A capacitor C7 is added between BO pin and ground to filter any noise, and to ensure a DC voltage. This capacitor value should be small enough, otherwise it may introduce a delay between input voltage collapsing and Power supply turn−off (a 10 nF ceramic capacitor gives good results). If we choose 20 kHz min for 160 W of output power at 110 Vdc, we obtain: LP 380 mH. To take tolerances into account, we can choose LP = 330 mH, and verify if it satisfies the second condition: For 80 W output power at 375 Vdc, TSW = 9 ms, i.e. FSW = 112 kHz. 4. Clamp We can calculate the overvoltage due to the leakage inductance: VOVLEAK IP 254 LCLEAK . TOT http://onsemi.com 3 AND8246/D 6. Overpower Protection when the input voltage is high: as a result overvoltage protection on VCC pin is lost. We have seen that full load maximum peak current at low input voltage is 6.5 A, but only 4.2 A at high input voltage. We need to create an offset on the current sense signal. As 500 mV on CS pin corresponds to 6.67 A, 2.3 A corresponds to a 172 mV offset. At 375 Vdc input voltage, BO voltage is 1.55 mV: as a result a 73.5 mA current flows out of CS pin during ON time. To create the desired 172 mV offset, it is necessary to insert a 2.34 kW resistor R6 in series. We choose a standard 2.2 kW value. Static Measurements Brown−Out Protection • Input voltage turn−ON level: • Input voltage turn−OFF level: 95 Vac 80 Vac Efficiency • At 230 Vac, 148 W IN for 135 W OUT • At 110 Vac, 154 W IN for 135 W OUT 7. Standby In order to reduce as much as possible the power wasted during standby mode, NCP1337 enters an efficient and quiet soft−skip mode. But because of the high output voltage of 135 V, any leakage current will create a significant output power, preventing the power supply to reach the requirement of less than 1 W standby power. This demonstration board thus includes a simple patented circuit that allows collapsing all unused outputs, while still powering the 5 V standby rail. This circuit is made of a regulated rectifier (around M1) connected between the high voltage output winding and the input of the 5 V linear regulator IC4, and of a switch (Q1) that changes the regulation setpoint. DZ2 is added to prevent voltage drops during transition from normal to standby mode. If the leakage current on the 135 V output is extremely low, this circuit can be omitted (see appendix schematic A). 91% 87% Standby Power • Noise−free • All outputs are low (135 V output is 12.7 V), except 5 V standby output which is maintained. IOUT consumption is taken on 5 V standby output. Controller is powered thanks to the Dynamic Self−Supply (DSS). IOUT VIN 0 10 20 30 40 230 Vac 390 mW 600 mW 780 mW 980 mW 1.18 W 110 Vac 230 mW 460 mW 700 mW 860 mW 975 mW • All outputs are low (135 V output is 12.7 V), except 5 V standby output which is maintained. IOUT consumption is taken on 5 V standby output. Controller is powered thanks to a forward−coupled auxiliary winding. 8. Controller Supply NCP1337 includes a DSS able to supply the controller without the help of any auxiliary supply. However this is possible only if the gate current is low, i.e. during standby in our case. So an auxiliary winding is necessary to supply the controller during normal mode, but DSS can be activated in standby, for instance in the case all voltages are decreased by the circuit described above. In order to minimize the power consumption of the DSS, HV pin can be connected to the half−wave rectified input voltage instead of the full−wave rectified bulk voltage. To further decrease the power consumed by the controller during standby, it may be interesting to prevent the DSS to turn on: this can be achieved by inverting the coupling of the auxiliary winding (see appendix schematic B). By creating the auxiliary supply from a forward winding instead of a flyback winding, it is possible to ensure a sufficient supply voltage even in standby mode with all voltages reduced. VCC voltage must then be clamped to protect the controller IOUT VIN 0 10 20 30 40 230 Vac 340 mW 470 mW 580 mW 730 mW 900 mW 110 Vac 140 mW 350 mW 540 mW 700 mW 820 mW • All outputs are at their nominal values. IOUT consumption is taken on 5 V standby output. Controller is powered thanks to the auxiliary winding. IOUT VIN 0 10 20 30 40 230 Vac 260 mW 380 mW 620 mW 740 mW 880 mW 110 Vac 180 mW 280 mW 400 mW 540 mW 690 mW http://onsemi.com 4 AND8246/D Static Measurements Soft−Start CS CS Drain Drain At 230 Vac, full load At 110 Vac, no load CS CS Drain Drain At 230 Vac, no load At 110 Vac, no load http://onsemi.com 5 AND8246/D Valley Switching At 230 Vac, full load At 110 Vac, full load At 230 Vac, half load At 110 Vac, half load Load Transients At 230 Vac, 20% to 80% load on 135 V output At 110 Vac, 20% to 80% load on 135 V output http://onsemi.com 6 AND8246/D Standby Vcc Vcc Standby burst at 110 Vac Vcc Vcc Standby burst at 230 Vac Transitions Between Modes 5V Standby 5V Standby 135V output 135V output Normal to Standby Transition Standby to Normal Transition http://onsemi.com 7 AND8246/D Board Layout 20V 12V 8V 5V 135V Standby switch: Left: normal mode Right: standby mode AC input Bill of Material IC1 IC2 IC3 IC4 X1 M1 Q1 T1 L1 F1 D1 D5, D10, D14, D16, D141 D6 D7 D11, D12, D111 D13 DZ2 DZ3 R1, R35 R2 Rbo Rbo1 Rbo2 Rhyst R3 R4 R5, R21 Rs1, Rs2 R6 NCP1337 TL431 SFH615A MC78L05 IRFIB6N60A BS108 BC547 TDK SRW42/15EC−X21V017, CLICK BCK4201−304 OREGA 47283900 RM4 2A 250V KBU4K 1N4007 1N4937 1N4148 MUR420 MUR460 3V9 6V2 1k 47k − −2W 2.7Meg 1.2Meg 15k − 47 15 33k 0.15 2.2k http://onsemi.com 8 R7 R8 R10 R11 R12 R13, R16 R17 R18, R31 R19 R33,R34 P1 C1, C2 C3 C4 C5 C7 C8 C9 C10 −4kV 10Meg − 330 150k 120k 5.6k 100k − 18k 1.5k 47k 1k 330p − −300Vac −X2 10p − −2kV − 220u −450V 450V 1u − −63V 10n − −630V − 33u − −25V C11, C13, C15, C25, C131 C12 C14, C16, C141 C17 C18 C20 C21 C23 C26 100n 330p − −1.5kV 1000u −35V 100u −25V 25V 1000u −16V 100u −200V 1n 2.2n −Y1 Y1 470n AND8246/D Board Picture Appendix Schematic A C19 R17 D13 135V 12 C20 0V 11 R2 D5 C8 R10 R1 T1 C4 6 D111 R31 4 D11 D7 Rbo1 Out12V 16 IC3 C14 C13 1 Rhyst C2 12V Out12V 17 8 D6 R11 C141 C131 C9 C5 Out20V 18 Rbo D1 20V Out20V 2 R3 8 2 7 3 6 8V Out8V Out8V 15 IC4 D12 IN C7 Rbo2 4 L1 5 C10 C11 R4 C1 C16 C15 C3 IC1 X1 R8 C12 F1 R19 R18 C25 Rs2 Rs1 IC3x C21 D10 mains R12 R7 C23 http://onsemi.com 9 5Vstby OUT C17 DZ2 13 R5 R6 Reg 5V ADJUST 14 IC2 P1 AND8246/D Appendix Schematic B C19 R17 D13 135V 12 C20 R2 D5 C8 R10 R1 D141 T1 C4 0V R33 11 D111 D14 M1 R34 Out20V 20V Out20V 2 8 R3 R31 D6 4 D11 C5 Rbo1 Rhyst C2 C7 L1 Rbo2 12V Out12V 17 6 D7 R11 C141 C131 C9 Rbo D1 R35 18 Out12V DZ3 C26 16 IC3 C13 1 8 2 7 DZ1 C14 8V Out8V Out8V 15 D12 IC4 D16 3 6 4 5 14 C11 C10 ADJUST C15 C3 IC1 5Vstby Reg 5V OUT IN C18 C16 C17 DZ2 13 X1 R4 R8 C12 C1 R19 SW1 R18 R5 C25 F1 R6 Rs2 Rs1 IC3x C21 D10 R12 mains R7 IC2 R13 P1 Q1 C23 R21 R16 ON Semiconductor and are registered trademarks of Semiconductor Components Industries, LLC (SCILLC). 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