Application Notes

APPLICATION NOTE
AN 00001
TEA6848H
A NICE RADIO
with
CIRCUMSTANTIAL
CONTROLLED
SELECTIVITY.
Version 1.2
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TEA 6848H A NICE RADIO
APPLICATION
NOTE
With Circumstantial Controlled Selectivity
AN
Abstract
The IC TEA 6848H is for small dimensioned Electronic Tuned AM/FM Car Radio
receiver, with advantage in application-area’s where the FM band is crowded.
They carry the following functions:
* AM receiver for long-, medium- and short- wave (up to 49 m) with
. reduced desensitization by cascode AGC
. noise-blanking and weak signal control.
* FM receiver with
. image cancelling on chip;
. dynamically controlled IF selectivity on chip
. keyed AGC;
. VCO for global application, with low side injection for Japan and Eastern Europe
. weather band included.
* A fast tuning Synthesizer with on chip control for inaudible RDS-AF updating.
* Digital Automatic Alignment for FM- RF circuit and for RF-fieldstrength indication.
AM and FM operate with worldwide application flexibility, given by peripheral
components and by software control via I2C-Bus.
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TEA 6848H
A NICE RADIO
with
Circumstantial Controlled Selectivity
Version 1.2
Author(s):
Sjakko Sandee, Gerrit van Werven
SLE-Eindhoven
System Laboratory Eindhoven,
The Netherlands>
Keywords
One chip AM/FM
Car Radio Receiver
On chip FM Image Rejection
and FM IF-selectivity
NICE/PACS
With acknowledgements for their valuable contribution to Joop Beunders, Hein van den Heuvel,
Kave Kianush, Jerry Lit, Oswald Moonen from SLE and to Wim van Dooremolen, Analog Advice.
Date: June 26th, 2000
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TEA 6848H A NICE RADIO
APPLICATION
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AN
Summary
The TEA 6848H in an LQFP80 package is a complete, highly-reliable/small dimensioned, AM/FM
Car Radio Receiver for global application.
It carries the following functions:
* AM receiver for LW/MW and SW (31 to 49m),
with double conversion, including
Ø linear AGC with high dynamic range, using cascode AGC and the AM pindiode BAQ806;
Ø fast level-detection;
Ø IF-output matched for AM stereo decoding;
Ø Noise Blanking Circuit at IF;
* FM receiver for broadcast frequencies 65 to 108MHz,
with double conversion, including
Ø image rejection on chip;
Ø controlled IF 2 selectivity (detecting adjacent channels / modulation index / frequency offset)
Ø digital auto alignment for the RF-tuned circuit;
Ø large AGC range with keyed agc feature;
Ø inaudible RDS updating feature
* Weather-band application 162.4 to 162.55MHz.
* Tuning Synthesizer, using
Ø fast tuning VCO with low phase noise
Ø a VCO, designed for global application, with low side injection for Japan and Eastern Europe
Ø a reference from a x-tal oscillator, designed for low interference.
* Interface, matched to Audio Signal Processors TEA6880H (CASP) and SAA7706/7709(CDSP)
* Bus Transceiver (I 2C), operating at 3.3 and 5 Volt,
Ø for tuning (PLL) and RF auto-alignment;
Ø programmable starting point / slope for AM/FM fieldstrength detection;
Ø for AM & FM wide band AGC-start;
Ø for alignment of: IF 2 filter-centre frequency, filter gain and frequency offset.
AM and FM double mixing goes via 10.7MHz to 450kHz.
The PLL Synthesizer has a fast tuning (for RDS AF-updating): < 1ms for a maximum step.
Special care has been taken for interference immunity, having the synthesizer on chip with the RF.
.
Excellent sensitivity figures can be achieved:
at AM, 30% modulation, typ. 43µV (from 15/60pF source) and
at FM, ∆f=±22.5kHz, typ. 1.4µV (at 75 Ohm source).
Large signal figures S/N: typ 60dB at AM (m=30%) and about 63dB at FM (∆f= ±22.5kHz).
High intercept points: At AM is IP3=130dBµV rms and at FM 117 dBµV rms for “ín-band signals”.
The peripheral components are limited,
Ø using just one x-tal as reference:
. for 2nd conversion,
. for Synthesizer reference,
. for Audio Signal Processor reference,
. for sequential RDS-updating circuitry and
. for IF-counter window.
Ø having wideband AM input (no RF- tuning or -switching);
Ø no LNA in FM front-end; only one single tuned circuit;
Ø only 2 standard ceramic filters and no demodulator coil or resonator.
With Circumstantial control for Precision Adaptive Channel Suppression (PACS), IF 2 gives performance
matching to local requirements. Performance setting by software gives matching to regional requirements.
The AM/FM receiver module for New In Car Entertainment (NICE) can be realised with small PCB
dimensions.
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TEA 6848H A NICE RADIO
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AN
Contents
SUMMARY
LIST OF FIGURES
1. INTRODUCTION.
2. FUNCTIONAL DESCRIPTION
3. FEATURES.
4. CIRCUIT DESCRIPTION
4.1 AM SIGNAL CHANNEL
4.1.1 RF INPUT AMPLIFIER
4.1.2 AM-MIXERS
4.1.3 IF AND DETECTION
4.1.4 AM NOISE BLANKING
4.1.5 SEARCH STOP INFO
4.2 FM SIGNAL CHANNEL
4.2.1 RF
4.2.2 IF AND DEMODULATION
4.2.3 IF BANDWIDTH CONTROL
4.2.4 SEARCH STOP INFO
4.3 OSCILLATORS
4.3.1 VCO
4.3.2 X-TAL OSCILLATOR
4.4 TUNING SYSTEM
4.4.1 DIGITAL AUTOMATIC ALIGNMENT
4.4.2 RDS UPDATING
4.4.3 ADAPTIVE SYNTHESIZER
4.5 I2C-BUS CONTROL
4.6 SUPPLY
5. LAYOUT GUIDELINES
6. APPLICATION
6.1. AM APPLICATION
6.2 FM APPLICATION
6.3 GLOBAL APPLICATIONS
6.4. OPTIONAL APPLICATIONS
Option1. AM- 49m reception
Option2. System applications
. RDS
. Weather Band
. Audio Signal Processing
APPENDIX 1. I2C-BUS DATA
APPENDIX 2. ALIGNMENTS
APPENDIX 3. a. Module PCB
b. Module application diagram
c. Components List
APPENDIX 4. Module Specification
APPENDIX 5. Weather-band receiver
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LIST OF FIGURES.
Fig. 1 TEA 6848H Simplified Block-diagram
Fig. 2 AM up/down conversion
Fig. 3 AM RF input amplifier for LW/MW
Fig. 4 AM Pin-diode characteristic
Fig. 5 AM Desensitization
Fig. 6 AM RF Aerial Filter response
Fig. 7 AM RF Bandpass filter
Fig. 8 AM 1st IF selectivity
Fig. 9 AM 2nd IF selectivity
Fig. 10 AM IF and Detection
Fig. 11 AM Level Voltage
Fig. 12 AM Noise Blanking Test Pulse
Fig. 13 FM functional Diagram
Fig. 14 FM RF Tuned Bandpass filter
Fig. 15 Quadrature Mixing
Fig. 16 FM Image Cancelling
Fig. 17 FM RF Wideband AGC
Fig. 18 Keyed RF-AGC at FM
Fig. 19 NICE / PACS IF2
Fig. 20 FM IF1 selectivity
Fig. 21 Resonance amplifier model
Fig. 22 FM IF2 selectivity = f(Bandwidth)
Fig. 23 FM Demodulator Circuit
Fig. 24 FM-IF Signal and Noise behaviour
Fig. 25 The FM-IF bandwidth control circuit
Fig. 26 IF2 Bandwidth
Fig. 27 Threshold Extension at FM
Fig. 28 Tuning System
Fig. 29 Inaudible mute behaviour
Fig. 30 RDS AF-check
Fig. 31 Adaptive Synthesizer block diagram
Fig. 32 TEA 6848H I2C Bus structure
Fig. 33 AM Gain distribution
Fig. 34 AM Signal and Noise behaviour
Fig. 35 AM Intermodulation characteristics
Fig. 36 FM Gain distribution
Fig. 37 FM Signal, Noise and Distortion
Fig. 38 AM LW/MW/SW-49m a 5th order Low Pass Filter
Fig. 39 AM SW - 49m Signal and noise behaviour
Fig. 40 Block-diagram of audio signal processor CASP
Fig. 41 FM Level Voltage
Fig. 42 TEA 6848H settings at FM
Fig. 43 Module PCB
Fig. 44 FM / AM-MW/LW application
Table 1
Table 2
Table 3
IF counter
Programmable Divider
Frequency-band Setting
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TEA 6848H A NICE RADIO
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1. INTRODUCTION
The TEA 6848H in an LQFP80 package is a complete, small dimensioned, AM/FM Car Radio
Receiver for global application. It carries the following functions:
* AM receiver for LW/MW and SW (31 to 49m),
with double conversion, including
Ø linear AGC high dynamic range, using cascode AGC and the AM pindiode BAQ 806;
Ø fast level-detection and a signal- ‘low distortion’ detector;
Ø IF-output matched for AM stereo decoding;
Ø Noise Blanking Circuit at IF;
* FM receiver for broadcast frequencies 65 to 108MHz,
with double conversion, including
Ø image rejection on chip for both frequency conversions;
Ø IF2 selectivity on chip, controlled via detection of adjacent channels/ modulation index / and
frequency offset;
Ø digital auto alignment for the RF-tuned circuit;
Ø large AGC range with keyed agc feature;
Ø inaudible RDS updating feature.
* Weather-band application 162.4 to 162.55MHz, with
Ø Image rejection and IF 2 channel-selectivity on chip;
Ø output current for front-end switching;
Ø audio gain compensation for standard output level.
* Tuning Synthesizer, using
Ø fast tuning VCO with low phase noise;
Ø a VCO, designed for global application, with low side injection for Japan and Eastern Europe;
Ø a reference from a x-tal oscillator, designed for low interference.
* Interface, matched to Audio Signal Processors TEA6880H (CASP) and SAA7706/7709(CDSP).
* Bus Transceiver (I 2C), operating at 3.3 and 5 Volt,
Ø for tuning (PLL) and RF auto-alignment;
Ø programmable starting point / slope for AM/FM fieldstrength detection and AM AGC start;
Ø for setting IF 2 filter-centre frequency, filter gain and frequency offset.
AM and FM double mixing goes via 10.7MHz to 450kHz, see Fig. 1
Fig. 1 Simplified Block Diagram of the TEA6848H Receiver Architecture.
Circumstantial control of the Precision Adaptive Channel Selectivity (PACS), IF 2, gives performance
matching to local requirements. Performance setting by software gives matching to regional requirements.
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This report describes the standard application FM and AM/MW based on TEA6848H.
2. FUNCTIONAL DESCRIPTION
The strong growth in the number of FM transmitters, stimulated by various services such as
R(B)DS and DARC, has increased the demand on the channel selectivity of the receivers. As a
consequence, the number of IF ceramic filters in a standard tuner application, in particular in Car
Radio, has grown from 2 to 3 and sometimes 4 and with narrower bandwidths than ever before.
This increases costs (more components that have to be matched) and reduces performance (higher
THD and poorer data reception). This trend is likely to continue into the next decennium.
A solution could be switching to a narrow ceramic filter. However, drawbacks are that the extra
narrow band filter will have to be carefully selected to match with the rest of the channel (filters and
the PLL crystal reference). In the narrow state, THD is high and data on ultrasonic sub-carriers is
lost, moreover actions of switching back and forth between the 2 states of selectivity causes audible
disturbances.
The IC described contains an integrated time-continuous adaptive FM-IF filter, whose instantaneous
bandwidth is determined by all relevant system parameters. The combination of the filter structure
and its bandwidth control algorithm deliver higher dynamic selectivity, improved sensitivity and low
THD at high frequency deviation without any audible artefacts. The automatic alignment of the filter
centre frequency eliminates IF channel tolerances and makes it suitable for global applications. Next
to this dynamic controlled IF-selectivity on chip, a reliable high performance concept with minimum
system price has been obtained, with special attention on interference reduction. To that end image
rejection is obtained by conversion to a high IF at AM and on-chip image-rejection at FM.
The AM Section is a double conversion receiver. The first IF is 10.7MHz, which allows a wide band
RF input stage without tracking requirements. The RF input has a wide dynamic range with a linear
AGC, using a cascode AGC at the RF-amplifier and the AM pin-diode BAQ806. The start of AGC
setting is Bus programmable by the set maker. The cost of IF filtering is kept low by a second
conversion to 450kHz. The AM IF stage provides soft mute, AM stereo compatibility and a fast stoplevel detection. Different antennas (capacitive / ohmic) are possible. AM noise detection with blanking
at IF is included.
The FM section has also double conversion architecture with the same IF frequencies as the AM
channel for maximum component sharing. The first conversion stage utilises a quadrature-input
stage combined with a wide band quadrature phase shift circuit for 30dB internal image rejection at
10.7MHz. The RF input filtering requirements are therefore reduced and can be met with a single
tuned stage. The RF Digital Automatic Alignment (DAA) block achieves the tracking of this tuned
circuit. The linear FM AGC has programmable start points and offers an optional Keyed AGC
function. The input quadrature mixers are designed for low noise and large signal handling so that
no FET Low Noise Amplifier (LNA) is required.
Only two relatively wide ceramic filters are required for the first IF selectivity. The second frequency
conversion provides quadrature signals at 450kHz, obtaining integrated IF2 image rejection. The rest
of the IF selectivity is then carried out by the integrated adaptive filter section, which has adjustable
centre frequency and bandwidth. The centre frequency is aligned by Bus, but the bandwidth is
dynamically controlled. The integrated resonator of the demodulator circuit is matched to and
aligned with the filter. The bandwidth control circuit determines the instantaneous bandwidth of the
filter for dynamic conditions. Combined with an Adjacent Channel Detector the IF bandwidth control
takes care for Precision Adjacent Channel Suppression (PACS).
The FM channel is prepared for Weather Band application. In the Weather band (WX) mode, the
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TEA 6848H A NICE RADIO
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integrated filter is automatically switched to its narrowest bandwidth to give adequate WX channel
selectivity.
Both AM and FM level outputs are aligned by Bus for start and slope. The alignment coefficients for
FM RF tracking and AM/FM level can be stored in a memory (e.g. EEPROM) for each individual
receiver.
The VCO has been defined such that all AM/FM-reception bands can be accessed without band
switching or any changes to the application. The wide band up-conversion AM input combined with
the programmable VCO AM dividers reduce the tuning range such that LW, MW and SW become a
continuous band without mechanical switching. The VCO FM dividers bring the required FM
frequency ranges for tuning in Western Europe, Eastern Europe, Japan, USA and Weather Band,
taking into account a low-side oscillator injection for Japan and Eastern Europe bands. Therefore
one VCO tunes to all bands in the same tuning voltage range.
The Adaptive PLL Tuning System combines low phase-noise and low reference spurious
breakthrough with a fast tuning response. During FM frequency jumps two charge pumps are active
enabling stability and fast tuning to be achieved. After the fast frequency jump only one pump is
active, resulting in a small loop bandwidth and low noise operation of the tuning system The crystal
oscillator operates in a linear-current mode to avoid interferences to the sensitive RF parts. This
oscillator generates all the necessary reference signals for the tuning operation and frequency
conversions.
The Mute circuitry. To provide a better reception, or other information, quality control of other signal
channels is used, for example in Radio Data System (R(B)DS) alternative frequency checking. This
usually causes audible breaks in the main channel, as the audio signal has to be muted while the
receiver is tuning to other frequencies. Muting actions are detected in two ways. Gaps in the audio
signal may be perceived if the muting time is not short enough. The other mechanism is the
distortion of the power spectrum, which is independent of the muting time. In practice, with actual
audio signals, muting times below 7ms with gentle slopes of 1ms are inaudible. To achieve FM
signal quality checks of 5ms, the tuning times have to be reduced to below 1ms, and the frequency
jumps have to be made independent of the slow Bus communication times. The first requirement
has to be accomplished by the tuning system, whereas the latter was solved by inclusion of 'local
intelligence' in the form of a sequential circuit that controls tuning operations during quality checks.
The I2C Bus makes different regional requirements programmable. It has specific building blocks in
order to perform inaudible frequency jumps: the sequential circuit, a shaped mute and the adaptive
PLL tuning system.
The IF2 filter-frequency alignment circuit centres the integrated filter for maximum RF level, thereby
eliminating both IC process and the PLL crystal reference tolerances.
The IF2 filter-gain alignment provides constant gain when the bandwidth is varied.
The frequency offset detector , which detects the offset between the momentary demodulator output
and the nominal output, is aligned to the nominal demodulator output.
The IC employs Digital Automatic Alignment of RF tuned circuit and level signals to avoid
mechanical alignments, using pre-aligned coils.
This, and the integration of FM-IF filtering, the FM image rejection, the tuning system and the AM
double conversion topology makes the application of TEA6848H simple.
3. FEATURES.
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1. Global Tuner concept to match on geographical requirements, including Weather Band
2. Modular, small dimensioned design; one chip receiver having few external components
3. Compatibility with both analogue and digital audio processors
4. Digital alignment
5. High performance with synthesizer on chip for high immunity and fast tuning
6. Fast Station detection and quality checking
7. Low interferences with FM image rejection/ AM IF Noise blanking and a linear Xtal-oscillator
8. Smooth operation with a.o. inaudible RDS updating
9. Circumstantial controlled FM-selectivity, to reduce the adjacent channel interferences
10. Flexibility by programmable settings, AM-stereo IF-output etc
11. High sensitivity even at very large signal conditions: high dynamic range (AM RF cascode AGC)
12. Innovative design towards low price with
. one X-tal oscillator for all reference frequencies
. standard IF-filters
. analogue AM-agc with pin-diode.
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4. CIRCUIT DESCRIPTION
See Fig. 44 for Total Circuit Diagram.
Note: In the description some application info is given in italic writing.
The main supply is 8.5 Volt; a 5 Volt supply is used for digital parts and some analogue
functions. The external voltages are stabilised, with ripple rejection of >50dB at 800Hz ripple,
creating internal reference voltages and currents.
4.1 AM-signal channel.
Main features:
*AM-RF input is voltage driven : Antenna to be capacitive or ohmic (active aerial);
*Linear RF-AGC (plop free), using an AM Pin-diode BAQ806 and a FET-amplifier with an
optional cascode transistor. Flexibility is realized by Bus-controlled setting of agc threshold.
*AM noise blanking with a noise detector at 1st mixer output and blanking at mixer 2.
*Fast station-level detection.
*AM-stereo compatibility
To avoid RF-selective circuitry for image rejection, or, to permit wideband RF-input, the input
frequencies are mixed to a 1st IF_Frequency of 10.7MHz. By doing so the main image
frequencies (image 1 in Fig.2) are 21.54 to 23.11 MHz, which is so far from the receiving band,
that they can be easily suppressed by a Low Pass Filter in front of the 1st Mixer.
AM up/down- conversion (via high to low IF)
Image3
Image2
IF2_image
Osc.2
IF2
Image1
Osc1
RF
IF1
Frequency (MHz)
0
5
10
15
20
25
Fig. 2 AM MW/LW conversion via 10.7 MHz to 450 kHz
The 1st IF-freq. (filtered with ceramic filter of 10.7MHz, common used with FM-IF) is mixed
down to 450kHz, a standard frequency where a low priced filter takes care for channel selectivity
before detection takes place. The image frequencies 2 and 3 are caused by this 2nd mixing, as
the VCO has transferred these image frequencies to 9.8MHz (here called IF2_image). This 9.8
MHz will be mixed to 450kHz by the 2nd mixer and therefor 9.8 MHz has to be suppressed in the
10.7MHz 1st IF filter. Suppression at 9.8MHz in a first IF selectivity acc. to Fig. 8 is about 65dB.
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4.1.1 RF Input Amplifier
The Aerial input (wideband-) amplifier, shown in Fig. 3, consists of:
* an input LC to separate AM from FM,
* the RF-amplifier with FET BF862 and a BC848C in cascode,
* an RF-AGC amplifier with pindiode BAQ806;
* surge-protection double diode BAV99;
* an output AM-bandpass-filter before entering the first mixer.
The aerial is capacitive loaded by about 90 pF, being the sum of
. the input capacitance of the FET BF862 (10pF, but dynamically about 60pF *));
. the zero capacitance of the pin-diode (about 5pF);
. the capacitive input of the FM-RF part (about 15pF) and
. the parasitic capacitance of aerial-connector and PCB (about 9pF).
With a capacitive (telescope) antenna, acting at 1MHz as a 15 to 60pF divider,
the total gain loss from dummy aerial input to FET-gate is about 20dB.
*
) Note: In cascode-input application this input capacitance is just 12.5pF (10+2.5pF).
Fig. 3 AM-RF Input Amplifier for LW/MW .
a. The AM-pin-diode BAQ 806
With the BAQ806 (see Fig. 4), special designed on high linearity, with slow operation for
AM frequencies, the RF-signal can be attenuated over a range of about 50dB.
The pin diode acts like a variable resistor for RF signals. Its linearity results in
excellent large signal capabilities. (In the given application IP2 and IP3 values are 140 and
130dBµV respectively.)
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However, by its virtue of behaviour as a
resistor, it’s a source of noise. As this
would result in loss of sensitivity
(desensitization) at the start of pindiode control, its control will be
delayed, using gain control in the
cascode RF amplifier first.
b. AM RF Amplifier:
The FET BF862 has a low noise of
Fig.4 AM Pin Diode Characteristic: BAQ806
0.8nV/√Hz. With its high trans- conductance the gain is Gm*Rload= 25dB. Gain control at
the gate is linear: without agc-plops and with large signal handling.
The concept matches to different antenna characteristics.
With this FET in the application of Fig.44, the overall AM sensitivity is typ Va= 43µV for S/N= 26dB
(at m=0.3), defined with a 15/60pF dummy antenna.
A S/N of 10 dB is reached at an input signal of Va = 7µV from a dummy antenna, see Fig. 34.
at Fa1=990 kHz; m=30%_1kHz
Wanted signal Va1 (dBµV)
for 26 dB SINAD
c. The RF-AGC
In the TEA6848H IC, the RF-signal at the mixer1 input (pins 22/23) is detected, to build up a
RF-AGC voltage available at pins 26 and 27. The gain control starts at RF-amplifier, the
bipolar transistor ‘on top’ of the FET, controlled by the AGC-signal delivered from pin 25,
followed by additional gain control with the pin-diode, to which end pin 28 sinks a current up
to 15mA.
The cascode-control lowers the drain voltage of the FET, in turn decreasing the FET
transconductance when the drain-source voltage has brought the FET in its linear region.
The gain control range of the cascode stage has to be limited to about 10dB to avoid
overloading the FET, special to avoid third-order/ cross-modulation at higher signals.
The BF862 maximum gate voltage related to cross-modulation performance is about
100mVrms (IP3 is 127dBµV).
A practical limitation is the drain-source voltage: not too low for reason of spread.
In the given application the gain
control of the cascode stage is 10
Desensitization of NICE at AM
dB (where the drain-source voltage
75
Pin-diode + Cascode control
ranges 4.1 to 0.26 Volt).
70
Pin-diode only
65
Notes:
No RF AGC
60
a. If a different type of bipolar
55
50
transistor (with higher Ft) is used
45
in the cascode stage, it is
40
possible that under certain
35
30
conditions the stage is showing a
Unwanted signal Va2 (dBµV) at Fa2=1500 kHz; m=0
spurious oscillation. This can be
avoided by ensuring that the
Fig.5 AM desensitization: SINAD influenced by an unwanted
decoupling line at the base is as
signal.
short as possible.
b. If, for cost reasons, the cascode AGC is not applied, the PIN-diode AGC will take over
at the original start of AGC.
91
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As an example desensitization by a 1500kHz signal at 990kHz tuning has been given in
Fig. 5 for ‘pin-diode control only’ and for ‘delayed pin-diode control’, using cascode stage
control as well. The last one behaves close to the situation of no RF-agc.
To avoid harmonic distortion at large in-band signals an additional IF2-detector can start
the RF-agc earlier (e.g. at V23=30mV) then in case of large 'out-of band signals', such to
keep the sensitivity for weak signals as high as possible in the last mentioned situation.
The wide-band agc starting point is programmable via the I2C-Bus (2-Bits).
For 80% modulated out of band large signals the RF-agc starts in TEA 6848H application
(Fig.44) at signals of 90/120/150/180mV , dependent on Bus setting.
With C= 22µF at pin 27 and C= 220nF at pin 26 the overall attack time of this AGC is
25ms; decay in 250ms, switching from 0.05mV to 0.5 Volt at AM 990kHz.
d. Input Filter:
The input filter takes care for
attenuation
of
undesired
frequencies.
e. The AM-Bandpass filter at
the FET-output
The output signal of the RFFET has to pass a fixed band
pass filter that suppresses the
image band before the signal
is converted to 10.7MHz in the
first mixer.
In
the
standard
LW/MW -application the low
pass filter has a cut-off
frequency of 2MHz, which 4th
order filter, see Fig.7, gives
>60dB suppression for images
1 and 2.
Fig. 6 AM RF aerial filter response
For
reception
of
the
49m-SW-band a filter with
cut-off frequency of 6MHz has
to be chosen
Fig. 7 AM RF Band pass filter response
f. The Surge protection.
The high -speed double diode BAV99 protects against static charging at the aerial.
The matching of the two diodes set them each at half the supply voltage to minimize
distortion by non-linear effects (note that capacitive coupling takes care for a stable dcmidpoint).
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4.1.2 AM Mixers
* The 1st Mixer, entered at pins 23/22, input resistance about 20 kOhm. The mixer
transconductance is 2.5mA/V.
To receive MW 530 to 1710 kHz an oscillator- frequency of 11.23 to 12.41MHz is required
at the 1st Mixer, to mix up to the 1st IF of10.7 MHz.
Important for the mixer is a low noise voltage, being 6nV/√Hz, and low intermodulation (IP3
is about 138dBµV at 2.8kOhm ac-load at mixer output). The mixer operates at a current of
2x6mA, having a large signal handling (-1dB compression) of > 500mVrms.
* The VCO (pins 49/50), delivers the required frequencies via an Oscillator-Freq.-Divider,
dividing the VCO-frequency by 20 at MW/LW operation; therefor the VCO operates at
216.88 to 248.2MHz. For SW the division ratio is 10.
It is done in this way to have one VCO for both AM and FM.
• The
1st
IF-filter,
symmetrical at the mixeroutput, pins 18/19, is a
tuned LC circuit with a
ceramic 10.7MHz filter,
common used with FM,
having behaviour as shown
in Fig.8.
An LC circuit with C= 150pF
and Q0= 55 is loaded, giving
a mixer output impedance of
2.8kOhm. With the 330 Ohm
ceramic filter via a coil turn
Fig.8 AM 1st IF selectivity
ratio 8 to 2, the gain of given
mixer1 is 17dB, resulting in about 1.5dB from mixer1 input to mixer2 input. The choice of
turn-ratio is weighted by AM-sensitivity and third order intermodulation, related to the noise
and IP3 contribution of the 2nd mixer and detector. (The larger the mixer gain, the better the
sensitivity, but at the cost of IP3). Moreover care has to been taken to 9.8MHz suppression
for image rejection.
After this 1st IF selectivity, the signal enters the 2nd Mixer at pins 14/15.
* The 2nd Mixer mixes the 1st IF of 10.7MHz down to 450kHz with an oscillator signal at
10.25MHz, obtained from a X-tal Oscillator. The mixer2 transconductance is 1.6mA/V; the
input resistance is 330 Ohm.
At 330 Ohm source its noise voltage is about 15nV/√Hz; biased for 2x4.5 mA current.
The mixer has a large signal handling (-1dB compression) of > 1.1Vpeak. IP3 is about
137dBµV at 1.5kOhm mixer output load, measured with signals at 50kHz distance.
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4.1.3 IF and detection
The Mixer output (pins 77/78) passes a 450kHz narrow band IF-filter (LC plus a 6_pole
ceramic filter, see Fig.9) and enters the IF section (at pins 6/3; pin 66 is AM-IF2 ground).
Including the losses in the 450
kHz filter, the gain from mixer2
input to IF2 amplifier input is
5dB, which makes the gain
from input-dummy to IF2 input
11dB.
Note: In the given application a
CFWS450H filter (6th order) is
used,
to
obtain
highest
performance as far as selectivity
and
stopband-attenuation
is
concerned.
Fig. 9 AM 2nd IF ceramic filter
selectivity
The AM-I.F. System (see Fig. 10), takes care
for:
. amplification with automatic gain control
. field strength level information
. a gain-controlled IF signal for AM-stereo
application
. AM-signal detection over a large dynamic
signal range, such with
. fast level detection
. smooth behaviour at small signal level
using soft mute.
* The AGC.
The IF-amplifier has a 3-stage gain control
with careful take-over behaviour to keep
distortion low. The input impedance of this IF2
amp. (2kOhm) has been matched to ceramic
filter applications. The equivalent noise
voltage is below 18nV/√Hz at 2kOhm source.
It can handle min. 1.0Vpeak with low distortion.
The 89dB agc-range starts at 20µV IF2 input
Fig. 10 AM IF and detection
signal (peak level). The time constant
(pin 79; commonly used with the time
constant of the FM frequency offset detector) in this AGC influences both settling time and lowfrequent modulation distortion. A 10µF capacitor gives
550 ms settling time with acceptable distortion of 0.3% for 400Hz/80% modulation (1.5% at 100Hz).
By Bus the settling time can be changed to 10 times faster (in test-mode).
This IF system is sensitive: V6-3 = 45µV for S/N=26dB at 30 % modulation.
At large signal a max S/N of 60dB is reached (30% modulation).
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* The detector
The envelope detector is with an on-chip 100 kHz low-pass filter to remove IF-frequency
components from the detected signal.
The A.F. -output level V56= 290mVrms at 30% modulation over an IF2 input signal range
V6-3 = 0.1 to 400mV; THD at m=80% is 0.3 % for a signal with 400Hz modulation.
* Mute
A mute function at the output of the detector gives a possibility for soft mute setting. Switched
by Bus, one can change the -10dB audio output from 6µV IF2 input signal towards 24µV (see
example in Fig. 34). This 12dB mute function is driven from the AGC-detector, not from the
level detector.
* The Level detection
To obtain fieldstrength information, the level detector delivers dc-information over a signal
range of about 20µV to 1 Volt at IF2 input (pins 6-3). The dc-information (see Fig. 11),
available at pin 70, is obtained via a second IF-channel (limiter / detector), such to have a fast
operating level detector. The slope and the starting point can be controlled by Bus for
customers’ flexibility as well as to match on product-spread: Digital Automatic Alignment DAA
(see Appendix 2). The slope, typical 800mV/20dB, will mostly not be aligned.
Special attention has been paid to the temperature compensation of the level info.
Fig. 11
Inside the IC the AM level information is only is only used to desensitize the AM noise blanker, which
occurs for levels >2V at pin 70.
* The AM-stereo info.
Mono/Stereo-controlled by the I2C-Bus; pin 56 can deliver (instead of mono a.f. output) a
limited, gain controlled, AM-IF2 signal to drive an AM Stereo decoder.
The IF2 output is 180mVrms at V6-3 = 5mV, where at pin 56 the output resistance is
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500 Ohm max.
The output is matched to the spec. of an AM-Stereo demodulator, like Motorola MC 13022.
4.1.4 AM Noise Blanking
At the output of the first
Repetition freq.: 500 Hz
mixer
(ignition-)
Pulse width
: 50 ns.
interferences
are
Rise-/fall time : < 5 ns.
detected, while blanking is
Peak-amplitude: 5 Volt
realised in the second
mixer. The noise blanker
is active only when the
(digital-) aligned level
Fig. 12 AM Noise Blanking Test Pulse
voltage at pin 70 is below
2 Volt, corresponding with e.g. Va <150µV (determined by DAA setting). The trigger sensitivity
can be modified by changing the voltage at pin 5. A resistor connected from pin 5 to Vcc or to
ground (e.g. 68kOhm) will increase respectively decrease sensitivity. The noise blanker will
be de-activated by adjusting the voltage at pin 5 to ~2V with a resistor to ground. Blanking
time typical 7.5µsec with C= 6.8nF at pin 55.
In Fig. 12 a definition of interference pulse, as used for testing, has been given.
4.1.5 Search -stop information
For station detection the signal quality is analysed in terms of fieldstrength and IF2 frequency.
At a search the AM the tuning step is 1kHz (at a reference frequency of 20kHz, with the VCOdivider M= N1*N2= 20).
The IF AGC-amplifier delivers the fieldstrength level information analogue to pin 70, to be
used in the Car Audio Signal Processor. For this and for the AM-noise blanker triggering the
starting point must be aligned with the help of the DAA.
Besides the fieldstrength level, the exactness of the IF can be used for stop-information :
An IF-counter counts the 450kHz IF signal with 8µV sensitivity (at aerial-dummy input for
m=0%).
In the AM-mode the counter counts the output signal of the IF-amplifier fast.
The resolution is
∆Fo = 1/tc = 500Hz for
tc = 2ms.
or
50Hz at
tc = 20ms; to be selected by Bus.
The I2C-Bus transmits this IF-count information to the µComputer; for IF=450kHz the readout
(Hex) is 084H with tc = 2ms and 028H at tc = 20ms.
The reference frequency for the counter window is obtained, via dividers, from the X-tal
oscillator.
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4.2 FM-signal channel.
The FM receiver has also a double conversion architecture with the same IF frequencies as the
AM channel for maximum component sharing. The 2nd oscillator, a crystal oscillator, operates in a
linear mode to avoid interferences to the sensitive RF parts.
Only two relatively wide ceramic filters are required for the first IF selectivity. The second frequency
conversion provides quadrature signals at 450kHz, obtaining integrated IF2 image rejection. The rest
of the IF selectivity is then carried out by the integrated adaptive filter section, which has adjustable
centre frequency and bandwidth. Fig. 13 gives this interesting part of the FM-channel.
The centre frequency is aligned by Bus, but the bandwidth is dynamically controlled. The bandwidth
control circuit determines the instantaneous bandwidth of the filter for dynamic conditions.
The FM-channel can be set to receive weather band. In the Weather band (WX) mode, the integrated
IF2-filter is automatically switched to its narrowest bandwidth to give adequate WX channel
selectivity.
10.7MHz
IF1
Downconverter
I
Soft- clipper
Integrated
IF-filter
Q
Limiter
I
Q
Demodulator
MPX-out
Level info
:2
Ref. Osc
OSCMHz
20.5
MHz
Σ
ACD
Threshold
Extension
(ACD-signal)
Modulation
detector
Frequency
offset
detector (DC-offset)
Bandwidth control
block
Fig. 13 FM-channel with IF functional diagram
Note that special functions are added for IF1- & IF2- image rejection / Digital RF-alignment /
Circumstantial IF2-Bandwidth Control and RDS AF- updating.
4.2.1 RF
The first conversion stage utilises a quadrature-input stage combined with a wide band phase shift
circuit for 30dB internal image rejection at 10.7MHz. The RF input filtering requirements are therefore
reduced and can be met with a single tuned stage. The RF Digital Alignment (DAA) block achieves
the tracking of this tuned circuit. The linear FM AGC has programmable start points and offers an
optional Keyed AGC function. The input quadrature mixers are designed for low noise and large
signal handling so that no FET Low Noise Amplifier (LNA) is required.
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The RF-part contains
. Aerial-input selectivity
. Mixer
. Image Frequency filter
. RF-agc
. Keyed AGC
* Aerial-input selectivity
The aerial signal has been
coupled to a single tuned
filter via a wideband
bandpass and the agcpin-diode circuitry. Having
passed the tuned filter
with varicap BB814, a
Fig. 14 FM RF Tuned bandpass Filter
transformer couples the
a-symmetrical rf-signal to
the symmetrical mixer input at pins 30/33. The tuned filter, having a quality figure Q of about
25, and a transfer characteristic as shown in Fig.14 (measured for the application of Fig.44)
is aligned automatically, see chapter 4.4.1. The tuned circuit has an additional rf notch filter,
using a printed coil to the midpoint tap of the coil of the parallel tuned circuit. This external
notch takes care for
>30 dB rejection at all image frequencies.
* Mixer
The RF-signal, which enters the IC symmetrically at pins 30/33 (pin 31 is the RF-ground),
passes the voltage to current converter, the mixer and a quadrature filter block (90° block in
Fig.15). The mixer, with a bias current of 12mA (having optimum source impedance of
~200Ω), has a noise figure of 3dB and a signal handling of 100mVpp for -1dB compression.
Input impedance 2.7kOhm // 4pF; output >100kOhm. Third order intermodulation IP3 is
117dBµV at the input of the mixer.
With its conversion transconductance of 12.5mA/V the mixer gain from dummy aerial to the
IF transformer input is 33dB and 16dB to the 1st IF amp. input. Such with the given 10.7MHz
IF selectivity, which, by the way, has been common used with AM 1st IF.
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20
Fig. 15 Quadrature
mixing
TEA 6848H A NICE RADIO
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AN
* Image Frequency filtering
To avoid the necessity of 'High
RF-selectivity
for
image
rejection',
the
image
frequencies are suppressed
on chip with a quadrature
mixer, driven by sin- and cososcillator signals. With a 90°
phase shifter and adder,
Fig.15, image cancellation of
30dB is realised (see Fig.16).
Note: A reference voltage for the
Q-mixer is decoupled by 22nF, pin
29; D.C 7.1V at FM (3.6V at AM).
Fig.16 FM image cancelling with quadrature
Mixer computer simulation)
The oscillator signals are
delivered from the VCO via a :2 divider.
* RF-agc
The RF-signal at the mixer
input has been detected for
RF-agc
(see
Fig.17),
delivering a current up to 11.5
mA (from pin 35) to control
pin-diodes in front of the tuned
RF-circuit.
The application of Fig. 44
shows a pin-diode-control
where parallel damping is
applied with two pin-diodes.
Note that for high stability in
the agc loop a series resistor
of at least 47 Ohm with a 47µF
decoupling capacitor at the
Fig. 17 The Wide band RF AGC
diode-current output (pin 28) is
recommended.
As RF-agc in front of the RF-stage is always a compromise between signal handling and
desensitization, the wide-band agc starting point can be influenced by Bus (2 bits) e.g. setting
starting points at 4 or 8 or 12 or 16 mVrms at mixer inputs.
By Bus the FM receiver can be set via this agc in local-mode at standard applications (USA/
Europe/Japan), giving a gain reduction (about 12dB in Fig.44 application) by 0.5mA current in
the pin-diodes. The local-mode can be used for search tuning; tuning to the strongest
stations only.
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In the application, Fig.44, the sensitivity is typ Va= 1.4µV for S/N= 26dB (∆f=±22.5kHz),
at input signal (from a 75 Ohm antenna) with 50µsec de-emphasis.
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AN
. Keyed AGC
RF gain control has to be done only if necessary. To that end the amount of agc can be
limited with the help of the narrow-band IF-level signal, see Fig.18.
- Influenced by a strong transmitter, the
weak signal is reduced till level voltage
is
decreased
to
0.95
Volt,
corresponding with about 4.5µV
antenna input-signal, dependent on
Level-DAA alignment.
- Then the wideband agc is fixed and
larger signals cannot drive the weak
signal further into noise.
- Although large signals can give
incidentally interferences (in case their
frequency difference equals IF) the
keyed agc can be preferred to
maintain
sensitivity
(minimum
desensitization by large signals).
- The keyed agc function can be
switched on/off by I2C Bus in case a
better Inter-modulation free dynamic
Fig. 18 The keyed RF-AGC at FM
range has performance priority.
Two AGC time constants are to be
connected at pins 36 and 37 respectively.
With one time constant, C=1µF for the wideband AGC at pin 36, the attack and decay timeconstants are about 5ms. With at pin 37 a C=1µF added (for keyed-AGC), the attack time is 90ms,
decay constant is 5ms.
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4.2.2 IF and demodulation
* The mixer output signal (pins 18/19) passes a tuned 10.7 MHz LC-filter and a ceramic SFE
filter, common used with AM-1st-IF, with bandwidth of 180kHz, and enters the IF at pins 14/15. To
minimise coupling with other functions the IF has its own supply pin (pin 13).
In the IF of the NICE/PACS system only two relatively wide ceramic filters are required. This
because the rest of the IF selectivity is carried out by an integrated adaptive filter section, which
centre frequency is adjustable by Bus and which bandwidth is dynamically controlled (see Fig. 19) by
circumstantial conditions. This eliminates the need for different filters for global applications.
FM-Tuner
RF & IF1
Downconverter
Soft-
clipper
Integrated
IF-filters
Limiter
ACD
PACS-system
Fig. 19 NICE / PACS IF2
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22
Demodulator
MPX- out
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With Circumstantial Controlled Selectivity
AN
* The first IF selectivity at
10.7MHz has two ceramic filters
(180kHz bandwidth, so no
special attention on group delay
character
or
on
centre
frequency tolerance is required).
They realize S200= >45dB, see
curve Fig. 20.
• An IF amplifier is used
between the filters, having a
high linearity and dynamic
range. The IF-amplifier (pin 14
to pin 12) has a Gain of 18dB
and >200mVpeak input for the
-1dB compression point.
At the input (pins 14/13) as well
as at the output (pins 12/11) the
impedance is matched for
Fig. 20 FM IF1 Selectivity
ceramic filters (330 Ohm).
Noise figure 10dB at 330 Ohm source; third order intermodulation (IP3) at 116dBµV.
• The 2nd FM-Mixer. To go for integrated dynamically controlled IF the 10.7MHz IF1 has been
converted to 450kHz in a mixer. To keep the power dissipation and chip area acceptable, the
IF2 frequency should not be much higher than the required filter bandwidth. 450kHz is chosen
for convenience; conversion signals are already available for AM. With mixer2-input at pins
8/10 and with the output direct coupled to the integrated FM-IF2 (via a soft clipper, to avoid
overload of the integrated filters) the IF2 performance will be defined from pins 8/10 onwards.
• The IF2 selectivity has been build with integrated time-continuous adaptive filters, whose
instantaneous bandwidth is determined by all relevant system parameters. The improved of
the filter structure and its bandwidth control algorithm deliver higher dynamic selectivity,
improved sensitivity and low THD at high frequency deviation without any audible artefacts.
The automatic alignment of the centre
frequency eliminates IF channel tolerances
(<1.3kHz using a 7 bits DAC) and makes it
suitable for global receiver applications. The
integrated filter of transconductance resonance
amplifier topology (see Fig. 21) gives the
possibility to adjust the centre frequency and
the bandwidth by currents. This because the
centre frequency is determined by Ft=Gt/2πC
and the bandwidth B=1/πRdC, with Rd=R/(1RGb). Complex realisation of the 4th order filter
takes care for image rejection, optimum groupdelay characteristic and symmetrical filter
Fig.21 Resonance Amplifier model
behaviour.
Fig. 22 gives an idea of the static selectivity of the integrated filter only.
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AN
S200
S100
IF2 Selectivity
Selectivity (dB)
80
70
60
50
40
30
20
10
0
25
60
90
130
-3dB Bandwidth (kHz)
155
Gain alignment with a 4 bits DAC takes
care for a constant gain during bandwidth
control (error typ ±0.35dB over the total
bandwidth in dynamic mode).
•
A limiter creates the quality signal
to drive the FM-demodulator and delivers
the signal for fieldstrength level detection.
The limiter starts (-3dB) at V8-10= 4.5µV.
AM-suppression over a signal range of V810 = 0.5mV to 300mV is 45dB.
Fig. 22 FM_IF2 selectivity = f(Bandwidth)
• The Quadrature Demodulator
has an integrated resonator circuit, matched to and aligned with the filter. The demodulator
circuit, shown in Fig. 23, utilizes
modulation feedback to reduce
distortion. The combination of the
above gives a superior receiver with
THD
performance
at
high
modulation
depths:
overall
(including 2 ceramic filters), 0.2% at
±75kHz deviation. The detector
output (FM-RDS_MPX) at pin 57 is
230mVrms at ±22.5kHz deviation.
Detector output signal is also
available via a mute function. Pin 58
Fig. 23 FM Demodulator circuit
delivers 230mVrms; bandwidth
>200kHz at Rload >20kOhm at pin
58. Mute depth 80dB; attack- and decay- times are 1ms., in case the mute time constant is
set by C=6.8nF at pin 55.
The IF and limiter signal
and noise behaviour from
pins 8/10 onwards are
shown in Fig.24.
4.2.3 IF Bandwidth Control
The IF2 bandwidth can be
fixed by Bus to narrow-/
medium-/ wide- bandwidth
(60/ 90/ 130 kHz) or to a
dynamic control (25 to
155kHz). The block diagram
of the bandwidth control
circuit is shown in Fig. 25.
The dynamic bandwidth
control operates different in
Fig.24 FM-IF Signal and Noise behaviour
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two areas: the area where R.F. input signal Va > 4.5µV, and the area below that Va level. The
definition of mentioned Va depends on the level-DAA alignment.
• At input signals Va >4.5µV, the bandwidth is controlled to reduce adjacent channel interferences.
The Adjacent Channel Detector delivers the information for the bandwidth control.
Fig. 25 The FM IF Bandwidth Control Circuit.
The adjacent channel detector (ACD) measures the ultrasonic residues in the demodulator
output in the 100kHz to 250kHz range, beat-signals caused by adjacent channel breakthrough.
The rectified signal, available at pin 65, will be compared with a dynamic threshold and, when
the threshold is exceeded, IF2 filter bandwidth is reduced in such a way that the dynamic
selectivity is constant. The sensitivity of the ACD can be influenced setting the threshold with the
voltage at pin 75. The nominal voltage of 380mV at pin 75 can be adjusted (if needed) with a
resistor to ground or one to +5V.
The example in Fig. 26a shows the bandwidth (to be monitored by a dc voltage at pin 65, ranging
from 2.2 to 0.3 Volt at IF2-bandwidth from 25 to 155kHz).
Care has been taken for control currents creating fast attack and slow decay to obtain graceful
bandwidth control. The capacitor at pin 69 influences these time constants.
• The Modulation Detector.
At low, noisy, RF levels and high modulation,
NICE PACS TEA6848H
IF2-Bandwidth(kHz)
BW (kHz)
the demodulator output generates its own
150
ultrasonic residues. This can cause a latch-up
140
130
effect in the bandwidth control circuit. To
120
110
prevent this, the threshold level for the ACD100
90
80
sensitivity consists of, next to a fixed setting, a
70
60
variable setting, controlled by modulation,
50
40
which, at high- (or over-) modulation, reduces
30
20
10
the sensitivity of the ACD-loop. To take care for
0
operation at ‘(stereo-)modulation frequencies’
0
0.5
1
1.5
2
2.5
3
Voltage at pin 65 (V)
only, the MPX signal passes external high- and
low-pass filters before entering the modulation
Fig. 26a. Bandwidth versus voltage at pin 65
detector input at pin 60 (Ri =40kOhm). At the
output, pin 68, a time constant of about 0.4 ms
makes the modulation detector an ‘average’ detector. At input signals Va <4.5µV (or the level
fixed by Level-DAA alignment), the deviation dependent threshold becomes not sufficient
anymore to avoid latch-up effects. Therefore the ACD-loop will be set out of order (setting
threshold voltage to maximum).
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• Threshold Extension: An extra benefit of
this control loop is that at low modulationdeviation the IF noise bandwidth can be
reduced at low RF levels, when permitted
(see Fig. 26b) by the modulation detector.
This has been done for input signals Va
<4.5µV.
As a consequence, the
demodulator threshold is lowered and the
effective receiver sensitivity is improved
(Fig. 27).
On/off switching of the threshold extension
can be done by Bus.
TEA6848H ThresholdExtension
IF2 Bandwidth (kHz)
160
150
140
130
120
110
100
90
80
70
60
50
40
30
20
0
10
20
30
40
DeviationDf(kHz)
Fig. 26b IF-Bandwidth = f (freq. deviation)
• The Frequency-offset Detector will
reduce the bandwidth of the IF-filter
when the detected frequency offset in
the demodulator is too large. This
avoids a kind of plop effect that could
occur under certain input signal
conditions. For example when tuned
to a (very) weak desired signal with a
strong undesired neighboring signal
at 100kHz with relatively high
deviation, the bandwidth will switch
continuously from maximum to
minimum and vice versa. To avoid
the resulting audible effect the
frequency-offset
detector
is
implemented at the demodulator
output of the TEA6848H. To measure
the offset, a large time constant has
been used (with C=10µF at pin 79,
commonly used with AM IF 2 AGC
Fig. 27 Threshold Extension at FM
amplifier). To cope with spread of the
demodulator, the frequency offset
must be aligned (a 4 bits DAC for matching within ±1.5kHz, typical).
Notes:
Bus switching to the freq.-offset alignment (with bit 4 of data byte 5) will set the offset detector
voltage to pin 62, where it can be monitored for minimum voltage. At dynamic control to a
bandwidth < 42kHz, pin 62 gives indication (a flag) which could be used in special radiosystem applications where large delay due to low bandwidths is not acceptable.
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4.2.4 Search -stop information
FM tuning steps of 100, 50, 25, 20 or 10kHz can be
chosen by Bus (reference frequency setting). With a
reference frequency of 100kHz and the VCO divided
by 2, the tuning step is 50kHz.
Station quality is detected on 2 items: fieldstrength
and IF-accuracy, necessary in areas where the FMband is crowded, illustrated in the figure.
Next to that this special NICE-IC detects adjacent
channels to control selectivity as explained in
previous chapter. Once stopped, a station far from
the wanted one will be neglected by IF-counting; a
station close to the wanted one will influence the IFbandwidth due to the ACD, thereby reducing its fieldstrength delivery.
a. Fieldstrength:
The IF limiter delivers a well defined fieldstrength-dependent DC-level information, analogue
available at pin 70, to be used in the audio signal processor
. for soft mute at weak signal handling
. for stereo blend
. for signal dependent response (high cut control etc.).
In a signal range V 8-10=10µV to 1 Volt the level-detector delivers 1 to 4 Volt dc.
Special attention has been paid to the temperature
behaviour of the level amplifier. Over the operating
temperature range, the level-change is just as much
as ±3dB RF-signal change.
Search stop sensitivity can be adapted with the help of
the Level DAA such to cope with spread on
fieldstrength level information.
For production starting point as well as the slope of the
level detector need alignments.
(Note that level depending parameters, like keyed AGC
and Threshold Extension, are influenced).
Example:
* FM level-start: The level-detector output is set to
940mV at a RF input level of 4.5µV. (Note: 940mV at
Level output signal
FM is the switch-off level of keyed AGC and start of
threshold extension, e.g. if selected by Bus keyed
AGC=on and threshold extension=on below 940mV).
* The level-slope is aligned in such a way that the
difference in level-detector output between RF levels of 20 and 200µV is 800mV with the level-start
value found in the first alignment. These alignments cannot be seen separately.
More about alignments in Appendix 2.
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b. IF-Counting:
Next to the fieldstrength level, the exactness of the IF frequency is counted for stop
information. To this end the TEA6848H has an 8-bit-IF-counter with a programmable counting
window of 2 or 20ms. The counter counts the output frequency of the limiter amplifier which is
divided in a programmable divider, the pre-scaler. For FM the dividing ratio N can be set to 10
or 40. The content of the counter can be read out via the I2C-Bus. It is not necessary to read
out the full value of the IF-frequency to get information about correct tuning. It is sufficient to
use only the 8 least significant bits. The counter resolution is given by the counting time and
the dividing factor of the pre-scaler.
The number of counted cycle’s n, counted during the counting window tc (2 or 20ms) is
Fif
n = ---- . tc
N
where N is the dividing factor of the pre-scaler and Fif is the output frequency of the IF amplifier.
The resolution ∆Fif of the system is the frequency difference, which corresponds to the least
significant bit of the counter (LSB).
N
∆Fif = ---tc
Next table gives an overview of the possible combinations of read back values and the
corresponding resolutions; not only for FM in different markets, but for weather-radio and AM
as well.
TABLE 1 IF counter
Application
FM-standard/-east/-weatherband
read out and IF count resolution
Tc
IF
Read out Resolution
prescaler value
(ms)
(N)
(Hex)
(kHz)
2
10
5A
5
FM-standard/-east/-weatherband
2
40
16
20
FM-standard/-east/-weatherband
20
10
84
0.50
FM-standard/-east/-weatherband
20
40
E1
2
LW / MW / SW
2
1
84
0.5
LW / MW / SW
20
1
28
0.05
The counter sensitivity voltage: 2µV antenna signal for a 30% modulated FM signal.
Note that the counter is reset after each Bus transmission, taking care that the count-info is
correct from reset onwards.
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4.3 Oscillators
4.3.1 VCO
The VCO, tunable from 159.9 to 248.2MHz, serves FM and AM and Weather-band
application on global scale. At FM the mixer is driven with ’high’ injection oscillator for
Europe/ USA and Weather band, where in Japan and Eastern Europe FM band the mixer is
driven with a ‘low’ injection oscillator.
Divider
VCO
Tuning Voltage
FM
Europe/USA
87.5 to 108MHz
2
196.4 to 237.4MHz
2.6 to 5.5V
Japan/Far East
76 to 91MHz
3
195.9 to 240.9MHz
2.5 to 5.8V
Eastern Europe(OIRT) 64 to 74MHz
3
159.9 to 189.9MHz
1.1 to 2.7V
Weather-band
AM
LW - MW
SW 49m
162.4 to 162.55 MHz
1
173.1 to 173.25 MHz
1.22V
144 to 1710kHz
20
216.88 to 248.2MHz
3.9 to 6.5V
5.73 to 6.295MHz
10
164.3 to 169.95MHz
0.8 to
1.06V
FM
AM
As the VCO at FM defines the final S/N ratio at full limited FM-channel, care has been taken
to the VCO Carrier to Noise Ratio. Therefor a high quality VCO-coil (Q0=130) has been used.
For a required (S+N)/N = 65dB, defined at ∆f = ±22.5kHz modulation at 50µsec de-emphasis,
the CNR at 10kHz distance has to be 101dBc/√Hz for the oscillator signal. The oscillator
signal is obtained from the VCO via a :2 divider. A VCO with, at 200MHz, a CNR of
97dBc/√Hz at 10kHz distance.
The target for AM is based on avoiding reciprocal mixing by interfering neighbouring
(∆=10kHz) signals. With a neighbouring signal 75dB attenuated and with 5kHz IF bandwidth
the oscillator signal CNR target at 10kHz distance becomes 75 + log(5000) = 112dBc/√Hz,
delivered from a VCO via 10 times divider (at SW).
So for the VCO 112-20 = 92dBc/√Hz is good enough for AM.
4.3.2 X-tal oscillator
The X-tal Oscillator (pins 71-73, with pin 72 for x-tal osc. ground)) operates at 20.5MHz,
having low interferences and using no additional components. The oscillator is fully balanced
with respect to the crystal pins, such to have low cross-talk towards sensitive receiver pins.
The current of the sinusoidal signal at the crystal pins is well defined by internal control to
obtain low power / low harmonics operation. The 5th harmonic at 102.5MHz is >70dB down.
A special circuit takes care for start-up of the oscillator using start-up current of 9mA and an
operating current 1.5mA.
The oscillator is used for
§
AM and FM second conversion,
§
synthesizer reference frequencies,
§
clock frequency generation for the sequential RDS-update circuit,
§
time-base for IF-Counter,
§
reference frequency of 75.368kHz for Car Audio Signal Processors (CASP);
the last with signal level 100mV from 50kOhm source at pin 45.
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Required is a crystal with the following specification:
. motional resistance (at start of operating): < 50 Ohm
. shunt capacitance:
< 3 pF
. load capacitance:
10 pF
. motional capacitance:
9 fF
resulting in ± 34 ppm pulling for ±1.25 pF capacitance variation.
Together with the other requirements on
Accuracy:
± 20 ppm
Ageing:
± 5 ppm
Temperature stability:
± 30 ppm,
the application of NICE with this x-tal oscillator permits a worst case max. deviation of
±1.8kHz (which is ±89ppm) from the 20.5MHz oscillator frequency.
4.4. Tuning System
The adaptive PLL tuning system combines low phase noise and reference spurious
breakthrough with a fast tuning response. The internal RDS, sequential, control circuit coordinates the tuning operation. The tuning algorithm combined with the mute circuit provides
inaudible signal quality checks on FM. The crystal oscillator generates all the necessary
reference signals for the tuning operation and frequency conversions.
Functional information on the tuning system is shown in Fig. 28.
4.4.1 Digital Automatic Alignment
In the application described, the design of the tuned input circuit with capacitance diode
BB814 is, in combination with VCO tank-circuit, containing a diode BB156, optimised for low
padding deviation by digital automatic alignment. Usually three alignments are necessary and
sufficient for a good tracking performance. (Padding max. 400kHz, where the Q of the RF
circuit is about 25), to which end the tuning voltage of the oscillator is converted in the DAA to
a controlled alignment voltage for the FM antenna circuit.
After having the phase lock loop of the NICE synthesiser locked to a new tuning position, the
analogue tuning voltage at the loop filter is used as reference for RF-tuning.
Starting with a certain input level at the selected input frequency, the level detector output is
measured and stored, where after the DAA value is increased by one. This sequence is
repeated for a certain time and from all measured values the maximum value is calculated.
When this value is stable for some measurements, the centre is calculated and the
corresponding DAA value is stored in the memory (EEPROM). This can be done for lower
limit- / upper limit- and mid-frequency of the frequency band.
A NICE alignment recipe ”Autonice” is available on request.
As the VCO charge pump may not be loaded, the DAA buffer input (pin 40) has very high
impedance (input current <10nA).
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Fig. 28 Tuning System
The input voltage of the DAA can be multiplied by 0.25 up to 1.75 by the 7 Bits setting of the
conversion gain.
The output voltage (>0.5 to <8 Volt) at pin 38 has a low noise level: <100µV, measured acc.
dB(A); ripple rejection is >50dB. The settling time of the DAA output at max. step is <30µsec
at 270pF load at pin 38.
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Next to the minimum leakage currents, low-noise and high ripple rejection, the temperature
dependency is an item. As the silicon varactor diode in the VCO is temperature dependent, a
compensating diode has been connected at pin 39. This diode is not on chip, such to have its
temperature behaviour the same as that of the varactor diode. Temperature drift over -40°C <
Tamb < 85°C is < ±8mV. The output voltage at pin 38 of the antenna DAA is
V38 = [2 x ( 0.75 x (n/128) + 0.25) x (V40 + V39)] – V39, where n=0 to127,
in which V40 is the DAA input voltage and V39 depends on the diode connected at pin 39 (V39
is about 0.46 Volt in case a diode has been used).
4.4.2 The RDS updating (Sequential circuit)
To provide best reception quality, a
control is used in car radio to
check for alternative frequencies
with equal programming; such with
the help of a system like RDS
(Radio Data System). This usually
can cause audible breaks in the
main channel received, as the
audio has to be muted for the
moment while the receiver is
tuning to other frequencies. Gaps
in the audio signal may be
perceived if the muting time is not
short enough. In practice, with
actual audio signals, muting times
below 5ms. with gentle slopes of
1ms are inaudible, see Fig.29.
Fig. 29 Inaudible mute behaviour
To achieve FM quality signal
checks of 5ms, the tuning times
have to be reduced to below 1ms. and the frequency jumps have to be made independent of
the (slow) Bus communication times. The first requirement has to be accomplished by the
tuning system, whereas the latter was solved by local intelligence in the form of a sequential
circuit that controls tuning operations during quality checks.
This sequential circuit responds on an AF-label in the frequency word (signifying a quality
check request) by
a. muting the audio with a 1ms slope
b. jumping the PLL to another frequency in less than 1ms.
c. sensing the quality of the new signal with the level- and IF-sensors in 2ms.
d. writing this information into latches
e. jumping back to the main frequency
f. de-muting the audio with the mentioned 1ms slope.
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Fig. 30 RDS Alternative Frequency check
A complete cycle (see Fig. 30):
• Starts with a Bus command to go to an alternative frequency;
• Next the AF-signal will be muted by reducing the audio-signal linear in 1ms.;
• Then the tuning voltage jumps due to an in between PLL-load command;
• After 1ms the new tuning position is reached and a quality check (level-info) can be done.
For this the counter-period is automatically switched to 2ms. The prescaler can be
chosen freely.
•
Then a PLL-load command can start Vtune to jump back to the original main-channel as
asked by Bus-data.
For application with audio processors (like CASP or CDSP) sample and hold info is available
from pins 53 and 54 respectively (Sample- like the ‘quality check’ and Hold-info like
‘mute/freeze’ in Fig. 30). The latched info can be read via the I2C Bus at any time with simple
software (with minimum load of the µC). Attention has to be paid to the timing of the maincommand and the fact that during AF-update no other Bus transmissions to the receiver are
permitted then those related to frequency and DAA-level. The time constant for mute
behaviour at RDS AF update is defined by the capacitor at pin 55.
4.4.3 Adaptive Synthesizer
The tuning system uses a PLL synthesizer, supplied via pin 44 (analogue 8.5 Volt) and at
pins 46/47 (digital 5 Volt)
The VCO frequency is divided in a programmable divider, controlled by the I2C Bus.
The Bus data define the divider ratio of the divider, N, which determines the RF at which the
system is tuned.
The divider ratio is
Fvco
.N = ----Fref
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AN
where Fvco= M * Fosc
= M * (Ftuned ± Fif ) with Fif= 10.7 MHz both for FM and AM (+
for high side and – for low side injection).
M is the divider ratio of the divider N1, which sets the oscillator frequency for the RF-Mixer.
In next table an overview is given for divider ratio calculation in different applications.
TABLE 2 Frequency Divider Ratio range
Application
Fif
Fref
M
Ftune
N
(MHz)
(kHz)
FM-standard
10.70
100
2
87.5-108
1964-2374
FM-Japan
10.70
100
3
76-91
1959-2409
FM-east (OIRT)
10.70
20
3
64-74
7995-9495
FM-weatherband
10.70
25
1
162.4 - 162.55
6924-6930
SW
10.70
10
10
5.85-9.99
16550-20690
LW
10.70
20
20
0.144-0.288
10844-10988
MW
10.70
20
20
0.53-1.71
11230-12410
The divider-output is connected to a phase detector, and the divided frequency is compared
with the reference frequency Fref. The output of the phase detector drives, via a charge
pump circuit (output pin 42/43), the loop filter (between pin 42/43 and 40), which in turn
delivers the VCO tuning voltage (at pin 40).
Fig. 31 Adaptive Synthesizer currents to different nodes of the loop filter.
Spectrum purity, small tuning steps and fast settling times are contradictory requirements for
the PLL synthesizer. With the adaptive PLL solution of Fig. 31 two loops work in parallel with
a smooth take-over to guarantee inaudibility. The phase detector outputs of the Loop-2 are
low-pass filtered before the high current charge pump CP2; CP2 is active only during tuning.
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AN
Some information in more detail:
The low-pass filters give a smooth transition into a well defined dead-zone when lock is being
achieved. The Loop-1 phase detector has no dead zone and directly steers the low current
charge pump CP1.
Good centring of the two charge-pump outputs (by careful symmetrical design etc) is
essential for low noise in lock. Additional freedom for optimisation of loop parameters is
obtained using two separate charge pump outputs, and by applying the charge pump.
During frequency jumps both CP1 and CP2 are active. The loop filter zero-gain frequency is
[1/(2π.Rb.Ca)] and lies at a high frequency, resulting in stability and fast tuning. After the
frequency jump only CP1 (to pin 43) is active. The loop filter zero-gain moves, without
switching of loop filter components, to a lower frequency [1/(2π.(Ra+Rb).Ca)], increasing the
in-lock phase margin. Furthermore, when the loop is in-lock, an extra pole is introduced
[1/(2π.Rc.Cc)] increasing the 100kHz reference breakthrough suppression by about 20dB.
To obtain a fast tuning step the charge pump CP2 (pin 42) can deliver 3mA current to the
loop filter. After tuning the active charge pump CP1 delivers 130µA at FM to 1mA at AM
(Weather-band and East-Europe FM at 300µA). The pre-set time for FM tuning to bandlimits
is within 1 ms (using 100kHz reference-freq. in the synthesizer, like in RDS AF-updating).
The loop-filter as shown in application, Fig.44, is optimum for fast PLL tuning (< 1ms for a
tuning-step 88 to 108MHz).
* The reference frequency, delivered by the 20.5MHz crystal oscillator, can be set by Bus.
For fast AF-updating at RDS, PLL control is on chip.
4.5 I2C-Bus control
For details: see APPENDIX 1.
The basic functions and the specification of this Bus system are described in a special
Philips brochure: "The I2C-Bus and how to use it" (December 1998, document order
number 9398 393 40011).
The I2C-Bus, with data and clock lines at pins 63/64, is structured as shown in Fig. 32.
The Bus communication starts with a "start"-signal given by the system controller. The first
transmitted byte is the address byte (byte 0). The following bytes (1 to 8) are used to transmit
information to the IC or to receive information from the IC. When the Bus communication is
used partially the transmission must be ended by a stop condition. In this case the remaining
bytes will contain the old information.
The complete information to set the IC TEA6848H consists of the address byte and 8 data
bytes.
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AN
Byte
0
1
2
3
4
5
Bits
7+R/W
1
7
8
1
7
1
3
1
3
1
2
1
1
Command
Address
AFmute
PLLFreq.
PLLFreq.
Mute
Ant
DAA
Counttime
Ref.
Freq.
IF-Prescaler
Band
Keyed
AGC
AGC
AM/FM
AM
Soft
mut
e
Lo
/
Dx
ms
kHz
2
10
10
20
20
40
FM
6
1
2
SoftIF2
ware band
flag width
mV
kHz
150 / 16
275 / 12
Dyna
mic
130
25
FMJapan
FMOIRT
WX
400 / 8
90
50
SW
525 / 4
60
100
MW
/LWmono
MW
/LWstereo
7
8
5
3
1
7
4
4
Level
DAA
start
Level
DAA
slope
FMthres
hold
IF2
centre
DAA
FMDemod
Offset
DAA
IF2
filter
Gain
DAA
Fig. 32 TEA6848H Bus-structure
The address byte is 1100001R/W, with a 2nd address 1100000 R/W, to be selected by connecting
pin 45 via 68kOhm to ground. For Read/Write: logic 1=read and logic 0 =write.
TABLE 3 Frequency Band Setting
Application
Bit 1
Bit 2
AM/FM
Bnd1
FM-standard
0
0
FM-Japan
0
1
FM-east (OIRT)
0
0
FM-weatherband
0
1
SW-mono
1
0
SW-stereo
1
1
LW/MW-mono
1
0
LW/MW-stereo
1
1
Bit 3
Bnd2
0
0
1
1
0
0
1
1
VCO-divider
2
3
3
1
10
10
20
20
Charge Pump
Current
130µA + 3mA
130µA + 3mA
1mA
300µA
1mA
1mA
1mA
1mA
4.6 Supply
The main supply Vcc1=8.5 Volt, pin 61, which has to deliver typical 65mA at FM and 50 mA
at AM. In addition 5 Volt supply is needed at pins 46/47 for digital functions and at pin 59 for
analogue functions with 33 to 46mA current consumption, application dependent.
The external voltages create internal reference voltages and currents, taking care for the
required stabilization and temperature behaviour.
Notes :
. Switching performance in this report refers to switching both 8.5 and 5 Volt supplies
simultaneously.
. Care has to be taken for a good ripple rejection of the VCO-supply (pins 51/48).
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AN
5. LAYOUT GUIDELINES
Application of the TEA 6848H simplifies the PCB design of a digital tuned AM/FM receiver
dramatically. With a complete tuner LSI for low design costs, special measures have been
taken during the IC design for good internal separation of the analogue-receiver and digitaltuning parts in order to minimize interferences.
Because of these measures, the PCB given in this Application Note (see Fig. 44) is rather
simple and a large list of layout tips is not necessary. However, being a radio application in
which the gain in several parts of the receiver is considerably high and where RF and
oscillator signals should not enter the final IF stages etc. still some attention has to be spend
on the PCB design. When the two-sided layout, given in this application note, is used,
problems are not to be expected (see Appendix 3 for a two-sided PCB, version TEA 6848H).
Some layout hints are:
VCO:
The VCO coil needs to be put close to the IC pins, also the grounding of the VCO varactor
diode (BB156) via the 270pF capacitor (C63); and the grounding of VCO coil and capacitor
C63 needs to be done directly to the VCO-GND pin 48.
FM-Mixer:
The connections of the FM input transformer to the mixer pins should have the same length.
The first FM PIN diode (D3) needs to be put close to the antenna connection to prevent large
signals from entering the PCB.
AM:
Using an FM intrusion trap (L3+C4), it needs to be placed close to the antenna connection
and its grounding.
Reference crystal:
The 20.5MHz crystal can best be put close to the IC-pins.
I2C-Bus tracks:
To suppress I2C-Bus interferences, 330 Ohm resistors are placed in the SCL and SDA lines.
It is important to keep these tracks away from the VCO coil (and its tracks connecting it to
the IC pins). An RC- filter in the I2C-Bus outputs of the embedded micro-controller has been
used to round-off the I2C-Bus pulses a little.
Coils:
Since the coils need no mechanical alignment you don't need physical holes. It is advised
however to have holes in the copper pattern below FM-coils L10, L14 and L15. See our demo
module-board. (A solid copper plain below these three coil- functions will act as a 'shortcircuited turn', so will deteriorate the coil-performance!).
SMT-components:
In order to minimise surface area, change over the coils to SMT types. Companies TOKO and
SAGAMI know the (leaded) coils and their specs needed for the Nice_Pacs tuner, so they
should be able to advise what's the best SMT replacement for them.
To help with this process please find below a short description of the used coils with some background info.
1
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TEA 6848H A NICE RADIO
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AN
FM aerial input; L12 & L18:
These coils are in series with the antenna input; therefor any series resistance of this coil will
have a negative effect on the FM sensitivity. In our current NICE tuner modules we use Murata
coils, Q0 >85, Fres >300MHz, Rseries <0.7 Ohm.
When a cheaper coil is desired an increase of the series resistance Rseries <2 Ohm giving a Q0
>30 could still be usable. The coil's resonance frequency however should be well above the FM
band so remain Fres >300MHz !
Note that the PIN diode decoupling capacitors of 1nF (C37, C38) need to be of the NP0 type
(0805 SMD’s have the best quality at 100MHz)
Tuned FM-RF coil (L14):
To have a good selectivity this coil needs to have a Q0 of at least 60.
With our tuned coil and with it's tap chosen as we have the SLINE coil is about 20 - 30nH. The
Q of this coil is not very critical as long as the image suppression of the front -end is about
30dB. (Please note that the rest of the needed image suppression is made by the I-Q mixer
system inside the chip!).
FM RF coil (L10)
The output impedance of this coil is about 200 Ohm (to match mixer input) so it doesn't need to
have a very high Q0. Loaded Q of about 20.
VCO coil (L15)
The coil used for the VCO needs to have a good Q of >120 to guarantee a good carrier to noise
ratio !! If required also an air-coil can be used (most air coils have a Q0 >300 !). Use always NP0
type capacitors in the VCO circuit.
FM mixer (L1)
This is a normal 10.7MHz mixer transformer with a Q0=50-70
AM mixer (L4)
This is a normal 450kHz mixer transformer with a Q0=50-70.
20.5MHz Xtal specification
• motional resistance (at start of operating):
< 50 Ohm
• shunt capacitance:
< 3 pF
• load capacitance:
10 pF
• motional capacitance:
9 fF
• Accuracy:
± 20 ppm
• Ageing:
± 5 ppm
• Temperature stability:
± 30 ppm
6. APPLICATION.
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TEA 6848H A NICE RADIO
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AN
6.1. Application AM
For the Module TEA6848H (Fig.43, application acc. Fig. 44) the Gain distribution in the AM-
channel is as shown in Fig. 33.
TEA 6848H AM –MW signal channel
Dummy- Input
Aerial selectivity
Measuring
Point:
Equivalent
Noise
Voltage
S/N= 26dB
at Vi =
Relative
Levels
Stage Gain
_
Preampl.
_
LowPass
Filter
_
Mixer
1
1st IF,
LC+SFE
_
Mixer
2
2nd IF,
LC+SFP
_
Result:
|
0
*
|
1
*
|
2
*
|
3
*
|
4
*
|
5
*
|
6
*
|
7
|
|
|
|
|
|
|
|
10
0.8
5.5
|
47
|
|
|
|
|
0
-20
5
4.5
|
|
-20
|
|
|
25
|
|
|
-0.5
|
|
14
|
|
39
|
|
|
18
|
-16.5
|
|
|
10
6
nV/√Hz
|
45
µV
|
6
Fig.33 AM Gain Distribution
1
|
IF 2
Ampl. /
Det.
-5
11
dB
|
|
dB
TEA 6848H A NICE RADIO
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With Circumstantial Controlled Selectivity
AN
Fig. 34 shows MW Signal and Noise behaviour as a function of fieldstrength with selectivity acc. to
Fig.6, 7, 8 and 9; the THD (Total Harmonic Distortion) behaviour is given too.
The noise limited Sensitivity: S/N=26dB at standard modulation
Dummy-antenna
S/N = 26 dB
at loaded generator at relative Va =
15 to 80pF
55µV
15 to 60pF
47µV
27 to 47pF
28µV
AM dummy aerial
50 Ohm
3.4µV
Fig. 34 AM Signal and Noise behaviour; 1=with and 2=without soft-mute
This sensitivity is constant
over the MW band. At LW
the value is higher: 70µV.
In case lower inter-station
noise is required (or lower
Figure Of Merit), one can
reduce gain, switching on
the AM soft mute function,
see curves 1 & 2 in Fig.
34.
Intermodulation:
Reception of sum and difference frequencies due to 2 strong signals (F2 and F3); combinations of it
cause IP3, cross-modulation related 3rd order non-linearity. The Intermodulation Points: see Fig.35,
with IP2 caused by 600 and 800kHz (F2 and F3) at 1400kHz (F1) tuning and IP3 caused by 1040
and 1090kHz (F2 and F3) at 990kHz (F1) tuning; as a function of the input voltage at the dummy
aerial.
Fig. 35 AM Intermodulation characteristic
1
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TEA 6848H A NICE RADIO
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AN
6.2. Application FM.
For the given application the FM-Gain Distribution is given below:
Dummy Input Trans-Aerial selec former
_ Mixer
1st
1
IF
I&Q LC+SFE
_
*
|
0
*
|
1
*
|
2
*
|
3
|
6
|
|
|
|
1.3
|
|
|
-tivity
Measuring
Point:
Noise
Figure
S/N= 26dB
at Vi =
3
-1dB Compression
|
|
|
|
|
Relative
Levels
Stage Gain
|
1
|
|
|
|
-0.5
_
IF2- Limiter
Filter
( PACS)
*
|
5
*
|
6
*
|
7
|
|
|
|
dB
|
|
|
|
|
10
|
µV
|
>400
|
|
mVpp
|
|
|
|
116
|
32.5
33
Mixer Ampl.Limite
2
r
I&Q
*
|
4
|
|
117
|
0
_
9.5
>70
IP3
IF- 2nd
ampl IF
SFE
|
-16.5
|
16
|
|
dBµVrms
|
34
|
18
-4.5
|
29.5
|
|
|
39.7
|
10.2
3.3
|
43
|
|
dB
0
Fig. 36 FM Gain Distribution
Fig. 37 shows Signal, Noise and SINAD characteristics / Total Harmonic distortion and
AM-Suppression and level-information in this application.
Fig.37 FM Signal, Noise and Distortion
1
41
dB
TEA 6848H A NICE RADIO
APPLICATION
NOTE
With Circumstantial Controlled Selectivity
AN
R_antenna
75 Ohm
-3dB Limiting
at Va:
S/N = 26 dB
at Va:
1µV
1.4µV
75 Ω / 6dB FM-dummy aerial
6.3 Global Applications :
1. Europe: Standard application;
FM-band 87.5 to 108MHz, channel grid 100kHz, de-emphasis 50µsec
AM- LW 144 to 288kHz
MW 522 to 1620kHz, channel grid 9kHz
2. USA
FM-Band 87.9 to 107.9MHz, channel grid 200kHz, de-emphasis 75µsec
AM-Band 530 to 1710kHz, channel grid 10kHz.
Some items, characteristic for USA applications, influencing choice of components:
• De-emphasis-C's to be matched to USA situation (75µs instead of 50µs); to be done by
software setting in the audio processor (CDSP or CASP).
• Narrow spaced FM-broadcast transmitters located at one position (city) ask special
attention for certain intermodulation products suppression and AM-interference
breakthrough (so called ‘FM intrusion’; see note below).
Note: To reduce FM intrusion e.g. for USA-markets, where two FM-transmitters F1 and F2
can have frequency difference |F1-F2| which can fall inside an AM-band, a special filter can
be designed to block the FM frequencies.
With such a filter included, 2 FM-signals having 800kHz freq. offset, need to deliver 450mV
aerial input level, before they give, at 800kHz AM-tuning, an audio-interference of 20dB below
standard (30% modulated) AM a.f. output.
Other signals to be attenuated at the AM-input are mains-interferences from high-tension
wires and ultra-sonor signals (used at deep-sea research). These signals are suppressed by
coil L18 at the input to ground, which improves the 50/60Hz suppression at some loss of
sensitivity at Long Wave.
6.4. Optional applications:
Option 1. AM - SW 49m reception.
Compared to the given MW/LW application, the main difference is that after the rf-prestage the signal passes a low-pass filter which can be for LW/MW/SW-49m a 5th order
filter, see Fig. 38 for a filter which passes short wave up to about 8MHz.
This filter gives additional 10.7MHz suppression (about 55dB by notch-filtering).
Moreover the VCO divider is set at 10, with the VCO in a range 164.3 to 169.95MHz to tune
5.73 to 6.295MHz. Tuning step 1kHz, using a synthesiser reference frequency of 10kHz.
1
42
TEA 6848H A NICE RADIO
APPLICATION
NOTE
With Circumstantial Controlled Selectivity
AN
Fig.38. AM LW/MW/SW-49m 5th order Low-Pass Filter.
Fig. 39 shows the S/N and distortion at SW 49m.
Fig. 39 AM SW - 49m Signal and noise behaviour
Option 2. System applications.
* RDS: After FM-demodulation, before entering the mute function, an MPX-RDS signal is
available, to drive the RDS-demodulator (like SAA6579). A sensitivity of 13µV (using 75Ohm
dummy) can be obtained; defined from 50% good blocks detection at RDS signal modulation
with ∆f=±2kHz.
1
43
TEA 6848H A NICE RADIO
APPLICATION
NOTE
With Circumstantial Controlled Selectivity
AN
* Weather-Band (WX-mode):
For FM Weather band applications at frequencies 162.4 to 162.55MHz,
the IC has to receive data byte 4 the bits 0,1,2 set to 011,
Then the Nice_Pacs concept provides:
1. Delivery, at pin 34, of a WX-flag for switching the rf-input from FM- to WX-band.
2. Setting of the divider N1 at N1=1 to use the VCO at (WX-IF1) = 173.1 to 173.25MHz;
3. Activation of a quadrature phase shift network to drive the quadrature mixer,
to achieve 25dB of integrated image rejection.
4. Switching the audio-amplifier at WX to 15 times higher gain
to obtain standard a.f.-output level at the small WX-deviation.
5. Switching of the integrated IF filter to minimum bandwidth, providing a selectivity of typ
23dB at ±25kHz and 10dB additional gain in front of integrated IF filter (see App.5).
More application information in Appendix 5.
* Audio Signal Processors (CASP/ CDSP)
The application of NICE with Audio signal processors CASP (TEA6880H) or CDSP
(SAA7709) gives extra functional advantages;
e.g. with CASP (see Fig. 40):
Fref (from Nice_Pacs, 75,4kHz)
Fig. 40 CASP Functional Block-diagram
• For RDS updating NICE delivers AF-sample and AF-hold output, taking care that RDS update
will be done with a mute according to timing and a behaviour which gives no audible
1
44
TEA 6848H A NICE RADIO
APPLICATION
NOTE
With Circumstantial Controlled Selectivity
AN
interferences. To that end at a start of AF-update the AF-hold freezes the status of the audio
weak-signal functions. When AF-sample arrives the audio signal processor starts detecting
signal quality and at the end AF-hold gives free the audio weak signal controls and tells the
processor that the outcome of the update check can be transferred by I2C-Bus.
• For FM-stereo decoding NICE delivers a 75.368kHz reference signal, pre-setting the
oscillator for sub-carrier regeneration. This reference signal has been used for all other
timing too.
• After a pre-cancelling of AM noise-interferences in NICE, CASP in turn cancels the rest of
spikes in the AM-audio signal. (In addition CASP delivers an AMHOLD pulse to operate the
gate into an external AM-stereo processor.)
2
2
• The I C-Bus interface in CASP has an I C-Bus output for NICE. As it is preferred to have the
NICE-Bus switched off if no NICE Bus commands are asked for (such to eliminate
interference risks) this can be done with one Bus, both for CASP and NICE.
• For weak signal management NICE delivers AM/FM fieldstrength levels, well defined in start
and slope points. Note that CASP has six signal quality detectors: noise/ fieldstrength/
multipath, and those both in average and peak detection.
Additional functions of interest in CASP are the Rear Seat Audio source-selector and a Chime
Adder circuit to sum the chime signal with audio.
1
45
TEA 6848H A NICE RADIO
APPLICATION
NOTE
With Circumstantial Controlled Selectivity
AN
I2C-BUS TEA6848H
Byte 0: Device address:
1
1
ADDR=0:
R/Wn=0:
0
0
Device address=$C0
Write mode
APPENDIX 1
0
0
ADDR
ADDR=1:
R/Wn=1:
R/Wn
Device address=$C2
Read mode
Byte 1:
AF
PCA6
PCA5
PCA4
PCA3
PCA2
AF=0:
PCA6..PCA0:
Normal operation
Upper byte PLL divider word
Byte2:
PCB7
PCB7..PCB0:
PCB6
PCB5
PCB4
Lower byte PLL divider word
PCB3
PCB2
Byte 3:
MUTE
MUTE=0:
ANT6
ANT5
Normal operation
ANT3
MUTE=1:
ANT2
ANT6..ANT0:
ANT4
AF=1:
PCA1
PCA0
AF update mode
PCB1
PCB0
ANT1
ANT0
FM MPX output muted,
Load progr. counter AM/FM
Setting of antenna DAA
Byte 4:
IFMT
IFMT=0:
REF2..REF0:
IFPR=0:
BND1..BND0:
AMFM=0:
REF2
0
1
0
1
0
1
0
1
REF2
REF1
REF0
IFPR
BND1
BND0
AMFM
IF measuring time=20ms
IFMT=1:
IF measuring time=2ms
Reference frequency
IF prescaler ratio=40
IFPR=1:
IF prescaler ratio=10
Band switch
FM
AMFM=1:
AM
Reference
VCO
REF1 REF0
BND1 BND0 AMFM Frequency band
Frequency
Div
0
0
100 kHz
0
0
0
FM standard
2
0
0
50
0
0
1
AM SW mono
10
1
0
25
0
1
0
FM Japan
3
1
0
20
0
1
1
AM SW stereo
10
0
1
10
1
0
0
FM OIRT
3
0
1
10
1
0
1
AM MW/LW mono
20
1
1
10
1
1
0
FM Weather
1
AM MW/LW
1
1
10
1
1
1
20
stereo
Byte 5:
KAGC
AGC1
KAGC=0:
AGC1..AGC0:
AMSM/FMBW=0:
LODX=0:
FLAG=0:
BW1..BW0:
AGC1
AGC0
0
0
0
1
1
0
1
1
Byte 6:
LST4
LST4..LST0:
AGC0
AMSM/
FMBW
FM keyed AGC=OFF
AM/FM wide band AGC
AM mode: AM soft mute =OFF
FM mode: standard
Local =OFF
Flag pin 21=HIGH
IF2 bandwidth setting
Start Wideband
AGC
AM
FM
150 mV
275
400
525
16 mV
12
8
4
LST3
LST2
LST1
Level starting point for level DAA
LODX
FLAG
BW1
BW0
KAGC=1:
FM keyed AGC=ON
AMSM/FMBW=1:
AM mode: AM soft mute =ON
FM mode: FM align mode, BW=minimum
Local =ON
Flag pin 21=LOW
LODX=1:
FLAG=1:
BW1
BW0
0
0
1
1
0
1
0
1
IF2
Dynamic
Wide
Medium
Narrow
LST0
LSL2
LSL2..LSL0:
1
46
Bandwidth
[kHz]
25..155
130
90
61
LSL1
LSL0
Level slope setting for level DAA
TEA 6848H A NICE RADIO
APPLICATION
NOTE
With Circumstantial Controlled Selectivity
AN
Byte 7:
TE
TE=0:
CF6..CF0:
Byte 8:
FOF3
FOF3..FOF0:
CF6
CF5
CF4
Threshold extension =OFF
Setting of IF2 centre frequency
CF3
TE=1:
CF2
FOF2
FOF1
FOF0
FGN3
Setting of Frequency offset detector
APPENDIX 2:
CF1
CF0
Threshold extension =ON
FGN2
FGN1
FGN0
FGN3..FGN0:
Setting of IF filter gain
ALIGNMENTS
The Nice_Pacs tuner concept requires a number of (software) alignments for optimum performance
to be done in the sequence as given below:
1. FM-IF2 Filter DAA alignment.
The FM-IF2 filter needs alignment for centre-frequency, gain alignment.
1.1 IF2 Centre Frequency alignment
Due to spread of the crystal frequency and spread in the integrated IF2_filter, the center frequency of
the IF filter/demodulator has to be aligned. Spread on the crystal frequency however causes spread
on both IF frequencies, 10.7MHz(IF1) and 450kHz(IF2), which is related to the tuned frequency.
It is recommended to align the centre frequency of the IF_ filter/demodulator in the middle of every
tuning band to the exact value. When the standard FM band is aligned on 97.8MHz, the worst case
frequency offset in this band is 560Hz, which adds to the alignment accuracy of 700Hz to a total
centre frequency accuracy in standard FM band of ±1.3kHz.
The centre frequency alignment goes for maximum dc-output at the level detector (pin 70).
1.2 IF2 Gain alignment
Bandwidth variations of the IF filter are wished to suppress neighboring channels or for increasing
sensitivity by threshold extension. Changing the bandwidth dynamic or fixed (by Bus) causes gain
variations in the IF filter of the TEA6848H. These gain variations will influence the field strength RF
level information and this can influence for example the level dependent weak signal handling
parameters of the audio backend processor.
A 4 bits filter gain alignment reduces the change in IF filter gain from ±5dB to ±0.35dB when the
bandwidth is changed in dynamic mode from maximum 155kHz to minimum 25kHz.
The procedure to align is as follows:
. Set at FM the IF2 bandwidth to dynamic; byte 5: bits 0, 1 and 4 (FMBW) at 0.
. With an RF input signal of 200µV and byte 5: bit 4 = 0 the IF2 bandwidth is maximum (155kHz).
. While varying bits 0 to 3 in Byte 8, read and store the dc output levels at pin 70.
. Set the FMBW bit (byte 5: bit4) at 1 (= align-mode), so the bandwidth will be minimum (=25kHz).
. Again vary bits 0 to 3 in Byte 8, read and store the dc output levels at pin 70.
. The proper setting of the IF2 Filter Gain (FGN bits in byte 8) is the value where the difference
between the two readings is at its minimum.
2. Antenna DAA alignment at FM.
The DAA values take care of the tracking between front-end and oscillator by applying an offset
between the tuning voltage to the front-end and the tuning voltage of the oscillator circuit. Usually only
three alignments are necessary and sufficient for a good tracking performance: lower band limit,
upper band limit and in the centre of the band.
1
47
TEA 6848H A NICE RADIO
APPLICATION
NOTE
With Circumstantial Controlled Selectivity
AN
In the user application the proper DAA value for any given tuning frequency may be interpolated from
the aligned values. When also the weather-band is included in the final application one extra
alignment is required.
The procedure to align the antenna DAA value is as follows:
Set a generator (no modulation) with an RF level of about 200 µV to the frequency to be aligned and
tune NICE to this frequency. Next ramp the DAA word from 0 to 127 while measuring the DC-output
of the level detector of NICE (pin 70) for each DAA value. The proper DAA value to be stored is the
DAA word where the level has its maximum value. To speed-up this process an ‘intelligent’ algorithm
can be used.
3. AM & FM level slope and level start.
The DC-output of the level detector (pin 70) is used to control the NICE_PACS tuner functions: FM
Keyed AGC, FM threshold extension and AM noise canceller inside the tuner, where weak signal
behaviour, search criteria etc. are controlled either by CASP (Car Analogue Signal Processor) or by
CDSP (Car Digital Signal Processor). For (re) production purposes the starting point of the level
detector output should be aligned as well as the slope of the level detector output. These level
alignments (done on different frequency bands to compensate for gain variations over frequency)
require three different steps:
FM:
• The level detector output is aligned to e.g. 950mV at an RF input level of 4.5µV.
• The level slope is aligned in such a way that the difference in level detector output between RF
levels of 20 and 200µV is 800mV with the level start value found in the first alignment.
• The level detector output at 4.5µV RF level is re-aligned to 950 mV with level slope at the value
found at the previous alignment.
(Note: 950mV at FM is switch-off level of keyed AGC as well as start of threshold extension).
Fig. 41 FM Level Voltage
AM:
• The level detector output is aligned to 2 Volt (switch-off level of the noise blanker) at an RF input
level of e.g.150µV.
• Dependent on backend (ASP) requirements, the level slope could be aligned, and then in such a
way that the difference in level detector output between RF levels of 20 and 200µV is 800 mV
with the level start value found in the first alignment.
• The level detector output at 150µV RF level is re-aligned to 2000mV with level slope at the value
found at the previous alignment.
The procedure to align the start and slope values is as follows:
1
48
TEA 6848H A NICE RADIO
APPLICATION
NOTE
With Circumstantial Controlled Selectivity
AN
Initially set level start to 11 and tune NICE to 97.5MHz (990kHz for AM). Set the frequency
generator to 97.5MHz (990kHz for AM) without modulation. In the first alignment the level start is
ramped down until the proper level detector output has been found (950mV at 4.5µV RF level at FM
or 2000mV at 150µV RF level for AM). In the second alignment the level slope is ramped down until
the difference in signal level output between 20 and 200µV RF level is 800mV (both for FM and AM
*)
). Finally the level start value is re-aligned to 950mV at 4.5µV RF level or 2000mV at 150µV RF
level for AM.
*)
Note:
Normally the AM slope alignment is of no importance for the performance of the NICE
system, so this alignment could be skipped and the slope may be set to e.g. 0 in the
final application. Only in case AM soft-mute feature in the CDSP is used the AM slope
alignment has to be done.
4. Frequency offset detector alignment.
When tuned to a (very) weak desired signal with a strong undesired neighboring signal at 100kHz
with relatively high deviation, the bandwidth could switch continuously from maximum to minimum
and vice versa (with resulting audible effects). To avoid this a frequency offset detector is
implemented in the TEA6848H to reduce the bandwidth of the IF filter when the detected frequency
offset in the demodulator is too large. This avoids the so-called pop effect what otherwise could be
present under certain input signal conditions.
Due to spread the frequency-offset detector itself must be aligned with 4 bits. This will reduce the
frequency offset due to spread and temperature to a low value. The ± 1.5kHz on resulting accuracy
with ± 5kHz on temperature dependency (so max. ± 7kHz) results in a good performance. It is
recommended to align the frequency offset detector in the middle of each frequency band covered
by the application.
The procedure to align:
. The FM bandwidth in align-mode by FMBW, byte 5_bit 4=1,such to have minimum IF2 bandwidth
(25 kHz). The frequency offset detector output will then be routed to pin 62.
. Vary bits 4 to 7 in Byte 8 (FOF-bits) until a minimum voltage is found at pin 62.
Note: We only put a separate alignment-data EEPROM on the tuner PCB so we can test and align
the tuner independently of the car radio (or CASP/CDSP demo boards). If that is not required it is
very well possible to use only one EEPROM. Because the NICE tuner needs about ten bytes MAX for
alignment data, it is nearly always possible to find that memory space in the main EEPROM.
Fig. 42 is an example of aligned settings in an FM-Europe application.
1
49
TEA 6848H A NICE RADIO
APPLICATION
NOTE
With Circumstantial Controlled Selectivity
AN
Fig. 42 Settings in an FM-Europe application
MODULE
APPENDIX 3.
a. Module PCB
Fig. 43
1
50
TEA 6848H A NICE RADIO
APPLICATION
NOTE
With Circumstantial Controlled Selectivity
AN
1
51
52
TEA 6848H A NICE RADIO
APPLICATION
NOTE
With Circumstantial Controlled Selectivity
AN
b. Module application diagram
Fig. 44
1
53
c. COMPONENTS
From the NICE-Module with TEA6848H, acc. to the given application, the components are:
ITEM
CNT
PART_NO
COMPONENT
SERIES TOLE
RANC
E
1
1
8222-411-39272b
BOARD
PR39272b
2
1
2322-732-63302
3.3K
RC12G
1%
3
4
5
1
3
4
CAP-CER-590-nF
2222-950-16654
2222-910-19854
XnF_0805
220nF
220nF
C0805-X7R
X7R
Y5V
6
7
8
9
10
11
1
1
1
3
1
1
2322-730-61271
PN-BAQ806
LQN1HR50K04
2222-861-12102
2222-861-12271
LAL03NA101K
270
BAQ806
500nH
1nF
270pF
100uH
RC11
Pin diode
LQH
NP0
NP0
LAL03NA
12
13
1
1
LAL03NA151K
LQN1HR21K04
150uH
215nH
LAL03NA
LQH
10% TAIYO_YUDEN
14
15
16
17
1
2
1
6
LAL02NA1ROK
2222-872-16663
2322-702-60102
2322-702-60103
1uH
1uF
1k
10k
LAL02NA
X7R
RC21
RC21
10% TAIYO_YUDEN
10%
25V
5%
0.063W
5%
0.063W
18
19
20
21
22
23
24
1
1
1
2
1
2
1
2322-702-60122
2322-702-60124
2322-702-60185
2322-702-60222
2322-702-60223
2322-702-60225
2322-702-60479
1.2k
120k
1.8M
2.2k
22k
2.2M
47
RC21
RC21
RC21
RC21
RC21
RC21
RC21
5%
5%
5%
5%
5%
5%
5%
0.063W
0.063W
0.063W
0.063W
0.063W
0.063W
0.063W
PHILIPS
PHILIPS
PHILIPS
PHILIPS
PHILIPS
PHILIPS
PHILIPS
R0603
R0603
R0603
R0603
R0603
R0603
R0603
25
8
2322-702-60229
22
RC21
5%
0.063W
PHILIPS
R0603
26
5
2322-702-60331
330
RC21
5%
0.063W
PHILIPS
27
28
29
30
31
32
1
1
1
2
1
9
2322-702-60391
2322-702-60472
2322-702-60479
2322-702-60561
2322-702-60563
2322-702-96001
390
4.7k
47
560
47k
0
RC21
RC21
RC21
RC21
RC21
RC21
5%
5%
5%
5%
5%
5%
0.063W
0.063W
0.063W
0.063W
0.063W
0.063W
PHILIPS
PHILIPS
PHILIPS
PHILIPS
PHILIPS
PHILIPS
R0603
R0603
R0603
R0603
R0603
R0603
'R23'
'R46'
'R9'
'R13' 'R5'
'R79'
'R34' 'R39' 'R4'
'R42' 'R45' 'R53'
'R73' 'R74' 'R77'
33
34
35
36
37
1
1
1
2
2
388BN-1211Z
LN-G102-587
9332-153-70212
LAL02NA3R3K
2222-134-35109
TOKO
NDK
PHILIPS
7PD_1
'L18'
'X1'
'D1'
'L17' 'L2'
'C16'
38
1
2222-134-55229
388BN-1211Z
7PS
20.5MHz
Crystal
BAV99 Gen. purpose
3.3uH
LAL02NA
10uF
RLP5 134
22uF
RLP5 134
54
RATING
VENDOR
GEOMETRY
PS-SLE
BOARD
0.1W
PHILIPS
R0805
10%
10%
-400
63V
16V
25V
PHILIPS
PHILIPS
PHILIPS
C0805
C36'
C0805 'C102' 'C28' 'C9'
C0805 'C12' 'C21' 'C51'
'C76'
5%
0.1W
PHILIPS
PHILIPS
muRata
PHILIPS
PHILIPS
R0805
'R19'
SOD106
'D2'
LQH1N
'L12'
C0805 'C37' 'C38' 'C59'
C0805
'C63'
uChoke_3e
'L9'
5%
50V
5%
50V
10% TAIYO_YUDEN
muRata
10% TAIYO_YUDEN
20%
16V
20%
16V
PHILIPS
PHILIPS
PHILIPS
uChoke_3e
LQH1N
REFERENCE
R78*
'L6'
'L16a'
uChoke_2e
'L16'
C1210
'C39' 'C42'
R0603
'R95'
R0603 'R11' 'R41' 'R49'
'R51' 'R80' 'R70'
'R36'
'R15'
'R12'
'R22' 'R38'
'R29'
'R14' 'R3'
'R24'
'R1' 'R10' 'R37'
'R40' 'R7' 'R75'
'R91'
R0603 'R21' 'R26' 'R27'
'R30' 'R31'
SOT23
uChoke_2e
PHILIPS CASE_R52_T
FA
PHILIPS CASE_R54_C
A
'C23’
TEA 6848H A NICE RADIO
APPLICATION
NOTE
With Circumstantial Controlled Selectivity
AN
39
4
2222-134-55479
47uF
RLP5 134
40
41
42
43
44
45
1
1
1
1
2
2
9340-555-19215
P826RC-5134N-S
PN-PCF8594--2T
PN-BB156
2222-596-16606
2222-596-16607
BF862
BB156
270pF
330pF
Fet
7PSG
IC Universal
Tuner Diode
X7R
X7R
10%
10%
50V
50V
PHILIPS
TOKO
PHILIPS
PHILIPS
PHILIPS
PHILIPS
46
47
48
49
50
51
52
53
1
1
1
1
2
3
1
6
2222-596-16614
2222-596-16621
2222-596-16622
2222-596-16625
2222-596-16626
2222-916-16736
2222-916-16738
2222-916-16741
1nF
3.3nF
3.9nF
6.8nF
8.2nF
10nF
18nF
22nF
X7R
X7R
X7R
X7R
X7R
X7R
X7R
X7R
10%
10%
10%
10%
10%
20%
20%
20%
50V
50V
50V
50V
50V
25V
25V
25V
PHILIPS
PHILIPS
PHILIPS
PHILIPS
PHILIPS
PHILIPS
PHILIPS
PHILIPS
C0603
'C19'
C0603
'C62'
C0603
'C56'
C0603
'C48'
C0603
'C75' 'C99'
C0603 'C14' 'C18' 'C67'
C0603
'C73'
C0603 'C11' 'C13' 'C5'
'C72' 'C74' 'C90'
54
55
3
9
2222-786-16745
2222-786-16749
47nF
100nF
X7R
X7R
20%
20%
16V
16V
PHILIPS
PHILIPS
C0603 'C71' 'C8' 'C91'
C0603 'C25' 'C30' 'C43'
'C49' 'C50' 'C53'
'C55' 'C57' 'C70'
56
5
2222-586-19807
22nF
Y5V
-400
50V
PHILIPS
C0603
'C1' 'C17' 'C29'
'C52' 'C6'
57
58
59
60
1
1
1
1
9334-606-20212
9335-896-40215
LAL02NA6R8K
2222-867-12108
PHILIPS
PHILIPS
PHILIPS
SOT23
SOT23
uChoke_2e
C0603
'D6'
'TR3'
'L5'
'C35'
61
62
63
1
1
1
2222-867-12129
2222-867-12151
2222-867-12188
PHILIPS
PHILIPS
PHILIPS
C0603
C0603
C0603
'C32'
'C66'
'C47'
64
65
1
1
2222-867-12189
2222-867-12278
PHILIPS
PHILIPS
C0603
C0603
'C26'
'C46'
66
1
2222-867-12338
PHILIPS
C0603
'C4'
67
2
2222-867-12398
PHILIPS
C0603
'C103' 'C33'
68
69
70
71
1
1
1
2
2222-867-12688
LAL02NA820K
PN-TEA6848
Q62702-A952
PHILIPS
PHILIPS
SIEMENS
C0603
uChoke_2e
SOT315
SOD323
'C100'
'L7'
'IC1'
'D3' 'D4'
72
73
1
1
Q62702-B372
CFWS450F
BAS16 Gen. Purpose
BC848C Gen. Purpose
6.8uH
LAL02NA
10% TAIYO_YUDEN
1pF
NP0 0.25p
50V
F
12pF
NP0
5%
50V
150pF
NP0
5%
50V
1.8pF
NP0 0.25p
50V
F
18pF
NP0
5%
50V
2.7pF
NP0 0.25p
50V
F
3.3pF
NP0 0.25p
50V
F
3.9pF
NP0 0.25p
50V
F
6.8pF
NP0 0.5pF
50V
82uH
LAL02NA
10% TAIYO_YUDEN
TEA6848
IC Universal
BA595
Pin diode
SEE REMARK
below
BB814
Tuner Diode
IF-Filter
SIEMENS
muRata
SOT23
SFR450H
'D5'
'FL3'
74
1
396INS.3076X
5KM
TOKO
TOKO_5km
'L10'
75
2
IF-Filter
muRata
SFE_3p
'FL1' 'FL2'
76
1
611SNS-1066Y
5KM
TOKO TOKO_5KM_
m2
'L14'
77
1
P7PSGAE-5078D=S
7KM
TOKO TOKO_7km_m
2_m5
'L4'
78
79
1
1
LAL02NAR27K
E543SNS-02010
uChoke_2e
MC137
'L3'
'L15'
SFE10.7MS3 SFE10.7MS3A1
0k-A
270nH
LAL02NA
MC137
1
55
20%
16V
PHILIPS CASE_R55_C 'C44' 'C58' 'C60'
A
‘C40’
10% TAIYO_YUDEN
TOKO
SOT23
7PS_p2
SOT96
SOD323
C0603
C0603
'TR1'
'L1'
'IC2'
'D7'
'C101' 'C20'
'C15' ‘C77’
TEA 6848H A NICE RADIO
APPLICATION
NOTE
With Circumstantial Controlled Selectivity
AN
REMARK: Alternative FM pin diode: KP2311E from TOKO.
1
56
TEA 6848H A NICE RADIO
APPLICATION
NOTE
With Circumstantial Controlled Selectivity
AN
TEA6848H Module Specification
APPENDIX 4
In a Car Radio application (see Fig. 44) TEA 6848H performs typical as given in next specification.
AM-MW reception:
At Fa
and at
With
=
530
to
Fif1 =
1710
kHz,
10.7
MHz,
Fosc1
=
11.23 to
12.41 MHz, obtained via divider
Fvco
=
224.6 to
248.2 MHz.
Fif2 =
Fosc2
450
=
N2
=:20 , so
N
=:2
kHz,
10.25 MHz, obtained via a divider
from
X-tal osc =
20.5
MHz.
FM reception (USA/W-Europe application):
At
Fa
and at
Fif1 =
With
=
87.5
to
108
10.7
MHz,
to
118.7 MHz, obtained via divider N=:2, so
Fosc1
=
98.2
Fvco
=
196.4 to
Fif2 =
Fosc2
450
=
MHz,
237.4 MHz,
kHz,
10.25 MHz, obtained via a divider N=:2
from
X-tal osc =
20.5
MHz.
RATINGS
Parameter
min
unit
typ
max
Supply Voltage: Operation
8
9
Volt
-40
+85
°C
Temperature : Operating
AM-SIGNAL-CHANNEL.
Test with dummy aerial 15/60pF from 50 Ohm source
Conditions (a.o. for standard output), unless otherwise specified :
Va= 10mV, F= 1MHz, fmod = 400Hz, m = 0.3;
Vsupply = 8.5 Volt/ Tamb = 25°C
AM Performance:
Typ.
Unit
10
dΒ
1. Sensitivity
Signal to Noise at Va = 6µV
:
1
57
TEA 6848H A NICE RADIO
APPLICATION
NOTE
With Circumstantial Controlled Selectivity
AN
Signal to Noise at Va = 45µV
:
26
dB
Typ.
Unit
:
58
dB
:
290
mV
Va1/Va2 :
61
dΒ
0.3
%
2. AM-Signal to noise ratio
Signal to Noise at Va = 10mV
3. A.F. output at m=30%
4. Figure of Merit at soft mute on
Va2 for ∆Vout_a.f. at = -10dB with respect to Va1 =5mV:
5. Distortion (THD) at m= 0.8
:
Further AM-performance:
Typ.
Unit
6. Selectivity S9
:
75
dB
7. Dynamic Selectivity (DS ±20)
:
73
dB
:
>75
dB
:
>80
dB
:
75
dB
Fa + 2*IF2
:
70
dB
Fa + 2*(IF1-IF2)
:
>100
dB
:
49
dB
:
>140
dBµV
:
130
dBµV
:
19
:
1
Vref= Vout at standard mod. of Va1.
When Va1 has m=0 and
Fa2= Fa1+20kHz resp -20kHz; m=30% 1kHz,
Va2/Va1 at Vout=Vref-10dB
8. IF-rejection (at 600kHz); Fa= IF1
Fa= IF2
9. Image rejection(at 1600kHz) Fa+ 2*IF1
10. Second Harmonic Rejection
Tuned at 1340 kHz for Vout1 at m= 30%;
Va2 (m= 30%) for Vout2=Vout1 at Fa2= 670kHz.
11. Large signal handling
at THD= 10% where m= 80%.
12. Intermodulation IP3 for in-band interference
(interfering transmitters at ± 100kHz offset)
13. Desensitization at Va=1Volt
dB
for Fa2= Fa1-40kHz at Fa1= 1310kHz
14. FM to AM switching time
FM-SIGNAL-CHANNEL
Test with 50 Ohm (gen.) + dummy aerial = 75 Ohm source.
Test conditions, unless otherwise specified :
Vi = 1mVrms , Fa = 98MHz,
1
58
sec
TEA 6848H A NICE RADIO
APPLICATION
NOTE
With Circumstantial Controlled Selectivity
AN
mono with fmod = 400Hz, ∆f= ±22.5 kHz. De-emphasis 50 µsec.
Vsupply = 8.5 Volt/ Tamb.= 25°C
FM PERFORMANCE
Typ.
Unit
26
dB
1. Sensitivity
at ∆f = ± 22.5kHz
- Signal to Noise ratio at Va = 1.4µV
:
2. Signal to noise ratio
- S/N at Va = 1mV
:
63
dB
3. A.F. output at ∆f= ±22.5kHz
:
230
mV
4. Distortion (THD) at ∆f = ±75kHz
:
0.3
%
5. AM signal suppression (m= 0.3)
:
>50
dB
Further FM-Performance:
Typ.
Unit
6. -3dB limiting at Vi
:
1
µV
7. Dynamic Selectivity DS±100
:
35
dB
Dynamic Selectivity DS±200
:
72
dB
:
<6
kHz
IF1
:
> 90
dB
IF2
:
>100
dB
Fa+ 2*IF1
:
70
dB
Fa + 2*IF2
:
>100
dB
Fa + 2*(IF1-IF2)
:
>100
dB
8. IF2 accuracy (incl. temperature influence)
9. IF-rejection (at 87.9MHz),
10. Image rejection (at 107.9MHz)
11. Fieldstrength dc-level info range
:
>80
dB
:
82
dB
:
71
62
dB
:
117
dBµV
:
34
dB
:
0.8
ms
12. Adjacent Channel Selectivity (static) SS±200
at Bandwidth 60 kHz
Bandwidth 90 kHz
Bandwidth 130 kHz
:
Standard signal at Va1= 100uV as reference,
then unmodulated and Fa2 at + resp. -200kHz,
causing a S/N= 30dB at a ratio Va2 to Va1
:
13. Intermodulation IP3
14. Desensitization on limiting sensitivity
by Fa2= Fa1+1.5MHz
15. PLL In-lock time 108 to 88MHz
1
59
dB
TEA 6848H A NICE RADIO
APPLICATION
NOTE
With Circumstantial Controlled Selectivity
AN
Weather-band Receiver
APPENDIX 5
The weather-band as currently in operation in the USA consists of a nation-wide network of
radio stations broadcasting continuous weather information direct from various (local) stations
throughout the US for general public use. The network is provided as a public service by the
National Oceanic & Atmospheric Administration (NOAA), and as such it is controlled by the
federal government. When necessary, warnings, watches and other hazard information can be
broadcast for the region involved in addition to weather forecasts. Broadcasts are found at
seven (narrow-band FM modulated) voice channel frequencies ranging from 162.400 MHz to
162.550 MHz.
A weather-band receiver with Nice_Pacs, using the integrated narrow-band IF2 filter
provides a cost-effective quality solution. The sensitivity achieved with Nice_Pacs module
including the IF2 filter set at 13 kHz is 2.0 µV for 20dB S/N ratio (@ ∆f= 1.5kHz and 110µsec
de-emphasis).
Weather-band
Frequency
Modulation
Deviation
De-emphasis Approx.
162.4 to 162.55 MHz
Narrow Band FM (NBFM)
∆f nominal = ±1.5kHz (∆f maximum = ±5kHz)
110 µsec (2nd order filter)
Weather-band reception with the TEA6848H
The Nice_Pacs IC TEA6848H has several on-board provisions for the reception of the
weather-band. The Nice_Pacs IC utilises the signal path used for the standard broadcast FM
band also in weather-band mode. This means the IF1 is at 10.7MHz, the local oscillator
operates at 173.1 to 173.25 MHz.
When programmed through the I2C Bus for weather band reception:
1. A weather-band flag is set (pin 34), indicating to switch the front-end filter to the weatherband frequency. A current switch at pin 34 is used to switch a coil in parallel with the LC
resonant circuit to move the tank resonant frequency to about 162MHz.
2. The internal local oscillator divider is switched off (division ratio one).
3. A quadrature phase shift network is activated to drive the quadrature mixer with image
cancelling.
4. The IF2 bandwidth is switched to its minimum (13kHz) with 10dB additional IF2 amplifier
gain *).
5. The MPX amplifier/buffer following the FM demodulator is switched to a 15x higher gain,
otherwise the demodulated FM signal level when receiving the weatherband should be
smaller than the demodulated wideband FM signals, due to the frequency deviation of the
narrow band FM signals in weatherband. So the A.F.-output in Wx mode is 230mV at ±1.5
kHz modulation.
*)
IF-filter switching: The allocation of the frequencies for the (neighbouring) transmitter
stations is such that adjacent channel interference is hardly anywhere present in the USA.
This means a larger bandwidth for the IF filter is tolerable. It is indeed of more importance,
judging from the frequency allocation scheme, that the rejection at the next adjacent
channel (at twice the channel spacing: 50kHz) is sufficient.
1
60
TEA 6848H A NICE RADIO
APPLICATION
NOTE
With Circumstantial Controlled Selectivity
AN
nd
A 2 order filter with a low pass response (-3dB point) at about 1400Hz can serve deemphasis and limit the audio bandwidth, thereby increasing the S/N ratio and enhancing the
audibility of the speech information.
Performance:
The following characteristics are measured with a 75Ω dummy antenna;
faf = 1 kHz; ∆f =± 1.5kHz; AF double pole filtering with –3dB at ~1400Hz.
Wide band FM AGC threshold is set to 4mV.
Sensitivity for 20dB S/N
Image rejection ratio
Distortion at ∆f = ±5kHz
Static selectivity : ± 25kHz
AM suppression
IF rejection
2.0µV
43.5 dB
< 0.5 % for Va >10µV
23 dB typ
> 25 dB
86 dB
1
61