InfiniBand Class I Power Supply Using the ISL6160 and HIP6006 ICs TM Application Note September 2001 Introduction The InfiniBand architecture represents a significant evolution of high-performance, switched-fabric interconnect systems. The goal is to provide a high-performance, reliable, and scalable way to connect high-end servers to each other and to I/O subsystems, routers and switches that connect to the world outside the data center. Available from Intersil is the ISL6160, an evolution of IC power sequencing, control and protection for InfiniBand I/O modules (IM). The ISL6160 is designed to address the unique power requirements of the InfiniBand (IB) industry initiative providing independent power control of both the VB (bulk) (+12V) and the VA (auxiliary) (+5V) power rails for a single port. This device can be implemented in both IB Class I (non isolated) and Class II (isolated) Power Topology applications. Intersil also provides the ISL6160EVAL2 concept evaluation platform. The ISL6160EVAL2 is a complete InfiniBand Class I (non isolated) power topology evaluation platform which highlights the operation of the ISL6160 and the HIP6006 single output PWM controller. See Figure 1 for a simplified block diagram of the ISL6160EVAL2 platform. This evaluation platform allows the InfiniBand Module (IM) power supply designer to evaluate the concept of this design and apply this concept to a specific IM power requirement. The evaluation platform is configured for 5V Vout and 3.5A max Iout capability where it exhibits an efficiency of 85%, all in a small 1.6 sq” area. See Figure 11 for a complete ISL6160EVAL2 schematic. Using the ISL6160EVAL2 Concept Board The ISL6160EVAL2 consists of a bus and load boards, representing the IB chassis and IM respectively. The bus board has terminals for VA and VB supplies. The load board with its staggered length connector fingers to emulate the IM connector then hot plugs into the socket as shown in Figure 2. When the load board is inserted into the bus board, the stagger on the connector fingers, first provides VX_RET, then VX connections, and finally the shortest finger emulates the VBx_En_L line connection. Once VB_In is connected the VB control portion of the circuit is biased but the VB Secondary Rail (TP1) is held off until the ISL6160 VB_ON pin is signaled high. Local power enable signaling is provided through the LCL_PWR_EN jumper either as a hard tie ‘high’ with the jumper installed or through an external input signal, on TP6 with the jumper removed. A single logic gate IC, provides for the XORing of the VBxEN _L and local power enable signals into the ISL6160 VB_ON pin. At the time VB_ON is asserted high the ISL6160 turns on the VB Secondary Rail in a soft start mode protecting the primary 1 AN9959 supply rail from sudden in-rush current. During turn-on, the external gate capacitor of the N-Channel MOSFET, Q3 (VB switch) is charged with a 20µA current source resulting in a programmable ramp (soft start turn-on). An internal charge pump supplies the gate drive for the 12V VB supply switch driving the MOSFET gate to VB +5V. Once the VB Secondary Rail ramps to 10V the DC-DC_En pin is pulled high thus enabling the accompanying voltage converter. The DC-DC converter then provides a well regulated output voltage to the load. For lab evaluation either an electronic or a passive load is suitable for suppling a load current. The ISL6160 VA undervoltage lockout feature prevents turn-on of VA until VA_In > 2.5V. It then enables the VA soft start and power up. The VA rising voltage output is a current limited ramp so that both the inrush current and voltage slew rate are limited, independent of load. This reduces supply droop due to surge and eliminates the need for additional external EMI filters. During operation, once a VA OC condition is detected the output current is limited to 1A for 12ms to allow transient conditions to pass. If VA is still in current limit after the current limit period has elapsed, the output is then latched off. The VA to the IM circuitry is latched off until reset by the disconnection and reconnection of the IM from the chassis backplane. VB_IN DC-DC_ON VB_ON VA_IN EN UGATE HIP6006 ISL6160 LGATE Vout RL CL VAout RLOAD FIGURE 1. ISL6160EVAL2 BLOCK DIAGRAM The VB Secondary Rail is enabled once the VB_ON (TP7) is signalled high (through the assertion of the local power enable), then the DC-DC En pin (TP2) is pulled high to VB. The RC network of R4, R15 and C14 allows for setting the DC-DC converter enabling signal level and ramp, thus customizing the time to DC-DC enabling. Once the DC-DC is enabled the output (TP3) ramps to 5V. The output is supplied with a banana jack for connecting to an external active or passive load. Figure 3 illustrates typical operational waveforms of the ISL6160EVAL2. These are accessible through the labeled test points (TPX) on the eval board. See Figures 4 and 5 for ISL6160EVAL2 turn-on and turn-off output voltage waveforms. CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures. 1-888-INTERSIL or 321-724-7143 | Intersil and Design is a trademark of Intersil Americas Inc. Copyright © Intersil Americas Inc. 2001, All Rights Reserved Application Note AN9959 LCL_PWR_EN (TP6) VB Secondary Rail (TP1) VGATE (TP4) DC-DC_EN (TP2) Vout (TP3) Iout (2A / DIV) 5V / DIV 2ms / DIV FIGURE 5. ISL6160EVAL2 Vout TURN-OFF ISL6160EVAL2 Performance FIGURE 2. ISL6160EVAL2 PHOTOGRAPH IOUT (1A/DIV) VB INPUT CURRENT (1A/DIV) Efficiency Figure 6 displays the ISL6160EVAL2 efficiency versus load current. It highlights the efficiency advantages of a switching regulator at a higher load current. The designed current limit of this evaluation bd is ~3.5A. The dashed portion of the curve was collected from a modified evaluation board with an increased overcurrent protection limit. The curve indicates maximum efficiency at about 4.5A of output current and approximately 25W of input power. VPHASE (10V/DIV) (TP9) VB RAIL 5V/DIV 90 85 VOUT (5V/DIV) (TP3) TIME (4µs/DIV) FIGURE 3. ISL6160EVAL2 OPERATIONAL WAVEFORMS EFFICIENCY (%) VB RAIL (5V/DIV) 80 25W 75 70 LCL_PWR_EN (TP6) MEAN CURRENT INTO VB SWITCH * VB MEAN LOAD CURRENT * VOUT 65 VGATE (TP4) 1 1.5 2 2.5 3 3.5 LOAD CURRENT (A) 4 4.5 5 FIGURE 6. ISL6160EVAL2 EFFICIENCY vs LOAD CURRENT VB Secondary Rail (TP1) DC-DC_EN (TP2) Vout (TP3) ISL6160 related efficiency improvements can only come from lowering the RDSon of the VB FET (Q3) switch and the threshold voltage across the sense resistor that invokes current regulation and shutdown. HIP6006 related efficiency improvements are explained in the Power Supply Design Considerations section of this document. Transient Response 5V / DIV 20ms / DIV FIGURE 4. ISL6160EVAL2 Vout TURN-ON 2 Figure 7 shows a laboratory oscillogram of the ISL6160EVAL2 in response to a 0-3.5A, 250A/ms load transient. The output voltage responds rapidly and is within 2% of its nominal value in less than 150µs. Application Note AN9959 This lower limit is based on the achieved efficiency of the down stream converter design and the max. power capability of that converter output. At point of failure, by providing a lower current regulation limit on the I/O module the risk of passing through any rail voltage disruptions is reduced or eliminated by not having to rely on the overhead capacity of the chassis supply. 5V VOUT (1V/DIV) VOUT (200mV/DIV) 5V HIP6006 OC Protection ILOAD (2A/DIV) 0A TIME (40us/DIV) FIGURE 7. ISL6160EVAL2 TRANSIENT RESPONSE Output Current and Voltage Ripple The output current and voltage ripple of the HIP6160EVAL2 is shown in Figure 8. The load current is 3.5A for this oscillogram. Peak-to-peak voltage ripple is about 60mV under these conditions. VOUT (50mV/DIV) The HIP6006 has a loss less overcurrent (OC) protection feature. This is accomplished via the current-sense function of the HIP600x family. The HIP6006 senses converter load current by monitoring the drop across the upper MOSFET (Q2a in the Figure 11 schematic) enhancing the converter’s efficiency and reducing cost by eliminating a current sensing resistor. The over-current function cycles the soft-start function in a hiccup mode to provide fault protection. A resistor (ROCSET, R6) programs the over-current trip level. An internal 200µA (typical) current sink develops a voltage across ROCSET that is referenced to the VB secondary rail. When the voltage across the upper MOSFET (also referenced to VB secondary rail) exceeds the voltage across R OCSET, the over-current function initiates a soft-start sequence. The soft-start function discharges CSS with a 10µA current sink and inhibits PWM operation. The soft-start function recharges CSS, and PWM operation resumes with the error amplifier clamped to the SS voltage. Should an overload occur while recharging CSS, the soft start function inhibits PWM operation while fully charging CSS to 4V to complete its cycle. The converter dissipates very little power with this method. ILOAD (50mA/DIV) TIME (2us/DIV) FIGURE 8. ISL6160EVAL2 OUTPUT RIPPLE OC Protection With the ISL6160EVAL2 Class I power supply concept there are two areas of OC protection. The ISL6160 limits the current into the port, whereas the HIP6006 will limit current to the load. The over-current function will trip at a peak inductor current (IPEAK) determined by: I OCSET • R OCSET I PEAK = --------------------------------------------------r DS ( ON ) where IOCSET is the internal OCSET current source (200µA - typical). The OC trip point varies mainly due to the MOSFETs rDS(ON) variations. To avoid over-current tripping in the normal operating load range, find the R OCSET resistor from the equation above with: 1. The maximum rDS(ON) at the highest junction temperature. ISL6160 OC Protection The ISL6160EVAL2 is designed to input current limit to 2.8A, the max. specified peak current for a 25W port. This allows a maximum output current of ~3.5A at 5V output voltage. As Iout increases above 3.5A the input current ripple peaks increase and are limited to 2.8A, beyond this point the ISL6160 reduces Q3 gate drive for current regulation (CR), causing a decrease in overall efficiency but protecting the VB primary rail. A lower limit based on the particular IM needs can be implemented to ‘tighten’ power budget control. 3 2. The minimum IOCSET from the specification table. 3. Determine IPEAK for I PEAK > I OU T ( MAX ) + ( ∆I ) ⁄ 2 where ∆I is the output inductor ripple current. , A small ceramic capacitor should be placed in parallel with ROCSET to smooth the voltage across ROCSET in the presence of switching noise on the input voltage. Figure 9 illustrates the ISL6160EVAL2 OC operational waveforms. The 5V DC-DC output is shorted and the Application Note AN9959 HIP6006 current limits it to 5A. When the Secondary Rail voltage decreases to 10V(TP1) the ISL6160 deasserts the DC-DC_EN pin (TP2) and shuts off the converter. The 1.6ms delay prevent spurious events from latching off the power supply. RFP45N06’s gain in switching losses offsets its decreased conduction losses at load currents up to about 9A. This data reinforces the need to consider both switching and conduction losses of the MOSFETs. This data is taken from the HIP6006EVAL1 platform. IOUT (1A/DIV) VB SECONDARY RAIL (5V/DIV) VB PRIMARY RAIL 5V/DIV EFFICIENCY (%) 90 85 RFP25N05 80 RFP45N06 75 5V VOUT (5V/DIV) DC-DC_EN (5V/DIV) 2 4 6 LOAD CURRENT (A) 8 10 TIME (0.4ms/DIV) FIGURE 9. ISL6160EVAL2 OVER-CURRENT OPERATION Power Supply Design Considerations The concept of the power supply demonstrated by the ISL6160EVAL2 can be scaled across the entire range of 1.3V to 12V of output voltage up to a 50W port power level. To encompass this entire range there are several component variables and trade-offs to consider. These variables and trade-offs are briefly discussed in this document, but for a more detailed and extensive explanation please refer to the several listed documents [2], [4], [5] on page 5. FIGURE 10. HIP6006EVAL1 EFFICIENCY WITH EITHER RFP25N05 OR RFP45N06 MOSFETs Setting the Output Voltage Simple resistor value changes allow for outputs as low as 1.3V or as high as the 12V input voltage. The steady-state DC output voltage can be set using the following simple formula: R8 V OUT = V R EF • 1 + -------- , where R5 VOUT = desired DC output voltage of the converter Input Capacitor Selection VREF = HIP6006 internal reference voltage (typically 1.27V) Use a mix of HIP6006 input (VB secondary rail) bypass capacitors to control the voltage overshoot across the MOSFETs. Use small ceramic capacitors for high frequency decoupling and bulk capacitors to supply the current needed each time Q1 turns on. The number of input capacitors and their capacitance are usually determined by their maximum RMS current rating. A conservative approach is to determine the converter maximum input RMS current, and assume it would all have to be supplied from the input capacitors. By providing enough capacitors to meet the required RMS current rating, one usually provides enough capacitance for proper power de-coupling. Output Capacitor Selection MOSFET Selection Effect on Efficiency [4] This section shows graphically that a larger, lower RDSon) MOSFET does not always improve converter efficiency. Figure 10 shows that smaller RFP25N05 MOSFETs are more efficient over most of the line and load range than larger RFP45N06 MOSFETs. The RFP25N05 has a rDS(ON) of 47mΩ (maximum at 25oC) versus 28mΩ for the RFP45N06. In comparison to the RFP25N05, the 4 Output capacitors are required to filter the output and supply the load transient current. The filtering requirements are a function of the switching frequency and the ripple current. The load transient requirements are a function of the slew rate (di/dt) and the magnitude of the transient load current. These requirements are generally met with a mix of capacitors and careful layout. As with the input capacitors, the number of output capacitors is determined by a parameter different than sheer capacitance. Based on the desired output ripple and output transient response, a maximum ESR can be determined. Based on the design’s dimensional restraints, an optimum compromise between the number and size of the output capacitors can be reached. Conservative approaches dictate using the data book’s maximum values for ESR; this way the design will still meet the initial criteria even at the end of capacitor’s active life. High frequency decoupling of the output may not be implemented if the application provides high frequency decoupling components at the load end of the output. In applications requiring good high frequency decoupling, the Application Note AN9959 output should be accordingly decoupled using a few ceramic capacitors. This measure is especially necessary if high ESL output capacitors are used. Output Inductor Selection The output inductor is selected to meet the output voltage ripple requirements and minimize the converter’s response time to the load transient. The inductor value determines the converter’s ripple current and the ripple voltage is a function of the ripple current. Increasing the value of inductance reduces the ripple current and voltage. However, the large inductance values reduce the converter’s response time to a load transient. One of the parameters limiting the converter’s response to a load transient is the time required to change the inductor current from an initial current value to the transient current level. During this interval the difference between the inductor current and the transient current level must be supplied by the output capacitor. Minimizing the response time can minimize the output capacitance required. Output Ripple Voltage The amount of ripple voltage on the output of the DC-DC converter varies with, switching frequency, output inductor, and output capacitors. by the output load is initially delivered from the output capacitors. This is due to the finite amount of time required for the inductor current to slew up to the level of output current required by the load, and results in a temporary dip (∆VLOW) in the output voltage, see Figure 7. Conversely, a sudden removal of the same output load, the energy stored in the inductor is dumped into the output capacitors, creating a temporary hump (∆VHIGH) in the output voltage. Conclusion The ISL6160EVAL2 board lends itself well to the evaluation of a complete single fixed voltage IB Class I power supply and providing a conceptual platform for your specific IM power control and supply needs. In addition, with the availability of multiple output voltage converters such as the IPM6220A coupled with the ISL6160, Intersil provides an application solution for almost every InfiniBand I/O module. References For Intersil documents available on the web, see http://www.intersil.com/ [1] ISL6160 Data Sheet, Intersil Corporation, FN9028 [2] HIP6006 Data Sheet, Intersil Corporation, FN4306. Output Load Transient Response [3] IPM6220A Data Sheet, Intersil Corporation, FN9032 The application of a sudden load requiring the converter to supply maximum output current, most of the energy required [4] Application Note, Intersil Corporation, AN9722. 5 [5] Application Note, Intersil Corporation, AN9761 Application Note AN9959 TP4 + VB_IN (12V) R2 0.01 Q3 C3 R3 50 R1 1.4K 47µF VB_Ret VB_In GATE ISEN U1 ISL6160 VA_En ISET VA_Fault VA_In VA_IN + (5V) C1 1000pF CTIM VA_Out C2 100µF TP8 VB_On VA_Ret C13 0.1µF DC-DC En R12 50 C14 1000pF TP7 VBxEN_L U3 VA_RET TP2 VB_RET TP1 TP6 Local Pwr En (jumper) C4 470µF R4 10K C5 1µF 1206 C7 1µF 1206 C6 VCC EN 6 SS 3 R15 6.98K D2 4148 1000pF 14 2 OCSET MONITOR AND PROTECTION R6 4.1K 10 BOOT RT 1 C8 0.1µF OSC U2 HIP6006 REF R5 1K C10 C11 R8 1.6K L1 8 PHASE 33pF 12 LGATE + 4 COMP 11 PGND 7 GND R7 0.001µF C9 20K SPARE R9 1.6K FIGURE 11. ISL6160EVAL2 SCHEMATIC 6 C15 0.1µF 5VOUT 13 PVCC + - FB 5 TP3 TP9 Q2a 9 UGATE Q2b C12 820µF VB_RET Application Note AN9959 Bill of Materials for HIP6006EVAL1 PART # DESCRIPTION PACKAGE QTY REF VENDOR ISL6160IB InfiniBand Power Controller 14NSOIC 1 U1 INTERSIL HIP6006CV Synchronous Rectified Buck Controller 14TSSOP 1 U2 INTERSIL SN74AHC1G86 or equiv Single XOR gate 5SOT-23 1 U3 Various Si4922DY or equiv Dual 8A, 30V, 0.018Ω, N-Channel MOSFET 8SOIC 1 Q2 Various ITF86130SK8T or equiv 14A, 40V, 0.008Ω, N-Channel MOSFET 8SOIC 1 Q3 Various 1N4148 Rectifier, 100mA, 75V DO35 1 D2 Various 10µH Output Filter Inductor Wound Toroid 1 L1 Various 1.4K VB Current Set Resistor, 1%, 1/16W 0603 1 R1 Various 0.01Ω VB Current Sense Resistor, 1%, 1W 2512 1 R2 Various 50Ω SMD Resistor, 5%, 1/16W 0603 2 R3, R10 Various 10KΩ SMD Resistor, 5%, 1/16W 0603 1 R4 Various 1KΩ SMD Resistor, 5%, 1/16W 0603 1 R5 Various 4.1Ω SMD Resistor, 5%, 1/16W 0603 1 R6 Various 20KΩ SMD Resistor, 5%, 1/16W 0603 1 R7 Various Various 3.0KΩ SMD Resistor, 5%, 1/16W 0603 1 R8 DNP SMD Resistor, 5%, 1/16W 0603 1 R9 Various 50Ω Through hole Resistor, 5%, 1W - 1 R12 Various 6.98KΩ SMD Resistor, 5%, 1/10W 0805 1 R15 Various 100µF Electrolytic Aluminum Capacitor, 16V Radial 1 C2 Various 470µF Electrolytic Aluminum Capacitor, 16V Radial 1 C4 Various 820µF Electrolytic Aluminum Capacitor, 16V Radial 1 C12 Various 1000pF Ceramic Capacitor, 50V 0603 2 C1, C6, C14 Various 47µF Ceramic Capacitor, 50V 0603 1 C3 Various 1µF Ceramic Capacitor, 50V 0603 2 C5, C7 Various 0.1µF Ceramic Capacitor, 50V 0603 2 C8, C13, C15 Various 33pF Ceramic Capacitor, 50V 0603 1 C10 Various 0.001µF Ceramic Capacitor, 50V 0603 1 C11 Various - Local power enable Jumper - 1 PWR_EN Various - Test Points - 9 TP1- TP9 Various 1314353-00 Scope Probe Test Point - 1 TP3, VOUT Tektronics - Banana Jacks - 4 Various VB_IN, VA_IN, VB_RET, VA_RET EZM06DRXH Edge Connector - 1 - Sullins All Intersil products are manufactured, assembled and tested utilizing ISO9000 quality systems. Intersil Corporation’s quality certifications can be viewed at www.intersil.com/design/quality Intersil products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design and/or specifications at any time without notice. Accordingly, the reader is cautioned to verify that data sheets are current before placing orders. Information furnished by Intersil is believed to be accurate and reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Intersil or its subsidiaries. For information regarding Intersil Corporation and its products, see www.intersil.com Sales Office Headquarters NORTH AMERICA Intersil Corporation 7585 Irvine Center Drive Suite 100 Irvine, CA 92618 TEL: (949) 341-7000 FAX: (949) 341-7123 Intersil Corporation 2401 Palm Bay Rd. Palm Bay, FL 32905 TEL: (321) 724-7000 FAX: (321) 724-7946 7 EUROPE Intersil Europe Sarl Ave. C - F Ramuz 43 CH-1009 Pully Switzerland TEL: +41 21 7293637 FAX: +41 21 7293684 ASIA Intersil Corporation Unit 1804 18/F Guangdong Water Building 83 Austin Road TST, Kowloon Hong Kong TEL: +852 2723 6339 FAX: +852 2730 1433