DATASHEET

ISL6313
®
Data Sheet
September 2, 2008
FN6448.2
Two-Phase Buck PWM Controller with
Integrated MOSFET Drivers for Intel VR11
and AMD Applications
Features
The ISL6313 two-phase PWM control IC provides a
precision voltage regulation system for advanced
microprocessors. The integration of power MOSFET drivers
into the controller IC marks a departure from the separate
PWM controller and driver configuration of previous
multi-phase product families. By reducing the number of
external parts, this integration is optimized for a cost and
space saving power management solution.
• Precision Core Voltage Regulation
- Differential Remote Voltage Sensing
- ±0.5% System Accuracy Over-Temperature
- Adjustable Reference-Voltage Offset
One outstanding feature of this controller IC is its
multiprocessor compatibility, allowing it to work with both Intel
and AMD microprocessors. Included are programmable VID
codes for Intel VR11 as well as AMD 5-bit and 6-bit DAC tables.
A circuit is provided for remote voltage sensing, compensating
for any potential difference between remote and local grounds.
The output voltage can also be positively or negatively offset
through the use of a single external resistor.
• Fully Differential, Continuous DCR Current Sensing
- Integrated Programmable Current Sense Resistors
- Accurate Load Line Programming
- Precision Channel Current Balancing
The ISL6313 also includes advanced control loop features
for optimal transient response to load apply and removal.
One of these features is highly accurate, fully differential,
continuous DCR current sensing for load line programming
and channel current balance. Active Pulse Positioning (APP)
Modulation and Adaptive Phase Alignment (APA) are two
other unique features, allowing for quicker initial response to
high di/dt load transients. With this quicker initial response to
load transients, the number of output bulk capacitors can be
reduced, helping to reduce cost.
Integrated into the ISL6313 are user programmable current
sense resistors, which require only a single external resistor
to set their values. No external current sense resistors are
required. Another unique feature of the ISL6313 is the
addition of a dynamic VID compensation pin that allows
optimizing compensation to be added for well controlled
dynamic VID response.
Protection features of this controller IC include a set of
sophisticated overvoltage, undervoltage, and overcurrent
protection. Furthermore, the ISL6313 includes protection
against an open circuit on the remote sensing inputs.
Combined, these features provide advanced protection for the
microprocessor and power system.
1
• Integrated Multi-Phase Power Conversion
- 2-Phase or 1-Phase Operation with Internal Drivers
• Optimal Transient Response
- Active Pulse Positioning (APP) Modulation
- Adaptive Phase Alignment (APA)
• Variable Gate Drive Bias: 5V to 12V
• Multi-Processor Compatible
- Intel VR11 Mode of Operation
- AMD Mode of Operation
• Microprocessor Voltage Identification Inputs
- 8-Bit DAC
- Selectable between Intel VR11, AMD 5-Bit, and AMD
6-Bit DAC tables
- Dynamic VID Technology
• Dynamic VID Compensation
• Overcurrent Protection
• Multi-tiered Overvoltage Protection
• Digital Soft-Start
• Selectable Operation Frequency up to 1.5MHz Per Phase
• Pb-Free (RoHS Compliant)
Ordering Information
PART
NUMBER
(Note)
PART
MARKING
ISL6313CRZ* ISL6313 CRZ
ISL6313IRZ*
ISL6313 IRZ
TEMP.
(°C)
PACKAGE
(Pb-Free)
PKG.
DWG. #
0 to +70 36 Ld 6x6 TQFN L36.6x6
-40 to +85 36 Ld 6x6 TQFN L36.6x6
*Add “-T” suffix for tape and reel. Please refer to TB347 for details on
reel specifications
NOTE: These Intersil Pb-free plastic packaged products employ
special Pb-free material sets, molding compounds/die attach
materials, and 100% matte tin plate plus anneal (e3 termination finish,
which is RoHS compliant and compatible with both SnPb and Pb-free
soldering operations). Intersil Pb-free products are MSL classified at
Pb-free peak reflow temperatures that meet or exceed the Pb-free
requirements of IPC/JEDEC J STD-020.
CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures.
1-888-INTERSIL or 1-888-468-3774 | Intersil (and design) is a registered trademark of Intersil Americas Inc.
Copyright Intersil Americas Inc. 2007, 2008. All Rights Reserved
All other trademarks mentioned are the property of their respective owners.
ISL6313
Pinout
EN
SS
FS
ISEN1-
ISEN1+
ISEN2+
ISEN2-
RSET
VCC
ISL6313
(36 LD TQFN)
TOP VIEW
36
35
34
33
32
31
30
29
28
PGOOD
1
27
PHASE1
VID7
2
26
UGATE1
VID6
3
25
BOOT1
VID5
4
24
LGATE1
23
PVCC
37
GND
BOOT2
VID1
8
20
UGATE2
VID0
9
19
PHASE2
10
11
12
13
14
15
16
17
18
RGND
21
VSEN
7
OFS
VID2
FB
LGATE2
COMP
22
APA
6
DVC
VID3
REF
5
IOUT
VID4
ISL6313 Integrated Driver Block Diagram
PVCC
BOOT
UGATE
20kΩ
PWM
GATE
CONTROL
LOGIC
SOFT-START
AND
SHOOTPHASE
THROUGH
PROTECTION
10kΩ
FAULT LOGIC
LGATE
2
FN6448.2
September 2, 2008
ISL6313
Block Diagram
EN
PGOOD
RGND
OPEN SENSE
LINE PREVENTION
0.85V
VCC
POWER-ON
RESET
PVCC
VSEN
UNDERVOLTAGE
DETECTION
LOGIC
SOFT-START
AND
FAULT LOGIC
OVERVOLTAGE
DETECTION
LOGIC
BOOT1
APA
SS
ADAPTIVE PHASE
ALLIGNMENT
CIRCUITRY
UGATE1
MOSFET
DRIVER
PHASE1
MODE / DAC
SELECT
LGATE1
CLOCK AND
MODULATOR
WAVEFORM
GENERATOR
VID7
FS
VID6
OCP
VID5
VID4
VID3
DYNAMIC
VID
D/A
PWM1
+
I_TRIP
VID2
-
VID1
VID0
BOOT2
+
∑
+
RGND
DVC
∑
PWM2
+
x2
2kΩ
UGATE2
MOSFET
DRIVER
PHASE2
∑
LGATE2
REF
E/A
FB
CHANNEL
DETECT
COMP
OFS
IOUT
FS
OFFSET
I_AVG
x1
CHANNEL
CURRENT
BALANCE
I_AVG
+
1
N
CH2
CURRENT
SENSE
ISEN2RISEN2
ISEN2+
∑
RSET
+
OCP
CH1
CURRENT
SENSE
RISEN1
ISEN1+
ISEN1-
VOCP
GND
3
FN6448.2
September 2, 2008
ISL6313
Typical Application - ISL6313
FB
DVC
VSEN
COMP
RGND
RSET
APA
+5V VCC
+12V
BOOT1
UGATE1
+5V
PHASE1
VCC
LGATE1
OFS
ISEN1ISEN1+
FS
REF
SS
LOAD
IOUT
ISL6313
VID7
VID6
VID5
VID4
VID3
VID2
VID1
VID0
+12V
PVCC
BOOT2
UGATE2
PHASE2
PGOOD
LGATE2
EN
ISEN2ISEN2+
GND
4
FN6448.2
September 2, 2008
ISL6313
Typical Application - ISL6313 with NTC Thermal Compensation
FB
DVC
VSEN
COMP
RGND
+5V VCC
RSET
+12V
APA
BOOT1
UGATE1
+5V
NTC
PHASE1
PLACE IN CLOSE
PROXIMITY
VCC
LGATE1
OFS
ISEN1ISEN1+
FS
REF
SS
LOAD
IOUT
ISL6313
VID7
VID6
VID5
VID4
VID3
VID2
VID1
VID0
+12V
PVCC
BOOT2
UGATE2
PHASE2
PGOOD
LGATE2
EN
ISEN2ISEN2+
GND
5
FN6448.2
September 2, 2008
ISL6313
Absolute Maximum Ratings
Thermal Information
Supply Voltage, VCC . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +6V
Supply Voltage, PVCC . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +15V
BOOT Voltage, VBOOT . . . . . . . . . . . . . . GND - 0.3V to GND + 36V
BOOT to PHASE Voltage, VBOOT-PHASE . . . . . . -0.3V to 15V (DC)
-0.3V to 16V (<10ns, 10µJ)
PHASE Voltage, VPHASE . . . . . . . GND - 0.3V to 15V (PVCC = 12)
GND - 8V (<400ns, 20µJ) to 24V (<200ns, VBOOT - PHASE = 12V)
UGATE Voltage, VUGATE. . . . . . . . VPHASE - 0.3V to VBOOT + 0.3V
VPHASE - 3.5V (<100ns Pulse Width, 2µJ) to VBOOT + 0.3V
LGATE Voltage, VLGATE . . . . . . . . . . . GND - 0.3V to PVCC + 0.3V
GND - 5V (<100ns Pulse Width, 2µJ) to PVCC + 0.3V
Input, Output, or I/O Voltage . . . . . . . . . GND - 0.3V to VCC + 0.3V
Thermal Resistance
θJA (°C/W)
θJC (°C/W)
TQFN Package (Notes 1, 2) . . . . . . . . .
32
2.0
Maximum Junction Temperature . . . . . . . . . . . . . . . . . . . . . . +150°C
Maximum Storage Temperature Range . . . . . . . . . .-65°C to +150°C
Pb-free Reflow Profile . . . . . . . . . . . . . . . . . . . . . . . . .see link below
http://www.intersil.com/pbfree/Pb-FreeReflow.asp
Recommended Operating Conditions
VCC Supply Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .+5V ±5%
PVCC Supply Voltage . . . . . . . . . . . . . . . . . . . . . . . +5V to 12V ±5%
Ambient Temperature
ISL6313CRZ . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 0°C to +70°C
ISL6313IRZ . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .-40°C to +85°C
CAUTION: Do not operate at or near the maximum ratings listed for extended periods of time. Exposure to such conditions may adversely impact product reliability and
result in failures not covered by warranty.
NOTES:
1. θJA is measured in free air with the component mounted on a high effective thermal conductivity test board with “direct attach” features. See
Tech Brief TB379.
2. For θJC, the “case temp” location is the center of the exposed metal pad on the package underside.
3. Limits established by characterization and are not production tested.
Electrical Specifications
Recommended Operating Conditions, Unless Otherwise Specified. Parameters with MIN and/or MAX limits are
100% tested at +25°C, unless otherwise specified. Temperature limits established by characterization and are
not production tested.
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNITS
BIAS SUPPLIES
Input Bias Supply Current
IVCC; EN = high
10
14
17
mA
Gate Drive Bias Current - PVCC Pin
IPVCC; EN = high
2
4.2
6
mA
VCC POR (Power-On Reset)
Threshold
VCC rising
4.25
4.38
4.50
V
VCC falling
3.75
3.87
4.00
V
PVCC POR (Power-On Reset)
Threshold
PVCC rising
4.25
4.38
4.50
V
PVCC falling
3.75
3.87
4.00
V
Oscillator Frequency Accuracy, FSW
(ISL6313CRZ)
RT = 100kΩ (±0.1%)
225
250
275
kHz
Oscillator Frequency Accuracy, FSW
(ISL6313IRZ)
RT = 100kΩ (±0.1%)
215
250
280
kHz
Adjustment Range of Switching
Frequency
(Note 3)
0.08
-
1.0
MHz
Oscillator Ramp Amplitude, VP-P
(Note 3)
-
1.50
-
V
0.84
0.85
0.88
V
-
100
-
mV
System Accuracy (1.000V to 1.600V)
-0.5
-
0.5
%
System Accuracy (0.600V to 1.000V)
-1.0
-
1.0
%
System Accuracy (0.375V to 0.600V)
-2.0
-
2.0
%
-
-
0.4
V
PWM MODULATOR
CONTROL THRESHOLDS
EN Rising Threshold
EN Hysteresis
REFERENCE AND DAC
DAC Input Low Voltage (INTEL)
6
FN6448.2
September 2, 2008
ISL6313
Electrical Specifications
Recommended Operating Conditions, Unless Otherwise Specified. Parameters with MIN and/or MAX limits are
100% tested at +25°C, unless otherwise specified. Temperature limits established by characterization and are
not production tested. (Continued)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNITS
0.8
-
-
V
DAC Input Low Voltage (AMD)
-
-
0.8
V
DAC Input High Voltage (AMD)
1.4
-
-
V
DAC Input High Voltage (INTEL)
PIN-ADJUSTABLE OFFSET
OFS Sink Current Accuracy (Negative
Offset)
ROFS = 32.4kΩ from OFS to VCC
47.0
50.0
53.0
µA
OFS Source Current Accuracy
(Positive Offset)
ROFS = 6.04kΩ from OFS to GND
47.0
50.0
53.0
µA
ERROR AMPLIFIER
DC Gain
RL = 10k to ground, (Note 3)
-
96
-
dB
Gain-Bandwidth Product
CL = 100pF, RL = 10k to ground,
(Note 3)
-
40
-
MHz
Slew Rate
CL = 100pF, Load = ±400µA, (Note 3)
-
20
-
V/µs
Maximum Output Voltage
Load = 1mA
3.90
4.20
-
V
Minimum Output Voltage
Load = -1mA
-
1.30
1.52
V
RS = 100kΩ
-
1.26
-
mV/µs
0.156
-
6.25
mV/µs
SOFT-START RAMP
Soft-Start Ramp Rate
Adjustment Range of Soft-Start Ramp
Rate (Note 3)
CURRENT SENSING
IOUT Current Sense Offset
RSET = 40.2kΩ, VISEN1+ = VISEN2+ = 0V
-2.5
0
2.5
µA
IOUT Current Sense Gain
(ISL6313CRZ)
RSET = 40.2kΩ, VISEN1 = VISEN2 = 24mV
76
80
84
µA
IOUT Current Sense Gain
(ISL6313IRZ)
RSET = 40.2kΩ, VISEN1 = VISEN2 = 24mV
73
80
84
µA
Normal operation (ISL6313CRZ)
88
100
112
µA
Normal operation (ISL6313IRZ)
82
100
118
µA
Dynamic VID change
114
140
166
µA
Normal operation
114
140
166
µA
Dynamic VID change
166
196
226
µA
1.97
2.02
2.07
V
OVERCURRENT PROTECTION
Overcurrent Trip Level - Average
Channel
Overcurrent Trip Level - Individual
Channel
IOUT Pin Overcurrent Trip Level
PROTECTION
Undervoltage Threshold
VSEN falling
VDAC - 325mV
VDAC - 350mV
VDAC - 375mV
V
Undervoltage Hysteresis
VSEN rising
85
100
125
mV
Overvoltage Threshold During
Soft-Start
VR11 and AMD
1.220
1.260
1.300
V
Overvoltage Threshold
VR11, VSEN rising
VDAC + 150mV VDAC + 175mV VDAC + 200mV
V
AMD, VSEN rising
VDAC + 200mV VDAC + 225mV VDAC + 250mV
V
Overvoltage Hysteresis
VSEN falling
7
-
100
-
mV
FN6448.2
September 2, 2008
ISL6313
Electrical Specifications
Recommended Operating Conditions, Unless Otherwise Specified. Parameters with MIN and/or MAX limits are
100% tested at +25°C, unless otherwise specified. Temperature limits established by characterization and are
not production tested. (Continued)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNITS
UGATE Rise Time
tRUGATE; VPVCC = 12V, 3nF load, 10% to
90%
-
26
-
ns
LGATE Rise Time
tRLGATE; VPVCC = 12V, 3nF load, 10% to
90%
-
18
-
ns
UGATE Fall Time
tFUGATE; VPVCC = 12V, 3nF load, 90% to
10%
-
18
-
ns
LGATE Fall Time
tFLGATE; VPVCC = 12V, 3nF load, 90% to
10%
-
12
-
ns
UGATE Turn-On Non-Overlap
tPDHUGATE; VPVCC = 12V, 3nF load,
adaptive
-
10
-
ns
LGATE Turn-On Non-Overlap
tPDHLGATE; VPVCC = 12V, 3nF load,
adaptive
-
10
-
ns
Upper Drive Source Resistance
VPVCC = 12V, 15mA source current
1.25
2.0
3.0
Ω
Upper Drive Sink Resistance
VPVCC = 12V, 15mA sink current
0.9
1.65
3.0
Ω
Lower Drive Source Resistance
VPVCC = 12V, 15mA source current
0.85
1.25
2.2
Ω
Lower Drive Sink Resistance
VPVCC = 12V, 15mA sink current
0.60
0.80
1.35
Ω
Thermal Shutdown Setpoint
-
160
-
°C
Thermal Recovery Setpoint
-
100
-
°C
SWITCHING TIME (Note 3)
GATE DRIVE RESISTANCE (Note 3)
OVER-TEMPERATURE SHUTDOWN (Note 3)
Timing Diagram
tPDHUGATE
tRUGATE
tFUGATE
UGATE
LGATE
tFLGATE
tRLGATE
tPDHLGATE
8
FN6448.2
September 2, 2008
ISL6313
Functional Pin Description
VCC
VCC is the bias supply for the ICs small-signal circuitry.
Connect this pin to a +5V supply and decouple using a
quality 0.1µF ceramic capacitor.
FB and COMP
These pins are the internal error amplifier inverting input and
output respectively. The FB pin, COMP pin, and the VSEN
pins are tied together through external R-C networks to
compensate the regulator.
DVC
PVCC
This pin is the power supply pin for the channel MOSFET
drivers, and can be connected to any voltage from +5V to
+12V depending on the desired MOSFET gate-drive level.
Decouple this pin with a quality 1.0µF ceramic capacitor.
GND
GND is the bias and reference ground for the IC.
EN
A series resistor and capacitor can be connected from the
DVC pin to the FB pin to compensate and smooth dynamic
VID transitions.
IOUT
The IOUT pin is the average channel-current sense output.
This pin is used as a load current indicator to monitor what
the output load current is.
This pin is a threshold-sensitive (approximately 0.85V) enable
input for the controller. Held low, this pin disables controller
operation. Pulled high, the pin enables the controller for
operation.
This pin can also be used to set the overcurrent protection
trip level if it desired that a lower level be used then the
internal trip point. Connecting this pin through a resistor to
ground allows the controller to set the alternate overcurrent
protection trip level.
FS
APA
A resistor, RS, tied to this pin sets the channel switching
frequency of the controller. Refer to Equation 47 for proper
resistor calculation.
This is the Adaptive Phase Alignment set pin. A 100µA
current flows into the APA pin and by tying a resistor from
this pin to COMP the trip level for the Adaptive Phase
Alignment circuitry can be set.
The FS pin also controls whether the internal IAVG current is
connected to the FB pin or not. Tying the RS resistor to
ground connects the IAVG current internally to the FB pin,
allowing the converter to incorporate output voltage droop
proportional to the output current. Tying the RS resistor to
VCC, disconnects the IAVG current internally from the FB
pin.
VID0, VID1, VID2, VID3, VID4, VID5, VID6, and VID7
These are the inputs for the internal DAC that provides the
reference voltage for output regulation. These pins respond to
TTL logic thresholds. These pins are internally pulled high, to
approximately 1.2V, by 40µA internal current sources for Intel
modes of operation, and pulled low by 20µA internal current
sources for AMD modes of operation. The internal pull-up
current decreases to 0 as the VID voltage approaches the
internal pull-up voltage. All VID pins are compatible with
external pull-up voltages not exceeding the IC’s bias voltage
(VCC).
VSEN
This pin senses the microprocessor’s CORE voltage. Connect
this pin to the CORE voltage sense pin or point of the
microprocessor.
RGND
This pin senses the local ground voltage of the
microprocessor and offsets the internal DAC by this sensed
voltage. Connect this pin to the Ground sense pin or point of
the microprocessor.
9
REF
The REF input pin is the positive input of the error amplifier. It
is internally connected to the DAC output through a 2kΩ
resistor. A capacitor is used between the REF pin and ground
to smooth the voltage transition during soft-start and Dynamic
VID transitions. This pin can also be bypassed to RGND if
desired.
RSET
Connect this pin to VCC through a resistor to set the effective
value of the internal RISEN current sense resistors. It is
recommended a 0.1µF ceramic capacitor be placed in
parallel with this resistor for noise immunization.
OFS
The OFS pin provides a means to program a dc current for
generating an offset voltage across the resistor between FB
and VSEN. The offset current is generated via an external
resistor and precision internal voltage references. The polarity
of the offset is selected by connecting the resistor to GND or
VCC. For no offset, the OFS pin should be left unconnected.
ISEN1-, ISEN1+, ISEN2-, and ISEN2+
These pins are used for differentially sensing the
corresponding channel output currents. The sensed currents
are used for channel balancing, protection, and load line
regulation.
Connect ISEN1- and ISEN2- to the node between the RC
sense elements surrounding the inductor of their respective
FN6448.2
September 2, 2008
ISL6313
channel. Tie the ISEN+ pins to the VCORE side of their
corresponding channel’s sense capacitor.
Tying ISEN2- to VCC programs the part for single phase
operation.
only multi-phase converters can accomplish. The ISL6313
controller helps simplify implementation by integrating vital
functions and requiring minimal external components. The
“Block Diagram” on page 3 provides a top level view of
multi-phase power conversion using the ISL6313 controller.
UGATE1 and UGATE2
Connect these pins to the corresponding upper MOSFET
gates. These pins are used to control the upper MOSFETs
and are monitored for shoot-through prevention purposes.
IL1 + IL2 + IL3, 7A/DIV
BOOT1 and BOOT2
IL3, 7A/DIV
These pins provide the bias voltage for the corresponding
upper MOSFET drives. Connect these pins to appropriately
chosen external bootstrap capacitors. Internal bootstrap
diodes connected to the PVCC pin provides the necessary
bootstrap charge.
PWM3, 5V/DIV
IL2, 7A/DIV
PWM2, 5V/DIV
IL1, 7A/DIV
PHASE1 and PHASE2
Connect these pins to the sources of the corresponding
upper MOSFETs. These pins are the return path for the
upper MOSFET drives.
LGATE1 and LGATE2
These pins are used to control the lower MOSFETs. Connect
these pins to the corresponding lower MOSFETs’ gates.
SS
A resistor, RSS, placed from SS to ground or VCC, will set
the soft-start ramp slope. Refer to Equations 20 and 21 for
proper resistor calculation.
The state of the SS pin also selects which of the available DAC
tables will be used to decode the VID inputs and puts the
controller into the corresponding mode of operation. For Intel
VR11 mode of operation the RSS resistor should be tied to
ground. AMD compliance is selected if the RSS resistor is tied
to VCC.
PGOOD
For Intel mode of operation, PGOOD indicates whether VSEN
is within specified overvoltage and undervoltage limits after a
fixed delay from the end of soft-start. If VSEN exceeds these
limits, an overcurrent event occurs, or if the part is disabled,
PGOOD is pulled low. PGOOD is always low prior to the end
of soft-start.
For AMD modes of operation, PGOOD will always be high
as long as VSEN is within the specified undervoltage,
overvoltage window and soft-start has ended. PGOOD only
goes low if VSEN is outside this window.
Operation
Multiphase Power Conversion
Microprocessor load current profiles have changed to the
point that using single-phase regulators is no longer a viable
solution. Designing a regulator that is cost-effective,
thermally sound, and efficient has become a challenge that
10
PWM1, 5V/DIV
1µs/DIV
FIGURE 1. PWM AND INDUCTOR-CURRENT WAVEFORMS
FOR 3-PHASE CONVERTER
Interleaving
The switching of each channel in a multi-phase converter is
timed to be symmetrically out of phase with each of the other
channels. In a 3-phase converter, each channel switches 1/3
cycle after the previous channel and 1/3 cycle before the
following channel. As a result, the three-phase converter has
a combined ripple frequency three times greater than the
ripple frequency of any one phase. In addition, the
peak-to-peak amplitude of the combined inductor currents is
reduced in proportion to the number of phases (Equations 1
and 2). Increased ripple frequency and lower ripple
amplitude mean that the designer can use less per-channel
inductance and lower total output capacitance for any
performance specification.
Figure 1 illustrates the multiplicative effect on output ripple
frequency. The three channel currents (IL1, IL2, and IL3)
combine to form the AC ripple current and the DC load
current. The ripple component has 3x the ripple frequency of
each individual channel current. Each PWM pulse is
terminated 1/3 of a cycle after the PWM pulse of the previous
phase. The peak-to-peak current for each phase is about 7A,
and the DC components of the inductor currents combine to
feed the load.
To understand the reduction of ripple current amplitude in the
multi-phase circuit, examine the equation representing an
individual channel peak-to-peak inductor current.
( V IN – V OUT ) ⋅ V OUT
I PP = --------------------------------------------------------L ⋅ fS ⋅ V
(EQ. 1)
IN
In Equation 1, VIN and VOUT are the input and output
voltages respectively, L is the single-channel inductor value,
and fS is the switching frequency.
FN6448.2
September 2, 2008
ISL6313
The output capacitors conduct the ripple component of the
inductor current. In the case of multi-phase converters, the
capacitor current is the sum of the ripple currents from each
of the individual channels. Compare Equation 1 to the
expression for the peak-to-peak current after the summation
of N symmetrically phase-shifted inductor currents in
Equation 2. Peak-to-peak ripple current decreases by an
amount proportional to the number of channels. Output
voltage ripple is a function of capacitance, capacitor
equivalent series resistance (ESR), and inductor ripple
current. Reducing the inductor ripple current allows the
designer to use fewer or less costly output capacitors.
( V IN – N ⋅ V OUT ) ⋅ V OUT
I C ( P-P ) = ------------------------------------------------------------------L ⋅ fS ⋅ V
(EQ. 2)
IN
Another benefit of interleaving is to reduce input ripple
current. Input capacitance is determined in part by the
maximum input ripple current. Multiphase topologies can
improve overall system cost and size by lowering input ripple
current and allowing the designer to reduce the cost of input
capacitance. The example in Figure 2 illustrates input
currents from a three-phase converter combining to reduce
the total input ripple current.
The converter depicted in Figure 2 delivers 1.5V to a 36A load
from a 12V input. The RMS input capacitor current is 5.9A.
Compare this to a single-phase converter also stepping down
12V to 1.5V at 36A. The single-phase converter has
11.9ARMS input capacitor current. The single-phase converter
must use an input capacitor bank with twice the RMS current
capacity as the equivalent three-phase converter.
INPUT-CAPACITOR CURRENT, 10A/DIV
which a PWM signal can occur is generated by an internal
clock, whose cycle time is the inverse of the switching
frequency set by the resistor connected to the FS pin. The
advantage of Intersil’s proprietary Active Pulse Positioning
(APP) modulator is that the PWM signal has the ability to
turn on at any point during this PWM time interval, and turn
off immediately after the PWM signal has transitioned high.
This is important because it allows the controller to quickly
respond to output voltage drops associated with current load
spikes, while avoiding the ring back affects associated with
other modulation schemes.
The PWM output state is driven by the position of the error
amplifier output signal, VCOMP minus the current correction
signal relative to the proprietary modulator ramp waveform
as illustrated in Figure 4. At the beginning of each PWM time
interval, this modified VCOMP signal is compared to the
internal modulator waveform. As long as the modified
VCOMP voltage is lower then the modulator waveform
voltage, the PWM signal is commanded low. The internal
MOSFET driver detects the low state of the PWM signal and
turns off the upper MOSFET and turns on the lower
synchronous MOSFET. When the modified VCOMP voltage
crosses the modulator ramp, the PWM output transitions
high, turning off the synchronous MOSFET and turning on
the upper MOSFET. The PWM signal will remain high until
the modified VCOMP voltage crosses the modulator ramp
again. When this occurs the PWM signal will transition low
again.
During each PWM time interval the PWM signal can only
transition high once. Once PWM transitions high it can not
transition high again until the beginning of the next PWM
time interval. This prevents the occurrence of double PWM
pulses occurring during a single period.
CHANNEL 3
INPUT CURRENT
EXTERNAL CIRCUIT
ISL6313 INTERNAL CIRCUIT
APA
CHANNEL 2
INPUT CURRENT
100µA
CAPA
RAPA
+
APA
VAPA,TRIP
+
-
CHANNEL 1
INPUT CURRENT
LOW
PASS
FILTER
COMP
TO APA
CIRCUITRY
1µs/DIV
+
ERROR
AMPLIFIER
-
FIGURE 2. CHANNEL INPUT CURRENTS AND
INPUT-CAPACITOR RMS CURRENT FOR
3-PHASE CONVERTER
FIGURE 3. ADAPTIVE PHASE ALIGNMENT DETECTION
Active Pulse Positioning (APP) Modulated PWM
Operation
Adaptive Phase Alignment (APA)
The ISL6313 uses a proprietary Active Pulse Positioning
(APP) modulation scheme to control the internal PWM
signals that command each channel’s driver to turn their
upper and lower MOSFETs on and off. The time interval in
To further improve the transient response, the ISL6313 also
implements Intersil’s proprietary Adaptive Phase Alignment
(APA) technique, which turns on all of the channels together
at the same time during large current step transient events.
11
FN6448.2
September 2, 2008
ISL6313
As Figure 3 shows, the APA circuitry works by monitoring the
voltage on the APA pin and comparing it to a filtered copy of
the voltage on the COMP pin. The voltage on the APA pin is
a copy of the COMP pin voltage that has been negatively
offset. If the APA pin exceeds the filtered COMP pin voltage
an APA event occurs and all of the channels are forced on.
The APA trip level is the amount of DC offset between the
COMP pin and the APA pin. This is the voltage excursion
that the APA and COMP pin must have during a transient
event to activate the Adaptive Phase Alignment circuitry.
This APA trip level is set through a resistor, RAPA, that
connects from the APA pin to the COMP pin. A 100µA
current flows across RAPA into the APA pin to set the APA
trip level as described in Equation 3. An APA trip level of
500mV is recommended for most applications. A 1000pF
capacitor, CAPA, should also be placed across the RAPA
resistor to help with noise immunity.
V APA ( TRIP ) = R APA ⋅ 100 × 10
–6
+
VCOMP
∑
FILTER
PWM1
+
MODULATOR
RAMP
WAVEFORM
-
TO GATE
CONTROL
LOGIC
-
f(s)
IER
∑ -
IAVG
∑
÷2
+
I2
+
+
I1
FIGURE 4. CHANNEL-1 PWM FUNCTION AND
CURRENT-BALANCE ADJUSTMENT
The filtered error signal modifies the pulse width
commanded by VCOMP to correct any unbalance and force
IER toward zero. The same method for error signal
correction is applied to each active channel.
(EQ. 3)
Number of Active Channels
The default number of active channels on the ISL6313 is two
for 2-phase operation. If single phase operation is desired
the ISEN2- pin should be tied to the VCC pin. This will
disable Channel 2, so only Channel 1 will fire. In single
phase operation all of the Channel 2 pins should be left
unconnected including the PHASE2, LGATE2, UGATE2,
BOOT2, and ISEN2+ pins.
Channel-Current Balance
One important benefit of multi-phase operation is the thermal
advantage gained by distributing the dissipated heat over
multiple devices and greater area. By doing this the designer
avoids the complexity of driving parallel MOSFETs and the
expense of using expensive heat sinks and exotic magnetic
materials.
In order to realize the thermal advantage, it is important that
each channel in a multi-phase converter be controlled to
carry equal amounts of current at any load level. To achieve
this, the currents through each channel must be sensed
continuously every switching cycle. The sensed currents, In,
from each active channel are summed together and divided
by the number of active channels. The resulting cycle
average current, IAVG, provides a measure of the total
load-current demand on the converter during each switching
cycle. Channel-current balance is achieved by comparing
the sensed current of each channel to the cycle average
current, and making the proper adjustment to each channel
pulse width based on the error. Intersil’s patented
current-balance method is illustrated in Figure 4, with error
correction for Channel 1 represented. In the figure, the cycle
average current, IAVG, is compared with the Channel 1
sensed current, I1 , to create an error signal IER.
12
PWM
SWITCHING PERIOD
IL
ISEN
TIME
FIGURE 5. CONTINUOUS CURRENT SAMPLING
Continuous Current Sensing
In order to realize proper current-balance, the currents in
each channel are sensed continuously every switching
cycle. During this time the current-sense amplifier uses the
ISEN inputs to reproduce a signal proportional to the
inductor current, IL. This sensed current, ISEN, is simply a
scaled version of the inductor current.
The ISL6313 supports inductor DCR current sensing to
continuously sense each channel’s current for
channel-current balance. The internal circuitry, shown in
Figure 6 represents channel n of an N-Channel converter.
This circuitry is repeated for each channel in the converter,
but may not be active depending on how many channels are
operating.
FN6448.2
September 2, 2008
ISL6313
VIN
I
UGATE
by the value of the effective sense resistance, RISEN, and
the DCR of the inductor.
L
L
DCR
MOSFET
INDUCTOR
+
+
VL(s)
R1
In
COUT
VC(s)
-
LGATE
-
DRIVER
C1
ISL6313 INTERNAL
CIRCUIT
SENSE
-
VC(s)
RISEN
ISEN
-
+
+
DCR
I SEN = I L ⋅ -----------------R
VOUT
ISEN-
(EQ. 6)
ISEN
The effective internal RISEN resistance is important to the
current sensing process because it sets the gain of the load
line regulation loop as well as the gain of the channel-current
balance loop and the overcurrent trip level. The effective
internal RISEN resistance is user programmable and is set
through use of the RSET pin. Placing a single resistor, RSET,
from the RSET pin to the VCC pin programs the effective
internal RISEN resistance according to Equation 7. It is
important to note that the RSET resistance value must be
between 20kΩ and 80kΩ for Equation 7 to be valid.
ISEN+
3
R ISEN = ---------- ⋅ R SET
400
(EQ. 7)
*Note: RSET must be between 20kΩ and 80kΩ
RSET
VCC
RSET
Output Voltage Setting
FIGURE 6. INDUCTOR DCR CURRENT SENSING
CONFIGURATION
Inductor windings have a characteristic distributed
resistance or DCR (Direct Current Resistance). For
simplicity, the inductor DCR is considered as a separate
lumped quantity, as shown in Figure 6. The channel current
IL, flowing through the inductor, passes through the DCR.
Equation 4 shows the s-domain equivalent voltage, VL,
across the inductor.
V L ( s ) = I L ⋅ ( s ⋅ L + DCR )
(EQ. 4)
A simple R-C network across the inductor (R1 and C)
extracts the DCR voltage, as shown in Figure 6. The voltage
across the sense capacitor, VC, can be shown to be
proportional to the channel current IL, shown in Equation 5.
s⋅L
⎛ ------------+ 1⎞
⎝ DCR
⎠
V C ( s ) = ----------------------------------------- ⋅ DCR ⋅ I L
( s ⋅ R1 ⋅ C1 + 1 )
(EQ. 5)
If the R1-C1 network components are selected such that
their time constant matches the inductor L/DCR time
constant, then VC is equal to the voltage drop across the
DCR.
The capacitor voltage VC, is then replicated across the
effective internal sense resistance RISEN. This develops a
current through RISEN which is proportional to the inductor
current. This current, ISEN, is continuously sensed and is
then used by the controller for load-line regulation,
channel-current balancing, and overcurrent detection and
limiting. Equation 6 shows that the proportion between the
channel-current, IL, and the sensed current, ISEN, is driven
13
The ISL6313 uses a digital to analog converter (DAC) to
generate a reference voltage based on the logic signals at
the VID pins. The DAC decodes the logic signals into one of
the discrete voltages shown in Tables 2, 3 or 4. In the Intel
VR11 mode of operation, each VID pin is pulled up to an
internal 1.2V voltage by a weak current source (40µA),
which decreases to 0A as the voltage at the VID pin varies
from 0 to the internal 1.2V pull-up voltage. In AMD modes of
operation the VID pins are pulled low by a week 20µA
current source. External pull-up resistors or active-high
output stages can augment the pull-up current sources, up to
a voltage of 5V.
The ISL6313 accommodates three different DAC ranges:
Intel VR11, AMD K8/K9 5-bit, and AMD 6-bit. The state of
the SS and VID7 pins decide which DAC version is active.
Refer to Table 1 for a description of how to select the desired
DAC version.
.
TABLE 1. ISL6313 DAC SELECT TABLE
DAC VERSION
SS PIN
VID7 PIN
INTEL VR11
RSS resistor tied to
GND
-
AMD 5-bit
RSS resistor tied to
VCC
high
AMD 6-bit
RSS resistor tied to
VCC
low
TABLE 2. VR11 VOLTAGE IDENTIFICATION CODES
VID7
VID6
VID5
VID4
VID3
VID2
VID1
VID0
VDAC
0
0
0
0
0
0
0
0
OFF
0
0
0
0
0
0
0
1
OFF
0
0
0
0
0
0
1
0
1.60000
0
0
0
0
0
0
1
1
1.59375
FN6448.2
September 2, 2008
ISL6313
TABLE 2. VR11 VOLTAGE IDENTIFICATION CODES (Continued)
TABLE 2. VR11 VOLTAGE IDENTIFICATION CODES (Continued)
VID7
VID6
VID5
VID4
VID3
VID2
VID1
VID0
VDAC
VID7
VID6
VID5
VID4
VID3
VID2
VID1
VID0
VDAC
0
0
0
0
0
1
0
0
1.58750
0
0
1
0
1
1
0
0
1.33750
0
0
0
0
0
1
0
1
1.58125
0
0
1
0
1
1
0
1
1.33125
0
0
0
0
0
1
1
0
1.57500
0
0
1
0
1
1
1
0
1.32500
0
0
0
0
0
1
1
1
1.56875
0
0
1
0
1
1
1
1
1.31875
0
0
0
0
1
0
0
0
1.56250
0
0
1
1
0
0
0
0
1.31250
0
0
0
0
1
0
0
1
1.55625
0
0
1
1
0
0
0
1
1.30625
0
0
0
0
1
0
1
0
1.55000
0
0
1
1
0
0
1
0
1.30000
0
0
0
0
1
0
1
1
1.54375
0
0
1
1
0
0
1
1
1.29375
0
0
0
0
1
1
0
0
1.53750
0
0
1
1
0
1
0
0
1.28750
0
0
0
0
1
1
0
1
1.53125
0
0
1
1
0
1
0
1
1.28125
0
0
0
0
1
1
1
0
1.52500
0
0
1
1
0
1
1
0
1.27500
0
0
0
0
1
1
1
1
1.51875
0
0
1
1
0
1
1
1
1.26875
0
0
0
1
0
0
0
0
1.51250
0
0
1
1
1
0
0
0
1.26250
0
0
0
1
0
0
0
1
1.50625
0
0
1
1
1
0
0
1
1.25625
0
0
0
1
0
0
1
0
1.50000
0
0
1
1
1
0
1
0
1.25000
0
0
0
1
0
0
1
1
1.49375
0
0
1
1
1
0
1
1
1.24375
0
0
0
1
0
1
0
0
1.48750
0
0
1
1
1
1
0
0
1.23750
0
0
0
1
0
1
0
1
1.48125
0
0
1
1
1
1
0
1
1.23125
0
0
0
1
0
1
1
0
1.47500
0
0
1
1
1
1
1
0
1.22500
0
0
0
1
0
1
1
1
1.46875
0
0
1
1
1
1
1
1
1.21875
0
0
0
1
1
0
0
0
1.46250
0
1
0
0
0
0
0
0
1.21250
0
0
0
1
1
0
0
1
1.45625
0
1
0
0
0
0
0
1
1.20625
0
0
0
1
1
0
1
0
1.45000
0
1
0
0
0
0
1
0
1.20000
0
0
0
1
1
0
1
1
1.44375
0
1
0
0
0
0
1
1
1.19375
0
0
0
1
1
1
0
0
1.43750
0
1
0
0
0
1
0
0
1.18750
0
0
0
1
1
1
0
1
1.43125
0
1
0
0
0
1
0
1
1.18125
0
0
0
1
1
1
1
0
1.42500
0
1
0
0
0
1
1
0
1.17500
0
0
0
1
1
1
1
1
1.41875
0
1
0
0
0
1
1
1
1.16875
0
0
1
0
0
0
0
0
1.41250
0
1
0
0
1
0
0
0
1.16250
0
0
1
0
0
0
0
1
1.40625
0
1
0
0
1
0
0
1
1.15625
0
0
1
0
0
0
1
0
1.40000
0
1
0
0
1
0
1
0
1.15000
0
0
1
0
0
0
1
1
1.39375
0
1
0
0
1
0
1
1
1.14375
0
0
1
0
0
1
0
0
1.38750
0
1
0
0
1
1
0
0
1.13750
0
0
1
0
0
1
0
1
1.38125
0
1
0
0
1
1
0
1
1.13125
0
0
1
0
0
1
1
0
1.37500
0
1
0
0
1
1
1
0
1.12500
0
0
1
0
0
1
1
1
1.36875
0
1
0
0
1
1
1
1
1.11875
0
0
1
0
1
0
0
0
1.36250
0
1
0
1
0
0
0
0
1.11250
0
0
1
0
1
0
0
1
1.35625
0
1
0
1
0
0
0
1
1.10625
0
0
1
0
1
0
1
0
1.35000
0
1
0
1
0
0
1
0
1.10000
0
0
1
0
1
0
1
1
1.34375
0
1
0
1
0
0
1
1
1.09375
14
FN6448.2
September 2, 2008
ISL6313
TABLE 2. VR11 VOLTAGE IDENTIFICATION CODES (Continued)
TABLE 2. VR11 VOLTAGE IDENTIFICATION CODES (Continued)
VID7
VID6
VID5
VID4
VID3
VID2
VID1
VID0
VDAC
VID7
VID6
VID5
VID4
VID3
VID2
VID1
VID0
VDAC
0
1
0
1
0
1
0
0
1.08750
0
1
1
1
1
1
0
0
0.83750
0
1
0
1
0
1
0
1
1.08125
0
1
1
1
1
1
0
1
0.83125
0
1
0
1
0
1
1
0
1.07500
0
1
1
1
1
1
1
0
0.82500
0
1
0
1
0
1
1
1
1.06875
0
1
1
1
1
1
1
1
0.81875
0
1
0
1
1
0
0
0
1.06250
1
0
0
0
0
0
0
0
0.81250
0
1
0
1
1
0
0
1
1.05625
1
0
0
0
0
0
0
1
0.80625
0
1
0
1
1
0
1
0
1.05000
1
0
0
0
0
0
1
0
0.80000
0
1
0
1
1
0
1
1
1.04375
1
0
0
0
0
0
1
1
0.79375
0
1
0
1
1
1
0
0
1.03750
1
0
0
0
0
1
0
0
0.78750
0
1
0
1
1
1
0
1
1.03125
1
0
0
0
0
1
0
1
0.78125
0
1
0
1
1
1
1
0
1.02500
1
0
0
0
0
1
1
0
0.77500
0
1
0
1
1
1
1
1
1.01875
1
0
0
0
0
1
1
1
0.76875
0
1
1
0
0
0
0
0
1.01250
1
0
0
0
1
0
0
0
0.76250
0
1
1
0
0
0
0
1
1.00625
1
0
0
0
1
0
0
1
0.75625
0
1
1
0
0
0
1
0
1.00000
1
0
0
0
1
0
1
0
0.75000
0
1
1
0
0
0
1
1
0.99375
1
0
0
0
1
0
1
1
0.74375
0
1
1
0
0
1
0
0
0.98750
1
0
0
0
1
1
0
0
0.73750
0
1
1
0
0
1
0
1
0.98125
1
0
0
0
1
1
0
1
0.73125
0
1
1
0
0
1
1
0
0.97500
1
0
0
0
1
1
1
0
0.72500
0
1
1
0
0
1
1
1
0.96875
1
0
0
0
1
1
1
1
0.71875
0
1
1
0
1
0
0
0
0.96250
1
0
0
1
0
0
0
0
0.71250
0
1
1
0
1
0
0
1
0.95625
1
0
0
1
0
0
0
1
0.70625
0
1
1
0
1
0
1
0
0.95000
1
0
0
1
0
0
1
0
0.70000
0
1
1
0
1
0
1
1
0.94375
1
0
0
1
0
0
1
1
0.69375
0
1
1
0
1
1
0
0
0.93750
1
0
0
1
0
1
0
0
0.68750
0
1
1
0
1
1
0
1
0.93125
1
0
0
1
0
1
0
1
0.68125
0
1
1
0
1
1
1
0
0.92500
1
0
0
1
0
1
1
0
0.67500
0
1
1
0
1
1
1
1
0.91875
1
0
0
1
0
1
1
1
0.66875
0
1
1
1
0
0
0
0
0.91250
1
0
0
1
1
0
0
0
0.66250
0
1
1
1
0
0
0
1
0.90625
1
0
0
1
1
0
0
1
0.65625
0
1
1
1
0
0
1
0
0.90000
1
0
0
1
1
0
1
0
0.65000
0
1
1
1
0
0
1
1
0.89375
1
0
0
1
1
0
1
1
0.64375
0
1
1
1
0
1
0
0
0.88750
1
0
0
1
1
1
0
0
0.63750
0
1
1
1
0
1
0
1
0.88125
1
0
0
1
1
1
0
1
0.63125
0
1
1
1
0
1
1
0
0.87500
1
0
0
1
1
1
1
0
0.62500
0
1
1
1
0
1
1
1
0.86875
1
0
0
1
1
1
1
1
0.61875
0
1
1
1
1
0
0
0
0.86250
1
0
1
0
0
0
0
0
0.61250
0
1
1
1
1
0
0
1
0.85625
1
0
1
0
0
0
0
1
0.60625
0
1
1
1
1
0
1
0
0.85000
1
0
1
0
0
0
1
0
0.60000
0
1
1
1
1
0
1
1
0.84375
1
0
1
0
0
0
1
1
0.59375
15
FN6448.2
September 2, 2008
ISL6313
TABLE 2. VR11 VOLTAGE IDENTIFICATION CODES (Continued)
VID7
VID6
VID5
VID4
VID3
VID2
VID1
VID0
VDAC
1
0
1
0
0
1
0
0
0.58750
1
0
1
0
0
1
0
1
0.58125
1
0
1
0
0
1
1
0
0.57500
1
0
1
0
0
1
1
1
0.56875
1
0
1
0
1
0
0
0
0.56250
1
0
1
0
1
0
0
1
0.55625
1
0
1
0
1
0
1
0
0.55000
1
0
1
0
1
0
1
1
0.54375
1
0
1
0
1
1
0
0
0.53750
1
0
1
0
1
1
0
1
0.53125
1
0
1
0
1
1
1
0
0.52500
1
0
1
0
1
1
1
1
0.51875
1
0
1
1
0
0
0
0
0.51250
1
0
1
1
0
0
0
1
0.50625
1
0
1
1
0
0
1
0
0.50000
1
1
1
1
1
1
1
0
OFF
1
1
1
1
1
1
1
1
OFF
TABLE 3. AMD 5-BIT VOLTAGE IDENTIFICATION
CODES
TABLE 3. AMD 5-BIT VOLTAGE IDENTIFICATION
CODES (Continued)
VID4
VID3
VID2
VID1
VID0
VDAC
0
1
0
1
1
1.275
0
1
0
1
0
1.300
0
1
0
0
1
1.325
0
1
0
0
0
1.350
0
0
1
1
1
1.375
0
0
1
1
0
1.400
0
0
1
0
1
1.425
0
0
1
0
0
1.450
0
0
0
1
1
1.475
0
0
0
1
0
1.500
0
0
0
0
1
1.525
0
0
0
0
0
1.550
TABLE 4. AMD 6-BIT VOLTAGE IDENTIFICATION
CODES
VID5
VID4
VID3
VID2
VID1
VID0
VDAC
0
0
0
0
0
0
1.5500
0
0
0
0
0
1
1.5250
0
0
0
0
1
0
1.5000
0
0
0
0
1
1
1.4750
VID4
VID3
VID2
VID1
VID0
VDAC
0
0
0
1
0
0
1.4500
1
1
1
1
1
Off
0
0
0
1
0
1
1.4250
1
1
1
1
0
0.800
0
0
0
1
1
0
1.4000
1
1
1
0
1
0.825
0
0
0
1
1
1
1.3750
1
1
1
0
0
0.850
0
0
1
0
0
0
1.3500
1
1
0
1
1
0.875
0
0
1
0
0
1
1.3250
1
1
0
1
0
0.900
0
0
1
0
1
0
1.3000
0
1
0
1
1
1.2750
1
1
0
0
1
0.925
0
1
1
0
0
0
0.950
0
0
1
1
0
0
1.2500
1
0
1
1
1
0.975
0
0
1
1
0
1
1.2250
1
0
1
1
0
1.000
0
0
1
1
1
0
1.2000
1
0
1
0
1
1.025
0
0
1
1
1
1
1.1750
0
1
0
0
0
0
1.1500
1
0
1
0
0
1.050
0
1
0
0
0
1
1.1250
1
0
0
1
1
1.075
0
1
0
0
1
0
1.1000
1
0
0
1
0
1.100
0
1
0
0
1
1
1.0750
1
0
0
0
1
1.125
0
1
0
1
0
0
1.0500
1
0
0
0
0
1.150
0
1
0
1
0
1
1.0250
0
1
1
1
1
1.175
0
1
0
1
1
0
1.0000
0
1
1
1
0
1.200
0
1
0
1
1
1
0.9750
0
1
1
0
1
1.225
0
1
1
0
0
0
0.9500
0
1
1
0
0
1.250
0
1
1
0
0
1
0.9250
16
FN6448.2
September 2, 2008
ISL6313
Voltage Regulation
TABLE 4. AMD 6-BIT VOLTAGE IDENTIFICATION
CODES (Continued)
The integrating compensation network shown in Figure 7
insures that the steady-state error in the output voltage is
limited only to the error in the reference voltage (output of
the DAC) and offset errors in the OFS current source,
remote-sense and error amplifiers. Intersil specifies the
guaranteed tolerance of the ISL6313 to include the
combined tolerances of each of these elements.
VID5
VID4
VID3
VID2
VID1
VID0
VDAC
0
1
1
0
1
0
0.9000
0
1
1
0
1
1
0.8750
0
1
1
1
0
0
0.8500
0
1
1
1
0
1
0.8250
0
1
1
1
1
0
0.8000
0
1
1
1
1
1
0.7750
1
0
0
0
0
0
0.7625
1
0
0
0
0
1
0.7500
1
0
0
0
1
0
0.7375
1
0
0
0
1
1
0.7250
1
0
0
1
0
0
0.7125
The output of the error amplifier, VCOMP, is used by the
modulator to generate the PWM signals. The PWM signals
control the timing of the Internal MOSFET drivers and
regulate the converter output so that the voltage at FB is
equal to the voltage at REF. This will regulate the output
voltage to be equal to Equation 8. The internal and external
circuitry that controls voltage regulation is illustrated in
Figure 7.
1
0
0
1
0
1
0.7000
V OUT = V REF – V OFS – V DROOP
1
0
0
1
1
0
0.6875
1
0
0
1
1
1
0.6750
1
0
1
0
0
0
0.6625
1
0
1
0
0
1
0.6500
1
0
1
0
1
0
0.6375
1
0
1
0
1
1
0.6250
1
0
1
1
0
0
0.6125
1
0
1
1
0
1
0.6000
1
0
1
1
1
0
0.5875
1
0
1
1
1
1
0.5750
1
1
0
0
0
0
0.5625
1
1
0
0
0
1
0.5500
1
1
0
0
1
0
0.5375
(EQ. 8)
The ISL6313 incorporates differential remote-sense
amplification in the feedback path. The differential sensing
removes the voltage error encountered when measuring the
output voltage relative to the controller ground reference
point resulting in a more accurate means of sensing output
voltage.
EXTERNAL CIRCUIT
ISL6313 INTERNAL CIRCUIT
COMP
CC
IAVG
RC
IOFS
FB
-
REF
1
1
0
0
1
1
0.5250
1
1
0
1
0
0
0.5125
1
1
0
1
0
1
0.5000
+
CREF
RFB
+
(VDROOP + VOFS)
-
VCOMP
ERROR
AMPLIFIER
2k
1
1
0
1
1
0
0.4875
1
1
0
1
1
1
0.4750
1
1
1
0
0
0
0.4625
1
1
1
0
0
1
0.4500
1
1
1
0
1
0
0.4375
1
1
1
0
1
1
0.4250
1
1
1
1
0
0
0.4125
1
1
1
1
0
1
0.4000
Load-Line (Droop) Regulation
1
1
1
1
1
0
0.3875
1
1
1
1
1
1
0.3750
Some microprocessor manufacturers require a
precisely-controlled output resistance. This dependence of
output voltage on load current is often termed “droop” or
“load line” regulation. By adding a well controlled output
impedance, the output voltage can effectively be level shifted
in a direction which works to achieve the load-line regulation
required by these manufacturers.
17
VSEN
+
VOUT
-
∑
+
VID
DAC
+
RGND
FIGURE 7. OUTPUT VOLTAGE AND LOAD-LINE
REGULATION WITH OFFSET ADJUSTMENT
FN6448.2
September 2, 2008
ISL6313
In other cases, the designer may determine that a more
cost-effective solution can be achieved by adding droop.
Droop can help to reduce the output-voltage spike that
results from fast load-current demand changes.
For Negative Offset (connect ROFS to VCC):
The magnitude of the spike is dictated by the ESR and ESL
of the output capacitors selected. By positioning the no-load
voltage level near the upper specification limit, a larger
negative spike can be sustained without crossing the lower
limit. By adding a well controlled output impedance, the
output voltage under load can effectively be level shifted
down so that a larger positive spike can be sustained without
crossing the upper specification limit.
For Positive Offset (connect ROFS to GND):
As shown in Figure 7, a current proportional to the average
current of all active channels, IAVG, flows from FB through a
load-line regulation resistor RFB. The resulting voltage drop
across RFB is proportional to the output current, effectively
creating an output voltage droop with a steady-state value
defined as Equation 9:
V DROOP = I AVG ⋅ R FB
1.6 ⋅ R FB
R OFS = -------------------------V OFFSET
(EQ. 12)
0.3 ⋅ R FB
R OFS = -------------------------V OFFSET
(EQ. 13)
VSEN
+
VOFS
-
RFB
VREF
E/A
FB
IOFS
(EQ. 9)
The regulated output voltage is reduced by the droop voltage
VDROOP. The output voltage as a function of load current is
derived by combining Equations 6, 8, and 9.
⎛ I OUT DCR
⎞
V OUT = V REF – V OFS – ⎜ ------------- ⋅ ------------------ ⋅ R FB⎟
N
R
⎝
⎠
ISEN
OFS
ISL6313
ROFS
-
0.3V
GND
VCC
GND
Therefore the equivalent loadline impedance, i.e. droop
impedance, is equal to:
1.6V
+
+
(EQ. 10)
In Equation 10, VREF is the reference voltage, VOFS is the
programmed offset voltage, IOUT is the total output current
of the converter, RISEN is the internal sense resistor
connected to the ISEN+ pin, RFB is the feedback resistor, N
is the active channel number, and DCR is the Inductor DCR
value.
R FB DCR
R LL = ------------ ⋅ -----------------N
R ISEN
-
FIGURE 8. POSITIVE OFFSET OUTPUT VOLTAGE
PROGRAMMING
VSEN
VOFS
+
RFB
VREF
E/A
FB
(EQ. 11)
IOFS
Output-Voltage Offset Programming
The ISL6313 allows the designer to accurately adjust the
offset voltage by connecting a resistor, ROFS, from the OFS
pin to VCC or GND. When ROFS is connected between OFS
and VCC, the voltage across it is regulated to 1.6V. This
causes a proportional current (IOFS) to flow into the OFS pin
and out of the FB pin, providing a negative offset. If ROFS is
connected to ground, the voltage across it is regulated to
0.3V, and IOFS flows into the FB pin and out of the OFS pin,
providing a positive offset. The offset current flowing through
the resistor between VSEN and FB will generate the desired
offset voltage which is equal to the product (IOFS x RFB).
These functions are shown in Figures 8 and 9.
VCC
-
ROFS
+
OFS
ISL6313
-
1.6V
+
0.3V
GND
VCC
FIGURE 9. NEGATIVE OFFSET OUTPUT VOLTAGE
PROGRAMMING
Once the desired output offset voltage has been determined,
use Equations 12 and 13 to set ROFS:
18
FN6448.2
September 2, 2008
ISL6313
Dynamic VID
Compensating Dynamic VID Transitions
Modern microprocessors need to make changes to their core
voltage as part of normal operation. They direct the ISL6313
to do this by making changes to the VID inputs. The ISL6313
is required to monitor the DAC inputs and respond to
on-the-fly VID changes in a controlled manner, supervising a
safe output voltage transition without discontinuity or
disruption. The DAC mode the ISL6313 is operating in,
determines how the controller responds to a dynamic VID
change.
During a VID transition, the resulting change in voltage on
the FB pin and the COMP pin causes an AC current to flow
through the error amplifier compensation components from
the FB to the COMP pin. This current then flows through the
feedback resistor, RFB, and can cause the output voltage to
overshoot or undershoot at the end of the VID transition. In
order to ensure the smooth transition of the output voltage
during a VID change, a VID-on-the-fly compensation
network is required. This network is composed of a resistor
and capacitor in series, RDVC and CDVC, between the DVC
and the FB pin.
INTEL DYNAMIC VID TRANSITIONS
When in Intel VR11 mode the ISL6313 checks the VID inputs
on the positive edge of an internal 5.5MHz clock. If a new
code is established and it remains stable for 3 consecutive
readings (0.36µs to 0.54µs), the ISL6313 recognizes the new
code and changes the internal DAC reference directly to the
new level. The Intel processor controls the VID transitions and
is responsible for incrementing or decrementing one VID step
at a time. In VR11 mode, the ISL6313 will immediately change
the internal DAC reference to the new requested value as
soon as the request is validated, which means the fastest
recommended rate at which a bit change can occur is once
every 1µs. If the VID code is changed by more then one step
at a time, the DAC will try to track it at a 5.5MHz step rate.
This will likely cause an overcurrent or overvoltage fault.
When running in AMD 5-bit or 6-bit modes of operation, the
ISL6313 responds differently to a dynamic VID change then
when in Intel VR11 mode. In the AMD modes the ISL6313 still
checks the VID inputs on the positive edge of an internal
5.5MHz clock. In these modes the VID code can be changed
by more than a 1-bit step at a time. If a new code is
established and it remains stable for 3 consecutive readings
(0.36µs to 0.54µs), the ISL6313 recognizes the change and
begins slewing the DAC in 6.25mV steps at a stepping
frequency of 345kHz until the VID and DAC are equal. Thus,
the total time required for a VID change, tDVID, is dependent
only on the size of the VID change (ΔVVID).
The time required for a ISL6313-based converter in AMD 5-bit
DAC configuration to make a 1.1V to 1.5V reference voltage
change is about 186µs, as calculated using Equation 14.
(EQ. 14)
VID “OFF” DAC CODES
Both the Intel VR11 and the AMD 5-bit VID tables include “Off”
DAC codes, which indicate to the controller to disable all
regulation. Recognition of these codes is slightly different in that
they must be stable for 4 consecutive readings of a 5.5MHz
clock (0.54µs to 0.72µs) to be recognized. Once an “Off” code
is recognized the ISL6313 latches off, and must be reset by
dropping the EN pin.
19
IDVC = IC
IC
IDVC
CC
CDVC
DVC
COMP
FB
x2
REF
+
CREF
VDAC + RGND
RC
RDVC
ERROR
AMPLIFIER
ISL6313 INTERNAL CIRCUIT
FIGURE 10. DYNAMIC VID COMPENSATION NETWORK
AMD DYNAMIC VID TRANSITIONS
ΔV VID
1
t DVID = -------------------------- ⋅ ⎛ ---------------------⎞
3 ⎝ 0.00625⎠
345 × 10
RFB
VSEN
This VID-on-the-fly compensation network works by
sourcing AC current into the FB node to offset the effects of
the AC current flowing from the FB to the COMP pin during a
VID transition. To create this compensation current the
ISL6313 sets the voltage on the DVC pin to be 2x the voltage
on the REF pin. Since the error amplifier forces the voltage
on the FB pin and the REF pin to be equal, the resulting
voltage across the series RC between DVC and FB is equal
to the REF pin voltage. The RC compensation components,
RDVC and CDVC, can then be selected to create the desired
amount of compensation current.
The amount of compensation current required is dependant
on the modulator gain of the system, K1, and the error
amplifier R-C components, RC and CC, that are in series
between the FB and COMP pins. Use Equations 15, 16, and
17 to calculate the RC component values, RDVC and CDVC,
for the VID-on-the-fly compensation network. For these
equations: VIN is the input voltage for the power train; VP-P
is the oscillator ramp amplitude (1.5V); and RC and CC are
the error amplifier R-C components between the FB and
COMP pins.
V IN
K1 = -----------V P-P
K1
A = ----------------K1 – 1
(EQ. 15)
R DVC = A × R C
(EQ. 16)
CC
C DVC = -------A
(EQ. 17)
FN6448.2
September 2, 2008
ISL6313
Advanced Adaptive Zero Shoot-Through Deadtime
Control (Patent Pending)
The integrated drivers incorporate a unique adaptive deadtime
control technique to minimize deadtime, resulting in high
efficiency from the reduced freewheeling time of the lower
MOSFET body-diode conduction, and to prevent the upper and
lower MOSFETs from conducting simultaneously. This is
accomplished by ensuring either rising gate turns on its
MOSFET with minimum and sufficient delay after the other has
turned off.
During turn-off of the lower MOSFET, the PHASE voltage is
monitored until it reaches a -0.3V/+0.8V (forward/reverse
inductor current). At this time the UGATE is released to rise. An
auto-zero comparator is used to correct the rDS(ON) drop in the
phase voltage preventing false detection of the -0.3V phase
level during rDS(ON) conduction period. In the case of zero
current, the UGATE is released after 35ns delay of the LGATE
dropping below 0.5V. When LGATE first begins to transition
low, this quick transition can disturb the PHASE node and
cause a false trip, so there is 20ns of blanking time once
LGATE falls until PHASE is monitored.
Once the PHASE is high, the advanced adaptive
shoot-through circuitry monitors the PHASE and UGATE
voltages during a PWM falling edge and the subsequent
UGATE turn-off. If either the UGATE falls to less than 1.75V
above the PHASE or the PHASE falls to less than +0.8V, the
LGATE is released to turn-on.
The bootstrap capacitor must have a maximum voltage
rating above PVCC + 4V and its capacitance value can be
chosen from Equation 18:
Q GATE
C BOOT_CAP ≥ -------------------------------------ΔV BOOT_CAP
(EQ. 18)
Q G1 ⋅ PVCC
Q GATE = ---------------------------------- ⋅ N Q1
V GS1
where QG1 is the amount of gate charge per upper MOSFET
at VGS1 gate-source voltage and NQ1 is the number of
control MOSFETs. The ΔVBOOT_CAP term is defined as the
allowable droop in the rail of the upper gate drive.
Gate Drive Voltage Versatility
The ISL6313 provides the user flexibility in choosing the
gate drive voltage for efficiency optimization. The controller
ties the upper and lower drive rails together. Simply applying
a voltage from 5V up to 12V on PVCC sets both gate drive
rail voltages simultaneously.
Initialization
Prior to initialization, proper conditions must exist on the EN,
VCC, PVCC and the VID pins. When the conditions are met,
the controller begins soft-start. Once the output voltage is
within the proper window of operation, the controller asserts
PGOOD.
ISL6313 INTERNAL CIRCUIT
EXTERNAL CIRCUIT
Internal Bootstrap Device
All three integrated drivers feature an internal bootstrap
schottky diode. Simply adding an external capacitor across
the BOOT and PHASE pins completes the bootstrap circuit.
The bootstrap function is also designed to prevent the
bootstrap capacitor from overcharging due to the large
negative swing at the PHASE node. This reduces voltage
stress on the boot to phase pins.
VCC
PVCC
POR
CIRCUIT
ENABLE
COMPARATOR
10.7kΩ
EN
+
1.6
+12V
1.4
1.40kΩ
0.85V
CBOOT_CAP (µF)
1.2
1.0
SOFT-START
AND
FAULT LOGIC
0.8
0.6
FIGURE 12. POWER SEQUENCING USING
THRESHOLD-SENSITIVE ENABLE (EN)
FUNCTION
QGATE = 100nC
0.4
50nC
0.2
20nC
0.0
0.0
0.1
Enable and Disable
0.2
0.3
0.4
0.5
0.6
0.7
0.8
0.9
1.0
ΔVBOOT_CAP (V)
FIGURE 11. BOOTSTRAP CAPACITANCE vs BOOT RIPPLE
VOLTAGE
20
While in shutdown mode, the LGATE and UGATE signals
are held low to assure the MOSFETs remain off. The
following input conditions must be met (for both Intel and
AMD modes of operation) before the ISL6313 is released
FN6448.2
September 2, 2008
ISL6313
from shutdown mode to begin the soft-start startup
sequence:
1. The bias voltage applied at VCC must reach the internal
power-on reset (POR) rising threshold. Once this
threshold is reached, proper operation of all aspects of
the ISL6313 is guaranteed. Hysteresis between the rising
and falling thresholds assure that once enabled, the
ISL6313 will not inadvertently turn off unless the bias
voltage drops substantially (see “Electrical
Specifications” on page 6).
2. The voltage on EN must be above 0.85V. The EN input
allows for power sequencing between the controller bias
voltage and another voltage rail. The enable comparator
holds the ISL6313 in shutdown until the voltage at EN
rises above 0.85V. The enable comparator has 110mV of
hysteresis to prevent bounce.
3. The driver bias voltage applied at the PVCC pin must
reach the internal power-on reset (POR) rising threshold.
Hysteresis between the rising and falling thresholds
assure that once enabled, the ISL6313 will not
inadvertently turn off unless the PVCC bias voltage drops
substantially (see “Electrical Specifications” on page 6).
For Intel VR11 and AMD 6-bit modes of operation these are
the only conditions that must be met for the controller to
immediately begin the soft-start sequence. If running in AMD
5-bit mode of operation there is one more condition that
must be met:
4. The VID code must not be 11111 in AMD 5-bit mode. This
code signals the controller that no load is present. The
controller will not allow soft-start to begin if this VID code
is present on the VID pins.
Once all of these conditions are met the controller will begin
the soft-start sequence and will ramp the output voltage up
to the user designated level.
output voltage reaches the VID voltage plus/minus any offset
or droop voltage.
The soft-start time is the sum of the 4 periods as shown in
Equation 19.
t SS = t d1 + t d2 + t d3 + t d4
(EQ. 19)
VOUT, 500mV/DIV
td1
td3
td2
The soft-start function allows the converter to bring up the
output voltage in a controlled fashion, resulting in a linear
ramp-up. The soft-start sequence for the Intel modes of
operation is slightly different then the AMD soft-start
sequence.
For the Intel VR11 mode of operation, the soft-start
sequence if composed of four periods, as shown in
Figure 13. Once the ISL6313 is released from shutdown and
soft-start begins (as described in “Enable and Disable” on
page 20), the controller will have a fixed delay period TD1 of
typically 1.10ms. After this delay period, the VR will begin
first soft-start ramp until the output voltage reaches 1.1V
VBOOT voltage. Then, the controller will regulate the VR
voltage at 1.1V for another fixed delay period td3, of typically
93µs. At the end of td3 period, ISL6313 will read the VID
signals. It is recommended that the VID codes be set no
later then 50µs into period td3. If the VID code is valid,
ISL6313 will initiate the second soft-start ramp until the
21
td5
EN
PGOOD
500µs/DIV
FIGURE 13. SOFT-START WAVEFORMS
During td2 and td4, ISL6313 digitally controls the DAC
voltage change at 6.25mV per step. The time for each step is
determined by the frequency of the soft-start oscillator which
is defined by the resistor RSS on the SS pin. The second
soft-start ramp time td2 and td4 can be calculated based on
Equations 20 and 21:
t d2 = 1.1 ⋅ R SS ⋅ 8 ⋅ 10
–3
( μs )
t d4 = V VID – 1.1 ⋅ R SS ⋅ 8 ⋅ 10
Intel Soft-Start
td4
(EQ. 20)
–3
( μs )
(EQ. 21)
For example, when VID is set to 1.5V and the RSS is set at
100kΩ, the first soft-start ramp time td2 will be 880µs and the
second soft-start ramp time td4 will be 320µs.
After the DAC voltage reaches the final VID setting, PGOOD
will be set to high with the fixed delay td5. The typical value
for td5 is 93µs.
AMD Soft-Start
For the AMD 5-bit and 6-bit modes of operation, the
soft-start sequence is composed of two periods, as shown in
Figure 14. At the beginning of soft-start, the VID code is
immediately obtained from the VID pins, followed by a fixed
delay period tdA of typically 1.10ms. After this delay period
the ISL6313 will begin ramping the output voltage to the
desired DAC level at a fixed rate of 6.25mV per step. The
time for each step is determined by the frequency of the
soft-start oscillator which is defined by the resistor RSS on
the SS pin. The amount of time required to ramp the output
FN6448.2
September 2, 2008
ISL6313
voltage to the final DAC voltage is referred to as tdB, and can
be calculated as shown in Equation 22:
⋅ 8 ⋅ 10
tDB = V VID ⋅ R
SS
–3
( μs )
the output drives are enabled at the end of the soft-start
period, leading to an abrupt correction in the output voltage
down to the DAC-set level.
(EQ. 22)
Fault Monitoring and Protection
At the end of soft-start, PGOOD will immediately go high if
the VSEN voltage is within the undervoltage and overvoltage
limits.
VOUT, 500mV/DIV
The ISL6313 actively monitors output voltage and current to
detect fault conditions. Fault monitors trigger protective
measures to prevent damage to a microprocessor load. One
common power good indicator is provided for linking to
external system monitors. The schematic in Figure 16
outlines the interaction between the fault monitors and the
power good signal.
IAVG
100µA
tdB
tdA
+
OCP
-
+
OCL
-
SS
140µA
REPEAT FOR
EACH CHANNEL
VDAC + RGND
EN
PGOOD
I1
IOUT
+
OCP
-
+175mV
OR
+225mV
VOCP
500µs/DIV
SOFT-START, FAULT
AND CONTROL LOGIC
FIGURE 14. SOFT-START WAVEFORMS
1.260V
Pre-Biased Soft-Start
The ISL6313 also has the ability to start up into a pre-charged
output, without causing any unnecessary disturbance. The FB
pin is monitored during soft-start, and should it be higher than
the equivalent internal ramping reference voltage, the output
drives hold both MOSFETs off. Once the internal ramping
OVP
+
PGOOD
VSEN
UV
OUTPUT PRECHARGED
ABOVE DAC LEVEL
-350mV
+
VDAC + RGND
ISL6313 INTERNAL CIRCUITRY
OUTPUT PRECHARGED
BELOW DAC LEVEL
FIGURE 16. POWER GOOD AND PROTECTION CIRCUITRY
Power Good Signal
GND>
VOUT (0.5V/DIV)
GND>
EN (5V/DIV)
t1 t2
t3
FIGURE 15. SOFT-START WAVEFORMS FOR ISL6313-BASED
MULTI-PHASE CONVERTER
reference exceeds the FB pin potential, the output drives are
enabled, allowing the output to ramp from the pre-charged
level to the final level dictated by the DAC setting. Should the
output be pre-charged to a level exceeding the DAC setting,
22
The power good pin (PGOOD) is an open-drain logic output
that signals whether or not the ISL6313 is regulating the
output voltage within the proper levels, and whether any fault
conditions exist. This pin should be tied through a resistor to
a voltage source that’s equal to or less then VCC.
For Intel mode of operation, PGOOD indicates whether VSEN
is within specified overvoltage and undervoltage limits after a
fixed delay from the end of soft-start. PGOOD transitions low
when an undervoltage, overvoltage, or overcurrent condition
is detected or when the controller is disabled by a reset from
EN, POR, or one of the no-CPU VID codes. In the event of
an overvoltage or overcurrent condition, or a no-CPU VID
code, the controller latches off and PGOOD will not return
high until EN is toggled and a successful soft-start is
FN6448.2
September 2, 2008
ISL6313
completed. In the case of an undervoltage event, PGOOD
will return high when the output voltage rises above the
undervoltage hysteresis level. PGOOD is always low prior to
the end of soft-start.
For AMD modes of operation, PGOOD will always be high
as long as VSEN is within the specified undervoltage,
overvoltage window and soft-start has ended. PGOOD only
goes low if VSEN is outside this window. Even if the
controller is shut down the PGOOD signal will still stay high
until VSEN falls below the undervoltage threshold.
Overvoltage Protection
The ISL6313 constantly monitors the difference between the
VSEN and RGND voltages to detect if an overvoltage event
occurs. During soft-start, while the DAC is ramping up, the
overvoltage trip level is the higher of a fixed voltage 1.260V or
DAC + 175mV for Intel modes of operation and DAC + 225mV
for AMD modes of operation. Upon successful soft-start, the
overvoltage trip level is only DAC + 175mV or DAC + 225mV
depending on whether the controller is running in Intel or AMD
mode. When the output voltage rises above the OVP trip level
actions are taken by the ISL6313 to protect the microprocessor
load.
At the inception of an overvoltage event, LGATE1 and
LGATE2 are commanded high and the PGOOD signal is
driven low. This turns on the all of the lower MOSFETs and
pulls the output voltage below a level that might cause
damage to the load. The LGATE outputs remain high until
VSEN falls 100mV below the OVP threshold that tripped the
overvoltage protection circuitry. The ISL6313 will continue to
protect the load in this fashion as long as the overvoltage
condition recurs. Once an overvoltage condition ends the
ISL6313 latches off, and must be reset by toggling EN, or
through POR, before a soft-start can be reinitiated.
There is an OVP condition that exists that will not latch off the
ISL6313. During a soft-start sequence, if the VSEN voltage is
above the OVP threshold an overvoltage event will occur, but
will be released once VSEN falls 100mV below the OVP
threshold. If VSEN then rises above the OVP trip threshold a
second time, the ISL6313 will be latched off and cannot be
restarted until the controller is reset.
Pre-POR Overvoltage Protection
Prior to PVCC and VCC exceeding their POR levels, the
ISL6313 is designed to protect the load from any overvoltage
events that may occur. This is accomplished by means of an
internal 10kΩ resistor tied from PHASE to LGATE, which
turns on the lower MOSFET to control the output voltage
until the overvoltage event ceases or the input power supply
cuts off. For complete protection, the low side MOSFET
should have a gate threshold well below the maximum
voltage rating of the load/microprocessor.
In the event that during normal operation the PVCC or VCC
voltage falls back below the POR threshold, the pre-POR
23
overvoltage protection circuitry reactivates to protect from
any more pre-POR overvoltage events.
Undervoltage Detection
The undervoltage threshold is set at DAC - 350mV of the
VID code. When the output voltage (VSEN - RGND) is below
the undervoltage threshold, PGOOD gets pulled low. No
other action is taken by the controller. PGOOD will return
high if the output voltage rises above DAC - 250mV.
Open Sense Line Prevention
In the case that either of the remote sense lines, VSEN or
GND, become open, the ISL6313 is designed to prevent the
controller from regulating. This is accomplished by means of
a small 5µA pull-up current on VSEN, and a pull-down
current on RGND. If the sense lines are opened at any time,
the voltage difference between VSEN and RGND will
increase until an overvoltage event occurs, at which point
overvoltage protection activates and the controller stops
regulating. The ISL6313 will be latched off and cannot be
restarted until the controller is reset.
Overcurrent Protection
The ISL6313 takes advantage of the proportionality between
the load current and the average current, IAVG, to detect an
overcurrent condition. Two different methods of detecting
overcurrent events are available on the ISL6313. The first
method continually compares the average sense current
with a constant 100µA OCP reference current as shown in
Figure 16. Once the average sense current exceeds the
OCP reference current, a comparator triggers the converter
to begin overcurrent protection procedures.
For this first method the overcurrent trip threshold is dictated
by the DCR of the inductors, the number of active channels,
and the RSET pin resistor, RSET. To calculate the
overcurrent trip level, IOCP, using this method use
Equation 23, where N is the number of active channels, DCR
is the individual inductor’s DCR, and RSET is the RSET pin
resistor value.
–6
100 ⋅ 10 ⋅ R SET ⋅ N ⋅ 3
I OCP = --------------------------------------------------------------DCR ⋅ 400
(EQ. 23)
During VID-on-the-fly transitions the overcurrent trip level for
this method is boosted to prevent false overcurrent trip
events that can occur. Starting from the beginning of a
dynamic VID transition, the overcurrent trip level is boosted
to 140µA. The OCP level will stay at this boosted level until
50µs after the end of the dynamic VID transition, at which
point it will return to the typical 100µA trip level.
The second method for detecting overcurrent events
continuously compares the voltage on the IOUT pin, VIOUT,
to the overcurrent protection voltage, VOCP, as shown in
Figure 16. The average channel sense current flows out the
IOUT pin and through RIOUT, creating the IOUT pin voltage
which is proportional to the output current. When the IOUT
FN6448.2
September 2, 2008
ISL6313
pin voltage exceeds the VOCP voltage of 2.0V, the
overcurrent protection circuitry activates. Since the IOUT pin
voltage is proportional to the output current, the overcurrent
trip level, IOCP, can be set by selecting the proper value for
RIOUT, as shown in Equation 24.
6 ⋅ R SET ⋅ N
I OCP = --------------------------------------------------DCR ⋅ R IOUT ⋅ 400
(EQ. 24)
Once the output current exceeds the overcurrent trip level,
VIOUT will exceed VOCP and a comparator will trigger the
converter to begin overcurrent protection procedures.
At the beginning of an overcurrent shutdown, the controller
turns off both upper and lower MOSFETs and lowers
PGOOD. The controller will then immediately attempt to softstart. If the overcurrent fault remains, the trip-retry cycles will
continue until either the controller is disabled or the fault is
cleared. If five overcurrent events occur without successfully
completing soft-start, the controller will latch off after the fifth
try and must be reset by toggling EN before a soft-start can
be reinitiated. Note that the energy delivered during trip-retry
cycling is much less than during full-load operation, so there
is no thermal hazard.
OUTPUT CURRENT, 50A/DIV
0A
OUTPUT VOLTAGE,
500mV/DIV
During VID-on-the-fly transitions the OCL trip level is
boosted to prevent false overcurrent limiting events that can
occur. Starting from the beginning of a dynamic VID
transition, the overcurrent trip level is boosted to 196µA. The
OCL level will stay at this boosted level until 50µs after the
end of the dynamic VID transition, at which point it will return
to the typical 140µA trip level.
General Design Guide
This design guide is intended to provide a high-level
explanation of the steps necessary to create a multi-phase
power converter. It is assumed that the reader is familiar with
many of the basic skills and techniques referenced below. In
addition to this guide, Intersil provides complete reference
designs that include schematics, bills of materials, and example
board layouts for all common microprocessor applications.
Power Stages
The first step in designing a multi-phase converter is to
determine the number of phases. This determination
depends heavily on the cost analysis which in turn depends
on system constraints that differ from one design to the next.
Principally, the designer will be concerned with whether
components can be mounted on both sides of the circuit
board, whether through-hole components are permitted, the
total board space available for power-supply circuitry, and
the maximum amount of load current. Generally speaking,
the most economical solutions are those in which each
phase handles between 25A and 30A. All surface-mount
designs will tend toward the lower end of this current range.
If through-hole MOSFETs and inductors can be used, higher
per-phase currents are possible. In cases where board
space is the limiting constraint, current can be pushed as
high as 40A per phase, but these designs require heat sinks
and forced air to cool the MOSFETs, inductors and
heat-dissipating surfaces.
MOSFETS
0V
FIGURE 17. OVERCURRENT BEHAVIOR IN HICCUP MODE
Individual Channel Overcurrent Limiting
The ISL6313 has the ability to limit the current in each
individual channel without shutting down the entire regulator.
This is accomplished by continuously comparing the sensed
currents of each channel with a constant 140µA OCL
reference current as shown in Figure 17. If a channel’s
individual sensed current exceeds this OCL limit, the UGATE
signal of that channel is immediately forced low, and the
LGATE signal is forced high. This turns off the upper
MOSFET(s), turns on the lower MOSFET(s), and stops the
rise of current in that channel, forcing the current in the
channel to decrease. That channel’s UGATE signal will not
be able to return high until the sensed channel current falls
back below the 140µA reference.
24
The choice of MOSFETs depends on the current each
MOSFET will be required to conduct, the switching frequency,
the capability of the MOSFETs to dissipate heat, and the
availability and nature of heat sinking and air flow.
LOWER MOSFET POWER CALCULATION
The calculation for power loss in the lower MOSFET is
simple, since virtually all of the loss in the lower MOSFET is
due to current conducted through the channel resistance
(rDS(ON)). In Equation 25, IM is the maximum continuous
output current, IP-P is the peak-to-peak inductor current (see
Equation 1), and d is the duty cycle (VOUT/VIN).
I L ( P-P )2⋅ ( 1 – d )
⎛ I M⎞ 2
P LOW ( 1 ) = r DS ( ON ) ⋅ ⎜ -----⎟ ⋅ ( 1 – d ) + ---------------------------------------12
⎝ N⎠
(EQ. 25)
An additional term can be added to the lower-MOSFET loss
equation to account for additional loss accrued during the dead
time when inductor current is flowing through the
FN6448.2
September 2, 2008
ISL6313
lower-MOSFET body diode. This term is dependent on the
diode forward voltage at IM, VD(ON), the switching frequency,
fS, and the length of dead times, td1 and td2, at the beginning
and the end of the lower-MOSFET conduction interval
respectively.
P
LOW ( 2 )
= V
D ( ON )
⋅f
S
⎛I
⎞
⎛I
⎞
⎟
M I P-P⎟ ⋅ t
M I----------⋅ ⎜ -----+ ⎜ -----+ ----------– P-P⎟ ⋅ t d2
d1
⎜
2 ⎠
⎝N
2 ⎠
⎝N
(EQ. 26)
The total maximum power dissipated in each lower MOSFET
is approximated by the summation of PLOW(1) and PLOW(2).
UPPER MOSFET POWER CALCULATION
In addition to rDS(ON) losses, a large portion of the
upper-MOSFET losses are due to currents conducted across
the input voltage (VIN) during switching. Since a substantially
higher portion of the upper-MOSFET losses are dependent on
switching frequency, the power calculation is more complex.
Upper MOSFET losses can be divided into separate
components involving the upper-MOSFET switching times,
the lower-MOSFET body-diode reverse-recovery charge, Qrr,
and the upper MOSFET rDS(ON) conduction loss.
When the upper MOSFET turns off, the lower MOSFET does
not conduct any portion of the inductor current until the
voltage at the phase node falls below ground. Once the
lower MOSFET begins conducting, the current in the upper
MOSFET falls to zero as the current in the lower MOSFET
ramps up to assume the full inductor current. In Equation 27,
the required time for this commutation is t1 and the
approximated associated power loss is PUP(1)..
I M I P-P⎞ ⎛ t 1 ⎞
P UP ( 1 ) ≈ V IN ⋅ ⎛ ----- ⋅ ⎜ ---- ⎟ ⋅ f
⎝ N- + --------2 ⎠ ⎝ 2⎠ S
(EQ. 27)
At turn on, the upper MOSFET begins to conduct and this
transition occurs over a time t2. In Equation 28, the
approximate power loss is PUP(2)..
⎛ I M I P-P⎞ ⎛ t 2 ⎞
P UP ( 2 ) ≈ V IN ⋅ ⎜ ----- – ----------⎟ ⋅ ⎜ ---- ⎟ ⋅ f S
2 ⎠ ⎝ 2⎠
⎝N
(EQ. 29)
Finally, the resistive part of the upper MOSFET is given in
Equation 30 as PUP(4)..
2
I P-P2
⎛ I M⎞
P UP ( 4 ) ≈ r DS ( ON ) ⋅ d ⋅ ⎜ -----⎟ + ---------12
⎝ N⎠
25
Package Power Dissipation
When choosing MOSFETs it is important to consider the
amount of power being dissipated in the integrated drivers
located in the controller. Since there are a total of three
drivers in the controller package, the total power dissipated
by all three drivers must be less than the maximum
allowable power dissipation for the QFN package.
Calculating the power dissipation in the drivers for a desired
application is critical to ensure safe operation. Exceeding the
maximum allowable power dissipation level will push the IC
beyond the maximum recommended operating junction
temperature of +125°C. The maximum allowable IC power
dissipation for the 6x6 QFN package is approximately 3.5W
at room temperature. See “Layout Considerations” on
page 31 for thermal transfer improvement suggestions.
When designing the ISL6313 into an application, it is
recommended that the following calculation is used to ensure
safe operation at the desired frequency for the selected
MOSFETs. The total gate drive power losses, PQg_TOT, due to
the gate charge of MOSFETs and the integrated driver’s
internal circuitry and their corresponding average driver current
can be estimated with Equations 31 and 32, respectively.
P Qg_TOT = P Qg_Q1 + P Qg_Q2 + I Q ⋅ VCC
(EQ. 31)
3
P Qg_Q1 = --- ⋅ Q G1 ⋅ PVCC ⋅ F SW ⋅ N Q1 ⋅ N PHASE
2
P Qg_Q2 = Q G2 ⋅ PVCC ⋅ F SW ⋅ N Q2 ⋅ N PHASE
(EQ. 28)
A third component involves the lower MOSFET
reverse-recovery charge, Qrr. Since the inductor current has
fully commutated to the upper MOSFET before the
lower-MOSFET body diode can recover all of Qrr, it is
conducted through the upper MOSFET across VIN. The
power dissipated as a result is PUP(3).
P UP ( 3 ) = V IN ⋅ Q rr ⋅ f S
The total power dissipated by the upper MOSFET at full load
can now be approximated as the summation of the results
from Equations 27, 28, 29 and 30. Since the power
equations depend on MOSFET parameters, choosing the
correct MOSFETs can be an iterative process involving
repetitive solutions to the loss equations for different
MOSFETs and different switching frequencies.
(EQ. 32)
3
I DR = ⎛ --- ⋅ Q G1 ⋅ N
+ Q G2 ⋅ N Q2⎞ ⋅ N PHASE ⋅ F SW + I Q
⎝2
⎠
Q1
In Equations 31 and 32, PQg_Q1 is the total upper gate drive
power loss and PQg_Q2 is the total lower gate drive power
loss; the gate charge (QG1 and QG2) is defined at the
particular gate to source drive voltage PVCC in the
corresponding MOSFET data sheet; IQ is the driver total
quiescent current with no load at both drive outputs; NQ1 and
NQ2 are the number of upper and lower MOSFETs per phase,
respectively; NPHASE is the number of active phases. The
IQ*VCC product is the quiescent power of the controller
without capacitive load and is typically 75mW at 300kHz.
(EQ. 30)
FN6448.2
September 2, 2008
ISL6313
Inductor DCR Current Sensing Component
Selection
BOOT
D
CGD
G
RG1
CDS
RGI1
CGS
Q1
S
PHASE
FIGURE 18. TYPICAL UPPER-GATE DRIVE TURN-ON PATH
Follow the steps below to choose the component values for
this RC network.
1. Choose an arbitrary value for C1. The recommended
value is 0.1µF.
PVCC
D
CGD
G
LGATE
RG2
RGI2
CGS
Q2
S
FIGURE 19. TYPICAL LOWER-GATE DRIVE TURN-ON PATH
The total gate drive power losses are dissipated among the
resistive components along the transition path and in the
bootstrap diode. The portion of the total power dissipated in
the controller itself is the power dissipated in the upper drive
path resistance, PDR_UP, the lower drive path resistance,
PDR_UP, and in the boot strap diode, PBOOT. The rest of the
power will be dissipated by the external gate resistors (RG1
and RG2) and the internal gate resistors (RGI1 and RGI2) of
the MOSFETs. Figures 18 and 19 show the typical upper and
lower gate drives turn-on transition path. The total power
dissipation in the controller itself, PDR, can be roughly
estimated as Equation 33:
P DR = P DR_UP + P DR_LOW + P BOOT + ( I Q ⋅ VCC )
P Qg_Q1
P BOOT = --------------------3
(EQ. 33)
2. Plug the inductor L and DCR component values, and the
value for C1 chosen in step 1 into Equation 34, to
calculate the value for R1.
L
R 1 = ------------------------DCR ⋅ C 1
(EQ. 34)
Once the R-C network components have been chosen, the
effective internal RISEN resistance must then be set. The
RISEN resistance sets the gain of the load line regulation
loop as well as the gain of the channel-current balance loop
and the overcurrent trip level. The effective internal RISEN
resistance is set through a single resistor on the RSET pin,
RSET.
VIN
UGATE
DRIVER
DCR
INDUCTOR
VL(s)
R1
In
VOUT
COUT
VC(s)
C1
ISL6313 INTERNAL
CIRCUIT
SENSE
+
R HI2
R LO2
⎛
⎞ P Qg_Q2
P DR_LOW = ⎜ -------------------------------------- + ----------------------------------------⎟ ⋅ --------------------R
+
R
R
+
R
2
⎝ HI2
EXT2
LO2
EXT2⎠
Q1
L
LGATE
R LO1
R HI1
⎛
⎞ P Qg_Q1
P DR_UP = ⎜ -------------------------------------- + ----------------------------------------⎟ ⋅ --------------------R
+
R
R
+
R
3
⎝ HI1
EXT1
LO1
EXT1⎠
R GI1
R EXT1 = R G1 + ------------N
IL
MOSFET
+
RLO2
CDS
R GI2
R EXT2 = R G2 + ------------N
-
VC(s)
RISEN
ISEN
-
RHI2
+
RLO1
+
UGATE
-
RHI1
The ISL6313 senses each individual channel’s inductor
current by detecting the voltage across the output inductor
DCR of that channel (As described in the “Continuous Current
Sensing” on page 12). As Figure 20 illustrates, an R-C
network is required to accurately sense the inductor DCR
voltage and convert this information into a current, which is
proportional to the total output current. The time constant of
this R-C network must match the time constant of the inductor
L/DCR.
-
PVCC
ISENISEN+
RSET
VCC
RSET
FIGURE 20. DCR SENSING CONFIGURATION
Q2
Use Equation 35 to calculate the value of RSET. In
this equation, DCR is the DCR of the output inductor at room
temperature, IOCP is the desired overcurrent trip level, and
N is the number of phases. It is recommended that the
26
FN6448.2
September 2, 2008
ISL6313
desired overcurrent trip level, IOCP, be chosen so that it’s
30% larger then the maximum load current expected.
I OCP 400
DCR
R SET = ---------------------------- ⋅ -------------- ⋅ ---------–6
N
3
100 × 10
(EQ. 35)
*Note: RSET must be between 20kΩ and 80kΩ
Due to errors in the inductance or DCR it may be necessary
to adjust the value of R1 to match the time constants
correctly. The effects of time constant mismatch can be seen
in the form of droop overshoot or undershoot during the
initial load transient spike, as shown in Figure 21. Follow the
steps below to ensure the R-C and inductor L/DCR time
constants are matched accurately.
1. Capture a transient event with the oscilloscope set to
about L/DCR/2 (sec/div). For example, with L = 1µH and
DCR = 1mΩ, set the oscilloscope to 500µs/div.
2. Record ΔV1 and ΔV2 as shown in Figure 21.
3. Select new values, R1(NEW), for the time constant
resistor based on the original value, R1(OLD), using
Equation 36.
ΔV 1
R 1 ( NEW ) = R 1 ( OLD ) ⋅ ---------ΔV
(EQ. 36)
2
4. Replace R1 with the new value and check to see that the
error is corrected. Repeat the procedure if necessary.
.
V DROOP
R LL = -----------------------I FL
(EQ. 37)
Based on the desired loadline, the loadline regulation
resistor, RFB, can be calculated from Equation 38.
R LL ⋅ N ⋅ R SET 3
R FB = --------------------------------------- ⋅ ---------DCR
400
(EQ. 38)
*Note: RFB must be not exceed 4.0kΩ
In Equation 38, RLL is the loadline resistance; N is the
number of active channels; DCR is the DCR of the individual
output inductors; and RSET is the RSET pin resistor.
If no loadline regulation is required, the resistor on the FS
pin, RT, should be connected to the VCC pin. To choose the
value for RFB in this situation, please refer to “Compensation
without load-line regulation” on page 28.
IOUT Pin Resistor
A copy of the average sense current flows out of the IOUT
pin, and a resistor, RIOUT, placed from this pin to ground can
be used to set the overcurrent protection trip level. Based on
the desired overcurrent trip threshold, IOCP, the IOUT pin
resistor, RIOUT, can be calculated from Equation 39.
R SET ⋅ N
6
R IOUT = -------------------------------- ⋅ ---------DCR ⋅ I OCP 400
(EQ. 39)
APA Pin Component Selection
ΔV2
ΔV1
VOUT
ITRAN
ΔI
A 100µA current flows into the APA pin and across RAPA to
set the APA trip level. A 1000pF capacitor, CAPA, should
also be placed across the RAPA resistor to help with noise
immunity. Use Equation 40 to set RAPA to get the desired
APA trip level. An APA trip level of 500mV is recommended
for most applications.
V APA ( TRIP )
500mV
R APA = --------------------------------- = ----------------------------- = 5kΩ
100 × 10 – 6
100 × 10 – 6
(EQ. 40)
Compensation
FIGURE 21. TIME CONSTANT MISMATCH BEHAVIOR
Loadline Regulation Resistor
If loadline regulation is desired, the resistor on the FS pin,
RT, should be connected to Ground in order for the internal
average sense current to flow out across the loadline
regulation resistor, labeled RFB in Figure 7. This resistor’s
value sets the desired loadline required for the application.
The desired loadline, RLL, can be calculated by Equation 37
where VDROOP is the desired droop voltage at the full load
current IFL.
27
The two opposing goals of compensating the voltage
regulator are stability and speed. Depending on whether the
regulator employs the optional load-line regulation as
described in Load-Line Regulation, there are two distinct
methods for achieving these goals.
COMPENSATION WITH LOAD-LINE REGULATION
The load-line regulated converter behaves in a similar
manner to a peak current mode controller because the two
poles at the output filter L-C resonant frequency split with the
introduction of current information into the control loop. The
final location of these poles is determined by the system
function, the gain of the current signal, and the value of the
compensation components, RC and CC.
FN6448.2
September 2, 2008
ISL6313
C2 (OPTIONAL)
Case 1:
RC
CC
1
-------------------------------- > f 0
2⋅π⋅ L⋅C
2 ⋅ π ⋅ f 0 ⋅ V P-P ⋅ L ⋅ C
R C = R FB ⋅ ---------------------------------------------------------V
COMP
IN
V IN
C C = ----------------------------------------------------2 ⋅ π ⋅ V P-P ⋅ R FB ⋅ f 0
FB
ISL6313
RFB
VSEN
Case 2:
V P-P ⋅ ( 2 ⋅ π ) 2 ⋅ f 02 ⋅ L ⋅ C
R C = R FB ⋅ -----------------------------------------------------------------V
FIGURE 22. COMPENSATION CONFIGURATION FOR
LOAD-LINE REGULATED ISL6313 CIRCUIT
Since the system poles and zero are affected by the values
of the components that are meant to compensate them, the
solution to the system equation becomes fairly complicated.
Fortunately, there is a simple approximation that comes very
close to an optimal solution. Treating the system as though it
were a voltage-mode regulator, by compensating the L-C
poles and the ESR zero of the voltage mode approximation,
yields a solution that is always stable with very close to ideal
transient performance.
Select a target bandwidth for the compensated system, f0.
The target bandwidth must be large enough to assure
adequate transient performance, but smaller than 1/3 of the
per-channel switching frequency. The values of the
compensation components depend on the relationships of f0
to the L-C pole frequency and the ESR zero frequency. For
each of the following three, there is a separate set of
equations for the compensation components.
The optional capacitor C2, is sometimes needed to bypass
noise away from the PWM comparator (see Figure 22). Keep
a position available for C2, and be prepared to install a
high-frequency capacitor of between 22pF and 150pF in
case any leading edge jitter problem is noted.
28
(EQ. 41)
IN
V IN
C C = -------------------------------------------------------------------------------------2
2
( 2 ⋅ π ) ⋅ f 0 ⋅ V P-P ⋅ R FB ⋅ L ⋅ C
Case 3:
1
f 0 > ------------------------------------2 ⋅ π ⋅ C ⋅ ESR
2 ⋅ π ⋅ f 0 ⋅ V P-P ⋅ L
R C = R FB ⋅ ---------------------------------------------V ⋅ ESR
IN
V IN ⋅ ESR ⋅ C
C C = ----------------------------------------------------------------2 ⋅ π ⋅ V P-P ⋅ R FB ⋅ f 0 ⋅ L
COMPENSATION WITHOUT LOAD-LINE REGULATION
The non load-line regulated converter is accurately modeled
as a voltage-mode regulator with two poles at the L-C
resonant frequency and a zero at the ESR frequency. A
type III controller, as shown in Figure 23, provides the
necessary compensation.
In Equation 41, L is the per-channel filter inductance divided
by the number of active channels; C is the sum total of all
output capacitors; ESR is the equivalent series resistance of
the bulk output filter capacitance; and VP-P is the
peak-to-peak sawtooth signal amplitude as described in the
“Electrical Specifications” on page 6.
Once selected, the compensation values in Equation 41
assure a stable converter with reasonable transient
performance. In most cases, transient performance can be
improved by making adjustments to RC. Slowly increase the
value of RC while observing the transient performance on an
oscilloscope until no further improvement is noted. Normally,
CC will not need adjustment. Keep the value of CC from
Equation 41 unless some performance issue is noted.
1
1
-------------------------------- ≤ f 0 < -----------------------------------2 ⋅ π ⋅ C ⋅ ESR
2⋅π⋅ L⋅C
C2
RC
CC
COMP
FB
C1
R1
ISL6313
RFB
VSEN
FIGURE 23. COMPENSATION CIRCUIT WITHOUT LOAD-LINE
REGULATION
The first step, is to choose the desired bandwidth, f0, of the
compensated system. Choose a frequency high enough to
assure adequate transient performance but not higher than
1/3 of the switching frequency. The type-III compensator has
an extra high-frequency pole, fHF. This pole can be used for
added noise rejection or to assure adequate attenuation at
the error-amplifier high-order pole and zero frequencies. A
FN6448.2
September 2, 2008
ISL6313
good general rule is to choose fHF = 10f0, but it can be
higher if desired. Choosing fHF to be lower than 10f0 can
cause problems with too much phase shift below the system
bandwidth.
C ⋅ ESR
R 1 = R FB ⋅ -------------------------------------------L ⋅ C – C ⋅ ESR
L ⋅ C – C ⋅ ESR
C 1 = -------------------------------------------R FB
di
ΔV ≈ ESL ⋅ ----- + ESR ⋅ ΔI
dt
V IN
C 2 = ---------------------------------------------------------------------------------------------------( 2 ⋅ π ) 2 ⋅ f 0 ⋅ f HF ⋅ ( L ⋅ C ) ⋅ R FB ⋅ V P-P
(EQ. 42)
2
V P-P ⋅ ⎛ 2π⎞ ⋅ f 0 ⋅ f HF ⋅ L ⋅ C ⋅ R FB
⎝ ⎠
R C = ----------------------------------------------------------------------------------------V ⋅ ( 2 ⋅ π ⋅ f HF ⋅ L ⋅ C – 1 )
IN
V IN ⋅ ( 2 ⋅ π ⋅ f HF ⋅ L ⋅ C – 1 )
C C = ---------------------------------------------------------------------------------------------------( 2 ⋅ π ) 2 ⋅ f 0 ⋅ f HF ⋅ ( L ⋅ C ) ⋅ R FB ⋅ V P-P
In the solutions to the compensation equations, there is a
single degree of freedom. For the solutions presented in
Equation 41, RFB is selected arbitrarily. The remaining
compensation components are then selected in Equation 42.
In Equation 42, L is the per-channel filter inductance divided
by the number of active channels; C is the sum total of all
output capacitors; ESR is the equivalent-series resistance of
the bulk output-filter capacitance; and VP-P is the
peak-to-peak sawtooth signal amplitude as described in the
Electrical Specifications on page 6.
Output Filter Design
The output inductors and the output capacitor bank together
to form a low-pass filter responsible for smoothing the
pulsating voltage at the phase nodes. The output filter also
must provide the transient energy until the regulator can
respond. Because it has a low bandwidth compared to the
switching frequency, the output filter limits the system
transient response. The output capacitors must supply or
sink load current while the current in the output inductors
increases or decreases to meet the demand.
In high-speed converters, the output capacitor bank is usually
the most costly (and often the largest) part of the circuit.
Output filter design begins with minimizing the cost of this part
of the circuit. The critical load parameters in choosing the
output capacitors are the maximum size of the load step, ΔI,
the load-current slew rate, di/dt, and the maximum allowable
output-voltage deviation under transient loading, ΔVMAX.
Capacitors are characterized according to their capacitance,
ESR, and ESL (equivalent series inductance).
At the beginning of the load transient, the output capacitors
supply all of the transient current. The output voltage will
29
initially deviate by an amount approximated by the voltage
drop across the ESL. As the load current increases, the
voltage drop across the ESR increases linearly until the load
current reaches its final value. The capacitors selected must
have sufficiently low ESL and ESR so that the total
output-voltage deviation is less than the allowable maximum.
Neglecting the contribution of inductor current and regulator
response, the output voltage initially deviates by an amount
as shown in Equation 43.
(EQ. 43)
The filter capacitor must have sufficiently low ESL and ESR
so that ΔV < ΔVMAX.
Most capacitor solutions rely on a mixture of high frequency
capacitors with relatively low capacitance in combination
with bulk capacitors having high capacitance but limited
high-frequency performance. Minimizing the ESL of the
high-frequency capacitors allows them to support the output
voltage as the current increases. Minimizing the ESR of the
bulk capacitors allows them to supply the increased current
with less output voltage deviation.
The ESR of the bulk capacitors also creates the majority of
the output-voltage ripple. As the bulk capacitors sink and
source the inductor AC ripple current (see “Interleaving” on
page 10 and Equation 2), a voltage develops across the bulk
capacitor ESR equal to IC(P-P)(ESR). Thus, once the output
capacitors are selected, the maximum allowable ripple
voltage, VP-P(MAX), determines the lower limit on the
inductance as shown in Equation 44.
⎛V – N ⋅ V
⎞
OUT⎠ ⋅ V OUT
⎝ IN
L ≥ ESR ⋅ -------------------------------------------------------------------f S ⋅ V IN ⋅ V P-P( MAX )
(EQ. 44)
Since the capacitors are supplying a decreasing portion of
the load current while the regulator recovers from the
transient, the capacitor voltage becomes slightly depleted.
The output inductors must be capable of assuming the entire
load current before the output voltage decreases more than
ΔVMAX. This places an upper limit on inductance.
Equation 45 gives the upper limit on L for the cases when
the trailing edge of the current transient causes a greater
output-voltage deviation than the leading edge. Equation 46
addresses the leading edge. Normally, the trailing edge
dictates the selection of L because duty cycles are usually
less than 50%. Nevertheless, both inequalities should be
evaluated, and L should be selected based on the lower of
the two results. In each equation, L is the per-channel
inductance, C is the total output capacitance, and N is the
number of active channels.
2 ⋅ N ⋅ C ⋅ VO
L ≤ --------------------------------- ⋅ ΔV MAX – ( ΔI ⋅ ESR )
( ΔI ) 2
(EQ. 45)
FN6448.2
September 2, 2008
ISL6313
0.3
(EQ. 46)
Switching Frequency
There are a number of variables to consider when choosing
the switching frequency, as there are considerable effects on
the upper MOSFET loss calculation. These effects are
outlined in “MOSFETs” on page 24, and they establish the
upper limit for the switching frequency. The lower limit is
established by the requirement for fast transient response
and small output-voltage ripple. Choose the lowest switching
frequency that allows the regulator to meet the
transient-response requirements.
Switching frequency is determined by the selection of the
frequency-setting resistor, RS. Figure 24 and Equation 47
are provided to assist in selecting the correct value for RS.
[10.61 – ( 1.035 ⋅ log ( f S ) ) ]
(EQ. 47)
RS (kΩ)
500
100
0.2
0.1
IL(P-P) = 0
IL(P-P) = 0.5 IO
IL(P-P) = 0.75 IO
0
0
0.2
0.4
0.6
0.8
1.0
DUTY CYCLE (VIN/VO)
FIGURE 25. NORMALIZED INPUT-CAPACITOR RMS
CURRENT FOR 2-PHASE CONVERTER
Select a bulk capacitor with a ripple current rating which will
minimize the total number of input capacitors required to
support the RMS current calculated. The voltage rating of
the capacitors should also be at least 1.25x greater than the
maximum input voltage. Figure 26 provides the same input
RMS current information for single-phase designs. Use the
same approach for selecting the bulk capacitor type and
number.
0.6
10
50
100
1k
2k
SWITCHING FREQUENCY (kHz)
FIGURE 24. RS vs SWITCHING FREQUENCY
Input Capacitor Selection
The input capacitors are responsible for sourcing the AC
component of the input current flowing into the upper
MOSFETs. Their RMS current capacity must be sufficient to
handle the AC component of the current drawn by the upper
MOSFETs which is related to duty cycle and the number of
active phases.
For a two-phase design, use Figure 25 to determine the
input-capacitor RMS current requirement set by the duty
cycle, maximum sustained output current (IO), and the ratio
of the peak-to-peak inductor current (IL(P-P)) to IO.
30
INPUT-CAPACITOR CURRENT (IRMS/IO)
R S = 10
INPUT-CAPACITOR CURRENT (IRMS/IO)
1.25 ⋅ N ⋅ C- ⋅ ΔV
⎛
⎞
L ≤ ---------------------------MAX – ( ΔI ⋅ ESR ) ⋅ ⎝ V IN – V O⎠
( ΔI ) 2
0.4
0.2
IL(P-P) = 0
IL(P-P) = 0.5 IO
IL(P-P) = 0.75 IO
0
0
0.2
0.4
0.6
0.8
1.0
DUTY CYCLE (VIN/VO)
FIGURE 26. NORMALIZED INPUT-CAPACITOR RMS
CURRENT FOR SINGLE-PHASE CONVERTER
Low capacitance, high-frequency ceramic capacitors are
needed in addition to the input bulk capacitors to suppress
leading and falling edge voltage spikes. The spikes result from
the high current slew rate produced by the upper MOSFET
turn on and off. Select low ESL ceramic capacitors and place
one as close as possible to each upper MOSFET drain to
minimize board parasitics and maximize suppression.
FN6448.2
September 2, 2008
ISL6313
Layout Considerations
MOSFETs switch very fast and efficiently. The speed with
which the current transitions from one device to another
causes voltage spikes across the interconnecting
impedances and parasitic circuit elements. These voltage
spikes can degrade efficiency, radiate noise into the circuit
and lead to device overvoltage stress. Careful component
selection, layout, and placement minimizes these voltage
spikes. Consider, as an example, the turnoff transition of the
upper PWM MOSFET. Prior to turnoff, the upper MOSFET
was carrying channel current. During the turnoff, current
stops flowing in the upper MOSFET and is picked up by the
lower MOSFET. Any inductance in the switched current path
generates a large voltage spike during the switching interval.
Careful component selection, tight layout of the critical
components, and short, wide circuit traces minimize the
magnitude of voltage spikes.
There are two sets of critical components in a DC/DC
converter using a ISL6313 controller. The power
components are the most critical because they switch large
amounts of energy. Next are small signal components that
connect to sensitive nodes or supply critical bypassing
current and signal coupling.
The power components should be placed first, which include
the MOSFETs, input and output capacitors, and the inductors. It
is important to have a symmetrical layout for each power train,
preferably with the controller located equidistant from each.
Symmetrical layout allows heat to be dissipated equally
across all power trains. Equidistant placement of the controller
to the power trains it controls through the integrated drivers
helps keep the gate drive traces equally short, resulting in
equal trace impedances and similar drive capability of all sets
of MOSFETs.
When placing the MOSFETs try to keep the source of the
upper FETs and the drain of the lower FETs as close as
thermally possible. Input Bulk capacitors should be placed
close to the drain of the upper FETs and the source of the lower
FETs. Locate the output inductors and output capacitors
between the MOSFETs and the load. The high-frequency input
and output decoupling capacitors (ceramic) should be placed
as close as practicable to the decoupling target, making use of
the shortest connection paths to any internal planes, such as
vias to GND next or on the capacitor solder pad.
The critical small components include the bypass capacitors
for VCC and PVCC, and many of the components
surrounding the controller including the feedback network
and current sense components. Locate the VCC/PVCC
bypass capacitors as close to the ISL6313 as possible. It is
especially important to locate the components associated
with the feedback circuit close to their respective controller
pins, since they belong to a high-impedance circuit loop,
sensitive to EMI pick-up.
31
A multi-layer printed circuit board is recommended. Figure 27
shows the connections of the critical components for the
converter. Note that capacitors Cxx(IN) and Cxx(OUT) could
each represent numerous physical capacitors. Dedicate one
solid layer, usually the one underneath the component side of
the board, for a ground plane and make all critical component
ground connections with vias to this layer.
Dedicate another solid layer as a power plane and break this
plane into smaller islands of common voltage levels. Keep the
metal runs from the PHASE terminal to output inductors short.
The power plane should support the input power and output
power nodes. Use copper filled polygons on the top and bottom
circuit layers for the phase nodes. Use the remaining printed
circuit layers for small signal wiring.
Routing UGATE, LGATE, and PHASE Traces
Great attention should be paid to routing the UGATE, LGATE,
and PHASE traces since they drive the power train MOSFETs
using short, high current pulses. It is important to size them as
large and as short as possible to reduce their overall
impedance and inductance. They should be sized to carry at
least one ampere of current (0.02” to 0.05”). Going between
layers with vias should also be avoided, but if so, use two vias
for interconnection when possible.
Extra care should be given to the LGATE traces in particular
since keeping their impedance and inductance low helps to
significantly reduce the possibility of shoot-through. It is also
important to route each channels UGATE and PHASE traces
in as close proximity as possible to reduce their inductances.
Current Sense Component Placement and Trace
Routing
One of the most critical aspects of the ISL6313 regulator
layout is the placement of the inductor DCR current sense
components and traces. The R-C current sense components
must be placed as close to their respective ISEN+ and
ISEN- pins on the ISL6313 as possible.
The sense traces that connect the R-C sense components to
each side of the output inductors should be routed on the
bottom of the board, away from the noisy switching
components located on the top of the board. These traces
should be routed side by side, and they should be very thin
traces. It’s important to route these traces as far away from
any other noisy traces or planes as possible. These traces
should pick up as little noise as possible.
Thermal Management
For maximum thermal performance in high current, high
switching frequency applications, connecting the thermal
GND pad of the ISL6313 to the ground plane with multiple
vias is recommended. This heat spreading allows the part to
achieve its full thermal potential. It is also recommended
that the controller be placed in a direct path of airflow if
possible to help thermally manage the part.
FN6448.2
September 2, 2008
ISL6313
RFB
R2
C2
CDVC
RDVC
KEY
HEAVY TRACE ON CIRCUIT PLANE LAYER
DVC
FB
VSEN
RGND
C1
C3
ISLAND ON POWER PLANE LAYER
ISLAND ON CIRCUIT PLANE LAYER
+5V VCC
VIA CONNECTION TO GROUND PLANE
R1
RSET
COMP
LOCATE CLOSE TO IC
(MINIMIZE CONNECTION PATH)
RSET
RAPA
+12V
APA
CBIN1
+5V
CBOOT1
LOCATE NEAR SWITCHING TRANSISTORS;
(MINIMIZE CONNECTION PATH)
BOOT1
VCC
(CF1)
UGATE1
ROFS
PHASE1
OFS
R1
FS
RT
C1
LGATE1
REF
ISEN1ISEN1+
CREF
SS
RSS
(CHFOUT)
VID7
VID6
VID5
CBOUT
+12V
PVCC
VID4
VID3
(CF2)
ISL6313
VID2
CBIN2
LOAD
CBOOT2
BOOT2
VID1
UGATE2
VID0
VRSEL
PHASE2
PGOOD
R1
C1
LOCATE NEAR LOAD;
(MINIMIZE CONNECTION
PATH)
LGATE2
EN
ISEN2ISEN2+
IOUT
RIOUT
GND
FIGURE 27. PRINTED CIRCUIT BOARD POWER PLANES AND ISLANDS
All Intersil U.S. products are manufactured, assembled and tested utilizing ISO9000 quality systems.
Intersil Corporation’s quality certifications can be viewed at www.intersil.com/design/quality
Intersil products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design, software and/or specifications at any time without
notice. Accordingly, the reader is cautioned to verify that data sheets are current before placing orders. Information furnished by Intersil is believed to be accurate and
reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result
from its use. No license is granted by implication or otherwise under any patent or patent rights of Intersil or its subsidiaries.
For information regarding Intersil Corporation and its products, see www.intersil.com
32
FN6448.2
September 2, 2008
ISL6313
Package Outline Drawing
L36.6x6
36 LEAD THIN QUAD FLAT NO-LEAD PLASTIC PACKAGE
Rev 5, 08/08
6.00
6
PIN #1 INDEX AREA
32x 0.50
A
B
36
28
6
27
6.00
PIN 1
INDEX AREA
1
4.15 +0.10/-0.15
4X 4.00
9
19
(4X)
0.15
10
18
36X 0.55 ± 0.10
36X 0.25 +0.05/-.07 4
0.10 M C A B
BOTTOM VIEW
TOP VIEW
( 5.65 )
( 4.15)
Exp. Dap.
SEE DETAIL "X"
( 5.65 )
0.10 C
Max 0.80
C
( 32x 0.50)
0.08 C
( 4.15)
Exp. Dap.
SIDE VIEW
(36X .25)
C
0 . 2 REF
5
0 . 00 MIN.
0 . 05 MAX.
( 4X 4.00)
(36X 0.75)
TYPICAL RECOMMENDED LAND PATTERN
DETAIL "X"
1. Dimensions are in millimeters.
2. Dimensioning and tolerancing conform to AMSEY14.5m-1994.
3. Unless otherwise specified, tolerance : Decimal ± 0.05
4. Dimension applies to the metallized terminal and is measured
between 0.15mm and 0.30mm from the terminal tip.
5. Tiebar shown (if present) is a non-functional feature.
6. The configuration of the pin #1 identifier is optional, but must be
located within the zone indicated. The pin #1 indentifier may be
either a mold or mark feature.
33
FN6448.2
September 2, 2008