[ /Title (HIP60 04) /Subject (Buck and SynchronousRectifier (PWM )Controller and Output Voltage Monitor) /Autho r () /Keywords (Intersil Corporation, Intelligent Power, Computer Power, Pentium Power, HIP6004 Data Sheet March 2000 Buck and Synchronous-Rectifier (PWM) Controller and Output Voltage Monitor The HIP6004 provides complete control and protection for a DC-DC converter optimized for high-performance microprocessor applications. It is designed to drive two N-Channel MOSFETs in a synchronous-rectified buck topology. The HIP6004 integrates all of the control, output adjustment, monitoring and protection functions into a single package. The output voltage of the converter is easily adjusted and precisely regulated. The HIP6004 includes a 5-input digital-to-analog converter (DAC) that adjusts the output voltage from 2.1VDC to 3.5VDC in 0.1V increments and from 1.3VDC to 2.1VDC in 0.05V steps. The precision reference and voltage-mode regulator hold the selected output voltage to within ±1% over temperature and line voltage variations. The HIP6004 provides simple, single feedback loop, voltage-mode control with fast transient response. It includes a 200kHz free-running triangle-wave oscillator that is adjustable from below 50kHz to over 1MHz. The error amplifier features a 15MHz gain-bandwidth product and 6V/μs slew rate which enables high converter bandwidth for fast transient performance. The resulting PWM duty ratio ranges from 0% to 100%. The HIP6004 monitors the output voltage with a window comparator that tracks the DAC output and issues a Power Good signal when the output is within ±10%. The HIP6004 protects against over-current conditions by inhibiting PWM operation. Built-in over-voltage protection triggers an external SCR to crowbar the input supply. The HIP6004 monitors the current by using the rDS(ON) of the upper MOSFET which eliminates the need for a current sensing resistor. HIP6004CB TEMP. RANGE (oC) 0 to 70 PACKAGE 20 Ld SOIC • Drives Two N-Channel MOSFETs • Operates from +5V or +12V Input • Simple Single-Loop Control Design - Voltage-Mode PWM Control • Fast Transient Response - High-Bandwidth Error Amplifier - Full 0% to 100% Duty Ratio • Excellent Output Voltage Regulation - ±1% Over Line Voltage and Temperature • 5-Bit Digital-to-Analog Output Voltage Selection - Wide Range . . . . . . . . . . . . . . . . . . . 1.3VDC to 3.5VDC - 0.1V Binary Steps. . . . . . . . . . . . . . . 2.1VDC to 3.5VDC - 0.05V Binary Step. . . . . . . . . . . . . . . 1.3VDC to 2.1VDC • Power-Good Output Voltage Monitor • Over-Voltage and Over-Current Fault Monitors - Does Not Require Extra Current Sensing Element, Uses MOSFETs rDS(ON) • Small Converter Size - Constant Frequency Operation - 200kHz Free-Running Oscillator Programmable from 50kHz to over 1MHz Applications • Power Supply for Pentium®, Pentium Pro, PowerPC™ and Alpha™ Microprocessors • High-Power 5V to 3.xV DC-DC Regulators • Low-Voltage Distributed Power Supplies Pinout This data sheet describes a pre-released product. Alpha Micro™ is a trademark of Digital Computer Equipment Corporation. Pentium® is a registered trademark of Intel Corporation. PowerPC™ is a registered trademark of IBM. 1 HIP6004 (SOIC) TOP VIEW PKG. NO. M20.3 4275.2 Features Ordering Information PART NUMBER File Number VSEN 1 20 RT OCSET 2 19 OVP SS 3 18 VCC VID0 4 17 LGATE VID1 5 16 PGND VID2 6 15 BOOT VID3 7 14 UGATE VID4 8 13 PHASE COMP 9 12 PGOOD FB 10 11 GND CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures. 1-888-INTERSIL or 1-888-468-3774 | Copyright © Intersil Corporation 2000 HIP6004 Typical Application +12V VIN = +5V or +12V VCC PGOOD OCSET MONITOR AND PROTECTION SS EN BOOT OVP RT VID0 VID1 VID2 VID3 VID4 OSC UGATE PHASE HIP6004 +VOUT D/A FB - LGATE + + - COMP PGND GND VSEN Block Diagram VCC VSEN POWER-ON RESET (POR) 110% + - PGOOD 90% + - 115% + OVERVOLTAGE 10μA OVP - SOFTSTART + - OCSET REFERENCE 200μA OVERCURRENT SS BOOT UGATE 4V PHASE VID0 VID1 VID2 VID3 VID4 D/A CONVERTER (DAC) DACOUT PWM COMPARATOR + - + - ERROR AMP FB GATE INHIBIT CONTROL LOGIC PWM LGATE PGND COMP GND OSCILLATOR RT 2 HIP6004 Absolute Maximum Ratings Thermal Information Supply Voltage, VCC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .+15V Boot Voltage, VBOOT - VPHASE . . . . . . . . . . . . . . . . . . . . . . . .+15V Input, Output or I/O Voltage . . . . . . . . . . . GND -0.3V to VCC +0.3V ESD Classification . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Class 2 Thermal Resistance (Typical, Note 1) θJA (oC/W) SOIC Package . . . . . . . . . . . . . . . . . . . . . . . . . . . . 118 Maximum Junction Temperature (Plastic Package) . . . . . . . .150oC Maximum Storage Temperature Range . . . . . . . . . -65oC to 150oC Maximum Lead Temperature (Soldering 10s) . . . . . . . . . . . . .300oC (SOIC - Lead Tips Only) Operating Conditions Supply Voltage, VCC . . . . . . . . . . . . . . . . . . . . . . . . . . . +12V ±10% Ambient Temperature Range. . . . . . . . . . . . . . . . . . . . . 0oC to 70oC Junction Temperature Range . . . . . . . . . . . . . . . . . . . 0oC to 125oC CAUTION: Stresses above those listed in “Absolute Maximum Ratings” may cause permanent damage to the device. This is a stress only rating and operation of the device at these or any other conditions above those indicated in the operational sections of this specification is not implied. NOTE: 1. θJA is measured with the component mounted on a low effective thermal conductivity test board in free air. See Tech Brief 379 for details. Electrical Specifications Recommended Operating Conditions, Unless Otherwise Noted PARAMETER SYMBOL TEST CONDITIONS MIN TYP MAX UNITS UGATE and LGATE Open - 5 - mA Rising VCC Threshold VOCSET = 4.5V - - 10.4 V Falling VCC Threshold VOCSET = 4.5V 8.2 - - V - 1.26 - V VCC SUPPLY CURRENT Nominal Supply ICC POWER-ON RESET Rising VOCSET Threshold OSCILLATOR Free Running Frequency RT = OPEN 185 200 215 kHz Total Variation 6kΩ < RT to GND < 200kΩ -15 - +15 % - 1.9 - VP-P -1.0 - +1.0 % - 88 - dB - 15 - MHz - 6 - V/μs 350 500 - mA ΔVOSC Ramp Amplitude RT = Open REFERENCE AND DAC DACOUT Voltage Accuracy ERROR AMPLIFIER DC Gain Gain-Bandwidth Product GBW Slew Rate SR COMP = 10pF GATE DRIVERS Upper Gate Source IUGATE VBOOT - VPHASE = 12V, VUGATE = 6V Upper Gate Sink RUGATE ILGATE = 0.3A Lower Gate Source ILGATE VCC = 12V, VLGATE = 6V Lower Gate Sink RLGATE ILGATE = 0.3A - 115 120 % OCSET Current Source IOCSET VOCSET = 4.5VDC 170 200 230 μA OVP Sourcing Current IOVP VSEN = 5.5V, VOVP = 0V 60 - - mA - 10 - μA VSEN Rising 106 - 111 % Lower Threshold (VSEN / DACOUT) VSEN Falling 89 - 94 % Hysteresis (VSEN / DACOUT) Upper and Lower Threshold - 2 - % IPGOOD = -5mA - 0.5 - V - 5.5 10 Ω 300 450 - mA - 3.5 6.5 Ω PROTECTION Over-Voltage Trip (VSEN/DACOUT) Soft Start Current ISS POWER GOOD Upper Threshold (VSEN / DACOUT) PGOOD Voltage Low VPGOOD 3 HIP6004 Typical Performance Curves 80 CGATE = 3300pF 70 60 RT PULLUP TO +12V ICC (mA) RESISTANCE (kΩ) 1000 100 50 CUPPER = CLOWER = CGATE 40 CGATE = 1000pF 30 10 20 RT PULLDOWN TO VSS CGATE = 10pF 10 10 100 1000 0 100 200 SWITCHING FREQUENCY (kHz) FIGURE 1. RT RESISTANCE vs FREQUENCY Functional Pin Description VSEN 1 OCSET 2 19 OVP SS 3 18 VCC VID0 4 17 LGATE VID1 5 16 PGND VID2 6 15 BOOT VID3 7 14 UGATE VID4 8 13 PHASE COMP 9 12 PGOOD 20 RT FB 10 11 GND 300 400 500 600 700 800 SWITCHING FREQUENCY (kHz) 900 1000 FIGURE 2. BIAS SUPPLY CURRENT vs FREQUENCY VID0-4 (Pins 4-8) VID0-4 are the input pins to the 5-bit DAC. The states of these five pins program the internal voltage reference (DACOUT). The level of DACOUT sets the converter output voltage. It also sets the PGOOD and OVP thresholds. Table 1 specifies DACOUT for the 32 combinations of DAC inputs. COMP (Pin 9) and FB (Pin 10) COMP and FB are the available external pins of the error amplifier. The FB pin is the inverting input of the error amplifier and the COMP pin is the error amplifier output. These pins are used to compensate the voltage-control feedback loop of the converter. GND (Pin 11) Signal ground for the IC. All voltage levels are measured with respect to this pin. VSEN (Pin 1) This pin is connected to the converters output voltage. The PGOOD and OVP comparator circuits use this signal to report output voltage status and for overvoltage protection. OCSET (Pin 2) Connect a resistor (ROCSET) from this pin to the drain of the upper MOSFET. ROCSET, an internal 200μA current source (IOCS), and the upper MOSFET on-resistance (rDS(ON)) set the converter over-current (OC) trip point according to the following equation: I OCS • R OCSET I PEAK = -------------------------------------------r DS ( ON ) PGOOD (Pin 12) PGOOD is an open collector output used to indicate the status of the converter output voltage. This pin is pulled low when the converter output is not within ±10% of the DACOUT reference voltage. PHASE (Pin 13) Connect the PHASE pin to the upper MOSFET source. This pin is used to monitor the voltage drop across the MOSFET for over-current protection. This pin also provides the return path for the upper gate drive. UGATE (Pin 14) An over-current trip cycles the soft-start function. SS (Pin 3) Connect a capacitor from this pin to ground. This capacitor, along with an internal 10μA current source, sets the softstart interval of the converter. 4 Connect UGATE to the upper MOSFET gate. This pin provides the gate drive for the upper MOSFET. BOOT (Pin 15) This pin provides bias voltage to the upper MOSFET driver. A bootstrap circuit may be used to create a BOOT voltage suitable to drive a standard N-Channel MOSFET. HIP6004 PGND (Pin 16) This is the power ground connection. Tie the lower MOSFET source to this pin. LGATE (Pin 17) Connect LGATE to the lower MOSFET gate. This pin provides the gate drive for the lower MOSFET. VCC (Pin 18) Provide a 12V bias supply for the chip to this pin. OVP (Pin 19) of increasing width that charge the output capacitor(s). This interval of increasing pulse width continues to t2 . With sufficient output voltage, the clamp on the reference input controls the output voltage. This is the interval between t2 and t3 in Figure 3. At t3 the SS voltage exceeds the DACOUT voltage and the output voltage is in regulation. This method provides a rapid and controlled output voltage rise. The PGOOD signal toggles ‘high’ when the output voltage (VSEN pin) is within ±5% of DACOUT. The 2% hysteresis built into the power good comparators prevents PGOOD oscillation due to nominal output voltage ripple. The OVP pin can be used to drive an external SCR in the event of an overvoltage condition. RT (Pin 20) This pin provides oscillator switching frequency adjustment. By placing a resistor (RT) from this pin to GND, the nominal 200kHz switching frequency is increased according to the following equation: PGOOD (2V/DIV.) 0V SOFT-START (1V/DIV.) 6 5 • 10 Fs ≈ 200kHz + --------------------R T ( kΩ ) OUTPUT VOLTAGE (1V/DIV.) (RT to GND) Conversely, connecting a pull-up resistor (RT) from this pin to VCC reduces the switching frequency according to the following equation: 0V 0V t1 (RT to 12V) t3 Initialization The HIP6004 automatically initializes upon receipt of power. Special sequencing of the input supplies is not necessary. The Power-On Reset (POR) function continually monitors the input supply voltages. The POR monitors the bias voltage at the VCC pin and the input voltage (VIN) on the OCSET pin. The level on OCSET is equal to VIN less a fixed voltage drop (see over-current protection). The POR function initiates soft start operation after both input supply voltages exceed their POR thresholds. For operation with a single +12V power source, VIN and VCC are equivalent and the +12V power source must exceed the rising VCC threshold before POR initiates operation. Soft Start The POR function initiates the soft start sequence. An internal 10μA current source charges an external capacitor (CSS) on the SS pin to 4V. Soft start clamps the error amplifier output (COMP pin) and reference input (+ terminal of error amp) to the SS pin voltage. Figure 3 shows the soft start interval with CSS = 0.1μF. Initially the clamp on the error amplifier (COMP pin) controls the converter’s output voltage. At t1 in Figure 3, the SS voltage reaches the valley of the oscillator’s triangle wave. The oscillator’s triangular waveform is compared to the ramping error amplifier voltage. This generates PHASE pulses 5 Over-Current Protection The over-current function protects the converter from a shorted output by using the upper MOSFETs on-resistance, rDS(ON) to monitor the current. This method enhances the converter’s efficiency and reduces cost by eliminating a current sensing resistor. SOFT-START Functional Description FIGURE 3. SOFT START INTERVAL 4V 2V 0V OUTPUT INDUCTOR 4 • 10 Fs ≈ 200kHz – --------------------R T ( kΩ ) t2 TIME (5ms/DIV.) 7 15A 10A 5A 0A TIME (20ms/DIV.) FIGURE 4. OVER-CURRENT OPERATION The over-current function cycles the soft-start function in a hiccup mode to provide fault protection. A resistor (ROCSET) programs the over-current trip level. An internal 200μA current sink develops a voltage across ROCSET that is referenced to VIN . When the voltage across the upper MOSFET (also HIP6004 referenced to VIN) exceeds the voltage across ROCSET , the over-current function initiates a soft-start sequence. The softstart function discharges CSS with a 10μA current sink and inhibits PWM operation. The soft-start function recharges CSS , and PWM operation resumes with the error amplifier clamped to the SS voltage. Should an overload occur while recharging CSS, the soft start function inhibits PWM operation while fully charging CSS to 4V to complete its cycle. Figure 4 shows this operation with an overload condition. Note that the inductor current increases to over 15A during the CSS charging interval and causes an over-current trip. The converter dissipates very little power with this method. The measured input power for the conditions of Figure 4 is 2.5W. I OCSET • R OCSET I PEAK = --------------------------------------------------r DS ( ON ) where IOCSET is the internal OCSET current source (200μA typical). The OC trip point varies mainly due to the MOSFETs rDS(ON) variations. To avoid over-current tripping in the normal operating load range, find the ROCSET resistor from the equation above with: 1. The maximum rDS(ON) at the highest junction temperature. 2. The minimum IOCSET from the specification table. 3. Determine IPEAK for I PEAK > I OUT ( MAX ) + ( ΔI ) ⁄ 2 , where ΔI is the output inductor ripple current. For an equation for the ripple current see the section under component guidelines titled ‘Output Inductor Selection’. The over-current function will trip at a peak inductor current (IPEAK) determined by: TABLE 1. OUTPUT VOLTAGE PROGRAM PIN NAME PIN NAME VID4 VID3 VID2 VID1 VID0 NOMINAL OUTPUT VOLTAGE DACOUT VID4 VID3 VID2 VID1 VID0 NOMINAL OUTPUT VOLTAGE DACOUT 0 1 1 1 1 1.30 1 1 1 1 1 2.0 0 1 1 1 0 1.35 1 1 1 1 0 2.1 0 1 1 0 1 1.40 1 1 1 0 1 2.2 0 1 1 0 0 1.45 1 1 1 0 0 2.3 0 1 0 1 1 1.50 1 1 0 1 1 2.4 0 1 0 1 0 1.55 1 1 0 1 0 2.5 0 1 0 0 1 1.60 1 1 0 0 1 2.6 0 1 0 0 0 1.65 1 1 0 0 0 2.7 0 0 1 1 1 1.70 1 0 1 1 1 2.8 0 0 1 1 0 1.75 1 0 1 1 0 2.9 0 0 1 0 1 1.80 1 0 1 0 1 3.0 0 0 1 0 0 1.85 1 0 1 0 0 3.1 0 0 0 1 1 1.90 1 0 0 1 1 3.2 0 0 0 1 0 1.95 1 0 0 1 0 3.3 0 0 0 0 1 2.00 1 0 0 0 1 3.4 0 0 0 0 0 2.05 1 0 0 0 0 3.5 NOTE: 0 = connected to GND or VSS, 1 = OPEN. A small ceramic capacitor should be placed in parallel with ROCSET to smooth the voltage across ROCSET in the presence of switching noise on the input voltage. Output Voltage Program The output voltage of a HIP6004 converter is programmed to discrete levels between 1.3VDC and 3.5VDC . The voltage identification (VID) pins program an internal voltage reference (DACOUT) with a 5-bit digital-to-analog converter (DAC). The level of DACOUT also sets the PGOOD and OVP thresholds. Table 1 specifies the DACOUT voltage for the 32 combinations of open or short 6 connections on the VID pins. The output voltage should not be adjusted while the converter is delivering power. Remove input power before changing the output voltage. Adjusting the output voltage during operation could toggle the PGOOD signal and exercise the overvoltage protection. The DAC function is a precision non-inverting summation amplifier shown in Figure 5. The resistor values shown are only approximations of the actual precision values used. Grounding any combination of the VID pins increases the DACOUT voltage. The ‘open’ circuit voltage on the VID pins is the band gap reference voltage, 1.26V. HIP6004 Figure 7 shows the circuit traces that require additional layout consideration. Use single point and ground plane construction for the circuits shown. Minimize any leakage current paths on the SS PIN and locate the capacitor, Css close to the SS pin because the internal current source is only 10μA. Provide local VCC decoupling between VCC and GND pins. Locate the capacitor, CBOOT as close as practical to the BOOT and PHASE pins. BAND GAP REFERENCE 1.26V 12kΩ 3.6kΩ VID4 VID3 2.7kΩ 5.4kΩ VID2 VID1 VID0 ERROR AMPLIFIER DACOUT + - + COMP - Figure 8 highlights the voltage-mode control loop for a synchronous-rectified buck converter. The output voltage (VOUT) is regulated to the Reference voltage level. The error amplifier (Error Amp) output (VE/A) is compared with the oscillator (OSC) triangular wave to provide a pulse-width modulated (PWM) wave with an amplitude of VIN at the PHASE node. The PWM wave is smoothed by the output filter (LO and CO). 1.7kΩ 10.7kΩ 21.5kΩ DAC Feedback Compensation FB 2.9kΩ FIGURE 5. DAC FUNCTION SCHEMATIC BOOT Application Guidelines As in any high frequency switching converter, layout is very important. Switching current from one power device to another can generate voltage transients across the impedances of the interconnecting bond wires and circuit traces. These interconnecting impedances should be minimized by using wide, short printed circuit traces. The critical components should be located as close together as possible, using ground plane construction or single point grounding. VIN HIP6004 VCC SS VOUT PHASE +12V GND FIGURE 7. PRINTED CIRCUIT BOARD SMALL SIGNAL LAYOUT GUIDELINES OSC Q1 LO PHASE PWM COMPARATOR VOUT CIN CO LO - DRIVER + PGND VE/A RETURN ZFB - FIGURE 6. PRINTED CIRCUIT BOARD POWER AND GROUND PLANES OR ISLANDS Figure 6 shows the critical power components of the converter. To minimize the voltage overshoot the interconnecting wires indicated by heavy lines should be part of ground or power plane in a printed circuit board. The components shown in Figure 6 should be located as close together as possible. Please note that the capacitors CIN and CO each represent numerous physical capacitors. Locate the HIP6004 within 3 inches of the MOSFETs, Q1 and Q2. The circuit traces for the MOSFETs’ gate and source connections from the HIP6004 must be sized to handle up to 1A peak current. ERROR AMP VOUT CO ESR (PARASITIC) ZIN + 7 PHASE LOAD D2 VIN DRIVER Δ VOSC Q2 CO Q2 CVCC CSS HIP6004 LGATE Q1 LO CBOOT Layout Considerations UGATE +VIN D1 LOAD 12kΩ REFERENCE DETAILED COMPENSATION COMPONENTS C2 C1 ZFB VOUT ZIN C3 R2 R3 R1 COMP - FB + HIP6004 DACOUT FIGURE 8. VOLTAGE-MODE BUCK CONVERTER COMPENSATION DESIGN HIP6004 Modulator Break Frequency Equations 1 F LC = --------------------------------------2π • L O • C O 1 F ESR = ---------------------------------------2π • ESR • C O The compensation network consists of the error amplifier (internal to the HIP6004) and the impedance networks ZIN and ZFB. The goal of the compensation network is to provide a closed loop transfer function with the highest 0dB crossing frequency (f0dB) and adequate phase margin. Phase margin is the difference between the closed loop phase at f0dB and 180 degrees. The equations below relate the compensation network’s poles, zeros and gain to the components (R1, R2, R3, C1, C2, and C3) in Figure 8. Use these guidelines for locating the poles and zeros of the compensation network: 1. Pick Gain (R2/R1) for desired converter bandwidth 2. Place 1STZero Below Filter’s Double Pole (~75% FLC) 3. Place 2ND Zero at Filter’s Double Pole 4. Place 1ST Pole at the ESR Zero 5. Place 2ND Pole at Half the Switching Frequency 6. Check Gain against Error Amplifier’s Open-Loop Gain 7. Estimate Phase Margin - Repeat if Necessary Compensation Break Frequency Equations 1 F Z1 = ---------------------------------2π • R2 • C1 1 F P1 = -----------------------------------------------------C1 • C2 2π • R 2 • ⎛ ----------------------⎞ ⎝ C1 + C2⎠ F Z2 = 2π • ( R1 + R3 ) • C3 1 F P2 = ---------------------------------2π • R3 • C3 Figure 9 shows an asymptotic plot of the DC-DC converter’s gain vs. frequency. The actual Modulator Gain has a high gain peak due to the high Q factor of the output filter and is not shown in Figure 9. Using the above guidelines should give a Compensation Gain similar to the curve plotted. The open loop error amplifier gain bounds the compensation gain. Check the compensation gain at FP2 with the capabilities of the error amplifier. The Closed Loop Gain is constructed on the log-log graph of Figure 9 by adding the Modulator Gain (in dB) to the Compensation Gain (in dB). This is equivalent to multiplying the modulator transfer function to the compensation transfer function and plotting the gain. The compensation gain uses external impedance networks ZFB and ZIN to provide a stable, high bandwidth (BW) overall loop. A stable control loop has a gain crossing with -20dB/decade slope and a phase margin greater than 45 degrees. Include worst case component variations when determining phase margin. 8 100 FZ1 FZ2 FP1 FP2 80 OPEN LOOP ERROR AMP GAIN 60 GAIN (dB) The modulator transfer function is the small-signal transfer function of VOUT/VE/A. This function is dominated by a DC Gain and the output filter (LO and CO), with a double pole break frequency at FLC and a zero at FESR. The DC Gain of the modulator is simply the input voltage (VIN) divided by the peak-to-peak oscillator voltage ΔVOSC . 40 20 20LOG (R2/R1) 0 20LOG (VIN/ΔVOSC) MODULATOR GAIN -20 -40 -60 COMPENSATION GAIN CLOSED LOOP GAIN FLC 10 100 1K FESR 10K 100K 1M 10M FREQUENCY (Hz) FIGURE 9. ASYMPTOTIC BODE PLOT OF CONVERTER GAIN Component Selection Guidelines Output Capacitor Selection An output capacitor is required to filter the output and supply the load transient current. The filtering requirements are a function of the switching frequency and the ripple current. The load transient requirements are a function of the slew rate (di/dt) and the magnitude of the transient load current. These requirements are generally met with a mix of capacitors and careful layout. Modern microprocessors produce transient load rates above 1A/ns. High frequency capacitors initially supply the transient and slow the current load rate seen by the bulk capacitors. The bulk filter capacitor values are generally determined by the ESR (effective series resistance) and voltage rating requirements rather than actual capacitance requirements. High frequency decoupling capacitors should be placed as close to the power pins of the load as physically possible. Be careful not to add inductance in the circuit board wiring that could cancel the usefulness of these low inductance components. Consult with the manufacturer of the load on specific decoupling requirements. For example, Intel recommends that the high frequency decoupling for the Pentium Pro be composed of at least forty (40) 1μF ceramic capacitors in the 1206 surface-mount package. Use only specialized low-ESR capacitors intended for switching-regulator applications for the bulk capacitors. The bulk capacitor’s ESR will determine the output ripple voltage and the initial voltage drop after a high slew-rate transient. An aluminum electrolytic capacitor’s ESR value is related to the case size with lower ESR available in larger case sizes. However, the equivalent series inductance (ESL) of these capacitors increases with case size and can reduce the usefulness of the capacitor to high slew-rate transient loading. Unfortunately, ESL is not a specified parameter. Work with your capacitor supplier and measure the capacitor’s impedance with frequency to select a suitable component. In HIP6004 most cases, multiple electrolytic capacitors of small case size perform better than a single large case capacitor. Output Inductor Selection The output inductor is selected to meet the output voltage ripple requirements and minimize the converter’s response time to the load transient. The inductor value determines the converter’s ripple current and the ripple voltage is a function of the ripple current. The ripple voltage and current are approximated by the following equations: ΔI = VIN - VOUT VOUT • Fs x L VIN ΔVOUT = ΔI x ESR Increasing the value of inductance reduces the ripple current and voltage. However, the large inductance values reduce the converter’s response time to a load transient. One of the parameters limiting the converter’s response to a load transient is the time required to change the inductor current. Given a sufficiently fast control loop design, the HIP6004 will provide either 0% or 100% duty cycle in response to a load transient. The response time is the time required to slew the inductor current from an initial current value to the transient current level. During this interval the difference between the inductor current and the transient current level must be supplied by the output capacitor. Minimizing the response time can minimize the output capacitance required. The response time to a transient is different for the application of load and the removal of load. The following equations give the approximate response time interval for application and removal of a transient load: tRISE = L x ITRAN VIN - VOUT tFALL = L x ITRAN VOUT where: ITRAN is the transient load current step, tRISE is the response time to the application of load, and tFALL is the response time to the removal of load. With a +5V input source, the worst case response time can be either at the application or removal of load and dependent upon the DACOUT setting. Be sure to check both of these equations at the minimum and maximum output levels for the worst case response time. With a +12V input, and output voltage level equal to DACOUT, tFALL is the longest response time. Input Capacitor Selection Use a mix of input bypass capacitors to control the voltage overshoot across the MOSFETs. Use small ceramic capacitors for high frequency decoupling and bulk capacitors to supply the current needed each time Q1 turns on. Place the small ceramic capacitors physically close to the MOSFETs and between the drain of Q1 and the source of Q2. The important parameters for the bulk input capacitor are the voltage rating and the RMS current rating. For reliable operation, select the bulk capacitor with voltage and current ratings above the maximum input voltage and largest RMS 9 current required by the circuit. The capacitor voltage rating should be at least 1.25 times greater than the maximum input voltage and a voltage rating of 1.5 times is a conservative guideline. The RMS current rating requirement for the input capacitor of a buck regulator is approximately 1/2 the DC load current. For a through hole design, several electrolytic capacitors (Panasonic HFQ series or Nichicon PL series or Sanyo MV-GX or equivalent) may be needed. For surface mount designs, solid tantalum capacitors can be used, but caution must be exercised with regard to the capacitor surge current rating. These capacitors must be capable of handling the surge-current at power-up. The TPS series available from AVX, and the 593D series from Sprague are both surge current tested. MOSFET Selection/Considerations The HIP6004 requires 2 N-Channel power MOSFETs. These should be selected based upon rDS(ON) , gate supply requirements, and thermal management requirements. In high-current applications, the MOSFET power dissipation, package selection and heatsink are the dominant design factors. The power dissipation includes two loss components; conduction loss and switching loss. The conduction losses are the largest component of power dissipation for both the upper and the lower MOSFETs. These losses are distributed between the two MOSFETs according to duty factor (see the equations below). Only the upper MOSFET has switching losses, since the Schottky rectifier clamps the switching node before the synchronous rectifier turns on. These equations assume linear voltage-current transitions and do not adequately model power loss due the reverse-recovery of the lower MOSFETs body diode. The gate-charge losses are dissipated by the HIP6004 and don't heat the MOSFETs. However, large gate-charge increases the switching interval, tSW which increases the upper MOSFET switching losses. Ensure that both MOSFETs are within their maximum junction temperature at high ambient temperature by calculating the temperature rise according to package thermal-resistance specifications. A separate heatsink may be necessary depending upon MOSFET power, package type, ambient temperature and air flow. PUPPER = Io2 x rDS(ON) x D + 1 Io x VIN x tSW x FS 2 2 PLOWER = Io x rDS(ON) x (1 - D) Where: D is the duty cycle = VOUT / VIN , tSW is the switching interval, and FS is the switching frequency. Standard-gate MOSFETs are normally recommended for use with the HIP6004. However, logic-level gate MOSFETs can be used under special circumstances. The input voltage, upper gate drive level, and the MOSFETs absolute gate-tosource voltage rating determine whether logic-level MOSFETs are appropriate. HIP6004 +12V VCC HIP6004 +12V DBOOT BOOT CBOOT UGATE HIP6004 Q1 - Q2 PGND D2 NOTE: VG-S ≈ VCC GND FIGURE 10. UPPER GATE DRIVE - BOOTSTRAP OPTION Figure 10 shows the upper gate drive (BOOT pin) supplied by a bootstrap circuit from VCC. The boot capacitor, CBOOT develops a floating supply voltage referenced to the PHASE pin. This supply is refreshed each cycle to a voltage of VCC less the boot diode drop (VD) when the lower MOSFET, Q2 turns on. Logic-level MOSFETs can only be used if the MOSFETs absolute gate-to-source voltage rating exceeds the maximum voltage applied to VCC . Figure 11 shows the upper gate drive supplied by a direct connection to VCC . This option should only be used in converter systems where the main input voltage is +5VDC or less. The peak upper gate-to-source voltage is approximately VCC less the input supply. For +5V main power and +12VDC for the bias, the gate-to-source voltage of Q1 is 7V. A logiclevel MOSFET is a good choice for Q1 and a logic-level MOSFET can be used for Q2 if its absolute gate-to-source voltage rating exceeds the maximum voltage applied to VCC . 10 Q1 PHASE NOTE: VG-S ≈ VCC -VD LGATE BOOT UGATE PHASE + +5V OR LESS VCC +5V or +12V + VD - - + LGATE PGND GND NOTE: VG-S ≈ VCC -5V Q2 D2 NOTE: VG-S ≈ VCC FIGURE 11. UPPER GATE DRIVE - DIRECT VCC DRIVE OPTION Schottky Selection Rectifier D2 is a clamp that catches the negative inductor swing during the dead time between turning off the lower MOSFET and turning on the upper MOSFET. The diode must be a Schottky type to prevent the lossy parasitic MOSFET body diode from conducting. It is acceptable to omit the diode and let the body diode of the lower MOSFET clamp the negative inductor swing, but efficiency will drop one or two percent as a result. The diode's rated reverse breakdown voltage must be greater than the maximum input voltage. HIP6004 HIP6004 DC-DC Converter Application Circuit Materials and circuit board description, can be found in Application Note AN9672. Figure 12 shows an application circuit of a DC-DC Converter for an Intel Pentium Pro microprocessor. Detailed information on the circuit, including a complete Bill-of- VIN = L1 - 1μH F1 +5V OR +12V C1 5x 1000μF 2x 1μF 2N6394 +12V 2K D1 0.1μF 1000pF VSEN 1 RT VID0 VID1 VID2 VID3 VID4 FB OVP 18 19 2 OCSET MONITOR AND PROTECTION SS 3 0.1μF VCC 20 4 5 6 7 8 1K 12 PGOOD 15 BOOT OSC 14 UGATE 13 PHASE HIP6004 - - 17 LGATE + + 16 PGND 9 11 COMP 2.2nF GND 20K 8.2nF 0.1μF 15 Component Selection Notes; C0 - 9 Each 1000μF 6.3W VDC, Sanyo MV-GX or Equivalent C1 - 5 Each 330μF 25W VDC, Sanyo MV-GX or Equivalent L2 - Core: Micrometals T50-52B; Each Winding: 10 Turns of 16AWG L1 - Core: Micrometals T50-52; Winding: 5 Turns of 18AWG D1 - 1N4148 or Equivalent D2 - 3A, 40V Schottky, Motorola MBR340 or Equivalent Q1, Q2 - Intersil MOSFET; RFP70N03 FIGURE 12. PENTIUM PRO DC-DC CONVERTER 11 L2 3μH D/A 10 1.33K 0.1μF Q1 Q2 D2 CO 9x 1000μF +VO HIP6004 Small Outline Plastic Packages (SOIC) M20.3 (JEDEC MS-013-AC ISSUE C) N INDEX AREA 20 LEAD WIDE BODY SMALL OUTLINE PLASTIC PACKAGE 0.25(0.010) M H B M INCHES E -B- 1 2 3 L SEATING PLANE -A- h x 45o A D -C- e A1 B 0.25(0.010) M C 0.10(0.004) C A M SYMBOL MIN MAX MIN MAX NOTES A 0.0926 0.1043 2.35 2.65 - A1 0.0040 0.0118 0.10 0.30 - B 0.013 0.0200 0.33 0.51 9 C 0.0091 0.0125 0.23 0.32 - D 0.4961 0.5118 12.60 13.00 3 E 0.2914 0.2992 7.40 7.60 4 e µα B S 0.050 BSC 1.27 BSC - H 0.394 0.419 10.00 10.65 - h 0.010 0.029 0.25 0.75 5 L 0.016 0.050 0.40 1.27 6 N α NOTES: MILLIMETERS 20 0o 20 8o 0o 7 8o Rev. 0 12/93 1. Symbols are defined in the “MO Series Symbol List” in Section 2.2 of Publication Number 95. 2. Dimensioning and tolerancing per ANSI Y14.5M-1982. 3. Dimension “D” does not include mold flash, protrusions or gate burrs. Mold flash, protrusion and gate burrs shall not exceed 0.15mm (0.006 inch) per side. 4. Dimension “E” does not include interlead flash or protrusions. Interlead flash and protrusions shall not exceed 0.25mm (0.010 inch) per side. 5. The chamfer on the body is optional. If it is not present, a visual index feature must be located within the crosshatched area. 6. “L” is the length of terminal for soldering to a substrate. 7. “N” is the number of terminal positions. 8. Terminal numbers are shown for reference only. 9. The lead width “B”, as measured 0.36mm (0.014 inch) or greater above the seating plane, shall not exceed a maximum value of 0.61mm (0.024 inch) 10. Controlling dimension: MILLIMETER. Converted inch dimensions are not necessarily exact. All Intersil semiconductor products are manufactured, assembled and tested under ISO9000 quality systems certification. Intersil semiconductor products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design and/or specifications at any time without notice. Accordingly, the reader is cautioned to verify that data sheets are current before placing orders. Information furnished by Intersil is believed to be accurate and reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Intersil or its subsidiaries. For information regarding Intersil Corporation and its products, see web site www.intersil.com Sales Office Headquarters NORTH AMERICA Intersil Corporation P. O. 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