INTERSIL HIP6004DCB

HIP6004D
®
Data Sheet
July 13, 2005
Buck and Synchronous-Rectifier (PWM)
Controller and Output Voltage Monitor
The HIP6004D provides complete control and protection for
a DC-DC converter optimized for high-performance
microprocessor applications. It is designed to drive two
N-Channel MOSFETs in a synchronous-rectified buck
topology. The HIP6004D integrates all of the control, output
adjustment, monitoring and protection functions into a single
package.
The output voltage of the converter is easily adjusted and
precisely regulated. The HIP6004D includes a fully TTLcompatible 5-input digital-to-analog converter (DAC) that
adjusts the output voltage from 1.1VDC to 1.85VDC in 25mV
increments steps. The precision reference and voltagemode regulator hold the selected output voltage to within
±1% over temperature and line voltage variations.
FN4855.3
Features
• Drives Two N-Channel MOSFETs
• Operates from +5V or +12V Input
• Simple Single-Loop Control Design
- Voltage-Mode PWM Control
• Fast Transient Response
- High-Bandwidth Error Amplifier
- Full 0% to 100% Duty Ratio
• Excellent Output Voltage Regulation
- ±1% Over Line Voltage and Temperature
• TTL-Compatible 5-Bit Digital-to-Analog Output
Voltage Selection
- 25mV Binary Steps . . . . . . . . . 1.100VDC to 1.850VDC
• Power-Good Output Voltage Monitor
The HIP6004D provides simple, single feedback loop,
voltage-mode control with fast transient response. It includes
a 200kHz free-running triangle-wave oscillator that is
adjustable from below 50kHz to over 1MHz. The error
amplifier features a 15MHz gain-bandwidth product and
6V/μs slew rate which enables high converter bandwidth for
fast transient performance. The resulting PWM duty ratio
ranges from 0% to 100%.
• Over-Voltage and Over-Current Fault Monitors
- Does Not Require Extra Current Sensing Element,
Uses MOSFET’s rDS(ON)
The HIP6004D monitors the output voltage with a window
comparator that tracks the DAC output and issues a Power
Good signal when the output is within ±10%. The HIP6004D
protects against over-current and overvoltage conditions by
inhibiting PWM operation. Additional built-in overvoltage
protection triggers an external SCR to crowbar the input
supply. The HIP6004D monitors the current by using the
rDS(ON) of the upper MOSFET which eliminates the need for
a current sensing resistor.
• QFN Package:
- Compliant to JEDEC PUB95 MO-220
QFN - Quad Flat No Leads - Package Outline
- Near Chip Scale Package footprint, which improves
PCB efficiency and has a thinner profile
Ordering Information
• High-Power DC-DC Regulators
PART NUMBER
TEMP.
RANGE (oC)
PACKAGE
PKG.
DWG. #
HIP6004DCB
0 to 70
20 Ld SOIC
M20.3
HIP6004DCBZ
(See Note)
0 to 70
20 Ld SOIC
(Pb-free)
M20.3
HIP6004DCR
0 to 70
20 Ld 5x5 QFN
L20.5x5
HIP6004DCRZ
(See Note)
0 to 70
20 Ld 5x5 QFN
(Pb-free)
L20.5x5
• Small Converter Size
- Constant Frequency Operation
- 200kHz Free-Running Oscillator Programmable from
50kHz to over 1MHz
• Pb-Free plus anneal available (RoHS compliant)
Applications
• Power Supply for K7™, and Other Microprocessors
• Low-Voltage Distributed Power Supplies
Add “-T” suffix for tape and reel.
NOTE: Intersil Pb-free plus anneal products employ special Pb-free material
sets; molding compounds/die attach materials and 100% matte tin plate
termination finish, which are RoHS compliant and compatible with both SnPb
and Pb-free soldering operations. Intersil Pb-free products are MSL classified
at Pb-free peak reflow temperatures that meet or exceed the Pb-free
requirements of IPC/JEDEC J STD-020.
1
CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures.
1-888-INTERSIL or 1-888-468-3774 | Intersil (and design) is a registered trademark of Intersil Americas Inc.
Copyright © Intersil Americas Inc. 2003, 2005. All Rights Reserved.
All other trademarks mentioned are the property of their respective owners.
HIP6004D
Pinouts
VSEN
1
OCSET
2
19 OVP
OCSET
VSEN
RT
OVP
HIP6004D (QFN)
TOP VIEW
SS
HIP6004D (SOIC, TSSOP)
TOP VIEW
SS
3
18 VCC
20
19
18
17
16
VID0
4
17 LGATE
VID1
5
16 PGND
VID2
6
15 BOOT
VID3
7
14 UGATE
VID4
8
13 PHASE
COMP
9
12 PGOOD
20 RT
VID1
2
15 VCC
14 LGATE
GND
21
VID3
4
12 BOOT
VID4
5
11 UGATE
6
7
8
9
10
PHASE
13 PGND
PGOOD
3
GND
VID2
FB
11 GND
1
COMP
FB 10
VID0
Typical Application
+12V
VCC
VIN = +5V OR +12V
HIP6004D
PGOOD
MONITOR AND
PROTECTION
SS
OVP
OCSET
EN
BOOT
RT
OSC
VID0
VID1
VID2
VID3
VID4
UGATE
PHASE
+VOUT
D/A
+
-
FB
COMP
2
LGATE
+
PGND
VSEN
GND
FN4855.3
July 13, 2005
HIP6004D
Block Diagram
VCC
VSEN
POWER-ON
RESET (POR)
110%
+
90%
+
-
OVERVOLTAGE
+
-
115%
1µA
OVP
SOFTSTART
+
- OVERCURRENT
OCSET
REFERENCE
VID0
VID1
VID2
VID3
VID4
PGOOD
TTL D/A
CONVERTER
(DAC)
200µA
DACOUT
BOOT
4V
UGATE
PWM
COMPARATOR
+
-
+
-
ERROR
AMP
FB
SS
PHASE
GATE
INHIBIT CONTROL
LOGIC
PWM
LGATE
PGND
COMP
GND
OSCILLATOR
RT
3
FN4855.3
July 13, 2005
HIP6004D
Absolute Maximum Ratings
Thermal Information
Supply Voltage, VCC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .+15V
Boot Voltage, VBOOT - VPHASE . . . . . . . . . . . . . . . . . . . . . . . .+15V
Input, Output or I/O Voltage . . . . . . . . . . . . GND -0.3V to VCC +0.3V
ESD Classification . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Class 2
Thermal Resistance
θJA (oC/W) θJC (oC/W)
SOIC Package (Note 1) . . . . . . . . . . . .
65
NA
QFN Package (Notes 2, 3). . . . . . . . . .
33
5
Maximum Junction Temperature . . . . . . . . . . . . . . . . . . . . . . 150oC
Maximum Storage Temperature Range . . . . . . . . . -65oC to 150oC
Maximum Lead Temperature (Soldering 10s) . . . . . . . . . . . . 300oC
(SOIC - Lead Tips Only)
Operating Conditions
Supply Voltage, VCC . . . . . . . . . . . . . . . . . . . . . . . . . . . +12V ±10%
Ambient Temperature Range. . . . . . . . . . . . . . . . . . . . . 0oC to 70oC
CAUTION: Stresses above those listed in “Absolute Maximum Ratings” may cause permanent damage to the device. This is a stress only rating and operation of the
device at these or any other conditions above those indicated in the operational sections of this specification is not implied.
NOTES:
1. θJA is measured with the component mounted on a high effective thermal conductivity test board in free air. See Tech Brief TB379 for details.
2. θJA is measured in free air with the component mounted on a high effective thermal conductivity test board with “direct attach” features. See
Tech Brief TB379.
3. For θJC, the “case temp” location is the center of the exposed metal pad on the package underside. See Tech Brief TB379
Electrical Specifications
Recommended Operating Conditions, Unless Otherwise Noted
PARAMETER
SYMBOL
TEST CONDITIONS
MIN
TYP
MAX
UNITS
UGATE and LGATE Open
-
5
-
mA
Rising VCC Threshold
VOCSET = 4.5V
-
-
10.4
V
Falling VCC Threshold
VOCSET = 4.5V
8.2
-
-
V
-
1.26
-
V
VCC SUPPLY CURRENT
Nominal Supply
ICC
POWER-ON RESET
Rising VOCSET Threshold
OSCILLATOR
Free Running Frequency
RT = OPEN
185
200
215
kHz
Total Variation
6kΩ < RT to GND < 200kΩ
-15
-
+15
%
-
1.9
-
VP-P
DAC (VID0-VID4) Input Low Voltage
-
-
0.8
V
DAC (VID0-VID4) Input High Voltage
2.0
-
-
V
DACOUT Voltage Accuracy
-1.0
-
+1.0
%
ΔVOSC
Ramp Amplitude
RT = Open
REFERENCE AND DAC
ERROR AMPLIFIER
DC Gain
Gain-Bandwidth Product
-
88
-
dB
-
15
-
MHz
-
6
-
V/μs
350
500
-
mA
-
5.5
10
Ω
300
450
-
mA
-
3.5
6.5
Ω
-
115
120
%
VOCSET = 4.5VDC
170
200
230
µA
VSEN = 5.5V, VOVP = 0V
60
-
-
mA
-
10
-
µA
GBWP
Slew Rate
SR
COMP = 10pF
GATE DRIVERS
Upper Gate Source
IUGATE
VBOOT - VPHASE = 12V, VUGATE = 6V
Upper Gate Sink
RUGATE
ILGATE = 0.3A
Lower Gate Source
ILGATE
VCC = 12V, VLGATE = 6V
Lower Gate Sink
RLGATE
ILGATE = 0.3A
PROTECTION
Over-Voltage Trip (VSEN/DACOUT)
OCSET Current Source
IOCSET
OVP Sourcing Current
IOVP
Soft Start Current
ISS
4
FN4855.3
July 13, 2005
HIP6004D
Electrical Specifications
Recommended Operating Conditions, Unless Otherwise Noted (Continued)
PARAMETER
SYMBOL
TEST CONDITIONS
MIN
TYP
MAX
UNITS
POWER GOOD
Upper Threshold (VSEN/DACOUT)
VSEN Rising
106
-
111
%
Lower Threshold (VSEN/DACOUT)
VSEN Falling
89
-
94
%
Hysteresis (VSEN/DACOUT)
PGOOD Voltage Low
VPGOOD
Upper and Lower Threshold
-
2
-
%
IPGOOD = -5mA
-
0.5
-
V
Typical Performance Curves
80
CGATE = 3300pF
70
60
RT PULLUP
TO +12V
ICC (mA)
RESISTANCE (kΩ)
1000
100
50
CUPPER = CLOWER = CGATE
40
CGATE = 1000pF
30
10
20
RT PULLDOWN TO VSS
CGATE = 10pF
10
10
100
1000
0
100
200
SWITCHING FREQUENCY (kHz)
FIGURE 1. RT RESISTANCE vs FREQUENCY
Functional Pin Descriptions
VSEN
1
20 RT
OCSET
2
19 OVP
300
400
500
600
700
800
the converter over-current (OC) trip point according to the
following equation:
I OCSET x R OCSET
I PEAK = ----------------------------------------------------r DS ( ON )
SS
3
18 VCC
4
17 LGATE
An over-current trip cycles the soft-start function.
VID1
5
16 PGND
VID2
6
15 BOOT
SS (Pin 3)
VID3
7
14 UGATE
VID4
8
13 PHASE
COMP
9
12 PGOOD
11 GND
VSEN (Pin 1)
This pin is connected to the converter’s output voltage. The
PGOOD and OVP comparator circuits use this signal to
report output voltage status and for overvoltage protection.
OCSET (Pin 2)
Connect a resistor (ROCSET) from this pin to the drain of the
upper MOSFET. ROCSET , an internal 200µA current source
(IOCS), and the upper MOSFET on-resistance (rDS(ON)) set
5
1000
FIGURE 2. BIAS SUPPLY CURRENT vs FREQUENCY
VID0
FB 10
900
SWITCHING FREQUENCY (kHz)
Connect a capacitor from this pin to ground. This capacitor,
along with an internal 10µA current source, sets the softstart interval of the converter.
VID0-4 (Pins 4-8)
VID0-4 are the input pins to the 5-bit DAC. The states of
these five pins program the internal voltage reference
(DACOUT). The level of DACOUT sets the converter output
voltage. It also sets the PGOOD and OVP thresholds. Table
1 specifies DACOUT for the all combinations of DAC inputs.
COMP (Pin 9) and FB (Pin 10)
COMP and FB are the available external pins of the error
amplifier. The FB pin is the inverting input of the error
amplifier and the COMP pin is the error amplifier output.
These pins are used to compensate the voltage-control
feedback loop of the converter.
FN4855.3
July 13, 2005
HIP6004D
GND (Pin 11)
Functional Description
Signal ground for the IC. All voltage levels are measured
with respect to this pin.
Initialization
PGOOD (Pin 12)
PGOOD is an open collector output used to indicate the
status of the converter output voltage. This pin is pulled low
when the converter output is not within ±10% of the
DACOUT reference voltage. Exception to this behavior is the
‘11111’ VID pin combination which disables the converter; in
this case PGOOD asserts a high level.
PHASE (Pin 13)
Connect the PHASE pin to the upper MOSFET source. This
pin is used to monitor the voltage drop across the MOSFET
for over-current protection. This pin also provides the return
path for the upper gate drive.
UGATE (Pin 14)
Connect UGATE to the upper MOSFET gate. This pin
provides the gate drive for the upper MOSFET.
BOOT (Pin 15)
This pin provides bias voltage to the upper MOSFET driver.
A bootstrap circuit may be used to create a BOOT voltage
suitable to drive a standard N-Channel MOSFET.
PGND (Pin 16)
This is the power ground connection. Tie the lower MOSFET
source to this pin.
LGATE (Pin 17)
Connect LGATE to the lower MOSFET gate. This pin
provides the gate drive for the lower MOSFET.
VCC (Pin 18)
Provide a 12V bias supply for the chip to this pin.
OVP (Pin 19)
The OVP pin can be used to drive an external SCR in the
event of an overvoltage condition. Output rising 15% more
than the DAC-set voltage triggers a high output on this pin
and disables PWM gate drive circuitry.
RT (Pin 20)
The HIP6004D automatically initializes upon receipt of power.
Special sequencing of the input supplies is not necessary. The
Power-On Reset (POR) function continually monitors the input
supply voltages. The POR monitors the bias voltage at the VCC
pin and the input voltage (VIN) on the OCSET pin. The level on
OCSET is equal to VIN less a fixed voltage drop (see overcurrent protection). The POR function initiates soft start
operation after both input supply voltages exceed their POR
thresholds. For operation with a single +12V power source, VIN
and VCC are equivalent and the +12V power source must
exceed the rising VCC threshold before POR initiates
operation.
Soft Start
The POR function initiates the soft start sequence. An internal
10µA current source charges an external capacitor (CSS) on
the SS pin to 4V. Soft start clamps the error amplifier output
(COMP pin) and reference input (+ terminal of error amp) to the
SS pin voltage. Figure 3 shows the soft start interval with
CSS = 0.1μF. Initially the clamp on the error amplifier (COMP
pin) controls the converter’s output voltage. At t1 in Figure 3,
the SS voltage reaches the valley of the oscillator’s triangle
wave. The oscillator’s triangular waveform is compared to the
ramping error amplifier voltage. This generates PHASE pulses
of increasing width that charge the output capacitor(s). This
interval of increasing pulse width continues to t2 . With sufficient
output voltage, the clamp on the reference input controls the
output voltage. This is the interval between t2 and t3 in Figure 3.
At t3 the SS voltage exceeds the DACOUT voltage and the
output voltage is in regulation. This method provides a rapid
and controlled output voltage rise. The PGOOD signal toggles
‘high’ when the output voltage (VSEN pin) is within ±10% of
DACOUT. The 2% hysteresis built into the power good
comparators prevents PGOOD oscillation due to nominal
output voltage ripple.
PGOOD
(2V/DIV)
0V
This pin provides oscillator switching frequency adjustment.
By placing a resistor (RT) from this pin to GND, the nominal
200kHz switching frequency is increased according to the
following equation:
SOFT-START
(1V/DIV)
OUTPUT
VOLTAGE
(1V/DIV)
6
5 x 10
Fs ≈ 200kHz + --------------------R T ( kΩ )
(RT to GND)
Conversely, connecting a pull-up resistor (RT) from this pin
to VCC reduces the switching frequency according to the
following equation:
0V
t1
t2
t3
TIME (5ms/DIV)
FIGURE 3. SOFT START INTERVAL
7
4 x 10
Fs ≈ 200kHz – --------------------R T ( kΩ )
0V
(RT to 12V)
6
FN4855.3
July 13, 2005
HIP6004D
2. The minimum IOCSET from the specification table.
The over-current function protects the converter from a
shorted output by using the upper MOSFET’s on-resistance,
rDS(ON) to monitor the current. This method enhances the
converter’s efficiency and reduces cost by eliminating a
current sensing resistor.
3. Determine IPEAK for I PEAK > I OUT ( MAX ) + ( ΔI ) ⁄ 2 ,
where ΔI is the output inductor ripple current.
SOFT-START
Over-Current Protection
4V
A small ceramic capacitor should be placed in parallel with
ROCSET to smooth the voltage across ROCSET in the
presence of switching noise on the input voltage.
Output Voltage Program
2V
0V
OUTPUT INDUCTOR
For an equation for the ripple current see the section under
component guidelines titled ‘Output Inductor Selection’.
15A
10A
5A
0A
TIME (20ms/DIV)
FIGURE 4. OVER-CURRENT OPERATION
The over-current function cycles the soft-start function in a
hiccup mode to provide fault protection. A resistor (ROCSET)
programs the over-current trip level. An internal 200μA current
sink develops a voltage across ROCSET that is referenced to
VIN . When the voltage across the upper MOSFET (also
referenced to VIN) exceeds the voltage across ROCSET , the
over-current function initiates a soft-start sequence. The softstart function discharges CSS with a 10μA current sink and
inhibits PWM operation. The soft-start function recharges
CSS , and PWM operation resumes with the error amplifier
clamped to the SS voltage. Should an overload occur while
recharging CSS , the soft start function inhibits PWM operation
while fully charging CSS to 4V to complete its cycle. Figure 4
shows this operation with an overload condition. Note that the
inductor current increases to over 15A during the CSS
charging interval and causes an over-current trip. The
converter dissipates very little power with this method. The
measured input power for the conditions of Figure 4 is 2.5W.
The output voltage of a HIP6004D converter is programmed
to discrete levels between 1.100VDC and 1.850VDC . The
voltage identification (VID) pins program an internal voltage
reference (DACOUT) with a TTL-compatible 5-bit digital-toanalog converter (DAC). The level of DACOUT also sets the
PGOOD and OVP thresholds. Table 1 specifies the DACOUT
voltage for the 32 different combinations of connections on the
VID pins. The output voltage should not be adjusted while the
converter is delivering power. Remove input power before
changing the output voltage. Adjusting the output voltage
during operation could toggle the PGOOD signal and exercise
the overvoltage protection.
‘11111’ VID pin combination resulting in a 0V output setting
activates the Power-On Reset function and disables the gate
drives circuitry. For this specific VID combination, though,
PGOOD asserts a high level. This unusual behavior has been
implemented in order to allow for operation in dualmicroprocessor systems where AND-ing of the PGOOD signals
from two individual power converters is implemented.
Application Guidelines
Layout Considerations
As in any high frequency switching converter, layout is very
important. Switching current from one power device to another
can generate voltage transients across the impedances of the
interconnecting bond wires and circuit traces. These
interconnecting impedances should be minimized by using
wide, short printed circuit traces. The critical components
should be located as close together as possible, using ground
plane construction or single point grounding.
The over-current function will trip at a peak inductor current
(IPEAK) determined by:
I OCSET x R OCSET
I PEAK = ----------------------------------------------------r DS ( ON )
where IOCSET is the internal OCSET current source (200μA
typical). The OC trip point varies mainly due to the
MOSFET’s rDS(ON) variations. To avoid over-current
tripping in the normal operating load range, find the ROCSET
resistor from the equation above with:
1. The maximum rDS(ON) at the highest junction
temperature.
7
FN4855.3
July 13, 2005
HIP6004D
TABLE 1. OUTPUT VOLTAGE PROGRAM
PIN NAME
PIN NAME
VID4
VID3
VID2
VID1
VID0
NOMINAL OUTPUT
VOLTAGE DACOUT
VID4
VID3
VID2
VID1
VID0
NOMINAL OUTPUT
VOLTAGE DACOUT
1
1
1
1
1
0
0
1
1
1
1
1.475
1
1
1
1
0
1.100
0
1
1
1
0
1.500
1
1
1
0
1
1.125
0
1
1
0
1
1.525
1
1
1
0
0
1.150
0
1
1
0
0
1.550
1
1
0
1
1
1.175
0
1
0
1
1
1.575
1
1
0
1
0
1.200
0
1
0
1
0
1.600
1
1
0
0
1
1.225
0
1
0
0
1
1.625
1
1
0
0
0
1.250
0
1
0
0
0
1.650
1
0
1
1
1
1.275
0
0
1
1
1
1.675
1
0
1
1
0
1.300
0
0
1
1
0
1.700
1
0
1
0
1
1.325
0
0
1
0
1
1.725
1
0
1
0
0
1.350
0
0
1
0
0
1.750
1
0
0
1
1
1.375
0
0
0
1
1
1.775
1
0
0
1
0
1.400
0
0
0
1
0
1.800
1
0
0
0
1
1.425
0
0
0
0
1
1.825
1
0
0
0
0
1.450
0
0
0
0
0
1.850
NOTE: 0 = connected to GND or VSS , 1 = connected to VDD through pull-up resistors.
current paths on the SS pin and locate the capacitor, CSS
close to the SS pin because the internal current source is
only 10μA. Provide local VCC decoupling between VCC and
GND pins. Locate the capacitor, CBOOT as close as
practical to the BOOT and PHASE pins.
VIN
HIP6004D
Q1
LO
PHASE
VOUT
LGATE
D2
CIN
CO
LOAD
BOOT
Q2
CBOOT
PGND
HIP6004D
RETURN
FIGURE 5. PRINTED CIRCUIT BOARD POWER AND
GROUND PLANES OR ISLANDS
D1
LO
VOUT
PHASE
VCC
SS
+VIN
Q1
+12V
Q2
LOAD
UGATE
CO
CVCC
CSS
GND
Figure 5 shows the critical power components of the converter.
To minimize the voltage overshoot the interconnecting wires
indicated by heavy lines should be part of ground or power
plane in a printed circuit board. The components shown in
Figure 5 should be located as close together as possible.
Please note that the capacitors CIN and CO each represent
numerous physical capacitors. Locate the HIP6004D within 3
inches of the MOSFETs, Q1 and Q2 . The circuit traces for the
MOSFETs’ gate and source connections from the HIP6004D
must be sized to handle up to 1A peak current.
Figure 6 shows the circuit traces that require additional
layout consideration. Use single point and ground plane
construction for the circuits shown. Minimize any leakage
8
FIGURE 6. PRINTED CIRCUIT BOARD SMALL SIGNAL
LAYOUT GUIDELINES
Feedback Compensation
Figure 7 highlights the voltage-mode control loop for a
synchronous-rectified buck converter. The output voltage
(VOUT) is regulated to the Reference voltage level. The
error amplifier (Error Amp) output (VE/A) is compared with
the oscillator (OSC) triangular wave to provide a pulsewidth modulated (PWM) wave with an amplitude of VIN at
the PHASE node.
FN4855.3
July 13, 2005
HIP6004D
-
DRIVER
+
VOUT
PHASE
CO
ESR
(PARASITIC)
ZFB
VE/A
7. Estimate Phase Margin - Repeat if Necessary.
LO
-
ZIN
+
REFERENCE
ERROR
AMP
DETAILED COMPENSATION COMPONENTS
ZFB
C2
C1
VOUT
ZIN
C3
R2
R3
R1
COMP
FB
+
HIP6004D
DACOUT
FIGURE 7. VOLTAGE-MODE BUCK CONVERTER
COMPENSATION DESIGN
The PWM wave is smoothed by the output filter (LO and CO).
The modulator transfer function is the small-signal transfer
function of VOUT/VE/A . This function is dominated by a DC
Gain and the output filter (LO and CO), with a double pole
break frequency at FLC and a zero at FESR . The DC Gain of
the modulator is simply the input voltage (VIN) divided by the
peak-to-peak oscillator voltage ΔVOSC .
Modulator Break Frequency Equations
1
F LC = ------------------------------------------2π x L O x C O
1
F ESR = -------------------------------------------2π x ESR x C O
The compensation network consists of the error amplifier
(internal to the HIP6004D) and the impedance networks ZIN
and ZFB. The goal of the compensation network is to provide
a closed loop transfer function with the highest 0dB crossing
frequency (f0dB) and adequate phase margin. Phase margin
is the difference between the closed loop phase at f0dB and
180 degrees. The equations below relate the compensation
network’s poles, zeros and gain to the components (R1 , R2 ,
R3 , C1 , C2 , and C3) in Figure 7. Use these guidelines for
locating the poles and zeros of the compensation network:
1. Pick Gain (R2/R1) for desired converter bandwidth.
2. Place 1ST Zero Below Filter’s Double Pole (~75% FLC).
3. Place 2ND Zero at Filter’s Double Pole.
4. Place 1ST Pole at the ESR Zero.
5. Place 2ND Pole at Half the Switching Frequency.
9
Compensation Break Frequency Equations
1
F Z1 = -----------------------------------2π x R 2 x C 1
1
F P1 = --------------------------------------------------------⎛ C 1 x C 2⎞
2π x R 2 x ⎜ ----------------------⎟
⎝ C1 + C2 ⎠
1
F Z2 = ------------------------------------------------------2π x ( R 1 + R 3 ) x C 3
1
F P2 = -----------------------------------2π x R 3 x C 3
Figure 8 shows an asymptotic plot of the DC-DC converter’s
gain vs frequency. The actual Modulator Gain has a high gain
peak due to the high Q factor of the output filter and is not
shown in Figure 8. Using the above guidelines should give a
Compensation Gain similar to the curve plotted. The open
loop error amplifier gain bounds the compensation gain.
Check the compensation gain at FP2 with the capabilities of
the error amplifier. The Closed Loop Gain is constructed on
the log-log graph of Figure 8 by adding the Modulator Gain (in
dB) to the Compensation Gain (in dB). This is equivalent to
multiplying the modulator transfer function to the
compensation transfer function and plotting the gain.
The compensation gain uses external impedance networks
ZFB and ZIN to provide a stable, high bandwidth (BW)
overall loop. A stable control loop has a gain crossing with
-20dB/decade slope and a phase margin greater than 45
degrees. Include worst case component variations when
determining phase margin.
100
FZ1 FZ2
FP1
FP2
80
OPEN LOOP
ERROR AMP GAIN
60
GAIN (dB)
PWM
COMPARATOR
ΔVOSC
6. Check Gain against Error Amplifier’s Open-Loop Gain.
VIN
DRIVER
OSC
40
20
20LOG
(R2/R1)
0
20LOG
(VIN/ΔVOSC)
MODULATOR
GAIN
-20
COMPENSATION
GAIN
-40
-60
CLOSED LOOP
GAIN
FLC
10
100
1K
FESR
10K
100K
1M
10M
FREQUENCY (Hz)
FIGURE 8. ASYMPTOTIC BODE PLOT OF CONVERTER GAIN
Component Selection Guidelines
Output Capacitor Selection
An output capacitor is required to filter the output and supply
the load transient current. The filtering requirements are a
function of the switching frequency and the ripple current.
The load transient requirements are a function of the slew
rate (di/dt) and the magnitude of the transient load current.
These requirements are generally met with a mix of
capacitors and careful layout.
FN4855.3
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HIP6004D
Modern microprocessors produce transient load rates above
1A/ns. High frequency capacitors initially supply the transient
and slow the current load rate seen by the bulk capacitors.
The bulk filter capacitor values are generally determined by
the ESR (Effective Series Resistance) and voltage rating
requirements rather than actual capacitance requirements.
High frequency decoupling capacitors should be placed as
close to the power pins of the load as physically possible. Be
careful not to add inductance in the circuit board wiring that
could cancel the usefulness of these low inductance
components. Consult with the manufacturer of the load on
specific decoupling requirements.
Use only specialized low-ESR capacitors intended for
switching-regulator applications for the bulk capacitors. The
bulk capacitor’s ESR will determine the output ripple voltage
and the initial voltage drop after a high slew-rate transient. An
aluminum electrolytic capacitor’s ESR value is related to the
case size with lower ESR available in larger case sizes.
However, the Equivalent Series Inductance (ESL) of these
capacitors increases with case size and can reduce the
usefulness of the capacitor to high slew-rate transient loading.
Unfortunately, ESL is not a specified parameter. Work with
your capacitor supplier and measure the capacitor’s
impedance with frequency to select a suitable component. In
most cases, multiple electrolytic capacitors of small case size
perform better than a single large case capacitor.
Output Inductor Selection
The output inductor is selected to meet the output voltage
ripple requirements and minimize the converter’s response
time to the load transient. The inductor value determines the
converter’s ripple current and the ripple voltage is a function
of the ripple current. The ripple voltage and current are
approximated by the following equations:
DI =
VIN - VOUT
Fs x L
x
VOUT
VIN
DVOUT = DI x ESR
Increasing the value of inductance reduces the ripple current
and voltage. However, the large inductance values reduce
the converter’s response time to a load transient.
One of the parameters limiting the converter’s response to
a load transient is the time required to change the inductor
current. Given a sufficiently fast control loop design, the
HIP6004D will provide either 0% or 100% duty cycle in
response to a load transient. The response time is the time
required to slew the inductor current from an initial current
value to the transient current level. During this interval the
difference between the inductor current and the transient
current level must be supplied by the output capacitor.
Minimizing the response time can minimize the output
capacitance required.
The response time to a transient is different for the
application of load and the removal of load. The following
10
equations give the approximate response time interval for
application and removal of a transient load:
tRISE =
L x ITRAN
VIN - VOUT
tFALL =
L x ITRAN
VOUT
where: ITRAN is the transient load current step, tRISE is the
response time to the application of load, and tFALL is the
response time to the removal of load. With a +5V input
source, the worst case response time can be either at the
application or removal of load and dependent upon the
DACOUT setting. Be sure to check both of these equations
at the minimum and maximum output levels for the worst
case response time. With a +12V input, and output voltage
level equal to DACOUT, tFALL is the longest response time.
Input Capacitor Selection
Use a mix of input bypass capacitors to control the voltage
overshoot across the MOSFETs. Use small ceramic
capacitors for high frequency decoupling and bulk capacitors
to supply the current needed each time Q1 turns on. Place the
small ceramic capacitors physically close to the MOSFETs
and between the drain of Q1 and the source of Q2 .
The important parameters for the bulk input capacitor are the
voltage rating and the RMS current rating. For reliable
operation, select the bulk capacitor with voltage and current
ratings above the maximum input voltage and largest RMS
current required by the circuit. The capacitor voltage rating
should be at least 1.25 times greater than the maximum
input voltage and a voltage rating of 1.5 times is a
conservative guideline. The RMS current rating requirement
for the input capacitor of a buck regulator is approximately
1/2 the DC load current.
For a through hole design, several electrolytic capacitors may
be needed. For surface mount designs, solid tantalum
capacitors can be used, but caution must be exercised with
regard to the capacitor surge current rating. These capacitors
must be capable of handling the surge-current at power-up.
Some capacitor series available from reputable manufacturers
are surge current tested.
MOSFET Selection/Considerations
The HIP6004D requires 2 N-Channel power MOSFETs. These
should be selected based upon rDS(ON) , gate supply
requirements, and thermal management requirements.
In high-current applications, the MOSFET power dissipation,
package selection and heatsink are the dominant design
factors. The power dissipation includes two loss components;
conduction loss and switching loss. The conduction losses are
the largest component of power dissipation for both the upper
and the lower MOSFETs. These losses are distributed between
the two MOSFETs according to duty factor (see the equations
below). Only the upper MOSFET has switching losses, since
the Schottky rectifier clamps the switching node before the
synchronous rectifier turns on. These equations assume linear
FN4855.3
July 13, 2005
HIP6004D
voltage-current transitions and do not adequately model power
loss due the reverse-recovery of the lower MOSFET’s body
diode. The gate-charge losses are dissipated by the HIP6004D
and don't heat the MOSFETs. However, large gate-charge
increases the switching interval, tSW which increases the upper
MOSFET switching losses. Ensure that both MOSFETs are
within their maximum junction temperature at high ambient
temperature by calculating the temperature rise according to
package thermal-resistance specifications. A separate heatsink
may be necessary depending upon MOSFET power, package
type, ambient temperature and air flow.
PUPPER = Io2 x rDS(ON) x D +
Figure 10 shows the upper gate drive supplied by a direct
connection to VCC . This option should only be used in
converter systems where the main input voltage is +5VDC or
less. The peak upper gate-to-source voltage is approximately
VCC less the input supply. For +5V main power and +12VDC
for the bias, the gate-to-source voltage of Q1 is 7V. A logiclevel MOSFET is a good choice for Q1 and a logic-level
MOSFET can be used for Q2 if its absolute gate-to-source
voltage rating exceeds the maximum voltage applied to VCC .
+12V
1 Io x V x t
IN SW x FS
2
+5V OR LESS
PLOWER = Io2 x rDS(ON) x (1 - D)
VCC
BOOT
Where: D is the duty cycle = VOUT / VIN ,
tSW is the switch ON time, and
HIP6004D
UGATE
FS is the switching frequency.
PHASE
Standard-gate MOSFETs are normally recommended for
use with the HIP6004D. However, logic-level gate MOSFETs
can be used under special circumstances. The input voltage,
upper gate drive level, and the MOSFET’s absolute gate-tosource voltage rating determine whether logic-level
MOSFETs are appropriate.
Figure 9 shows the upper gate drive (BOOT pin) supplied by a
bootstrap circuit from VCC. The boot capacitor, CBOOT
develops a floating supply voltage referenced to the PHASE
pin. This supply is refreshed each cycle to a voltage of VCC
less the boot diode drop (VD) when the lower MOSFET, Q2
turns on. Logic-level MOSFETs can only be used if the
MOSFET’s absolute gate-to-source voltage rating exceeds
the maximum voltage applied to VCC.
+12V
VCC
DBOOT
+5V OR +12V
+ VD -
Q1
-
+
LGATE
PGND
GND
NOTE:
VG-S ≈ VCC -5V
Q2
D2
NOTE:
VG-S ≈ VCC
FIGURE 10. UPPER GATE DRIVE - DIRECT VCC DRIVE OPTION
Schottky Selection
Rectifier D2 is a clamp that catches the negative inductor
swing during the dead time between turning off the lower
MOSFET and turning on the upper MOSFET. The diode
must be a Schottky type to prevent the lossy parasitic
MOSFET body diode from conducting. It is acceptable to
omit the diode and let the body diode of the lower MOSFET
clamp the negative inductor swing, but efficiency will drop
one or two percent as a result. The diode’s rated reverse
breakdown voltage must be greater than the maximum
input voltage.
BOOT
CBOOT
HIP6004D
UGATE
Q1
PHASE
-
+
NOTE:
VG-S ≈ VCC -VD
Q2
LGATE
PGND
D2
NOTE:
VG-S ≈ VCC
GND
FIGURE 9. UPPER GATE DRIVE - BOOTSTRAP OPTION
11
FN4855.3
July 13, 2005
HIP6004D
HIP6004D DC-DC Converter Application Circuit
description, can be found in Application Note AN9672.
Although the Application Note details the HIP6004, the same
evaluation platform can be used to evaluate the HIP6004D.
Figure 11 shows an application circuit of a DC-DC Converter
for a microprocessor. Detailed information on the circuit,
including a complete Bill-of-Materials and circuit board
VIN =
+5V
OR
+12V
L1 - 1μH
F1
2 x 1μF
2N6394
CIN
5x 1000μF
+12V
2K
D1
0.1μF
1000pF
VCC
18
2 OCSET
MONITOR
AND
PROTECTION
SS 3
0.1μF
OVP
19
12 PGOOD
15 BOOT
VSEN 1
RT 20
VID0
VID1
VID2
VID3
VID4
FB
4
5
6
7
8
1K
0.1μF
OSC
14 UGATE
Q1
L2
3μH
13 PHASE
HIP6004D
D/A
-
10
-
Q2
16 PGND
9
2.2nF
17 LGATE
+
+
COMP
+VOUT
D2
COUT
9x 1000μF
11
GND
20K
8.2nF
0.1μF
1.33K
15
Component Selection Notes:
COUT - Each 1000µF 6.3W VDC, Sanyo MV-GX or Equivalent.
CIN - Each 330µF 25W VDC, Sanyo MV-GX or Equivalent.
L2 - Core: Micrometals T50-52B; Winding: 10 Turns of 16AWG.
L1 - Core: Micrometals T50-52; Winding: 5 Turns of 18AWG.
D1 - 1N4148 or Equivalent.
D2 - 3A, 40V Schottky, Motorola MBR340 or Equivalent.
Q1 , Q2 - Intersil MOSFET; RFP70N03.
FIGURE 11. MICROPROCESSOR DC-DC CONVERTER
12
FN4855.3
July 13, 2005
HIP6004D
Small Outline Plastic Packages (SOIC)
M20.3 (JEDEC MS-013-AC ISSUE C)
20 LEAD WIDE BODY SMALL OUTLINE PLASTIC PACKAGE
N
INDEX
AREA
H
0.25(0.010) M
B M
INCHES
E
MILLIMETERS
SYMBOL
MIN
MAX
MIN
MAX
NOTES
A
0.0926
0.1043
2.35
2.65
-
A1
0.0040
0.0118
0.10
0.30
-
B
0.014
0.019
0.35
0.49
9
C
0.0091
0.0125
0.23
0.32
-
D
0.4961
0.5118
12.60
13.00
3
E
0.2914
0.2992
7.40
7.60
4
-B1
2
3
L
SEATING PLANE
-A-
A
D
h x 45°
-C-
α
e
A1
B
0.25(0.010) M
e
C
0.10(0.004)
C A M
B S
1.27 BSC
-
H
0.394
0.419
10.00
10.65
-
h
0.010
0.029
0.25
0.75
5
L
0.016
0.050
0.40
1.27
6
N
α
NOTES:
0.050 BSC
20
0°
20
8°
0°
7
8°
Rev. 2 6/05
1. Symbols are defined in the “MO Series Symbol List” in Section
2.2 of Publication Number 95.
2. Dimensioning and tolerancing per ANSI Y14.5M-1982.
3. Dimension “D” does not include mold flash, protrusions or gate
burrs. Mold flash, protrusion and gate burrs shall not exceed
0.15mm (0.006 inch) per side.
4. Dimension “E” does not include interlead flash or protrusions.
Interlead flash and protrusions shall not exceed 0.25mm (0.010
inch) per side.
5. The chamfer on the body is optional. If it is not present, a visual
index feature must be located within the crosshatched area.
6. “L” is the length of terminal for soldering to a substrate.
7. “N” is the number of terminal positions.
8. Terminal numbers are shown for reference only.
9. The lead width “B”, as measured 0.36mm (0.014 inch) or greater
above the seating plane, shall not exceed a maximum value of
0.61mm (0.024 inch)
10. Controlling dimension: MILLIMETER. Converted inch
dimensions are not necessarily exact.
13
FN4855.3
July 13, 2005
HIP6004D
Quad Flat No-Lead Plastic Package (QFN)
Micro Lead Frame Plastic Package (MLFP)
L20.5x5
20 LEAD QUAD FLAT NO-LEAD PLASTIC PACKAGE
MILLIMETERS
SYMBOL
MIN
NOMINAL
MAX
NOTES
A
0.80
0.90
1.00
-
A1
-
0.02
0.05
-
A2
-
0.65
1.00
9
0.38
5, 8
A3
b
0.20 REF
0.23
0.30
9
D
5.00 BSC
-
D1
4.75 BSC
9
D2
2.95
E
E1
E2
3.10
3.25
7, 8
5.00 BSC
-
4.75 BSC
2.95
e
3.10
9
3.25
7, 8
0.65 BSC
-
k
0.20
-
-
-
L
0.35
0.60
0.75
8
N
20
2
Nd
5
3
Ne
5
3
P
-
-
0.60
9
θ
-
-
12
9
Rev. 4 11/04
NOTES:
1. Dimensioning and tolerancing conform to ASME Y14.5-1994.
2. N is the number of terminals.
3. Nd and Ne refer to the number of terminals on each D and E.
4. All dimensions are in millimeters. Angles are in degrees.
5. Dimension b applies to the metallized terminal and is measured
between 0.15mm and 0.30mm from the terminal tip.
6. The configuration of the pin #1 identifier is optional, but must be
located within the zone indicated. The pin #1 identifier may be
either a mold or mark feature.
7. Dimensions D2 and E2 are for the exposed pads which provide
improved electrical and thermal performance.
8. Nominal dimensions are provided to assist with PCB Land Pattern
Design efforts, see Intersil Technical Brief TB389.
9. Features and dimensions A2, A3, D1, E1, P & θ are present when
Anvil singulation method is used and not present for saw
singulation.
10. Compliant to JEDEC MO-220VHHC Issue I except for the "b"
dimension.
All Intersil U.S. products are manufactured, assembled and tested utilizing ISO9000 quality systems.
Intersil Corporation’s quality certifications can be viewed at www.intersil.com/design/quality
Intersil products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design, software and/or specifications at any time without
notice. Accordingly, the reader is cautioned to verify that data sheets are current before placing orders. Information furnished by Intersil is believed to be accurate and
reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result
from its use. No license is granted by implication or otherwise under any patent or patent rights of Intersil or its subsidiaries.
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14
FN4855.3
July 13, 2005