LT1249 Power Factor Controller U FEATURES ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ DESCRIPTIO The 8-pin LT ®1249 provides active power factor correction for universal offline power systems with very few external parts. By using fixed high frequency PWM current averaging without the need for slope compensation, the LT1249 achieves far lower line current distortion, with a smaller magnetic element than systems that use either peak current detection or zero current switching approach, in both continuous and discontinuous modes of operation. Standard 8-Pin Packages High Power Factor Over Wide Load Range with Line Current Averaging International Operation Without Switches Instantaneous Overvoltage Protection Minimal Line Current Dead Zone Typical 250µA Start-Up Supply Current Rejects Line Switching Noise Synchronization Capability Low Quiescent Current: 9mA Fast 1.5A Peak Current Gate Driver The LT1249 uses a multiplier containing a square gain function from the voltage amplifier to reduce the AC gain at light output load and thus maintains low line current distortion and high system stability. The LT1249 also provides filtering capability to reject line switching noise which can cause instability when fed into the multiplier. Line current dead zone is minimized with low bias voltage at the current input to the multiplier. U APPLICATIO S ■ ■ Universal Power Factor Corrected Power Supplies Preregulators up to 1500W The LT1249 provides many protection features including peak current limiting and overvoltage protection. The switching frequency is internally set at 100kHz. While the LT1249 simplifies PFC design with minimal parts count, the LT1248 provides flexibilities in switching frequency, overvoltage and current limit. , LTC and LT are registered trademarks of Linear Technology Corporation. W BLOCK DIAGRA VSENSE 7.5V 6 IAC 4 VAOUT MOUT CAOUT GND 5 3 2 1 RMOUT 4k + EA – IA IB MULTIPLIER 32k IM I 2I IM = A B 2 200µA VCC + 16V/10V – 7 RUN + – CA R – 15µA 1V 250µA MAX VCC 7.5V VREF + + gm = 1/3k RUN Q GTDR S 8 0.7V + M1 – 44µA 22µA + 4k SYNC OSC – 20µA 16V 35pF 1249 BD 1 LT1249 W U U W W W Supply Voltage ....................................................... 27V GTDR Current Continuous ..................................... 0.5A GTDR Output Energy (Per Cycle) ............................. 5µJ IAC Input Current ................................................. 20mA VSENSE Input Voltage ............................................ VMAX MOUT Input Current.............................................. ± 5mA Operating Junction Temperature Range LT1249C ................................................ 0°C to 100°C LT1249I ........................................... – 40°C to 125°C Thermal Resistance (Junction-to-Ambient) N8 Package ................................................ 100°C/W S8 Package ................................................. 120°C/W Storage Temperature Range ..................–65°C to 150°C Lead Temperature (Soldering, 10 sec)................. 300°C U ABSOLUTE MAXIMUM RATINGS PACKAGE/ORDER INFORMATION ORDER PART NUMBER TOP VIEW GND 1 8 GTDR CAOUT 2 7 VCC MOUT 3 6 VSENSE IAC 4 5 VAOUT LT1249CN8 LT1249IN8 LT1249CS8 LT1249IS8 N8 PACKAGE 8-LEAD PDIP S8 PART MARKING S8 PACKAGE 8-LEAD PLASTIC SO TJMAX = 125°C, θJA = 100°C/W (N8) TJMAX = 125°C, θJA = 120°C/W (S8) 1249 1249I Consult factory for Military grade parts. ELECTRICAL CHARACTERISTICS The ● denotes specifications which apply over the operating temperature range, otherwise specifications are at TA = 25°C. Maximum operating voltage (VMAX) = 25V, VCC = 18V, IAC = 100µA, CAOUT = 3.5V, VAOUT = 5V, no load on any outputs, unless otherwise noted. PARAMETER Overall Supply Current (VCC in Undervoltage Lockout) Supply Current, On VCC Turn-On Threshold VCC Turn-Off Threshold Voltage Amplifier VSENSE Bias Current Voltage Amp Gain Voltage Amp Unity-Gain Bandwidth Voltage Amp Output High Voltage Amp Output Low Voltage Amp Source Current Voltage Amp Sink Current Threshold Voltage Amp Sink Current Hysteresis Current Amplifier Current Amp Offset Voltage Current Amp Transconductance Current Amp Voltage Gain Current Amp Source Current Current Amp Sink Current Current Amp Output High Current Amp Output Low 2 CONDITIONS VCC = Lockout Voltage – 0.2V 11.5V ≤ VCC ≤ VMAX, CAOUT = 1V TYP MAX UNITS 15.5 9.5 0.25 9 16.5 10.5 0.45 12 17.5 11.5 mA mA V V –25 100 1.5 12 0.1 260 44 22.5 –250 nA dB MHz V V µA µA µA ±2 320 1000 145 95 8.1 1.2 ±15 550 ● ● ● ● VSENSE = 0V to 7V MIN ● 70 0 ≤ Source Current ≤ 50µA 0 ≤ Sink Current ≤ 5µA ● ● Linear Operation, 2V < VAOUT < 10V 2V < VAOUT < 10V 10 ● ● ● 130 33 14 ● ∆ICAOUT = ±40µA 2.5V ≤ VCAOUT ≤ 7.5V VMOUT = 1V, IM = 0µA VMOUT = – 0.3V, IM = 0µA ● 150 500 100 67 7.4 0.4 450 57 30 220 125 2 mV µmho V/V µA µA V V LT1249 ELECTRICAL CHARACTERISTICS The ● denotes specifications which apply over the operating temperature range, otherwise specifications are at TA = 25°C. Maximum operating voltage (VMAX) = 25V, VCC = 18V, IAC = 100µA, CAOUT = 3.5V, VAOUT = 5V, no load on any outputs, unless otherwise noted. PARAMETER Reference Reference Output Voltage Reference Output Voltage Worst Case Reference Output Voltage Line Regulation Multiplier Multiplier Output Current Multiplier Output Current Offset Multiplier Max Output Current (IM(MAX)) Multiplier Max Output Voltage (IM(MAX) • RMOUT) Multiplier Gain Constant (Note 3) IAC Input Resistance Oscillator Oscillator Frequency Control Pin (CAOUT) Threshold Synchronization Frequency Range Gate Driver Max GTDR Output Voltage GTDR Output High GTDR Output Low (Device Unpowered) GTDR Output Low (Device Active) Peak GTDR Current GTDR Rise and Fall Time GTDR Max Duty Cycle CONDITIONS TA = 25°C, Measured at VSENSE Pin All Line, Temperature VLOCKOUT < VCC < VMAX ● ● IAC = 100µA, VAOUT = 5V RAC = 1M from IAC to GND IAC = 450µA, VAOUT = 7V (Note 2) IAC = 450µA, VAOUT = 7V (Note 2) MIN TYP MAX UNITS 7.39 7.32 – 20 7.5 7.5 5 7.6 7.68 20 V V mV – 375 – 1.25 35 – 0.05 – 250 – 1.1 0.035 32 ● ● ● IAC from 50µA to 1mA 15 ● Duty Cycle = 0 Synchronizing Pulse Low ≤ 0.35V on CAOUT ● 0mA Load, 18V < VCC < VMAX (Note 4) – 200mA Load, 11.5V ≤ VCC ≤ 15V VCC = 0V, 50mA Load (Sinking) 200mA Load (Sinking) 10nF from GTDR to GND 1nF from GTDR to GND ● ● ● 75 1.3 127 100 1.8 125 2.3 160 kHz V kHz 12 VCC – 3.0 15 17.5 0.9 0.5 1.5 25 96 1.5 1 V V V V A ns % ● ● 90 Note 1: Absolute Maximum Ratings are those values beyond which the life of a device may be impaired. Note 2: Current amplifier is in linear mode with 0V input common mode. 50 µA µA µA V –2 V kΩ – 0.5 – 150 – 0.96 Note 3: Multiplier Gain Constant: K = IM IAC (VAOUT – 1.5)2 Note 4: Maximum GTDR output voltage is internally clamped for higher VCC voltages. U W TYPICAL PERFORMANCE CHARACTERISTICS Transconductance of Current Amplifier Voltage Amplifier Open-Loop Gain and Phase 0 80 400 40 –60 20 –80 PHASE 0 –20 –100 10 100 1k 10k 100k FREQUENCY (Hz) 1M –120 10M 0 300 –20 250 –40 200 –60 150 –80 100 –100 50 –120 0 1k 10k 1M 100k FREQUENCY (Hz) PHASE (DEG) –40 PHASE (DEG) GAIN (dB) GAIN 60 20 θ gm 350 –20 TRANSCONDUCTANCE (µmho) 100 –140 10M 1249 G01 1249 G02 3 LT1249 U W TYPICAL PERFORMANCE CHARACTERISTICS Reference Voltage vs Temperature Multiplier Current 7.536 300 VAOUT = 5V VAOUT = 6.5V 7.512 VAOUT = 6V 7.500 VAOUT = 5.5V IM (µA) REFERENCE VOLTAGE (V) 7.524 7.488 7.476 VAOUT = 4.5V 150 VAOUT = 4V 7.464 VAOUT = 3.5V 7.452 VAOUT = 3V 7.440 7.428 –75 –50 –25 0 25 50 75 100 125 150 JUNCTION TEMPERATURE (°C) 0 0 VAOUT = 2.5V VAOUT = 2V 500 250 IAC (µA) 1249 G04 1249 G03 18.5 10 7 18.0 TJ = 25°C 17.5 0.9 17.0 0.8 TJ = 125°C 6 5 4 3 VCC = 18V 16.5 16.0 15.5 TJ = 125°C 15.0 TJ = 25°C 14.5 2 14.0 1 13.5 0 1.0 GTDR VOLTAGE (V) 8 1.1 TJ = –55°C GTDR VOLTAGE (V) 9 SUPPLY CURRENT (mA) GTDR Sink Current GTDR Source Current Supply Current vs Supply Voltage 13.0 10 12 14 16 18 20 22 24 26 28 30 SUPPLY VOLTAGE (V) 0.7 0.6 0.5 TA = –55°C 0.4 0.3 TJ = –55°C 0.2 TA = 25°C 0.1 –120 –180 –240 – 60 SOURCE CURRENT (mA) 0 0 –300 TA = 125°C 0 60 120 180 240 SINK CURRENT (mA) 1249 G06 1249 G05 1249 G07 Start-Up Supply Current vs Supply Voltage GTDR Rise and Fall Time 400 300 Switching Frequency 550 140 500 130 450 FALL TIME 200 RISE TIME 100 0 10 20 30 40 LOAD CAPACITANCE (nF) 300 –55°C 25°C 250 200 125°C 120 110 100 90 150 80 50 50 1249 G08 4 350 100 NOTE: GTDR SLEWS BETWEEN 1V AND 16V 0 400 FREQUENCY (kHz) SUPPLY CURRENT (µA) TIME (ns) 300 0 0 2 4 6 8 10 12 14 16 18 20 SUPPLY VOLTAGE (V) 1249 G09 70 25 50 75 –75 –50 –25 0 TEMPERATURE (°C) 100 125 1249 G10 LT1249 U W TYPICAL PERFORMANCE CHARACTERISTICS Transconductance of Current Amplifier Over Temperature MOUT Pin Characteristics 1.2 0.9 1.0 0.8 0.8 0.7 0.6 0.6 0.5 0.4 0.3 0.2 400 125°C 25°C –50°C 350 TRANSCONDUCTANCE (µmho) 1.0 MOUT CURRENT (mA) SYNCHRONIZATION THRESHOLD (V) Synchronization Threshold at CAOUT 0.4 0.2 0 –0.2 –0.4 –0.6 0.1 –25 0 25 50 75 TEMPERATURE (°C) –1.0 125 100 –2.4 0 –1.2 1.2 MOUT VOLTAGE (V) 60 100 0 –50 –25 2.4 0 25 50 75 100 125 TEMPERATURE (°C) 1249 G13 Maximum Duty Cycle 100 –1.30 99 –1.25 50 98 40 30 DOWN THRESHOLD 20 –1.20 DUTY CYCLE (%) IM(MAX) × RMOUT (V) UP THRESHOLD CURRENT (µA) 150 Maximum Multiplier Output Voltage (IM(MAX) • RMOUT) Voltage Amp Sink Current Limits (Threshold) –1.15 –1.10 –1.05 –1.00 10 100 125 NOTE: THESE SINK CURRENT THRESHOLDS ARE FOR OVERVOLTAGE PROTECTION FUNCTION. –0.90 –75 –50 97 96 95 94 93 92 –0.95 –25 0 25 50 75 TEMPERATURE (°C) 200 1249 G12 1249 G11 0 –75 –50 250 50 –0.8 0 –50 300 91 –25 0 25 50 75 100 125 TEMPERATURE (°C) 1249 G15 90 –50 –25 0 25 50 75 TEMPERATURE (°C) 100 125 1249 G16 1249 G14 U U U PIN FUNCTIONS GND (Pin 1): Ground. CAOUT (Pin 2): This is the output of the current amplifier that senses and forces the line current to follow the reference signal that comes from the multiplier by commanding the pulse width modulator. When CAOUT is low, the modulator has zero duty cycle. MOUT (Pin 3): The multiplier current goes out of this pin through the 4k resistor RMOUT. The voltage developed across RMOUT is the reference voltage of the current loop and it is limited to 1.1V. The noninverting input of the current amplifier is also tied to RMOUT. In operation, MOUT is normally at negative potential and only AC signals appear at the noninverting input of the current amplifier. IAC (Pin 4): This is the AC line voltage sensing input to the multiplier. It is a current input that is biased at 2V to minimize the crossover dead zone caused by low line voltage. A 32k resistor is in series with the current input, so that a small external capacitor can be used to filter out the switching noise from the high impedance lines. VAOUT (Pin 5): This is the output of the voltage error amplifier. The output is clamped at 12V. When the output goes below 1.5V, the multiplier output current is zero. 5 LT1249 U U U PIN FUNCTIONS VSENSE (Pin 6): This is the inverting input to the voltage amplifier. capacitor in parallel with a low ESR electrolytic capacitor, 56µF or higher is required in close proximity to IC GND. VCC (Pin 7): This is the supply of the chip. The LT1249 has a very fast gate driver required to fast charge high power MOSFET gate capacitance. High current spikes occur during charging. For good supply bypass, a 0.1µF ceramic GTDR (Pin 8): The MOSFET gate driver is a 1.5A fast totem pole output. It is clamped at 15V. Capacitive loads like MOSFET gates may cause overshoot. A gate series resistor of at least 5Ω will prevent the overshoot. U W U U APPLICATIONS INFORMATION Error Amplifier Multiplier The error amplifier has a 100dB DC gain and 1.5MHz unitygain frequency. It is internally clamped at 12V. The noninverting input is tied to the 7.5V reference. The multiplier is a current multiplier with high noise immunity in a high power switching environment. The current gain is: The multiplier output current IM flows out of the MOUT pin through the 4k resistor RMOUT and develops the reference signal to the current loop that is controlled by the current amplifier. Current gain is the ratio of RMOUT to line current sense resistor. The current amplifier is a transconductance amplifier. Typical gm is 320µmho and gain is 60dB with no load. The inverting input is internally tied to GND. The noninverting input is tied to the multiplier output. The output is internally clamped at 8V. Output resistance is about 4M; DC loading should be avoided because it will lower the gain and introduce offset voltage at the inputs which becomes a false reference signal to the current loop and can distort line current. Note that in the current averaging operation, high gain at twice the line frequency is necessary to minimize line current distortion. Because CAOUT may need to swing 5V over one line cycle at high line condition, 11mV will be present at the inputs of the current amplifier if gain is rolled off to 450 at 120Hz (1nF in series with 10k at CAOUT). At light load, when (IM)(RMOUT) can be less than 100mV, lower gain will distort the current loop reference signal and line current. If signal gain at the 100kHz switching frequency is too high, the system behaves more like a current mode system and can cause subharmonic oscillation. Therefore, the current amplifier should be compensated to have a gain of less than 15 at 100kHz and more than 300 at 120Hz. 6 IM = (IAC)(IEA2)/(200µA)2, and IEA = (VAOUT – 1.5V)/25k With a square function, because of the lower gain at light power load, system stability is maintained and line current distortion caused by the AC ripple fed back to the error amplifier is minimized. Note that switching ripple on the high impedance lines could get into the multiplier from the IAC pin and cause instability. The LT1249 provides an internal 25k resistor in series with the low impedance multiplier current input so that only a capacitor from the IAC pin to GND is needed to filter out the noise. Maximum multiplier output current is limited to 250µA. Figure 1 shows the multiplier transfer curves. 300 VAOUT = 5V VAOUT = 6.5V VAOUT = 6V VAOUT = 5.5V IM (µA) Current Amplifier VAOUT = 4.5V 150 VAOUT = 4V VAOUT = 3.5V VAOUT = 3V 0 0 250 IAC (µA) VAOUT = 2.5V VAOUT = 2V 500 1249 G04 Figure 1. Multiplier Current IM vs IAC and VAOUT LT1249 U W U U APPLICATIONS INFORMATION Line Current Limiting Maximum voltage across RMOUT is internally limited to 1.1V. Therefore, line current limit is 1.1V divided by the sense resistor RS. With a 0.2Ω sense resistor RS line current limit is 5.5A. As a general rule, RS is chosen according RS = With ts = 30ns, fs = 130kHz, VC = 3V and R2 = 10k, offset voltage shift is ≈5mV. Note that this offset voltage will add slight distortion to line current at light load. CAOUT 1N5712 R2 10k K (1.414)POUT(MAX) Synchronization The LT1249 can be externally synchronized in a frequency range of 127kHz to 160kHz. Figure 2 shows the synchronizing circuit. Synchronizing occurs when CAOUT pin is pulled below 0.5V with an external transistor and a Schottky diode. The Schottky diode and the 10k pull-up resistor are necessary for the required fast slewing back up to the normal operating voltage on CAOUT after the transistor is turned off. Positive slewing on CAOUT should be faster than the oscillator ramp rate of 0.5V/µs. The width of the synchronizing pulse should be under 60ns. The synchronizing pulses introduce an offset voltage on the current amplifier inputs, according to: V − 0.5 (ts)(fs)IC + C R2 ∆VOS = gm ts = pulse width fs = pulse frequency IC = CAOUT source current (≈ 150µA) VC = CAOUT operating voltage (1.8V to 6.8V) R2 = resistor for the midfrequency “zero” in the current loop gm = current amplifier transconductance (≈ 320µmho) R1 10k 80pF (IM(MAX) )(RMOUT )(VLINE(MIN)) where POUT(MAX) is the maximum power output and K is usually between 1.1 and 1.3 depending on efficiency and resistor tolerance. When the output is overloaded and line current reaches limit, output voltage VOUT will drop to keep line current constant. System stability is still maintained by the current loop which is controlled by the current amplifier. Further load current increase results in further VOUT drop and clipping of the line current, which degrades power factor. VCC 5V 0V 2N2369 2k 1nF 1249 F02 Figure 2. Synchronizing the LT1249 Overvoltage Protection In Figure 3, R1 and R2 set the regulator output DC level: VOUT = VREF[(R1 + R2)/R2]. With R1 = 1M and R2 = 20k, VOUT is 382V. Because of the slow loop response necessary for power factor correction, output overshoot can occur with sudden load removal or reduction. To protect the power components and output load, the LT1249 voltage error amplifier senses the output voltage and quickly shuts off the current switch when overvoltage occurs. When overshoot occurs on VOUT, the overcurrent from R1 will go through VAOUT because amplifier feedback keeps VSENSE locked at 7.5V. When this overcurrent reaches 44µA amplifier sinking limit, the amplifier loses feedback and its output snaps low to turn the multiplier off. Overvoltage trip level: ∆VOUT = (44µA)(R1) 0.047µF VOUT C1 0.47µF R1 1M R3 330k VSENSE VAOUT – EA + R2 20k VREF 7.5V 44µA MULTIPLIER 22µA LT1249 1249 F03 Figure 3. Overvoltage Protection 7 LT1249 U W U U APPLICATIONS INFORMATION The Figure 3 circuit therefore has 382V on VOUT, and an overvoltage level = (VOUT + 44V), or 426V. With a 22µA hysteresis, VOUT then has to drop 22V to 404V before feedback recovers and the switch turns back on. MOUT is a high impedance current output. In the current loop, offset line current is determined by multiplier offset current and input offset voltage of the current amplifier. A negative 4mV current amplifier VOS translates into 20mA line current and 5W input power for 250V line if 0.2Ω sense resistor is used. Under no load or when the load power is less than this offset input power, VOUT would slowly charge up to an overvoltage state because the overvoltage comparator can only reduce multiplier output current to zero. This does not guarantee zero output current if the current amplifier has offset. To regulate VOUT under this condition, the amplifier M1 (see Block Diagram), becomes active in the current loop when VAOUT goes down to 1V. The M1 can put out up to 15µA to the 4k resistor at the inverting input to cancel the current amplifier negative VOS and keep VOUT error to within 2V. Undervoltage Lockout The LT1249 turns on when VCC is higher than 16V and remains on until VCC falls below 10V, whereupon the chip enters the lockout state. In the lockout state, the LT1249 only draws 250µA, the oscillator is off, the VREF and the GTDR pins remain low to keep the power MOSFET off. Start-Up and Supply Voltage The LT1249 draws only 250µA before the chip starts at 16V on VCC. To trickle start, a 90k resistor from the power line to VCC supplies the trickle current and C4 holds the VCC up while switching starts (see Figure 4). Then the auxiliary winding takes over and supplies the operating current. Note that D3 and the large value C3, in both Figures 4 and 5, are only necessary for systems that have sudden large load variation down to minimum load and/or very light load conditions. Under these conditions, the loop may exhibit a start/restart mode because switching remains off long enough for C4 to discharge below 10V. The C3 will hold VCC up until switching resumes. For less severe load variations, D3 is replaced with a short and C3 is omitted. The turns ratio between the primary winding and the 8 LINE MAIN INDUCTOR NP NS R1 90k 1W D1 D3 VCC + D2 C1 2µF + + C3 390µF C2 2µF + C4 56µF ALL CAPACITORS ARE RATED 35V 1249 F04 Figure 4. Power Supply for LT1249 C2 1000pF 450V MAIN INDUCTOR LINE D2 R1 90k 1W D3 + D1 C3 390µF 35V 18V + VCC C4 56µF 35V 1249 F05 Figure 5. Power Supply for LT1249 auxiliary winding determines VCC according to: VOUT/(VCC – 2V) = NP/NS. For 382V VOUT and 18V VCC, NP/NS ≈ 19. In Figure 5 a new technique for supply voltage eliminates the need for an extra inductor winding. It uses capacitor charge transfer to generate a constant current source which feeds a Zener diode. Current to the Zener is equal to (VOUT – VZ)(C)(f), where VZ is Zener voltage and f is switching frequency. For VOUT = 382V, VZ = 18V, C = 1000pF and f = 100kHz, Zener current will be 36mA. This is enough to operate the LT1249, including the FET gate drive. Output Capacitor The peak-to-peak 120Hz output ripple is determined by: VP-P = (2)(ILOADDC)(Z) where ILOAD DC: DC load current Z: capacitor impedance at 120Hz For 180µF at 300W load, ILOADDC = 300W/385V = 0.78A, LT1249 U U W U APPLICATIONS INFORMATION VP-P = (2)(0.78A)(7.4Ω) = 11.5V. If less ripple is desired, higher capacitance should be used. The selection of the output capacitor should also be based on the operating ripple current through the capacitor. The ripple current can be divided into three major components. The first is at 120Hz whose RMS value is related to the DC load current as follows: I1RMS ≈ (0.71)(ILOADDC) I2RMS = 0.82A at 120VAC, 200W The third component is the switching ripple from the load, if the load is a switching regulator. I3RMS ≈ ILOADDC For United Chemicon KMH 400V capacitor series, ripple current multiplier for currents at 100kHz is 1.43. The equivalent 120Hz ripple current can then be found: 2 I I + 2RMS + 3RMS 1.43 1.43 (I ) 2 1RMS 2 2 0.82A 0.52A 0.37A + + = 0.77A 1.43 1.43 ( ) where LO = hours of load life at rated ripple current and rated ambient temperature ∆TK = capacitor internal temperature rise at rated condition. ∆TK = (I2R)/(KA), where I is the rated current, R is capacitor ESR, and KA is a volume constant. TAMB = operating ambient temperature ∆TO = capacitor internal temperature rise at operating condition In our example, LO = 2000 hours and ∆TK = 10°C at rated 0.95A. ∆TO can then be calculated from: 2 2 I 0.77A ∆TO = RMS (∆TK ) = (10°C ) = 6.6°C 0.95A 0.95A Assuming the operating ambient temperature is 60°C, the approximate life time is: (105°C +10°C)–(60°C + 6.6°C) 10 ILOADDC = 0.52A I1RMS ≈ (0.71)(0.52A) = 0.37A I2RMS ≈ 0.82A at 120VAC I3RMS ≈ ILOADDC = 0.52A IRMS = (105°C + ∆TK )–( TAMB + ∆TO ) 10 LO ≈ (2000)(2) ≈ 57,000 Hrs. 2 For a typical system that runs at an average load of 200W and 385V output: 2 L = (LO )(2) L = expected life time The second component contains the PF switching frequency ripple current and its harmonics. Analysis of this ripple is complicated because it is modulated with a 120Hz signal. However, computer numerical integration and Fourier analysis approximate the RMS value reasonably close to the bench measurements. The RMS value is about 0.82A at a typical condition of 120VAC, 200W load. This ripple is line voltage dependent, and the worst case is at low line. IRMS = The 120Hz ripple current rating at 105°C ambient is 0.95A for the 180µF KMH 400V capacitor. The expected life of the output capacitor may be calculated from the thermal stress analysis: For longer life, capacitor with higher ripple current rating or parallel capacitors should be used. Protection Against Abnormal Current Surge Conditions The LT1249 has an upper limit on the allowed voltage across the current sense resistor. The voltage into the MOUT pin connected to this resistor must not exceed – 6V while the chip is running and –12V under any conditions. The LT1249 gate drive will malfunction if the MOUT pin voltage exceeds – 6V while VCC is powered, destroying the power FET. The 12V absolute limit is imposed by ESD clamps on the MOUT pin. Large currents will flow at 9 LT1249 U W U U APPLICATIONS INFORMATION voltages above 8V and the 12V limit is only for surge conditions. resistor, the standard LT1249 application will not be affected because the chip is not yet powered. Problems are only created if the VCC pin is powered from some external housekeeping supply that remains powered when bridge power is switched off. In normal operation, the voltage into MOUT does not exceed 1.1V, but under surge conditions, the voltage could temporarily go higher. To date, no field failures due to surges have been reported for normal LT1249 configurations, but if the possibility exists for extremely large current surges, please read the following discussion. A huge line voltage surge, beyond the normal worst-case limits, can also create a large current surge. The peak of the line voltage must significantly exceed the storage capacitor voltage (typically 380V) for this to occur, so peak line voltage would probably have to exceed 450V. Such excessive surges might occur if a very large mains load was suddenly removed, with a resulting line “kickback”. If the surge results in voltage at the MOUT pin greater than 6V, it must also last more than 30µs (three switch cycles) to cause FET problems. Offline switching power supplies can create large current surges because of the high value storage capacitor used. The surge can be the result of closing the line switch near the peak of the AC line voltage, or because of a large transient in the line itself. These surges are well known in the power supply business, and are normally controlled with a negative temperature coefficient thermistor in series with the rectifier bridge. When power is switched on, the thermistor is cold (high resistance) and surges are limited. Current flow in the thermistor causes it to heat and resistance drops to the point where overall efficiency loss in the resistor is acceptable. External Clamp The external clamp shown in Figure 6 will protect the LT1249 MOUT pin against extremely large line current surges (see above). Protection is provided for all VCC power methods. The 100Ω resistor and three diodes limit the peak negative voltage into MOUT to less than 3V. Current sense gain is attenuated by only 100Ω/4000Ω = 2.5%. Three diodes are used because the peak negative voltage into MOUT in normal operation could go as high as –1.1V and the diodes should not conduct more than a few microamps under this condition. This basic protection mechanism can be partially defeated if the power supply is switched off for a few seconds, then turned back on. The thermistor has not had time to cool significantly and if the subsequent turn-on catches the AC line near its peak, the resulting surge is much higher than normal. Even if this surge current generates a voltage greater than 6V (but less than 12V) across the sense THERMISTOR + BRIDGE SURGE PATH + STORAGE CAPACITOR RS – 100Ω MOUT LT1249 Figure 6. Protecting MOUT from Extremely High Current Surges 10 LT1249 U PACKAGE DESCRIPTION Dimensions in inches (millimeters) unless otherwise noted. N8 Package 8-Lead PDIP (Narrow 0.300) (LTC DWG # 05-08-1510) 0.400* (10.160) MAX 8 7 6 5 1 2 3 4 0.255 ± 0.015* (6.477 ± 0.381) 0.300 – 0.325 (7.620 – 8.255) 0.065 (1.651) TYP 0.009 – 0.015 (0.229 – 0.381) ( +0.035 0.325 –0.015 8.255 +0.889 –0.381 0.130 ± 0.005 (3.302 ± 0.127) 0.045 – 0.065 (1.143 – 1.651) ) 0.125 (3.175) 0.020 MIN (0.508) MIN 0.018 ± 0.003 0.100 (2.54) BSC (0.457 ± 0.076) N8 1098 *THESE DIMENSIONS DO NOT INCLUDE MOLD FLASH OR PROTRUSIONS. MOLD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.010 INCH (0.254mm) S8 Package 8-Lead Plastic Small Outline (Narrow 0.150) (LTC DWG # 05-08-1610) 0.189 – 0.197* (4.801 – 5.004) 8 7 6 5 0.150 – 0.157** (3.810 – 3.988) 0.228 – 0.244 (5.791 – 6.197) 1 0.010 – 0.020 × 45° (0.254 – 0.508) 0.008 – 0.010 (0.203 – 0.254) 0.053 – 0.069 (1.346 – 1.752) 0°– 8° TYP 0.016 – 0.050 (0.406 – 1.270) 0.014 – 0.019 (0.355 – 0.483) TYP *DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE **DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE 2 3 4 0.004 – 0.010 (0.101 – 0.254) 0.050 (1.270) BSC Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. SO8 1298 11 LT1249 U TYPICAL APPLICATION MURH860 750µH* + 90V TO 270V VOUT EMI FILTER 6A – + IRF840 0.047µF RS 0.2Ω 0.47µF 20k 100pF 1nF 330k 180µF 1M 10Ω 10k 5 7.5V VAOUT 3 EA VSENSE 1M CAOUT 2 1 †V GND 7 4 IA MULTIPLIER IB I 2I IM = A B 2 200µA MAX 250µA IM VCC + 16V/10V – 15µA 1V RUN + – – CA 32k IAC CC 7.5V VREF RMOUT 4k + – 6 MOUT + gm = 1/3k RUN 0.7V + R + Q S 8 GTDR – M1 – + SYNC OSC – 4.7nF 44µA 22µA 4k 20µA 16V ** 1N5819 35pF * 1. COILTRONICS CTX02-12236 (TYPE 52 CORE) AIR MOVEMENT NEEDED AT POWER LEVEL GREATER THAN 250W. 2. COILTRONICS CTX02-12295 (MAGNETICS Kool Mµ® 77930 CORE) ** THIS SCHOTTKY DIODE IS TO CLAMP GTDR WHEN MOS SWITCH TURNS OFF. PARASITIC INDUCTANCE AND GATE CAPACITANCE MAY TURN ON CHIP SUBSTRATE DIODE AND CAUSE ERRATIC OPERATIONS IF GTDR IS NOT CLAMPED. 1249 TA01 † SEE APPLICATIONS INFORMATION SECTION FOR CIRCUITRY TO SUPPLY POWER TO VCC. RELATED PARTS PART NUMBER LT1103 LT1248 LT1508 LT1509 DESCRIPTION Off-Line Switching Regulator Full Feature Average Current Mode Power Factor Controller Power Factor and PWM Controller Power Factor and PWM Controller COMMENTS Universal Off-Line Inputs with Outputs to 100W Provides All Features in 16-Lead Package Simplified PFC Design Complete Solution for Universal Off-Line Switching Power Supplies Kool Mµ is a registered trademark of Magnetics, Inc. 12 Linear Technology Corporation sn1249 1249fbs LT/TP 0799 2K REV B • PRINTED IN USA 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408)432-1900 ● FAX: (408) 434-0507 ● www.linear-tech.com LINEAR TECHNOLOGY CORPORATION 1994