AN1088 APPLICATION NOTE ® L6234 THREE PHASE MOTOR DRIVER by Domenico Arrigo INTRODUCTION The L6234 is a DMOSs triple half-bridge driver with input supply voltage up 52V and output current of 5A. It can be used in a very wide range of applications. It has been realized in Multipower BCD60II technology which allows the combination of isolated DMOS transistors with CMOS and Bipolar circuits on the same chip. It is available in Power DIP 20 (16+2+2) and in Power SO 20 packages. All the inputs are TTL/CMOS compatible and each half bridge can be driven by its own dedicated input and enable. The DMOS structure has an intrinsic free wheeling body diode so the use of external diodes, which are necessary in the bipolar configuration, can be avoided. The DMOS structure allows a very low quiescent current of 6.5 mA typ. at Vs=42V , irrespective of the load. DEVICE DESCRIPTION The device is composed of three channels. Each channel is composed of a half bridge with two power DMOS switches ( typ. Rdson of 300mW @ 25°C) and intrinsic free wheeling diodes. Each channel includes two TTL/CMOS and uP compatible comparators, and a logic block to interface the inputs with the drivers. The device includes an internal bandgap reference of 1.22V, a 10V voltage reference to supply the internal circuitry of the device, a central charge pump to drive the upper DMOS switch, thermal shutdown protection and an internal hysteretic function which turns off the device when the junction temperature exceeds approximately 160 °C. Hysteresys is about 20 °C. Figure 1. L6234 Block Diagram C4 220nF C3 10nF VCP C5 1µF VREF D2 1N4148 D1 1N4148 VBOOT Vs VREF 10V CHARGE PUMP Vs IN1 T1 Vs C2 100nF C1 100µF OUT1 EN1 T2 IN2 BRUSHLESS MOTOR WINDINGS T3 OUT2 EN2 T4 SENSE1 THERMAL PROTECTION IN3 T5 OUT3 EN3 T6 SENSE2 RSENSE GND D98IN940A April 2001 1/14 AN1088 APPLICATION NOTE PIN DESCRIPTION. Figure 2. Vs ( INPUT SUPPLY VOLTAGE PINS). VS These are the two input supply voltage pins. The unregulated input DC voltage can range from 7V to 52V. T1 T3 ON/OFF ON VF With inductive loads the recommended operating maximum supply voltage is 42V to prevent overvoltage applied to the L B C OFF -VF (VS+VF) DMosfets. In fact considering a full bridge configuration (see fig. 2), when the bridge is switched off (ENABLE CHOPPING) the current recirculation produces a negative voltage to the ON/OFF T2 T4 source of the lower DMOS switches (point A). In this condiA S tion the drain-source voltage of T1 and T4 is VS + VF + Vsense . -VSENSE Dinamically VF can be same Volts depending on the current Rsense slope, dI/dt, and also Vsense, depending on the parasitic inD98IN938A ductance and current slope can be some Volts. So the drainsource voltage of T1 and T4 DMOS switches can reach more than 10V over the VS voltage. The input capacitors C1 and C2 are chosen in order to reduce overvoltage due to current decay and to parasitic inductance. For this reason C2 has to be placed as closed as possible to VS and GND pins. The device can sustain a 4A DC input current for each of the two Vs pins, in accordance with the power dissipation. Figure 3. Reference Voltage vs. Junction Temperature. OUT1, OUT2, OUT3 (OUTPUTS). These are the output pins that correspond to the mid point of each half bridge. They are Vref [V] designed to sustain a DC current of 4A. 11 Vs = 52V 10 Vs = 24V 9 SENSE1, SENSE2. Vs = 10V SENSE1 is the common source of the lower DMOS of the half 8 bridge 1 and 2. 7 SENSE2 is the source of the lower DMOS of the half bridge 6 Vs = 7V 3. 5 Each of these pins can handle a current of 5A. 4 A resistance, Rsense, connected to these pins provides feed- 3 back for motor current control. 2 Care must be taken with the negative voltage applied to these 1 pins : negative DC voltage lower than -1V could damage the device. For duration lower than 300ns the device can sustain 0-50 -25 0 25 50 75 100 125 pulsed negative voltage up to -4V. Tj [°C] For example, if enable chopping current control method is used, negative voltage pulses appear to these pins, due to the Figure 4. Reference Voltage vs. Supply Voltage. current recirculation through the sensing resistor. Vref [V] 12 Vref ( Voltage Reference). This is the internal 10V voltage reference pin to bias the logic and the low voltage circuitry of the device. A 1µF electrolytic capacitor connected from this pin to GND ensures the stability of the DMOS drive circuit. This pin can be externally loaded up to 5mA . Figure 3 and 4 show the typical behavior of the Vref pin. 10 8 6 4 Tj = 25°C Vcp ( CHARGE PUMP ). This is the internal oscillator output pin for the charge pump. The oscillator supplied by the 10V Voltage Reference switches from GND to 10V with a typical frequency of 2/14 2 0 0 10 20 30 Vs [V] 40 50 150 AN1088 APPLICATION NOTE 1.2MHz (see fig 4). When the oscillator output is at ground , C3 is charged by Vs through D1. When it rises to 10V, D1 is reverse biased and the charge flows from C3 to C4 through D2, so the Vboot pin after a few cycles reaches the maximum voltage of Vs + 10V - VD1- VD2. Vboot ( BOOTSTRAP). This is the input bootstrap pin which gives the overvoltage necessary to drive all the upper DMOS of the three half bridges (see fig 5). Figure 5. Charge Pump Circuit. Vs C2 0.1µF Vs+Vref-VD1-VD2 Vs+Vref-VD1 C1 100µF D1 1N4148 Vs-VD1 f=1.2 MHz VCP C3 10nF D2 1N4148 C4 0.22µF VBOOT Vs Vref f=1.2 MHz CHARGE HIGH SIDE DRIVER OUT SENSE Vref 10V PUMP LOGIC INPUTS PINS. EN1, EN2, EN3 (ENABLES). These pins are TTL/CMOS and µP compatible. Each half bridge can be enabled by its own dedicated pin with a logic HIGH. The logic LOW on these pins switches off the related half bridge (see Fig. 6). The maximum switching frequency is 50kHz. Figure 6. Control logic for each half bridge. Figure 7. Cross Conduction Protection. high level INPUT high level high level INPUT PIN low level low level time high level ENABLE low level low level high level time low level low level DMOS ON time UPPER DMOS DMOS OFF DMOS OFF DMOS ON UPPER DMOS DMOS OFF DMOS OFF DMOS OFF 300ns LOWER DMOS LOWER DMOS time tdelay 300ns DMOS ON time DMOS ON tdelay DMOS ON DMOS OFF DMOS OFF DMOS OFF DMOS OFF time time IN1, IN2, IN3 (INPUTS). These pins are TTL/CMOS and µP compatible. They allow switching on the upper DMOS ( INPUT at high logic level) or the lower Dmos (INPUT at low logic level) in each half bridge (see Fig. 6). 3/14 AN1088 APPLICATION NOTE Cross conduction protection (see Fig. 7) avoids simultaneously turning on both the upper and lower DMOS of each half bridge. There is a fixed delay time of 300ns between the turn on and the turn off of the two DMOS switches in each half bridge. The switching operating frequency is up 50kHz. High commutation frequency permits the reduction of ripple of the output current but increases the device’s power dissipation, however low commutation frequency causes high ripple of the output current. The switching frequency should be higher than 16kHz to avoid acoustic noises. The sink current at the INPUTS and ENABLES pins is approximately 30µA if the voltage to these pins is at least 1V less than the Vref voltage (see Fig. 3 and Fig. 4). To avoid overload of the logic INPUTS and ENABLES , voltage should be applied to Vs prior to the logic signal inputs. POWER DISSIPATION An evaluation of the power dissipation of the IC driving a three phase motor in a chopping current control application follows. With a simplified approach it can be distinguished three periods (see Fig. 8) : Figure 8. Rise Time, Tr, period. This is the rise time period, Tr, in which the current switches from one winding to another. In this time a DMOS is switched on and the current increases up to the peak value Ipk with the law i(t) = (Ipk/Tr) t. The energy lost for the rise time in the period T is : Tchop Ipk Iload Ival Tr Erise = ∫ Rdson ⋅ i2(t)dt = Rdson ⋅ I2pk ⋅ 0 Trise Tload Tr 3 Fall Time,Tf, period. When the current switches from one winding to another, there is a fall time in which the current that flows in the intrisic diode of the DMOS decreases from Ipk to zero. If VD is the voltage fall of the diode, the energy lost is : Tfall tf Efall = ∫ VD(t) ⋅ i(t)dt 0 Tload During this time the current that flows in the winding is limited by the chopping current control. The energy dissipated due to the ON resistance of the DMOS is : Eload = Rdson ⋅ (Irms)2 ⋅ Tload In the formula, Irms is the RMS load current, given by : Irms = Ipk − Ival 2 √ (Iload)2 + 3 √ and Iload is the average load current. When the switch is ON, the energy dissipated due to the commutation of the chopping current control in the DMOS can be assumed to be: tcom Eon = Vs ⋅ Ival ⋅ 2 where tcom is the commutation time of the DMOS switch. 4/14 AN1088 APPLICATION NOTE When the switch is OFF : Eoff = Vs ⋅ Ipk ⋅ tcom 2 The energy lost by commutation in a chopping period, given by Eon + Eoff, is : Ecom = Vs ⋅ Iload ⋅ tcom The energy lost by commutation during the Tload time is given by : Ecom = Vs ⋅ Iload ⋅ tcom ⋅ Tload ⋅ fchop Quiescent Power Dissipation, Pq. The power dissipation due to the quiescent current is Pq = Vs ⋅ Iq , in which Iq is the quiescent current at the chopping frequency, fchop = 1/Tchop. Total Power Dissipation. Let’s evaluate the power dissipation of the device driving a three phase brushless motor in chopping current control. In the driving sequence only one upper DMOS and a lower one are on at the same time (see fig. 9 and 10). The total power dissipation is given by : Ptot = 2 ⋅ (Erise + Efall + Eload + Ecom) + Pq T Figure 11 shows the total power dissipation, Pd, of the L6234 driving a three phase brushless motor in input chopping current control at different chopping frequency. EVALUATION BOARD. The L6234 Power SO20 board has been realized to evaluate the device driving, in closed loop control, a three phase brushless motor with open collector Hall effect sensors. Figure 9. Input chopping current circulation. _ PHASE 12 CHOPPING INPUT I1A half bridge 1 Vs half bridge 2 ILOAD I1A ON/OFF OFF OUT1 OUT2 ILOAD OFF/ON I1B ON I2B I1B IOFF I2B 5/14 AN1088 APPLICATION NOTE Figure 10. Three Phase Brushless motor control sequence. IOUT1 BRUSHLESS MOTOR OUT1 OUT2 L6234 OUT3 IOUT2 T IOUT3 The device soldered on the copper heat dissipating Figure 11. L6234 Power Dissipation in Three area on the board ,without any additional heat sink, Phase Brushless Motor Control. can sustain a DC load current of 2.3 A at Tamb of Pd [W] approximately 40 °C. INPUT CHOPPING Vs=36V fchop=30kHz The board provides a closed loop speed and torque 15 L=2mH fchop=50kHz control, with a constant TOFF chopping current conT=2ms Tj=100C trol method. It allows the user to change the direction and brake the motor. DC 10 Constant tOFF Chopping Current Control. When the current through the motor exceeds the threshold, fixed by the ratio between the control 5 voltage Vcontrol and the sensing resistor, Rsense, an error signal is generated, the output of the LM393 comparator switches to ground. This state is maintained by the monostable (M74HC123) for a 0 constant delay time ( tOFF ) generating a PWM sig2 0 1 3 4 5 nal that achieves the chopping current control. The ILOAD [A] PWM signal is used for chopping the INPUT pattern. During the toff in chopping current control, the current flows in the low side loop ( see fig. 9 ) and does not flow through the sensing resistor. The tOFF value can be set by the R9 and C11 to values shown in the table 1. A suitable value of toff for the majority of applications is 30µs. The larger the tOFF, the higher is the current ripple. If the tOFF is too large the ripple current becomes excessive . On the other hand if the tOFF is too small the winding current cannot decrease under the threshold and current regulation is not guaranteed. 6/14 AN1088 APPLICATION NOTE Figure 12. Application board Schematic Circuit. Vs=8V to 42 +5V OUT 3 T1 1 L7805 IN J7 C1 100uF 60V 2 C7 10uF GND C6 220nF Z1 18V 10k R5 1N4148 1N4148 D1 D2 C3 10nF C2 100nF Vs HALL EFFECT SIGNALS IN1 IN2 IN2 IN3 IN3 14 EN1 8 EN2 3 EN3 13 CONTROL EN1 LOGIC EN2 BRAKE DIR 9 12 IN1 EN3 PWM Vcp 17 18 CONSTANT toff CHOPPING CURRENT CONTROL R11 10k B PWM C10 100nF 16 _ Q 2 3 1 10 11 20 16 2 19 C5 1uF TORQUE & SPEED CONTROL A M74HC123 4 monostable 15 14 R2 R3 R4 1Ω 1Ω 1Ω REFERENCE SPEED Reference Speed Table 1. toff selection 2 + 4 3 R6 1K Vsense C8 470pF R7 11 k 8 LM393 R9 R1 1Ω Hall effect signal 1 BRUSHLESS MOTOR SENSE Vref 8 1 OUT3 15 R10 10K each Vcontrol C9 100nF 5 POWER SO20 +5V +5V OUT2 L6234 Figure 13. Constant toff current control. +5V OUT1 6 +5V PWM HALL EFFECT SENSORS Vboot 7 4 GND Vsense C4 220nF toff 20µs 30µs R9 100k 100k C11 270pF 330pF 45µs 70µs 100k 100k 560pF 1nF +5V J1 +5V 100k Vcontrol R8 1K C11 330pF Torque & Speed Closed Loop Control. The motor’s rotational speed is determined by the frequency of the Hall effect signals. The speed control loop has been achieved by comparing this frequency with a frequency of a reference oscillator (see fig. 14) that corresponds to a desired speed limit. Figure 14. PLL Motor Control. REFERENCE FEEDBACK PHASE/ FREQUENCY DETECTOR Amp. Vcontrol PWM MOTOR COMPENSATION NETWORK HALL SENSORS D01IN1209 7/14 AN1088 APPLICATION NOTE Figure 15. Oscillator for Reference Speed. When the hall effect signal frequency is lower than the reference frequency, the control voltage is maintained to a value that sets the motor current limit and therefore the torque control limit. The peak current limit is given by Ipeak = Vcontrol/Rsense. When the frequency from the Hall Effect sensors exceeds the reference frequency and an error signal is generated by the PLL (see Fig. 14). An LM358 comparator, a loop amplifier and an auxiliary OP-AMP ensure the right gain and filtering to guarantee the stability (see fig.16). The error signal causes Vcontrol decrease to a value that sets the PWM chopping current control in order to reduce the torque and set the desired speed. The motor speed is regulated to within ± 0.02 % of the desired speed. Reference Speed +5V 4 8 3 NE555 1 R26 36K R27 7 C21 100nF 16K 6 5 2 C19 100nF C20 100nF Figure 16. Phase Locked Loop and filtering. +5V +5V R17 10M LM358 C12 100nF 8 R14 47K 3 Vcontrol + - 1 2 BAT47 R13 47K +5V 4 R12 47K 33K +5V R20 Output 11 Aux. OP-AMP 2.5V 8 9 Loop Amplifier +VIN 13 C14 100nF C13 1uF R21 91K C17 47nF 270K R15 47K P2 1K R19 91K R16 P1 5K +5V GND C16 220nF C15 100nF R18 14 HALL1 (Speed feedback) 6 12 10 5 3635 Phase/ Frequency Detector 3 2 7 15 1 Reference Speed TP8 Figure 17. Control Logic Circuit. +5V +5V R22 R23 10k 10k MOTOR HALL EFFECT SIGNALS R29 10k SW2 R24 10k HALL1 1 7 19 2 18 HALL2 HALL3 DIR BRAKE PWM 3 17 4 GAL 16V8 16 5 15 6 14 10 20 EN1 EN2 EN3 IN1 IN2 IN3 EN1 EN2 EN3 IN1 IN2 IN3 PWM +5V R25 10k R26 10k C18 100nF DIRECTION CHANGE DIR =0 GND : BACK ROTATION DIR = 5V : FORWARD ROTATION BRAKE GND SW1 DIR J1 BRAKE FUNCTION BRAKE = GND : BRAKE BRAKE = 5V : GO 8/14 4 16 GND Control Logic Circuit. The logic sequence to the motor is generated by a GAL16V8, which decodes the Hall Effect signals and generates the INPUT and ENABLE pattern shown in Fig. 18. The brake function is obtained by setting the input pattern to logic low and thus turning on the lower DMOS switches of the enabled halfbridges. The PWM signal is used for chopping the INPUT pattern. The control logic circuit decodes Hall effect sensors having different phasing. With the DIR jumper opened the application achieves forward rotation for motors having 60° and 120° Hall Effect sensor electrical phasing and the reverse rotation for motors having 300° and 240° Hall Effect sensor phasing. Connecting the DIR jumper to ground sets the reverse rotation for motors having 60° and 120° Hall sensors phasing and the forward rotation for motors having 300° and 240° Hall sensor phasing. The SW2 switch performs the startstop function. AN1088 APPLICATION NOTE Figure 17. 0˚ ELECTRICAL DEGREES 360˚ HALL1 SENSOR SIGNALS HALL2 HALL3 EN1 ENABLE EN2 EN3 IN1 FORWARD ROTATION IN2 IN3 IN1 REVERSE ROTATION IN2 IN3 IOUT1 0 MOTOR DRIVE CURRENT IN FORWARD ROTATION IOUT2 0 IOUT3 0 NO PWM PWM CONSTANT tOFF D98IN912 9/14 AN1088 APPLICATION NOTE Layout Considerations. Special attention must be taken to avoid overvoltages at Vs and additional negative voltages to the SENSE pins and noise due to distributed inductance. Thus the input capacitor must be connected close to the Vs pins with symmetrical paths. The paths between the SENSE pins and the input capacitor ground have to be minimized and symmetrical . The sensing resistors must be non-inductive. The device GND has to be connected with a separate path to the input capacitor ground. Figure 19. Application Board Layout. Figure 20. Component side. 10/14 AN1088 APPLICATION NOTE Figure 21. Copper side. APPLICATION IDEAS. The L6234 can be used in many different applications. Typical examples are a half bridge driver using one channel and a full bridge driver using two channels. In addition, the bridges can be paralleled to reduce the RDSon and the device dissipation. The paralleled configuration can also be used to increase output current capability. Channel 1 can be paralleled with Channel 3 or Channel 2 can be paralleled with Channel 3. Channel 1 should not be paralleled with Channel 2 because the sources of their low side DMOSs are connected to the same SENSE1 pin . Application ideas for the L6234 follow. Figure 22. Constant frequency current control Vs 1N4148 1N4148 100uF 220nF 100nF CONTROL LOGIC EN1 EN1 EN2 EN2 EN3 EN3 IN1 IN1 10nF Vs Vcp Vboot OUT1 OUT2 L6234 POWER SO20 IN2 OUT3 IN3 IN2 GND Vref SENSE1 IN3 SENSE2 Reset 1uF S Vsense Q R RSENSE S +5V Rx Cx OSC Q +5V R Vcontrol +5V 100nF L6506 Constant frequency Current Control 1 Fchop= __________ 0.69 Rx Cx for Rx>10kOhm 11/14 AN1088 APPLICATION NOTE Low Cost Application with Speed and Torque Control Loops. Figure 23. Complete three phase brushless motor application with speed and torque control. VS IN1 HALL EFFECT SIGNALS 7 IN2 17 18 L6234 8 EN2 +5V 16 2 19 GND 1 7 1/2 M74HC123 MONOSTABLE Q 12 11 6 + RSENSE 0.3W VSENSE 5 V5=+5V 1/4 TSM221 6 7 BRUSHLESS MOTOR SENSE V5=+5V A OUT3 15 13 1,10,11,20 11 OUT2 5 POWER SO20 3 EN3 10 OUT1 6 14 EN1 PWM 12 4 IN3 CONTROL LOGIC 9 HALL EFFECT SENSORS 1/fe C1 200nF V5 3 Vm +5V Rx2 R2 1M 1 Cx2 VCONTROL R3 4K R4 1K 1/4 TSM221 4 + 2 R1 100K 3 Ton 1Q Vref (Reference Speed Voltage) 16 2 1/2 M74HC123 MONOSTABLE 13 15 1 14 1B HALL EFFECT SIGNAL 8 +5V Rx1 100K Cx1 D01IN1210 SPEED LOOP A low cost solution to obtain a complete three phase brushless motor control application with speed and torque closed control loop is shown in Fig. 23. This simple low cost solution is useful when high dynamic performances and accuracy of the speed loop are not required. The current regulation limit, which determines the torque , is given by Vcontrol/Rsense. The constant toff of the PWM is fixed by Rx2 and Cx2. The speed loop is realised using a Hall effect signal, whose frequency is proportional to the motor speed. At each positive transition of the Hall effect sensors the monostable maintains the pulse for a constant time , Ton, with a fixed amplitude, V5. The average value of this signal is proportional to the frequency of the Hall effect signal and the motor speed . An OP-AMP configured as an integrator , filters this signal and compares it with a reference voltage, Vref, which sets the speed . The generated error signal is the control voltage, Vcontrol, of the currrent loop. Therefore the current loop modifies the produced torque in order to regulate the speed at the desired value. The values of Cf and R2 should be chosen to obtain a nearly ripple free op-amp output, even at low motor speed. This constrain limits the system bandwidth and so limits the response time of the loop. The regulated speed, for a rotor with n pairs of permanent magnetic poles , is given by : R1 1 + R2 ωm = ⋅ Vref ⋅ 60 1 V5 ⋅ Ton ⋅ n + KG with KG = [RPM] R4 1 1 R2 ⋅ Kt ⋅ ⋅ ⋅ Rsense B R1 R3 + R4 in which Kt, expressed in [Nm/A] , is the Motor Torque constant and B, expressed in [Nms], is the Total Viscous Friction. In most cases 1/KG can be neglected. 12/14 AN1088 APPLICATION NOTE The Ton values, given by KCx1Rx1, must be less than the period of the Hall effect electrical signal at the desired motor speed , so Ton must meet the requirement of 1.1 : ( 1.1 ) Ton < 60 n ⋅ ωm For the motor and the load used in this application, which have the following parameters : Jt = 10-4 [Kg ⋅ m2] (Motor plus Load Inertia Moment); Kt = 10-2 [Nm/A] ; n=4 ; R1=100k +/- 10% [kΩ] ; R2=1M ±1[kW] ; B = 10-5 [Nms] Cf=220n [F] A regulated speed of 6000RPM can be obtained with an accuracy of around +/-3%, considering Ton accuracy of +/-1% , the V5 and Vref mismatch of +/-1% . If the speed is 6000RPM, there are 100 rotor revolution for second, with n=4, the Hall effect frequency is 400Hz. Therefore Ton has to be lower than 2.5ms (according to equation 1.1). The phase margin is about 45° and the response time of the speed loop for a speed step variation is around 200ms . 6X6 BRUSHLESS APPLICATION Figure 24. 6x6 Three Phase Brushlees Application Circuit Vs 100uF 100nF SPEED AND POSITION FEEDBACK IN1 IN2 IN3 EN1 EN2 EN3 EN2 Vcp Vref THREE PHASE BRUSHLESS MOTOR Vboot OUT1 OUT1A OUT2 OUT1B L6234 GND CONTROL LOGIC 220nF 10nF Vs IN1A IN1B IN2A EN1 1N4148 1N4148 OUT3 OUT2A SENSE1 SENSE2 OUT2B 1uF Vs 100nF 100uF 1N4148 1N4148 220nF 10nF Vs IN3A IN1 IN2 IN3 EN1 EN2 EN3 IN3B IN2B EN3 Constant Toff PWM Current Control. Two M74HC124 plus an LM339 Vcp Vboot OUT1 L6234 OUT3 GND Vref OUT3A OUT2 OUT3B SENSE1 SENSE2 Comparator & monostable +5V 100nF +5V 8 B 100nF PWM3 16 2 3 1 A M74HC123 monostable 15 _ 4 14 8 Q 1 1uF +5V LM339 + 1K Vsense3 470pF 4 Vcontrol +5V Comparator & monostable Vsense2 PWM2 Comparator & monostable Vsense1 PWM1 13/14 AN1088 APPLICATION NOTE Information furnished is believed to be accurate and reliable. However, STMicroelectronics assumes no responsibility for the consequences of use of such information nor for any infringement of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of STMicroelectronics. Specification mentioned in this publication are subject to change without notice. This publication supersedes and replaces all information previously supplied. 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