LT1913 25V, 3.5A, 2.4MHz Step-Down Switching Regulator U DESCRIPTIO FEATURES ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ The LT®1913 is an adjustable frequency (200kHz to 2.4MHz) monolithic buck switching regulator that accepts input voltages up to 25V. A high efficiency 95m switch is included on the die along with a boost Schottky diode and the necessary oscillator, control, and logic circuitry. Current mode topology is used for fast transient response and good loop stability. Shutdown reduces input supply current to less than 1μA while a resistor and capacitor on the RUN/SS pin provide a controlled output voltage ramp (soft-start). A power good flag signals when VOUT reaches 91% of the programmed output voltage. The LT1913 is available in 10-Pin 3mm × 3mm DFN packages with exposed pads for low thermal resistance. Wide Input Range: 3.6V to 25V 3.5A Maximum Output Current Adjustable Switching Frequency: 200kHz to 2.4MHz Low Shutdown Current: IQ < 1μA Integrated Boost Diode Synchronizable Between 250kHz to 2MHz Power Good Flag Saturating Switch Design: 95m On-Resistance 0.790V Feedback Reference Voltage Output Voltage: 0.79V to 25V Thermal Protection Soft-Start Capability Small 10-Pin (3mm × 3mm) DFN Packages , LT, LTC and LTM are registered trademarks of Linear Technology Corporation. All other trademarks are the property of their respective owners. U APPLICATIO S ■ ■ ■ ■ ■ Automotive Battery Regulation Power for Portable Products Distributed Supply Regulation Industrial Supplies Wall Transformer Regulation U TYPICAL APPLICATIO 5V Step-Down Converter Efficiency VOUT 5V 3.5A VIN 6.5V TO 25V VIN = 12V BD VIN OFF ON RUN/SS 90 BOOST EFFICIENCY (%) 0.47μF 4.7μH 15k VC 10μF LT1913 SW RT 680pF 100 80 VIN = 24V 70 PG 63.4k SYNC 60 536k GND VOUT = 5V L = 4.7μH f = 600kHz FB 47μF 100k 50 0 0.5 2 1.5 1 2.5 OUTPUT CURRENT (A) 3 3.5 1913 TA01a 1913 G01 1913f 1 LT1913 W W U W ABSOLUTE AXI U RATI GS (Note 1) VIN, RUN/SS Voltage .................................................25V BOOST Pin Voltage ...................................................50V BOOST Pin Above SW Pin.........................................25V FB, RT, VC Voltage .......................................................5V PG, BD Voltage .........................................................25V SYNC Voltage ............................................................20V Operating Junction Temperature Range (Note 2) LT1913E ............................................. –40°C to 125°C LT1913I .............................................. –40°C to 125°C Storage Temperature Range................... –65°C to 150°C PIN CONFIGURATION TOP VIEW 10 RT BD 1 BOOST 2 SW 3 VIN 4 7 PG RUN/SS 5 6 SYNC 9 VC 11 8 FB DD PACKAGE 10-LEAD (3mm × 3mm) PLASTIC DFN θJA = 45°C/W, θJC = 10°C/W EXPOSED PAD (PIN 11) IS GND, MUST BE SOLDERED TO PCB ORDER INFORMATION LEAD FREE FINISH TAPE AND REEL PART MARKING* PACKAGE DESCRIPTION TEMPERATURE RANGE LT1913EDD#PBF LT1913EDD#TRPBF LDJW 10-Lead (3mm × 3mm) Plastic DFN –40°C to 125°C LT1913IDD#PBF LT1913IDD#TRPBF LDJW 10-Lead (3mm × 3mm) Plastic DFN –40°C to 125°C Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container. Consult LTC Marketing for information on non-standard lead based finish parts. For more information on lead free part marking, go to: http://www.linear.com/leadfree/ For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/ ELECTRICAL CHARACTERISTICS The ● denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN = 10V, VRUN/SS = 10V, VBOOST = 15V, VBD = 3.3V unless otherwise noted. (Note 2) PARAMETER CONDITIONS ● Minimum Input Voltage Quiescent Current from VIN VRUN/SS = 0.2V VBD = 3V, Not Switching Quiescent Current from BD ● TYP MAX UNITS 3 3.6 V 0.01 0.5 μA 0.45 1.2 mA VBD = 0, Not Switching 1.3 2.3 mA VRUN/SS = 0.2V 0.01 0.5 μA 0.9 1.8 mA 1 10 μA 2.7 3 V VBD = 3V, Not Switching VBD = 0, Not Switching Minimum Bias Voltage (BD Pin) MIN ● 1913f 2 LT1913 ELECTRICAL CHARACTERISTICS The ● denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN = 10V, VRUN/SS = 10V, VBOOST = 15V, VBD = 3.3V unless otherwise noted. (Note 2) PARAMETER CONDITIONS Feedback Voltage ● FB Pin Bias Current (Note 3) VFB = 0.8V, VC = 0.4V FB Voltage Line Regulation 4V < VIN < 25V MIN TYP MAX UNITS 780 775 790 790 800 805 mV mV 10 40 nA 0.002 0.01 %/V ● Error Amp gm 525 Error Amp Gain 2000 μMho VC Source Current 60 μA VC Sink Current 60 μA VC Pin to Switch Current Gain 5.3 A/V VC Clamp Voltage 2.0 V Switching Frequency RT = 8.66k RT = 29.4k RT = 187k 2.2 1.0 200 ● Minimum Switch Off-Time 4.6 2.45 1.1 230 2.7 1.25 260 MHz MHz kHz 60 150 nS 5.4 6.0 A Switch Current Limit Duty Cycle = 5% Switch VCESAT ISW = 3.5A 335 Boost Schottky Reverse Leakage VSW = 10V, VBD = 0V 0.02 2 ● mV μA 1.5 2.0 V BOOST Pin Current ISW = 1A 35 60 mA RUN/SS Pin Current VRUN/SS = 2.5V 5 8 μA 2.5 V Minimum Boost Voltage (Note 4) RUN/SS Input Voltage High RUN/SS Input Voltage Low PG Threshold Offset from Feedback Voltage 0.2 VFB Rising PG Hysteresis PG Leakage VPG = 5V PG Sink Current VPG = 0.4V SYNC Low Threshold V 65 mV 10 mV 0.1 ● 200 800 V 0.8 VSYNC = 0V Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Note 2: The LT1913E is guaranteed to meet performance specifications from 0°C to 125°C. Specifications over the –40°C to 125°C operating temperature range are assured by design, characterization and correlation with statistical process controls. The LT1913I specifications are guaranteed over the –40°C to 125°C temperature range. μA μA 0.5 SYNC High Threshold SYNC Pin Bias Current 1 0.1 V μA Note 3: Bias current flows out of the FB pin. Note 4: This is the minimum voltage across the boost capacitor needed to guarantee full saturation of the switch. 1913f 3 LT1913 U W TYPICAL PERFOR A CE CHARACTERISTICS Efficiency Efficiency Efficiency 100 100 VIN = 12V EFFICIENCY (%) 80 VIN = 24V 70 60 50 0 0.5 80 VIN = 24V 70 60 VOUT = 5V L = 4.7μH f = 600kHz 2 1.5 1 2.5 OUTPUT CURRENT (A) 3 3.5 EFFICIENCY (%) VIN = 12V 90 50 0.5 2 1.5 1 2.5 OUTPUT CURRENT (A) 3 1913 G01 90 2.5 80 2.0 70 1.5 VIN = 12V VOUT = 5V L = 4.7μH f = 600kHz 50 3.5 0 LOAD CURRENT (A) 3.5 VOUT = 3.3V TA = 25°C L = 4.7μH f = 600kHz 3.0 5 10 4.5 MINIMUM 4.0 VOUT = 5V TA = 25°C L = 4.7μH f = 600kHz 3.5 2.5 20 15 INPUT VOLTAGE (V) SWITCH CURRENT LIMIT(A) TYPICAL 5.0 4.0 20 15 INPUT VOLTAGE (V) 1913 G06 6.0 4.0 3.5 120 600 105 500 400 300 200 100 25 50 75 100 125 150 TEMPERATURE (°C) 1913 G09 60 40 DUTY CYCLE (%) 80 100 Boost Pin Current 700 3.0 2.5 20 1913 G08 BOOST PIN CURRENT (mA) VOLTAGE DROP (mV) DUTY CYCLE = 90 % 0 0 25 DUTY CYCLE = 10 % 5.0 2.0 –50 –25 4.0 Switch Voltage Drop 6.5 4.5 4.5 1913 G07 Switch Current Limit 5.5 5.0 3.0 10 5 5.5 3.5 3.0 25 0.5 Switch Current Limit TYPICAL MINIMUM 3.5 3 6.0 5.5 4.5 2 1.5 1 2.5 OUTPUT CURRENT (A) 1913 G03 Maximum Load Current Maximum Load Current 5.0 0.5 1.0 1913 G02 5.5 LOAD CURRENT (A) 3.0 60 VOUT = 3.3V L = 3.3μH f = 600kHz 0 100 TOTAL POWER LOSS (W) EFFICIENCY (%) 90 SWITCH CURRENT LIMIT (A) TA = 25°C unless otherwise noted. 90 75 60 45 30 15 0 0 1 2 4 3 SWITCH CURRENT (A) 5 1913 G10 0 0 1 2 3 SWITCH CURRENT (A) 4 5 1913 G11 1913f 4 LT1913 U W TYPICAL PERFOR A CE CHARACTERISTICS TA = 25°C unless otherwise noted. Switching Frequency Feedback Voltage 840 Frequency Foldback 1200 1.20 RT = 34.0k RT = 34.0k SWITCHING FREQUENCY (kHz) 820 1.10 FREQUENCY (MHz) FEEDBACK VOLTAGE (mV) 1.15 800 780 1.05 1.00 0.95 0.90 1000 0.85 760 –50 –25 0 0.80 –50 –25 25 50 75 100 125 150 TEMPERATURE (°C) 25 50 75 100 125 150 TEMPERATURE (°C) 6 100 80 60 40 100 200 300 400 500 600 700 800 900 FB PIN VOLTAGE (mV) 0 1913 G14 RUN/SS Pin Current 12 RUN/SS PIN CURRENT (μA) 120 SWITCH CURRENT LIMIT (A) MINIMUM SWITCH ON TIME (ns) 7 5 4 3 2 0 0 25 50 75 100 125 150 TEMPERATURE (°C) 10 8 6 4 2 1 20 0.5 2.5 2 1.5 RUN/SS PIN VOLTAGE (V) 1 3 0 3.5 0 5 20 15 10 RUN/SS PIN VOLTAGE (V) 1913 G16 1913 G15 Minimum Input Voltage 50 1.4 25 1913 G17 Error Amp Output Current Boost Diode 5.0 40 1.2 4.5 1.0 0.8 0.6 0.4 20 INPUT VOLTAGE (V) VC PIN CURRENT (μA) 30 BOOST DIODE VF (V) 200 Soft-Start Minimum Switch On-Time 10 0 –10 –20 –30 0.2 0 400 1913 G13 140 0 600 0 0 1913 G12 0 –50 –25 800 4.0 3.5 3.0 2.5 –40 0 0.5 1.0 1.5 BOOST DIODE CURRENT (A) 2.0 1913 G18 –50 –200 2.0 –100 100 0 FB PIN ERROR VOLTAGE (mV) 200 1913 G19 VOUT = 3.3V TA = 25°C L = 4.7μH f = 600kHz 1 10 100 1000 LOAD CURRENT (mA) 10000 1913 G20 1913f 5 LT1913 U W TYPICAL PERFOR A CE CHARACTERISTICS VC Voltages 6.5 2.50 6.0 2.00 Power Good Threshold 95 THRESHOLD VOLTAGE (%) Minimum Input Voltage CURRENT LIMIT CLAMP VC VOLTAGE (V) INPUT VOLTAGE (V) TA = 25°C unless otherwise noted. 5.5 5.0 VOUT = 5V TA = 25 °C L = 4.7μH f = 600kHz 4.5 1.00 SWITCHING THRESHOLD 85 80 0.50 4.0 1 1.50 90 10 100 1000 LOAD CURRENT (mA) 10000 0 –50 –25 0 25 50 75 100 125 150 TEMPERATURE (°C) 75 –50 –25 0 25 50 75 100 125 150 TEMPERATURE (°C) 1913 G22 1913 G21 Switching Waveforms; Discontinuous Operation 1913 G23 Switching Waveforms; Continuous Operation VSW 5V/DIV VSW 5V/DIV IL 0.2A/DIV IL 0.5A/DIV VOUT 10mV/DIV VOUT 10mV/DIV VIN = 12V VOUT = 3.3V ILOAD = 110mA 1μs/DIV 1913 G25 VIN = 12V VOUT = 3.3V ILOAD = 1A 1μs/DIV 1913 G26 1913f 6 LT1913 U U U PI FU CTIO S BD (Pin 1): This pin connects to the anode of the boost Schottky diode. BD also supplies current to the internal regulator. BOOST (Pin 2): This pin is used to provide a drive voltage, higher than the input voltage, to the internal bipolar NPN power switch. SW (Pin 3): The SW pin is the output of the internal power switch. Connect this pin to the inductor, catch diode and boost capacitor. VIN (Pin 4): The VIN pin supplies current to the LT1913’s internal regulator and to the internal power switch. This pin must be locally bypassed. RUN/SS (Pin 5): The RUN/SS pin is used to put the LT1913 in shutdown mode. Tie to ground to shut down the LT1913. Tie to 2.5V or more for normal operation. If the shutdown feature is not used, tie this pin to the VIN pin. RUN/SS also provides a soft-start function; see the Applications Information section. SYNC (Pin 6): This is the external clock synchronization input. Ground this pin when not used. Tie to a clock source for synchronization. Clock edges should have rise and fall times faster than 1μs. Do not leave pin floating. See synchronizing section in Applications Information. PG (Pin 7): The PG pin is the open collector output of an internal comparator. PG remains low until the FB pin is within 9% of the final regulation voltage. PG output is valid when VIN is above 3.6V and RUN/SS is high. FB (Pin 8): The LT1913 regulates the FB pin to 0.790V. Connect the feedback resistor divider tap to this pin. VC (Pin 9): The VC pin is the output of the internal error amplifier. The voltage on this pin controls the peak switch current. Tie an RC network from this pin to ground to compensate the control loop. RT (Pin 10): Oscillator Resistor Input. Connecting a resistor to ground from this pin sets the switching frequency. Exposed Pad (Pin 11): Ground. The Exposed Pad must be soldered to PCB. 1913f 7 LT1913 BLOCK DIAGRAM VIN 4 VIN – + C1 INTERNAL 0.79V REF 5 10 RUN/SS ∑ SLOPE COMP BD SWITCH LATCH BOOST 2 C3 R RT OSCILLATOR 200kHz TO 2.4MHz RT Q S SW 6 1 SYNC L1 VOUT 3 C2 D1 SOFT-START 7 PG ERROR AMP + – + – 0.7V GND 11 FB VC CLAMP VC 9 CC RC CF 8 R2 R1 1913 BD 1913f 8 LT1913 OPERATION The LT1913 is a constant frequency, current mode stepdown regulator. An oscillator, with frequency set by RT, enables an RS flip-flop, turning on the internal power switch. An amplifier and comparator monitor the current flowing between the VIN and SW pins, turning the switch off when this current reaches a level determined by the voltage at VC. An error amplifier measures the output voltage through an external resistor divider tied to the FB pin and servos the VC pin. If the error amplifier’s output increases, more current is delivered to the output; if it decreases, less current is delivered. An active clamp on the VC pin provides current limit. The VC pin is also clamped to the voltage on the RUN/SS pin; soft-start is implemented by generating a voltage ramp at the RUN/SS pin using an external resistor and capacitor. An internal regulator provides power to the control circuitry. The bias regulator normally draws power from the VIN pin, but if the BD pin is connected to an external voltage higher than 3V bias power will be drawn from the external source (typically the regulated output voltage). This improves efficiency. The RUN/SS pin is used to place the LT1913 in shutdown, disconnecting the output and reducing the input current to less than 0.5μA. The switch driver operates from either the input or from the BOOST pin. An external capacitor and diode are used to generate a voltage at the BOOST pin that is higher than the input supply. This allows the driver to fully saturate the internal bipolar NPN power switch for efficient operation. The oscillator reduces the LT1913’s operating frequency when the voltage at the FB pin is low. This frequency foldback helps to control the output current during startup and overload. The LT1913 contains a power good comparator which trips when the FB pin is at 91% of its regulated value. The PG output is an open-collector transistor that is off when the output is in regulation, allowing an external resistor to pull the PG pin high. Power good is valid when the LT1913 is enabled and VIN is above 3.6V. 1913f 9 LT1913 APPLICATIONS INFORMATION FB Resistor Network The output voltage is programmed with a resistor divider between the output and the FB pin. Choose the 1% resistors according to: V R1= R2 OUT – 1 0.79V Reference designators refer to the Block Diagram. Setting the Switching Frequency The LT1913 uses a constant frequency PWM architecture that can be programmed to switch from 200kHz to 2.4MHz by using a resistor tied from the RT pin to ground. A table showing the necessary RT value for a desired switching frequency is in Figure 1. SWITCHING FREQUENCY (MHz) RT VALUE (kΩ) 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0 1.2 1.4 1.6 1.8 2.0 2.2 2.4 215 140 100 78.7 63.4 53.6 45.3 39.2 34 26.7 22.1 18.2 15 12.7 10.7 9.09 Figure 1. Switching Frequency vs. RT Value Operating Frequency Tradeoffs Selection of the operating frequency is a tradeoff between efficiency, component size, minimum dropout voltage, and maximum input voltage. The advantage of high frequency operation is that smaller inductor and capacitor values may be used. The disadvantages are lower efficiency, lower maximum input voltage, and higher dropout voltage. The highest acceptable switching frequency (fSW(MAX)) for a given application can be calculated as follows: fSW (MAX ) = VD + VOUT tON(MIN ) ( VD + VIN – VSW ) where VIN is the typical input voltage, VOUT is the output voltage, VD is the catch diode drop (~0.5V) and VSW is the internal switch drop (~0.5V at max load). This equation shows that slower switching frequency is necessary to safely accommodate high VIN/VOUT ratio. Also, as shown in the next section, lower frequency allows a lower dropout voltage. The reason input voltage range depends on the switching frequency is because the LT1913 switch has finite minimum on and off times. The switch can turn on for a minimum of ~150ns and turn off for a minimum of ~150ns. Typical minimum on time at 25°C is 80ns. This means that the minimum and maximum duty cycles are: DCMIN = fSW tON(MIN ) DCMAX = 1– fSW tOFF (MIN ) where fSW is the switching frequency, the tON(MIN) is the minimum switch on time (~150ns), and the tOFF(MIN) is the minimum switch off time (~150ns). These equations show that duty cycle range increases when switching frequency is decreased. A good choice of switching frequency should allow adequate input voltage range (see next section) and keep the inductor and capacitor values small. Input Voltage Range The maximum input voltage for LT1913 applications depends on switching frequency and Absolute Maximum Ratings of the VIN and BOOST pins (25V and 50V respectively). While the output is in start-up, short-circuit, or other overload conditions, the switching frequency should be chosen according to the following equation: VIN(MAX ) = VOUT + VD –V +V fSW tON(MIN ) D SW where VIN(MAX) is the maximum operating input voltage, VOUT is the output voltage, VD is the catch diode drop (~0.5V), VSW is the internal switch drop (~0.5V at max load), fSW is the switching frequency (set by RT), and tON(MIN) is the minimum switch on time (~100ns). Note that a higher switching frequency will depress the maximum 1913f 10 LT1913 APPLICATIONS INFORMATION operating input voltage. Conversely, a lower switching frequency will be necessary to achieve safe operation at high input voltages. If the output is in regulation and no short-circuit, startup, or overload events are expected, then input voltage transients of up to 25V are acceptable regardless of the switching frequency. In this mode, the LT1913 may enter pulse skipping operation where some switching pulses are skipped to maintain output regulation. In this mode the output voltage ripple and inductor current ripple will be higher than in normal operation. The minimum input voltage is determined by either the LT1913’s minimum operating voltage of ~3.6V or by its maximum duty cycle (see equation in previous section). The minimum input voltage due to duty cycle is: VIN(MIN ) = VOUT + VD –V +V 1– fSW tOFF (MIN ) D SW where VIN(MIN) is the minimum input voltage, and tOFF(MIN) is the minimum switch off time (150ns). Note that higher switching frequency will increase the minimum input voltage. If a lower dropout voltage is desired, a lower switching frequency should be used. Inductor Selection For a given input and output voltage, the inductor value and switching frequency will determine the ripple current. The ripple current ΔIL increases with higher VIN or VOUT and decreases with higher inductance and faster switching frequency. A reasonable starting point for selecting the ripple current is: ΔIL = 0.4(IOUT(MAX)) where IOUT(MAX) is the maximum output load current. To guarantee sufficient output current, peak inductor current must be lower than the LT1913’s switch current limit (ILIM). The peak inductor current is: IL(PEAK) = IOUT(MAX) + ΔIL/2 where IL(PEAK) is the peak inductor current, IOUT(MAX) is the maximum output load current, and ΔIL is the inductor ripple current. The LT1913’s switch current limit (ILIM) is 5.5A at low duty cycles and decreases linearly to 4.5A at DC = 0.8. The maximum output current is a function of the inductor ripple current: IOUT(MAX) = ILIM – ΔIL/2 Be sure to pick an inductor ripple current that provides sufficient maximum output current (IOUT(MAX)). The largest inductor ripple current occurs at the highest VIN. To guarantee that the ripple current stays below the specified maximum, the inductor value should be chosen according to the following equation: V +V V +V L = OUT D 1– OUT D VIN(MAX) fSW IL where VD is the voltage drop of the catch diode (~0.4V), VIN(MAX) is the maximum input voltage, VOUT is the output voltage, fSW is the switching frequency (set by RT), and L is in the inductor value. The inductor’s RMS current rating must be greater than the maximum load current and its saturation current should be about 30% higher. To keep the efficiency high, the series resistance (DCR) should be less than 0.05 , and the core material should be intended for high frequency applications. Table 1 lists several vendors and suitable types. Table 1. Inductor Vendors VENDOR URL PART SERIES TYPE Murata www.murata.com LQH55D Open TDK www.componenttdk.com SLF10145 Shielded Toko www.toko.com D75C D75F Shielded Open Sumida www.sumida.com CDRH74 CR75 CDRH8D43 Shielded Open Shielded NEC www.nec.com MPLC073 MPBI0755 Shielded Shielded Of course, such a simple design guide will not always result in the optimum inductor for your application. A larger value inductor provides a slightly higher maximum load current and will reduce the output voltage ripple. If your 1913f 11 LT1913 APPLICATIONS INFORMATION load is lower than 3.5A, then you can decrease the value of the inductor and operate with higher ripple current. This allows you to use a physically smaller inductor, or one with a lower DCR resulting in higher efficiency. There are several graphs in the Typical Performance Characteristics section of this data sheet that show the maximum load current as a function of input voltage and inductor value for several popular output voltages. Low inductance may result in discontinuous mode operation, which is okay but further reduces maximum load current. For details of maximum output current and discontinuous mode operation, see Linear Technology Application Note 44. Finally, for duty cycles greater than 50% (VOUT/VIN > 0.5), there is a minimum inductance required to avoid subharmonic oscillations. See AN19. Input Capacitor Bypass the input of the LT1913 circuit with a ceramic capacitor of X7R or X5R type. Y5V types have poor performance over temperature and applied voltage, and should not be used. A 10μF to 22μF ceramic capacitor is adequate to bypass the LT1913 and will easily handle the ripple current. Note that larger input capacitance is required when a lower switching frequency is used. If the input power source has high impedance, or there is significant inductance due to long wires or cables, additional bulk capacitance may be necessary. This can be provided with a lower performance electrolytic capacitor. Step-down regulators draw current from the input supply in pulses with very fast rise and fall times. The input capacitor is required to reduce the resulting voltage ripple at the LT1913 and to force this very high frequency switching current into a tight local loop, minimizing EMI. A 10μF capacitor is capable of this task, but only if it is placed close to the LT1913 and the catch diode (see the PCB Layout section). A second precaution regarding the ceramic input capacitor concerns the maximum input voltage rating of the LT1913. A ceramic input capacitor combined with trace or cable inductance forms a high quality (under damped) tank circuit. If the LT1913 circuit is plugged into a live supply, the input voltage can ring to twice its nominal value, possibly exceeding the LT1913’s voltage rating. This situation is easily avoided (see the Hot Plugging Safety section). For space sensitive applications, a 4.7μF ceramic capacitor can be used for local bypassing of the LT1913 input. However, the lower input capacitance will result in increased input current ripple and input voltage ripple, and may couple noise into other circuitry. Also, the increased voltage ripple will raise the minimum operating voltage of the LT1913 to ~3.7V. Output Capacitor and Output Ripple The output capacitor has two essential functions. Along with the inductor, it filters the square wave generated by the LT1913 to produce the DC output. In this role it determines the output ripple, and low impedance at the switching frequency is important. The second function is to store energy in order to satisfy transient loads and stabilize the LT1913’s control loop. Ceramic capacitors have very low equivalent series resistance (ESR) and provide the best ripple performance. A good starting value is: COUT = 100 VOUT fSW where fSW is in MHz, and COUT is the recommended output capacitance in μF. Use X5R or X7R types. This choice will provide low output ripple and good transient response. Transient performance can be improved with a higher value capacitor if the compensation network is also adjusted to maintain the loop bandwidth. A lower value of output capacitor can be used to save space and cost but transient performance will suffer. See the Frequency Compensation section to choose an appropriate compensation network. When choosing a capacitor, look carefully through the data sheet to find out what the actual capacitance is under operating conditions (applied voltage and temperature). A physically larger capacitor, or one with a higher voltage 1913f 12 LT1913 APPLICATIONS INFORMATION Table 2. Capacitor Vendors VENDOR PHONE URL PART SERIES Panasonic (714) 373-7366 www.panasonic.com Ceramic, COMMANDS Polymer, EEF Series Tantalum Kemet (864) 963-6300 www.kemet.com Ceramic, Tantalum Sanyo (408) 749-9714 www.sanyovideo.com T494, T495 Ceramic, Polymer, POSCAP Tantalum Murata (408) 436-1300 AVX www.murata.com Ceramic www.avxcorp.com Ceramic, Tantalum Taiyo Yuden (864) 963-6300 www.taiyo-yuden.com rating, may be required. High performance tantalum or electrolytic capacitors can be used for the output capacitor. Low ESR is important, so choose one that is intended for use in switching regulators. The ESR should be specified by the supplier, and should be 0.05 or less. Such a capacitor will be larger than a ceramic capacitor and will have a larger capacitance, because the capacitor must be large to achieve low ESR. Table 2 lists several capacitor vendors. Catch Diode The catch diode conducts current only during switch off time. Average forward current in normal operation can be calculated from: ID(AVG) = IOUT (VIN – VOUT)/VIN where IOUT is the output load current. The only reason to consider a diode with a larger current rating than necessary for nominal operation is for the worst-case condition of shorted output. The diode current will then increase to the TPS Series Ceramic typical peak switch current. Peak reverse voltage is equal to the regulator input voltage. Use a schottky diode with a reverse voltage rating greater than the input voltage. Table 3 lists several Schottky diodes and their manufacturers. Table 3. Diode Vendors PART NUMBER VR (V) IAVE (A) VF AT 3A (mV) On Semiconductor MBRA340 40 3 500 Diodes Inc. B330 B320 30 20 3 3 500 450 Frequency Compensation The LT1913 uses current mode control to regulate the output. This simplifies loop compensation. In particular, the LT1913 does not require the ESR of the output capacitor for stability, so you are free to use ceramic capacitors to achieve low output ripple and small circuit size. Frequency compensation is provided by the components tied to the 1913f 13 LT1913 APPLICATIONS INFORMATION Capacitor C3 and the internal boost Schottky diode (see the Block Diagram) are used to generate a boost voltage that is higher than the input voltage. In most cases a 0.47μF capacitor will work well. Figure 2 shows three ways to arrange the boost circuit. The BOOST pin must be LT1913 CURRENT MODE POWER STAGE gm = 5.3mho SW ERROR AMPLIFIER OUTPUT R1 CPL FB gm = 525μmho + Loop compensation determines the stability and transient performance. Designing the compensation network is a bit complicated and the best values depend on the application and in particular the type of output capacitor. A practical approach is to start with one of the circuits in this data sheet that is similar to your application and tune the compensation network to optimize the performance. Stability should then be checked across all operating conditions, including load current, input voltage and temperature. The LT1375 data sheet contains a more thorough discussion of loop compensation and describes how to test the stability using a transient load. Figure 2 shows an equivalent circuit for the LT1913 control loop. The error amplifier is a transconductance amplifier with finite output impedance. The power section, consisting of the modulator, power switch and inductor, is modeled as a transconductance amplifier generating an output current proportional to the voltage at the VC pin. Note that the output capacitor integrates this current, and that the capacitor on the VC pin (CC) integrates the error amplifier output current, resulting in two poles in the loop. In most cases a zero is required and comes from either the output capacitor ESR or from a resistor RC in series with CC. This simple model works well as long as the value of the inductor is not too high and the loop crossover frequency is much lower than the switching frequency. A phase lead capacitor (CPL) across the feedback divider may improve the transient response. Figure 3 shows the transient response when the load current is stepped from 1A to 3A and back to 1A. BOOST and BIAS Pin Considerations – VC pin, as shown in Figure 2. Generally a capacitor (CC) and a resistor (RC) in series to ground are used. In addition, there may be lower value capacitor in parallel. This capacitor (CF) is not part of the loop compensation but is used to filter noise at the switching frequency, and is required only if a phase-lead capacitor is used or if the output capacitor has high ESR. ESR 0.8V C1 + 3M C1 VC CF POLYMER OR TANTALUM GND RC CERAMIC R2 CC 1913 F02 Figure 2. Model for Loop Response VOUT 100mV/DIV IL 1A/DIV 10μs/DIV 1913 F03 Figure 3. Transient Load Response of the LT1913 Front Page Application as the Load Current is Stepped from 1A to 3A. VOUT = 5V 1913f 14 LT1913 APPLICATIONS INFORMATION VOUT BD BOOST VIN VIN LT1913 GND 4.7μF C3 SW (4a) For VOUT > 2.8V VOUT D2 BD BOOST VIN VIN LT1913 GND 4.7μF is more efficient because the BOOST pin current and BD pin quiescent current comes from a lower voltage source. You must also be sure that the maximum voltage ratings of the BOOST and BD pins are not exceeded. The minimum operating voltage of an LT1913 application is limited by the minimum input voltage (3.6V) and by the maximum duty cycle as outlined in a previous section. For proper startup, the minimum input voltage is also limited by the boost circuit. If the input voltage is ramped slowly, or the LT1913 is turned on with its RUN/SS pin when the output is already in regulation, then the boost capacitor may not be fully charged. Because the boost capacitor is C3 SW 6.0 (4b) For 2.5V < VOUT < 2.8V VOUT INPUT VOLTAGE (V) 5.5 BD TO START (WORST CASE) 5.0 4.5 4.0 TO RUN 3.5 3.0 VOUT = 3.3V TA = 25°C L = 8.2μH f = 600kHz BOOST VIN VIN LT1913 2.5 C3 2.0 4.7μF GND SW 10 1 100 1000 LOAD CURRENT (mA) 10000 8.0 1913 FO4 more than 2.3V above the SW pin for best efficiency. For outputs of 3V and above, the standard circuit (Figure 4a) is best. For outputs between 2.8V and 3V, use a 1μF boost capacitor. A 2.5V output presents a special case because it is marginally adequate to support the boosted drive stage while using the internal boost diode. For reliable BOOST pin operation with 2.5V outputs use a good external Schottky diode (such as the ON Semi MBR0540), and a 1μF boost capacitor (see Figure 4b). For lower output voltages the boost diode can be tied to the input (Figure 4c), or to another supply greater than 2.8V. The circuit in Figure 4a INPUT VOLTAGE (V) Figure 4. Three Circuits For Generating The Boost Voltage TO START (WORST CASE) 7.0 (4c) For VOUT < 2.5V 6.0 5.0 TO RUN 4.0 VOUT = 5V TA = 25°C L = 8.2μH f = 600kHz 3.0 2.0 1 10 100 1000 LOAD CURRENT (mA) 10000 1913 F05 Figure 5. The Minimum Input Voltage Depends on Output Voltage, Load Current and Boost Circuit 1913f 15 LT1913 APPLICATIONS INFORMATION charged with the energy stored in the inductor, the circuit will rely on some minimum load current to get the boost circuit running properly. This minimum load will depend on input and output voltages, and on the arrangement of the boost circuit. The minimum load generally goes to zero once the circuit has started. Figure 5 shows a plot of minimum load to start and to run as a function of input voltage. In many cases the discharged output capacitor will present a load to the switcher, which will allow it to start. The plots show the worst-case situation where VIN is ramping very slowly. For lower start-up voltage, the boost diode can be tied to VIN. At light loads, the inductor current becomes discontinuous and the effective duty cycle can be very high. This reduces the minimum input voltage to approximately 300mV above VOUT. At higher load currents, the inductor current is continuous and the duty cycle is limited by the maximum duty cycle of the LT1913, requiring a higher input voltage to maintain regulation. Soft-Start The RUN/SS pin can be used to soft-start the LT1913, reducing the maximum input current during start-up. The RUN/SS pin is driven through an external RC filter to create a voltage ramp at this pin. Figure 6 shows the startup and shut-down waveforms with the soft-start circuit. By choosing a large RC time constant, the peak start-up current can be reduced to the current that is required to regulate the output, with no overshoot. Choose the value of the resistor so that it can supply 20μA when the RUN/SS pin reaches 2.5V. Synchronization Synchronizing the LT1913 oscillator to an external frequency can be done by connecting a square wave (with 20% to 80% duty cycle) to the SYNC pin. The square wave amplitude should have valleys that are below 0.3V and peaks that are above 0.8V (up to 6V). The LT1913 may be synchronized over a 250kHz to 2MHz range. The RT resistor should be chosen to set the LT1913 switching frequency 20% below the lowest synchronization IL 1A/DIV RUN 15k RUN/SS 0.22μF VRUN/SS 2V/DIV GND VOUT 2V/DIV 2ms/DIV 1913 F06 Figure 6. To Soft-Start the LT1913, Add a Resisitor and Capacitor to the RUN/SS Pin input. For example, if the synchronization signal will be 250kHz and higher, the RT should be chosen for 200kHz. To assure reliable and safe operation the LT1913 will only synchronize when the output voltage is near regulation as indicated by the PG flag. It is therefore necessary to choose a large enough inductor value to supply the required output current at the frequency set by the RT resistor. See Inductor Selection section. It is also important to note that slope compensation is set by the RT value: When the sync frequency is much higher than the one set by RT, the slope compensation will be significantly reduced which may require a larger inductor value to prevent subharmonic oscillation. Shorted and Reversed Input Protection If the inductor is chosen so that it won’t saturate excessively, an LT1913 buck regulator will tolerate a shorted output. There is another situation to consider in systems where the output will be held high when the input to the LT1913 is absent. This may occur in battery charging applications or in battery backup systems where a battery or some other supply is diode OR-ed with the LT1913’s output. If the VIN pin is allowed to float and the RUN/SS pin is held high (either by a logic signal or because it is tied to VIN), then the LT1913’s internal circuitry will pull its quiescent current through its SW pin. This is fine if your system can tolerate a few mA in this state. If you ground the RUN/SS pin, the SW pin current will drop to essentially zero. However, if the VIN pin is grounded while 1913f 16 LT1913 APPLICATIONS INFORMATION the output is held high, then parasitic diodes inside the LT1913 can pull large currents from the output through the SW pin and the VIN pin. Figure 7 shows a circuit that will run only when the input voltage is present and that protects against a shorted or reversed input. The Exposed Pad on the bottom of the package must be soldered to ground so that the pad acts as a heat sink. To keep thermal resistance low, extend the ground plane as much as possible, and add thermal vias under and near the LT1913 to additional ground planes within the circuit board and on the bottom side. PCB Layout For proper operation and minimum EMI, care must be taken during printed circuit board layout. Figure 8 shows the recommended component placement with trace, ground plane and via locations. Note that large, switched currents flow in the LT1913’s VIN and SW pins, the catch diode (D1) and the input capacitor (C1). The loop formed by these components should be as small as possible. These components, along with the inductor and output capacitor, should be placed on the same side of the circuit board, and their connections should be made on that layer. Place a local, unbroken ground plane below these components. The SW and BOOST nodes should be as small as possible. Finally, keep the FB and VC nodes small so that the ground traces will shield them from the SW and BOOST nodes. Hot Plugging Safely The small size, robustness and low impedance of ceramic capacitors make them an attractive option for the input bypass capacitor of LT1913 circuits. However, these capacitors can cause problems if the LT1913 is plugged into a live supply (see Linear Technology Application Note 88 for a complete discussion). The low loss ceramic capacitor, combined with stray inductance in series with the power source, forms an under damped tank circuit, and the voltage at the VIN pin of the LT1913 can ring to twice the nominal input voltage, possibly exceeding the LT1913’s rating and damaging the part. If the input supply is poorly controlled or the user will be plugging the LT1913 into an energized supply, the input network should be designed D4 MBRS140 VIN VIN L1 BOOST C2 VOUT LT1913 RUN/SS VOUT SW RRT VC CC GND FB BACKUP RC R2 1913 F07 Figure 7. Diode D4 Prevents a Shorted Input from Discharging a Backup Battery Tied to the Output. It Also Protects the Circuit from a Reversed Input. The LT1913 Runs Only When the Input is Present R1 D1 C1 GND RPG 1913 F08 VIAS TO LOCAL GROUND PLANE VIAS TO VOUT VIAS TO SYNC VIAS TO RUN/SS VIAS TO PG VIAS TO VIN OUTLINE OF LOCAL GROUND PLANE Figure 8. A Good PCB Layout Ensures Proper, Low EMI Operation 1913f 17 LT1913 APPLICATIONS INFORMATION to prevent this overshoot. Figure 9 shows the waveforms that result when an LT1913 circuit is connected to a 24V supply through six feet of 24-gauge twisted pair. The first plot is the response with a 4.7μF ceramic capacitor at the input. The input voltage rings as high as 50V and the input current peaks at 26A. A good solution is shown in Figure 9b. A 0.7 resistor is added in series with the input to eliminate the voltage overshoot (it also reduces the peak input current). A 0.1μF capacitor improves high frequency filtering. For high input voltages its impact on efficiency is minor, reducing efficiency by 1.5 percent for a 5V output at full load operating from 24V. CLOSING SWITCH SIMULATES HOT PLUG IIN VIN High Temperature Considerations The PCB must provide heat sinking to keep the LT1913 cool. The Exposed Pad on the bottom of the package must be soldered to a ground plane. This ground should be tied to large copper layers below with thermal vias; these layers will spread the heat dissipated by the LT1913. Place additional vias can reduce thermal resistance further. With these steps, the thermal resistance from die (or junction) to ambient can be reduced to JA = 35°C/W or less. With 100 LFPM airflow, this resistance can fall by another 25%. Further increases in airflow will lead to lower thermal re- DANGER VIN 20V/DIV RINGING VIN MAY EXCEED ABSOLUTE MAXIMUM RATING LT1913 + 4.7μF LOW IMPEDANCE ENERGIZED 24V SUPPLY IIN 10A/DIV STRAY INDUCTANCE DUE TO 6 FEET (2 METERS) OF TWISTED PAIR 20μs/DIV (9a) 0.7Ω LT1913 VIN 20V/DIV + 0.1μF 4.7μF IIN 10A/DIV (9b) LT1913 + 22μF 35V AI.EI. 20μs/DIV VIN 20V/DIV + 4.7μF IIN 10A/DIV (9c) 20μs/DIV 1913 F09 Figure 9. A Well Chosen Input Network Prevents Input Voltage Overshoot and Ensures Reliable Operation when the LT1913 is Connected to a Live Supply 1913f 18 LT1913 APPLICATIONS INFORMATION sistance. Because of the large output current capability of the LT1913, it is possible to dissipate enough heat to raise the junction temperature beyond the absolute maximum of 125°C. When operating at high ambient temperatures, the maximum load current should be derated as the ambient temperature approaches 125°C. Power dissipation within the LT1913 can be estimated by calculating the total power loss from an efficiency measurement and subtracting the catch diode loss and inductor loss. The die temperature is calculated by multiplying the LT1913 power dissipation by the thermal resistance from junction to ambient. Other Linear Technology Publications Application Notes 19, 35 and 44 contain more detailed descriptions and design information for buck regulators and other switching regulators. The LT1376 data sheet has a more extensive discussion of output ripple, loop compensation and stability testing. Design Note 100 shows how to generate a bipolar output supply using a buck regulator. TYPICAL APPLICATIONS 5V Step-Down Converter VOUT 5V 3.5A VIN 6.5V TO 25V VIN BD RUN/SS ON OFF BOOST 0.47μF VC 10μF LT1913 SW D RT 15k L 4.7μH PG SYNC 63.4k 680pF f = 600kHz 536k GND FB 47μF 100k 1913 TA02 D: ON SEMI MBRA340 L: NEC MPLC0730L4R7 1913f 19 LT1913 TYPICAL APPLICATIONS 3.3V Step-Down Converter VOUT 3.3V 3.5A VIN 4.8V TO 25V VIN BD RUN/SS ON OFF BOOST L 3.3μH 0.47μF VC 4.7μF SW LT1913 D RT 19k PG SYNC 63.4k 316k FB GND 680pF 47μF 100k f = 600kHz 1913 TA03 D: ON SEMI MBRA340 L: NEC MPLC0730L3R3 2.5V Step-Down Converter VOUT 2.5V 3.5A VIN 4V TO 25V VIN BD RUN/SS ON OFF D2 BOOST 1μF VC 4.7μF LT1913 SW D1 RT 15.4k L 3.3μH PG 215k SYNC 63.4k 680pF f = 600kHz GND FB 47μF 100k 1913 TA04 D1: ON SEMI MBRA340 D2: MBR0540 L: NEC MPLC0730L3R3 1913f 20 LT1913 TYPICAL APPLICATIONS 5V, 2MHz Step-Down Converter VOUT 5V 2.5A VIN 8.6V TO 22V VIN BD RUN/SS ON OFF BOOST L 2.2μH 0.47μF VC 4.7μF LT1913 SW D RT 15k PG 536k SYNC 12.7k FB GND 680pF 22μF 100k f = 2MHz 1913 TA05 D: ON SEMI MBRA340 L: NEC MPLC0730L2R2 12V Step-Down Converter VOUT 12V 3.5A VIN 15V TO 25V VIN BD RUN/SS ON OFF BOOST 0.47μF VC 10μF LT1913 SW D RT 17.4k L 8.2μH PG 715k SYNC 63.4k GND 680pF f = 600kHz FB 47μF 50k 1913 TA06 D: ON SEMI MBRA340 L: NEC MBP107558R2P 1913f 21 LT1913 TYPICAL APPLICATIONS 1.8V Step-Down Converter VOUT 1.8V 3.5A VIN 3.6V TO 25V VIN BD RUN/SS ON OFF BOOST 0.47μF VC 4.7μF LT1913 SW D RT 16.9k L 3.3μH PG SYNC 78.7k 680pF f = 500kHz 127k GND FB 47μF 100k 1913 TA08 D: ON SEMI MBRA340 L: NEC MPLC0730L3R3 1913f 22 LT1913 PACKAGE DESCRIPTION DD Package 10-Lead Plastic DFN (3mm × 3mm) (Reference LTC DWG # 05-08-1699) 0.675 ±0.05 3.50 ±0.05 1.65 ±0.05 2.15 ±0.05 (2 SIDES) PACKAGE OUTLINE 0.25 ± 0.05 0.50 BSC 2.38 ±0.05 (2 SIDES) RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS 3.00 ±0.10 (4 SIDES) R = 0.115 TYP 6 0.38 ± 0.10 10 1.65 ± 0.10 (2 SIDES) PIN 1 TOP MARK (SEE NOTE 6) (DD) DFN 1103 5 0.200 REF 1 0.25 ± 0.05 0.50 BSC 0.75 ±0.05 0.00 – 0.05 2.38 ±0.10 (2 SIDES) BOTTOM VIEW—EXPOSED PAD NOTE: 1. DRAWING TO BE MADE A JEDEC PACKAGE OUTLINE M0-229 VARIATION OF (WEED-2). CHECK THE LTC WEBSITE DATA SHEET FOR CURRENT STATUS OF VARIATION ASSIGNMENT 2. DRAWING NOT TO SCALE 3. ALL DIMENSIONS ARE IN MILLIMETERS 4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE 5. EXPOSED PAD SHALL BE SOLDER PLATED 6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON THE TOP AND BOTTOM OF PACKAGE 1913f 23 LT1913 U TYPICAL APPLICATIO 1.2V Step-Down Converter VOUT 1.2V 3.5A VIN 3.6V TO 25V VIN BD RUN/SS ON OFF BOOST 0.47μF VC 4.7μF LT1913 SW D RT 17k L 3.3μH PG 52.3k SYNC 78.7k GND 470pF FB 100k 100μF f = 500kHz 1913 TA09 D: ON SEMI MBRA340 L: NEC MPLC0730L3R3 RELATED PARTS PART NUMBER DESCRIPTION COMMENTS LT1766 60V, 1.2A (IOUT), 200kHz, High Efficiency Step-Down DC/DC Converter VIN: 5.5V to 60V, VOUT(MIN) = 1.2V, IQ = 2.5mA, ISD = 25μA, TSSOP16/E Package LT1933 500mA (IOUT), 500kHz Step-Down Switching Regulator in SOT-23 VIN: 3.6V to 36V, VOUT(MIN) = 1.2V, IQ = 1.6mA, ISD < 1μA, ThinSOTTM Package LT1936 36V, 1.4A (IOUT), 500kHz, High Efficiency Step-Down DC/DC Converter VIN: 3.6V to 36V, VOUT(MIN) = 1.2V, IQ = 1.9mA, ISD < 1μA, MS8E Package LT1940 Dual 25V, 1.4A (IOUT), 1.1MHz, High Efficiency Step-Down DC/DC Converter VIN: 3.6V to 25V, VOUT(MIN) = 1.2V, IQ = 3.8mA, ISD < 30μA, TSSOP16E Package LT1976/LT1967 60V, 1.2A (IOUT), 200kHz/500kHz, High Efficiency Step-Down DC/DC Converters with Burst Mode Operation VIN: 3.3V to 60V, VOUT(MIN) = 1.2V, IQ = 100μA, ISD < 1μA, TSSOP16E Package LT3434/LT3435 60V, 2.4A (IOUT), 200kHz/500kHz, High Efficiency Step-Down DC/DC Converters with Burst Mode Operation VIN: 3.3V to 60V, VOUT(MIN) = 1.2V, IQ = 100μA, ISD < 1μA, TSSOP16 Package LT3437 60V, 400mA (IOUT), Micropower Step-Down DC/DC Converter with Burst Mode Operation VIN: 3.3V to 60V, VOUT(MIN) = 1.25V, IQ = 100μA, ISD < 1μA, 3mm × 3mm DFN10 and TSSOP16E Packages LT3480 36V with Transient Protection to 60V, 2A (IOUT), 2.4MHz, High Efficiency Step-Down DC/DC Converter with Burst Mode Operation VIN: 3.6V to 38V, VOUT(MIN) = 0.78V, IQ = 70μA, ISD < 1μA, 3mm × 3mm DFN10 and MSOP10E Packages LT3481 34V with Transient Protection to 36V, 2A (IOUT), 2.8MHz, High Efficiency Step-Down DC/DC Converter with Burst Mode Operation VIN: 3.6V to 34V, VOUT(MIN) = 1.26V, IQ = 50μA, ISD < 1μA, 3mm × 3mm DFN10 and MSOP10E Packages LT3493 36V, 1.4A (IOUT), 750kHz High Efficiency Step-Down DC/DC Converter VIN: 3.6V to 36V, VOUT(MIN) = 0.8V, IQ = 1.9mA, ISD < 1μA, 2mm × 3mm DFN6 Package LT3505 36V with Transient Protection to 40V, 1.4A (IOUT), 3MHz, High Efficiency Step-Down DC/DC Converter VIN: 3.6V to 34V, VOUT(MIN) = 0.78V, IQ = 2mA, ISD = 2μA, 3mm × 3mm DFN8 and MSOP8E Packages LT3508 36V with Transient Protection to 40V, Dual 1.4A (IOUT), 3MHz, High Efficiency Step-Down DC/DC Converter VIN: 3.7V to 37V, VOUT(MIN) = 0.8V, IQ = 4.6mA, ISD = 1μA, 4mm × 4mm QFN24 and TSSOP16E Packages LT3680 36V, 3.5A(IOUT), 2.4MHz High Efficiency Step-Down DC/DC Converter VIN: 3.6V to 36V, VOUT(MIN) = 0.79V, IQ = 75μA, ISD < 1μA, 3mm × 3mm DFN, MSOP10E LT3684 34V with Transient Protection to 36V, 2A (IOUT), 2.8MHz, High Efficiency Step-Down DC/DC Converter VIN: 3.6V to 34V, VOUT(MIN) = 1.26V, IQ = 850μA, ISD < 1μA, 3mm × 3mm DFN10 and MSOP10E Packages LT3685 36V with Transient Protection to 60V, Dual 2A (IOUT), 2.4MHz, High Efficiency Step-Down DC/DC Converter VIN: 3.6V to 38V, VOUT(MIN) = 0.78V, IQ = 70μA, ISD < 1μA, 3mm × 3mm DFN10 and MSOP10E Packages 1913f 24 Linear Technology Corporation LT 1207 • PRINTED IN USA 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408) 432-1900 ● FAX: (408) 434-0507 ● www.linear.com © LINEAR TECHNOLOGY CORPORATION 2007