LINER LT1913EDD

LT1913
25V, 3.5A, 2.4MHz
Step-Down Switching Regulator
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DESCRIPTIO
FEATURES
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The LT®1913 is an adjustable frequency (200kHz to
2.4MHz) monolithic buck switching regulator that accepts
input voltages up to 25V. A high efficiency 95m switch
is included on the die along with a boost Schottky diode
and the necessary oscillator, control, and logic circuitry.
Current mode topology is used for fast transient response
and good loop stability. Shutdown reduces input supply
current to less than 1μA while a resistor and capacitor on
the RUN/SS pin provide a controlled output voltage ramp
(soft-start). A power good flag signals when VOUT reaches
91% of the programmed output voltage. The LT1913 is
available in 10-Pin 3mm × 3mm DFN packages with exposed pads for low thermal resistance.
Wide Input Range: 3.6V to 25V
3.5A Maximum Output Current
Adjustable Switching Frequency: 200kHz to 2.4MHz
Low Shutdown Current: IQ < 1μA
Integrated Boost Diode
Synchronizable Between 250kHz to 2MHz
Power Good Flag
Saturating Switch Design: 95m On-Resistance
0.790V Feedback Reference Voltage
Output Voltage: 0.79V to 25V
Thermal Protection
Soft-Start Capability
Small 10-Pin (3mm × 3mm) DFN Packages
, LT, LTC and LTM are registered trademarks of Linear Technology Corporation.
All other trademarks are the property of their respective owners.
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APPLICATIO S
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Automotive Battery Regulation
Power for Portable Products
Distributed Supply Regulation
Industrial Supplies
Wall Transformer Regulation
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TYPICAL APPLICATIO
5V Step-Down Converter
Efficiency
VOUT
5V
3.5A
VIN
6.5V TO 25V
VIN = 12V
BD
VIN
OFF ON
RUN/SS
90
BOOST
EFFICIENCY (%)
0.47μF
4.7μH
15k
VC
10μF
LT1913
SW
RT
680pF
100
80
VIN = 24V
70
PG
63.4k
SYNC
60
536k
GND
VOUT = 5V
L = 4.7μH
f = 600kHz
FB
47μF
100k
50
0
0.5
2
1.5
1
2.5
OUTPUT CURRENT (A)
3
3.5
1913 TA01a
1913 G01
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LT1913
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ABSOLUTE
AXI U RATI GS
(Note 1)
VIN, RUN/SS Voltage .................................................25V
BOOST Pin Voltage ...................................................50V
BOOST Pin Above SW Pin.........................................25V
FB, RT, VC Voltage .......................................................5V
PG, BD Voltage .........................................................25V
SYNC Voltage ............................................................20V
Operating Junction Temperature Range (Note 2)
LT1913E ............................................. –40°C to 125°C
LT1913I .............................................. –40°C to 125°C
Storage Temperature Range................... –65°C to 150°C
PIN CONFIGURATION
TOP VIEW
10 RT
BD
1
BOOST
2
SW
3
VIN
4
7 PG
RUN/SS
5
6 SYNC
9 VC
11
8 FB
DD PACKAGE
10-LEAD (3mm × 3mm) PLASTIC DFN
θJA = 45°C/W, θJC = 10°C/W
EXPOSED PAD (PIN 11) IS GND, MUST BE SOLDERED TO PCB
ORDER INFORMATION
LEAD FREE FINISH
TAPE AND REEL
PART MARKING*
PACKAGE DESCRIPTION
TEMPERATURE RANGE
LT1913EDD#PBF
LT1913EDD#TRPBF
LDJW
10-Lead (3mm × 3mm) Plastic DFN
–40°C to 125°C
LT1913IDD#PBF
LT1913IDD#TRPBF
LDJW
10-Lead (3mm × 3mm) Plastic DFN
–40°C to 125°C
Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container.
Consult LTC Marketing for information on non-standard lead based finish parts.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
ELECTRICAL CHARACTERISTICS
The ● denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 10V, VRUN/SS = 10V, VBOOST = 15V, VBD = 3.3V unless otherwise
noted. (Note 2)
PARAMETER
CONDITIONS
●
Minimum Input Voltage
Quiescent Current from VIN
VRUN/SS = 0.2V
VBD = 3V, Not Switching
Quiescent Current from BD
●
TYP
MAX
UNITS
3
3.6
V
0.01
0.5
μA
0.45
1.2
mA
VBD = 0, Not Switching
1.3
2.3
mA
VRUN/SS = 0.2V
0.01
0.5
μA
0.9
1.8
mA
1
10
μA
2.7
3
V
VBD = 3V, Not Switching
VBD = 0, Not Switching
Minimum Bias Voltage (BD Pin)
MIN
●
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LT1913
ELECTRICAL CHARACTERISTICS
The ● denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 10V, VRUN/SS = 10V, VBOOST = 15V, VBD = 3.3V unless otherwise
noted. (Note 2)
PARAMETER
CONDITIONS
Feedback Voltage
●
FB Pin Bias Current (Note 3)
VFB = 0.8V, VC = 0.4V
FB Voltage Line Regulation
4V < VIN < 25V
MIN
TYP
MAX
UNITS
780
775
790
790
800
805
mV
mV
10
40
nA
0.002
0.01
%/V
●
Error Amp gm
525
Error Amp Gain
2000
μMho
VC Source Current
60
μA
VC Sink Current
60
μA
VC Pin to Switch Current Gain
5.3
A/V
VC Clamp Voltage
2.0
V
Switching Frequency
RT = 8.66k
RT = 29.4k
RT = 187k
2.2
1.0
200
●
Minimum Switch Off-Time
4.6
2.45
1.1
230
2.7
1.25
260
MHz
MHz
kHz
60
150
nS
5.4
6.0
A
Switch Current Limit
Duty Cycle = 5%
Switch VCESAT
ISW = 3.5A
335
Boost Schottky Reverse Leakage
VSW = 10V, VBD = 0V
0.02
2
●
mV
μA
1.5
2.0
V
BOOST Pin Current
ISW = 1A
35
60
mA
RUN/SS Pin Current
VRUN/SS = 2.5V
5
8
μA
2.5
V
Minimum Boost Voltage (Note 4)
RUN/SS Input Voltage High
RUN/SS Input Voltage Low
PG Threshold Offset from Feedback Voltage
0.2
VFB Rising
PG Hysteresis
PG Leakage
VPG = 5V
PG Sink Current
VPG = 0.4V
SYNC Low Threshold
V
65
mV
10
mV
0.1
●
200
800
V
0.8
VSYNC = 0V
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: The LT1913E is guaranteed to meet performance specifications
from 0°C to 125°C. Specifications over the –40°C to 125°C operating
temperature range are assured by design, characterization and correlation
with statistical process controls. The LT1913I specifications are
guaranteed over the –40°C to 125°C temperature range.
μA
μA
0.5
SYNC High Threshold
SYNC Pin Bias Current
1
0.1
V
μA
Note 3: Bias current flows out of the FB pin.
Note 4: This is the minimum voltage across the boost capacitor needed to
guarantee full saturation of the switch.
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LT1913
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TYPICAL PERFOR A CE CHARACTERISTICS
Efficiency
Efficiency
Efficiency
100
100
VIN = 12V
EFFICIENCY (%)
80
VIN = 24V
70
60
50
0
0.5
80
VIN = 24V
70
60
VOUT = 5V
L = 4.7μH
f = 600kHz
2
1.5
1
2.5
OUTPUT CURRENT (A)
3
3.5
EFFICIENCY (%)
VIN = 12V
90
50
0.5
2
1.5
1
2.5
OUTPUT CURRENT (A)
3
1913 G01
90
2.5
80
2.0
70
1.5
VIN = 12V
VOUT = 5V
L = 4.7μH
f = 600kHz
50
3.5
0
LOAD CURRENT (A)
3.5
VOUT = 3.3V
TA = 25°C
L = 4.7μH
f = 600kHz
3.0
5
10
4.5
MINIMUM
4.0
VOUT = 5V
TA = 25°C
L = 4.7μH
f = 600kHz
3.5
2.5
20
15
INPUT VOLTAGE (V)
SWITCH CURRENT LIMIT(A)
TYPICAL
5.0
4.0
20
15
INPUT VOLTAGE (V)
1913 G06
6.0
4.0
3.5
120
600
105
500
400
300
200
100
25 50 75 100 125 150
TEMPERATURE (°C)
1913 G09
60
40
DUTY CYCLE (%)
80
100
Boost Pin Current
700
3.0
2.5
20
1913 G08
BOOST PIN CURRENT (mA)
VOLTAGE DROP (mV)
DUTY CYCLE = 90 %
0
0
25
DUTY CYCLE = 10 %
5.0
2.0
–50 –25
4.0
Switch Voltage Drop
6.5
4.5
4.5
1913 G07
Switch Current Limit
5.5
5.0
3.0
10
5
5.5
3.5
3.0
25
0.5
Switch Current Limit
TYPICAL
MINIMUM
3.5
3
6.0
5.5
4.5
2
1.5
1
2.5
OUTPUT CURRENT (A)
1913 G03
Maximum Load Current
Maximum Load Current
5.0
0.5
1.0
1913 G02
5.5
LOAD CURRENT (A)
3.0
60
VOUT = 3.3V
L = 3.3μH
f = 600kHz
0
100
TOTAL POWER LOSS (W)
EFFICIENCY (%)
90
SWITCH CURRENT LIMIT (A)
TA = 25°C unless otherwise noted.
90
75
60
45
30
15
0
0
1
2
4
3
SWITCH CURRENT (A)
5
1913 G10
0
0
1
2
3
SWITCH CURRENT (A)
4
5
1913 G11
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LT1913
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TYPICAL PERFOR A CE CHARACTERISTICS
TA = 25°C unless otherwise noted.
Switching Frequency
Feedback Voltage
840
Frequency Foldback
1200
1.20
RT = 34.0k
RT = 34.0k
SWITCHING FREQUENCY (kHz)
820
1.10
FREQUENCY (MHz)
FEEDBACK VOLTAGE (mV)
1.15
800
780
1.05
1.00
0.95
0.90
1000
0.85
760
–50 –25
0
0.80
–50 –25
25 50 75 100 125 150
TEMPERATURE (°C)
25 50 75 100 125 150
TEMPERATURE (°C)
6
100
80
60
40
100 200 300 400 500 600 700 800 900
FB PIN VOLTAGE (mV)
0
1913 G14
RUN/SS Pin Current
12
RUN/SS PIN CURRENT (μA)
120
SWITCH CURRENT LIMIT (A)
MINIMUM SWITCH ON TIME (ns)
7
5
4
3
2
0
0
25 50 75 100 125 150
TEMPERATURE (°C)
10
8
6
4
2
1
20
0.5
2.5
2
1.5
RUN/SS PIN VOLTAGE (V)
1
3
0
3.5
0
5
20
15
10
RUN/SS PIN VOLTAGE (V)
1913 G16
1913 G15
Minimum Input Voltage
50
1.4
25
1913 G17
Error Amp Output Current
Boost Diode
5.0
40
1.2
4.5
1.0
0.8
0.6
0.4
20
INPUT VOLTAGE (V)
VC PIN CURRENT (μA)
30
BOOST DIODE VF (V)
200
Soft-Start
Minimum Switch On-Time
10
0
–10
–20
–30
0.2
0
400
1913 G13
140
0
600
0
0
1913 G12
0
–50 –25
800
4.0
3.5
3.0
2.5
–40
0
0.5
1.0
1.5
BOOST DIODE CURRENT (A)
2.0
1913 G18
–50
–200
2.0
–100
100
0
FB PIN ERROR VOLTAGE (mV)
200
1913 G19
VOUT = 3.3V
TA = 25°C
L = 4.7μH
f = 600kHz
1
10
100
1000
LOAD CURRENT (mA)
10000
1913 G20
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LT1913
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TYPICAL PERFOR A CE CHARACTERISTICS
VC Voltages
6.5
2.50
6.0
2.00
Power Good Threshold
95
THRESHOLD VOLTAGE (%)
Minimum Input Voltage
CURRENT LIMIT CLAMP
VC VOLTAGE (V)
INPUT VOLTAGE (V)
TA = 25°C unless otherwise noted.
5.5
5.0
VOUT = 5V
TA = 25 °C
L = 4.7μH
f = 600kHz
4.5
1.00
SWITCHING THRESHOLD
85
80
0.50
4.0
1
1.50
90
10
100
1000
LOAD CURRENT (mA)
10000
0
–50 –25
0
25 50 75 100 125 150
TEMPERATURE (°C)
75
–50 –25
0
25 50 75 100 125 150
TEMPERATURE (°C)
1913 G22
1913 G21
Switching Waveforms;
Discontinuous Operation
1913 G23
Switching Waveforms;
Continuous Operation
VSW
5V/DIV
VSW
5V/DIV
IL
0.2A/DIV
IL
0.5A/DIV
VOUT
10mV/DIV
VOUT
10mV/DIV
VIN = 12V
VOUT = 3.3V
ILOAD = 110mA
1μs/DIV
1913 G25
VIN = 12V
VOUT = 3.3V
ILOAD = 1A
1μs/DIV
1913 G26
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LT1913
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PI FU CTIO S
BD (Pin 1): This pin connects to the anode of the boost
Schottky diode. BD also supplies current to the internal
regulator.
BOOST (Pin 2): This pin is used to provide a drive
voltage, higher than the input voltage, to the internal bipolar
NPN power switch.
SW (Pin 3): The SW pin is the output of the internal power
switch. Connect this pin to the inductor, catch diode and
boost capacitor.
VIN (Pin 4): The VIN pin supplies current to the LT1913’s
internal regulator and to the internal power switch. This
pin must be locally bypassed.
RUN/SS (Pin 5): The RUN/SS pin is used to put the
LT1913 in shutdown mode. Tie to ground to shut down
the LT1913. Tie to 2.5V or more for normal operation. If
the shutdown feature is not used, tie this pin to the VIN
pin. RUN/SS also provides a soft-start function; see the
Applications Information section.
SYNC (Pin 6): This is the external clock synchronization
input. Ground this pin when not used. Tie to a clock source
for synchronization. Clock edges should have rise and
fall times faster than 1μs. Do not leave pin floating. See
synchronizing section in Applications Information.
PG (Pin 7): The PG pin is the open collector output of an
internal comparator. PG remains low until the FB pin is
within 9% of the final regulation voltage. PG output is valid
when VIN is above 3.6V and RUN/SS is high.
FB (Pin 8): The LT1913 regulates the FB pin to 0.790V.
Connect the feedback resistor divider tap to this pin.
VC (Pin 9): The VC pin is the output of the internal error
amplifier. The voltage on this pin controls the peak switch
current. Tie an RC network from this pin to ground to
compensate the control loop.
RT (Pin 10): Oscillator Resistor Input. Connecting a resistor
to ground from this pin sets the switching frequency.
Exposed Pad (Pin 11): Ground. The Exposed Pad must
be soldered to PCB.
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LT1913
BLOCK DIAGRAM
VIN
4
VIN
–
+
C1
INTERNAL 0.79V REF
5
10
RUN/SS
∑
SLOPE COMP
BD
SWITCH
LATCH
BOOST
2
C3
R
RT
OSCILLATOR
200kHz TO 2.4MHz
RT
Q
S
SW
6
1
SYNC
L1
VOUT
3
C2
D1
SOFT-START
7
PG
ERROR AMP
+
–
+
–
0.7V
GND
11
FB
VC CLAMP
VC
9
CC
RC
CF
8
R2
R1
1913 BD
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LT1913
OPERATION
The LT1913 is a constant frequency, current mode stepdown regulator. An oscillator, with frequency set by RT,
enables an RS flip-flop, turning on the internal power
switch. An amplifier and comparator monitor the current
flowing between the VIN and SW pins, turning the switch
off when this current reaches a level determined by the
voltage at VC. An error amplifier measures the output
voltage through an external resistor divider tied to the FB
pin and servos the VC pin. If the error amplifier’s output
increases, more current is delivered to the output; if it
decreases, less current is delivered. An active clamp on the
VC pin provides current limit. The VC pin is also clamped to
the voltage on the RUN/SS pin; soft-start is implemented
by generating a voltage ramp at the RUN/SS pin using an
external resistor and capacitor.
An internal regulator provides power to the control circuitry.
The bias regulator normally draws power from the VIN pin,
but if the BD pin is connected to an external voltage higher
than 3V bias power will be drawn from the external source
(typically the regulated output voltage). This improves
efficiency. The RUN/SS pin is used to place the LT1913
in shutdown, disconnecting the output and reducing the
input current to less than 0.5μA.
The switch driver operates from either the input or from
the BOOST pin. An external capacitor and diode are used
to generate a voltage at the BOOST pin that is higher than
the input supply. This allows the driver to fully saturate
the internal bipolar NPN power switch for efficient operation.
The oscillator reduces the LT1913’s operating frequency
when the voltage at the FB pin is low. This frequency
foldback helps to control the output current during startup
and overload.
The LT1913 contains a power good comparator which trips
when the FB pin is at 91% of its regulated value. The PG
output is an open-collector transistor that is off when the
output is in regulation, allowing an external resistor to pull
the PG pin high. Power good is valid when the LT1913 is
enabled and VIN is above 3.6V.
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LT1913
APPLICATIONS INFORMATION
FB Resistor Network
The output voltage is programmed with a resistor divider
between the output and the FB pin. Choose the 1% resistors according to:
V
R1= R2 OUT – 1
0.79V Reference designators refer to the Block Diagram.
Setting the Switching Frequency
The LT1913 uses a constant frequency PWM architecture
that can be programmed to switch from 200kHz to 2.4MHz
by using a resistor tied from the RT pin to ground. A table
showing the necessary RT value for a desired switching
frequency is in Figure 1.
SWITCHING FREQUENCY (MHz)
RT VALUE (kΩ)
0.2
0.3
0.4
0.5
0.6
0.7
0.8
0.9
1.0
1.2
1.4
1.6
1.8
2.0
2.2
2.4
215
140
100
78.7
63.4
53.6
45.3
39.2
34
26.7
22.1
18.2
15
12.7
10.7
9.09
Figure 1. Switching Frequency vs. RT Value
Operating Frequency Tradeoffs
Selection of the operating frequency is a tradeoff between
efficiency, component size, minimum dropout voltage, and
maximum input voltage. The advantage of high frequency
operation is that smaller inductor and capacitor values may
be used. The disadvantages are lower efficiency, lower
maximum input voltage, and higher dropout voltage. The
highest acceptable switching frequency (fSW(MAX)) for a
given application can be calculated as follows:
fSW (MAX ) =
VD + VOUT
tON(MIN ) ( VD + VIN – VSW )
where VIN is the typical input voltage, VOUT is the output
voltage, VD is the catch diode drop (~0.5V) and VSW is the
internal switch drop (~0.5V at max load). This equation
shows that slower switching frequency is necessary to
safely accommodate high VIN/VOUT ratio. Also, as shown
in the next section, lower frequency allows a lower dropout
voltage. The reason input voltage range depends on the
switching frequency is because the LT1913 switch has finite
minimum on and off times. The switch can turn on for a
minimum of ~150ns and turn off for a minimum of ~150ns.
Typical minimum on time at 25°C is 80ns. This means that
the minimum and maximum duty cycles are:
DCMIN = fSW tON(MIN )
DCMAX = 1– fSW tOFF (MIN )
where fSW is the switching frequency, the tON(MIN) is the
minimum switch on time (~150ns), and the tOFF(MIN) is
the minimum switch off time (~150ns). These equations
show that duty cycle range increases when switching
frequency is decreased.
A good choice of switching frequency should allow adequate input voltage range (see next section) and keep
the inductor and capacitor values small.
Input Voltage Range
The maximum input voltage for LT1913 applications
depends on switching frequency and Absolute Maximum Ratings of the VIN and BOOST pins (25V and 50V
respectively).
While the output is in start-up, short-circuit, or other
overload conditions, the switching frequency should be
chosen according to the following equation:
VIN(MAX ) =
VOUT + VD
–V +V
fSW tON(MIN ) D SW
where VIN(MAX) is the maximum operating input voltage,
VOUT is the output voltage, VD is the catch diode drop
(~0.5V), VSW is the internal switch drop (~0.5V at max
load), fSW is the switching frequency (set by RT), and
tON(MIN) is the minimum switch on time (~100ns). Note that
a higher switching frequency will depress the maximum
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LT1913
APPLICATIONS INFORMATION
operating input voltage. Conversely, a lower switching
frequency will be necessary to achieve safe operation at
high input voltages.
If the output is in regulation and no short-circuit, startup, or overload events are expected, then input voltage
transients of up to 25V are acceptable regardless of the
switching frequency. In this mode, the LT1913 may enter
pulse skipping operation where some switching pulses
are skipped to maintain output regulation. In this mode
the output voltage ripple and inductor current ripple will
be higher than in normal operation.
The minimum input voltage is determined by either the
LT1913’s minimum operating voltage of ~3.6V or by its
maximum duty cycle (see equation in previous section).
The minimum input voltage due to duty cycle is:
VIN(MIN ) =
VOUT + VD
–V +V
1– fSW tOFF (MIN ) D SW
where VIN(MIN) is the minimum input voltage, and tOFF(MIN)
is the minimum switch off time (150ns). Note that higher
switching frequency will increase the minimum input
voltage. If a lower dropout voltage is desired, a lower
switching frequency should be used.
Inductor Selection
For a given input and output voltage, the inductor value
and switching frequency will determine the ripple current.
The ripple current ΔIL increases with higher VIN or VOUT
and decreases with higher inductance and faster switching frequency. A reasonable starting point for selecting
the ripple current is:
ΔIL = 0.4(IOUT(MAX))
where IOUT(MAX) is the maximum output load current. To
guarantee sufficient output current, peak inductor current
must be lower than the LT1913’s switch current limit (ILIM).
The peak inductor current is:
IL(PEAK) = IOUT(MAX) + ΔIL/2
where IL(PEAK) is the peak inductor current, IOUT(MAX) is
the maximum output load current, and ΔIL is the inductor
ripple current. The LT1913’s switch current limit (ILIM) is
5.5A at low duty cycles and decreases linearly to 4.5A at
DC = 0.8. The maximum output current is a function of
the inductor ripple current:
IOUT(MAX) = ILIM – ΔIL/2
Be sure to pick an inductor ripple current that provides
sufficient maximum output current (IOUT(MAX)).
The largest inductor ripple current occurs at the highest
VIN. To guarantee that the ripple current stays below the
specified maximum, the inductor value should be chosen
according to the following equation:
V +V V +V L = OUT D 1– OUT D VIN(MAX) fSW IL where VD is the voltage drop of the catch diode (~0.4V),
VIN(MAX) is the maximum input voltage, VOUT is the output
voltage, fSW is the switching frequency (set by RT), and
L is in the inductor value.
The inductor’s RMS current rating must be greater than the
maximum load current and its saturation current should be
about 30% higher. To keep the efficiency high, the series
resistance (DCR) should be less than 0.05 , and the core
material should be intended for high frequency applications.
Table 1 lists several vendors and suitable types.
Table 1. Inductor Vendors
VENDOR
URL
PART SERIES
TYPE
Murata
www.murata.com
LQH55D
Open
TDK
www.componenttdk.com
SLF10145
Shielded
Toko
www.toko.com
D75C
D75F
Shielded
Open
Sumida
www.sumida.com
CDRH74
CR75
CDRH8D43
Shielded
Open
Shielded
NEC
www.nec.com
MPLC073
MPBI0755
Shielded
Shielded
Of course, such a simple design guide will not always result in the optimum inductor for your application. A larger
value inductor provides a slightly higher maximum load
current and will reduce the output voltage ripple. If your
1913f
11
LT1913
APPLICATIONS INFORMATION
load is lower than 3.5A, then you can decrease the value
of the inductor and operate with higher ripple current. This
allows you to use a physically smaller inductor, or one
with a lower DCR resulting in higher efficiency. There are
several graphs in the Typical Performance Characteristics
section of this data sheet that show the maximum load
current as a function of input voltage and inductor value
for several popular output voltages. Low inductance may
result in discontinuous mode operation, which is okay
but further reduces maximum load current. For details of
maximum output current and discontinuous mode operation, see Linear Technology Application Note 44. Finally,
for duty cycles greater than 50% (VOUT/VIN > 0.5), there
is a minimum inductance required to avoid subharmonic
oscillations. See AN19.
Input Capacitor
Bypass the input of the LT1913 circuit with a ceramic
capacitor of X7R or X5R type. Y5V types have poor
performance over temperature and applied voltage, and
should not be used. A 10μF to 22μF ceramic capacitor is
adequate to bypass the LT1913 and will easily handle the
ripple current. Note that larger input capacitance is required
when a lower switching frequency is used. If the input
power source has high impedance, or there is significant
inductance due to long wires or cables, additional bulk
capacitance may be necessary. This can be provided with
a lower performance electrolytic capacitor.
Step-down regulators draw current from the input supply in pulses with very fast rise and fall times. The input
capacitor is required to reduce the resulting voltage
ripple at the LT1913 and to force this very high frequency
switching current into a tight local loop, minimizing EMI.
A 10μF capacitor is capable of this task, but only if it is
placed close to the LT1913 and the catch diode (see the
PCB Layout section). A second precaution regarding the
ceramic input capacitor concerns the maximum input
voltage rating of the LT1913. A ceramic input capacitor
combined with trace or cable inductance forms a high
quality (under damped) tank circuit. If the LT1913 circuit
is plugged into a live supply, the input voltage can ring to
twice its nominal value, possibly exceeding the LT1913’s
voltage rating. This situation is easily avoided (see the Hot
Plugging Safety section).
For space sensitive applications, a 4.7μF ceramic capacitor can be used for local bypassing of the LT1913 input.
However, the lower input capacitance will result in increased input current ripple and input voltage ripple, and
may couple noise into other circuitry. Also, the increased
voltage ripple will raise the minimum operating voltage
of the LT1913 to ~3.7V.
Output Capacitor and Output Ripple
The output capacitor has two essential functions. Along
with the inductor, it filters the square wave generated by the
LT1913 to produce the DC output. In this role it determines
the output ripple, and low impedance at the switching
frequency is important. The second function is to store
energy in order to satisfy transient loads and stabilize the
LT1913’s control loop. Ceramic capacitors have very low
equivalent series resistance (ESR) and provide the best
ripple performance. A good starting value is:
COUT =
100
VOUT fSW
where fSW is in MHz, and COUT is the recommended
output capacitance in μF. Use X5R or X7R types. This
choice will provide low output ripple and good transient
response. Transient performance can be improved with a
higher value capacitor if the compensation network is also
adjusted to maintain the loop bandwidth. A lower value
of output capacitor can be used to save space and cost
but transient performance will suffer. See the Frequency
Compensation section to choose an appropriate compensation network.
When choosing a capacitor, look carefully through the
data sheet to find out what the actual capacitance is under
operating conditions (applied voltage and temperature).
A physically larger capacitor, or one with a higher voltage
1913f
12
LT1913
APPLICATIONS INFORMATION
Table 2. Capacitor Vendors
VENDOR
PHONE
URL
PART SERIES
Panasonic
(714) 373-7366
www.panasonic.com
Ceramic,
COMMANDS
Polymer,
EEF Series
Tantalum
Kemet
(864) 963-6300
www.kemet.com
Ceramic,
Tantalum
Sanyo
(408) 749-9714
www.sanyovideo.com
T494, T495
Ceramic,
Polymer,
POSCAP
Tantalum
Murata
(408) 436-1300
AVX
www.murata.com
Ceramic
www.avxcorp.com
Ceramic,
Tantalum
Taiyo Yuden
(864) 963-6300
www.taiyo-yuden.com
rating, may be required. High performance tantalum or
electrolytic capacitors can be used for the output capacitor.
Low ESR is important, so choose one that is intended for
use in switching regulators. The ESR should be specified by the supplier, and should be 0.05 or less. Such a
capacitor will be larger than a ceramic capacitor and will
have a larger capacitance, because the capacitor must be
large to achieve low ESR. Table 2 lists several capacitor
vendors.
Catch Diode
The catch diode conducts current only during switch off
time. Average forward current in normal operation can
be calculated from:
ID(AVG) = IOUT (VIN – VOUT)/VIN
where IOUT is the output load current. The only reason to
consider a diode with a larger current rating than necessary
for nominal operation is for the worst-case condition of
shorted output. The diode current will then increase to the
TPS Series
Ceramic
typical peak switch current. Peak reverse voltage is equal
to the regulator input voltage. Use a schottky diode with a
reverse voltage rating greater than the input voltage. Table
3 lists several Schottky diodes and their manufacturers.
Table 3. Diode Vendors
PART NUMBER
VR
(V)
IAVE
(A)
VF AT 3A
(mV)
On Semiconductor
MBRA340
40
3
500
Diodes Inc.
B330
B320
30
20
3
3
500
450
Frequency Compensation
The LT1913 uses current mode control to regulate the
output. This simplifies loop compensation. In particular, the
LT1913 does not require the ESR of the output capacitor
for stability, so you are free to use ceramic capacitors to
achieve low output ripple and small circuit size. Frequency
compensation is provided by the components tied to the
1913f
13
LT1913
APPLICATIONS INFORMATION
Capacitor C3 and the internal boost Schottky diode (see
the Block Diagram) are used to generate a boost voltage that is higher than the input voltage. In most cases
a 0.47μF capacitor will work well. Figure 2 shows three
ways to arrange the boost circuit. The BOOST pin must be
LT1913
CURRENT MODE
POWER STAGE
gm = 5.3mho
SW
ERROR
AMPLIFIER
OUTPUT
R1
CPL
FB
gm =
525μmho
+
Loop compensation determines the stability and transient
performance. Designing the compensation network is a bit
complicated and the best values depend on the application
and in particular the type of output capacitor. A practical
approach is to start with one of the circuits in this data
sheet that is similar to your application and tune the compensation network to optimize the performance. Stability
should then be checked across all operating conditions,
including load current, input voltage and temperature. The
LT1375 data sheet contains a more thorough discussion of
loop compensation and describes how to test the stability using a transient load. Figure 2 shows an equivalent
circuit for the LT1913 control loop. The error amplifier is a
transconductance amplifier with finite output impedance.
The power section, consisting of the modulator, power
switch and inductor, is modeled as a transconductance
amplifier generating an output current proportional to
the voltage at the VC pin. Note that the output capacitor
integrates this current, and that the capacitor on the VC pin
(CC) integrates the error amplifier output current, resulting
in two poles in the loop. In most cases a zero is required
and comes from either the output capacitor ESR or from
a resistor RC in series with CC. This simple model works
well as long as the value of the inductor is not too high
and the loop crossover frequency is much lower than the
switching frequency. A phase lead capacitor (CPL) across
the feedback divider may improve the transient response.
Figure 3 shows the transient response when the load current is stepped from 1A to 3A and back to 1A.
BOOST and BIAS Pin Considerations
–
VC pin, as shown in Figure 2. Generally a capacitor (CC)
and a resistor (RC) in series to ground are used. In addition, there may be lower value capacitor in parallel. This
capacitor (CF) is not part of the loop compensation but
is used to filter noise at the switching frequency, and is
required only if a phase-lead capacitor is used or if the
output capacitor has high ESR.
ESR
0.8V
C1
+
3M
C1
VC
CF
POLYMER
OR
TANTALUM
GND
RC
CERAMIC
R2
CC
1913 F02
Figure 2. Model for Loop Response
VOUT
100mV/DIV
IL
1A/DIV
10μs/DIV
1913 F03
Figure 3. Transient Load Response of the LT1913 Front Page
Application as the Load Current is Stepped from 1A to 3A.
VOUT = 5V
1913f
14
LT1913
APPLICATIONS INFORMATION
VOUT
BD
BOOST
VIN
VIN
LT1913
GND
4.7μF
C3
SW
(4a) For VOUT > 2.8V
VOUT
D2
BD
BOOST
VIN
VIN
LT1913
GND
4.7μF
is more efficient because the BOOST pin current and BD
pin quiescent current comes from a lower voltage source.
You must also be sure that the maximum voltage ratings
of the BOOST and BD pins are not exceeded.
The minimum operating voltage of an LT1913 application
is limited by the minimum input voltage (3.6V) and by the
maximum duty cycle as outlined in a previous section. For
proper startup, the minimum input voltage is also limited
by the boost circuit. If the input voltage is ramped slowly,
or the LT1913 is turned on with its RUN/SS pin when the
output is already in regulation, then the boost capacitor
may not be fully charged. Because the boost capacitor is
C3
SW
6.0
(4b) For 2.5V < VOUT < 2.8V
VOUT
INPUT VOLTAGE (V)
5.5
BD
TO START
(WORST CASE)
5.0
4.5
4.0
TO RUN
3.5
3.0
VOUT = 3.3V
TA = 25°C
L = 8.2μH
f = 600kHz
BOOST
VIN
VIN
LT1913
2.5
C3
2.0
4.7μF
GND
SW
10
1
100
1000
LOAD CURRENT (mA)
10000
8.0
1913 FO4
more than 2.3V above the SW pin for best efficiency. For
outputs of 3V and above, the standard circuit (Figure 4a)
is best. For outputs between 2.8V and 3V, use a 1μF boost
capacitor. A 2.5V output presents a special case because it
is marginally adequate to support the boosted drive stage
while using the internal boost diode. For reliable BOOST pin
operation with 2.5V outputs use a good external Schottky
diode (such as the ON Semi MBR0540), and a 1μF boost
capacitor (see Figure 4b). For lower output voltages the
boost diode can be tied to the input (Figure 4c), or to
another supply greater than 2.8V. The circuit in Figure 4a
INPUT VOLTAGE (V)
Figure 4. Three Circuits For Generating The Boost Voltage
TO START
(WORST CASE)
7.0
(4c) For VOUT < 2.5V
6.0
5.0
TO RUN
4.0
VOUT = 5V
TA = 25°C
L = 8.2μH
f = 600kHz
3.0
2.0
1
10
100
1000
LOAD CURRENT (mA)
10000
1913 F05
Figure 5. The Minimum Input Voltage Depends on
Output Voltage, Load Current and Boost Circuit
1913f
15
LT1913
APPLICATIONS INFORMATION
charged with the energy stored in the inductor, the circuit
will rely on some minimum load current to get the boost
circuit running properly. This minimum load will depend
on input and output voltages, and on the arrangement of
the boost circuit. The minimum load generally goes to
zero once the circuit has started. Figure 5 shows a plot
of minimum load to start and to run as a function of input
voltage. In many cases the discharged output capacitor
will present a load to the switcher, which will allow it to
start. The plots show the worst-case situation where VIN
is ramping very slowly. For lower start-up voltage, the
boost diode can be tied to VIN.
At light loads, the inductor current becomes discontinuous and the effective duty cycle can be very high. This
reduces the minimum input voltage to approximately
300mV above VOUT. At higher load currents, the inductor
current is continuous and the duty cycle is limited by the
maximum duty cycle of the LT1913, requiring a higher
input voltage to maintain regulation.
Soft-Start
The RUN/SS pin can be used to soft-start the LT1913,
reducing the maximum input current during start-up.
The RUN/SS pin is driven through an external RC filter to
create a voltage ramp at this pin. Figure 6 shows the startup and shut-down waveforms with the soft-start circuit.
By choosing a large RC time constant, the peak start-up
current can be reduced to the current that is required to
regulate the output, with no overshoot. Choose the value
of the resistor so that it can supply 20μA when the RUN/SS
pin reaches 2.5V.
Synchronization
Synchronizing the LT1913 oscillator to an external frequency can be done by connecting a square wave (with
20% to 80% duty cycle) to the SYNC pin. The square
wave amplitude should have valleys that are below 0.3V
and peaks that are above 0.8V (up to 6V).
The LT1913 may be synchronized over a 250kHz to 2MHz
range. The RT resistor should be chosen to set the LT1913
switching frequency 20% below the lowest synchronization
IL
1A/DIV
RUN
15k
RUN/SS
0.22μF
VRUN/SS
2V/DIV
GND
VOUT
2V/DIV
2ms/DIV
1913 F06
Figure 6. To Soft-Start the LT1913, Add a Resisitor
and Capacitor to the RUN/SS Pin
input. For example, if the synchronization signal will be
250kHz and higher, the RT should be chosen for 200kHz.
To assure reliable and safe operation the LT1913 will only
synchronize when the output voltage is near regulation
as indicated by the PG flag. It is therefore necessary to
choose a large enough inductor value to supply the required
output current at the frequency set by the RT resistor. See
Inductor Selection section. It is also important to note that
slope compensation is set by the RT value: When the sync
frequency is much higher than the one set by RT, the slope
compensation will be significantly reduced which may
require a larger inductor value to prevent subharmonic
oscillation.
Shorted and Reversed Input Protection
If the inductor is chosen so that it won’t saturate excessively, an LT1913 buck regulator will tolerate a shorted
output. There is another situation to consider in systems
where the output will be held high when the input to the
LT1913 is absent. This may occur in battery charging applications or in battery backup systems where a battery
or some other supply is diode OR-ed with the LT1913’s
output. If the VIN pin is allowed to float and the RUN/SS
pin is held high (either by a logic signal or because it is
tied to VIN), then the LT1913’s internal circuitry will pull
its quiescent current through its SW pin. This is fine if
your system can tolerate a few mA in this state. If you
ground the RUN/SS pin, the SW pin current will drop to
essentially zero. However, if the VIN pin is grounded while
1913f
16
LT1913
APPLICATIONS INFORMATION
the output is held high, then parasitic diodes inside the
LT1913 can pull large currents from the output through
the SW pin and the VIN pin. Figure 7 shows a circuit that
will run only when the input voltage is present and that
protects against a shorted or reversed input.
The Exposed Pad on the bottom of the package must be
soldered to ground so that the pad acts as a heat sink. To
keep thermal resistance low, extend the ground plane as
much as possible, and add thermal vias under and near
the LT1913 to additional ground planes within the circuit
board and on the bottom side.
PCB Layout
For proper operation and minimum EMI, care must be
taken during printed circuit board layout. Figure 8 shows
the recommended component placement with trace,
ground plane and via locations. Note that large, switched
currents flow in the LT1913’s VIN and SW pins, the catch
diode (D1) and the input capacitor (C1). The loop formed
by these components should be as small as possible. These
components, along with the inductor and output capacitor,
should be placed on the same side of the circuit board,
and their connections should be made on that layer. Place
a local, unbroken ground plane below these components.
The SW and BOOST nodes should be as small as possible.
Finally, keep the FB and VC nodes small so that the ground
traces will shield them from the SW and BOOST nodes.
Hot Plugging Safely
The small size, robustness and low impedance of ceramic
capacitors make them an attractive option for the input
bypass capacitor of LT1913 circuits. However, these capacitors can cause problems if the LT1913 is plugged into a
live supply (see Linear Technology Application Note 88 for
a complete discussion). The low loss ceramic capacitor,
combined with stray inductance in series with the power
source, forms an under damped tank circuit, and the
voltage at the VIN pin of the LT1913 can ring to twice the
nominal input voltage, possibly exceeding the LT1913’s
rating and damaging the part. If the input supply is poorly
controlled or the user will be plugging the LT1913 into an
energized supply, the input network should be designed
D4
MBRS140
VIN
VIN
L1
BOOST
C2
VOUT
LT1913
RUN/SS
VOUT
SW
RRT
VC
CC
GND FB
BACKUP
RC
R2
1913 F07
Figure 7. Diode D4 Prevents a Shorted Input from
Discharging a Backup Battery Tied to the Output. It Also
Protects the Circuit from a Reversed Input. The LT1913
Runs Only When the Input is Present
R1
D1
C1
GND
RPG
1913 F08
VIAS TO LOCAL GROUND PLANE
VIAS TO VOUT
VIAS TO SYNC
VIAS TO RUN/SS
VIAS TO PG
VIAS TO VIN
OUTLINE OF LOCAL
GROUND PLANE
Figure 8. A Good PCB Layout Ensures Proper, Low EMI Operation
1913f
17
LT1913
APPLICATIONS INFORMATION
to prevent this overshoot. Figure 9 shows the waveforms
that result when an LT1913 circuit is connected to a 24V
supply through six feet of 24-gauge twisted pair. The
first plot is the response with a 4.7μF ceramic capacitor
at the input. The input voltage rings as high as 50V and
the input current peaks at 26A. A good solution is shown
in Figure 9b. A 0.7 resistor is added in series with the
input to eliminate the voltage overshoot (it also reduces
the peak input current). A 0.1μF capacitor improves high
frequency filtering. For high input voltages its impact on
efficiency is minor, reducing efficiency by 1.5 percent for
a 5V output at full load operating from 24V.
CLOSING SWITCH
SIMULATES HOT PLUG
IIN
VIN
High Temperature Considerations
The PCB must provide heat sinking to keep the LT1913
cool. The Exposed Pad on the bottom of the package must
be soldered to a ground plane. This ground should be tied
to large copper layers below with thermal vias; these layers will spread the heat dissipated by the LT1913. Place
additional vias can reduce thermal resistance further. With
these steps, the thermal resistance from die (or junction)
to ambient can be reduced to JA = 35°C/W or less. With
100 LFPM airflow, this resistance can fall by another 25%.
Further increases in airflow will lead to lower thermal re-
DANGER
VIN
20V/DIV
RINGING VIN MAY EXCEED
ABSOLUTE MAXIMUM RATING
LT1913
+
4.7μF
LOW
IMPEDANCE
ENERGIZED
24V SUPPLY
IIN
10A/DIV
STRAY
INDUCTANCE
DUE TO 6 FEET
(2 METERS) OF
TWISTED PAIR
20μs/DIV
(9a)
0.7Ω
LT1913
VIN
20V/DIV
+
0.1μF
4.7μF
IIN
10A/DIV
(9b)
LT1913
+
22μF
35V
AI.EI.
20μs/DIV
VIN
20V/DIV
+
4.7μF
IIN
10A/DIV
(9c)
20μs/DIV
1913 F09
Figure 9. A Well Chosen Input Network Prevents Input Voltage Overshoot and
Ensures Reliable Operation when the LT1913 is Connected to a Live Supply
1913f
18
LT1913
APPLICATIONS INFORMATION
sistance. Because of the large output current capability of
the LT1913, it is possible to dissipate enough heat to raise
the junction temperature beyond the absolute maximum of
125°C. When operating at high ambient temperatures, the
maximum load current should be derated as the ambient
temperature approaches 125°C.
Power dissipation within the LT1913 can be estimated by
calculating the total power loss from an efficiency measurement and subtracting the catch diode loss and inductor
loss. The die temperature is calculated by multiplying the
LT1913 power dissipation by the thermal resistance from
junction to ambient.
Other Linear Technology Publications
Application Notes 19, 35 and 44 contain more detailed
descriptions and design information for buck regulators
and other switching regulators. The LT1376 data sheet
has a more extensive discussion of output ripple, loop
compensation and stability testing. Design Note 100
shows how to generate a bipolar output supply using a
buck regulator.
TYPICAL APPLICATIONS
5V Step-Down Converter
VOUT
5V
3.5A
VIN
6.5V TO 25V
VIN
BD
RUN/SS
ON OFF
BOOST
0.47μF
VC
10μF
LT1913
SW
D
RT
15k
L
4.7μH
PG
SYNC
63.4k
680pF
f = 600kHz
536k
GND
FB
47μF
100k
1913 TA02
D: ON SEMI MBRA340
L: NEC MPLC0730L4R7
1913f
19
LT1913
TYPICAL APPLICATIONS
3.3V Step-Down Converter
VOUT
3.3V
3.5A
VIN
4.8V TO 25V
VIN
BD
RUN/SS
ON OFF
BOOST
L
3.3μH
0.47μF
VC
4.7μF
SW
LT1913
D
RT
19k
PG
SYNC
63.4k
316k
FB
GND
680pF
47μF
100k
f = 600kHz
1913 TA03
D: ON SEMI MBRA340
L: NEC MPLC0730L3R3
2.5V Step-Down Converter
VOUT
2.5V
3.5A
VIN
4V TO 25V
VIN
BD
RUN/SS
ON OFF
D2
BOOST
1μF
VC
4.7μF
LT1913
SW
D1
RT
15.4k
L
3.3μH
PG
215k
SYNC
63.4k
680pF
f = 600kHz
GND
FB
47μF
100k
1913 TA04
D1: ON SEMI MBRA340
D2: MBR0540
L: NEC MPLC0730L3R3
1913f
20
LT1913
TYPICAL APPLICATIONS
5V, 2MHz Step-Down Converter
VOUT
5V
2.5A
VIN
8.6V TO 22V
VIN
BD
RUN/SS
ON OFF
BOOST
L
2.2μH
0.47μF
VC
4.7μF
LT1913
SW
D
RT
15k
PG
536k
SYNC
12.7k
FB
GND
680pF
22μF
100k
f = 2MHz
1913 TA05
D: ON SEMI MBRA340
L: NEC MPLC0730L2R2
12V Step-Down Converter
VOUT
12V
3.5A
VIN
15V TO 25V
VIN
BD
RUN/SS
ON OFF
BOOST
0.47μF
VC
10μF
LT1913
SW
D
RT
17.4k
L
8.2μH
PG
715k
SYNC
63.4k
GND
680pF
f = 600kHz
FB
47μF
50k
1913 TA06
D: ON SEMI MBRA340
L: NEC MBP107558R2P
1913f
21
LT1913
TYPICAL APPLICATIONS
1.8V Step-Down Converter
VOUT
1.8V
3.5A
VIN
3.6V TO 25V
VIN
BD
RUN/SS
ON OFF
BOOST
0.47μF
VC
4.7μF
LT1913
SW
D
RT
16.9k
L
3.3μH
PG
SYNC
78.7k
680pF
f = 500kHz
127k
GND
FB
47μF
100k
1913 TA08
D: ON SEMI MBRA340
L: NEC MPLC0730L3R3
1913f
22
LT1913
PACKAGE DESCRIPTION
DD Package
10-Lead Plastic DFN (3mm × 3mm)
(Reference LTC DWG # 05-08-1699)
0.675 ±0.05
3.50 ±0.05
1.65 ±0.05
2.15 ±0.05 (2 SIDES)
PACKAGE
OUTLINE
0.25 ± 0.05
0.50
BSC
2.38 ±0.05
(2 SIDES)
RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS
3.00 ±0.10
(4 SIDES)
R = 0.115
TYP
6
0.38 ± 0.10
10
1.65 ± 0.10
(2 SIDES)
PIN 1
TOP MARK
(SEE NOTE 6)
(DD) DFN 1103
5
0.200 REF
1
0.25 ± 0.05
0.50 BSC
0.75 ±0.05
0.00 – 0.05
2.38 ±0.10
(2 SIDES)
BOTTOM VIEW—EXPOSED PAD
NOTE:
1. DRAWING TO BE MADE A JEDEC PACKAGE OUTLINE M0-229 VARIATION OF (WEED-2).
CHECK THE LTC WEBSITE DATA SHEET FOR CURRENT STATUS OF VARIATION ASSIGNMENT
2. DRAWING NOT TO SCALE
3. ALL DIMENSIONS ARE IN MILLIMETERS
4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE
MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE
5. EXPOSED PAD SHALL BE SOLDER PLATED
6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON THE
TOP AND BOTTOM OF PACKAGE
1913f
23
LT1913
U
TYPICAL APPLICATIO
1.2V Step-Down Converter
VOUT
1.2V
3.5A
VIN
3.6V TO 25V
VIN
BD
RUN/SS
ON OFF
BOOST
0.47μF
VC
4.7μF
LT1913
SW
D
RT
17k
L
3.3μH
PG
52.3k
SYNC
78.7k
GND
470pF
FB
100k
100μF
f = 500kHz
1913 TA09
D: ON SEMI MBRA340
L: NEC MPLC0730L3R3
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
LT1766
60V, 1.2A (IOUT), 200kHz, High Efficiency Step-Down DC/DC
Converter
VIN: 5.5V to 60V, VOUT(MIN) = 1.2V, IQ = 2.5mA, ISD = 25μA,
TSSOP16/E Package
LT1933
500mA (IOUT), 500kHz Step-Down Switching Regulator in SOT-23
VIN: 3.6V to 36V, VOUT(MIN) = 1.2V, IQ = 1.6mA, ISD < 1μA,
ThinSOTTM Package
LT1936
36V, 1.4A (IOUT), 500kHz, High Efficiency Step-Down DC/DC
Converter
VIN: 3.6V to 36V, VOUT(MIN) = 1.2V, IQ = 1.9mA, ISD < 1μA,
MS8E Package
LT1940
Dual 25V, 1.4A (IOUT), 1.1MHz, High Efficiency Step-Down DC/DC
Converter
VIN: 3.6V to 25V, VOUT(MIN) = 1.2V, IQ = 3.8mA, ISD < 30μA,
TSSOP16E Package
LT1976/LT1967
60V, 1.2A (IOUT), 200kHz/500kHz, High Efficiency Step-Down
DC/DC Converters with Burst Mode Operation
VIN: 3.3V to 60V, VOUT(MIN) = 1.2V, IQ = 100μA, ISD < 1μA,
TSSOP16E Package
LT3434/LT3435
60V, 2.4A (IOUT), 200kHz/500kHz, High Efficiency Step-Down
DC/DC Converters with Burst Mode Operation
VIN: 3.3V to 60V, VOUT(MIN) = 1.2V, IQ = 100μA, ISD < 1μA,
TSSOP16 Package
LT3437
60V, 400mA (IOUT), Micropower Step-Down DC/DC Converter with
Burst Mode Operation
VIN: 3.3V to 60V, VOUT(MIN) = 1.25V, IQ = 100μA, ISD < 1μA,
3mm × 3mm DFN10 and TSSOP16E Packages
LT3480
36V with Transient Protection to 60V, 2A (IOUT), 2.4MHz, High
Efficiency Step-Down DC/DC Converter with Burst Mode Operation
VIN: 3.6V to 38V, VOUT(MIN) = 0.78V, IQ = 70μA, ISD < 1μA,
3mm × 3mm DFN10 and MSOP10E Packages
LT3481
34V with Transient Protection to 36V, 2A (IOUT), 2.8MHz, High
Efficiency Step-Down DC/DC Converter with Burst Mode Operation
VIN: 3.6V to 34V, VOUT(MIN) = 1.26V, IQ = 50μA, ISD < 1μA,
3mm × 3mm DFN10 and MSOP10E Packages
LT3493
36V, 1.4A (IOUT), 750kHz High Efficiency Step-Down
DC/DC Converter
VIN: 3.6V to 36V, VOUT(MIN) = 0.8V, IQ = 1.9mA, ISD < 1μA,
2mm × 3mm DFN6 Package
LT3505
36V with Transient Protection to 40V, 1.4A (IOUT), 3MHz,
High Efficiency Step-Down DC/DC Converter
VIN: 3.6V to 34V, VOUT(MIN) = 0.78V, IQ = 2mA, ISD = 2μA,
3mm × 3mm DFN8 and MSOP8E Packages
LT3508
36V with Transient Protection to 40V, Dual 1.4A (IOUT), 3MHz,
High Efficiency Step-Down DC/DC Converter
VIN: 3.7V to 37V, VOUT(MIN) = 0.8V, IQ = 4.6mA, ISD = 1μA,
4mm × 4mm QFN24 and TSSOP16E Packages
LT3680
36V, 3.5A(IOUT), 2.4MHz High Efficiency Step-Down DC/DC
Converter
VIN: 3.6V to 36V, VOUT(MIN) = 0.79V, IQ = 75μA, ISD < 1μA,
3mm × 3mm DFN, MSOP10E
LT3684
34V with Transient Protection to 36V, 2A (IOUT), 2.8MHz,
High Efficiency Step-Down DC/DC Converter
VIN: 3.6V to 34V, VOUT(MIN) = 1.26V, IQ = 850μA, ISD < 1μA,
3mm × 3mm DFN10 and MSOP10E Packages
LT3685
36V with Transient Protection to 60V, Dual 2A (IOUT), 2.4MHz,
High Efficiency Step-Down DC/DC Converter
VIN: 3.6V to 38V, VOUT(MIN) = 0.78V, IQ = 70μA, ISD < 1μA,
3mm × 3mm DFN10 and MSOP10E Packages
1913f
24 Linear Technology Corporation
LT 1207 • PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 ● FAX: (408) 434-0507
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www.linear.com
© LINEAR TECHNOLOGY CORPORATION 2007