NCV890130 1.2 A, 2 MHz Automotive Buck Switching Regulator The NCV890130 is a fixed−frequency, monolithic, Buck switching regulator intended for Automotive, battery−connected applications that must operate with up to a 36 V input supply. The regulator is suitable for systems with low noise and small form factor requirements often encountered in automotive driver information systems. The NCV890130 is capable of converting the typical 4.5 V to 18 V automotive input voltage range to outputs as low as 3.3 V at a constant switching frequency above the sensitive AM band, eliminating the need for costly filters and EMI countermeasures. The NCV890130 also provides several protection features expected in Automotive power supply systems such as current limit, short circuit protection, and thermal shutdown. In addition, the high switching frequency produces low output voltage ripple even when using small inductor values and an all−ceramic output filter capacitor − forming a space−efficient switching regulator solution. http://onsemi.com MARKING DIAGRAMS 1 DFN8 CASE 506BY 8 8 Internal N−Channel Power Switch Low VIN Operation Down to 4.5 V High VIN Operation to 36 V Withstands Load Dump to 45 V 2 MHz Free−running Switching Frequency Logic level Enable Input Can be Directly Tied to Battery 1.4 A (min) Cycle−by−Cycle Peak Current Limit Short Circuit Protection enhanced by Frequency Foldback ±1.75% Output Voltage Tolerance Output Voltage Adjustable Down to 0.8 V 1.4 Millisecond Internal Soft−Start Thermal Shutdown (TSD) Low Shutdown Current Wettable Flanks − DFN NCV Prefix for Automotive and Other Applications Requiring Unique Site and Control Change Requirements; AEC−Q100 Qualified and PPAP Capable CDRV V890130 ALYWXG G 1 SOIC−8 EP CASE 751AC Features • • • • • • • • • • • • • • • V8901 30 ALYWG G A L Y W G 1 = Assembly Location = Wafer Lot = Year = Work Week = Pb−Free Device (Note: Microdot may be in either location) ORDERING INFORMATION See detailed ordering and shipping information in the package dimensions section on page 17 of this data sheet. • These Devices are Pb−Free and are RoHS Compliant Applications • • • • Audio Infotainment Safety − Vision Systems Instrumentation DBST NCV890130 L1 VIN 1 VIN SW 8 CIN 2 DRV BST 7 3 GND FB 6 CBST DFW VOUT COUT RFB1 EN 4 EN COMP 5 RFB2 RCOMP CCOMP Figure 1. Typical Application © Semiconductor Components Industries, LLC, 2014 October, 2014 − Rev. 3 1 Publication Order Number: NCV890130/D NCV890130 CDRV DBST VIN VIN SW L1 CIN 3V Reg CBST DRV BST PWM LOGIC ON OFF Oscillator 1.2 A + +S + − FB GND + − TSD Soft−Start RESET + COMP VOLTAGES MONITORS RCOMP Enable EN CCOMP Figure 2. NCV890130 Block Diagram http://onsemi.com 2 DFW COUT NCV890130 MAXIMUM RATINGS Rating Symbol Min/Max Voltage VIN Max Voltage VIN to SW Min/Max Voltage SW Min Voltage SW − 20ns Value Unit −0.3 to 45 V 45 V −0.7 to 40 V −3.0 V Min/Max Voltage BST −0.3 to 40 Min/Max Voltage BST to SW −0.3 to 3.6 V Min/Max Voltage on EN −0.3 to 40 V Min/Max Voltage COMP −0.3 to 2 V Min/Max Voltage FB −0.3 to 18 V Min/Max Voltage DRV −0.3 to 3.6 V 50 °C/W −55 to +150 °C Thermal Resistance, 3x3 DFN Junction−to−Ambient* RqJA Storage Temperature range TJ −40 to +150 °C VESD 2.0 200 >1.0 kV V kV Moisture Sensitivity, DFN8 MSL Level 1 Moisture Sensitivity, SOIC−8 EP MSL Level 2 Operating Junction Temperature Range ESD withstand Voltage Human Body Model Machine Model Charge Device Model Peak Reflow Soldering Temperature 260 °C Stresses exceeding those listed in the Maximum Ratings table may damage the device. If any of these limits are exceeded, device functionality should not be assumed, damage may occur and reliability may be affected. *Mounted on 1 sq. in. of a 4−layer PCB with 1 oz. copper thickness. http://onsemi.com 3 NCV890130 VIN 1 8 SW DRV 2 7 BST GND 3 6 FB EN 4 5 COMP (Top View) Figure 3. Pin Connections PIN FUNCTION DESCRIPTIONS Pin No. Symbol Description 1 VIN Input voltage from battery. Place an input filter capacitor in close proximity to this pin. 2 DRV Output voltage to provide a regulated voltage to the Power Switch gate driver. 3 GND Battery return, and output voltage ground reference. 4 EN 5 COMP 6 FB Feedback input pin to program output voltage, and detect pre−charged or shorted output conditions. 7 BST Bootstrap input provides drive voltage higher than VIN to the N−channel Power Switch for optimum switch RDS(on) and highest efficiency. 8 SW Switching node of the Regulator. Connect the output inductor and cathode of the freewheeling diode to this pin. Exposed Pad This TTL compatible Enable input allows the direct connection of Battery as the enable signal. Grounding this input stops switching and reduces quiescent current draw to a minimum. Error Amplifier output, for tailoring transient response with external compensation components. Connect to Pin 3 (electrical ground) and to a low thermal resistance path to the ambient temperature environment. http://onsemi.com 4 NCV890130 ELECTRICAL CHARACTERISTICS (VIN = 4.5 V to 28 V, VEN = 5 V, VBST = VSW + 3.0 V, CDRV = 0.1 mF, Min/Max values are valid for the temperature range −40°C ≤ TJ ≤ 150°C unless noted otherwise, and are guaranteed by test, design or statistical correlation.) Parameter Symbol Conditions Quiescent Current, shutdown IqSD Quiescent Current, enabled Min Typ Max Unit VIN = 13.2 V, VEN = 0 V, TJ = 25°C 10 mA IqEN VIN = 13.2 V 3.0 mA UVLO Start Threshold VUVLSTT VIN rising 4.1 4.5 V UVLO Stop Threshold VUVLSTP VIN falling 3.9 4.4 V UVLO Hysteresis VUVLOHY 0.1 0.2 V Logic Low VENLO 0.8 Logic High VENHI QUIESCENT CURRENT UNDERVOLTAGE LOCKOUT − VIN (UVLO) ENABLE (EN) Input Current IEN 8.0 tSS 0.8 V 2.0 V 30 mA 1.4 2.0 ms 0.8 0.814 V 1.0 mA SOFT−START (SS) Soft−Start Completion Time VOLTAGE REFERENCE FB Pin Voltage during regulation VFBR COMP shorted to FB 0.786 IFBBIAS VFB = 0.8 V 0.25 gm VCOMP = 1.3 V 4.5 V < VIN < 18 V 20 V < VIN < 28 V 0.6 0.3 ERROR AMPLIFIER FB Bias Current Transconductance gm(HV) Output Resistance COMP Source Current Limit COMP Sink Current Limit Minimum COMP voltage mmho ROUT 1.0 0.5 1.5 0.75 1.4 ISOURCE ISINK MW mA VFB = 0.63 V, VCOMP = 1.3 V 4.5 V < VIN < 18 V 20 V < VIN < 28 V 75 40 VFB = 0.97 V, VCOMP = 1.3 V 4.5 V < VIN < 18 V 20 V < VIN < 28 V 75 40 mA VCMPMIN VFB = 0.97 V 0.2 FSW FSW(HV) 4.5 < VIN < 18 V 20 V < VIN < 28 V 1.8 0.9 0.7 V 2.2 1.1 MHz OSCILLATOR Frequency 2.0 1.0 VIN FREQUENCY FOLDBACK MONITOR VFB = 0.63 V V Frequency Foldback Threshold VIN rising VIN falling VFLDUP VFLDDN 18.4 18 Frequency Foldback Hysteresis VFLDHY 0.2 Overvoltage Stop Threshold VOVSTP Overvoltage Start Threshold Overvoltage Hysteresis 20 19.8 0.3 0.4 V 32.4 36 V VOVSTT 30 35.4 V VOVHY 0.6 2.4 V 1.3 0.6 A/ms VIN OVERVOLTAGE SHUTDOWN MONITOR 1.5 SLOPE COMPENSATION Ramp Slope (Note 1) (With respect to switch current) Sramp Sramp(HV) 4.5 < VIN < 18 V 20 V < VIN < 28 V 1. Not tested in production. Limits are guaranteed by design. http://onsemi.com 5 0.7 0.25 NCV890130 ELECTRICAL CHARACTERISTICS (VIN = 4.5 V to 28 V, VEN = 5 V, VBST = VSW + 3.0 V, CDRV = 0.1 mF, Min/Max values are valid for the temperature range −40°C ≤ TJ ≤ 150°C unless noted otherwise, and are guaranteed by test, design or statistical correlation.) Parameter Symbol Conditions ON Resistance RDSON Leakage current VIN to SW Min Typ Max Unit VBST = VSW + 3.0 V 650 mW ILKSW VEN = 0 V, VSW = 0, VIN = 18 V 10 mA Minimum ON Time tONMIN Measured at SW pin 70 ns Minimum OFF Time tOFFMIN Measured at SW pin At FSW = 2 MHz (normal) At FSW = 500 kHz (max duty cycle) POWER SWITCH 45 ns 30 30 50 70 1.4 1.55 1.7 A 400 200 24 500 250 32 600 300 40 kHz VDRV 3.1 3.3 3.5 V DRV POR Start Threshold VDRVSTT 2.7 2.9 3.05 V DRV POR Stop Threshold VDRVSTP 2.5 2.8 3.0 V 45 mA 50 mV PEAK CURRENT LIMIT Current Limit Threshold ILIM SHORT CIRCUIT FREQUENCY FOLDBACK Lowest Foldback Frequency Lowest Foldback Frequency − High Vin Hiccup Mode FSWAF FSWAFHV FSWHIC VFB = 0 V, 4.5 V < VIN < 18 V VFB = 0 V, 20 V < VIN < 28 V VFB = 0 V GATE VOLTAGE SUPPLY (DRV pin) Output Voltage DRV Current Limit IDRVLIM VDRV = 0 V 16 OUTPUT PRECHARGE DETECTOR VSSEN 20 Activation Temperature (Note 1) TSD 150 190 °C Hysteresis (Note 1) THYS 5 20 °C Threshold Voltage 35 THERMAL SHUTDOWN 1. Not tested in production. Limits are guaranteed by design. Product parametric performance is indicated in the Electrical Characteristics for the listed test conditions, unless otherwise noted. Product performance may not be indicated by the Electrical Characteristics if operated under different conditions. http://onsemi.com 6 NCV890130 IqEN. ENABLED QUIESCENT CURRENT (mA) 8 VIN = 13.2 V 7 6 5 4 3 2 1 0 −50 −25 0 25 50 75 100 125 150 2.4 2.2 2.0 1.8 1.6 1.4 1.2 1.0 −50 25 50 75 100 125 Figure 5. Enabled Quiescent Current vs. Junction Temperature 4.5 4.4 4.3 4.2 4.1 4.0 −25 0 25 50 75 100 125 TJ. JUNCTION TEMPERATURE (°C) 150 150 4.6 4.5 4.4 4.3 4.2 4.1 4.0 3.9 3.8 3.7 −50 −25 Figure 6. UVLO Start Threshold vs. Junction Temperature 0 25 50 75 100 125 TJ. JUNCTION TEMPERATURE (°C) 150 Figure 7. UVLO Stop Threshold vs. Junction Temperature 2.4 0.85 VFBR. FB REGULATION VOLTAGE (V) tSS. SOFT−START DURATION (ms) 0 Figure 4. Shutdown Quiescent Current vs. Junction Temperature 4.6 2.2 2.0 1.8 1.6 1.4 1.2 1.0 0.8 0.6 −50 −25 TJ. JUNCTION TEMPERATURE (°C) 4.7 3.9 −50 2.6 TJ. JUNCTION TEMPERATURE (°C) VUVLSTP. UVLO STOP THRESHOLD (V) VUVLSTT. UVLO START THRESHOLD (V) IqSD. SHUTDOWN QUIESCENT CURRENT (mA) TYPICAL CHARACTERISTICS CURVES −25 0 25 50 75 100 125 150 0.84 0.83 0.82 0.81 0.80 0.79 0.78 0.77 0.76 0.75 −50 −25 0 25 50 75 100 125 TJ. JUNCTION TEMPERATURE (°C) TJ. JUNCTION TEMPERATURE (°C) Figure 8. Soft−Start Duration vs. Junction Temperature Figure 9. FB Regulation Voltage vs. Junction Temperature http://onsemi.com 7 150 NCV890130 TYPICAL CHARACTERISTICS CURVES 100 1.2 1.0 VIN = 4.5 V 0.8 0.6 VIN = 28 V 0.4 0.2 −50 −25 0 25 50 75 90 ISOURCE. ERROR AMPLIFIER SOURCING CURRENT (mA) gm. ERROR AMPLIFIER TRANSCONDUCTANCE (mS) 1.4 100 125 VIN = 4.5 V 80 70 60 50 VIN = 28 V 40 30 20 −50 150 −25 TJ. JUNCTION TEMPERATURE (°C) Figure 10. Error Amplifier Transconductance vs. Junction Temperature FSW. OSCILLATOR FREQENCY (MHz) ISINK. ERROR AMPLIFIER SINKING CURRENT (mA) 90 VIN = 4.5 V 70 60 50 VIN = 28 V 40 30 20 −50 −25 0 25 50 75 100 75 100 125 150 125 150 VIN = 13.2 V 2.0 1.8 1.6 1.4 1.2 VIN = 28 V 1.0 0.8 −50 −25 0 25 50 75 100 125 150 TJ. JUNCTION TEMPERATURE (°C) Figure 12. Error Amplifier Max Sinking Current vs. Junction Temperature Figure 13. Oscillator Frequency vs. Junction Temperature 19.6 900 19.4 19.2 RDS(on). POWER SWITCH ON RESISTANCE (mW) VFLDUP. VFLDDN, FREQ. FOLDBACK THRESHOLD (V) 50 2.2 TJ. JUNCTION TEMPERATURE (°C) VFLDUP 19.0 VFLDDN 18.8 18.6 18.4 18.2 −50 25 Figure 11. Error Amplifier Max Sourcing Current vs. Junction Temperature 100 80 0 TJ. JUNCTION TEMPERATURE (°C) −25 0 25 50 75 100 125 TJ. JUNCTION TEMPERATURE (°C) 150 800 700 600 500 400 300 200 100 0 −50 Figure 14. Rising Frequency Foldback Threshold vs. Junction Temperature −25 0 25 50 75 100 125 TJ. JUNCTION TEMPERATURE (°C) Figure 15. Power Switch RDS(on) vs. Junction Temperature http://onsemi.com 8 15 0 NCV890130 80 75 75 70 tOFFMIN. MINIMUM TIME (ns) tONMIN. MINIMUM TIME (ns) TYPICAL CHARACTERISTICS CURVES 70 65 60 55 50 45 40 −50 −25 0 25 50 75 100 125 65 60 55 50 45 40 35 −50 150 −25 TJ. JUNCTION TEMPERATURE (°C) Figure 16. Minimum On Time vs. Junction Temperature FSWAF. FOLDBACK MODE SWITCHING FREQUENCY (kHz) ILIM. MINIMUM TIME (ns) 1.60 1.55 1.50 1.45 75 100 125 150 −25 0 25 50 75 100 125 VIN = 4.5 V 550 500 450 400 350 300 VIN = 28 V 250 200 −50 150 −25 TJ. JUNCTION TEMPERATURE (°C) 0 25 50 75 100 125 150 TJ. JUNCTION TEMPERATURE (°C) Figure 18. Current Limit Threshold vs. Junction Temperature Figure 19. Short−Circuit Foldback Frequency vs. Junction Temperature 40 3.50 38 3.45 VDRV. DRV VOLTAGE (V) FSWHC. HICCUP MODE FREUQNCY (kHz) 50 600 1.65 36 34 32 30 28 3.40 3.35 IDRV = 0 mA 3.30 IDRV = 16 mA 3.25 3.20 3.15 26 24 −50 25 Figure 17. Minimum Off Time vs. Junction Temperature 1.70 1.40 −50 0 TJ. JUNCTION TEMPERATURE (°C) −25 0 25 50 75 100 125 3.10 −50 150 TJ. JUNCTION TEMPERATURE (°C) −25 0 25 50 75 100 125 TJ. JUNCTION TEMPERATURE (°C) Figure 20. Hiccup Mode Switching Frequency vs. Junction Temperature Figure 21. DRV Voltage vs. Junction Temperature http://onsemi.com 9 150 NCV890130 TYPICAL CHARACTERISTICS CURVES IDRVLIM. DRV CURRENT LIMIT (mA) 30 3.0 2.9 VDRVSTT 2.8 VDRVSTP 2.7 2.6 2.5 −50 −25 0 25 50 75 100 125 TJ. JUNCTION TEMPERATURE (°C) 150 29 28 27 26 25 24 23 22 21 −50 Figure 22. DRV Reset Threshold vs. Junction Temperature −25 0 25 50 75 100 125 TJ. JUNCTION TEMPERATURE (°C) Figure 23. DRV Current Limit vs. Junction Temperature 55 VSSEN. OUTPUT PRECHARGE DETECTOR THRESHOLD (V) VDRVSTT. VDRVSTP, DRV RESET THRESHOLDS (V) 3.1 50 45 40 35 30 25 20 −50 −25 0 25 50 75 100 125 TJ. JUNCTION TEMPERATURE (°C) Figure 24. Output Precharge Detector Threshold vs. Junction Temperature http://onsemi.com 10 150 150 NCV890130 GENERAL INFORMATION INPUT VOLTAGE SLOPE COMPENSATION An Undervoltage Lockout (UVLO) circuit monitors the input, and inhibits switching and resets the Soft−start circuit if there is insufficient voltage for proper regulation. The NCV890130 can regulate a 3.3 V output with input voltages above 4.5 V and a 5.0 V output with an input above 6.5 V. The NCV890130 automatically terminates switching if input voltage exceeds VOVSTP (see Figure 25), and withstands input voltages up to 45 V. To limit the power lost in generating the drive voltage for the Power Switch, the switching frequency is reduced by a factor of 2 when the input voltage exceeds the VIN Frequency Foldback threshold VFLDUP (see Figure 25). Frequency reduction is automatically terminated when the input voltage drops back below the VIN Frequency Foldback threshold VFLDDN. A fixed slope compensation signal is generated internally and added to the sensed current to avoid increased output voltage ripple due to bifurcation of inductor ripple current at duty cycles above 50%. The fixed amplitude of the slope compensation signal requires the inductor to be greater than a minimum value, depending on output voltage, in order to avoid sub−harmonic oscillations. For 3.3 V and 5 V output voltages, the recommended inductor value is 4.7 mH. SHORT CIRCUIT FREQUENCY FOLDBACK During severe output overloads or short circuits, the NCV890130 automatically reduces its switching frequency. This creates duty cycles small enough to limit the peak current in the power components, while maintaining the ability to automatically reestablish the output voltage if the overload is removed. If the current is still too high after the switching frequency folds back to 500 kHz, the regulator enters an auto−recovery burst mode that further reduces the dissipated power. Fsw (MHz) 2 CURRENT LIMITING Due to the ripple on the inductor current, the average output current of a buck converter is lower than the peak current setpoint of the regulator. Figure 26 shows − for a 4.7 mH inductor − how the variation of inductor peak current with input voltage affects the maximum DC current the NCV890130 can deliver to a load. 1 1.4 18 20 30 36 45 MINIMUM CURRENT LIMIT (A) 4 VIN (V) Figure 25. NCV890130 Switching Frequency Reduction at High Input Voltage ENABLE The NCV890130 is designed to accept either a logic level signal or battery voltage as an Enable signal. EN low induces a ’sleep mode’ which shuts off the regulator and minimizes its supply current to a couple of mA typically (IqSD) by disabling all functions. Upon enabling, voltage is established at the DRV pin, followed by a soft−start of the switching regulator output. 1.3 (3.3 VOUT) 1.2 1.1 (5 VOUT) 1.0 0.9 0.8 0.7 0.6 0 5 10 15 20 25 30 35 40 INPUT VOLTAGE (V) Figure 26. NCV890130 Load Current Capability with 4.7 mH Inductor SOFT−START Upon being enabled or released from a fault condition, and after the DRV voltage is established, a soft−start circuit ramps the switching regulator error amplifier reference voltage to the final value. During soft−start, the average switching frequency is lower than its normal mode value (typically 2 MHz) until the output voltage approaches regulation. http://onsemi.com 11 NCV890130 BOOTSTRAP In order for the bootstrap capacitor to stay charged, the Switch node needs to be pulled down to ground regularly. In very light load condition, the NCV890130 skips switching cycles to ensure the output voltage stays regulated. When the skip cycle repetition frequency gets too low, the bootstrap voltage collapses and the regulator stops switching. Practically, this means that the NCV890130 needs a minimum load to operate correctly. Figure 27 shows the minimum current requirements for different input and output voltages. At the DRV pin an internal regulator provides a ground−referenced voltage to an external capacitor (CDRV), to allow fast recharge of the external bootstrap capacitor (CBST) used to supply power to the power switch gate driver. If the voltage at the DRV pin goes below the DRV UVLO Threshold VDRVSTP, switching is inhibited and the Soft−start circuit is reset, until the DRV pin voltage goes back up above VDRVSTT. 16 MINIMUM OUTPUT CURRENT (mA) MINIMUM OUTPUT CURRENT (mA) 50 40 L = 2.2 mH 30 20 L = 4.7 mH 10 0 5.2 6.2 7.2 8.2 L = 4.7 mH 10 8 6 4 L = 2.2 mH 2 9.2 4.2 4.7 5.2 5.7 6.2 INPUT VOLTAGE (V) INPUT VOLTAGE (V) Minimum Load 5 V Out Minimum Load 3.3 V Out 20 6.7 50 MINIMUM OUTPUT CURRENT (mA) MINIMUM OUTPUT CURRENT (mA) 12 0 4.2 18 16 14 12 L = 2.2 mH 10 8 6 14 L = 4.7 mH 4 2 0 4.2 4.7 5.2 5.7 6.2 6.7 7.2 45 L = 2.2 mH 40 35 30 25 L = 4.7 mH 20 15 10 5 0 4.2 6.2 8.2 INPUT VOLTAGE (V) INPUT VOLTAGE (V) Minimum Load 3.7 V Out Minimum Load 5.5 V Out Figure 27. Minimum Load Current with Different Input and Output Voltages http://onsemi.com 12 10.2 7.2 NCV890130 7 OUTPUT PRECHARGE DETECTION Prior to Soft−start, the FB pin is monitored to ensure the SW voltage is low enough to have charged the external bootstrap capacitor (CBST). If the FB pin is higher than VSSEN, restart is delayed until the output has discharged. Figure 28 shows the IC starts to switch after the voltage on FB pin reaches VSSEN, even the EN pin is high. After the IC is switching, the FB pin follows the soft starts reference to reach the final set point. VOUT = 5 V 6 VIN (V) 5 4 3 2 1 EN 0 0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0 1.1 1.2 IOUT (A) Time Figure 29. Minimum Input Voltage vs. Output Current EXPOSED PAD FB The exposed pad (EPAD) on the back of the package must be electrically connected to the electrical ground (GND pin) for proper, noise−free operation. VSSEN Time DESIGN METHODOLOGY The NCV890130 being a fixed−frequency regulator with the switching element integrated, is optimized for one value of inductor. This value is set to 4.7 mH, and the slope compensation is adjusted for this inductor. The only components left to be designed are the input and output capacitor and the freewheeling diode. Please refer to the design spreadsheet www.onsemi.com NCV890130 page that helps with the calculation. Output capacitor: The minimum output capacitor value can be calculated based on the specification for output voltage ripple: SW Figure 28. Output Voltage Detection Time THERMAL SHUTDOWN A thermal shutdown circuit inhibits switching, resets the Soft−start circuit, and removes DRV voltage if internal temperature exceeds a safe level. Switching is automatically restored when temperature returns to a safe level. C OUT min + MINIMUM DROPOUT VOLTAGE DI L 8 @ DV OUT @ F SW With − DIL the inductor ripple current: When operating at low input voltages, two parameters play a major role in imposing a minimum voltage drop across the regulator: the minimum off time (that sets the maximum duty cycle), and the on state resistance. When operating in continuous conduction mode (CCM), the output voltage is equal to the input voltage multiplied by the duty ratio. Because the NCV890130 needs a sufficient bootstrap voltage to operate, its duty cycle cannot be 100%: it needs a minimum off time (tOFFmin) to periodically re−fuel the bootstrap capacitor CBST. This imposes a maximum duty ratio DMAX = 1 − tOFFmin.FSW(min), with the switching frequency being folded back down to FSW(min) = 500 kHz to keep regulating at the lowest input voltage possible. The drop due to the on−state resistance is simply the voltage drop across the Switch resistance RDSON at the given output current: VSWdrop = IOUT.RDSon. Which leads to the maximum output voltage in low Vin condition: VOUT = DMAX.VIN(min) − VSWdrop ǒ V OUT @ 1 * DI L + V Ǔ (eq. 1) OUT V IN (eq. 2) L @ F SW − DVOUT the desired voltage ripple. However, the ESR of the output capacitor also contributes to the output voltage ripple, so to comply with the requirement, the ESR cannot exceed RESRmax: R ESR max + DV OUT @ L @ F SW ǒ V OUT 1 * V Ǔ OUT V IN (eq. 3) Finally, the output capacitor must be able to sustain the ac current (or RMS ripple current): I OUTac + DI L 2 Ǹ3 (eq. 4) Typically, with the recommended 4.7 mH inductor, two ceramic capacitors of 10 mF each in parallel give very good results. http://onsemi.com 13 NCV890130 Input capacitor: The input capacitor must sustain the RMS input ripple current IINac: Freewheeling diode: The diode must be chosen according to its maximum current and voltage ratings, and to thermal considerations. As far as max ratings are concerned, the maximum reverse voltage the diode sees is the maximum input voltage (with some margin in case of ringing on the Switch node), and the maximum forward current the peak current limit of the NCV890130, ILIM. The power dissipated in the diode is PDloss: ǒ P Dloss + I OUT @ 1 * Ǔ V OUT V IN I INac + Ǹ (eq. 7) It can be designed in combination with an inductor to build an input filter to filter out the ripple current in the source, in order to reduce EMI conducted emissions. For example, using a 4.7 mH input capacitor, it is easy to calculate that an inductor of 200 nH will ensure that the input filter has a cut−off frequency below 200 kHz (low enough to attenuate the 2 MHz ripple). @ V F ) I DRMS @ R D (eq. 5) with: − IOUT the average (dc) output current − VF the forward voltage of the diode − IDRMS the RMS current in the diode: I DRMS + D Ǹ 3 2 DI L Error Amplifier and Loop Transfer Function ǒ (1 * D) I OUT 2 ) DI L 12 Ǔ The error amplifier is a transconductance type amplifier. The output voltage of the error amplifier controls the peak inductor current at which the power switch shuts off. The Current Mode control method employed allows the use of a simple, type II compensation to optimize the dynamic response according to system requirements. Figure 30 shows the error amplifier with the compensation components and the voltage feedback divider. 2 (eq. 6) − RD the dynamic resistance of the diode (extracted from the V/I curve of the diode in its datasheet). Then, knowing the thermal resistance of the package and the amount of heatsinking on the PCB, the temperature rise corresponding to this power dissipation can be estimated. VOUT RFB1 VCOMP VFB V RCOMP Cp RFB2 RO gm * V CCOMP Vref Figure 30. Feedback Compensator Network Model The transfer function from VOUT to VCOMP is the product of the feedback voltage divider and the error amplifier. RFB2 Gdivider(s) + RFB1 ) RFB2 s 1 ) wz Gerr amp(s) + gm @ Ro @ 1) s 1) s wpl wph ǒ wz + Ǔǒ 1 RCOMP @ CCOMP Ǔ wpl + 1 Ro @ CCOMP wph + (eq. 8) 1 RCOMP @ Cp (eq. 11) (eq. 12) The output resistor Ro of the error amplifier is 1.4 MW and gm is 1 mA/V. The capacitor Cp is for rejecting noise at high frequency and is integrated inside the IC with a value of 18 pF. The power stage transfer function (from Vcomp to output) is shown below: (eq. 9) (eq. 10) s 1 ) wz 1 (eq. 13) Gps(s) + Rload @ @ s @ Fh(s) Ri 1 ) Rload@Tsw @ [Mc @ (1 * D) * 0.5] 1 ) wp L 1 wp + (eq. 14) Resr @ Cout Mc @ (1 * D) * 0.5 1 (eq. 15) wp + ) L @ Cout @ Fsw Rload @ Cout http://onsemi.com 14 NCV890130 where Mc + 1 ) Se Sn (eq. 16) Sn + Vin * Vout @ Ri L (eq. 17) The bode plots of the open loop transfer function will show the gain and phase margin of the system. The compensation network is designed to make sure the system has enough phase margin and bandwidth. Design of the Compensation Network Ri represents the equivalent sensing resistor which has a value of 0.29 W, Se is the compensation slope which is 291.9 kV/S, Sn is the slope of the sensing resistor current during on time. Fh(s) represents the sampling effect from the current loop which has two poles at one half of the switching frequency: 1 2 s ) s 1) wn@Qp wn 2 wn + p @ Fsw Fh(s) + Qp + The function of the compensation network is to provide enough phase margin at crossover frequency to stabilize the system as well as to provide high gain at low frequency to eliminate the steady state error of the output voltage. Please refer to the design spreadsheet www.onsemi.com NCV890130 page that helps with the calculation. The design steps will be introduced through an example. Example: Vin = 15.5 V, Vout = 3.3 V, Rload = 2.75 W, Iout = 1.2 A, L = 4.7 mH, Cout = 20 mF (Resr = 7 mW) The reference voltage of the feedback signal is 0.8 V and to meet the minimum load requirements, select RFB1 = 100 W, RFB2 = 31.6 W. From the specification, the power stage transfer function can be plotted as below: (eq. 18) 1 p @ [Mc @ (1 * D) * 0.5] (eq. 19) The total loop transfer function is the product of power stage and feedback compensation network. (dB) Gloop(s) + Gdivider(s) @ Gerr amp(s) @ Gps(s) (eq. 20) 20‧ log⎣ Gps ⎣f ( m ) ⎦ 90 180 45 90 0 0 − 45 − 90 100 arg(Gps (f m ))‧ 180 p − 90 3 4 1⋅ 10 − 180 6 1⋅ 10 5 1⋅ 10 1⋅ 10 fm (Hz) Figure 31. Power Stage Bode Plots The crossover frequency is chosen to be Fc = 70 kHz, the power stage gain at this frequency is −8 dB (0.398) from calculation. Then the gain of the feedback compensation network must be 8 dB. Next is to decide the locations of one zero and one pole of the compensator. The zero is to provide phase boost at the crossover frequency and the pole is to reject the noise of high frequency. In this example, a zero is placed at 1/10 of the crossover frequency and a pole is placed at 1/5 of the switching frequency (Fsw = 2 MHz): Fz = 7000 Hz, Fp = 400000 Hz, RCOMP, CCOMP and Cp can be calculated from the following equations: RCOMP + Fp @ gm @ Vout @ (Fp * Fz) @ |Gps(Fc)| Vref CCOMP + Cp + Ǹ ǒ Ǔ 2 1 ) Fc Fp Ǹ 1 2p @ Fz @ RCOMP 1 2p @ Fp @ RCOMP (eq. 21) ǒ Ǔ 1 ) Fz Fc 2 (eq. 22) (eq. 23) Note: there is an 18 pF capacitor at the output of the OTA integrated in the IC, and if a larger capacitor needs to be used, subtract this value from the calculated Cp. Figure 32 shows Cp is split into two capacitors. Cint is the 18 pF in the IC. Cext is the extra capacitor added outside the IC. http://onsemi.com 15 NCV890130 From the calculation: So the feedback compensation network is as below: RCOMP = 10.6 KW, CCOMP = 2 nF, Cp = 37 pF VOUT 100 W RFB1 VCOMP VFB V RFB2 31.6 W RCOMP 10 KW 18 pF RO gm*V Cint Vref 0.8 V CCOMP 19 pF Cext 2 nF Figure 32. Example of the Feedback Compensation Network (dB) Figure 33 shows the bode plot of the OTA compensator 20 ‧ log⎣ Gerr_amp ⎣f ( m )⎦ 90 180 45 90 ⎦ 0 − 45 − 90 100 arg(Gerr_amp (f m ))‧ 0 180 p − 90 1 ⋅ 10 3 1 ⋅ 10 4 1 ⋅ 10 5 − 180 6 1 ⋅ 10 fm (Hz) Figure 33. Bode Plot of the OTA Compensator (dB) The total loop bode plot is as below: 20‧ log⎣ Gloop ⎣f ( m )⎦ 90 180 45 90 ⎦ 0 0 − 45 − 90 − 90 100 1⋅ 10 3 1⋅ 10 4 1⋅ 10 5 fm (Hz) Figure 34. Bode Plot of the Total Loop The crossover frequency is at 70 KHz and phase margin is 75 degrees. http://onsemi.com 16 − 180 6 1⋅ 10 arg(Gloop (f m ))‧ 180 p NCV890130 Freewheeling diode ³ inductor ³ Output capacitor ³ return through ground − Minimize the length of high impedance signals, and route them far away from the power loops: ♦ Feedback trace ♦ Comp trace ♦ PCB LAYOUT RECOMMENDATION As with any switching power supplies, there are some guidelines to follow to optimize the layout of the printed circuit board for the NCV890130. However, because of the high switching frequency extra care has to be taken. − Minimize the area of the power current loops: ♦ Input capacitor ³ NCV890130 switch ³ Inductor ³ output capacitor ³ return through Ground ORDERING INFORMATION Package Shipping† NCV890130MWTXG DFN8 with wettable flanks (Pb−Free) 3000 / Tape & Reel NCV890130PDR2G SOIC−8 EP (Pb−Free) 2500 / Tape & Reel Device †For information on tape and reel specifications, including part orientation and tape sizes, please refer to our Tape and Reel Packaging Specifications Brochure, BRD8011/D. http://onsemi.com 17 NCV890130 PACKAGE DIMENSIONS SOIC−8 EP CASE 751AC ISSUE B 2X NOTES: 1. DIMENSIONS AND TOLERANCING PER ASME Y14.5M, 1994. 2. DIMENSIONS IN MILLIMETERS (ANGLES IN DEGREES). 3. DIMENSION b DOES NOT INCLUDE DAMBAR PROTRUSION. ALLOWABLE DAMBAR PROTRUSION SHALL BE 0.08 MM TOTAL IN EXCESS OF THE “b” DIMENSION AT MAXIMUM MATERIAL CONDITION. 4. DATUMS A AND B TO BE DETERMINED AT DATUM PLANE H. 0.10 C A-B D 8 E1 2X 0.10 C D PIN ONE LOCATION DETAIL A D A 5 ÉÉ ÉÉ ÉÉ 1 F EXPOSED PAD 5 8 G E h 2X 4 4 0.20 C e 8X b 0.25 C A-B D B 1 BOTTOM VIEW A END VIEW TOP VIEW A 0.10 C A2 0.10 C b1 GAUGE PLANE L SEATING PLANE C SIDE VIEW ÇÇ ÉÉ ÉÉ ÇÇ ÉÉ ÇÇ c H 8X A A1 0.25 (L1) DETAIL A q c1 (b) SECTION A−A SOLDERING FOOTPRINT* 2.72 0.107 1.52 0.060 7.0 0.275 Exposed Pad 4.0 0.155 2.03 0.08 0.6 0.024 1.270 0.050 SCALE 6:1 mm Ǔ ǒinches *For additional information on our Pb−Free strategy and soldering details, please download the ON Semiconductor Soldering and Mounting Techniques Reference Manual, SOLDERRM/D. http://onsemi.com 18 DIM A A1 A2 b b1 c c1 D E E1 e L L1 F G h q MILLIMETERS MIN MAX 1.35 1.75 0.00 0.10 1.35 1.65 0.31 0.51 0.28 0.48 0.17 0.25 0.17 0.23 4.90 BSC 6.00 BSC 3.90 BSC 1.27 BSC 0.40 1.27 1.04 REF 2.24 3.20 1.55 2.51 0.25 0.50 0_ 8_ NCV890130 PACKAGE DIMENSIONS DFN8, 3x3, 0.5P CASE 506BY ISSUE A A B D L NOTES: 1. DIMENSIONING AND TOLERANCING PER ASME Y14.5M, 1994. 2. CONTROLLING DIMENSION: MILLIMETERS. 3. DIMENSION b APPLIES TO PLATED TERMINAL AND IS MEASURED BETWEEN 0.15 AND 0.30mm FROM THE TERMINAL TIP. 4. COPLANARITY APPLIES TO THE EXPOSED PAD AS WELL AS THE TERMINALS. 5. FOR DEVICE OPN CONTAINING W OPTION, DETAIL B ALTERNATE CONSTRUCTION IS NOT APPLICABLE. L L1 PIN ONE REFERENCE 2X 0.10 C 0.10 C 2X ÉÉÉ ÉÉÉ ÉÉÉ DETAIL A ALTERNATE CONSTRUCTIONS E ÉÉ ÇÇ ÇÇ EXPOSED Cu TOP VIEW (A3) DETAIL B 0.05 C A A3 MOLD CMPD A1 ÇÇ ÉÉ DETAIL B ALTERNATE CONSTRUCTIONS 0.05 C NOTE 4 SIDE VIEW C SEATING PLANE 1 MILLIMETERS MIN MAX 0.80 1.00 0.00 0.05 0.20 REF 0.25 0.35 3.00 BSC 2.20 2.40 3.00 BSC 1.40 1.60 0.65 BSC 0.20 −−− 0.20 0.40 0.00 0.15 RECOMMENDED SOLDERING FOOTPRINT* D2 DETAIL A 8X A1 DIM A A1 A3 b D D2 E E2 e K L L1 ÇÇÇÇÇÇÇ ÇÇÇÇÇÇÇ 4 2.46 L E2 8X 0.53 3.30 1.66 8X 8 K 5 e/2 e BOTTOM VIEW 8X ÇÇÇÇÇÇÇ ÇÇÇÇÇÇÇ b 0.10 C A B 0.05 C 1 NOTE 3 0.65 PITCH 8X 0.40 DIMENSIONS: MILLIMETERS *For additional information on our Pb−Free strategy and soldering details, please download the ON Semiconductor Soldering and Mounting Techniques Reference Manual, SOLDERRM/D. ON Semiconductor and the are registered trademarks of Semiconductor Components Industries, LLC (SCILLC) or its subsidiaries in the United States and/or other countries. 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