AND8257/D Implementing a Medium Power AC−DC Converter with the NCP1395 Prepared by: Roman Stuler ON Semiconductor http://onsemi.com APPLICATION NOTE INTRODUCTION during the light load conditions. Standby consumption of whole supply can thus be significantly decreased. This document describes all the necessary design steps that need to be evaluated when designing the NCP1395 controller in an LLC resonant converter topology. A 240 W AC−DC converter has been selected for the typical application. The design requirements for our 240 W AC−DC converter example are as follows: Requirement Min Max Unit Input Voltage 90 265 Vac Output Voltage − 24 Vdc Output Power 0 240 W Operating Frequency 65 125 kHz Efficiency Under Full Load 90 − % No Load Consumption − 1000 mW Timer Based Fault Protection The converter stops operation after a programmed delay when this input is activated. This protection can be implemented as a cumulative or integrating characteristic. Thus under transient load conditions the converter output will not be turned off, unless the extreme load condition exceeds the timeout. Internal Transconductance Amplifier This internal transconductance amplifier can be used to create effective overload protection. As the result the power supply can be operated in either CV or CC mode. This feature is very useful for the battery chargers applications. Common Collector Optocoupler Connection The open collector output allows multiple inputs to the feedback pin, for example overcurrent sensing circuit, overtemperature sensor, etc. The additional input can pull up the feedback voltage level and take over the voltage feedback loop. Please refer to the NCP1395A/B data sheet for a detailed description of all the functions. LLC series resonant converter topology has been selected to meet efficiency requirements. The NCP1395 resonant mode controller is a very attractive solution for such designs because it offers the following features. Brownout Protection Input Demo Board Connection Description The schematic for the 240 W demo board is shown in Figure 1. The demo board contains three blocks: a PFC front stage (which is necessary for the required power level and to restrict the bulk voltage operating range of the downstream resonant converter), an LLC converter, and an auxiliary buck converter which provides bias power for PFC and LLC controllers. The NCP1653A PFC controller is used to control the PFC front stage. Capacitors C1−C5 together with BALUNs L1, L2 and varistor VDR1 forms the EMI filter which suppresses noise conducted to the mains. The divided down input voltage of the converter is permanently monitored by the Brownout pin (pin name). If the voltage on the bulk capacitor falls outside of the desired operating range, the controller drive output will be shut off. This feature is necessary for an LLC topology because it is usually optimized to operate over a narrow range of bulk voltages. Immediately Fault Input This input can be used as a shutdown input in some applications (LCD television SMPS, etc.). It can also be used to induce skip mode operation of the LLC converter, © Semiconductor Components Industries, LLC, 2006 September, 2006 − Rev. 2 1 Publication Order Number: AND8257/D AND8257/D E11 + R45 R46 220 m/63 V R40 R39 C24 D13 NU 1000 m/35 V 1000 m/35 V C23 NU 33 n NU D7 NU 1000 m/35 V D15B D8 MBRF20100CT MBRF20100CT TR1 R34 100 mH R36 1k2 STP12NM50FP R37 M2 R32 IC5 PC817C NU C22 220 p L4 1m5 15 V E3 2u2/450 V D1 R27 5k6 220 k R25 C21 10 n 0R R23 R30 C7 C6 NINV 330 n R1 0.1R R9 330 n 11 k R5 2M2 R7 R6 470 k CV275K10B1 2n2/Y1 2n2/Y1 Rt 56 k NCP1653 R8 3k3 KBU810 R22 IC1 C17 C9 1n C12 C13 39 n 1n C14 100 n C4 3k3 L2 C3 470 n C2 NU 3m3 L1 C1 470 n Figure 1. Schematic of the NCP1395 Demo Board http://onsemi.com 2 R21 C16 150 k 4u7 NU C15 1u R18 R17 100 k NU N AC INPUT 90−265 VAC L T3, 15 A 100 n 330 k 680 k F1 3m3 C18 R20 R19 RTH1 NU 2M2 C5 IC2 100 n VCC 10 k NCP1395 DRV Gnd VDD 4R7 VDR1 4n7 R12 C11 R11 1N5408 B1 100 n R16 R10 L3 650 mH 56 k C20 1 n + E12 NU 2k7 Voff IN C D6 D5 MURA160 R15 D2 470 k 750 k 750 k CSD0660A SPA20N60C3 M1 PC817 + R14 FB C8 1 n 1M2 NU C19 C10 Gnd Drain IC3 FB IC4 R47 300 k 470 k 470 k + E2 150u/450 V + E1 150u/450 V R2 R3 R4 D3 E4 VCC 2k2 PGnd D4 R13 1 k 47 m + MURA160 NCP1012AP065 220 m/25 V + E5 R33 33 k NU 7V5 10 k MURA160 4M7 D12 D16 10 k G_HI S_HI G_LO S_LO GND VCC DRIVER MODULE DRV_LO DRV_HI R48 AGnd R31 2n2/Y1 R26 BC846BLT1 R29 Q1 R24 180 NU C25 B A L5 STP12N M50FP M3 R35 IC6 TL431 C28 1k R28 100 R D11 NU C26 R38 NU D10 D9 3k3 BO D14B D15A TR2 CST1−100LB− COILCRAFT R41 2k7 22 n VCC D14A 1000 m/35 V 1000 m/35 V R42 FB Ctimer E6 R44 1k OUT S.F. F.F. E7 C290 1 n Fmax DT CSS E10 E9 E8 + + + + + 5k6 NU 18 k L6 2.2 mH R43 C27 NU PE AND8257/D A bridge rectifier is used to rectify the input AC line voltage. Capacitors C6 and C7 filter the high frequency ripple current, which is generated by the PFC stage. In this application a Classical PFC boost topology is used. The PFC power stage is formed by: inductor L3, MOSFET switch M1, SiC diode D2, bulk capacitors E1, E2 and inrush current bypassing diode D1. The current in the PFC stage is monitored by current sense network R1, R8//R9. The output voltage of the PFC stage is regulated to a nominal 400 Vdc via feedback loop components R2−R4 and C9. Part of the rectified input voltage is taken by the divider R5−R7 and capacitors C13, C14 to create over power protection of the PFC switch. Capacitor C12 filters the control voltage and sets the PFC feedback loop bandwidth. Devices connected to Pin 5 of the NCP1653A set the operation mode of the PFC stage to CCM and also dictate over power protection level. Please refer to the application note AND8184/D for detailed explanation how to design a PFC using the NCP1653A controller. An ON Semiconductor NCP1012 monolithic switcher is used for the auxiliary buck converter. This converter provides a stable Vcc to assures proper operation of the PFC and LCC controller under all operating conditions and fault conditions, such as when a short−circuit is applied to the output of the LLC converter. The NCP1012 is connected as a high side switch. Diode D3 is the freewheeling diode. The input power for the buck converter is supplied from the rectified mains via diode D5 and electrolytic capacitor E3. Feedback is done via diode D6, optocoupler IC4 and capacitor C8. Supply voltage of the NCP1012 is maintained on the capacitor E4 using diode D4, resistor R13 and internal DSS architecture. The internal dynamic self supply block is inactive during steady state operation decreasing the power consumption of the buck converter. The output voltage of the buck converter is regulated to 16 V allowing some margin above the PFC controller VccON level, which is 15 V maximum. Please refer to application note AND8191/D for other information regarding NCP10xx products. As it was previously mentioned the NCP1395 resonant mode SMPS controller is used to control the LLC converter power stage, provide output voltage regulation, and fault protection. The power stage of the LLC converter is formed by bulk capacitors E1, E2, MOSFETs M2, M3, resonant inductor L5, transformer TR1 and resonant capacitor C23. A center tapped winding is used on the secondary side to increase efficiency of the converter. Electrolytic capacitors E6−E11 together with inductor L6 serve as an output filter. The output voltage is set at 24 Vdc using a TL431 (IC6) for feedback. The resistive divider formed by R46, R42 and R43 provide a sample of the output voltage to the reference pin of the TL431 (2.5 V). The control loop feedback compensation is done with the series combination of capacitor C26 and resistor R41. The biasing current for the IC6 is provided by the resistor R44. The optocoupler IC current is set by the series resistor R40. The current through the feedback optocoupler is translated to a voltage on the primary side by the resistor R33. The voltage is then applied to the NCP1395 feedback pin. A Zener diode, D12, clamps the maximum feedback voltage and resistor R28 limits the current through D12. The output power level at which the controller enters skip mode is set by the voltage divider R26, R27. The fast fault input is filtered by the capacitor C20. Resistor R30 sets the voltage gain on the output of the operational transconductance amplifier, capacitor C21 is used for the current feedback loop compensation. The voltage on the OTA output is clamped to 7.5 V maximum with the resistor R47 and Zener diode D16. The clamp is necessary because a higher voltage could cause the controller to enter skip mode during startup, or during overload. The output current level for which the skip mode takes place (during the overload conditions) is set by the divider R24, R25. The primary current for the LLC power stage is sensed by current transformer TR2 along with diodes D8−D11, resistors R35, R36, and capacitor C22. An alternative to the current transformer is to sense the primary current with the sensing circuit formed by the resistors R38, R39 diodes D7, D13 and capacitor C24. This alternative is included in the demo board layout. The minimum operating frequency of the converter is set by the resistor R18, and the maximum operating frequency is set by resistor R19. The dead time between the outputs A and B is set by the resistor R20. The soft−start duration is set by capacitor C15, and the timer duration is set by capacitor C16 together with resistor R21. The Brown Out circuit monitors the bulk capacitor voltage with the resistive divider set by R14, R15, R16, R22, and capacitor C18. When the bulk capacitor voltage drops outside the desired operating range of the LCC converter, the output drives are turned off. A decoupling capacitor, C19, is used between the ground and Vcc pin of the controller to improve the noise immunity. The switches for the Half Bridge (M2 and M3) are driven from a High Side Driver module which is mounted vertically on the converter’s main board. This arrangement allows the designer to test the entire driver topology quickly and easily. Two versions of the driver are available as demo board accessories. Schematics for both versions are shown in Figures 2 and 3. One version uses the NCP5181 − integrated high voltage driver and is tailored for consumer applications where the price is important. http://onsemi.com 3 AND8257/D R6 22R D9 G−Hi Q1 BC817−40LT1 MMSD4148 R7 10R +Vcc T1 D1 Input A R3 10k R1 150R D2 C1 100 n Q6 BC807−40LT1 D11 18 V D10 MMSD4148 D12 18 V Q2 BC807−40LT1 S−Hi G−Lo MMSD4148 D1−D4 Q3 BC817−40LT1 MBR140 R5 22R R8 10R D5 D3 Q5 BC807−40LT1 D7 18 V D6 MMSD4148 D8 18 V Input B R4 10k R2 150R C2 100 n D4 GND S−Lo Q4 BC807−40LT1 Figure 2. Connection of the MOSFET Driver with Transformer D1 MURA160 R1 18R R5 0R Input A IC1 IN_Hi Input B R4 0R VBOOT C1 100n R2 10R IN_Lo DRV−Hi G−Hi Bridge S−Hi GND Vcc CRV_LO NCP5181 +Vcc C2 100n R3 10R G−Lo S−Lo GND Figure 3. Connection of the MOSFET Driver with NCP5181 − Integrated Driver http://onsemi.com 4 AND8257/D • Same component count as half bridge topology: The Design steps for resonant tank components values are described in details below. component count is nearly the same between an LLC converter and a classical HB configuration. LLC Converter Stage Design An LLC resonant converter is an attractive topology in comparison to a traditional half bridge for the following reasons: • An LCC converter is capable of ZVS while operating over the entire set of anticipated load conditions. With ZVS the switches are turned on when its drain voltages are zero, the result is nearly zero turn−on losses and a reduction in the EMI signature. The classic half bridge topology, which uses hard switching, can have significant turn−on switching losses and this can increase the EMI signature. • Low turn−off current: switches are turned off under low current and thus the turn− off losses are also lowered in comparison to a classical half bridge topology. • Zero current turn−off of the secondary diodes: when the converter is operating under full load condition, the output rectifiers are turned of under zero current which results in an reduced EMI signature. Disadvantages of the LLC converter lie in these features: • Higher peak and RMS currents in the primary and secondary windings in comparison to the classical HB topology. Thus this topology isn’t very attractive for very high output current levels. • Narrow input voltage operating range: LLC converter has to be optimized for narrow input voltage range, if one wants to take full advantage of the topologies benefits. • Changeable operating frequency: operating frequency of the LLC converter has to vary to keep the output regulated. LLC Resonant Converter Operation Description The LLC resonant converter power stage is shown in Figure 4. + C1 M1 Control Circuit T1 D1 C2 LR + RL LM M2 CR D2 Figure 4. Power Stage of the LLC Resonant Converter One can observe three resonant components in this topology: LR–resonant inductance, LM–magnetizing inductance of the transformer and CR− resonant capacitor. We can define two different resonant frequencies using Thompson’s Equations 1 and 2: fr1 + fr2 + 1 2 · p · ǸLr · Cr 1 2·p· Ǹ(Lr ) Lm) · Cr This topology behaves like a frequency dependent divider which is shown in the Figure 5 schematic. CR (eq. 1) VIN LR LM RAC (eq. 2) Figure 5. Substitutive Schematic of the LLC Resonant Converter http://onsemi.com 5 VOUT AND8257/D One can find the gain transfer function of the divider if the fundamental analysis is used [4], [5]. The main idea of this analysis is that only the fundamental frequency is passed through the resonant tank. As a result of this simplification, the real loading resistance needs to be converted to equivalent loading resistance Rac using Equation 3. Rac + 8 · RL p2 · n2 · h Three operating areas can be identified in these characteristics: 1. In this area (above the resonant frequency fr1) the converter works as series resonant converter. The transformer magnetizing inductance never participates on the resonance because it is clamped by the converter output – one of the secondary diode is conducting for entire switching period. ZVS condition is naturally assured in this operation area for the entire load. 2. In this area the LLC converter works like a multi resonant converter. Let us go through one switching cycle (please refer to the timing diagram depicted in Figure 8). (eq. 3) Where: RL is the real loading resistance n is the transformer turns ratio h is expected efficiency Using an equivalent load resistance (Rac) we can calculate the gain transfer characteristic of the LLC converter and obtain the characteristic for any given value of load resistance, resonant tank components, and quality factor of the resonant circuit. The fastest way to do it is to use SPICE simulator. The result of such simulation can be seen in Figure 6. Figure 8. Typical waveforms of the LLC converter operating in the 2nd area of the gain characteristic. Figure 6. Typical Gain Characteristic of the LLC Resonant Converter The load will change in the real application so it is necessary to plot the characteristic for several different load conditions in one graph. We can use parametric analysis to produce Figure 7. One of the MOSFET switches is turned on. The resonant inductance Lr resonates together with the resonant capacitor Cr while the magnetizing inductance is clamped by the converter output – one of the secondary diode is conducting. When the resonant current decreases to a low value (the same as the magnetizing current) the output diode stop conducting and magnetizing inductance comes to play. The resonant circuit has thus been reconfigured to Lr+Lm – Cr. The resonant frequency fr2 is low in comparison to the resonant frequency fr1. Thus the primary current slowly increases. This current stores energy in the magnetic components − mainly in the magnetizing inductance. Now the switching cycle is forced (by the driver) to be finished. The energy stored in the magnetic components causes ZVS for the opposite MOSFET switch using its body diode. 3. In this area the converter entering to the zero current switching mode. This could happen when converter is overloaded. Some protection circuit (passive or active) has to be implemented to prevent entering this area. Figure 7. Couple of Gain Characteristic for Different Load Conditions http://onsemi.com 6 AND8257/D d) Minimum output current: 0 A – skip mode has to be implemented to assure low power consumption under no load. e) Operating frequency limits: from 65 kHz to 125 kHz. f) Converter should work under series resonant frequency (fr1) for 400 VDC nominal input voltage and full loaded output. This frequency should be below 100 kHz. Based on the input requirement f) we can calculate turns ratio of the LLC transformer. Voltages on the resonant capacitor Cr and resonant inductor Lr are the same values but opposite orientation when converter works at resonant frequency fr1. Thus the gain of the converter is given only by the transformer turns ratio value under this operating condition (assume that the leakage inductance of the transformer is low in comparison with magnetizing inductance). We can calculate needed turns ratio value using Equation 4: The big advantage of the LLC series resonant converter is the fact that the output can be regulated over the entire load range, while the change of the operating frequency is not as high as for other types of resonant converters. Design of the resonant tank components i.e. Lr, Lm, and Cr is always compromise between maximum load changes, maximum acceptable operating frequency excursion, value of the circulating energy in the resonant circuit and short circuit characteristic. Behavior under short circuit is not an issue when some overcurrent protection is used – like with the NCP1395 controller. The best way to operate an LLC resonant converter from the efficiency and EMI point of view is to let it work directly at the resonant frequency fr1. Under these conditions the switching losses are minimized, circulating energy in the resonant tank is also low and secondary diodes are turned off under zero current so there are nearly no reverse recovery losses. This optimal operating point can be reached only for one given input voltage and load resistance value. Thus in the practice the LLC converter is usually designed using these operating conditions i.e. under series resonant frequency fr1 for full load and nominal bulk voltage (which is given by the PFC stage). When the load is decreased the operating frequency is increased by the feedback loop to keep the output voltage regulated. On the other hand converter has also to cope with the bulk voltage drops, due to transient loading (PFC regulation loop is very slow). Also hold up time requirements come into play. The converter will also operate under resonant frequency fr1 in these special cases. The minimum operating bulk voltage of the LLC converter can be effectively limited using the NCP1395 Brown Out input pin. One important thing that has to be taken into account during the LLC designing is the fact that the manufacturing tolerances of the inductors and capacitors are pretty high for standard production. If one wants to have good repeatability for resonant converter design then higher accuracy of these components has to be specified or the LLC converter has to be designed with higher margins. n+ Np Ns + Vin 400 + [ 8 (eq. 4) 2 · (24 ) 0.8) 2 · (Vout ) Vf) Where: Np is the primary turns count Ns is the secondary turns count Vin is the input voltage (PFC output) Vout is the wanted output voltage Vf is the secondary diode voltage drop ETD29 core has been selected for the transformer construction. The primary turns count can be calculated using Equation 5: Np + Vin 8 · DB max · fswmin · Ae (eq. 5) 400 + [ 40 8 · 0.25 · 65 · 103 · 76 · 10−6 where: DBmax is the maximum flux density excursion fswmin is the minimum operating frequency of the converter Ae is the core effective area The minimum switching frequency will be reached only in special cases − overload and bulk voltage dropouts. The flux density excursion will be always lower for normal steady state operation mode – hysteresis losses. The secondary turns count can be calculated using Equation 6: Transformer and Resonant Tank Components Design Based on the mentioned recommendations we can start with the LLC resonant tank components design. Input variables: a) Input voltage range i.e. PFC output voltage: 350–420 VDC, 400 VDC nominal. b) Output voltage: 24 VDC. c) Max output current: 10 A continuous, overcurrent protection with 115% threshold. Np Ns + n + 40 + 5 8 (eq. 6) Needed copper area of the primary and secondary windings can be calculated based on the RMS current values. http://onsemi.com 7 AND8257/D Now we can calculate resonant components values. There are many variables that can be chosen by the designer, however, poor choices can result in converter efficiency degradation or too wide operating frequency range. It is necessary to take the following into account, during the design: 1. Quality factor of the resonant tank Q (and also its characteristic impedance) will significantly affect the gain characteristic and thus also the operating frequency range. If the Q of the resonant circuit is high, the gain characteristic will be narrow and the operating frequency range will be low. However, if the Q is too high the characteristic impedance will be low and converter operation under overload will be degraded. 2. The Lm/Lr = k ratio will also significantly influence shape of the gain characteristics and thus the ZVS region borders [5], [6]. Based on the previous simulations and calculations Q of 3 and k = Lm/Lr = 6 ratio have been chosen. Now we can calculate characteristic impedance Z0 of the resonant circuit using Equation 7: Z0 + 2 n2 · RL + 8 · 2.4 + 51.2 [ 51 W 3 Q Cr = 33 nF fop = fr1 = 85.1 kHz Now we can start simulation of the resonant converter to obtain gain characteristic for full loaded converter. Simulation schematic, which includes the leakage inductance of the transformer, is depicted in Figure 9. V1 1V L4 R4 33 n 100 mH 0.1 TX1 R1 (Rac) 24 V Output 0 0 Figure 9. Simulation Schematic of the Proposed Converter The gain values we are looking for can be calculated from Equations 11 through 13: (eq. 7) The resonant capacitor value can be calculated from Equation 8. Let us select the nominal operating frequency fop = fr1 = 85 kHz for full load and nominal bulk voltage: 1 Cr + 2 · p · fr · Z0 1 + + 36.7 nF 2 · 3.14 · 85 · 103 · 51 Cs (eq. 8) G min + 2 · (Vout ) Vf) 2 · (24 ) 0.8) + + 0.118 (eq. 11) 420 Vin max Gnom + 2 · (Vout ) Vf) 2 · (24 ) 0.8) + + 0.124 (eq. 12) 400 Vinnom G max + 2 · (Vout ) Vf) 2 · (24 ) 0.8) + + 0.142 (eq. 13) 350 Vin min Operating frequencies for these gains can be read from the simulated gain characteristic which is depicted in Figure 10. An E6 standard value capacitor of 33 nF has been chosen. The characteristic impedance will then change to 56.7 W and Q will be 2.71. Now we can easily calculate the resonant inductance value, which is given by Equation 9: Lr + Z02 · Cr + 56.72 · 33 · 10−9 + 106 mH (eq. 9) Use standard E6 value of Lr = 100 mH. The magnetizing inductance is given by the selected k ratio (Equation 10): Lm + k · Lr + 6 · 100 · 10−6 + 600 mH (eq. 10) Now we have nearly all needed for the simulation and obtain gain characteristic. However, for accurate results the leakage inductance of the transformer has to be taken into account. This inductance will affect on the resonance and it will also change gain of the converter somewhat. We have selected the ETD29 transformer core with classical winding technique so we can assume that the leakage inductance will be around 1% of the magnetizing inductance i.e. Llk = 6.0 mH. Let us now summarize real values of the resonant tank components: Figure 10. Simulated Gain Characteristic of Proposed LLC Converter – Full Load Conditions As can be seen the operating frequency range is 61 kHz to 101 kHz, for a fully loaded converter output and with the input voltage range. The simulated operating frequency for the nominal input voltage and full load current is 88 kHz, which is very near to the calculated resonant frequency fr1. One can see that there is enough gain margin to keep the converter in the ZVS region (Region 2 in Figure 7) even during a slight overload on the output. Lr’ = Lr + Llk = 106 mH Lm = 600 mH http://onsemi.com 8 AND8257/D 0.95 0.93 When the load resistance is increased the gain of the converter will increase too, because of the Q changes. Moreover, the drops in the transformer copper and secondary diodes are lower for lower loading current. For these reasons the feedback loop will try to increase the operating frequency to lower the converter gain. If maximum operating frequency limit is set too low, the feedback loop will not be able to regulate the output. This is exactly what we need to assure skip mode for light load. Under such conditions the feedback pin voltage goes up and triggers the fast fault input via resistor divider R26, R27. Simulation can again be used to find the maximum needed frequency for light load regulation. If the maximum operating frequency is limited to 125 kHz skip mode will be automatically implemented. Let us summarize the simulations and design results: 1. Converter should operate at fop [ fr1 = 88 kHz for full load and nominal bulk voltage. 2. Operating frequency range is 61 kHz to 101 kHz, at full load, over the input voltage range. 3. Operating frequency goes above fr1 frequency for light loads. 4. For very light loads, the converter will reach the maximum frequency limit and the feedback will activate fast fault input. Skip mode will thus take place. Vin = 230 V/50 Hz EFFICIENCY (%) 0.91 Vin = 110 V/60 Hz 0.89 0.87 0.85 0.83 0.81 0.79 0.77 0.75 1 2 3 4 5 6 7 8 9 10 Iout (Adc) Figure 11. Efficiency of the Designed Converter versus Output Current EFFICIENCY (%) 0.93 0.92 0.91 0.90 Overcurrent Protection Circuitry The Q of the resonant tank can drop to a very low value during overload conditions. To overcome incorrect operation in the ZCS region, the primary current has to be limited by the overcurrent protection circuit. This circuit has been already mentioned. The onboard OTA takes over and pulls up the feedback pin via transistor Q1 when input current goes over the desired maximum value. 0.89 90 110 130 150 170 190 210 230 250 270 Vsc in (Vrms) Figure 12. Efficiency of the Example Converter versus Input Voltage for Full Loaded Output Results Summarization Operating frequency of the real demo board prototype is 83 kHz for full load and 395 VDC input voltage which is very close to the theoretical results. Output current level for which the skip mode takes place has been set to 400 mA using resistor divider R26, R27. Skip level is given by the feedback voltage and thus it is very sensitive to the resonant component values and output voltage setup accuracy! Measured efficiency for different input voltages and load conditions can be seen in Figures 11 and 12. Loading characteristic of the prototype can be seen in Figure 13. http://onsemi.com 9 AND8257/D OUTPUT VOLTAGE (Vdc) 25 20 15 10 5 0 1 2 3 4 5 6 7 8 9 10 11 12 OUTPUT CURRENT (Adc) Figure 15. Detail of the ZVS Condition – Rising Edge Figure 13. Loading Characteristic of the Proposed Converter One can see that the CV operation is possible up to 10 A load current. Then the overcurrent circuit starts to limit output power and finally the hiccup mode takes place when output current goes over 11.4 A. Several snapshots taken from the prototype can be seen in Figures 14 through 22. Standby consumption is below 1.0 W for both input voltage levels, i.e. 230 VAC and 110 VAC. Figure 16. Detail of the ZVS Condition – Falling Edge Figure 14. Primary Current and Waveforms for Full Loaded Converter http://onsemi.com 10 AND8257/D Figure 17. Load Regulation for 230 V Input Voltage Figure 20. Operating Under Short Circuit Figure 18. Output Ripple Under Full Load Figure 21. Full Load to Short Circuit Transition Figure 19. Output Ripple During the Skip Mode Figure 22. Operation in the Skip Mode http://onsemi.com 11 AND8257/D Layout Consideration Leakage inductance on the primary side is not very critical for LLC converters compared to other topologies, because it will only slightly modify the resonant frequency. However it is well to keep the areas of each power loop as small as possible due to radiated EMI noise. A two−sided PCB with one side a ground plane helps (see Figures 23 through 26). Special care has to be taken with Pins 1, 2 and 3 of the NCP1395 controller because these are high impedance pins. Ensure that these pins are not near high voltages and high dV/dt or use some ground shielding. Literature 1. NCP1395A/B data sheet. 2. Application note AND8184/D. 3. Application note AND8191/D. 4. Application note AND8255. 5. Bo Yang − Topology Investigation for Front End DC−DC Power Conversion for Distributed Power System. 6. Milan Jovanovich – Resonant converters training brochure. 7. M. B. Borage, S. R. Tiwari and S. Kotaiah − Design Optimization for an LCL – Type Series Resonant Converter. CAUTION This board is intended only for demonstration and evaluation purpose. Board is designed for free air operation. Temperature damage of its components can occur if the board will be placed under the cover without forced air cooling. Figure 23. Conducted EMI Signature of the Board http://onsemi.com 12 AND8257/D Figure 24. Component Placement on the Top Side (Top View) Figure 25. Top Side (Top View) http://onsemi.com 13 AND8257/D Figure 26. Component Placement on the Bottom Side (Bottom View) Figure 27. Bottom Side (Bottom View) http://onsemi.com 14 AND8257/D Figure 28. Photo of the Designed Prototype (Real Dimensions are 183 x 122 mm) http://onsemi.com 15 AND8257/D NCP1395 Demo Board Parts List Designator B1 C1,C3 Description 8.0 A Bridge Rectifier EMI Suppression Capacitors (MKP) C2 Value Manufacturer Part Number KBU810 KBU810 470n B81130B1474M NU C4, C5 Safety Ceramic Disc Capacitor 2n2 KZH 2200PF M 2E3 500 Y1/X1 B1 C6,C7 Polyester Chip Capacitor 330n R46X−0,33UF 15 275V M M1 00 C8, C9, C13 SMD Capacitor 1nF VJ0806Y102KXBAx C10 SMD Capacitor 4n7 VJ1206Y472KXBAx C12 SMD Capacitor 39n VJ0806Y393KXBAx C14 SMD Capacitor 100n VJ1206Y104KXBAx C15 SMD Capacitor 1u VJ1206Y105KXBAx C16 SMD Capacitor 4u7 VJ1206Y475KXBAx C17 NU C11, C18, C19 SMD Capacitor 100n VJ1206Y104KXBAx C20, C29 SMD Capacitor 1n VJ1206Y102KXBAx C21 SMD Capacitor 10n VJ1206Y103KXBAx C22 SMD Capacitor 220p VJ1206Y221KXBAx C23 Polypropylene Capacitor 33n R73−0,033UF 15 630V J 00 AA C24 NU C25 Capacitor, Y1 Class C26 SMD Capacitor 2n2/Y1 WKP222 22n VJ1206Y223KXBAx C27 NU C28 NU D1 3.0 A 1000 V Standard Recovery Rectifier D2 600 V; 4.0 A; Zero Recovery Rectifier D3, D4, D5 D6 1N5408 1N5408 CSD04060A CSD04060A MURA160 MURA160 MMSZ15T1 MMSZ15T1 Surface Mount Ultrafast Power Rectifier Zener Diode 500 mW 15 V D7 D8, D9, D10, D11 NU Small Signal Switch Diode 100 V MMSD4148T1 D12 NU D13 NU D14A,B; D15A,B D16 20 A 100 V Schottky Rectifier Zener Diode 500 mW 7.0, 5.0 V MMSD4148T1 MBRF20100CT MBRF20100CT MMSZ7V5T1 MMSZ7V5T1 E1,E2 Radial Lead Electrolytic Capacitor 150u/450V K05 105°C 1500 mF/ 450V 0514 13592 E3 Radial Lead Electrolytic Capacitor 2u2/450V CERA−2,2/450 10x12,5 KMG E4 Radial Lead Electrolytic Capacitor 47u/25V CERA−47/25 5x11 CD268 E5 Radial Lead Electrolytic Capacitor 220u/25V CERA−220/25 8x12 LXZ E6, E7, E8, E9, E10 Radial Lead Electrolytic Capacitor 1000u/35V CERA−1000/35 12,5x25 LXZ E11 Radial Lead Electrolytic Capacitor 220u/63V CERA−220/63 10x16 A KMG E12 NU http://onsemi.com 16 AND8257/D NCP1395 Demo Board Parts List Designator Description F1 Fuse IC1 Compact Fixed−Frequency Current−Mode PFC Controller IC2 Resonant Mode SMPS Controller IC3 Self−Supplied Monolithic Switcher for Low Standby−Power Offline SMPS IC4, IC5 IC6 L1, L2 Value Manufacturer Part Number T3,15A T3,15A NCP1653DR2 NCP1653DR2 NCP1395A NCP1395A NCP1012AP065 NCP1012AP065 PC817P PC817P TL431BILP TL431BILP 2m7 PMEC103/V 2m7 Optocoupler Adjustable Shunt Regulator 2.5−36 V/ 1.0−100 mA Common Mode Inductor L3 Inductor 650uH IND−PFC−260W−v1 L4 Inductor 1m5 RFB0810−152L L5 Inductor 100uH IND−LLC−v1 L6 Inductor 2.2uH IND−FLT−2u2−10A M1 N−Channel 600 V 0.140 W−20 A TO−220 SPA20N60C3 SPA20N60C3 M2, M3 N−Channel 550 V @ Tjmax−0.30 W − 12 A TO−220FP STP12NM50FP STP12NM50FP Q1 General Purpose Transistors NPN Silicon BC846BLT1 BC846BLT1 R1 Axial Lead Resistor 3.0 W 0.1R 3W R2, R7, R15, R16 SMD Resistor 470k CRCW1206 R3, R4 SMD Resistor 750k CRCW1206 R5, R6 SMD Resistor 2M2 CRCW1206 R8, R22, R42 SMD Resistor 3k3 CRCW1206 R9 SMD Resistor 11k CRCW1206 R10 SMD Resistor 4R7 CRCW1206 R12 SMD Resistor 56k CRCW1206 R13, R40, R44 SMD Resistor 1k CRCW1206 R14 SMD Resistor 300k CRCW1206 R17 NU R18 SMD Resistor 100k CRCW1206 R19 SMD Resistor 680k CRCW1206 R20 SMD Resistor 330k CRCW1206 R21 SMD Resistor 150k CRCW1206 R23, STRAP 1, 2 SMD Resistor 0R CRCW1206 R24 SMD Resistor 1M2 CRCW1206 R25 SMD Resistor 220k CRCW1206 R26 SMD Resistor 33k CRCW1206 R27 SMD Resistor 5k6 CRCW1206 R28 SMD Resistor 100R CRCW1206 R29 NU R30 SMD Resistor 56k CRCW1206 R11, R31, R32 SMD Resistor 10k CRCW1206 http://onsemi.com 17 AND8257/D NCP1395 Demo Board Parts List Designator R33 Description SMD Resistor R34 Value Manufacturer Part Number 2k2 CRCW1206 NU CRCW1206 R35 SMD Resistor 180R CRCW1206 R36 SMD Resistor 1k2 CRCW1206 R37 NU R38 NU R39 NU R41, R47 SMD Resistor 2k7 CRCW1206 R43 SMD Resistor 5k6 CRCW1206 R45 NU R46 SMD Resistor 18k CRCW1206 R48 High Ohmic, High Voltage Resistor 4M7 0.5W VR37 Series RTH1 VDR1 NU Voltage Dependent Resistor CV 275 K 10 B1 CV 275 K 10 B 1 TR1 Transformer TRAFO ETD29 TR−LLC−v3−24V TR2 CS Transformer CST1−100LB − CS TR. CST1−100LB − CS Tr. H1 PCB Connector NA SVPS CEE7.5/3 Grey H2 PCB Connector NA SVPS MV 252/5.08 Green HS1 Heat Sink NA CHL01−BLK HS2 Heat Sink NA SK 505 30 SA Driver Module with NCP5181 R1 SMD Resistor 18R CRCW1206 R2, R3 SMD Resistor 10R CRCW1206 R4, R5 SMD Resistor C1, C2 SMD Capacitor 0R CRCW1206 100n VJ1206Y104KXBAx D1 Surface Mount Ultrafast Power Rectifier MURA160 MURA160 IC1 Integrated Half Bridge Driver NCP5181 NCP5181 Driver Module with Transformer R1, R2 SMD Resistor 150R CRCW1206 R3, R4 SMD Resistor 10k CRCW1206 R5, R6 SMD Resistor 22R CRCW1206 R7, R8 SMD Resistor 8R2 CRCW1206 STRAP SMD Resistor 0R CRCW1206 C1, C2 SMD Capacitor 100n VJ1206Y104KXBAx D1, D2, D3, D4 0.5 A 40 V Schottky Rectifier D5, D6, D9, D10 Small Signal Switch Diode 100 V D7, D8, D11, D12 Zener Diode 500 mW 18 V MBR0540T1 MBR0540T1 MMSD4148T1 MMSD4148T1 MMSZ18T1 MMSZ18T1 http://onsemi.com 18 AND8257/D NCP1395 Demo Board Parts List Driver Module with Transformer Q1, Q3 General Purpose Transistors NPN Silicon BC817−40LT1 BC817−40LT1 Q2, Q4, Q5, Q6 General Purpose Transistors PNP Silicon BC807−40LT1 BC807−40LT1 NA TR−DRW−01 TR1 Transformer Please see the NCP1395A/B product folder on www.onsemi.com for PCB Gerber files and other collateral information regarding this demo board. 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