CS5111 1.4 A Switching Regulator with 5.0 V, 100 mA Linear Regulator with Watchdog, RESET and ENABLE The CS5111 is a dual output power supply integrated circuit. It contains a 5.0 V ± 2%, 100 mA linear regulator, a watchdog timer, a linear output voltage monitor to provide a Power On Reset (POR) and a 1.4 A current mode PWM switching regulator. The 5.0 V linear regulator is comprised of an error amplifier, reference, and supervisory functions. It has low internal supply current consumption and provides 1.2 V (typical) dropout voltage at maximum load current. The watchdog timer circuitry monitors an input signal (WDI) from the microprocessor. It responds to the falling edge of this watchdog signal. If a correct watchdog signal is not received within the externally programmable time, a reset signal is issued. The externally programmable active reset circuit operates correctly for an output voltage (VLIN) as low as 1.0 V. During power up, or if the output voltage shifts below the regulation limit, RESET toggles low and remains low for the duration of the delay after proper output voltage regulation is restored. Additionally a reset pulse is issued if the correct watchdog is not received within the programmed time. Reset pulses continue until the correct watchdog signal is received. The reset pulse width and frequency, as well as the Power On Reset delay, are set by one external RC network. The current mode PWM switching regulator is comprised of an error amplifier with selectable feedback inputs, a current sense amplifier, an adjustable oscillator, and a 1.4 A output power switch with anti–saturation control. The switching regulator can be configured in a variety of topologies. The CS5111 is load dump capable and has protection circuitry which includes overvoltage shutdown, current limit on the linear and switcher outputs, and an overtemperature limiter. Features • Linear Regulator – 5.0 V ± 2% @ 100 mA • Switching Regulator – 1.4 A Peak Internal Switch – 120 kHz Maximum Switching Frequency – 5.0 V to 26 V Operating Supply Range • Smart Functions – Watchdog – RESET – ENABLE • Protection – Overvoltage – Overtemperature – Current Limit • 54 V Peak Transient Capability • Internally Fused Leads in SO–24L Package Semiconductor Components Industries, LLC, 2001 January, 2001 – Rev. 9 1 http://onsemi.com SO–24L DWF SUFFIX CASE 751E 24 1 MARKING DIAGRAM 24 CS5111 AWLYYWW 1 A WL, L YY, Y WW, W = Assembly Location = Wafer Lot = Year = Work Week PIN CONNECTIONS VIN NC NC VSW GND GND GND GND VFB1 VFB2 SELECT COMP 1 24 ENABLE VREG VLIN IBIAS GND GND GND GND RESET CDelay WDI COSC ORDERING INFORMATION Device Package Shipping CS5111YDWF24 SO–24L 31 Units/Rail CS5111YDWFR24 SO–24L 1000 Tape & Reel Publication Order Number: CS5111/D CS5111 VIN Switcher Error Amplifier VFB1 VFB2 SELECT Multiplexer – + VSW COMP Base Drive Logic COMP Current Sense Amplifier IBIAS 1.4 A + – GND Oscillator COSC + – Switcher Shutdown ENABLE VREG Overvoltage + – Linear Error Amplifier VLIN Current Limit Bandgap Reference Over Temperature 1.25 V RESET RESET & Watchdog Timer CDELAY WDI Figure 1. Block Diagram ABSOLUTE MAXIMUM RATINGS* Rating Logic Inputs/Outputs (ENABLE, SELECT, WDI, RESET) VLIN VIN, VREG: Value Unit –0.3 to VLIN V –0.3 to 10 DC Input Voltage Peak Transient Voltage (40 V Load Dump @ 14 V VIN) –0.3 to 26 –0.3 to 54 V V 54 V –0.3 to VLIM V Power Dissipation Internally Limited – VLIN Output Current Internally Limited – VSW Output Current Internally Limited – RESET Output Sink Current 5.0 mA ESD Susceptibility (Human Body Model) 2.0 kV ESD Susceptibility (Machine Model) 200 V –65 to 150 °C 230 peak °C VSW Peak Transient Voltage COSC, CDelay, COMP, VFB1, VFB2 Storage Temperature Lead Temperature Soldering: Reflow: (SMD styles only) (Note 1.) 1. 60 second maximum above 183°C. *The maximum package power dissipation must be observed. http://onsemi.com 2 CS5111 ELECTRICAL CHARACTERISTICS (5.0 V ≤ VIN ≤ 26 V and –40°C ≤ TJ ≤ 150°C, COUT = 100 µF (ESR ≤ 8.0 Ω), CDelay = 0.1 µF, RBIAS = 64.9 kΩ, COSC = 390 pF, CCOMP = 0.1 µF; unless otherwise specified.) Test Conditions Characteristic Min Typ Max Unit General IIN Off Current 6.6 V ≤ VIN ≤ 26 V, ISW = 0 A – – 2.0 mA IIN On Current 6.6 V ≤ VIN ≤ 26 V, ISW = 1.4 A – 30 70 mA IREG Current ILIN = 100 mA, 6.6 V ≤ VIN ≤ 26 V – – 6.0 mA Thermal Limit Guaranteed by Design 160 – 210 °C VLIN Output Voltage 6.6 V ≤ VREG ≤ 26 V, 1.0 mA ≤ ILIN ≤ 100 mA 4.9 5.0 5.1 V Dropout Voltage (VREG – VLIN) @ ILIN = 100 mA – 1.2 1.5 V 30 34 38 V 5.0 V Regulator Section Overvoltage Shutdown – Line Regulation 6.6 V ≤ VREG ≤ 26 V, ILIN = 5.0 mA – 5.0 25 mV Load Regulation VREG = 19 V, 1.0 mA ≤ ILIN ≤ 100 mA – 5.0 25 mV Current Limit 6.6 V ≤ VREG ≤ 26 V 120 – – mA DC Ripple Rejection 14 V ≤ VREG ≤ 24 V 60 75 – dB RESET Section Low Threshold (VRTL) VLIN Decreasing 4.05 4.25 4.45 V High Threshold (VRTH) VLIN Increasing 4.2 4.45 4.7 V Hysteresis VRTH – VRTL 140 190 240 mV Active High VLIN > VRTH, IRESET = –25 µA VLIN – 0.5 – – V Active Low VLIN = 1.0 V, 10 kΩ Pull–Up from RESET to VLIN VLIN = 4.0 V, IRESET = 1.0 mA – – 0.4 V – – 0.7 V Delay Invalid WDI 6.25 8.78 11 ms Power On Delay VLIN Crossing VRTH 6.25 – – ms – – 2.0 V 0.8 – – V Watchdog Input (WDI) VIH Peak WDI Needed to Activate RESET VIL – Hysteresis Note 2. 25 50 – mV Pull–Up Resistor WDI = 0 V 20 50 100 kΩ Low Threshold – 6.25 8.78 11 ms Floating Input Voltage – 3.5 – – V WDI Pulse Width – – – 5.0 µs – – – 5.0 V Switcher Section Minimum Operating Input Voltage Switching Frequency Refer to Figure 5 80 95 110 kHz Switch Saturation Voltage ISW = 1.4 A 0.7 1.1 1.6 V 1.4 – 2.5 A 120 – – kHz Output Current Limit Max Switching Frequency – VSW = 7.5 V with 50 Ω Load, Refer to Figure 5 2. Guaranteed by design, not 100% tested in productions. http://onsemi.com 3 CS5111 ELECTRICAL CHARACTERISTICS (continued) (5.0 V ≤ VIN ≤ 26 V and –40°C ≤ TJ ≤ 150°C, COUT = 100 µF (ESR ≤ 8.0 Ω), CDelay = 0.1 µF, RBIAS = 64.9 kΩ, COSC = 390 pF, CCOMP = 0.1 µF; unless otherwise specified.) Characteristic Test Conditions Min Typ Max Unit VFB1 Regulation Voltage – 1.206 1.25 1.294 V VFB2 Regulation Voltage – 1.206 1.25 1.294 V Switcher Section (continued) VFB1, VFB2 Input Current VFB1 = VFB2 = 5.0 V – – 1.0 µA Oscillator Charge Current COSC = 0 V 35 40 45 µA Oscillator Discharge Current COSC = V40 270 320 370 µA CDelay Charge Current CDelay = 0 V 35 40 45 µA Switcher Max Duty Cycle VSW = 5.0 V with 50 Ω Load, VFB1 = VFB2 = 1.0 V 72 85 95 % Current Sense Amp Gain ISW = 2.3 A – 7.0 – V/V Error Amp DC Gain – – 67 – dB Error Amp Transconductance – – 2700 – µA/V VIL – 0.8 1.24 – V VIH – – 1.3 2.0 V Hysteresis – – 60 – mV Input Impedance – 10 20 40 kΩ ENABLE Input Select Input VIL (Selects VFB1) 4.9 ≤ VLIN ≤ 5.1 0.8 1.25 – V VIH (Selects VFB2) 4.9 ≤ VLIN ≤ 5.1 – 1.25 2.0 V SELECT Pull–Up SELECT = 0 V 10 24 50 kΩ 3.5 4.5 – V Floating Input Voltage – PIN FUNCTION DESCRIPTION PACKAGE PIN # SO–24L PIN SYMBOL 1 VIN Supply voltage. 2, 3 NC No connection. 4 VSW Collector of NPN power switch for switching regulator section. 5, 6, 7, 8, 17, 18, 19, 20 GND Connected to the heat removing leads. 9 VFB1 Feedback input voltage 1 (referenced to 1.25 V). 10 VFB2 Feedback input voltage 2 (referenced to 1.25 V). 11 SELECT 12 COMP Output of the transconductance error amplifier. 13 COSC A capacitor connected to GND sets the switching frequency. Refer to Figure 5. 14 WDI Watchdog input. Active on falling edge. 15 CDelay A capacitor connected to GND sets the Power On Reset and Watchdog time. 16 RESET RESET output. Active low if VLIN is below the regulation limit. If watchdog timeout is reached, a reset pulse train is issued. FUNCTION Logic level input that selects either VFB1 or VFB2. An open selects VFB2. Connect to GND to select VFB1. http://onsemi.com 4 CS5111 PIN FUNCTION DESCRIPTION (continued) PACKAGE PIN # SO–24L PIN SYMBOL 21 IBIAS A resistor connected to GND sets internal bias currents as well as the COSC and CDelay charge currents. 22 VLIN Regulated 5.0 V output from the linear regulator section. 23 VREG Input voltage to the linear regulator and the internal supply circuitry. 24 ENABLE FUNCTION Logic level input to shut down the switching regulator. TYPICAL PERFORMANCE CHARACTERISTICS 0 –10 IIN (mA) IREG – ILIN (mA) 4.5 4.0 –20 –30 3.5 0 20 60 40 80 –40 100 0 0.5 1.5 1.0 ILIN (mA) 2.0 ISW (A) Figure 2. 5.0 V Regulator Bias Current vs. Load Current Figure 3. Supply Current vs. Switch Current 1.4 180 160 1.2 140 Frequency (kHz) VSW (V) 1.0 0.8 0.6 0.4 120 100 80 60 40 0.2 0 20 0 0.5 1.0 1.5 0 2.0 0 ISW (A) 500 1000 1500 2000 2500 COSC (pF) Figure 4. Switch Saturation Voltage Figure 5. Oscillator Frequency (kHz) vs. COSC (pF), Assuming RBIAS = 64.9 k http://onsemi.com 5 3000 CS5111 CIRCUIT DESCRIPTION VREG Overvoltage + – R1 Linear Error Amplifier Q2 Q3 Q1 VLIN COUT = 100 µF ESR < 8.0 Ω R2 Current Limit R3 IBIAS Bandgap Reference RBIAS 64.9 kΩ 1.25 V Over Temperature R4 R5 CDelay RESET & Watchdog Timer RESET WDI Figure 6. Block Diagram of 5.0 V Linear Regulator Portion of the CS5111 5.0 V LINEAR REGULATOR edge of this watchdog signal which it expects to see within an externally programmable time (see Figure 7). The watchdog time is given by: The 5.0 V linear regulator consists of an error amplifier, bandgap voltage reference, and a composite pass transistor. The 5.0 V linear regulator circuitry is shown in Figure 6. When an unregulated voltage greater than 6.6 V is applied to the VREG input, a 5.0 V regulated DC voltage will be present at VLIN. For proper operation of the 5.0 V linear regulator, the IBIAS lead must have a 64.9 kΩ pull down resistor to ground. A 100 µF or larger capacitor with an ESR < 8.0 Ω must be connected between VLIN and ground. To operate the 5.0 V linear regulator as an independent regulator (i.e. separate from the switching supply), the input voltage must be tied to the VREG lead. As the voltage at the VREG input is increased, Q1 is turned on. Q1 provides base drive for Q2 which in turn provides base current for Q3. As Q3 is turned on, the output voltage, VLIN, begins to rise as Q3’s output current charges the output capacitor, COUT. Once VLIN rises to a certain level, the error amplifier becomes biased and provides the appropriate amount of base current to Q1. The error amplifier monitors the scaled output voltage via an internal voltage divider, R2 through R5, and compares it to the bandgap voltage reference. The error amplifier output or error signal is an output current equal to the error amplifier’s input differential voltage times the transconductance of the amplifier. Therefore, the error amplifier varies the base current to Q1, which provides bias to Q2 and Q3, based on the difference between the reference voltage and the scaled VLIN output voltage. tWDI 1.353 CDelayRBIAS Using CDelay = 0.1 µF and RBIAS = 64.9 kΩ gives a time ranging from 6.25 ms to 11 ms assuming ideal components. Based on this, the software must be written so that the watchdog arrives at least every 6.25 ms. In practice, the tolerance of CDelay and RBIAS must be taken into account when calculating the minimum watchdog time (tWDI). VREG RESET WDI VLIN tPOR Normal Operation Figure 7. Timing Diagram for Normal Regulator Operation If a correct watchdog signal is not received within the specified time a reset pulse train is issued until the correct watchdog signal is received. The nominal reset signal in this case is a 5 volt square wave with a 50% duty cycle as shown in Figure 8. CONTROL FUNCTIONS The watchdog timer circuitry monitors an input signal (WDI) from the microprocessor. It responds to the falling http://onsemi.com 6 CS5111 The POR delay (tPOR) is given by: 50% Duty Cycle VREG tPOR 1.353 CDelayRBIAS RESET CURRENT MODE PWM SWITCHING CIRCUITRY The current mode PWM switching voltage regulator contains an error amplifier with selectable feedback inputs, a current sense amplifier, an adjustable oscillator and a 1.4 A output power switch with antisaturation control. The switching regulator and external components, connected in a boost configuration, are shown in Figure 11. The switching regulator begins operation when VREG and VIN are raised above 5 volts. VREG is required since the switching supply’s control circuitry is powered through VLIN. VIN supplies the base drive to the switcher output transistor. The output transistor turns on when the oscillator starts to charge the capacitor on COSC. The output current will develop a voltage drop across the internal sense resistor (RS). This voltage drop produces a proportional voltage at the output of the current sense amplifier, which is compared to the output of the error amplifier. The error amplifier generates an output voltage which is proportional to the difference between the scaled down output boost voltage (VFB1 or VFB2) and the internal bandgap voltage reference. Once the current sense amplifier output exceeds the error amplifier’s output voltage, the output transistor is turned off. The energy stored in the inductor during the output transistor on time is transferred to the load when the output transistor is turned off. The output transistor is turned back on at the next rising edge of the oscillator. On a cycle by cycle basis, the current mode controller in a discontinuous mode of operation charges the inductor to the appropriate amount of energy, based on the energy demand of the load. Figure 12 shows the typical current and voltage waveforms for a boost supply operating in the discontinuous mode. WDI VLIN tPOR A B A: Watchdog waiting for low–going transition on WDI B: RESET stays low for tWDI time Figure 8. Timing Diagram When WDI Fails to Appear Within the Preset Time Interval, tWDI The RESET signal frequency is given by: 1 fRESET 2(tWDI) The Power On Reset (POR) and low voltage RESET use the same circuitry and issue a reset when the linear output voltage is below the regulation limit. After VLIN rises above the minimum specified value, RESET remains low for a fixed period tPOR as shown in Figures 9 and 10. VLIN 4.45 V 4.25 V RESET VR(LO) VR(PEAK) Notes: tPOR 1. Refer to Figure 5 to determine oscillator frequency. 2. The switching regulator can be disabled by providing a logic high at the ENABLE input. 3. The boost output voltage can be controlled dynamically by the feedback select input. If select is open, VFB2 is selected. If select is low, then VFB1 is selected. Figure 9. The Power On Reset Time Interval (tPOR) Begins When VLIN Rises Above 4.45 V (Typical) VLIN 5.0 V 4.25 V RESET 5.0 V tPOR Figure 10. RESET Signal Is Issued Whenever VLIN Falls Below 4.25 V (Typical) http://onsemi.com 7 CS5111 VIN VLIN VOUT VSW COMP IBIAS RBIAS 64.9 kΩ Base Drive Logic Current Sense Amplifier COSC Oscillator + – RS VREG Switcher Shutdown 1.25 V COMP Switcher Error Amplifier GND + – ENABLE Overvoltage COUT 1.4 A Bandgap Reference R1 VFB1 – + Multiplexer R2 VFB2 R3 SELECT Figure 11. Block Diagram of the 1.4 A Current Mode Control Switching Regulator Portion of the CS5111 in a Boost Configuration PROTECTION CIRCUITRY VSW If the input voltage at VREG is increased above the overvoltage threshold, the drive to the linear and switcher output transistors is shut off. Therefore, VLIN is disabled and VSW can not be pulled low. The current out of VLIN is sensed in order to limit excessive power dissipation in the linear output transistor over the output range of 0 V to regulation. Also, the current into VSW is sensed in order to provide the current limit function in the switcher output transistor. If the die temperature is increased above 160°C, either due to excessive ambient temperature or excessive power dissipation, the drive to the linear output transistor is reduced proportionally with increasing die temperature. Therefore, VLIN will decrease with increasing die temperature above 160°C. Since the switcher control circuitry is powered through VLIN, the switcher performance, including current limit, will be affected by the decrease in VLIN. VOUT VIN VSAT 0 t ISW IPeak t 0 ID IPeak t 0 Figure 12. Voltage and Current Waveforms for Boost Topology in CS5111 http://onsemi.com 8 CS5111 APPLICATION NOTES VINtON (VOUT VIN)tOFF DESIGN PROCEDURE FOR BOOST TOPOLOGY This section outlines a procedure for designing a boost switching power supply operating in the discontinuous mode. where the maximum on time is: VIN(MIN) tON(MAX) 1 VOUT(MAX) Step 1 Determine the output power required by the load. POUT IOUTVOUT (1) 2 f V 2 t L(MAX) SW(MIN) IN (MIN) ON (MAX) 2POUT Step 3 (2) FB1 Step 6 REQ VFB1 VOUT R1 REQ Determine the peak inductor current at the minimum inductance, minimum VIN and maximum on time to make sure the inductor current doesn’t exceed 1.4 A. (3) V t IPK IN(MIN) ON(MAX) L(MIN) and then calculate R2 where: V V VFB2 R2 R2 FB1 IR2 VFB1REQ Step 7 R3 REQ R2 Determine the minimum output capacitance and maximum ESR based on the allowable output voltage ripple. (5) COUT(MIN) VOUT ESR(MIN) R1 VFB1 IPK 8fVRIPPLE (10) VRIPPLE IPK (11) In practice, it is normally necessary to use a larger capacitance value to obtain a low ESR. By placing capacitors in parallel, the equivalent ESR can be reduced. R2 VFB2 REQ (9) (4) Then find R3, where: VR2 (8) where η = efficiency. Usually η = 0.75 is a good starting point. The IC’s power dissipation should be calculated after the peak current has been determined in Step 6. If the efficiency is less than originally assumed, decrease the efficiency and recalculate the maximum inductance and peak current. Next select the output voltage feedback sense resistor divider as follows (Figure 13). For VFB1 active, choose a value for R1 and then solve for REQ where: (7) Calculate the maximum inductance allowed for discontinuous operation: Choose COSC based on the target oscillator frequency with an external resistor value, RBIAS = 64.9 kΩ. (See Figure 5). For VFB2 active, find: 1 fSW(MIN) Step 5 Step 2 R REQ VOUT 1 1 V (6) Step 8 Compensate the feedback loop to guarantee stability under all operating conditions. To do this, we calculate the modulator gain and the feedback resistor network attenuation and set the gain of the error amplifier so that the overall loop gain is 0 dB at the crossover frequency, fCO. In addition, the gain slope should be –20 dB/decade at the crossover frequency. The low frequency gain of the modulator (i.e. error amplifier output to output voltage) is: R3 Figure 13. Feedback Sense Resistor Divider Connected Between VOUT and Ground Step 4 Determine the maximum on time at the minimum oscillator frequency and VIN. For discontinuous operation, all of the stored energy in the inductor is transferred to the load prior to the next cycle. Since the current through the inductor cannot change instantaneously and the inductance is constant, a volt–second balance exists between the on time and off time. The voltage across the inductor during the on cycle is VIN and the voltage across the inductor during the off cycle is VOUT – VIN. Therefore: IPK(MAX) VOUT VEA(MAX) VEA Lf RLOAD 2 (12) where: V G 2.4 V7 IPK(MAX) EA(MAX) CSA 2.3 A (13) 150 m RS The VOUT/VEA transfer function has a pole at: http://onsemi.com 9 CS5111 fp 1(RLOADCOUT) VOUT (14) and a zero due to the output capacitor’s ESR at: R1 fz 1(2ESR(COUT)) Since the error amplifier reference voltage is 1.25 V, the output voltage must be divided down or attenuated before being applied to the input of the error amplifier. The feedback resistor divider attenuation is: R2 M U X VFB2 Error Amplifier The error amplifier in the CS5111 is an operational transconductance amplifier (OTA), with a gain given by: GOTA gmZOUT + – R3 1.25 V VOUT C1 R4 SELECT Figure 14. RC Network Used to Compensate the Error Amplifier (OTA) IOUT VIN (17) A pole at point C: For the CS5111, gm = 2700 µA/V typical. One possible error amplifier compensation scheme is shown in Figure 14. This gives the error amplifier a gain plot as shown in Figure 15. For the error amplifier gain shown in Figure 15, a low frequency pole is generated by the error amplifier output impedance and C1 . This is shown by the line AB with a –20 dB/decade slope in Figure 15. The slope changes to zero at point B due to the zero at: fz 1(2R4C1) fp 1(R4C2) Step 9 Finally the watchdog timer period and Power on Reset time is determined by: tDelay 1.353 CDelayRBIAS (18) G A (19) offsets the zero set by the ESR of the output capacitors. An alternative scheme uses a single capacitor as shown in Figure 16, to roll the gain off at a relatively low frequency. Pole due to error amplifier output impedance and C1 fz = 1/(2πR4C1) fp = 1/(πR4C2) +G C B error amplifier gain Gain (dB) C2 (16) where: gm 1.25 V VFB1 (15) –20 dB/dec fp = 1/(πRLOADCOUT) fCO 0 modulator gain + feedback resistor divider attenuation –G fz = 1/(2πESR(COUT)) Figure 15. Bode Plot of Error Amplifier (OTA) Gain and Modulator Gain Added to the Feedback Resistor Divider Attenuation http://onsemi.com 10 (20) CS5111 VIN VOUT = 18 V, Select > 2.0 V VOUT = 16 V, Select < 0.8 V L = 33 µH COUT 88 µF VIN ENABLE NC VREG NC VLIN VSW IBIAS GND GND (2) GND 5.0 V 100 µF ESR < 8.0 Ω RBIAS 64.9 kΩ GND CS5111 R1 100 kΩ R2 946 Ω GND GND GND GND VFB1 RESET VFB2 CDelay Microprocessor (1) R3 7.5 kΩ SELECT COMP CDelay 0.1 µF WDI COSC COSC 390 pF CCOMP 0.33 µF Figure 16. A Typical Application Diagram with External Components Configured in a Boost Topology http://onsemi.com 11 CS5111 LINEAR REGULATOR OUTPUT CURRENT VS. INPUT VOLTAGE 100 ILIN (mA) ILIN (mA) 100 75 ΘJA = 55°C/W VIN = 14 V Max Total Power = 1.18 W 50 25 0 75 ΘJA = 35°C/W VIN = 14 V Max Total Power = 1.86 W 50 25 0 5 10 15 20 25 0 30 0 5 10 VREG (V) 15 20 25 30 VREG (V) Figure 17. The Shaded Area Shows the Safe Operating Area of the CS5111 as a Function of ILIN, VREG, and JA. Refer to Table 1 for Typical Loads and Voltages. Table 1. Worst Case Switcher Power Available (ΘJA = 55°C/W) (W) Worst Case Switcher Power Available (ΘJA = 35°C/W) (W) VREG (V) VIN (V) ILIN (mA) Linear Power Dissipation (W) 20 14 25 0.44 0.74 1.42 20 14 50 0.83 0.35 1.03 20 14 75 1.22 * 0.64 20 14 100 1.60 * 0.26 25 14 25 0.60 0.58 1.26 25 14 50 1.11 0.07 0.75 25 14 75 1.62 * 0.24 25 14 100 2.14 * * *Subjecting the CS5111 to these conditions will exceed the maximum total power that the part can handle, thereby forcing it into thermal limit. http://onsemi.com 12 CS5111 PACKAGE DIMENSIONS SO–24L DWF SUFFIX CASE 751E–04 ISSUE E –A– 24 NOTES: 1. DIMENSIONING AND TOLERANCING PER ANSI Y14.5M, 1982. 2. CONTROLLING DIMENSION: MILLIMETER. 3. DIMENSIONS A AND B DO NOT INCLUDE MOLD PROTRUSION. 4. MAXIMUM MOLD PROTRUSION 0.15 (0.006) PER SIDE. 5. DIMENSION D DOES NOT INCLUDE DAMBAR PROTRUSION. ALLOWABLE DAMBAR PROTRUSION SHALL BE 0.13 (0.005) TOTAL IN EXCESS OF D DIMENSION AT MAXIMUM MATERIAL CONDITION. 13 –B– 12X P 0.010 (0.25) 1 M B M 12 24X D J 0.010 (0.25) M T A S B S DIM A B C D F G J K M P R F R X 45 C –T– SEATING PLANE M 22X G K MILLIMETERS MIN MAX 15.25 15.54 7.40 7.60 2.35 2.65 0.35 0.49 0.41 0.90 1.27 BSC 0.23 0.32 0.13 0.29 0 8 10.05 10.55 0.25 0.75 PACKAGE THERMAL DATA Parameter SO–24L Unit RΘJC Typical 9 °C/W RΘJA Typical 55 °C/W http://onsemi.com 13 INCHES MIN MAX 0.601 0.612 0.292 0.299 0.093 0.104 0.014 0.019 0.016 0.035 0.050 BSC 0.009 0.013 0.005 0.011 0 8 0.395 0.415 0.010 0.029 CS5111 Notes http://onsemi.com 14 CS5111 Notes http://onsemi.com 15 CS5111 ON Semiconductor and are trademarks of Semiconductor Components Industries, LLC (SCILLC). SCILLC reserves the right to make changes without further notice to any products herein. SCILLC makes no warranty, representation or guarantee regarding the suitability of its products for any particular purpose, nor does SCILLC assume any liability arising out of the application or use of any product or circuit, and specifically disclaims any and all liability, including without limitation special, consequential or incidental damages. “Typical” parameters which may be provided in SCILLC data sheets and/or specifications can and do vary in different applications and actual performance may vary over time. All operating parameters, including “Typicals” must be validated for each customer application by customer’s technical experts. SCILLC does not convey any license under its patent rights nor the rights of others. SCILLC products are not designed, intended, or authorized for use as components in systems intended for surgical implant into the body, or other applications intended to support or sustain life, or for any other application in which the failure of the SCILLC product could create a situation where personal injury or death may occur. Should Buyer purchase or use SCILLC products for any such unintended or unauthorized application, Buyer shall indemnify and hold SCILLC and its officers, employees, subsidiaries, affiliates, and distributors harmless against all claims, costs, damages, and expenses, and reasonable attorney fees arising out of, directly or indirectly, any claim of personal injury or death associated with such unintended or unauthorized use, even if such claim alleges that SCILLC was negligent regarding the design or manufacture of the part. SCILLC is an Equal Opportunity/Affirmative Action Employer. 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