Micronote 1401 (AN-13) LX1686 Backlight CCFL Direct Drive Design Reference (353.87 kB)

Microsemi Application Note
ANAN-13
LX1686 Direct Drive
CCFL Inverter Design
Reference
George Henry
Microsemi – Power Management Group
L I N F I N I T Y
Copyright © 2000
Rev. 1.5, 9.01
D I V I S I O N
AN-13
LX1686 Direct Drive CCFL Inverter Design Reference
Contents
1.0 INTRODUCTION ........................................................................................................................................... 4
RangeMAX vs. Analog Dimming ................................................................................................................................................. 4
2.0 WHAT IS DIRECT DRIVE? ........................................................................................................................... 5
High Voltage Transformers for Direct Drive ................................................................................................................................ 5
Designing for Cost ........................................................................................................................................................................ 5
Lamp Strike Voltage Generation .................................................................................................................................................. 5
Standard Safety Features ............................................................................................................................................................ 6
RangeMAX Digital Dimming ........................................................................................................................................................ 6
3.0 A COMPARISON OF CONVENTIONAL “BUCK/ROYER” INVERTERS WITH
MICROSEMI’S “DIRECT DRIVE” TOPOLOGY ............................................................................................ 7
4.0 THE LX1686 CCFL CONTROLLER IC ......................................................................................................... 8
Block Diagram .............................................................................................................................................................................. 8
Dimming Control .......................................................................................................................................................................... 8
PWM Ramp Generator .................................................................................................................................................................. 8
PWM Controllers ........................................................................................................................................................................... 9
BIAS Generator ............................................................................................................................................................................. 9
Output Drivers .............................................................................................................................................................................. 9
Timing Diagrams and Wavforms ............................................................................................................................................... 10
4.2 FUNCTIONAL PIN DESCRIPTION ............................................................................................................. 11
4.3 PACKAGE DIMENSIONS ........................................................................................................................... 12
4.4 BIAS & TIMING EQUATIONS REFERENCE .............................................................................................. 12
5.0 DESIGNING INVERTER MODULES WITH THE LX1686 ............................................................................ 13
Getting Started ............................................................................................................................................................................ 13
Generating An On-Board VDD Supply For The LX1686 ........................................................................................................... 13
Driving The ENABLE Input ......................................................................................................................................................... 14
Logic Input Threshold Voltage ................................................................................................................................................... 14
Setting Main Oscillator Frequency ............................................................................................................................................ 14
Setting TRI_C Ramp Frequency ................................................................................................................................................ 14
Driving The BRITE Input ............................................................................................................................................................. 14
BRITE Pin DC Input Operating Voltage Range ......................................................................................................................... 15
Pointers Using PWM Inputs For Brightness Control ............................................................................................................... 15
Synchronizing The Digital Dimming Burst Rate ....................................................................................................................... 15
AFD Circuit Response Time ....................................................................................................................................................... 15
Maximum VCO Frequency ......................................................................................................................................................... 15
Compensating The Phase Detector ........................................................................................................................................... 16
PLL Damping Factor ................................................................................................................................................................... 16
High Frequency Attenuation Capacitor C7 ............................................................................................................................... 16
Selecting U2, The Power FET ..................................................................................................................................................... 16
AOUT And BOUT Connections .................................................................................................................................................. 16
The High Current Power Stage .................................................................................................................................................. 16
Configuring The High Voltage Output ....................................................................................................................................... 16
High Voltage Feedback ............................................................................................................................................................... 17
Short-Circuit Protection ............................................................................................................................................................. 17
Open Lamp Sensing ................................................................................................................................................................... 17
2
Copyright © 2000
Rev. 1.5 9/01
AN-13
LX1686 Direct Drive CCFL Inverter Design Reference
Lamp Current Regulation Loop ................................................................................................................................................. 18
Compensating The Error Amplifiers .......................................................................................................................................... 18
Lowest Cost Configuration ........................................................................................................................................................ 18
BRITE Input Signal Conditioning .............................................................................................................................................. 19
A Broken Lamp Shut Down Circuit ........................................................................................................................................... 19
6.0 TRANSFORMER DESIGN and SELECTION CRITERIA............................................................................ 21
Bifilar-Wound Primary ................................................................................................................................................................ 21
Leakage Inductance ................................................................................................................................................................... 21
Open Circuit Series Resonance ................................................................................................................................................ 21
Sufficient “Q” at Resonance ...................................................................................................................................................... 21
Transformer Turns Ratio ............................................................................................................................................................. 21
Estimating Minimum Primary Voltage ....................................................................................................................................... 21
Estimating Maximum Secondary Running Voltage*................................................................................................................. 22
Turns Ratio .................................................................................................................................................................................. 22
Required Number of Secondary Turns ...................................................................................................................................... 23
Required Number of Primary Turns ........................................................................................................................................... 23
Determination of Primary Wire Gauge ...................................................................................................................................... 23
Conclusion of Winding Analysis ............................................................................................................................................... 24
Copyright © 2000
Rev. 1.5 9/01
3
AN-13
LX1686 Direct Drive CCFL Inverter Design Reference
1.0 INTRODUCTION
Two equally important but technically diverse components determine
overall performance, reliability and cost of CCFL inverters. One is the
control integrated circuit; the other is the high voltage transformer.
Since Microsemi is the only manufacturer that designs and manufactures
both of these critical components, we alone are able to develop
completely optimized modular solutions.
Microsemi Microelectronics has created the next generation in CCFL
inverter topology with its patented new LX1686 Direct Drive integrated
circuit architecture. The LX1686 backlight controller IC provides all
the control functions necessary to implement Microsemi Direct Drive
CCFL inverters . CCFLs are used for back or edge lighting of liquid
crystal flat panel displays (LCD’s) and typically find application in
notebook computers, web browsers, automotive and industrial
instrumentation, and entertainment systems. This IC can be used to
control single or multiple lamp configurations.
LX1686 FEATURES
• RangeMAXTM Wide Range Dimming (>100:1)
• Dimming Burst Rate Synchronizable to Display Video Frequency
• High Voltage Feedback Loop Regulates Maximum Open Lamp and
Minimum Strike Voltages
• Transformer Protected from Overheating During Lamp Striking
• Micro-Amp Sleep Mode
• User-Programmable Fixed Frequency Operation
• Under-Voltage Lockout Feature with Power-Up Reset
• Built-In Soft-Start Feature
• Operates with 3.3V or 5V Power Supplies
• 50mA Output Drive Capability
APPLICATIONS
This application note provides a guide for using this new control circuit
in your module and magnetic designs. A comlete functional description
of the chip is included, along with an applications information section
in which typical module schematics are explained as an example to
designers.
•
•
•
•
Direct Drive offers distinct advantages over conventional Buck/Royer
inverters. These advantages are illustrated with practical examples
and comparisons so that you may make informed decisions about
which technology is best for your application.
BENEFITS
Safety and reliability features include dual feedback loops that permit
regulation of maximum strike voltage as well as lamp current.
Regulating maximum lamp voltage permits the designer to
simultaneously provide for ample worst case lamp voltage while
conservatively limiting maximum open circuit voltage.
An innovative new strike voltage generation technique enables the
module designer to optimize high voltage transformer design for
maximum efficiency while the lamp is ignited. The high voltage drop
on the output ballast capacitor needed for Royer oscillators is much
less, reducing transformer size and power dissipation.
• Extremely High Efficiency from 3.3V, 5V and 12V Power Supplies
• Lower Cost than Conventional Buck/Royer Inverter Topologies
• Patented Strike Voltage Generation Method Ensures Lamp Ignition
While Increasing Efficiency
• Fool-Proof Output Voltage Regulation Prevents Over-Voltage
Failures
RangeMAX vs. Analog Dimming
5
Power Used (Watts)
The LX1686 backlight controller IC includes a PWM (Pulse Width
Modulation) controlled lamp current burst circuit that can provide
greater than a 100:1 dimming range from a simple zero to 2.5V
potentiometer input. The lamp current burst rate may be easily
synchronized to the LCD panel’s frame rate to prevent interference
from optical beating between the two frequencies.
Notebooks
Instrumentation Displays
Desktop Computer Monitors
Low Ambient Light Displays (Used in Aircraft, Automobiles and
Hand-Held Equipment)
Standard Analog
Dimming Inverter
4
3
Microsemi's
RangeMAX
2
1
0
30
45
60
75
90
105
Light Output (Nits)
4
Copyright © 2000
Rev. 1.5 9/01
AN-13
LX1686 Direct Drive CCFL Inverter Design Reference
2.0 WHAT IS DIRECT DRIVE?
“Direct Drive” refers to the ability of Microsemi's new architecture to
eliminate the inductor and resonant capacitors necessary to implement
a conventional Royer oscillator based inverter solution. Instead, Direct
Drive topology uses a fixed frequency PWM control circuit connected
directly to a high voltage transformer primary via a pair of N-FET drivers.
Removing these costly and power-hungry components simultaneously
improves module cost, efficiency and size.
A two transistor N channel drive scheme was selected over popular
bipolar and complementary P/N channel FET drives for three significant
reasons:
• Using ground referenced transistors in conjunction with “push-pull”
transformer operation permits the IC to be implemented with a low
cost 5V fabrication process such as CMOS. This process permits
the smallest die size, very high performance, and direct compatibility
with 5V and 3.3V system power buses. The IC can interface through
external N-FETs to any system voltage desired by simply changing
the high voltage transformer turns ratio. Thus, an operating input
voltage range for the module from 3V to more than 50V is possible.
• N-FETs are significantly more efficient switches than bipolar
transistors or P-FETs of equal size and cost.
• Dual N-FETs are readily available in small surface mount packages
at prices that compete favorably with the installed cost of bipolar
transistors and their required additional circuit components.
Direct Drive topology is a non-resonant, fixed frequency PWM
regulation method for operating CCFLs. The LX1686 allows a wide
choice of operating frequencies to match the lamp’s most efficient
operating point, and to minimize high frequency interference.
High Voltage Transformers for Direct Drive
Direct Drive’s push-pull transformer design provides important technical
advantages when used in low voltage applications such as notebook
computers and hand-held battery operated products.
Because of its dual primary winding construction, the voltage
impressed across the primary in Direct Drive modules is twice the
supply voltage. This is electrically equivalent to a four transistor “full
or H-bridge” drive configuration and makes for very efficient operation
at low voltages. Direct Drive topology enables efficient inverters that
can run directly from 3.3 and 5V logic supplies now common in LCD
panels.
Unlike Royer oscillator implementations, Microsemi's new approach for
lamp strike voltage generation relieves the transformer from operating
continuously at full lamp strike voltage once the lamp is ignited. Direct
Drive transformers can be optimized for normal operation where they
spend most of their life. Transformers can be smaller while system
reliability is improved because the extra high voltage required to strike
a CCFL is only present for the instant it is required. High voltage
corona discharge, which gradually destroys insulation material, can
be more easily avoided with Direct Drive designs (see the detailed
description of strike voltage generation below).
Copyright © 2000
Rev. 1.5
9/01
Microsemi has developed completely new transformers to be used with
our LX1686 control IC. The first two are 6 watt units that can drive
lamps having strike voltages of up to 1800Vrms. The smaller of these
permits modules to be built with maximum profiles of 10.5mm wide by
6mm high while operating at a very conservative 18V per mil voltage
stress in air. Microsemi develops and works with other world class
manufacturers to develop transformers for other applications as
needed. A two lamp magnetic now in development will support up to
2500Vrms strike and 12W operating power. A magnetic design guide
is included in this note to help you design your own magnetics if ours
do not meet your exact needs.
Designing for Cost
Continuous cost reduction is a way of life in the computer industry.
Higher performance at ever decreasing cost is fundamental for success.
Direct Drive topology takes this principle and applies it to LCD
backlighting. For fixed input voltage dimmable inverter applications,
the Microsemi solution provides the most efficient, least expensive, and
smallest size available. Coupled with a step-down input voltage
regulator, Direct Drive can handle extremely wide input voltage range
applications while providing higher reliability and more features per
dollar than older technologies.
Lamp Strike Voltage Generation
Prior to the introduction of Direct Drive technology, the need to generate
strike voltages more than twice the operating voltage of a lamp have
limited transformer size reduction. The number of lines of flux required
to generate strike voltage governs transformer minimum size. The
number of lines of flux available in a magnetic structure is directly
proportional to both core cross-section (Ae), and operating frequency.
Instead of increasing the physical size of the transformer, Microsemi
chose to increase frequency, but only during lamp strike time when
very high voltage output is needed. This allows the transformer to be
sized for normal run voltages, resulting in a smaller design for a given
power level. Smaller core structures have lower losses which helps
improve inverter efficiency.
The LX1686 integrated circuit ramps operating frequency slowly up
and down over a user adjustable range when the open lamp sense
input (OLSNS) indicates the lamp is not ignited. The high voltage
transformer and output capacitance have an unloaded self-resonant
frequency that is higher than normal operating frequency. As strike
frequency is increased, unloaded resonance is approached, resulting
in a resonant rise of voltage across the output capacitance of the
lamp, and the lamp is ignited. Since the unloaded resonance circuit
has a high Q, typically in the range of 5 to 10, it is easy to generate
very high strike voltages. At the same time, transformer flux density is
maintained at low levels due to the higher frequency, preventing
magnetic saturation.
This technique also solves the problem of the lamp’s parasitic
capacitance to ground forming a voltage divider with the ballast
capacitor placed between the transformer secondary and the lamp.
5
AN-13
LX1686 Direct Drive CCFL Inverter Design Reference
In Royer oscillator designs, the lamp ballast capacitor is usually in the
range of 12 to 22 pico Farad in order to drop excess transformer output
voltage after the lamp has ignited. Direct Drive designs reduce
transformer output voltage after ignition, permitting ballast capacitance
values to be much larger.
The 100nF ballast (In this application it is really a DC bypass cap)
typically used in Direct Drive transfers far more of the available
transformer voltage to the lamp. For example, if a 22pF the available
transformer voltage to the lamp. For example, if a 22pF ballast capacitor
is used for a backlight assembly with 10pF of parasitic capacitance
across the lamp, only 69% of the voltage generated by the transformer
is available to strike the lamp. If this same back light assembly uses a
100nF ballast, 100% of available voltage is impressed across the
lamp at strike time. Said another way, the Direct Drive transformer
needs to develop only 6% more output than maximum lamp voltage,
while the Royer design must develop 31% more! Clearly, strike capability
is enhanced while transformer size and reliability are improved.
Standard Safety Features
Microsemi's new Direct Drive controller includes an active open circuit
voltage regulation feedback loop to prevent hazardous voltages, even
if the lamp is removed from the circuit. When an open lamp condition
is detected, the controller automatically enters strike mode as
described above, and at the same time operates at 50% duty cycle to
limit average power dissipation of the module to levels that can be
maintained indefinitely. The module designer can select the open
circuit voltage regulation point by choosing values of a non-dissipative
capacitor voltage divider placed on the output terminals.
In Digital Dimming mode, the same analog brightness control voltage
is processed to modulate lamp current duty cycle. In this case, lamp
current is either “on” at a user determined rms value, or off. The ontime duty cycle determines brightness of the lamp. Duty cycle can be
controlled from nearly zero to 100% by the analog control voltage.
Lamp brightness adjustment by duty cycle control is possible without
the turn-on stress that occurs when the lamp is initially ignited, because
the burst repetition rate is high enough to prevent lamp gasses from
de-ionizing between current pulses.
The LX1686 “bursts” current on and off in a smooth and precisely
controlled manner, eliminating turn-on overshoot that could shorten
lamp life and pulse-to-pulse jitter that could cause flickering. An onchip PLL can synchronize lamp current bursts to an external sync
pulse connected to the FVERT pin. This feature is important in high
quality displays because it prevents optically visible beat frequencies
between the lamp burst rate and the video frame rate. When an
external sync pulse is used, the PLL multiplies the external frequency
by 2 resulting in a burst frequency twice the sync input frequency. If
the FVERT pin is left floating, lamp current burst frequency will free
run at approximately 260Hz.
Lamp short circuit protection is inherent because lamp current is
regulated even when lamp voltage is zero. Shorts from either lamp
terminal to ground may also be protected by sensing all or part of the
lamp current at the low voltage end of the transformer secondary
winding.
RangeMAX Digital Dimming
The LX1686 provides both current amplitude modulated dimming
circuitry, which typically achieves between 3:1 and 5:1 dimming range
and “Range MAX” time modulated lamp current circuitry that can easily
achieve dimming ratios of 100:1. While duty cycle control of CCFL
lamp current is not new, the LX1686 is the first CCFL control IC to
completely integrate the function. The user interface for both dimming modes is the familiar DC control voltage or potentiometer.
Amplitude modulation brightness control differences a 0 to 2.5V (or
optionally a 2.5 to 0V) analog brightness control voltage from a voltage
developed by sampling current flow through the lamp. The resulting
error signal is used to regulate lamp current amplitude between user
definable minimum and maximum values.
6
Copyright © 2000
Rev. 1.5 9/01
AN-13
LX1686 Direct Drive CCFL Inverter Design Reference
3.0 A COMPARISON OF CONVENTIONAL “BUCK/ROYER” INVERTERS WITH
MICROSEMI'S “DIRECT DRIVE” TOPOLOGY
BUCK / ROYER ADVANTAGES:
ROYER OSCILLATOR
BALLAST
CAP 22pF
V BATTERY
PPS CAP
LAMP
1. Step-down voltage regulator (Buck regulator) adjusts input voltage to Royer
oscillator, providing both line and load regulation.
2. Lamp brightness is extremely insensitive to both static and dynamic input
voltage changes. Conversion efficiency is fairly constant across input
voltage range.
3. Self-resonant Royer oscillator (transformer, ballast cap, PPS cap) provides
low crest factor sine wave current waveform to the lamp. CCFLs operate
most efficiently with low (1.4 to 1.6) crest factors.
BUCK / ROYER DISADVANTAGES:
1. Requires more components, resulting in higher cost and larger size.
2. Four power semiconductors, two power inductors and one PPS (Poly
Phenelene Sulfide) high current capacitor burn power, reducing conversion
efficiency. Very high quality components can be selected to regain some
of these power losses, but high cost premiums and larger size must be
paid.
3. Open circuit voltage is very difficult to limit due to multiple resonances in
the power circuit, lack of voltage feedback, and lack of frequency control
from the controller IC. These circuits are prone to arcing and self destruction
when operated open circuit (lamp unplugged or broken).
4. Buck regulator and Royer oscillator operate asynchronously at different
frequencies, making EMI and RFI more difficult to control.
INDUCTOR
SCHOTTKY
I MONITOR
SHTDWN
BRT
CONTROLLER
BUCK REGULATOR
Figure 1: Buck/Royer Simplified Circuit
DIRECT DRIVE ADVANTAGES:
BALLAST CAP
100nF
V BATTERY
LAMP
1. Single stage PWM power conversion requires only two power transistors
to provide line and load regulation. A very high efficiency dual N-Channel
FET in an SO-8 package handles up to 10W.
2. Non-resonant, fixed frequency drive eliminates inductor and high current
resonant capacitor, reducing cost and size while increasing conversion
efficiency.
3. Simplified high voltage transformer has one less winding and two less
pins (smaller form factor = lower cost).
4. Higher value ballast cap requires less transformer output voltage while
lamp is ignited for additional efficiency and/or smaller size transformer is
realized.
5. Active open circuit output voltage regulation is achieved via non-dissipative
capacitor feedback from the transformer, eliminating open circuit hazards
forever without costly (and sometimes ineffective) thermal protection devices
being thermally bonded to the power transformer.
V MONITOR
V SYNC
ENABLE
I MONITOR
BRITE
CONTROLLER
DIRECT DRIVE DISADVANTAGES:
1. PWM regulation causes high lamp current crest factors when input voltage
Figure 2: Direct Drive Simplified Circuit
is more than 1.5 times the design minimum. This reduces nits/watt efficiency
at high-line to approximately the same as a typical Buck/Royer design. At
low-line (battery voltage levels) efficiency is 10-30% higher than Buck/
Royer inverters. Direct Drive efficiency is so high at 5V and 3.3V that it
makes up for Buck regulator losses in notebook systems, yielding
comparable overall system efficiency while completely eliminating the
disadvantage of high crest factors at high input voltages.
Copyright © 2000
Rev. 1.5 9/01
7
LX1686 Direct Drive CCFL Inverter Design Reference
AN-13
4.0 THE LX1686 CCFL CONTROLLER IC
C
Block Diagram
B R T
IS N S
P D _ C R
C
R
P D
C
P D
F V E R T
2 .5 V
A u to
F re q .
D e te c t
A F D _ C
C
V C O M P
V o lta g e
C o m p a ra to r
O u tp u t
S te e r in g
L o g ic
Q
C L K
O U T
1 .5 V
O p e n L a m p
C o m p a ra to r
V D D
T r ia n g le
W a v e
G e n e ra to r
0 -3 V
Ig n ite
E N A
P W R _ G D
2 .5 0 V
V D D _ S W
3 0 0 m V
2 5 5 m V
P W M
R a m p
G e n e ra to r
D im m in g C o n tr o l
R A M P _ C
C
R A M P
T R I_ C
R A M P _ R
R
R
R A M P 2
V S S _ P
V D D
E N A B L E
I_ R
R i
B R T _ P O S
D IG _ D IM
B O U T
B ia s G e n e r a to r ,
U V L O , a n d V R EF
5 0 k
V D D
B
R A M P _ D O U T
R A M P _ C
B R IT E
5 0 k
A O U T
Q
IC O M P
C u rre n t
C o m p a ra to r
B u rs t
C o m p a ra to r
E N A
V D D _ P
A
V C O M P
IA M P
0 .5 - 2 .5 V
V C O
O u tp u t
D r iv e r s
P W M
C o n tr o lle r s
C u rre n t
E rro r A m p
V C O _ C
IC O M P
V A M P
1 .2 5 V
0 V
V C O
A F D
C
IC O M P
V o lta g e
E rro r
A m p
P h a s e
D e te c to r
P D C
C
V C O M P
V S N S
C
V S S
O L S N S
T R I
R A M P 1
V D D
Figure 3: LX1686 Simplified Block Diagram
Dimming Control
• Performs the frequency locking for a 2VPP ramp oscillator with
oscillation frequency that locks to the LCD display vertical scanning
frequency. Frequency locking is done by a 3rd order phase-lock
loop formed by the phase detector, VCO and the divide-by-2 TF/F.
The phase detector is a 3-state charge pump type that allows easy
filtering and loop compensation. The VCO is a voltage-controlled
ramp oscillator with fixed output voltage levels (0.5 to 2.5V). When
there is no external FVERT detected by the Auto-Freq-Detector,
the VCO will oscillate at 5.5/10.5 its maximum frequency. The VCO
can be forced to oscillate at 5.5/10.5 of its maximum oscillation
frequency by grounding Pin AFD_C.
• Performs BRITE voltage inversion (determined by BRT_POS pin
voltage) and passes the converted voltage to BRT for analog and
digital dimming. The BRITE signal is buffered by a MUX before
connection to the internal op amp so that during SLEEP Mode or
Power-Down, BRITE is isolated from internal circuitry to ensure no
chip biasing occurs through the BRITE pin.
• Performs the generation of a digital dimming PWM signal derived
from the VCO’s ramp output with the burst comparator and the
external analog DC input (BRITE) as shown in the timing diagram
Figure 4. When digital dimming is selected, this signal is used to
generate a 2.5V pulse muxed to port BRT. When analog dimming
is selected, then either BRITE’s DC voltage or its inversion will be
muxed to the BRT port.
8
• Pin BRT_POS is used to control BRITE polarity during digital and
analog dimming. When BRT_POS is “1”, the regulated lamp current
will be proportional to BRITE’s voltage. When “0”, the current will
be inversely proportional to BRITE’s voltage.
PWM Ramp Generator
• Performs the generation of a 2 VPP ramp oscillator whose oscillation frequency can be changed through the voltage imposed
on the input port TRI_C as shown in Figure 5. When TRI_C voltage
is below 1.5V, the ramp oscillates at the normal run frequency.
When TRI_C is above 1.5V, the ramp oscillation frequency
will be proportional to VTRI_C. When TRI_C is at 2.25V, the ramp
will oscillate at a higher frequency, depending on the R RAMP value.
This ramp oscillator provides 2 timing signals for the controller:
analog ramp output RAMP_C and digital ramp output RAMP_DOUT.
Both of these outputs are used by the controller PWM block, in
conjunction with the dimming output BRT, to create two digital PWM
outputs (AOUT and BOUT) that operate in push-pull fashion.
• Performs frequency sweeping during CCFL startup mode (pre-strike)
and run mode (post-strike). The tasks are performed and controlled
by an approximately 10Hz triangular wave generator. The triangular
analog output (VTRI_C) swings from 0.75 to 2.25V. Normally, VTRI_C
will be reset at 0V, waiting to be ramped up. When the lamp is off,
Copyright © 2000
Rev. 1.5 9/01
AN-13
LX1686 Direct Drive CCFL Inverter Design Reference
detected by pin OLSNS, the triangular wave generator will start to
ramp up. It takes about 25ms to ramp up to 1.5V; this time is used
for the PWM block to settle out such that it can perform lamp current
regulation. After reaching 1.5V, VTRI_C will continue to ramp up and
increase the RAMP_C frequency. Increase in ramp frequency will
increase lamp voltage due to the CCFL backlight transformer
parasitic LC resonance. Increase in lamp voltage will in turn improve
lamp ignition probability. After the lamp ignites, detected again by
OLSNS, VTRI_C will start ramping down and decrease the RAMP
frequency slowly back to nominal. Lamp ignition is indicated by an
internal signal called IGNITE. The lamp striking timing and sequence
are illustrated in Figure 6. If the lamp fails to ignite when VTRI_C
reaches 2.25V, VTRI_C will ramp down. During this ramp down time
AOUT and BOUT signals are turned OFF to prevent transformer
overheating.
• Both the ramp generator and the triangular wave generator are reset
during low power supply voltage.
PWM Controllers
• Performs voltage and current regulation functions. Timing is shown
in Figure 7.
• Performs transformer output voltage regulation by comparator
VCOMP and error amplifier VAMP. The error amplifier generates
an error voltage derived from the voltage difference between VSNS
and the internal 1.25V reference. The error voltage is compared
with RAMP_C by VCOMP to generate two PWM signals (A, B) that
drive two output buffers. These two buffers drive two external power
FET switches that can increase or decrease the transformer output
voltage. By feeding back the transformer voltage through pin VSNS,
a negative feedback loop is formed to regulate the maximum
transformer output voltage to a predetermined value.
• Performs lamp current regulation by comparator ICOMP and the
error amplifier IAMP. The error amplifier generates an error voltage
derived from the voltage difference between ISNS and BRT. The
error voltage is compared by ICOMP with RAMP_C to generate
two PWM signals (A, B) that drive the same two output buffers. By
feeding back the lamp current through a sense resistor network to
pin ISNS, a negative feedback loop is formed to regulate the lamp
current to a set value determined by BRT voltage.
• All PWM flip-flops are reset and the VCOMP pin is discharged during
power-up.
Copyright © 2000
Rev. 1.5 9/01
BIAS Generator
• Performs the power-down functions controlled by the ENABLE input.
The power-down mode is activated by forcing ENABLE low. During
the power-down mode, internal power supply voltage VDD_SW is
turned off and no DC power is available to any internal circuitry
except the ENABLE circuitry. The main source of Sleep Mode
operating current is from the ENABLE pin internal pull-up resistor
(100k to VDD).
• Performs the Under-Voltage-Lock-Out (UVLO) function. Internal
ports PWR_BD and PWR_GD are used to indicate whether VDD
voltage is acceptable for reliable internal circuit operation. PWR_BD
and PWR_GD are valid when VDD voltage is above 1.3V. VDD_P
voltage is not monitored.
• Performs voltage and current bias generation. An internal 2.5V
voltage regulator for precision voltage biasing is generated by a
bandgap circuit. An internal precision current bias is generated
through an external resistor RI at pin I_R. This precision current
bias is copied four times and distributed to four circuit blocks.
Output Drivers
• Performs buffering function for the signals A and B from the PWM
block.
• Port PWR_BD is used to ensure pins AOUT and BOUT will stay
low during power startup or when VDD voltage is below the UVLO
threshold (about 2.8 V).
• Pins VDD_P and GND_P are used to isolate high-current power
and ground from the low signal power and ground terminals (VDD,
GND). This is done to reduce switching noise coupling.
9
LX1686 Direct Drive CCFL Inverter Design Reference
AN-13
Timing Diagrams and Wavforms
F V E R T
D IV -2
(In te rn a l N o d e )
V C O _ C L K
(In te rn a l N o d e )
B R IT E
V C O _ C
B R T
Figure 4: Digital Dimming Timing Diagram
2 8 0 k
F re q u e n c y (H z )
3 2 0 k
2 8 0 k
2 4 0 k
2 .6 2 5
2 0 0 k
2 .2 5
1 6 0 k
1 2 0 k
2 4 0 k
1 .8 7 5
R A M P _ C
2 0 0 k
1 .1 2 5
V tr i_ c
1 6 0 k
0 .7 5
1 2 0 k
L a m p D id N o t Ig n ite L
T R I_ D N
L a m p Ig n ite d
0 .3 7 5
(In te rn a l N o d e )
0
V tr i_ c (v o lts )
R a m p F re q u e n c y
3 6 0 k
0L
0 .5L
1 .0L
1 .5L
2 .0L
2 .5
T r ia n g u la r W a v e G e n e r a to r
O u t p u t V o l t a g e ( V T R I_ C ) , V D C
Figure 5: RAMP_C Frequency vs. VTRI_C
3 .0
IG N IT E
(In te rn a l N o d e )
0
5 0 m s L
1 0 0 m s L
1 5 0 m s L
2 0 0 m s
T im e
Figure 6: Lamp Ignition Timing Diagram
IC O M P
o r
V C O M P
R A M P _ C
R A M P _ D O U T
(In te rn a l N o d e )
A O U T
B O U T
Figure 7: Direct Drive PWM Regulation TIming Diagram
10
Copyright © 2000
Rev. 1.5 9/01
LX1686 Direct Drive CCFL Inverter Design Reference
AN-13
4.2 FUNCTIONAL PIN DESCRIPTION
Pin
Number
Pin
Designator
Description
1
AOUT
Output driver A.
2
VSS_P
Power ground for output drivers only.
3
VSS
4
AFD_C
5
RAMP_C
Connects to external capacitor CRAMP for setting Direct Drive PWM operating frequency.
6
RAMP_R
Connects to external resistor RRAMP for setting Direct Drive PWM operating frequency.
7
FVERT
Vertical frequency reference digital input. Has internal pull down.
8
PD_CR
Phase Detector Filter. Part of phase lock loop. Connects to external capacitor and resistor network.
9
VCO_C
Connects to external capacitor CVCO.
10
BRT_POS
11
BRITE
12
DIG_DIM
Digital Dimming Enable internal pullup. Leave open or pull up to VDD for operating in digital dimming mode.
Connect to ground for analog dimming mode.
13
ENABLE
Chip Enable internal pullup. High enables the chip. Low disables.
14
I_R
Current Reference Resistor. External resistor to ground (Ri) determines internal reference currents.
15
BRT
Current Error Amplifier non-inverting input.
16
VCOMP
17
VSNS
Voltage Error Amplifier inverting input.
18
ICOMP
Current Error Amplifier output. Connects to external frequency compensation capacitor CICOMP. CICOMP not needed
for amplifier stability.
19
ISNS
20
OLSNS
Open Lamp Sense Input. Lamp assumed ingnited if VOLSNS ≥ 300mV.
21
TRI_C
Connects to external capacitor CTRI for setting strike frequency ramp slope.
22
VDD
23
VDD_P
Dedicated VDD for output buffers only.
24
BOUT
Output driver B.
Signal ground.
Connects to an external cap, CAFD. Forcing to ground or VDD will make the VCO oscillate at approximately 50% of
the maximum VCO frequency. Forcing to VDD/2 will make the VCO oscillate at 2x the FVERT frequency.
Brightness polarity control. Has internal pullup. Leave open or pull up to VDD for dimming brightness proportional
to BRITE voltage, connect to ground for brightness inversely proportional to BRITE voltage.
Analog voltage input for brightness control.
Voltage Error Amplifier output. Connects to external frequency compensation capacitor CVCOMP. Controls soft-start
timing. CVCOMP not needed for amplifier stability.
Current Error Amplifier inverting input.
VDD
AOUT
VSS_P
VSS
AFD_C
RAMP_C
RAMP_R
FVERT
PD_CR
VCO_C
BRT_POS
BRITE
DIG_DIM
1
24
2
23
3
22
4
21
5
20
6
19
7
18
8
17
9
16
10
15
11
14
12
13
BOUT
VDD_P
VDD
TRI_C
OLSNS
ISNS
ICOMP
VSNS
VCOMP
BRT
I_R
ENABLE
PW PACKAGE
(Top View)
Figure 8: Pin Diagram
Copyright © 2000
Rev. 1.5
9/01
11
LX1686 Direct Drive CCFL Inverter Design Reference
AN-13
4.3 PACKAGE DIMENSIONS
3 2 1
E P
E
D
F
A H
SEATING PLANE
B
L
G
M
C
Dim
Millimeters
Min.
Max.
Inches
Min.
Max.
A
B
C
D
E
F
G
H
L
M
P
*LC
0.85
0.95
0.19
0.3
0.09
0.2
7.7
7.9
4.3
4.5
0.65 BSC
0.05
0.15
1.1
0.5
0.75
0
8
6.4 BSC
0.1
0.033
0.037
0.007
0.012
0.0035
0.008
0.303
0.311
0.169
0.177
0.025 BSC
0.002
0.005
0.0433
0.02
0.03
0
8
0.252 BSC
0.004
* Lead Coplanarity
Note:
1. Dimensions do not include mold flash or protrusions; these shall not exceed 0.15mm (.006") on any side. Lead dimension shall not include solder
coverage.
Figure 9: PW (TSSOP) 24-pin Package Dimensions
4.4 BIAS & TIMING EQUATIONS REFERENCE
Equation 1: I_PD = I_VCO = I_RAMP = I_TRI
1.0V
[A]
Ri
Where I_PD, I_VCO, I_RAMP and I_TRI are internal nodes. Capacitor
charge currents flowing out of PD_CR, VCO_G, RAMP_C and TRI_C
are various multiples of the current flowing in Ri. These multiples are
taken into account in the following equations.
Equation 5: Maximum VCO Frequency
1
[Hz]
(5 ⋅ Ri ⋅ C VCO )
FVCOMAX =
Equation 6: Maximum Phase Detector Output Current
IPD =
Equation 2: Triangular Wave Generator Frequency
1
[A]
10 ⋅ Ri
Equation 7: PLL PD High-Frequency Attenuation Capacitor
FTRI
1
[Hz]
=
(30 ⋅ Ri ⋅ C TRI )
Equation 3: Ramp Generator Frequency, FRAMP in Hz
If VTRI_C ≤ 1.5V then
(1 + VDD/20)
= FO
FRAMP = 0.72 ⋅
(Ri ⋅ CRAMP + 0.5 sec)
CPDC = 0.1 · CPD
Equation 8: PLL Zero Time Constant
TZ = RPD · CPD [sec]
Equation 9: PLL VCO Gain
K VCO = 2 ⋅
If VTRI_C > 1.5V then
FRAMP = (1 + 0.75 ⋅ (VTRI_C − 1.5) ⋅ (N − 1)) ⋅ FO
FRAMP _ MAX = N ⋅ FO
where
FO = FRAMP during run-mode (at VTRI_C < 1.5V)
N = Maximum FRAMP multiplier
Equation 4: N = Maximum Frequency Multiplier
π
(Ri ⋅ C VCO )
Equation 10: PLL Natural Frequency
WN = (1/2)⋅
WN =
(2 ⋅Ri ⋅
K VCO
[radian]
(2 ⋅ π ⋅ N ⋅ Ri ⋅ CPD )
1
N ⋅ C VCO ⋅ CPD
) [Hz]
Equation 11: PLL Damping Factor
VDD ⋅ Ri
RRAMP1 =
4(N − 2)
RRAMP2 =
12
8
VDD ⋅ Ri
3 (N − 1)⋅ VDD − 4 ⋅ (N − 2)
T 
DF =  Z  ⋅ WN
 2 
DF = 0.5 · RPD · CPD · WN [see Equation 8]
Copyright © 2000
Rev. 1.5 9/01
LX1686 Direct Drive CCFL Inverter Design Reference
AN-13
Equation 12: PLL Approximated Pull-In Time
TP =
(16 ⋅ π )
[sec]
WN
Equation 13: Auto-Frequency Detection Response Time
TD_AFD = 2,000,000 · CAFD [sec]
Equation 14: Soft-Start Time
TSS = 450,000 · CVCOMP [sec]
Equation 15: Minimum Error Amp Bandwidth
BWEA _ MIN =
0.000048
[Hz]
CICOMP
5.0 DESIGNING INVERTER MODULES
WITH THE LX1686
This section gives the design procedure for a typical inverter module
similar to the Microsemi standard LXM1612-12-01. It is designed to
operate from a single 10.8 to 13.2 volt power supply, and will drive a
single lamp with a run voltage in the range of 550 to 750VRMS. The inverter
includes an on-board regulator to provide VDD for the LX1686, a zero
power sleep mode circuit, and signal filtering on the BRITE input that
will accept either DC voltages or a PWM logic input to control lamp
current. Lamp current may be optionally controlled in either analog
(amplitude modulated) or digital (duty cycle modulated) fashion. The
schematic for this circuit is shown in Figure 10.
This design specifies an available Microsemi designed high voltage
transformer. Please see Appendix A for information about designing
custom transformers. Equations for calculating circuit component
values are summarized in Section 4.5. Additional module variations
follow this basic design.
Getting Started
First, select the bias current resistor for the LX1686. The value of R21
determines magnitude of four internal current sources that set timing
parameters. These are phase detector slew rate; digital dimming burst
oscillator frequency, triangle generator frequency, and main oscillator
frequency. R21 is normally set around 40K ohms, but may be in the
range of 20k to 60K. This design uses 43.2K 1%. While its value is
not critical, R21 temperature coefficient should be very low to prevent
parameter drift. Bias current is 1.00 volt divided by R21. Various
ratios of this current will be sourced from pins 5, 8, 9, and 21. Note
that this resistor affects all four functions stated, so it must be selected
first.
Next decide if you will be using the analog or digital dimming method,
and the polarity of the dimming signal. If you use digital dimming,
connect pin 12 (DIG_DIM) to VDD. It may alternately be left open
because it has an internal 100K pull-up to VDD. If you choose analog
dimming mode, connect pin 12 to ground. This design has an optional
jumper ( R20 ) in the design to permit easily changing methods.
Copyright © 2000
Rev. 1.5 9/01
The LX1686 includes an amplifier that can be used to invert the sense
of the BRITE pin. If pin 10 (BRITE_POS) is pulled up to VDD, lamp
current will increase for increasing voltage on BRITE. If grounded,
lamp current will increase for decreasing voltage on BRITE. This pin
has an internal 100K pull-up as well, but may be terminated directly to
VDD. This design also includes an optional jumper ( R19 ) to permit
easily reversing BRITE sense.
Now is a good time to consider filtering the IC power input pins. There
are two VDD inputs, pin 22 (VDD) feeds all analog signals, and pin 23
(VDD_P) feeds only the output drive stage. These need to be filtered
separately. We recommend 220nF ceramic capacitors on each pin
(C2 And C3) although 100nF is sufficient if the trace layouts are very
short and wide. VDD needs a 47 ohm resistor (R8) to help filter power
stage switching transients from the analog control circuits. This is
particularly important when using digital dimming, as any noise on
the supply can act as an unwanted input at the BRITE pin and cause
lamp current jitter. This will appear as lamp flicker.
C1 is added to reduce conducted emissions onto the power line. It
should be a ceramic type with good high frequency characteristics,
and must be located directly across the input connector power pins to
be effective.
Generating An On-Board VDD Supply For The LX1686
The LX1686 can operate with VDD and VDD_P from 3.0 to 6.0V. If a
supply in this range is not available, it can be generated on the inverter
module. For best efficiency VDD should be high enough to fully
enhance the FET (U2) at turn on. This design uses a 4.5 volt VGS FET
and sets VDD nominal at 5.3 volts to insure full enhancement under
worst case component tolerances. If you are using less than a five
volt supply, select a 2.7 volt VGS FET for U2.
The VDD linear regulator must supply 7mA maximum operating current
for the LX1686 plus another 5 to 10mA average for switching the FETs.
Design for 17 mA to insure that worst case conditions are met.
A transistor switch circuit that is driven with the ENABLE input precedes
the regulator. It removes VDD from the LX1686, eliminating its
quiescent current drain when ENABLE is low. Remaining battery load
is limited to leakage current of the three transistors and the filter
capacitors.
Q1 and Q2 form a non-inverting high side switch that draws no current
when the base of Q1 is less than 0.5 volts. R1 and R2, together with
Q1 VBE set input threshold to 2.0 volts. When Q1 turns on, Q2 saturates.
R5 provides positive feedback to Q1 generating hysterisis to prevent
partial turn on if input rise time is very slow. R6 and D1 establish a 6.2
volt reference that is buffered by emitter follower Q3. R7 is placed in
series with Q3 collector to reduce its power dissipation. C2 is an X7R
ceramic filter capacitor for VDD_P. R8 and C3, another X7R, filter
switching noise from the LX1686 VDD input. Both C2 and C3 must be
located so their connections to U1 use very short and wide traces.
VDD = VD1 - Q1VBE - R8 · IVDD
VDD ≅ 6.2 - 0.6 - 0.3
VDD ≅ 5.3V
13
LX1686 Direct Drive CCFL Inverter Design Reference
AN-13
Driving The ENABLE Input
In the example described directly above, ENABLE is connected to
VDD so the LX1686 will be turned on when its power is applied.
ENABLE may also be driven by a CMOS or TTL logic signal to cause
the LX1686 to enter sleep mode. A low on this line is sleep and a high
is active. If this later method is used, the switch circuit comprised of
Q1 and Q2 is not needed. Instead, connect the open end of R6 to
Vbat. Sleep current in this configuration will be approximately 1mA.
Logic Input Threshold Voltage
It is possible to have different voltages on VDD for the LX1686 and on
the logic chip that drives ENABLE or other logic inputs such as FVERT
and BRT_POS (as used in Figure 10). A common example is 5.3V on
the LX1686 and 3.3V on its driver IC. Since logic threshold voltage for
the LX1686 is VDD/2 + 0.6V, it will be necessary to provide a positive
bias to the LX1686 inputs. Adding a 6.8K series resistor at the input
terminal and a 10K pull-up to VDD at the LX1686 pin easily does this.
R29 and R30 illustrate this circuit on the FVERT pin. Voltage on the
FVERT pin will be 2.62V when the SYNC input is 0.8 volts and 3.93V
when SYNC is at 3.0V. These values center the 3.3V FVERT threshold
between worst case input levels from the driver.
Setting Main Oscillator Frequency
The main oscillator controls lamp current frequency. It is a linear ramp
generator that runs at twice the frequency of the output stage. Two
frequency limits must be programmed for this oscillator, FRAMP and
FRAMP_MAX. When the inverter is operating normally with the lamp ignited,
OLSNS will be greater than 300mV indicating lamp ignition.
This locks the oscillator at FRAMP. FRAMP is programmed by the values
of R21 and C5. When the inverter is in ‘strike mode’ (lamp is not ignited)
OLSNS will be less than 260mV, and oscillator frequency is placed
under control of the voltage ramp on the TRI_C pin. As TRI_C voltage
increases above 1.5 volts, frequency is slowly ramped to FRAMP_MAX.
The values of R13 and R14 determine FRAMP_MAX. The equations below
are written for a value N that is the ratio of maximum over minimum
frequency. Most inverter designs should be set up for N = 3 to insure
adequate strike voltage can be produced, e.g., open circuit resonance
of the high voltage transformer and parasitic load capacitance can be
reached during strike mode. N may be increased to as high as 5 for
designs using very high turn’s ratio transformers. These are generally
needed for striking 1400Vrms or higher lamps from a very low input
voltage such as a single cell Lithium-ion battery. It is a good idea to
measure the resonant frequency of your transformer wired to its actual
load before choosing ‘N’. Units in the following equations are volts,
ohms, farads, and Hertz.
FRAMP =
 VDD
0.72 1 +


20 
R21⋅ C5 + 0.5 ⋅ 10− 6
 VDD
0.72 1 +


20  0.5 ⋅ 10 −6
C5 =
−
R21 (FRAMP )
R21
14
FRAMP_MAX= N· FO ,
R13 =
R14 =
N = Maximum FRAMP multiplier
VDD ⋅ R21
4 (N − 2)
8
VDD ⋅ R21
VDD
(N − 1) − 4 (N − 2)
3
Using these equations to calculate values:
VDD = 5.3V (determined previously)
R21 = 43.2K (determined previously)
FRAMP = 130KHz, N = 3 (assumed)
C5 = 152.36pF (Use 150pF, 5%, COG to allow for
PCB trace capacitance and low T.C.)
R13 = 57.2K
(Use 57.6K, 1% or 56K, 5%)
R14 = 9.44K
(Use 9.53K, 1% or 10K, 5%)
Setting TRI_C Ramp Frequency
The signal at pin 21, TRI_C, controls the main oscillator frequency
during the lamp striking process. TRI_C is low, near zero volts, when
the lamp is operating. When in strike mode, TRI_C is a triangular
waveform that varies between .75 and 2.25 volts. The frequency of
the triangle wave is proportional to the value of C11, the capacitor on
pin 21. Frequency should be in the range of 2 to 20Hz. For
convenience, we use a 100nF X7R capacitor that gives an operating
frequency of:
FTRI =
1
30 ⋅ R21⋅ C11
FTRI =
1
= 7.7Hz
30 ⋅ 43.2K ⋅ 0.1 F
C11 =
1
30 ⋅ R21⋅ FTRI
Driving The BRITE Input
The BRITE input connects to an inverting op amp (Av = -1) and a
comparator. Depending on the state of BRT_POS, the comparator
will be connected to either the input or the output of the op amp. In
either case, the op amp output is connected to the BRITE pin through
its feedback resistors. The resistors are each 50K ohms (See Figure
3 Block Diagram). Since the op amp will force its inverting input to be
1.5 volts independent of BRITE’s voltage, the equivalent BRITE input
circuit becomes 50K ohms terminated to a 1.5V power supply. Input
current to the BRITE pin ranges from 0 to +/- 30µA, and flows into the
pin for signals lower than 1.5V. In order to minimize this loading effect
on dimming linearity and range, source impedance of the BRITE signal
should be much lower than 50K. While op amp gain accuracy is
excellent due to 1% matching of these two feedback resistors, their
absolute values can vary up to +/- 30% including lot to lot process
Copyright © 2000
Rev. 1.5 9/01
AN-13
LX1686 Direct Drive CCFL Inverter Design Reference
variations and temperature coefficient of resistance. This needs to be
considered in applications requiring extreme accuracy and repeatability
of brightness control, and usually requires the BRITE input be driven
with a low impedance op amp. In this design we use a simple 2K ohm
RC low pass filter that gives performance suitable for a notebook
computer display. This filter can convert a high frequency PWM signal
to a DC voltage compatible with the BRITE pin, or pass a DC voltage
directly from a potentiometer or DAC.
BRITE Pin DC Input Operating Voltage Range
The DC input operating voltage range for the BRITE pin is 0 to 3 volts
if in analog dimming mode (DIG_DIM is low) and 0.5 to 2.5 volts if in
digital dimming mode (DIG_DIM is high). The abbreviated digital
dimming range insures the burst comparator can produce both 0%
and 100% duty cycles by making the voltage ramp on its inverting
input have top and bottom limits that can be over ridden by the BRITE
input voltage of 0 to 3 volts. The top and bottom points of the ramp
are set at 2.5 and 0.5V respectively by an internal 2% reference voltage
to provide good accuracy and repeatability for the VBRITE vs. lamp current
transfer function. Dimming ratios of 100:1 are possible.
Pointers Using PWM Inputs For Brightness Control
Many microprocessors have PWM output channels available for
controlling external analog functions. These channels may be
programmed to output variable width pulses at various repetition rates.
Since the LX1686 needs a DC control voltage at its bright input, it is
necessary to change the PWM signal to DC with a low pass filter. In
Figure 10, the BRITE filter has a time constant of 2K x 100nF = 200µs.
This will not adequately filter a PWM input signal (If PWM frequency
is less than 75 KHz). If using a PWMinput, increase the value of C9
to make its time constant with R15 atleast 20 times the period of the
PWM pulses. For example, 100µFgenerates a 200ms time constant
that will adequately filter 100Hzand above. If filtering is inadequate,
ripple voltage on the BRITE pin will cause lamp current modulation at
the PWM repetition rate, and ifripple frequency is lower than 90Hz,
will appear as visible flicker.
Remember also, that DC voltage output from the filter is proportional
to input duty cycle and input voltage amplitude. If BRITE signal
amplitude is not constant, additional circuitry is needed to prevent its
variations from changing the lamp current. A universal conditioning
circuit is shown in Figure 13 that clamps any pulse amplitude above
2.5 volts while still passing a DC input without changing it. The circuit
has 500K input impedance so it is more tolerant of high source
impedance. Its large time constant integrator can filter PWM inputs
as slow as 50Hz, but it uses low frequency op amps which limit
maximum input frequency to 1 or 2KHz. High speed op amps can be
substituted if higher PWM frequencies are desired.
The optional resistor divider, R16 and R17 (Figure 10) can be used to
set a lower limit on BRITE voltage, even if the source voltage goes to
ground. This can guarantee a minimum brightness level, and, in the
case digital dimming is used, provide a hardware ‘stop’ that will keep
the BRITE input far enough above the VCO valley voltage to prevent
entering strike mode. This is discussed in more detail when we deal
with the OLSNS input circuit.
Copyright © 2000
Rev. 1.5 9/01
Synchronizing The Digital Dimming Burst Rate
The LX1686 has a complete PLL on chip to permit an external signal
to control burst rate. An AFD circuit (Auto Frequency Detect)
automatically looks for a logic signal input on the FVERT pin. If one is
detected, the PLL locks to it and generates an internal burst rate at
exactly twice its frequency. Doubling burst frequency insures an ability
to synchronize to low 50 and 60Hz frame rates while still having a
burst rate above visual detection. Actually, the PLL can lock to any
frequency between 40 and 200Hz. The polarity and duty cycle of
FVERT may be anything as long as a minimum pulse width of 300ns
is presented. Pulse polarity can even change on the fly without loosing
phase lock. This is important in portable applications because the
video sync signal can have different width and polarity in various
programs. A classic case is during boot up on a notebook where
DOS and Windows generate different VSYNC polarity.
AFD Circuit Response Time
AFD circuit response time is controlled by C4. An internal current
source alternately charges and discharges C4 with a square wave
that is generated by dividing the signal at FVERT by two. If FVERT is
present, the voltage at AFD_C will stabilize at VDD/2. If FVERT is
static high or low, the voltage at AFD_C will stabilize at either VDD of
VSS accordingly. Comparators sample this voltage to determine if a
signal is present. Response time is a function of the current source
amplitude set by R21 and the value of C4 as follows:
TD-AFD = 50 · R21 · C4
C4 =
TD − AFD
50 ⋅ R21
A 22nF X7R capacitor will result in about 0.48 second response, a
good place to start.
Maximum VCO Frequency
Digital dimming burst frequency is the same as VCO frequency.
Maximum VCO frequency (Fvco_max) must be designed for two times
the maximum expected frequency on the FVERT pin, e.g., Fvco_max
= 2 FVERTmax. If no signal is detected on FVERT, the VCO defaults
to ½ its maximum design value.
This feature can be used to set nominal burst frequency when no
synchronization input is used.
FVCO _ MAXI = 2 FVERTMAX =
1
5 ⋅ R21⋅ C8
For FVERT expected maximum equals 100Hz, FVCO_MAX = 200Hz.
Then;
C8 =
1
5 ⋅ R21⋅ FVCO _ MAX
C8 =
1
= 23nF
5 ⋅ 43.2K ⋅ 200
15
LX1686 Direct Drive CCFL Inverter Design Reference
AN-13
The closest smaller standard value is 22nF, which gives actual
maximum VCO frequency of 210Hz. If no input is placed on the FVERT
pin, typical burst frequency is 210 / 2 = 105Hz. This is about as low as
you should go to prevent visual detection of lamp current bursts. If
you want to calculate C8 to provide a specific ‘unsynchronized’ burst
frequency;
1
C8 =
5 ⋅ R21⋅ 2 ⋅ FVCO _ TYPICAL
1
= 10nF
5 ⋅ 43.2K ⋅ 2 ⋅ 200
Compensating The Phase Detector
C6, C7, and R18 values determine phase detector pull-in time (time
to lock to a new frequency) and damping factor. The phase detector
amplifier is a Gm type and has a maximum output current, IPD = 1 / 10
· R21. This current is in the range of 2µA, so low leakage ceramic
capacitors must be used for C6 and C7 to prevent erratic operation.
Pull-in time in the range of 0.1 to 1 second is usually acceptable. If
too long, however, the lamp may flash on power up as the VCO changes
from the initial default frequency to the lock frequency (remember, the
VCO will default to FVCO_MAX / 2 until PLL lock is achieved). The damping
factor is usually set to 0.707 (critically damped).
Pull-in time:
16 ⋅ π
, where WN is the natural PLL frequency.
WN
1
WN =
2 ⋅ R21⋅ 10 ⋅ C8 ⋅ C6
TP =
TP = 32 ⋅ π ⋅ R21⋅ 10 ⋅ C8 ⋅ C6
C6 =
TP 2
(32 ⋅ π ⋅ R21)2 ⋅ (10 ⋅ C8)
C6 = 220nF
For Pull-in time of 0.5 seconds,
C6 = 100nF
PLL Damping Factor
T 
DF =  Z  ⋅ WN , where T is the PLL zero time constant,
Z
 2
TZ = R18 · C6
10 ⋅ C6 ⋅ C8
C6
For Damping factor = 0.707, C6 = 220nF, C8 = 10nF, and R21 = 43.2K,
R18 = 82K
16
C7 =
C6
10
For C6 = 220nF, C7 = 22nF
If you are not using synchronized digital dimming do not install C4,
C6, C7, and R18. Instead, connect pins 4 and 8 to analog ground.
Drain to source voltage rating for U2 should be 3 to 4 times the
maximum power supply voltage. In this push – pull topology, operating
VDS is twice VBAT due to the dual primary winding on the high voltage
transformer. Because these windings are impossible to perfectly
match, especially with respect to leakage inductance, there will be an
inductive spike at transistor turn off. Its amplitude is proportional the
leakage inductance difference of the two winding halves. VGS needs
to be high enough to survive these spikes. Some design trade-offs
are possible. VGS can be increased to permit more imbalance in the
transformer construction, or snubbers can be added on the transistor
drains to suppress the transients. Snubbers generally dissipate power,
and in doing so reduce efficiency. Another possibility is to select a
FET having an adequate repetitive avalanche power rating to absorb
transient energy without overheating. This will also reduce efficiency.
On state resistance of the FET should be less than 200 milli ohms for
a typical 4 to 6 Watt load. It can be increased for lighter loads.
Experience shows that reducing Rds_on below 100 milli ohms
increases efficiency very little. For highest efficiency, it is better to
select a FET with very low total gate charge. At 60 to 80KHz, gate
losses can be larger than the I2R losses of a 100 milli-ohm FET pair.
VGS is an important selection parameter. The LX1686 drive voltage to
the FET gates is equal to the VDDP supply. Select FET’s that are fully
enhanced at the minimum supply voltage. This design uses an on
board generated VDDP that operates at 5.6V nominal to give the FET’s
a bit more drive.
AOUT And BOUT Connections
For Pull-in time of 0.7 seconds:
R18 = 4 ⋅ DF ⋅ R21 ⋅
This capacitor reduces high frequency gain of the loop.
Selecting U2, The Power FET
Assuming FVCO_TYPICAL is 230Hz;
C8 =
High Frequency Attenuation Capacitor C7
The 10K resistors (R10 and R12) to ground on these pins pull down
the FET gates when input supply voltage is too low to guarantee the
state of the LX1686 outputs. This typically occurs below 2.5V. If the
supply is increased very slowly through the 1 to 2.5 volt levels, it is
possible to inadvertently turn the FET’s on at 100% duty. This could
blow the input fuse or destroy the FET’s. If slow power on cannot
happen in your system, the 10K resistors may be omitted.
Thirty nine ohm resistors R9 and R11 are connected in series with
the gates and work with FET input capacitance to slow down their
transition time. This reduces high frequency emissions and helps
control EMI. Care must be taken to not make R9 and 11 too high, as
slowing transition time too much will cause the FET’s to overheat.
Power losses are highest during transitions since current and voltage
are both high then. If you choose a different FET, re-optimize R9 and
R11 values.
Copyright © 2000
Rev. 1.5 9/01
AN-13
LX1686 Direct Drive CCFL Inverter Design Reference
The High Current Power Stage
U2, T1 and C10 are the high current power stage components. Square
wave switching currents pass through each of them. They must be
designed for low resistance to minimize I2R losses, and must be
physically positioned close together with extremely short conductors
to prevent ground noise from interfering with the BRITE input, and to
minimize radiated and conducted emissions. Further PC layout should
have separate power and signal ground paths. Analog signal ground
should connect to power ground only at the negative end of C10.
C10 is a low ESR tantalum in this design. Always use D or E case
parts because they have the lowest ESR. Ceramic capacitors in the 2
to 10µF range also work well but are more expensive.
Configuring The High Voltage Output
High voltage is delivered to the lamp through C14, a series DC Bypass
capacitor. Use a 50VDCor higher rated XR7 type capacitor. Maximum
voltage appears across the capacitor when the output is shorted,
because lamp current increases under short circuit conditions. The
high voltage end of the lamp must be connected to this capacitor to
maintain high efficiency. Actually the lamp itself is non-polarized, but
when mounted in a panel, it takes on an important characteristic.
One terminal will have a very short wire resulting in very low parasitic
capacitance. The other’s will be long, creating much higher parasitic
capacitance. The concern is leakage currents flowing into this
parasitic load, since they serve no good function, reduce efficiency,
and complicate lamp current regulation. Since there is little voltage
to flow parasitic current, the long wire is returned to the low voltage
rectifier circuit.
The purpose of the ballast capacitor is to prevent any DC current in
the lamp, as this drastically reduces its life, and to act as real impedance
against the negative impedance of the lamp. Its value is chosen as
high as possible to minimize voltage drop across it. Voltage across
C14 is at a 90 degree phase shift with lamp voltage, so must be added
vector wise to find total voltage required from the transformer. Note
that with direct drive topology, it is not necessary to drop large voltages
on the ballast capacitor, as is the case with Royer circuits.
One precaution: direct drive, as with all other known topologies, can
experience a low frequency amplitude modulation of lamp current
under certain conditions. This is caused by the negative resistance
and extreme voltage dependency on temperature of the lamp operating
in a current regulated closed loop. Check for modulation, usually in
the 5 to 15KHz range while varying lamp current and power supply
input voltage. Modulation can usually be eliminated by optimizing
operating frequency.
High Voltage Feedback
Lamp voltage is divided across capacitors C16 and C17 and fed back
to the LX1686 VSNS input through emitter follower Q5. R24 DC
restores C17 to keep the waveform at Q5 base sinusoidal. Q5 acts as
Copyright © 2000
Rev. 1.5 9/01
a rectifier and buffer, presenting a known fraction of peak lamp voltage
to VSNS.
VSNS is the inverting input of the voltage error amplifier. Its
characteristics are identical to, and its output is processed the same
as the current error amplifier. Its non-inverting input is connected to a
precision 1.25 volt DC reference. If the peak lamp voltage exceeds a
preset limit by choosing the ratio of C16 and C17, AOUT and BOUT
pulse widths will be reduced as needed to regulate peak lamp voltage.
Choose C16 in the range of 2 to 5pF, keeping it small to minimize
leakage current and to keep unloaded resonant frequency below
300KHz so strike voltage is easily developed. Next choose C17 as a
multiple of C16 so maximum peak voltage is equal to 1.25 volts plus
Q5 VBE. If more freedom from Q5 temperature variation is desired,
add a 3.3K resistor in series with R23 and ground to form a voltage
divider. Connect the resistor tap to VSNS. Now VPK max will be 3.75V
plus Q5 VBE.
Microsemi has designed many inverters that use a PCB capacitor for
C16. If space allows, this will save the relatively high cost of a low
pico farad high voltage capacitor. Standard formulas for parallel plate
capacitors apply, and we have found computer grade PCB materials
to be a very acceptable dielectric. If you do this be sure to round all
sharp corners and leave at least 3mm spacing between low and high
voltage conductors to prevent arcing.
Short-Circuit Protection
Underwriters Laboratory requires maximum peak current, for even a
single cycle, be limited to a specific value when any of the output
terminals and ground are shorted with a 2K ohm resistor. This can be
a problem when shorting either lamp terminal to ground because the
current sensing mechanism in the control loop is shorted. Under this
situation, the loop will put out maximum possible current, which will
always exceed UL’s requirements. Q4 and R22 are added to this design
to meet UL requirements for a “limited current device”. R22 is selected
so the voltage across it will regulate maximum short circuit current
through Q4 and into the VSNS circuit, just like high voltage is regulated.
Since R22 is on the inside leg of the transformer, a short at the
connector will not disable its function. Set R22 empirically since total
current in it consists of both parasitic currents and actual lamp currents.
Parasitic currents should be measured on the actual LCD panel with
a production level inverter installed.
Open Lamp Sensing
The open lamp sense circuit is a peak detector that monitors lamp
current. When lamp current flows, the positive half cycles will charge
C18 through R25. During negative half cycles C18 slowly discharges
through R25 and R26. By keeping the ratio of R26 to R25 very high,
a highly filtered peak voltage is presented to the OLSNS input pin.
OLSNS is the input to a voltage comparator with positive threshold of
300mV and negative (low going) threshold of 260mV. When OLSN is
< 260mV the lamp is considered not ignited since little or no current is
17
AN-13
LX1686 Direct Drive CCFL Inverter Design Reference
flowing through it. Component values are set up so a very small number
of lamp current cycles (typically 6 or 8) will raise OLSNS above the
300mV level to indicate the lamp is ignited.
Setting proper component values gets tricky for digital dimming
operation if very low brightness levels are expected. A fine line exists
between the lamp being considered ‘on and dim’ rather than ‘off’. The
problem is that when the lamp is considered off, the LX1686 enters
strike mode where it actually operates at a 10Hz, 50% duty cycle
causing the lamp to flash at that rate! Preventing this undesirable
situation requires the designer to carefully consider the worst case
capability of the OLSNS detector compared to the minimum brightness
setting. In other words, if the detector can reliably detect the presence
of only 8 current cycles, don’t allow the BRITE input voltage to request
less than eight.
Reducing the value of R25 or C18 decreases delay time to detect
OLSNS going high and will lower the number of detectable current
cycles. Increasing the C18 / R26 discharge time constant permits the
inverter to operate at a slower dimming burst rate.
Lamp Current Regulation Loop
Lamp current is rectified by the diodes in D2 (pin 1) and D3 (pin 2).
D2 pin 2 is part of the OLSNS peak detector, and D3 pin 1 is a voltage
clamp that prevents the common mode input range of the current error
amplifier from being exceeded. The R/C network R28/C20 converts
lamp current to a filtered DC voltage that is passed to the error amplifier
at the ISNS pin. The filter time constant should be just large enough
to prevent feedback voltage peaks from exceeding error amplifier input
common mode voltage. R27 (Optional) on the outside of the rectifiers
shunts some of the lamp current from the error amplifier and is selected
to keep lamp light output constant over the input voltage range. C19
provides a low impedance ground path to parasitic RF currents during
open circuit or strike conditions.
To adjust rms lamp current amplitude, adjust the values of R27 and
R28. Increasing resistance decreases rms current. Typically the values
of R27 and R28 can be equal with the following exceptions:
1) If inverter input voltage is fixed within ±5%, R27 may be omitted;
2) If input voltage varies widely, then R27 / R28 can be varied to
achieve constant light output. Decreasing R28 while increasing
R27 increases lamp current at high line.
Compensating The Error Amplifiers
Both error amps are unconditionally stable without external capacitors
at their outputs. Compensation capacitors on these transconductance
amplifier outputs are selected specifically to control the respective
loop time constants.
The ideal current error amplifier compensation depends on whether
using digital or analog dimming. With analog dimming, loop response
can be quite slow, but for digital dimming it must be fast so lamp current
gets to full amplitude with in 2 or 3 current cycles. If response is
slower than this, light output at low brightness settings will not be
satisfactory, exhibiting uneven light along the length of the tube.
18
Uncompensated error amp response will allow overshoot on even the
first current cycle after the BRITE input goes high. Enough
compensation should be used to limit overshoot to the lamps maximum
rated rms current.
The only other consideration for selecting C12 is to aid in controlling
lamp current modulation as described in the section titled ‘Configuring
the high voltage output’.
Minimum Error amp bandwidth:
BWEA _ MIN =
48
C12
BW in Hz, C12 in µF
Voltage error amplifier compensation should also be fast so that open
circuit output voltage overshoot is controlled. The objects of this
compensation is to slowly increase output voltage in the case of a
cold turn-on and after each strike attempt while in strike mode. The
compensation capacitor C13 is discharged to zero after each power
down and between strike attempts.
Soft start time:
TSS = 0.45 · C13
TSS in seconds, C13 in µF
Lowest Cost Configuration
Figure 11 shows a similar inverter as described above (ref: Figure
10) designed for lowest cost. The transformer style has changed
from a 6W topolopy to a 4W with a corresponding decrease in output
voltage from nominally 650VRMS to 550VRMS . The difference is in
support circuitry. By using the system 5V supply and simplifying the
digital dimming interface, a total of 27 components can be removed from
the design of Figure 10.
The digital dimming feature is maintained, but is not synchronized by
the LX1686. It is possible to synchronize the PWM brightness control
signal on the system board, but many users do not require this be
done. By examining Figure 11 and Figure 3, the LX1686 block diagram,
it can be seen that the BRT_POS input is used to dynamically switch
the burst comparators non-inverting input between VDD and ground.
Since BRITE is connected to VDD, its compliment will be zero volts.
Both of these levels exceed VCO_C ramps peak and valley voltages,
so the comparators output follows its non-inverting input. As a result,
the non-inverting input to the current error amplifier (pin BRT) sees
the same 2.5V PWM signal that it would in the synchronized digital
dimming mode illustrated in Figure 10.
Because the LX1686 operates on 5V, there is no need for the onboard voltage regulator. Removing the zero power sleep switch
increases sleep current to 200µA plus current in R13, but this would
not be a problem if the 5V system supply were switched off in sleep
mode, a standard feature in note book computers.
This design assumes the ENABLE input and PWM input to BRT_POS
are generated from 5 volt logic so their ‘one’ logic levels are compatible
with the LX1686 input threshold voltages. If they are generated with
3.3 volt logic, it will be necessary to add the two resistor bias networks
described for the FVERT pin in Figure 10.
Copyright © 2000
Rev. 1.5 9/01
AN-13
LX1686 Direct Drive CCFL Inverter Design Reference
Pull down resistors on the FET gates are omitted because the +5V
supply will have a controlled rise and fall time. The 39 ohm EMI
reduction resistors are removed, but C1, the EMI reduction capacitor
is left in. The fuse has been omitted since an adequate fuse is
assumed on the +5V supply.
This same circuit configuration may be used with a 3.3 volt supply.
Replace U2 with VGS = 2.7V FET, and change the tantalum filter
capacitor to 220µF, 10V. You also need to make sure that lamp feed
back peak voltages to ISNS and VSNS do not exceed 3.3 volts since
this would prevent proper current and voltage regulation. Finally, the
transformer turns ratio needs to be adjusted for the reduced input
voltage.
Low Power Configuration
In Figure 12 find an example of a 1W single lamp inverter that is
designed to work from a single cell Li-ion battery. In this configuration a +5V logic input of 10mA continuous is required to power
the control IC. Since a 5V logic level supply is used the peak ISNS
and VSNS feedback voltages allowed is greater then described above
in a 3.3V only design.
This schematic is also an example of an analog only non burst
mode inverter. The BRITE input can either be a DC voltage
between 0 and 2.5V with 2.5V being full brightness, or a 2.5V
logic level PWM signal greater than or equal to 50KHz. This
particular design uses a lower profile FET and magnetics and is
optimized for 290V ±15% lamps running at 3mA output current.
The second amplifier has an integrating capacitor across the feedback
resistor, so it filters 50Hz PWM input pulses sufficiently to convert
them to a DC level. This is needed by the BRITE input. The voltage
on this amplifiers non-inverting input offsets the input signal in a positive
direction to prevent BRITE voltage from going below the minimum for
2% duty cycle on lamp current. A 0.73V offset is needed, so 1.37V is
placed on the non-inverting input. When the first amplifier output is at
2.0 volts, corresponding to BRITE INPUT = 0V, BRITE OUTPUT will
be at 0.74V. Similarly, a zero volt output from the first stage gives 2.74
volts at the BRITE OUTPUT which guarantees 100% duty cycle
(maximum brightness) can be achieved.
The low bandwidth LM324 limits maximum input repetition rate to about
1KHz. If higher input frequencies are used, replace it with a higher
performance part.
A Broken Lamp Shut Down Circuit
Some applications such as automotive displays, require the inverter
to shut off for personal safety reasons if the lamp fails to strike in a
reasonable time. The circuit of Figure 14 will accomplish this. It utilizes
the fact that signal TRI_C is always above 0.7 volts when the lamp is
in strike mode, and at zero volts when the lamp is ignited. This signal
enables the first comparator output to go high if the lamp is not ignited,
allowing the 330K resistor to charge the 10µF capacitor. After about 3
seconds, the 3.5V threshold of comparator two is reached, driving its
output low. This pulls down the VCOMP pin on the LX1686, forcing
output pulses at AOUT and BOUT to turn off. The capacitor is reset
by bringing ENABLE or the 5V supply low.
BRITE Input Signal Conditioning
Figure 13 shows a dual op-amp circuit that can be used as signal
conditioning for the brightness control signal. It will condition a PWM
logic input or a DC voltage as designed. Two functions are performed:
First, any voltage exceeding 2V will be ignored, and second, a zero to
2+ volt input signal will be shifted up to guarantee the inverter will be
at a well defined ‘minimum brightness’ with a zero volt input. This later
feature also insures the inverter will not enter strike mode as a result
of a low brightness input voltage. Minimum voltage at the LX1686
BRITE input (Pin 11) should be 0.73V. Maximum should be at least
2.58V.
Both amplifiers have voltage gain of minus one so there is no net gain
or inversion through the circuit. The first amplifiers’ purpose is to clamp
any input signal above 2 volts so amplitude variation does not change
lamp brightness. Refer to ‘Pointers on using PWM inputs for brightness
control’ above. Since the LM324 output goes very near ground, output
voltage will be essentially zero with two volts input. Further increases
in input voltage cannot drive the output more negative, therefor the
input is effectively clamped at plus two volts.
Setting the non-inverting input voltage of this amplifier determines the
maximum unclamped input voltage. If a wider dynamic input range is
wanted, increase voltage on the non-inverting input. Dynamic input
range is 2 times non-inverting input voltage.
Copyright © 2000
Rev. 1.5 9/01
19
VBAT
VBAT
GND
GND
SLEEP
BRITE
BRITE RTN
VSYNC
CN1
1
2
3
4
5
6
7
8
VBAT Power Input
Functional: 10.2V to 16V
Nominal: 12.0V ± 10%
SLEEP Input
(2.5V to 5.0V Logic Level)
Enable 'ON': Logic 'HI' >= 2.0V
Disable 'OFF': Logic 'LO' <= 0.80V
C1
100nF
25V
1A
F1 24V
SLEEP
R1 10K
C2
220nF
10V
20%
22nF
16V
150pF
5%
16VCOG
C7 22nF
16V
C8 10nF
16V
R19
Jumper
Option
R20
Jumper
Option
C5
C4
VDDP
47
Ohm
R18 82K
R14
10K
C3
220nF
20%
10V
R8
VDD
R13
56K
R17
Optional
VDD
C6
220nF
R16
10V
20%
Optional
R30
10K
R29 6.8K
R15
2.0K
SLEEP
BRITE Input
(Linear DCV Range)
Max. Bright: 'HI' >= 2.60V
Min. Bright: 'LO' >= 0.75V
(2.5V PWM Logic Level)
Max. Bright Positive Duty Cycle: 100%
Min. Bright Positive Duty Cycle: 30%
PWM Frequency: >= 75kHz
C9
100nF
16V
NOTES, UNLESS OTHERWISE SPECIFIED:
VSYNC Input
(2.5V to 5.0V Logic Level)
Logic 'HI' > 1.8V
Logic 'LO' < 0.8V
Min. Pulse Width: 300nsec
Frequency Range: 50 to 275Hz
Free Run Burst Rate: 310Hz ± 13%
1. RESISTOR VALUES EXPRESSED IN OHMS, 5% 1/16 WATT
2. CAPACITOR VALUES EXPRESSES AS CERAMIC, 10% X7R
R6 10K
1
Q2
MMBT2907A
2
3
R5
200K
1
2
3
R7
R3
4.3K
OLSNS
13
14
15
16
17
18
19
20
21
22
23
24
R4
27K
Q1
BC847A
BOUT
VDD_P
VDD
RAMP_C
ISNS
TRI_C
RAMP_R
VSNS
ICOMP
PD_CR
ENABLE
I_R
BRT
VCOMP
DIG_DIM
BRITE
BRT_POS
VCO_C
FVERT
AFD_C
VSS
VSS_P
AOUT
U1
R2
4.3K
1
2
3
4
5
6
7
8
9
10
11
12
LX1686CPW
1
VDDP
Q3
BC847
SOT-23
A
330 Ohm
1/8W
3
2
D1
MMSZ6V2
VDDP
VDD
R9
39 Ohm
R11 39 Ohm
C18 100nF
16V
C11 100nF
16V
OLSNS
C20 3.3nF 5% 16V
COG
C10
47uF 20%
25V
TANT
1
U2
Si9945 or NDS9945
8
C15 10nF
16V
7
5
R10
10K 2
4
C13 10nF
16V
R12
10K
6
43.2K
R21 1%
3
C12 2.2nF
16V
ISNS
VSNS
VDD
1
2
1
2
VSNS
Q4
BC847A
3
Q5
BC847A
3
R23 10K
R24 10K
5
T1
SGE2642-1
4
R22
82 Ohm
3
2
1
LMT3811
1:39
VDD
P
VDD
P
C14
100nF 20%
50V
1.50K
R28 1%
3
D2
BAV99
ISNS
1
2
R25 10K
2
3
OLSNS
R26
1.0 Meg
R27 N/U
VDD
1
Lamp LO
Lamp HI
D3
BAW56
CN2
2
1
Operational RMS lamp
condition @ 6.0mA,
65kHz:
650V ± 15%
C19 3.3nF 5% 25V
COG
C17
2.2nF 5% 25V
COG
C16
2.2pF
PCB
CAP.
C16 PCB CAP. copper areas to
equal approximately 32²mm for
0.80mm thick PCB
Copyright © 2000
Rev. 1.5 9/01
Figure 10: Standard LXM1612-12 Digital Single Lamp Inverter
20
LX1686 Direct Drive CCFL Inverter Design Reference
AN-13
VBAT
VBAT
GND
GND
SLEEP
BRITE
CN1
1
2
3
4
5
6
R2
10K
VDD
R3
47 Ohm
VDDP
C1
100nF 16V
SLEEP
R1
51K
NOTES, UNLESS OTHERWISE SPECIFIED:
C4
100nF 16V
1
2
3
12
11
10
9
8
7
6
4
C2
120pF 5% 16V
COG
5
C3
10nF 16V
VDD
1. RESISTOR VALUES EXPRESSED IN OHMS, 5% 1/16 WATT
2. CAPACITOR VALUES EXPRESSES AS CERAMIC, 10% X7R
AOUT
VDD_P
BOUT
U1
VSS_P
VDD
TRI_C
VSS
AFD_C
OLSNS
ENABLE
I_R
BRT
VCOMP
VSNS
ICOMP
ISNS
RAMP_C
RAMP_R
FVERT
PD_CR
VCO_C
BRT_POS
BRITE
DIG_DIM
LX1686CPW
24
23
22
21
20
19
18
17
VDDP
VDD
OLSNS
ISNS
VSNS
C14
C13
2.2nF 16V
3.3nF 5% 16V
COG
100nF 16V
100nF 16V
C6
10nF 16V
C5
C10
10nF 16V
1
C8
100uF 20% 10V
TANT
U2
Si9945 or NDS9945
8
7
C7
2
5
16
4
6
R4 44.2K 1%
3
SLEEP
15
14
13
1
2
1
2
Q1
BC847A
3
R6 10K
3
Q2
BC847A
VSNS
6
7
R7 10K
T1
SGE2687-1
R5
82 Ohm
1
2
3
4
5
LMT2110
1:80
VDDP
R10
1.37K 1%
R11
3
D1
BAV99
ISNS
1
2
R8 10K
R9
1.0 Meg
C15
2
3
OLSNS
N/U
VDDP
1
Lamp LO
Lamp HI
D2
BAW56
CN2
2
1
C11 PCB CAP. copper areas to
equal approximately 32²mm for
0.80mm thick PCB
550V ± 15%
Operational RMS lamp
condition @ 6.0mA, 80kHz:
2.2nF 5% 25V
COG
C12
2.2nF 5% 25V
COG
C11
2.2pF
PCB CAP.
C9
100nF 20% 50V
VBAT Power Input
Functional: 5.0V ± 10%
Nominal: 5.0V ± 5%
SLEEP Input
(3.3V to 5.0V Logic Level)
Enable 'ON': Logic 'HI' >= 3.2V
Disable 'OFF': Logic 'LO' <= 0.80V
BRITE Input
(3.3V to 5.0V PWM Logic Level)
Max. Bright Positive Duty Cycle: 100%
Min. Bright Positive Duty Cycle: 5%
PWM Frequency: 100Hz to 500Hz
21
Copyright © 2000
Rev. 1.5 9/01
LX1686 Direct Drive CCFL Inverter Design Reference
AN-13
Figure 11: Low Cost Direct PWM Input, Single Lamp Inverter
VBAT
GND
+5V
SLEEP
BRITE
CN1
1
2
3
4
5
VBAT Power Input
Functional: 3.0V to 5.5V
Nominal: 3.7V ± 13.5%
C1
100nF 16V
SLEEP
+5V Logic Level Input
Functional: 5.0V ± 10%
Nominal: 5.0V ± 4%
I Logic: 10mA Continuous
SLEEP Input
(3.3V to 5.0V Logic Level)
Enable 'ON': Logic 'HI' >= 3.1V
Disable 'OFF': Logic 'LO' <= 0.80V
BRITE Input
(Linear DCV Range)
Max. Bright: 'HI' = 2.5V
Min. Bright: 'LO' >= 0.0V
(2.5V PWM Logic Level)
Max. Bright Positive Duty Cycle: 100%
Min. Bright Positive Duty Cycle: 0%
PWM Frequency: >= 50kHz
NOTES, UNLESS OTHERWISE SPECIFIED:
VDDP
C16
2.2uF 10V
R2
10K
VDD
C3
10nF 16V
1
2
3
12
11
10
9
8
7
6
4
C2
100pF 5% 16V
COG
5
C4
100nF 16V
VDD
R3 47 Ohm
R1
51K
R14
4.7K
C17
100nF 16V
1. RESISTOR VALUES EXPRESSED IN OHMS, 5% 1/16 WATT
2. CAPACITOR VALUES EXPRESSES AS CERAMIC, 10% X7R
AOUT
VDD_P
BOUT
U1
VSS_P
VDD
TRI_C
VSS
AFD_C
OLSNS
ENABLE
I_R
BRT
VCOMP
VSNS
ICOMP
ISNS
RAMP_C
RAMP_R
FVERT
PD_CR
VCO_C
BRT_POS
BRITE
DIG_DIM
LX1686CPW
24
23
22
21
20
19
18
17
16
15
14
13
VDDP
VDD
OLSNS
ISNS
VSNS
SLEEP
C13
2.2nF 5% 16V
COG
100nF 16V
100nF 16V
C14
10nF 16V
C5
C6
10nF 16V
10nF 16V
C10
C7
R4 46.4K 1%
C8
100uF 20% 10V
TANT
4
5
U2
Si3948DV or FDC6561AN
1
6
R12
10K
2
R13
10K
3
1
2
1
2
Q1
BC847A
3
Q2
BC847A
3
R6 10K
VSNS
6
7
R7 10K
T1
SGE2694-1
RTN
R5
82 Ohm
1
2
3
4
5
RTN
LMT1310
1:56
VDDP
R10
4.32K 1%
R11
3
D1
BAV99
ISNS
1
2
R8 10K
R9
1.0 Meg
C15
2
3
OLSNS
N/U
VDDP
1
Lamp LO
Lamp HI
D2
BAW56
CN2
2
1
C11 PCB CAP. copper areas to
equal approximately 40²mm for
0.80mm thick PCB
290V ± 15%
Operational RMS lamp
condition @ 3.0mA, 90kHz:
2.2nF 5% 25V
COG
C12
2.2nF 5% 25V
COG
C11
2.7pF
PCB CAP.
C9
100nF 20% 50V
Copyright © 2000
Rev. 1.5 9/01
Figure 12: Low Cost Analog Single Lamp Inverter
22
LX1686 Direct Drive CCFL Inverter Design Reference
AN-13
LX1686 Direct Drive CCFL Inverter Design Reference
AN-13
2.2uF *
Bipolar Capacitor
* for 50Hz PWM input. May be
lower for higher frequencies.
499K 1%
499K 1%
499K 1%
499K 1%
BRITE INPUT
BRITE OUTPUT
9
1.00V
10
6
8
C
5
1.37V
LM324
B
7
LM324
Figure 13: Brite Input Conditioning Circuit
Clamps and filters PWM BRITE inputs making lamp brightness insensitive to input pulse amplitude,
and shifts 0 to 2V DC inputs to match LX1686 levels.
8
+5V
330K
0.6V
3
TRI-C
2
A
1
LM393
3.5V
4
1
3
ENABLE
LM393
5
6
B
7
VCOMP
2
+
BAW56
10uF
16V 20%
TANT
Figure 14: Open Circuit Time Out
Shuts off inverter after attempting to strike the lamp for more than 3 seconds.
6.0 TRANSFORMER DESIGN and
SELECTION CRITERIA
Bifilar-Wound Primary
Open Circuit Series Resonance
The transformer must have a bifilar-wound primary so that the leakage
inductance of each half of the primary is approximately equal. Leakage
inductance of the two halves must be within 5%. The purpose is to
balance the energy released each time the FET switch is opened. If
the leakages are equal, the smallest possible voltage spike is imposed
upon the FET switch. When unequal, the voltage spikes are higher
causing high-voltage stress from drain to source of the FETs.
During lamp strike mode operation of the LX1686 controller, the
switching frequency is swept from its “base” operating frequency to a
maximum of about three times this base frequency. For example, if
the base operating frequency is set to 60kHz, during lamp strike mode
the switching frequency will sweep from 60 to 180kHz. The transformer
must exhibit additional “gain” at the higher frequencies so it will supply
a high enough output voltage to strike the lamp. It is best to program
the controller’s swept range to match the transformer’s characteristic
when installed on the inverter. This enables the strike voltage to be
available for the maximum duration.
Leakage Inductance
The leakage inductance must also be sufficient to limit the inrushpeak current that the FET must withstand. Loosely coupled designs
offer the best option for this required characteristic. Designs which
have closely coupled primary and secondary typically have very low
leakage inductance. Typically, the leakage should be sufficient to limit
the slew rate of the switching current such that the maximum required
primary current is reached in 1/4 of the total switching period. Most
sector-wound high voltage transformers in this power class have
sufficient leakage inductance to limit the peak current to acceptable
levels. Excessive leakage will limit the output current ability of the
transformer, but is seldom encountered in practice.
Copyright © 2000
Rev. 1.5 9/01
Open circuit (lamp not ignited) resonance of the transformer must be
within the frequency range of the sweep. The series resonant circuit
is composed of the leakage inductance, the secondary winding
capacitance, and load capacitance. Since the reactances cancel at
resonant frequency, maximum current is obtained. This maximum
current produces the maximum voltage across the load.
Load capacitance includes a capacitive voltage divider used in the
over-voltage protection scheme that typically adds 2-5pF. The lamp
high-lead capacitance-to-ground also adds 1-3pF. Ensure that these
capacitances are taken into account when evaluating a design.
23
AN-13
LX1686 Direct Drive CCFL Inverter Design Reference
Sufficient “Q” at Resonance
The strike voltage developed at resonance must be sufficient to ensure
lamp ignition. Typically, strike voltage is about twice (2X) the run voltage
at low temperature. It generally increases as the CCFL ages. “Q” of
the resonant circuit must be high enough that the required
“magnification factor” is achieved. The higher the L/C ratio, the higher
the “Q”. Thus, minimizing the lumped secondary capacitance
maximizes Q and strike voltage. All standard information regarding
resonant circuit behavior and simplified transformer equivalent circuits
is directly applicable to this topology. Most sector-wound high voltage
transformers we have evaluated have series resonances in the range
of 90 - 200kHz when installed on the inverter. Please refer to
transformer design discussion and equations.
Transformer Turns Ratio
The copper trace resistance is small enough to be neglected. The
“RDS ON” of the FET can be estimated from its datasheet; it is a function
of the gate drive voltage. The resistance of the primary winding will
not be known at this point but a rough estimate will suffice. It must be
remembered that the primary is a center-tapped configuration and
therefore each switching cycle “sees” 1/2 the total primary resistance.
for RDS ON = 0.075 Ohms & RPRI EST = 0.500 Ohms:
RPRI HALF = 0.5 Ohms · RPRI EST = 0.25 Ohms
RTOTAL = RDS ON + RPRI HALF = 0.325 Ohms
VFET XF DROP = IIN · RTOTAL = 0.122V
VPRIMARY = VMIN - VWIRE DROP - VFET XF DROP = 14.11V
1. The minimum input voltage available to the primary winding and,
This value of VPRIMARY is the peak voltage of the PWM square wave
that will be applied alternately to each half of the primary. It is not an
exact known value, but rather an estimate with which to predict the
required turns ratio of the first pass design. Now the maximum required
voltage of the secondary must be estimated.
2. The maximum output voltage required from the secondary winding.
Estimating Maximum Secondary Running Voltage*
Estimating Minimum Primary Voltage
The output section is composed of a ballast capacitor, a voltage divider
capacitance (used for open circuit protection), the parasitic lamp highlead capacitance to ground, the lamp itself and the parasitic
capacitance of the ignited lamp to ground.
Calculation of the required turns ratio involves the estimation of two
critical factors:
The minimum input voltage is obtained from the value of the lowest
supply voltage, the maximum resistive voltage drop of the wiring from
the supply to the inverter and the maximum voltage loss on the inverter
from the input connector to the primary. The primary winding resistance
itself must also be considered.
Example:
for VIN 15V with Tolerance ±5%:
VMIN = VIN · 0.95 = 14.25V
Wiring Voltage Drop:
for #22 AWG wire, resistance at 20°C is 0.0162 Ohms per foot. Two
wires (+VIN & Power Ground) of 18 inch Length are used for a total
length of 36 inches or 3 feet. At 20°C
RWIRE = Length · Resistance = 0.049 Ohms
In order to estimate the current flowing through the wiring, you must
know the required output power of the inverter and its efficiency.
Assume a worst-case efficiency of 75% & an output power of 4 Watts.
 1 
PIN =   ⋅ POUT = 5.333 W
 Eff 
IIN =
PIN
= 0.374 A
VIN
VWIREDROP = IIN · RWIRE = 0.018V
The ballast capacitance used in this topology is generally 220pF. The
voltage divider capacitance is generally 4pF. Lamp wiring capacitance
to ground of the high lead is typically 2-3pF for high quality voltage
wire of short length (measure it if in question). The ignited lamp has a
capacitance to ground that may be determined empirically by
measuring the panel ground return current and the lamp’s operating
voltage simultaneously. Knowing the operating frequency of the drive
voltage you can calculate the capacitive resistance value of this
parasitic and then “back-calculate” for the capacitance. Most quality
14 inch (diagonal) displays we have seen are close to 20pF. Make the
measurement if the data is not available from the panel supplier.
CBAL = 220 · 10-12 CDIV = 4 · 10-12 CWIRING = 3 · 10-12 CLAMP = 20 · 10-12
(all values in Farads)
The operating frequency of the inverter must now be known to complete
the estimate. Most CCFL inverters are designed to operate in the
range of 50 to 100kHz. For this example, 65kHz will be used. The
ballast capacitor is connected to the output high terminal (series) with
the remaining capacitors appearing in parallel with the lamp. Thus,
the voltage developed across the ballast capacitor is a function of the
lamp and the capacitor (divider, wiring and lamp) currents at the
operating frequency. The lamp voltage appears across these three
capacitances; this enables us to easily calculate the magnitude of this
“parasitic” current.
CEQ = CDIV + CWIRING + CLAMP = 27pF
Next, we must estimate the voltage loss on the inverter itself. The
losses here are the resistance of the copper traces, the “RDS ON” of the
FET switch and the resistance of the primary winding of the transformer.
24
X CEQ =
with f = 65KHz
1
= 90kΩ
(2 ⋅ π ⋅ f ⋅ C EQ )
Copyright © 2000
Rev. 1.5 9/01
LX1686 Direct Drive CCFL Inverter Design Reference
AN-13
Now the lamp operating voltage must be known. Lamp voltage
increases over time. For example, a lamp listed as 550 V typical at
7.5 mA might have an initial production tolerance of ±10% which would
translate to an initial maximum of 605V .
At “end-of-life” for a given number of hours, this might be as high as
760 V (example only!). For our example, we will use a lamp voltage of
850 V. The current that will flow through the lumped capacitance that
is across the lamp would be:
ICEQ =
VLAMPMAX
= 0.009 A
X CEQ
If the lamp is to be driven at 7.5 mA and is considered to be mainly
resistive in nature, we may now add (in quadrature due to the 90°
phase relationship) the capacitive current for this to arrive at the total
magnitude of the current flowing through the ballast capacitor which
must be supplied by the transformer secondary.
ITOTAL = ILAMP 2 + ICEQ 2 = 0.012 A
Since we know the operating frequency, the ballast capacitance and
the magnitude of the current through this capacitance, we can now
determine the magnitude of the voltage across the ballast.
X CBAL =
1
= 1.113 ⋅ 10 4
(2 ⋅ π ⋅ f ⋅ C BAL )
VCBAL = ITOTAL ⋅ X CBAL = 133 V
The last item to be addressed is the voltage drop of the secondary
resistance when delivering the required current. Most CCFL inverter
output transformers in the 3 to 8 Watt power range have secondary
winding resistances in the range of 300 to 800 Ohms. We will assume
500 Ohms here.
VSEC DROP = RSEC · ITOTAL = 6 V
Added to the lamp voltage (linearly as the two are considered “in
phase”) gives total resistive component which can be added in
quadrature (assumed 90° out of phase with capacitive voltage) to the
ballast capacitor’s voltage to give the total required output voltage.
Knowing this, we can estimate the required turns ratio of the
transformer.
2
VTOTAL = (VLAMPMAX + VSECDROP ) 2 + VCBAL
VPRIMARY = 14.11
VSECONDARY = VTOTAL = 866 V
At this point in the design process, you should have some idea of the
core geometry and size required for your application. From the core
constants provided by the core manufacturer, you will need the “Ae”
or effective core cross sectional area. Knowing the voltage to be supplied by the secondary and assuming it will be closer to a sine wave
than a triangular wave, we can estimate the peak flux density required
to support this voltage. We will use a core with an effective crosssectional area of 0.092 square centimeters for this example.
For most applications involving a relatively aggressive (i.e. small) size,
the flux density is in the neighborhood of 2000 Gauss (BPK= 2000).
Utilizing the common expression for peak flux density generated by a
sinusoidal voltage impressed upon a coil of “N” turns around a core of
“Ae” cross-section, the required number of turns to meet this flux density “limit” would be:
N SEC =
V SECONDARY ⋅ 10 8
= 1631
B PK ⋅ 4.44 ⋅ ƒ ⋅ Ae
Required Number of Primary Turns
Now that the turns ratio and number of secondary turns have been
estimated, simply divide the number of secondary turns by the turns
ratio to arrive at the number of primary turns. Since the primary is to
be center-tapped for the push-pull topology, we will need to round
down to the nearest even integer number of turns. If you choose, you
can round up, since there may be enough margin in the estimates
performed thus far to support the subsequent required maximum secondary voltage. Another choice is to alter the secondary turns count
based upon the rounded number of primary turns.
NSEC = 1631
NPRI =
NPRI SEC = 30.68
NSEC
= 53.2
NPRISEC
Depending on your choice of rounding, the primary could be 52 or 54
turns using the data supplied for this example. If you chose 5 then
you could subsequently recalculate the secondary based upon this
primary turns count:
NSEC 52TPRI = 52 · NPRI SEC = 1.591 · 103
and for 54 turns:
NSEC 54TPRI = 54 · NPRI SEC = 1.652 · 103
Turns Ratio
Turns Ratio: N =
Required Number of Secondary Turns
VSECONDARY
= 61.37
VPRIMARY
The primary voltage used in these calculations was applied to 1/2 of
the primary. Therefore, the ratio of TOTAL primary turns to TOTAL
secondary turns is 1/2 the “N” above. The primary to secondary turns
ratio is therefore:
Determination of Primary Wire Gauge
Based upon your particular bobbin (coil-former) selection, you will have
a maximum winding width and height for the bobbin used. It is
important to remember that the winding must be such that an even
number of layers must be wound to bring the center-tap back to the
NPRI SEC = 0.5N = 30.68
Copyright © 2000
Rev. 1.5 9/01
25
LX1686 Direct Drive CCFL Inverter Design Reference
AN-13
primary terminations (pins). All bobbins we have worked with for CCFL
transformer construction have the primary terminations on one end of
the bobbin. This means that the total number of layers must be in
multiples of 2 to meet the termination requirements.
This restriction limits the number of choices in terms of wire gauge
selection and primary winding volume utilization. For instance: If the
primary sector height is 0.040 inches and the design requires 4 layer
construction, then the maximum layer height would be 0.010 inches.
Referring to a standard wire chart of sizes and choosing “single”
insulation build magnet-wire, #31 AWG (max. diameter of 0.010 inches)
would just fit. In reality, you would probably want to go up one gauge
(down one size) to #32 AWG (max. dia. of 0.0091) to account for the
bowing out of the wire as it is wound around the bobbin.
Working with the available width of the primary sector, you must determine if the required number of turns will fit distributed among 4
layers (or however many layers are required). If the sector width is
0.125 inches, for example, then the maximum number of turns per
layer is the width divided by the wire diameter. In practice, it will be
somewhat less as some space will inevitably be wasted due to winding technique or slight bends in the wire itself. This is generally referred to as a “space factor” and might reduce the available width use
by 10% (i.e. a 90% space factor). Also, on the first and third layer, the
entry of the wires into the bobbin will reduce the theoretical turns count
by 1 turn on each of those layers. This is accounted for in the calculation of primary turns based on size, below.
Example:
HSECTOR = 0.040
HLAYER =
Layers = 4
HSECTOR
= 0.01
Layers
WSECTOR = 0.125
Turns per layer:
Apply your desired “space factor” (SF): SF = 0.90
TurnsCORRECTED = Turns PRI · SF = 53.75
The choice of primary turns is now best made in favor of the 52 turn
center-tapped winding wound in four layers of AWG#33 single insulation build. As calculated earlier, this puts the required number of secondary turns at 1591. This will result in a slightly higher flux density
than the example outlined. This should be of little consequence, but it
is up to the designer to evaluate all the trade-offs made in the design
process.
The final step in this procedure is to determine the secondary winding
constraints and apply these limits to the selection of wire gauge. The
differences in method are subtle. The secondary will most likely be
“random” wound as, even with traversing wire-guides, the secondary
windings will “wander” a bit. The copper loss of the secondary is usually small so the selection of wire gauge depends on the limits of the
transformer manufacturer’s capabilities.
Number of secondary sectors: SectorsSEC
Width of each sector: WSEC SECT = 0.08
WTOTAL = WSEC SECT · SectorsSEC = 0.4
Choose a wire gauge; this can be done by an acceptable current density or by choosing what will fit in the space allotted. You will still have
to evaluate the implications of your selection electronically and mechanically. Current density limits vary so widely that we suggest you
use what you are comfortable with based upon your own design experience. Consider, though, that the heat path from the inside of the
winding to the outside is generally short in CCFL inverter transformers. This enables operation at higher current densities.
Assumed Current Density Limitation: 500 circular mils per Ampere
Choose wire size: DiaAWG32 = 0.0091
W
TPL = SECTOR = 13.736
Dia AWG32
TurnsPRI = ((Layers - 2) · TPL) + (2 · (TPL - 1)) = 52.945
CDLIM = 500
ITOTAL = 0.012
Pick a wire-gauge that would fit your limit (i.e. AWG#43, 5.84 circular
mils or CMAWG43 = 5.84)
CD ACTUAL =
CM AWG43
= 486.6 cm / A
ITOTAL
Apply your desired “space factor” (SF): SF = 0.90
TurnsCORRECTED = Turns PRI · SF = 47.651
It is obvious that the desired 52 or 54 turn winding would not fit. Thus,
the recommendation would be to go up a wire gauge (drop a wire
size). AWG#33 single insulation build has a maximum diameter of
0.081 inches. Let’s check if it fits:
DiaAWG33 = 0.0081
TPL =
WSECTOR
= 15.432
Dia AWG33
TurnsPRI = ((Layers - 2) · TPL) + (2 · (TPL - 1)) = 59.728
26
This is a higher current density than the limit set because there is less
cross-sectional area of copper available for each Ampere of current
flowing. Now determine if this gauge will fit in the space allotted:
DiaAWG43T = 0.0031
Layers =
TPL =
WTOTAL
= 129.032
Dia AWG43T
NSEC52TPRI
= 12.326
TPL
Build = Layers · DiaAWG43T = 0.038
Copyright © 2000
Rev. 1.5 9/01
LX1686 Direct Drive CCFL Inverter Design Reference
AN-13
Assuming the secondary sectors are of the same height as the primary (0.040 inches), the build just calculated (0.038) will probably not
fit when the “space factor” is accounted for. The recommendation
would be to go up a wire gauge (down in size) to AWG#44.
DiaAWG43T = 0.0029
Layers =
TPL =
WTOTAL
= 137.931
Dia AWG44T
NSEC52TPRI
= 11.531
TPL
Build = Layers · DiaAWG43T = 0.036
Build 90%SF = 1.1 · Build = 0.039
It appears this will fit.
Conclusion of Winding Analysis
The intent of all of the above was to arrive at a “first-pass” design for a
given amount of winding space. The turns ratio and total number of
secondary turns are such that a maximum flux density will not be
exceeded when the transformer is required to supply the maximum
output voltage set by the operating conditions of the inverter. The
assumptions made are to be varied by the designer as are the
acceptable limits of flux density, current density and space factor of
the windings. Insulation build of the wire sizes chosen are a matter of
acceptable voltage stresses and winding capacitance.
Copyright © 2000
Rev. 1.5 9/01
27