LT3692 - Monolithic Dual Tracking 3.5A Step-Down Switching Regulator

LT3692
Monolithic Dual Tracking
3.5A Step-Down
Switching Regulator
Description
Features
Wide Input Range:
– Operation from 3V to 36V
– OVLO Protects Circuit Through 60V Transients
n Independent Supply, Shutdown, Soft-Start,
Programmable Current Limit and Programmable
Power Good for Each 3.5A Regulator
n Die Temperature Monitor
n Adjustable/Synchronizable Fixed Frequency
Operation from 250kHz to 2.25MHz with
Synchronized Clock Output
n Independent Synchronized Switching Frequencies
Optimize Component Size
n Antiphase Switching
n Outputs Can Be Paralleled
n Flexible Output Voltage Tracking
n Enhanced Short-Circuit Protection
n Low Dropout: 95% Maximum Duty Cycle
n5mm × 5mm QFN Package
The LT®3692 is a dual current mode PWM step-down
DC/DC converter with two internal 3.8A switches. Independent input voltage, shutdown, feedback, soft-start, current
limit and comparator pins for each channel simplify complex power supply tracking and sequencing requirements.
n
To optimize efficiency and component size, both converters have a programmable maximum current limit and are
synchronized to either a common external clock input,
or a resistor settable fixed 250kHz to 2.25MHz internal
oscillator. A frequency divider is provided for channel 1
to further optimize component size. At all frequencies, a
180° phase relationship between channels is maintained,
reducing voltage ripple and component size. A clock output
is available for synchronizing multiple regulators.
Minimum input to output voltage ratios are improved by
allowing the switch to stay on through multiple clock cycles
only switching off when the boost capacitor needs recharging. Independent channel operation can be programmed
using the SHDN pin. Disabling both converters reduces
the total quiescent to <10µA.
Applications
Automotive Supplies
Distributed Supply Regulation
n
L, LT, LTC, LTM, Burst Mode, Linear Technology and the Linear logo are registered trademarks
of Linear Technology Corporation. All other trademarks are the property of their respective
owners.
n
Typical Application
Independent Synchronized Switching Frequencies
Extend Full Frequency Input Range
Dual 1.8V/3.5A and 5V/3.5A Step-Down Converter
VIN
6V TO 32V
4.7µF
×2
VOUT1
1.8V 3.5A
500kHz/
250kHz
143k
7.15k
6.8µH
0.47µF
SW1
VIN2
VIN1
SHDN1
BST1
VOUT2
CLKOUT
SHDN2
BST2
SW1
SW2
IND1
IND2
SW2
6.8µH
1µF
LT3692
100µF
×2 8.06k
10k
FB1
SS2
60.4k
ILIM2
CMPI2
CMPO1
CMPO2
SS1
680pF
47pF
17.8k
13.0k
42.2k
100k
8.06k
PG2
VIN = 12V
1µs/DIV
3692 TA01b
VIN = 24V
1µs/DIV
3692 TA01c
CLKOUT
SS2
ILIM1
VC1
60.4k
FB2
CMPI1
DIV
VOUT2
5V 3.5A
47µF 500kHz
VOUT2
VOUT1
ILIM2
CLKOUT
VC2
RT/SYNC
GND
TJ
0.1µF
CLOCKOUT
500kHz
330pF
36.5k
SW1
60.4k
0.1µF
33pF
3692 TA01a
For more information www.linear.com/3692
SW2
3692fa
1
LT3692
SS1
VIN1
SW1
DNC*
IND1
DNC*
TOP VIEW
VOUT1
32 31 30 29 28 27 26 25
BST1 1
24 ILIM1
CMPO1 2
23 VC1
22 RT/SYNC
CMPI1 3
FB1 4
21 CLKOUT
33
GND
FB2 5
20 TJ
CMPI2 6
19 DIV
CMPO2 7
18 VC2
17 ILIM2
BST2 8
SS2
SHDN2
VIN2
SW2
DNC*
9 10 11 12 13 14 15 16
IND2
VIN1/2, SHDN1/2, CMPO1/2............................. 40V/–0.3V
VIN1/2 Transient (Note 2)................................. 60V/–0.3V
SW1/2.....................................................................VIN1/2
BST1/2............................................................ 60V/–0.3V
BST1/2 Pin Above SW1/2...........................................25V
IND1/2, VOUT1/2..........................................40V/–0.3V, 7A
FB1/2, CMPI1/2, SS1/2.................................................5V
RT/SYNC......................................................................5V
DIV, ILIM1/2..............................................................2.5V
VC1/2, TJ............................................................... ±100µA
Operating Junction Temperature Range (Note 3)
LT3692EUH......................................... –40°C to 125°C
LT3692IUH.......................................... –40°C to 125°C
Storage Temperature Range................... –65°C to 150°C
SHDN1
Pin Configuration
DNC*
(Note 1)
VOUT2
Absolute Maximum Ratings
UH PACKAGE
32-LEAD (5mm × 5mm) PLASTIC QFN
θJA = 35°C/W
EXPOSED PAD (PIN 33) IS GND, MUST BE SOLDERED TO PCB
*DO NOT CONNECT
Order Information
LEAD FREE FINISH
TAPE AND REEL
PART MARKING*
PACKAGE DESCRIPTION
TEMPERATURE RANGE
LT3692EUH#PBF
LT3692EUH#TRPBF
3692
–40°C to 125°C
32-Lead (5mm × 5mm) Plastic QFN
LT3692IUH#PBF
LT3692IUH#TRPBF
3692
–40°C to 125°C
32-Lead (5mm × 5mm) Plastic QFN
Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container.
Consult LTC Marketing for information on non-standard lead based finish parts.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
The
l denotes the specifications which apply over the full operating
Electrical
Characteristics
temperature range, otherwise specifications are at TA = 25°C. VVIN1/2 = 15V unless otherwise specified. (Note 3)
PARAMETER
CONDITIONS
SHDN Voltage Threshold Ch 1/2 (Note 4)
l
SHDN Input Current Ch 1/2
VSHDN = 1.3V
VIN1 Undervoltage Lockout (Note 5)
VFB1/2 = 0V, VVOUT1/2 = 0V, VIND1/2 = 0V
VIN Overvoltage Lockout Ch 1/2 (Note 6)
MIN
TYP
MAX
1.24
1.32
1.4
UNITS
V
1.5
5
µA
2.5
2.8
3.1
V
36
38
41
V
VIN1 Shutdown Current
VSHDN = 0V
l
6
10
µA
VIN2 Shutdown Current
VSHDN = 0V
l
0
2
µA
VIN1 Quiescent Current
VFB1/2 = 0.9V
3
4
5
mA
VIN2 Quiescent Current
VFB1/2 = 0.9V
400
630
1000
µA
Feedback Voltage Ch 1/2
VVC1/2 = 1V
790
806
822
mV
2
l
3692fa
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LT3692
Electrical
Characteristics
The
l denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VVIN1/2 = 15V unless otherwise specified. (Note 3)
PARAMETER
CONDITIONS
Feedback Voltage Regulation
VVIN1/2 = 3 to 40V, VVC1/2 = 0.6 to 1.6V
Feedback Voltage Offset Ch 1 to Ch 2
VVC1/2 = 1V
Feedback Bias Current Ch 1/2
VFB1/2 = 0.8V, VVC1/2 = 1V
TJ Output Voltage (Note 7)
TJ = 25°C, ITJ = 25µA, Temperature = 25°C
ITJ = 25µA, Temperature = 125°C
ITJ = 25µA, Temperature = –40°C
TJ Error
MIN
TYP
MAX
UNITS
l
780
806
830
mV
l
–12
0
12
mV
l
0
85
200
250
1.23
–380
nA
mV
V
mV
–60
0
60
mV
Error Amp gm Ch 1/2
VVC1/2 = 1V, IVC1/2 = ±5µA
350
400
450
µMho
Error Amp Source Current Ch 1/2
VFB1/2 = 0.6V, VVC1/2 = 1V
19
25
31
µA
Error Amp Sink Current Ch 1/2
VFB1/2 =1V, VVC1/2 = 1V
22
28
34
µA
Error Amp High Clamp Ch 1/2
VFB1/2 = 0.6V
1.7
1.9
2.1
V
0.75
0.9
1.05
V
9.5
12
14.5
µA
l
Error Amp Switching Threshold Ch 1/2
VFB1/2 = 0.6V
Soft-Start Source Current Ch 1/2
VFB1/2 = 0.9V, VSS1/2 = 0.05V
l
Soft-Start VOH Ch 1/2
VFB1/2 = 0.9V
1.9
2.15
2.4
Soft-Start Sink Current Ch 1/2
VFB1/2 = 0.4V, VSS1/2 = 2V
0.9
1.4
2
mA
Soft-Start VOL Ch 1/2
VFB1/2 = 0V
120
160
200
mV
Soft-Start to Feedback Offset Ch 1/2
VVC1/2 = 1V, VSS1/2 = 0.4V
–12
0
12
mV
Soft-Start Sink Current Ch 1/2 POR
VFB1/2 = 0V, VSS1/2 = 0.12V (Note 8)
150
400
600
µA
Soft-Start POR Threshold Ch 1/2
VFB1/2 = 0V (Note 8)
70
90
120
mV
Soft-Start SW Disable Ch 1/2
VFB1/2 = 0V (Note 8)
95
115
150
mV
CMPI Bias Current Ch 1/2
VCMPI1/2 = 0.8V
CMPO Leakage Ch 1/2
VCMP1/2 = 0.8V, VCMPO1/2 = 25V,
CMPI Threshold Ch 1/2
VCMPI1/2 Rising
CMPI Threshold Ch 1/2 of VFB1/2
VCMPI1/2 Rising (Note 9)
86
90
94
%
CMPI Hysteresis Ch 1/2
VCMPI1/2
35
60
85
mV
CMPO Sink Current Ch 1/2
VCMPI1/2 = 0.6V, VCMPO1/2 = 0.2V,
RT/SYNC Reference Current
VFB1/2 = 0.9V, VRT/SYNC = 0.5V
l
–100
l
l
700
V
0
100
nA
70
200
nA
720
740
mV
200
300
400
µA
11.3
12
12.7
µA
Minimum Switching Frequency
RRT/SYNC =0Ω
50
110
150
kHz
Switching Frequency
RRT/SYNC = 28k
925
1
1075
MHz
Maximum Switching Frequency
RRT/SYNC =100k
2.25
2.5
2.75
MHz
Switching Phase Angle Ch 1 ≥ Ch 2
DIV Reference Current
185
VFB1/2 = 0.9V, VDIV = 0.5V
l
Deg
10.5
12
13.5
µA
CH1 DIV 2 Threshold
RRT/SYNC = 0V
0.51
0.58
0.61
V
CH1 DIV 4 Threshold
RRT/SYNC = 0V
0.9
1.05
1.1
V
1.45
CH1 DIV 8 Threshold
RRT/SYNC = 0V
CLKOUT VOL
ICLKOUT = –100µA
1.55
V
0.25
V
CLKOUT VOH
ICLKOUT = 100µA
2
V
CLKOUT to SW1ON Delay ( tDCLKOSW1)
CLKOUT Rising
60
ns
CLKOUT to SW2ON Delay ( tDCLKOSW2)
CLKOUT Falling
30
ns
3692fa
For more information www.linear.com/3692
3
LT3692
Electrical
Characteristics
The
l denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VVIN1/2 = 15V unless otherwise specified. (Note 3)
PARAMETER
CONDITIONS
RT/SYNC to CLKOUT Delay ( tDRTSYNCH)
VRT/SYNC = 0V to 2V Rising Edge
MIN
RT/SYNC to CLKOUT Delay ( tDRTSYNCL)
VRT/SYNC = 2V to 0V Falling Edge
MAX
300
250
SYNC Frequency = 250kHz
UNITS
ns
150
SYNC Frequency Range
SYNC Phase Angle Ch 1 to Ch 2
TYP
ns
2000
kHz
180
Deg
Minimum Switch On-Time Ch 1/2
180
ns
Minimum Switch Off-Time Ch 1/2
200
ns
Minimum Boost for 100% DC Ch 1/2 (Note 10)
1.4
IND + VOUT Current Ch 1/2
VVOUT1/2 = 0V
VVOUT1/2 = 5V
ILIM1/2 Reference Current
VFB1/2 = 0.9V, VILIM = 0.4V
IND to VOUT Maximum Current Ch 1/2
2.2
V
1.5
0.5
5
5
µA
µA
l
10
12
14
µA
VILIM1/2 = 0V, VVOUT = 1V (Note 11)
VILIM1/2 = 0V, VVOUT = 5V (Note 11)
VILIM1/2 = 0.5V, VVOUT = 1V (Note 11)
VILIM1/2 = 0.5V, VVOUT = 5V (Note 11)
VILIM1/2 = 1.5V, VVOUT = 1V (Note 11)
VILIM1/2 = 1.5V, VVOUT = 5V (Note 11)
l
l
1
1.25
1.6
1.8
3.8
3.8
1.8
2
2.6
2.8
4.8
4.8
2.6
2.75
3.6
3.8
5.8
5.8
A
A
A
A
A
A
l
1
5.0
Switch Leakage Current Ch 1/2
VSW1/2 = 0V
Switch Saturation Voltage Ch 1/2
ISW1/2 = 500mA, VBST1/2 = 18V
ISW1/2 = 3A, VBST1/2 = 18V
Boost Current Ch 1/2
ISW1/2 = 500mA, VBST1/2 = 18V
ISW1/2 = 3A, VBST1/2 = 18V
Minimum Boost Voltage Ch1/2 (Note 12)
ISW1/2 = 3A, VBST1/2 = 18V
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: Absolute Maximum Voltage at VIN1/2 and SHDN1/2 pins is 60V for
nonrepetitive 1 second transients and 40V for continuous operation.
Note 3: The LT3692EUH is guaranteed to meet performance specifications
from 0°C to 125°C junction temperature. Specifications over the –40°C
to 125°C operating junction temperature range are assured by design,
characterization and correlation with statistical process controls. The
LT3692IUH is guaranteed over the full –40°C to 125°C operating junction
temperature range.
Note 4: The SHDN pins can be connected to VIN or driven by a logic-level
source with a series current limiting resistor.
Note 5: VIN undervoltage lockout is defined as the voltage which the VIN
pin must exceed for operation. The threshold guarantees that internal bias
lines are regulated and switching frequency is constant. Actual minimum
input voltage to maintain a regulated output will depend upon output
voltage and load current. See Applications Information.
Note 6: VIN overvoltage lockout is defined as the voltage when exceeded
halts converter operation. See Applications Information.
4
1.8
100
300
µA
mV
mV
9
40
13
55
17
70
mA
mA
1.75
2.0
2.5
V
Note 7: The TJ output voltage represents the temperature at the center
of the die while dissipating quiescent power. Due to switch power
dissipation and temperature gradients across the die, the TJ output
voltage measurement does not guarantee that absolute maximum junction
temperature will not be exceeded.
Note 8: An internal power on reset (POR) latch is set on the positive
transition of the SHDN1/2 pin through its threshold, thermal shutdown or
overvoltage lockout. The output of the latch activates current sources on
each SS pin which typically sink 400µA and discharge the SS capacitor.
The latch is reset when both SS pins are driven below the soft-start POR
threshold or the SHDN pin is taken below its threshold.
Note 9: The threshold is expressed as a percentage of the feedback
reference voltage for the channel.
Note 10: To enhance dropout operation, the output switch will be turned
off for the minimum off-time only when the voltage across the boost
capacitor drops below the minimum boost for 100% duty cycle threshold.
Note 11: The IND to VOUT maximum current is defined as the value of
current flowing from the IND pin to the VOUT pin which resets the switch
latch when the VC pin is at its high clamp.
Note 12: This is the minimum voltage across the boost capacitor needed
to guarantee full saturation of the internal power switch.
3692fa
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LT3692
Typical Performance Characteristics
50
3.5
39.5
VIN = 15V
VSHDN = 15V
40
2.5
2.0
SHUTDOWN
THRESHOLD VOLTAGE
1.5
25
20
0
0
–50 –25
25 50 75 100 125 150
TEMPERATURE (°C)
0
35.0
–50 –25
6000
10
9
5000
FB Voltage and CH1-CH2 FB
Offset vs Temperature
820
VSHDN1 = VSHDN2 = 0V
2000 125°C
FEEDBACK VOLTAGE (V)
CURRENT (µA)
–50°C
IQ1
6
5
4
3
2
1000
0.2
0.4
1
0.6 0.8
VOLTAGE (V)
1.2
0
–50 –25
1.4
3692 G04
0.25 RTJ = 30k
TO GND
0
RTJ = 30k TO –1V
0
3692 G07
–2
CH2
805
–4
CH1
0
–8
25 50 75 100 125 150
TEMPERATURE (°C)
3692 G06
Soft-Start-to-Feedback Offset
vs Temperature
4
VSS = 0.4V
3
415
2
410
405
400
395
390
385
1
0
–1
–2
380
–3
375
25 50 75 100 125 150
TEMPERATURE (°C)
0
800
–50 –25
25 50 75 100 125 150
TEMPERATURE (°C)
VOLTAGE (mV)
TRANSCONDUCTANCE (µMho)
0.50
–0.50
–50 –25
0
420
1.25
–0.25
IQ2
425
0.75
810
Error Amplifier Transconductance
vs Temperature
1.50
2
OFFSET
3692 G05
TJ Output Voltage vs Temperature
1.00
4
–6
1
0
815
OFFSET VOLTAGE (mV)
7
8
6
8
25°C
25 50 75 100 125 150
TEMPERATURE (°C)
3692 G03
Shutdown Quiescent Current
vs Temperature
4000
0
3692 G02
VIN1 Quiescent Current
vs SHDN1 Voltage
3000
37.0
35.5
25 50 75 100 125 150
TEMPERATURE (°C)
3692 G01
150°C
37.5
36.0
VIN = 15V
VSHDN = 1.3V
5
0
–50 –25
38.0
36.5
10
0.5
IQ (µA)
38.5
30
15
1.0
VOLTAGE (V)
39.0
VOLTAGE (V)
35
CURRENT (µA)
VOLTAGE (V)
40.0
45
MINIMUM
INPUT VOLTAGE
3.0
0
VIN Overvoltage Threshold
vs Temperature
Shutdown Pin Input Current
vs Temperature
Shutdown Threshold and Minimum
Input Voltage vs Temperature
370
–50 –30 –10 10 30 50 70 90 110 130 150
TEMPERATURE (°C)
3692 G08
–4
–50 –25
0
25 50 75 100 125 150
TEMPERATURE (°C)
3692 G09
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5
LT3692
Typical Performance Characteristics
Comparator Sink Current
vs Temperature
Comparator Thresholds
vs Temperature
750
400
740
RISING
THRESHOLD
CURRENT (µA)
700
690
680
0
200
150
1200
800
600
400
0
CLKOUT-to-SW1 Delay
vs Temperature
RT/SYNC-to-CLKOUT and SW1
Delay vs Temperature
400
191
140
350
189
130
187
120
DELAY (ns)
450
150
DELAY (ns)
160
193
110
100
200
150
179
80
100
177
70
50
60
–50 –25
0
MAXIMUM RT/SYNC DUTY CYCLE
÷4
1.0
0.8
÷2
0.6
MINIMUM RT/SYNC DUTY CYCLE
0.4
0
–50 –25
0
25 50 75 100 125 150
TEMPERATURE (°C)
3692 G16
0
–50 –25
200
ISW = 1A
150
100
ISW = 500mA
50
0.2
10
6
1.2
VOLTAGE (mV)
40
ISW = 3A
250
1.4
50
20
300
÷8
1.6
70
30
Switch Saturation Voltage
vs Temperature
1.8
RT/SYNC FREQUENCY = 1MHz
60
25 50 75 100 125 150
TEMPERATURE (°C)
3695 G15
DIV Voltage Threshold
vs Temperature
VOLTAGE (V)
DUTY CYCLE (%)
80
0
3692 G14
Synchronization Duty Cycles
vs Temperature
90
0
–50 –25
25 50 75 100 125 150
TEMPERATURE (°C)
3692 G13
100
CLKOUT
250
90
25 50 75 100 125 150
TEMPERATURE (°C)
SW1
300
181
0
25 50 75 100 125 150
TEMPERATURE (°C)
3692 G12
195
175
–50 –25
0
3692 G11
Switching Phase vs Temperature
183
RRT/SYNC = 0k
0
–50 –25
25 50 75 100 125 150
TEMPERATURE (°C)
3692 G10
185
RRT/SYNC = 13.0k
200
0
–50 –25
25 50 75 100 125 150
TEMPERATURE (°C)
RRT/SYNC = 28.0k
1000
50
660
650
–50 –25
250
100
FALLING
THRESHOLD
670
1400
300
710
RRT/SYNC = 44.2k
1600
FREQUENCY (kHz)
730
VOLTAGE (mV)
1800
SINK CURRENT AT VCMPO = 0.4V
350
720
PHASE (DEG)
Switching Frequency
vs Temperature
0
25 50 75 100 125 150
TEMPERATURE (°C)
3695 G17
0
–50 –25
0
25 50 75 100 125 150
TEMPERATURE (°C)
3692 G18
3692fa
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LT3692
Typical Performance Characteristics
Switch Peak Current
vs Temperature
5.0
90
4.5
80
4.0
70
3.5
3A
50
40
30
10
0
–50 –25
0
2.50
3.0
VILIM = 0.5V
2.5
VILIM = 0V
1.5
0
–50 –25
0
90
2500
Efficiency, TJ vs Load Current
88
EFFICIENCY
84
20
82
TJ
80
30 40 50 60
RESISTANCE (kΩ)
70
80
90
78
VIN = 12V
VOUT = 5V
FREQ = 1MHz
0
0.5
1
1.5 2
2.5 3
LOAD CURRENT (A)
Efficiency, TJ vs Load Current
3.5
30
72
20
VIN = 12V
VOUT = 2.5V
FREQ = 1MHz
1
1.5 2 2.5 3
LOAD CURRENT (A)
3.5
10
4
0
EFFICIENCY (%)
TJ
0.5
TJ
40
30
76
10
72
0
70
20
VIN = 12V
VOUT = 3.3V
FREQ = 1MHz
0
0.5
1
1.5 2 2.5 3
LOAD CURRENT (A)
4
0
3692 G24
Efficiency, TJ vs Load Current
60
EFFICIENCY
50
74
40
72
TJ
70
30
68
20
66
64
VIN = 12V
VOUT = 1.8V
FREQ = 1MHz
62
60
3.5
10
TJ (°C)
40
TJ (°C)
76
0
50
78
74
76
50
70
4
60
EFFICIENCY
80
20
78
60
EFFICIENCY
78
68
80
70
74
82
70
3692 G23
82
80
60
30
76
70
84
40
3692 G22
EFFICIENCY (%)
10
70
50
74
0
Efficiency, TJ vs Load Current
86
TJ (°C)
EFFICIENCY (%)
86
72
0
25 50 75 100 125 150
TEMPERATURE (°C)
3692 G21
80
TJ (°C)
FREQUENCY (kHz)
2000
500
0
3692 G20
Switching Frequency
vs RT/SYNC Resistance
1000
1.75
1.00
–50 –25
25 50 75 100 125 150
TEMPERATURE (°C)
3692 G19
1500
2.00
1.25
0.5
25 50 75 100 125 150
TEMPERATURE (°C)
2.25
1.50
1.0
0.5A
ISW = 3A
2.75
VILIM = 1.5V
2.0
1A
20
3.00
EFFICIENCY (%)
60
Minimum Boost Voltage
vs Temperature
VOLTAGE (V)
100
PEAK CURRENT (A)
BOOST CURRENT (mA)
Boost Current vs Temperature
0
0.5
3692 G25
1
1.5 2
2.5 3
LOAD CURRENT (A)
3.5
10
4
0
3692 G26
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LT3692
Pin Functions
BST1/2: The BST pin provides a higher than VIN base
drive to the power NPN to ensure a low switch drop. If the
voltage between the BST pin and the VIN pin is less than
the voltage required to fully turn on the power NPN, the
power switch is turned off to recharge the BST capacitor.
CMPI1/2: The CMPI pin is an input to a comparator with a
threshold of 720mV and 60mv of hysteresis. Connecting
the CMPI pin to the FB pin will generate a power good
signal when the output is within 90% of its regulated value.
CMPO1/2: The CMPO pin is an open-collector output that
sinks current when the CMPI pin falls below its threshold.
For a typical input voltage above 2.8V, its output state remains true, although during shutdown, VIN1 undervoltage
lockout or thermal shutdown, its current sink capability is
reduced. The COMPO pins can be left open circuit or tied
together to form a single power good signal.
DIV: The voltage present at the DIV pin determines the ratio
of channel 1 frequency to the master clock frequency set
by the RT/SYNC pin. The DIV pin is driven by an internal
current source with a typical value of 12µA which allows
a single resistor from the DIV pin to ground to set the
DIV voltage and resulting channel 1 frequency divider.
Ratios of 1, 2, 4 and 8 are available. See the Applications
Information section for more information.
DNC: Do Not Connect.
GND: The exposed pad pin is the only ground connection for the device. The exposed pad should be soldered
to a large copper area to reduce thermal resistance. The
GND pin is common to both channels and also serves as
small-signal ground. For ideal operation all small-signal
ground paths should connect to the GND pin at a single
point avoiding any high current ground returns.
FB1/2: The FB pin is the negative input to the error amplifier.
The output switches to regulate this pin to 806mV with
respect to the exposed ground pad. Bias current flows
out of the FB pin.
8
ILIM1/2: The voltage present at the ILIM pin determines
the peak inductor current for the channel. The ILIM pin is
driven by an internal current source with a typical value
of 12µA. A resistor from the ILIM pin to ground sets the
ILIM voltage. The maximum current limit range is 4.8A to
2A when the ILIM voltages are 1.5V and 0V respectively.
IND1/2: The IND pin is the input to the internal sense resistor
that measures current flowing in the inductor. When the
current in the resistor exceeds the current dictated by the
VC pin, the SW latch is held in reset, disabling the output
switch. Bias current flows out of the IND pin.
RT/SYNC: The voltage present at the RT/SYNC pin determines the constant switching frequency. The RT/SYNC
pin is driven by an internal current source with a typical
value of 12µA which allows a single resistor from the RT/
SYNC pin to ground to set the RT/SYNC voltage and resulting switching frequency. Minimum switching frequency
is typically 110kHz when VRT/SYNC is 0V and maximum
switching frequency is typically 2.5MHz when VRT/SYNC
is above 950mV.
Driving the RT/SYNC pin with an external clock signal will
synchronize the switch to the applied frequency. Synchronization occurs on the rising edge of the clock signal after
the clock signal is detected. Each rising clock edge initiates
an oscillator ramp reset. A gain control loop servos the
oscillator charging current to maintain constant oscillator
amplitude. Hence, the slope compensation and channel
phase relationship remain unchanged. If the clock signal
is removed, the oscillator reverts to resistor mode after
the synchronization detection circuitry times out. The clock
source impedance should be set such that the current out
of the RT/SYNC pin in resistor mode generates a frequency
roughly equivalent to the synchronization frequency. See
the Applications Information section for more information.
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LT3692
pin functions
SHDN1/2: The shutdown pin is used to control each
channel’s operation. In addition to controlling channel 1,
the SHDN1 pin also activates control circuitry for both
channels and must be present for channel 2 to operate.
When SHDN1 is below its threshold, switching on both
channels is halted. Further reducing the SHDN1 voltage
to 0.6V reduces the quiescent current to a typical value of
6µA. If the shutdown features are not used, the SHDN pin
can be tied to VIN. If SHDN pin is driven by a logic signal,
a series resistor is required. See Applications Information.
SS1/2: Current flowing out the SS pin into an external
capacitor defines the rise time of the output voltage. When
the SS pin is lower than the 0.8V reference, the feedback
is regulated to the SS voltage. When the SS pin exceeds
the reference voltage, the output will regulate the FB pin
voltage to 0.8V and the SS pin will continue to rise until
its clamp voltage. During an output overload, the VC pin is
driven above the maximum switch current level activating
its voltage clamp. When the VC clamp is activated, the SS
pin is discharged until the output reaches a regulation point
that the maximum output current can maintain. When the
overload condition is removed, the output soft starts from
that voltage. In the case of a SHDN or thermal shutdown
event, a power on reset latch ensures the capacitors on
both channels are fully discharged before either is released.
Connecting both SS pins together ensures the outputs
track together.
CLKOUT: The CLKOUT pin generates a square wave of 0V
to 2.5V which is synchronized to the internal oscillator. If
the switching frequency is set by an external resistor the
resultant clock duty cycle will be 50%. If the RT/SYNC pin
is driven by an external clock source, the resultant CLKOUT
duty cycle will mirror the external source.
TJ: The TJ pin outputs a voltage proportional to junction
temperature. The pin is 250mV for 25°C and has a slope
of 10mV/°C. See the Applications Information section for
more information.
VC1/2: The VC pin is the output of the error amplifier and the
input to the peak switch current comparator. It is normally
used for frequency compensation, but can also be used
as a current clamp or control loop override. If the error
amplifier drives VC above the maximum switch current
level, a voltage clamp activates. This indicates that the
output is overloaded and current is pulled from the SS
pin reducing the regulation point.
VIN1: The VIN1 pin powers the internal control circuitry for
both channels and is monitored by overvoltage/undervoltage lockout comparators. The VIN1 pin is also connected
to the collector of channel 1’s on-chip power NPN switch.
The VIN1 pin has high dI/dt edges and must be decoupled
to ground close to the pin of the device.
VIN2: The VIN2 pin powers the output stage for channel 2
and is monitored by overvoltage/undervoltage lockout
comparators. VIN1 voltage must be greater than 2.8V for
VIN2 operation. The VIN2 pin is also the collector of channel
2’s on-chip power NPN switch. The VIN2 pin has high dI/
dt edges and must be decoupled to ground close to the
pin of the device.
VOUT1/2: The VOUT pin is the output to the internal sense
resistor that measures current flowing in the inductor.
When the current in the resistor exceeds the current dictated by the VC pin, the SW latch is held in reset disabling
the output switch. Bias current flows out of the VOUT pin.
SW1/2: The SW pin is the emitter of the internal power
NPN. At switch off, the inductor will drive this pin below
ground with a high dV/dt. An external Schottky catch
diode to ground, close to the SW pin and respective VIN
decoupling capacitor’s ground, must be used to prevent
this pin from excessive negative voltages.
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9
LT3692
Block Diagram
VIN1
1.3V
SHDN1
+
–
+
–
VIN1
38V
CHANNEL 1
THERMAL
SHUTDOWN
2.5V
12µA
SS1
S
–
+
90mV
–
+
S
PRE
PRE
Q
R
DROPOUT
ENHANCEMENT
BST1
DRIVER
CIRCUITRY
SW1
+
–
Q
R
IND1
VOUT1
VC1
R1
FB1
2.5V
ILIM1
RLIM
CMPI1
12µA
SLOPE
COMPENSATION
2.5V
0.72V
CMPO1
+
–
TJ
CLK1
2.5V
DIV
0.806V
12µA
RT/SYNC
R3
+
+
–
R2
12µA
OSCILLATOR
AND AGC
MASTER CLOCK
VIN1
CLK2 TO CHANNEL 2
2.8V
+
–
INTERNAL
REGULATOR
AND
REFERENCES
2.5V
CLKOUT
GND
RDIV
3692 F01
Figure 1. LT3692 Block Diagram
Applications Information
The LT3692 is a dual channel, constant frequency, current
mode buck converter with internal 3.5A switches. Each
channel can be independently controlled with the exception
that VIN1 must be above the 2.8V undervoltage lockout
threshold to power the common internal regulator, oscillator and thermometer circuitry.
If the SHDN1 pin is taken below its 1.3V threshold switching on both channels will be disabled. Further reducing the
SHDN1 below a typical value of 0.6V will place the LT3692
in a low quiescent current mode. In this mode the LT3692
typically draws 6µA from VIN1 and <1µA from VIN2. When
the SHDN pin is driven above 1.3V, the internal bias circuits
turn on generating an internal regulated voltage, 0.806VFB,
12µA RT/SYNC, DIV and ILIM current references, and a
POR signal which sets the soft-start latch.
10
Once the internal reference reaches its regulation point,
the internal oscillator will start generating a master clock
signal for the two regulators at a frequency determined
by the voltage present at the RT/SYNC pin. The channel 1
clock is then divided by 1, 2, 4 or 8 depending on the
voltage present at the DIV pin. Channel 2’s clock runs at
the master clock frequency with a 180° phase shift from
channel 1.
Alternatively, if a synchronization signal is detected by
the LT3692 the RT/SYNC pin, the master clock will be
generated at the incoming frequency on the rising edge
of the synchronization pulse with channel 1 in phase with
the synchronization signal. Frequency division and phase
remains the same as the internally generated master clock.
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LT3692
Applications Information
In addition, the internal slope compensation will be automatically adjusted to prevent subharmonic oscillation
during synchronization. In either mode of oscillator operation, a square wave with the master clock frequency,
synchronized to channel 1 is present at the CLKOUT pin.
The two regulators are constant frequency, current mode
step-down converters. Current mode regulators are controlled by an internal clock and two feedback loops that
control the duty cycle of the power switch. In addition to
the normal error amplifier, there is a current sense amplifier
that monitors switch current on a cycle-by-cycle basis.
This technique means that the error amplifier commands
current to be delivered to the output rather than voltage.
A voltage fed system will have low phase shift up to the
resonant frequency of the inductor and output capacitor,
then an abrupt 180° shift will occur. The current fed system will have 90° phase shift at a much lower frequency,
but will not have the additional 90° shift until well beyond
the LC resonant frequency. This makes it much easier to
frequency compensate the feedback loop and also gives
much quicker transient response.
The Block Diagram in Figure 1 shows only one of the
switching regulators whose operation will be discussed
below. The additional regulator will operate in a similar
manner with the exception that its clock will be 180° out
of phase with the other regulator.
When, during power-up, an internal POR signal sets the
soft-start latch, both SS pins will be discharged to ground
to ensure proper start-up operation. When the SS pin voltage drops below 90mV, the VC pin is driven low disabling
switching and the soft-start latch is reset. Once the latch
is reset the soft-start capacitor starts to charge with a
typical value of 12µA.
As the voltage rises above 115mV on the SS pin, the VC
pin will be driven high by the error amplifier. When the
voltage on the VC pin exceeds 0.8V, the clock set-pulse sets
the driver flip-flop, which turns on the internal power NPN
switch. This causes current from VIN, through the NPN
switch, inductor and internal sense resistor to increase.
When the voltage drop across the internal sense resistor
exceeds a predetermined level set by the voltage on the
VC pin, the flip-flop is reset and the internal NPN switch
is turned off. Once the switch is turned off the inductor
will drive the voltage at the SW pin low until the external
Schottky diode starts to conduct, decreasing the current
in the inductor. The cycle is repeated with the start of each
clock cycle. However, if the internal sense resistor voltage
exceeds the predetermined level at the start of a clock cycle,
the flip-flop will not be set resulting in a further decrease
in inductor current. Since the output current is controlled
by the VC voltage, output regulation is achieved by the
error amplifier continually adjusting the VC pin voltage.
The error amplifier is a transconductance amplifier that
compares the FB voltage to the lowest voltage present at
either the SS pin or an internal 806mV reference. Compensation of the loop is easily achieved with a simple capacitor
or series resistor/capacitor from the VC pin to ground.
The regulators’ maximum output current occurs when the
VC pin is driven to its maximum clamp value by the error
amplifier. The value of the maximum switch current can be
programmed from 4.8A to 2A by placing a resistor from
the ILIM pin to ground.
Since the SS pin is driven by a constant current source, a
single capacitor on the soft-start pin will generate controlled
linear ramp on the output voltage.
If the current demanded by the output exceeds the maximum current dictated by the VC pin clamp, the SS pin
will be discharged, lowering the regulation point until the
output voltage can be supported by the maximum current.
Once the overload condition is removed, the regulator will
soft-start from the overload regulation point.
Shutdown control, VIN overvoltage, or thermal shutdown
will set the soft-start latch, resulting in a complete softstart sequence.
The switch driver operates from either the VIN or BST voltage. An external diode and capacitor are used to generate
a drive voltage higher than VIN to saturate the output NPN
and maintain high efficiency. If the BST capacitor voltage
is sufficient, the switch is allowed to operate to 100% duty
cycle. If the boost capacitor discharges towards a level
insufficient to drive the output NPN, a BST pin comparator forces a minimum cycle off time, allowing the boost
capacitor to recharge.
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LT3692
Applications Information
A comparator with a threshold of 720mV and 60mV of
hysteresis is provided for detecting error conditions. The
CMPO output is an open-collector NPN that is off when
the CMPI pin is above the threshold allowing a resistor
to pull the CMPO pin to a desired voltage.
The voltage present at the TJ pin is proportional to the
junction temperature of the LT3692. The TJ pin will be
250mV for a die temperature of 25°C and will have a
slope of 10mV/°C.
Choosing the Output Voltage
The output voltage is programmed with a resistor divider
between the output and the FB pin. Choose the 1% resistors according to:
V

R1= R2 •  OUT – 1
 0.806 
the RT/SYNC pin which determines the master oscillator
frequency as illustrated in Figure 2. A 0V to 2.5V square
wave with the same frequency as the master oscillator
and in phase with channel 1 is output via the CLKOUT pin.
The CLKOUT signal can be used to synchronize multiple
switching regulators.
To alleviate duty cycle restrictions due to minimum switchon times, channel 1’s switching frequency can be divided
from the master clock by 1, 2, 4 or 8 determined by resistor
RDIV in Figure 1. Channel 2’s switching frequency is not
affected by the DIV pin. The DIV pin is driven by a 12µA
current source. Setting resistor RDIV sets the voltage present at the DIV pin which determines the divisor as shown
in Table 1. The DIV pin doesn’t have any input hysteresis
near the ratio thresholds.
Table 1. Channel 1 Divisor vs VDIV
R2 should be 10k or less to avoid bias current errors. Reference designators refer to the Block Diagram in Figure 1.
2500
FREQUENCY (kHz)
1500
1000
RDIV
1
0
0.5V < VDIV < 1.0V
2
62k
1.0V < VDIV < 1.5V
4
100k
1.5V < VDIV
8
150k
The maximum recommended frequency can be approximated by the equation:
500
0
10
20
30 40 50 60
RESISTANCE (kΩ)
70
80
90
3692 F02
Figure 2. Switching Frequency vs RT/SYNC Resistance
Choosing the Switching Frequency
The LT3692 switching frequency is set by resistor R3 in
Figure 1. The RT/SYNC pin is driven by a 12µA current
source. Setting resistor R3 sets the voltage present at
12
FREQUENCY RATIO
VDIV < 0.5V
The switching frequency is typically set as high as possible to reduce overall solution size. The LT3692 employs
techniques to enhance dropout at high frequencies but
efficiency and maximum input voltage decrease due to
switching losses and minimum switch on times.
2000
0
DIV VOLTAGE
Frequency (Hz) =
VOUT + VD
1
•
VIN – VSW + VD tON(MIN)
where VD is the forward voltage drop of the catch diode (D1
Figure 2), VSW is the voltage drop of the internal switch,
and tON(MIN) in the minimum on-time of the switch.
The following example along with the data in Table 2
illustrates the trade-offs of switch frequency selection for
a single input voltage system.
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LT3692
Applications Information
Table 2. Efficiency and Size Comparisons for Different RRT/SYNC Values, 3.3V Output
FREQUENCY
RT/SYNC
EFFICIENCY
VVIN1/2 = 12V
VIN(MAX)†
L*
C*
C + L (Area)
250kHz
5.90kΩ
77.8%
38V
12µH
120µF
59.8mm2
500kHz
13.0kΩ
81.2%
31V
6.8µH
60µF
54.6mm2
1000kHz
28.0kΩ
80.5%
16V
3.3µH
30µF
51.9mm2
1500kHz
44.2kΩ
79.3%
10V
1.5µH
22µF
46.9mm2
2250kHz
71.5kΩ
76.7%
6.5V
0.82µH
15µF
19.1mm2
†V
IN(MAX) is defined as the highest input voltage that maintains constant output voltage ripple.
*Inductor and capacitor values chosen for stability and constant ripple current.
Example.
VIN = 25V, VOUT = 3.3V, IOUT = 2.5A, tON(MIN) = 250ns,
VD = 0.6V, VSW = 0.4V:
Max Frequency =
3.3 + 0.6
1
•
~ 600kHz
25 – 0.4 + 0.6 250e-9
RT/SYNC ~ 15.8kΩ (Figure 2 )
has the same effect as lowering the clock frequency for a
fixed off time, resulting in a higher duty cycle and lower
minimum input voltage. The resultant duty cycle depends
on the charging times of the boost capacitor and can be
approximated by the following equation:
DCMAX =
Input Voltage Range
Once the switching frequency has been determined, the
input voltage range of the regulator can be determined. The
minimum input voltage is determined by either the LT3692’s
minimum operating voltage of ~2.8V, or by its maximum
duty cycle. The duty cycle is the fraction of time that the
internal switch is on during a clock cycle. Unlike most
fixed frequency regulators, the LT3692 will not switch off
at the end of each clock cycle if there is sufficient voltage
across the boost capacitor (C3 in Figure 1) to fully saturate the output switch. Forcing switch off for a minimum
time will only occur at the end of a clock cycle when the
boost capacitor needs to be recharged. This operation
1
B
This leads to a minimum input voltage of:
VIN(MIN) =
VOUT + VD
– VD + VSW
DCMAX
where VSW is the voltage drop of the internal switch.
Figure 4 shows a typical graph of minimum input voltage
vs load current for Figure 19, the 3.3V and 1.8V application.
6
VOUT = 3.3V
5
START-UP
VOLTAGE (V)
SW1
tP
SW2
tP/2
RUNNING
4
3
2
1
tP
CLKOUT
tDCLKOSW1
1+
where B is 3A divided by the typical boost current from
the Electrical Characteristics table.
tP
tP/2
1
0
3692 F03
tDCLKOSW2
0
500 1000 1500 2000 2500 3000 3500
CURRENT (mA)
3692 F04
Figure 3. Timing Diagram RT/SYNC = 28.0k, tP = 1µs, VDIV = 0V
Figure 4. Minimum Input Voltage vs Load Current
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LT3692
Applications Information
The maximum input voltage is determined by the absolute
maximum ratings of the VIN and BST pins and by the
frequency and minimum duty cycle. The minimum duty
cycle is defined as:
DCMIN = tON(MIN) • Frequency
Maximum input voltage as:
VIN(MAX) =
VOUT + VD
– VD + VSW
DCMIN
Example:
Note that the LT3692 will regulate if the input voltage is
taken above the calculated maximum voltage as long as
maximum ratings of the VIN and BST pins are not violated.
However operation in this region of input voltage will
exhibit pulse skipping behavior.
Example:
VOUT = 3.3V, IOUT = 1A, Frequency = 1MHz, Temperature
= 25°C, VSW = 0.1V, B = 50 (from boost characteristics
specification), VD = 0.4V, tON(MIN) = 225ns:
DCMAX =
1
1+
VIN(MIN) =
1
50
DCMIN1 = tON(MIN1) • Frequency/2 = 0.1125
A good first choice for the LT3692 inductor value is:
VOUT
f
If the maximum load for a single channel is lower than
2.5A, then you can decrease the value of the inductor and
operate with higher ripple current, or you can adjust the
maximum switch current for the channel via the ILIM pin.
This allows you to use a physically smaller inductor, or one
with a lower DCR resulting in higher efficiency.
SW1
tP
SW2
The peak inductor and switch current is:
tP
CLKOUT
3692 F05
tDCLKOSW2
Figure 5. Timing Diagram RT/SYNC = 28.0k,
tP = 1µs, VDIV = 0.75V
14
L=
With this value the maximum load current will be ~3.5A,
independent of input voltage. The inductor’s RMS current rating must be greater than your maximum load
current and its saturation current should be higher than
the maximum peak switch current, and will reduce the
output voltage ripple.
2 • tP
tDCLKOSW1
3.3 + 0.4
– 0.4 + 0.1= 32.6V
0.1125
where f is frequency in MHz and L is in µH.
3.3 + 0.4
– 0.4 + 0.1= 16.1V
0.225
tP/2
VIN1(MAX) =
Inductor Selection and Maximum Output Current
3.3 + 0.4
– 0.4 + 0.1= 3.48V
0.98
1/(2 • tP)
VOUT = 3.3V, IOUT = 1A, Frequency = 1MHz, Temperature
= 25°C, VSW = 0.1V, B = 50 (from boost characteristics
specification), VD = 0.4V, tON(MIN) = 225ns. VDIV = 0.75V.
= 98%
DCMIN = tON(MIN) • Frequency = 0.225
VIN(MAX) =
In cases where multiple input voltages are present, or the
VIN/VOUT ratio for channel 1 is significantly different than
channel 2, channel 1’s frequency can be divided by a factor
of 2, 4 or 8 from the programmed value by setting the DIV
pin resistor to the appropriate value. Dividing channel 1’s
frequency will increase the maximum input voltage by the
same ratio. Channel 1’s external components will have to
be chosen according to the resulting frequency.
ISW(PK) = IL(PK) = IOUT +
∆IL
2
To maintain output regulation, this peak current must be
less than the LT3692’s switch current limit, ILIM. ILIM
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LT3692
Applications Information
can be set between 2A and 4.8A for each channel via
a resistor from the ILIM pin to ground. The ILIM pin is
driven by a 12µA current source. Setting resistor R LIM
sets the voltage present at the ILIM pin which determines
the maximum switch current as illustrated in Figure 6. A
capacitor from the ILIM pin to ground, or a resistor divider
from the output, can be used to limit the peak current during start-up. If a capacitor is used it must be discharged
before power-up to ensure proper operation (see 3.3V and
1.8V 2-Stage Dual Step-Down Multi-Frequency Converter
in Typical Applications).
Referring to Figure 6, as the peak current limit is reduced,
slope compensation further reduces the peak current with
increasing duty cycle.
When the ILIM pin is used to reduce the peak switch current, the equation for inductor choice becomes:
L=
50 • VOUT
f • RILIM
where f is frequency in MHz, L in µH and R in kΩ.
4.5
When the LT3692’s input supplies are operated at different
input voltages, an input capacitor sized for that channel
should be placed as close as possible to the respective
VIN pins.
A caution regarding the use of ceramic capacitors at the
input. A ceramic input capacitor can combine with stray
inductance to form a resonant tank circuit. If power is
applied quickly (for example by plugging the circuit into
a live power source) this tank can ring, doubling the input voltage and damaging the LT3692. The solution is to
either clamp the input voltage or dampen the tank circuit
by adding a lossy capacitor in parallel with the ceramic
capacitor. For details, see Application Note 88.
Output Capacitor Selection
Typically step-down regulators are easily compensated with
an output crossover frequency that is 1/10 of the switching frequency. This means that the time that the output
capacitor must supply the output load during a transient
step is ~2 or 3 switching periods. With an allowable 1%
drop in output voltage during the step, a good starting
value for the output capacitor can be expressed by:
PEAK SWITCH CURRENT (A)
4.0
CVOUT =
Max Load Step
Frequency • 0.01• VOUT
3.5
3.0
Example:
2.5
VOUT = 3.3V, Frequency = 1MHz, Max Load Step = 2A.
2.0
1.5
1.0
0 10 20 30 40 50 60 70 80 90 100
ILIM PIN RESISTOR (kΩ)
3692 F06
Figure 6. Peak Switch Current vs ILIM Resistor
Input Capacitor Selection
Bypass the inputs of the LT3692 circuit with a 4.7µF or
higher ceramic capacitor of X7R or X5R type. A lower
value or a less expensive Y5V type can be used if there
is additional bypassing provided by bulk electrolytic or
tantalum capacitors.
CVOUT =
2
= 60µF
1E6 • 0.01• 3.3V
The calculated value is only a suggested starting value.
Increase the value if transient response needs improvement
or reduce the capacitance if size is a priority. The output
capacitor filters the inductor current to generate an output
with low voltage ripple. It also stores energy in order to
satisfy transient loads and to stabilize the LT3692’s control
loop. The switching frequency of the LT3692 determines
the value of output capacitance required. Also, the current
mode control loop doesn’t require the presence of output
capacitor series resistance (ESR). For these reasons, you
are free to use ceramic capacitors to achieve very low
output ripple and small circuit size.
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15
LT3692
Applications Information
You can also use electrolytic capacitors. The ESRs of most
aluminum electrolytics are too large to deliver low output
ripple. Tantalum and newer, lower ESR organic electrolytic
capacitors intended for power supply use, are suitable
and the manufacturers will specify the ESR. The choice of
capacitor value will be based on the ESR required for low
ripple. Because the volume of the capacitor determines
its ESR, both the size and the value will be larger than a
ceramic capacitor that would give you similar ripple performance. One benefit is that the larger capacitance may
give better transient response for large changes in load
current. Table 3 lists several capacitor vendors.
BST Pin Considerations
Table 3
VENDOR
TYPE
Taiyo Yuden
Ceramic X5R, X7R
SERIES
AVX
Ceramic X5R, X7R
Tantalum
Kemet
Tantalum
TA Organic
AL Organic
T491, T494, T495
T520
A700
Sanyo
TA/AL Orgainic
POSCAP
Panasonic
AL Organic
SP CAP
TDK
Ceramic X5R, X7R
Catch Diode
The diode D1 conducts current only during switch off
time. Use a Schottky diode to limit forward voltage drop to
increase efficiency. The Schottky diode must have a peak
reverse voltage that is equal to regulator input voltage and
sized for average forward current in normal operation.
Average forward current can be calculated from:
ID(AVG) =
IOUT
• ( VIN – VOUT )
VIN
With a shorted condition, diode current will increase to the
typical value determined by the peak switch current limit
of the LT3692 set by the ILIM pin. This is safe for short
periods of time, but it would be prudent to check with the
diode manufacturer if continuous operation under these
conditions can be tolerated.
16
The capacitor and diode tied to the BST pin generate a
voltage that is higher than the input voltage. In most cases
a 0.47µF capacitor and a small Schottky diode (such as the
CMDSH-4E) will work well. To ensure optimal performance
at duty cycles greater than 80%, use a 0.5A Schottky
diode (such as a PMEG4005). Almost any type of film or
ceramic capacitor is suitable, but the ESR should be <1Ω
to ensure it can be fully recharged during the off time of
the switch. The capacitor value can be approximated by:
CBST =
IOUT(MAX) • VOUT
5 • VIN ( VOUT – 2) • f
where IOUT(MAX) is the maximum load current.
Figure 7 shows four ways to arrange the boost circuit. The
BST pin must be more than 3V above the SW pin for full
efficiency. Generally, for outputs of 3.3V and higher the
standard circuit (Figure 7a) is the best. For lower output
voltages the boost diode can be tied to the input (Figure 7b). The circuit in Figure 7a is more efficient because
the BST pin current comes from a lower voltage source.
Figure 7c shows the boost voltage source from available
DC sources that are greater than 3V. The highest efficiency
is attained by choosing the lowest boost voltage above 3V.
For example, if you are generating 3.3V and 1.8V and the
3.3V is on whenever the 1.8V is on, the 1.8V boost diode
can be connected to the 3.3V output. In any case, you
must also be sure that the maximum voltage at the BST
pin is less than the maximum specified in the Absolute
Maximum Ratings section.
The boost circuit can also run directly from a DC voltage
that is higher than the input voltage by more than 3V, as
in Figure 7d. The diode is used to prevent damage to the
LT3692 in case VX is held low while VIN is present. The
circuit saves several components (both BST pins can be
tied to D2). However, efficiency may be lower and dissipation in the LT3692 may be higher. Also, if VX is absent, the
LT3692 will still attempt to regulate the output, but will do
so with very low efficiency and high dissipation because
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LT3692
Applications Information
D2
VIN
VIN
BST
C3
D2
VIN
SW
VIN
LT3692
GND
(7b)
D2
VX > VIN + 3V
VIN
BST
C3
VIN
SW
LT3692
BST
VIN
SW
LT3692
IND
VOUT
VBST – VSW = VX
VBST(MAX) = VIN + VX
VX(MIN) = 3V
VOUT < 3V
GND
VBST – VSW = VIN
VBST(MAX) = 2 • VIN
D2
VIN
SW
IND
VOUT
VOUT
(7a)
VX = LOWEST VIN
OR VOUT > 3V
C3
LT3692
IND
VOUT
VBST – VSW = VOUT
VBST(MAX) = VIN + VOUT
BST
IND
VOUT
VOUT < 3V
GND
(7c)
VOUT < 3V
GND
VBST – VSW = VX
VBST(MAX) = VX
VX(MIN) = VIN + 3V
3692 F07
(7d)
Figure 7. BST Pin Considerations
the switch will not be able to saturate, dropping 1.5V to
2V in conduction.
The minimum input voltage of an LT3692 application is
limited by the minimum operating voltage (<3V) and by
the maximum duty cycle as outlined above. For proper
start-up, the minimum input voltage is also limited by
the boost circuit. If the input voltage is ramped slowly, or
the LT3692 is turned on with its SS pin when the output
is already in regulation, then the boost capacitor may not
be fully charged. Because the boost capacitor is charged
with the energy stored in the inductor, the circuit will rely
on some minimum load current to get the boost circuit
running properly. This minimum load will depend on input
and output voltages, and on the arrangement of the boost
circuit. The Typical Performance Characteristics section
shows plots of the minimum load current to start and to
run as a function of input voltage for 3.3V outputs. In many
cases the discharged output capacitor will present a load
to the switcher which will allow it to start. The plots show
the worst-case situation where VIN is ramping very slowly.
Use a Schottky diode for the lowest start-up voltage.
Outputs Greater Than 6V
For outputs greater than 6V, add a resistor of 1k to 2.5k
across the inductor to damp the discontinuous ringing of
the SW node, preventing unintended SW current. The 12V
output circuit in the Typical Applications section shows
the location of this resistor.
Frequency Compensation
The LT3692 uses current mode control to regulate the
output. This simplifies loop compensation. In particular, the
LT3692 does not require the ESR of the output capacitor
for stability so you are free to use ceramic capacitors to
achieve low output ripple and small circuit size. Frequency
compensation is provided by the components tied to the
VC pin. Generally a capacitor and a resistor in series to
ground determine loop gain. In addition, there is a lower
value capacitor in parallel. This capacitor is not part of
the loop compensation but is used to filter noise at the
switching frequency.
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LT3692
Applications Information
Loop compensation determines the stability and transient
performance. Designing the compensation network is a bit
complicated and the best values depend on the application
and in particular the type of output capacitor. A practical
approach is to start with one of the circuits in this data
sheet that is similar to your application and tune the compensation network to optimize the performance. Stability
should then be checked across all operating conditions,
including load current, input voltage and temperature.
The LT1375 data sheet contains a more thorough discussion of loop compensation and describes how to test the
stability using a transient load.
Figure 8 shows an equivalent circuit for the LT3692 control
loop. The error amp is a transconductance amplifier with
finite output impedance. The power section, consisting of
the modulator, power switch and inductor, is modeled as
a transconductance amplifier generating an output current proportional to the voltage at the VC pin. Note that
the output capacitor integrates this current, and that the
capacitor on the VC pin (CC) integrates the error amplifier
output current, resulting in two poles in the loop. In most
cases a zero is required and comes from either the output
capacitor ESR or from a resistor in series with CC.
This simple model works well as long as the value of the
inductor is not too high and the loop crossover frequency
is much lower than the switching frequency. A phase lead
capacitor (CPL) across the feedback divider may improve
the transient response.
Synchronization
The RT/SYNC pin can also be used to synchronize the
regulators to an external clock source. Driving the RT/SYNC
resistor with a clock source triggers the synchronization
detection circuitry. Once synchronization is detected, the
rising edge of SW1 will be synchronized to the rising edge
of the RT/SYNC signal and the rising edge of SW2 synchronized to the falling edge of the RT/SYNC signal (see
Figures 10 and 11). During synchronization, a 0V to 2.4V
square wave with the same frequency and duty cycle as
the synchronization signal is output via the CLKOUT pin
with a typical propagation delay of 250ns. In addition, an
internal AGC loop will adjust slope compensation to avoid
subharmonic oscillation. If the synchronization signal is
halted, the synchronization detection circuitry will timeout
in typically 10µs at which time the LT3692 reverts to the
free-running frequency based on the RT/SYNC pin voltage.
The synchronizing clock signal input to the LT3692 must
have a frequency between 200kHz and 2MHz, a duty cycle
between 20% and 80%, a low state below 0.5V and a high
state above 1.6V. Synchronization signals outside of these
parameters will cause erratic switching behavior. If the
RT/SYNC pin is held above 1.6V at any time, switching
will be disabled.
If the synchronization signal is not present during regulator start-up (for example, the synchronization circuitry
is powered from the regulator output) the RT/SYNC pin
must remain below 1V until the synchronization circuitry
is active for proper start-up operation.
LT3692
CURRENT MODE
POWER STAGE
gm = 4.8mho
OUTPUT
gm = 400µmho
3.6M
RC
CF
CC
+
–
VC
ERROR
AMP
R1
CPL
ESR
FB
C1
+
0.806V
R2
C1
CERAMIC
TANTALUM
OR
POLYMER
3692 F08
Figure 8. Model for Loop Response
18
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LT3692
Applications Information
If the synchronization signal powers up in an undetermined
state (VOL, VOH, Hi-Z), connect the synchronization clock
to the LT3692 as shown in Figure 9. The circuit as shown
will isolate the synchronization signal when the output
voltage is below 90% of the regulated output. The LT3692
will start up with a switching frequency determined by the
resistor from the RT/SYNC pin to ground.
If the synchronization signal powers up in a low impedance
state (VOL), connect a resistor between the RT/SYNC pin
and the synchronizing clock. The equivalent resistance
seen from the RT/SYNC pin to ground will set the startup frequency.
If the synchronization signal powers up in a high impedance state (Hi-Z), connect a resistor from the RT/SYNC
pin to ground. The equivalent resistance seen from the
RT/SYNC pin to ground will set the start-up frequency.
VOUT1
VCC
LT3692
PG1
SYNCHRONIZATION
CIRCUITRY
RT/SYNC
CLK
3692 F09
Figure 9. Synchronous Signal Powered from Regulator’s Output
tP
tP
SW1
tP
SW2
tPON
tP
CLKOUT
tDCLKOSW1
tDCLKOSW2
tPON
tP
RT/SYNC
3692 F11
tDRTSYNCH
tDRTSYNCH
Figure 11. Timing Diagram RT/SYNC = 1MHz, Duty Cycle > 50%
Reducing Input Ripple Voltage
Synchronizing the switches to the rising and falling edges
of the synchronization signal provides the unique ability to
reduce input ripple currents in systems where VIN1 and VIN2
are connected to the same supply. Decreasing the input
current ripple reduces the required input capacitance. For
example, the input ripple voltage shown in Figure 12 for
a typical antiphase dual 14.4V to 8.5V and 14.4V to 3.3V
regulator is decreased from a peak of 472mV to 160mV
as shown in Figure 13 by driving the LT3692 with a 71%
duty cycle synchronization signal.
SW1
SW1
SW2
tP/2
tP
INPUT
RIPPLE V
SW2
tP
tP/2
RT/SYNC
3692 F12
CLKOUT
Figure 12. Dual 14.4V/8.5V, 14.4V/3.3V with 180° Phase
tDCLKOSW1
tDCLKOSW2
tP/2
SW1
tP
RT/SYNC
SW2
3692 F10
tDRTSYNC
Figure 10. Timing Diagram RT/SYNC = 1MHz, Duty Cycle = 50%
INPUT
RIPPLE V
RT/SYNC
3692 F13
Figure 13. Dual 14.4V/8.5V, 14.4V/3.3V with 256° Phase
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LT3692
Applications Information
Shutdown and Undervoltage/Overvoltage Lockout
Typically, undervoltage lockout (UVLO) is used in situations where the input supply is current limited, or has a
relatively high source resistance. A switching regulator
draws constant power from the source, so source current increases as source voltage drops. This looks like a
negative resistance load to the source and can cause the
source to current limit or latch low under low source voltage
conditions. UVLO prevents the regulator from operating
at source voltages where these problems might occur.
Overvoltage lockout (OVLO) is typically used to shut down
the switching regulator during potentially harmful input
voltage transients.
Referring to Figure 14, if the SHDN pin is connected to
VIN, then the overvoltage lockout threshold is set to the
typical maximum value of 38V.
There is no hysteresis on the SHDN pins. If the SHDN pins
are not connected to VIN, then an internal clamp regulates
the SHDN pin voltage to 2.5V.
If the SHDN pin is driven by a logic signal greater than
2.5V, a series resistor is required to limit the current into
the SHDN pin to no more than 10µA.
Referring to Figure 15, a 249k resistor will suffice for
a typical logic-level signal. If the logic signal is 5V or
greater, choose a current limiting resistor equal to RSHDN =
(VLOGIC – 2.5V)/10µA. Place a small Schottky diode (such
as a BAT54) in parallel to the current-limiting resistor as
shown in Figure 15.
Keep the connections from any series resistors to the SHDN
pins short and make sure that the interplane or surface
capacitance to switching nodes is minimized.
Additionally, an internal comparator will force both channels into shutdown below the minimum VIN1 of 2.8V.
This feature can be used to prevent excessive discharge
of battery-operated systems. In addition to the VIN1 undervoltage lockout, both channels will be disabled when
SHDN1 is less than 1.3V.
Programmable UVLO may be implemented using an input
voltage divider and one of the internal comparators (see
the Typical Applications section).
When the SHDN pin is taken above 1.3V, its respective
channel is allowed to operate. When the SHDN pin is driven
below 1.3V, its channel is disabled. Taking SHDN1 below
0.6V will place the LT3692 in a low quiescent current mode.
A graph of quiescent current vs SHDN1 voltage can be
found in the Typical Performance Characteristics section.
1.3V
SHDN1
+
–
+
–
38V
THERMAL
SHUTDOWN
LT3692
CHANNEL
DISABLE
3692 F15
Figure 14. Connect SHDN to VIN to Select Default OVLO and UVLO
RSHDN
+
–
SHDN
LT3692
SCHOTTKY
OPTIONAL
IF VLOGIC < 5V
3692 F15
Figure 15. External Control of the SHDN Pin
Soft-Start
The output of the LT3692 regulates to the lowest voltage
present at either the SS pin or an internal 0.806V reference.
A capacitor from the SS pin to ground is charged by an
internal 12µA current source resulting in a linear output
ramp from 0V to the regulated output whose duration is
given by:
VINX
20
VLOGIC
tRAMP =
CSS • 0.806V
12µA
At power-up, a reset signal sets the soft-start latch and
discharges both SS pins to approximately 0V to ensure
proper start-up. When both SS pins are fully discharged
the latch is reset and the internal 12µA current source
starts to charge the SS pin.
When the SS pin voltage is below 115mV, the VC pin is
pulled low which disables switching. This allows the SS
pin to be used as an individual shutdown for each channel.
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3692fa
LT3692
Applications Information
As the SS pin voltage rises above 90mV, the VC pin is released and the output is regulated to the SS voltage. When
the SS pin voltage exceeds the internal 0.806V reference,
the output is regulated to the reference. The SS pin voltage
will continue to rise until it is clamped at 2V.
The comparators can be used to monitor input and output
voltages as well as die temperature. See the Typical Applications circuit collection for examples.
In the event of a VIN1 undervoltage lockout, the soft-start
latch is set for both channels, triggering a full start-up
sequence. If a channel’s SHDN pin is driven below 1.3V,
its overvoltage lockout is enabled, or the internal die
temperature for its power switch exceeds its maximum
rating during normal operation, the soft-start latch is set
for that channel.
Complex output tracking and sequencing between channels
can be implemented using the LT3692’s SS and CMPO
pins. Figure 16 shows several configurations for output
tracking/sequencing for a 3.3V and 1.8V application.
In addition, if the load exceeds the maximum output switch
current, the output will start to drop causing the VC pin
clamp to be activated. As long as the VC pin is clamped,
the SS pin will be discharged. As a result, the output will
be regulated to the highest voltage that the maximum
output current can support. For example, if a 6V output is
loaded by 1Ω the SS pin will drop to 0.48V, regulating the
output at 4.8V ( 4.8A • 1Ω ). Once the overload condition
is removed, the output will soft start from the temporary
voltage level to the normal regulation point.
Ratiometric tracking is achieved in Figure 16b by connecting both SS pins together. In this configuration, the
SS pin source current is doubled (24µA) which must be
taken into account when calculating the output rise time.
Since the SS pin is clamped at 2V and has to discharge
to 0.806V before taking control of regulation, momentary
overload conditions will be tolerated without a soft-start
recovery. The typical time before the SS pin takes control is:
C • 1.2V
tSS(CONTROL) = SS
1.4mA
Open-Collector Comparators
The CMPO pin is the open-collector output of an internal
comparator. The comparator compares the CMPI pin voltage to 90% of the reference voltage (0.72V) with 60mV
of hysteresis.
The CMPO pin has a typical sink capability of 300µA when
the CMPI pin is below the threshold and can withstand 38V
when the threshold is exceeded. The CMPO pin is active
(sink capability is reduced in shutdown and undervoltage
lockout mode) as long as the VIN1 pin voltage exceeds 2.8V.
Output Tracking/Sequencing
Independent soft-start for each channel is shown in Figure 16a. The output ramp time for each channel is set by
the soft-start capacitor as described in the soft-start section.
By connecting a feedback network from VOUT1 to the SS2
pin with the same ratio that sets VOUT2 voltage, absolute
tracking shown in Figure 16c is implemented. The minimum value of the top feedback resistor (R1) should be set
such that the SS pin can be driven all the way to ground
with 1.4mA of sink current when VOUT1 is at its regulated
voltage. In addition, a small VOUT2 voltage offset will be
present due to the SS2 12µA source current. This offset
can be corrected for by slightly reducing the value of R2.
Figure 16d illustrates output sequencing. When VOUT1 is
within 10% of its regulated voltage, CMPO1 releases the
SS2 soft-start pin allowing VOUT2 to soft-start. In this case
CMPO1 will be pulled up to 2V by the SS pin. If a greater
voltage is needed for CMPO1 logic, a pull-up resistor to
VOUT1 can be used. This will decrease the soft-start ramp
time and increase tolerance to momentary shorts.
If precise output ramp up and down is required, drive the
SS pins as shown in Figure 16e. The minimum value of
resistor (R3) should be set such that the SS pin can be
driven all the way to ground with 1.4mA of sink current
during power-up and fault conditions.
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21
LT3692
Applications Information
Independent Start-Up
Ratiometric Start-Up
Absolute Start-Up
VOUT1
0.5V/DIV
VOUT1
0.5V/DIV
PG1
VOUT2
0.5V/DIV
VOUT2
0.5V/DIV
5ms/DIV
LT3692
FB1
CMPI1
0.72V
+
–
0.1µF
CMPO1
12µA
0.72V
+
–
FB1
CMPI1
2.5V
12µA
0.72V
SS1
R2
2.5V
PG2
12µA
0.72V
SS2
12µA
PG1
0.72V
+
–
R4
R5
FB2
CMPI2
2.5V
PG2
12µA
0.72V
SS2
R8
0.22µF
(16a)
PG1
VOUT2
R6
CMPO2
R3
R2
CMPO1
+
–
0.1µF
R4
FB2
CMPI2
FB1
CMPI1
2.5V
VOUT2
R5
CMPO2
R1
SS1
R6
VOUT1
R3
CMPO1
+
–
0.1µF
FB2
CMPI2
SS2
R1
PG1
R4
LT3692
VOUT1
R3
R2
VOUT2
2.5V
10ms/DIV
LT3692
R1
SS1
PG2
10ms/DIV
VOUT1
2.5V
VOUT2
0.5V/DIV
PG2
PG2
12µA
VOUT1
0.5V/DIV
PG1
PG1
CMPO2
+
–
R6
R5
PG2
R7
(16b)
(16c)
Output Sequencing
Controlled Power Up and Down
VOUT1
0.5V/DIV
VOUT1
0.5V/DIV
PG1/PG2
VOUT2
0.5V/DIV
VOUT2
0.5V/DIV
PG1
SS1/2
PG2
10ms/DIV
10ms/DIV
LT3692
LT3692
VOUT1
VOUT1
R1
FB1
CMPI1
2.5V
12µA
0.72V
SS1
+
–
0.1µF
R1
R2
2.5V
CMPO1
R4
12µA
SS2
0.72V
+
–
CMPO2
0.72V
SS1
+
–
VOUT2
2.5V
12µA
PG1
R5
FB2
CMPI2
FB1
CMPI1
+
–
R2
CMPO1
PG1
VOUT2
R4
R6
R5
FB2
CMPI2
2.5V
PG2
R3
12µA
SS2
0.72V
+
–
0.22µF
R6
R5
CMPO2
PG2
3692 F16
(16d)
(16e)
Figure 16. SS Pin Configurations
22
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LT3692
Applications Information
Single Step Down:
Application Optimization
In multiple channel applications requiring large VIN to
VOUT ratios, the maximum frequency and resulting inductor size is determined by the channel with the largest
ratio. The LT3692’s multi-frequency operation allows the
user to minimize component size for each channel while
maintaining constant frequency operation. The circuit in
Figure 17 illustrates this approach. A 2-stage step-down
approach coupled with multi-frequency operation will
further reduce external component size by allowing an
increase in frequency for the channel with the lower VIN
to VOUT ratio. The drawback to this approach is that the
output power capability for the first stage is determined
by the output power drawn from the second stage. The
dual step-down application in Figure 18 steps down the
input voltage (VIN1) to the highest output voltage then
uses that voltage to power the second output (VIN2). VOUT1
must be able to provide enough current for its output plus
VOUT2 maximum load. Note that the VOUT1 voltage must
be above VIN2’s minimum input voltage as specified in the
Electrical Characteristics (2.8V) when the second channel
starts to switch. Delaying channel 2 can be accomplished
by either independent soft-start capacitors or sequencing
with the CMP01 output.
Frequency (Hz) =
L1=
(36V – 3.3) • 3.3 ≥ 12µH
L2 =
(36V – 1.8) • 1.8 ≥ 6.8µH
Frequency (Hz) =
L=
VOUT + VD
1
•
VIN – VSW + VD tON(MIN)
( VIN – VOUT ) • VOUT
36V • 250kHz
36V • 250kHz
2-Stage Step-Down:
Frequency (Hz) =
3.3 + 0.6
1
•
≅ 400kHz
36V – 0.4 + 0.6 225ns
L1=
(36V – 3.3) • 3.3 ≥ 7.5µH
L2 =
(3.3 – 1.8) • 1.8 ≥ 2µH
36V • 400kHz
3.3 • 400kHz
2-Stage Step-Down Multi-Frequency:
RDIV = 100k, FREQ1 = 400kHz, FREQ2 = 1600kHz.
For example, assume a maximum input of 36V:
VIN = 36V, VOUT1 = 3.3V at 1.5A and VOUT2 = 1.8V at 1.5A.
1.8 + 0.6
1
•
≅ 250kHz
36V – 0.4 + 0.6 225ns
L1=
(36V – 3.3) • 3.3 ≥ 7.5µH
L2 =
(3.3 – 1.8) • 1.8 ≥ 500nH
36V • 400kHz
3.3 • 2MHz
In addition, RILIM2 = 40.2k reduces the peak current limit
on channel to 2.5A, which reduces inductor size and catch
diode requirements.
VIN • f
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23
LT3692
Applications Information
VIN
5V TO 36V
4.7µF
4.7µF
VIN2
VIN1
SHDN1
BST1
8.2µH
0.47µF
SHDN2
BST2
SW1
SW2
IND1
IND2
8.2µH
0.47µF
LT3692
VOUT1
1.8V 3A
200kHz
VOUT2
VOUT1
100µF
×3 8.06k
100pF
10k
FB1
FB2
CMPI1
CMPI2
CMPO1
CMPO2
SS1
ILIM2
VC2
RT/SYNC
680pF
DIV
10k
10.2k
CLKOUT
TJ
GND
33pF
120k
100k
8.06k
PG
SS2
ILIM1
VC1
0.1µF
100µF
100pF
24.9k
VOUT2
3.3V 3A
400kHz
CLOCKOUT
400kHz
0.1µF
62k
470pF
0.1µF
33pF
49.9k
16.5k
120k
3692 F17
Figure 17. 3.3V and 1.8V Dual Step-Down Multi-Frequency Converter
VIN
5V TO 36V
4.7µF
SHDN1
BST1
8.2µH
0.47µF
VOUT1
3.3V 2.5A
400kHz 100µF
SHDN2
BST2
SW1
SW2
IND1
IND2
0.22µF
100pF
24.9k
FB1
FB2
CMPI1
CMPI2
CMPO1
CMPO2
33pF
16.5k
ILIM2
VC2
RT/SYNC
DIV
47.5k
100k
10k
100pF
100k
8.06k
PG
SS2
ILIM1
VC1
470pF
VOUT2
1.8V 1A
47µF 1600kHz
VOUT2
VOUT1
SS1
120k
1µH
LT3692
8.06k
0.1µF
1µF
VIN2
VIN1
CLKOUT
GND
TJ
0.1µF
CLOCKOUT
1600kHz
49.9k
0.1µF
330pF
33pF
36.5k
40.2k
3692 F18
Figure 18. 3.3V and 1.8V 2-Stage Dual Step-Down Multi-Frequency Converter
24
3692fa
For more information www.linear.com/3692
LT3692
Applications Information
Shorted and Reverse Input Protection
PCB Layout
If the inductor is chosen so that it won’t saturate excessively, an LT3692 step-down regulator will tolerate a shorted
output. There is another situation to consider in systems
where the output will be held high when the input to the
LT3692 is absent. This may occur in battery charging
applications or in battery back-up systems where a battery
or some other supply is diode OR-ed with the LT3692’s
output. If the VIN1/2 pin is allowed to float and the SHDN
pin is held high (either by a logic signal or because it is
tied to VIN), then the LT3692’s internal circuitry will pull its
quiescent current through its SW pin. This is fine if your
system can tolerate a few mA in this state. If you ground
the SHDN pin, the SW pin current will drop to essentially
zero. However, if the VIN pin is grounded while the output
is held high, then parasitic diodes inside the LT3692 can
pull large currents from the output through the SW pin
and the VIN1/2 pin. Figure 19 shows a circuit that will run
only when the input voltage is present and that protects
against a shorted or reversed input.
For proper operation and minimum EMI, care must be
taken during printed circuit board (PCB) layout. Figure 20
shows the high di/dt paths in the buck regulator circuit.
Note that large switched currents flow in the power switch,
the catch diode and the input capacitor. The loop formed
by these components should be as small as possible.
PARASITIC DIODE
D4
VIN
VIN1/2
SW
VOUT1/2
LT3692
3692 F19
Figure 19. Diode D4 Prevents a Shorted Input from Discharging a
Backup Battery Tied to the Output
These components, along with the inductor and output
capacitor, should be placed on the same side of the circuit
board and their connections should be made on that layer.
Place a local, unbroken ground plane below these components, and tie this ground plane to system ground at
one location, ideally at the ground terminal of the output
capacitor C2. Additionally, the SW and BST traces should
be kept as short as possible.
Thermal Considerations
The PCB must also provide heat sinking to keep the
LT3692 cool. The exposed metal on the bottom of the
package must be soldered to a ground plane. This ground
should be tied to other copper layers below with thermal
vias; these layers will spread the heat dissipated by the
LT3692. Place additional vias near the catch diodes. Adding
more copper to the top and bottom layers and tying this
copper to the internal planes with vias can further reduce
thermal resistance. With these steps, the thermal resistance from die (or junction) to ambient can be reduced to
θJA = 35°C/W.
VIN LT3692 SW
VIN LT3692 SW
VIN LT3692 SW
GND
GND
GND
3692 F20
(20a)
(20b)
(20c)
Figure 20. Subtracting the Current when the Switch is On (20a) from the Current when the Switch is Off (20b) Reveals the Path of the
High Frequency Switching Current (20c). Keep this Loop Small. The Voltage on the SW and BST Traces will Also Be Switched; Keep
These Traces as Short as Possible. Finally, Make Sure the Circuit is Shielded with a Local Ground Plane
3692fa
For more information www.linear.com/3692
25
LT3692
Applications Information
The power dissipation in the other power components
such as catch diodes, boost diodes and inductors, cause
additional copper heating and can further increase what
the IC sees as ambient temperature. See the LT1767 data
sheet’s Thermal Considerations section.
Die Temperature and Thermal Shutdown
The LT3692 TJ pin outputs a voltage proportional to the
internal junction temperature. The TJ pin typically outputs
250mV for 25°C and has a slope of 10mV/°C. Without the
aid of external circuitry, the TJ pin output is valid from 20°C
to 150°C (200mV to 1.5V) with a maximum load of 100µA.
Generating a Negative Regulated Voltage
The simple charge pump circuit in Figure 22 uses the
CLKOUT pin output to generate a negative voltage, eliminating the need for an external regulated supply. Surface
mount capacitors and dual-package Schottky diodes
minimize the board area needed to implement the negative voltage supply.
LT3692
TJ
R1
VNEG
GND
To extend the operating temperature range of the TJ output below 20°C, connect a resistor from the TJ pin to a
negative supply as shown in Figure 21. The negative rail
voltage and TJ pin resistor may be calculated using the
following equations:
+
Full Temperature Range Measurement
3692 F23
Figure 21. Circuit to Extend the TJ Pin Operating Range
2 • TEMP(MIN)°C
100
|V |
R1 ≤ NEG
33µA
VNEG ≤
TJ
LT3692
30k
330pF
D4
CLKOUT
GND
where:
D3, D4: ZETEX BAT54S
TEMP(MIN)°C is the minimum temperature where a
valid TJ pin output is required.
0.1µF
D3
3692 F23
Figure 22. Circuit to Generate the Negative Voltage Rail to
Extend the TJ Pin Operating Range
VNEG = Regulated negative voltage supply.
For example:
TEMP(MIN)°C = –40°C
VNEG ≤ –0.8V
VNEG = –1, R1 ≤ |VNEG|/33µA = 30.2kΩ
26
3692fa
For more information www.linear.com/3692
LT3692
Applications Information
As a safeguard, the LT3692 has an additional thermal
shutdown threshold set at a typical value of 163°C for each
channel. Each time the threshold is exceeded, a power on
sequence for that channel will be initiated. The sequence
will then repeat until the thermal overload is removed.
It should be noted that the TJ pin voltage represents
a steady-state temperature and should not be used to
guarantee that maximum junction temperatures are
not exceeded. Instantaneous power along with thermal
gradients and time constants may cause portions of the
die to exceed maximum ratings and thermal shutdown
thresholds. Be sure to calculate die temperature rise for
steady state (>1Min) as well as impulse conditions.
CLKOUT Capacitive Loading
A minor drawback to generating a negative rail from the
CLKOUT pin is that the charge pump adds capacitance to
the CLKOUT pin, resulting in an output synchronization
clock signal phase delay. Figures 23 and 24 show the impact of capacitive loading on the CLKOUT signal rise and
fall times. Note that a typical 10:1 150MHz oscilloscope
probe contributes significant capacitance to the CLKOUT
node, necessitating a low capacitance probe for accurate
measurements. Applications requiring CLKOUT to generate
the negative supply voltage and provide the synchronization clock to other regulators may benefit from buffering
CLKOUT prior to the charge pump circuitry.
Other Linear Technology Publications
Application notes AN19, AN35 and AN44 contain more
detailed descriptions and design information for buck
regulators and other switching regulators. The LT1376
data sheet has a more extensive discussion of output
ripple, loop compensation and stability testing. Design
Note DN100 shows how to generate a dual (+ and –)
output supply using a buck regulator.
SCOPE PROBE: 15pF
500mV/DIV
CHARGE PUMP
SCOPE PROBE: 15pF
SYNCHRONIZED LT3692
RT/SYNC PIN
FET PROBE: 2pF
40ns/DIV
FREQUENCY: 1.000MHz
3692 F23
CHARGE PUMP
SYNCHRONIZED
LT3692 RT/SYNC PIN
500mV/DIV
FET PROBE: 2pF
20ns/DIV
FREQUENCY: 1.000MHz
Figure 23. CLKOUT Rise Time
3692 F24
Figure 24. CLKOUT Fall Time
3692fa
For more information www.linear.com/3692
27
VIN
7V TO 36V
28
VOUT1
5V 2A
500kHz
8.06k
100pF
0.47µF
For more information www.linear.com/3692
120k
0.1µF 33pF
47µF
×2
4.7µF
36.5k
28.0k
330pF
42.2k
8.2µH
VIN2
62k
DIV
GND
0.1µF
TJ
CLKOUT
ILIM1
VC1
RT/SYNC
SS2
ILIM2
VC2
SS1
CMPO2
CMPO1
FB2
VOUT2
CMPI2
LT3692
IND2
SW2
SHDN2
BST2
CMPI1
FB1
VOUT1
IND1
SW1
SHDN1
BST1
VIN1
49.9k
33pF
16.9k
2.2µH
25.5k
330pF
100pF
0.22µF
8.06k
4.7µF
49.9k
PG
100k
8.06k
10k
CLOCKOUT
1MHz
0.1µF
47µF
VOUT2
2.5V 1A
1MHz
8.06k
4.02k
VOUT3
1.2V 1A
1MHz 100µF
49.9k
33pF
8.06k
100pF
0.22µF
17.8k
470pF
4.02k
2.2µH
2.2µF
DIV
SW2
ILIM2
VC2
SS2
CMPO2
CMPI2
FB2
VOUT2
IND2
0.1µF
TJ
CLKOUT
GND
RT/SYNC
ILIM1
VC1
SS1
VIN2
SHDN2
BST2
LT3692
CMPO1
CMPI1
FB1
VOUT1
IND1
SW1
SHDN1
BST1
VIN1
Quad Output 5V, 2.5V, 1.8V and 2V Multi-Frequency Synchronized, 2-Stage Converter
with Output Sequencing, Absolute Tracking and Current Limiting
10k
2.2µH
8.06k
49.9k
33pF
3692 TA03
49.9k
VOUT4
1.8V 1A
100µF 1MHz
330pF
25.5k
0.22µF
100pF
4.7µF
LT3692
Typical Applications
3692fa
LT3692
TYPICAL APPLICATIONS
3.3V and 1.8V 2-Stage Dual Step-Down Multi-Frequency Converter
VIN
5V TO 36V
4.7µF
SHDN1
BST1
8.2µH
0.47µF
1µF
VIN2
VIN1
SHDN2
BST2
SW1
SW2
IND1
IND2
1µH
0.22µF
LT3692
VOUT1
3.3V 2.5A
400kHz 100µF
100pF
24.9k
8.06k
FB1
FB2
CMPI1
CMPI2
CMPO1
CMPO2
SS1
0.1µF
16.5k
ILIM2
VC2
RT/SYNC
DIV
47.5k
CLKOUT
TJ
GND
33pF
120k
10k
100pF
100k
8.06k
PG
SS2
ILIM1
VC1
470pF
VOUT2
1.8V 1A
47µF 1600kHz
VOUT2
VOUT1
0.1µF
CLOCKOUT
1600kHz
33pF
49.9k
100k
330pF
36.5k
0.1µF
40.2k
3692 TA02
12V to 3.3V and 2.5V Converter with Start-Up Current Limiting
VIN
12V
4.7µF
4.7µF
VIN2
VIN1
SHDN1
BST1
2.2µH
0.22µF
SHDN2
BST2
SW1
SW2
IND1
IND2
0.22µF
LT3692
VOUT1
2.5V 3A
1MHz
VOUT2
VOUT1
47µF
8.06k
100pF
16.9k
FB1
FB2
CMPI1
CMPI2
CMPO1
CMPO2
ILIM2
VC2
ILIM1
VC1
0.1µF
RT/SYNC
330pF
DIV
33pF
0.1µF
21.0k
28.0k
100pF
24.9k
8.06k
VOUT2
3.3V 3A
47µF 1MHz
SS2
SS1
120k
2.2µH
CLKOUT
GND
TJ
CLOCKOUT
1MHz
0.1µF
33pF
0.1µF
330pF
49.9k
24.9k
0.1µF
120k
3692 TA04
3692fa
For more information www.linear.com/3692
29
LT3692
TYPICAL APPLICATIONS
3.3V/6A Single Output with UVLO and 100°C Temperature Warning
VIN
6V TO 30V
4.7µF
4.7µF
80.6k
VIN2
VIN1
6.5V UVLO RISING
6V UVLO FALLING
SHDN1
BST1
10k
6.8µH
0.22µF
SHDN2
BST2
SW1
SW2
IND1
IND2
6.8µH
0.22µF
VOUT
3.3V 6A
800kHz
EFFECTIVE RIPPLE
LT3692
VOUT
VOUT2
VOUT1
47µF
100pF
8.06k
24.9k
FB1
FB2
CMPI1
CMPI2
CMPO1
SS1
CMPO2
SS2
ILIM2
VC2
ILIM1
VC1
0.1µF
RT/SYNC
680pF
DIV
10k
100°C TEMP FLAG
SS1
ILIM1
VC1
CLOCKOUT 400kHz
TJ
GND
33pF
60.4k
CLKOUT
47µF
100k
FB1
28k
R3
73.2k
0.1µF
10.2k
3692 TA05
Power Supply Dual Input Single 3.3V/4A Output Step-Down Converter
VIN1
5V TO 36V
4.7µF
SHDN1
BST1
5.6µH
0.22µF
VOUT1
3.3V
4A
2.2µF
VIN2
VIN1
VIN2
5V TO 36V
1A MAX
SHDN2
BST2
SW1
SW2
IND1
IND2
6.8µH
0.22µF
LT3692
100pF
8.06k
24.9k
FB1
CMPI1
CMPI2
CMPO1
SS1
CMPO2
DIV
33pF
120k
24.9k
10.2k
FB1
100k
PG
SS2
ILIM2
VC2
RT/SYNC
330pF
47µF
FB2
ILIM1
VC1
0.1µF
VOUT1
VOUT2
VOUT1
47µF
CLKOUT
GND
VC1
CLOCKOUT 400kHz
TJ
0.1µF
49.9k
0.1µF
3692 TA07
30
3692fa
For more information www.linear.com/3692
LT3692
TYPICAL APPLICATIONS
5V and 1.8V Dual 2-Stage Converter
VIN
6V TO 36V
1µF
4.7µF
VIN2
VIN1
SHDN1
BST1
0.22µF
VOUT1
5V 2A
500kHz
8.2µH
SHDN2
BST2
SW1
SW2
IND1
IND2
1µH
0.22µF
LT3692
VOUT2
VOUT1
47µF
×2
R9
100k
220pF
42.2k
8.06k
PG
FB1
CMPI2
CMPO1
CMPO2
SS1
0.1µF
47pF
120k
36.5k
61.9k
100k
100pF
47µF
8.06k
ILIM2
VC2
RT/SYNC
DIV
FB1
SS2
ILIM1
VC1
330pF
10k
FB2
CMPI1
VOUT2
1.8V 2A
2MHz
CLKOUT
GND
33pF
TJ
0.1µF
0.1µF
330pF
49.9k
39.2k
120k
3692 TA10
3692fa
For more information www.linear.com/3692
31
32
VOUT1
12V 1A
750kHz
For more information www.linear.com/3692
120k
0.1µF
47µF
30.1k
33pF
330pF
8.06k
100pF
0.47µF
VIN
15V TO 36V
120k
113k
1k
8.2µH
4.7µF
DIV
GND
0.1µF
TJ
CLKOUT
ILIM1
VC1
RT/SYNC
SS2
ILIM2
VC2
SS1
CMPO2
CMPO1
FB2
VOUT2
CMPI2
LT3692
IND2
SW2
SHDN2
BST2
VIN2
CMPI1
FB1
VOUT1
IND1
SW1
SHDN1
BST1
VIN1
24.9k
4.7µH
49.9k
33pF
2.2µF
8.06k
330pF
24.9k
100pF
0.22µF
133k
0.1µF
49.9k
PG
47µF
100k
VOUT2
3.3V 2A
750kHz
OUT1
OUT2
MOD
GND
SET
LTC6908-1
V+
0.1µF
VOUT3
1.2V 3A
375kHz 100µF
×2
120k
33pF
8.06k
220pF
0.22µF
20k
470pF
4.02k
3.3µH
1µF
62.0k
DIV
SS2
ILIM2
VC2
0.1µF
TJ
CLKOUT
GND
RT/SYNC
ILIM1
VC1
SS1
CMPO2
CMPI2
FB2
VOUT2
IND2
SW2
SHDN2
BST2
VIN2
LT3692
CMPO1
CMPI1
FB1
VOUT1
IND1
SW1
SHDN1
BST1
VIN1
12V, 3.3V, 2.5V, 1.2V Quad Output with External Synchronization, Output Sequencing and Tracking Application
16.9k
4.7µH
30.1k
33pF
680pF
1µF
8.06k
100pF
100µF
VOUT4
2.5V 2A
750kHz
3692 TA07
49.9k
0.22µF
LT3692
TYPICAL APPLICATIONS
3692fa
LT3692
TYPICAL APPLICATIONS
5V, 3.3V, 2.5V, 1.8V Synchronized Quad Output
VOUT1
5V
2A
2.2µF
SVIN
PVIN2
PVIN
DDR
4.7µF
×2
RUN LT3612
MODE
RT
VIN2
VIN1
SHDN1
BST1
6.8µH
0.22µF
SW1
SW2
IND1
IND2
6.8µH
2.2µF
0.22µF
42.2k
8.06k
FB1
FB2
CMPI1
CMPI2
CMPO1
SS1
CMPO2
SS2
ILIM2
VC2
ILIM1
VC1
0.1µF
RT/SYNC
330pF
DIV
33pF
60.4k
24.9k
16k
FB
270k
ITH
TRACK/SS
SGND
PGND
SVIN
PVIN2
60.4k
TJ
0.1µF
RUN LT3612
100k
SS1
ILIM1
VC1
MODE
100°C
TEMP FLAG
RT
3.3µH
VOUT3
2.5V
100µF 2A
SW
PGOOD
47µF
FB1
CLKOUT
GND
PVIN
DDR
VOUT2
VOUT1
100pF
VOUT2
3.3V
100µF 2A
SHDN2
BST2
LT3692
47µF
3.3µH
SW
PGOOD
VIN1
7V TO 36V
FB
191k
ITH
TRACK/SS
SGND
PGND
SVIN
PVIN2
60.4k
CLOCKOUT
600kHz
2.2µF
28k
73.2k
R14
49.9k
PVIN
DDR
PGOOD
RUN LT3612
MODE
RT
SGND
0.1µF
3.3µH
VOUT4
1.8V
100µF 2A
SW
FB
121k
ITH
TRACK/SS
PGND
60.4k
3692 TA08
3692fa
For more information www.linear.com/3692
33
LT3692
Package Description
Please refer to http://www.linear.com/designtools/packaging/ for the most recent package drawings.
UH Package
32-Lead Plastic QFN (5mm × 5mm)
(Reference LTC DWG # 05-08-1693 Rev D)
0.70 ±0.05
5.50 ±0.05
4.10 ±0.05
3.50 REF
(4 SIDES)
3.45 ± 0.05
3.45 ± 0.05
PACKAGE OUTLINE
0.25 ± 0.05
0.50 BSC
RECOMMENDED SOLDER PAD LAYOUT
APPLY SOLDER MASK TO AREAS THAT ARE NOT SOLDERED
5.00 ± 0.10
(4 SIDES)
BOTTOM VIEW—EXPOSED PAD
0.75 ± 0.05
R = 0.05
TYP
0.00 – 0.05
R = 0.115
TYP
PIN 1 NOTCH R = 0.30 TYP
OR 0.35 × 45° CHAMFER
31 32
0.40 ± 0.10
PIN 1
TOP MARK
(NOTE 6)
1
2
3.50 REF
(4-SIDES)
3.45 ± 0.10
3.45 ± 0.10
(UH32) QFN 0406 REV D
0.200 REF
NOTE:
1. DRAWING PROPOSED TO BE A JEDEC PACKAGE OUTLINE
M0-220 VARIATION WHHD-(X) (TO BE APPROVED)
2. DRAWING NOT TO SCALE
3. ALL DIMENSIONS ARE IN MILLIMETERS
4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE
MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.20mm ON ANY SIDE
5. EXPOSED PAD SHALL BE SOLDER PLATED
6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION
ON THE TOP AND BOTTOM OF PACKAGE
34
0.25 ± 0.05
0.50 BSC
3692fa
For more information www.linear.com/3692
LT3692
Revision History
REV
DATE
DESCRIPTION
A
3/13
Clarified Shutdown Quiescent Current vs Temperature Graph
PAGE NUMBER
5
Clarified SHDN1/2 Pin Function description
9
Clarified SHDN pin operation in Applications Information operation in general description
10
Clarified SHDN pin operation description in Applications Information
20
3692fa
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection
its circuits
as described
herein will not infringe on existing patent rights.
Forofmore
information
www.linear.com/3692
35
LT3692
Typical Application
1.2MHz, 12V to 3.3V at 3A, 5V at 3A
VIN
12V
4.7µF
VIN1
SHDN1
BST1
2.2µH
0.22µF
4.7µF
VIN2
SHDN2
BST2
SW1
SW2
IND1
IND2
2.2µH
0.22µF
LT3692
VOUT1
3.3V 3A
1.2MHz
100pF
47µF
24.9k
100k
8.06k
PG1
FB1
CMPI2
CMPO1
CMPO2
120k
24.9k
ILIM2
VC2
RT/SYNC
DIV
33pF
8.06k
22µF
100k
PG
SS2
ILIM1
VC1
330pF
100pF
42.2k
FB2
CMPI1
SS1
0.1µF
VOUT2
5V 3A
1.2MHz
VOUT2
VOUT1
CLKOUT
GND
42.2k
TJ
CLOCKOUT
1.2MHz
0.1µF
330pF
0.1µF
33pF
49.9k
30.1k
120k
3692 TA09
Related Parts
PART
DESCRIPTION
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LT3508
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LT3680
36V, 3A, 2.4MHz High Efficiency Micropower Step-Down DC/DC Converter
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LT3693
36V, 3A, 2.4MHz High Efficiency Step-Down DC/DC Converter
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36V with Transient Protection to 60V, 2A (IOUT), 2.4MHz, High Efficiency
Step-Down DC/DC Converter with Burst Mode® Operation
VIN: 3.6V to 38V, Transients to 60V, VOUT(MIN) = 0.78V,
IQ = 70µA, ISD < 1µA, (3mm × 3mm) DFN-10, MSOP-10E
LT3980
58V with Transient Protection to 80V, 2A (IOUT), 2.4MHz, High Efficiency
Step-Down DC/DC Converter with Burst Mode Operation
VIN: 3.6V to 58V, Transients to 80V, VOUT(MIN) = 0.79V,
IQ = 75µA, ISD < 1µA, (3mm × 4mm) DFN-16, MSOP-16E
LT3971
38V, 1.2A (IOUT), 2MHz, High Efficiency Step-Down DC/DC Converter with
Only 2.8µA of Quiescent Current
VIN: 4.2V to 38V, VOUT(MIN) = 1.2V, IQ = 2.8µA, ISD < 1µA,
(3mm × 3mm) DFN-10, MSOP-10E
LT3991
55V, 1.2A (IOUT), 2MHz, High Efficiency Step-Down DC/DC Converter with
Only 2.8µA of Quiescent Current
VIN: 4.2V to 55V, VOUT(MIN) = 1.2V, IQ = 2.8µA, ISD < 1µA,
(3mm × 3mm) DFN-10, MSOP-10E
36 Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
For more information www.linear.com/3692
(408) 432-1900 ● FAX: (408) 434-0507
●
www.linear.com/3692
3692fa
LT 0313 REV A • PRINTED IN USA
 LINEAR TECHNOLOGY CORPORATION 2011