AN-573: OP07 Is Still Evolving (Rev. B) PDF

AN-573
APPLICATION NOTE
One Technology Way • P.O. Box 9106 • Norwood, MA 02062-9106, U.S.A. • Tel: 781.329.4700 • Fax: 781.461.3113 • www.analog.com
OP07 Is Still Evolving
by Reza Moghimi
to +2.7 V with single rail operation. The OP777/OP727/OP747
data sheet characterizes the parts with rails of +5 V and ±15 V.
The OP7x7 family’s true single-supply capability enables designers
to operate down to the negative supply or ground in both singleand dual-supply applications.
INTRODUCTION
The OP07 has been tinkered with over the years, and versions of it
are still available in plastic packages.
This application note highlights some of the major features that
the OP7x7 brings into new designs. A number of applications
using these features are presented.
Figure 1 shows that the gain of the instrumentation amplifier
(made up of U3 and U4) is set for 100. The AD589 establishes
1.235 V, while the U1 amplifier servos the bridge and maintains the
voltage across the parallel combination of 2.55 MΩ and 6.19 kΩ
to generate a 200 μA current source. This current splits evenly,
flows into both halves of the bridge, eventually through RTD, and
establishes an output voltage based upon its value.
SINGLE-SUPPLY OPERATION
One of the biggest problems with the part in today’s environment is
that the OP07 requires dual supplies. This family of amplifiers from
Analog Devices, Inc., addresses this problem while still giving a
close replica of the original specifications. The OP777 single,
OP727 dual, and OP747 quad operational amplifiers allow supplies
from ±15 V down to ±1.35 V with split rails and from +30 V down
5V
R4
26.7kΩ
AD589
2
1
U1
N D1
V–
1/4
OP747
V2
GAIN = 100 (V2 – V1)
R7
100Ω
R2
200Ω
RTD
100Ω
V1
R5
26.7kΩ
R8
2.55MΩ
R9
6.19kΩ
R12
1MΩ
R4
10.1kΩ
U3
VOUT
U4
1/4
OP747
R14
10.1kΩ
1/4
OP747
R15
1MΩ
Figure 1. Low Power Single-Supply RTD Amplifier
Rev. B | Page 1 of 8
02380-001
R3
37.4kΩ
V+
3
AN-573
APPLICATION NOTE
TABLE OF CONTENTS
Introduction ...................................................................................... 1 Rail-to-Rail Output ...........................................................................6 Single-Supply Operation.................................................................. 1 Negative Rail Input ............................................................................6 Revision History ............................................................................... 2 3 V Over the Input ............................................................................7 Much Lower Supply Currents ......................................................... 4 Design Reminders for Achieving High Performances .................7 Absence of Clamping Diodes at the Inputs ................................... 5 REVISION HISTORY
3/10—Rev. A to Rev. B
Changes to Format ............................................................. Universal
Changes to Introduction Section and Single-Supply Operation
Section ................................................................................................ 1
Changes to Figure 2 and Figure 4 ................................................... 3
Changes to Much Lower Supply Currents Section ...................... 4
Changes to Absence of Clamping Diodes at the Inputs Section
and Figure 10 ..................................................................................... 5
Changes to Figure 14 and Figure 16 ............................................... 6
Changes to 3 V Over the Input Section ......................................... 7
6/03—Rev. 0 to Rev. A
11/02—Revision 0: Initial Version
Rev. B | Page 2 of 8
APPLICATION NOTE
AN-573
VIN 0V TO 3V
R23
10kΩ
R24
100kΩ
R21
182kΩ
R20
1.21MΩ
V+
3
2
VIN
R26
100Ω
OP777
1
2
R22
1kΩ
12V TO
30V
T1
3
R25
220Ω
V–
TRIM
Q1
2N1711
VOUT
5
TWIST
PAIR
R29
100Ω
4mA TO 20mA
6
GND
C2
220pF
2
1
HP5082-2800
REF-02A/D
4
02380-002
D2
R27
100kΩ
R28
100kΩ
Figure 2. Self-Powered 4 mA to 20 mA Current Loop Transmitter
+VS
2
VS
+VS
REF192
OUTPUT
GND
4
6
3
V+
1
C7
0.1µF
2
V–
1/4
OP747
R1
R1 (1 + δ)
A = 300
AR1 × VREF
VOUT =
δ + 2.5V
2R2
VOUT
R91
10.1kΩ
R1 (1 + δ)
1/4
OP747
+VS
R1
1/4
OP747
2
R83
1MΩ
VS
REF192
R2
OUTPUT
R84
6 1MΩ
R82
10.1kΩ
02380-003
GND
R85
10kΩ
4
Figure 3. Single-Supply Linear Response Bridge
2.67kΩ
As shown in Figure 2, the circuit floats up from the single-supply
(12 V to 30 V) return. It consumes only 1.5 mA, leaving 2.5 mA
available to the user for powering other signal conditioning
circuitry.
The OP7x7 is very useful in many bridge applications. Figure 3
shows a single-supply bridge circuit whose output is linearly
proportional to the fractional deviation (δ) of the bridge.
+3V
V+
3
2
V–
100kΩ
1µF
1
A1
1/4
OP747
1µF
2.67kΩ
2.67kΩ
2µF
1.33kΩ
2.67kΩ
A2
1kΩ
1kΩ
ΔR
R
VOUT
1/4
OP747
+3V
1MΩ
To process ac signals in single-supply systems, it is often best
to use a false-ground biasing scheme. In Figure 4, this is done by
Amplifier A3. The user should replace the 2.67 kΩ Twin-T
section with a 3.16 kΩ resistor to reject 50 Hz. Sensitivity is due to
the relative matching of the capacitors and resistors in the Twin-T
section. Use Mylar (5%) and 1% resistors for satisfactory results.
1µF
1MΩ
1/4
OP747
499Ω
A3
0.01µF
100kΩ
1µF
02380-004
Note that δ =
VIN
Figure 4. 3 V Single-Supply 50 Hz/60 Hz Active Notch Filter with False Ground
Rev. B | Page 3 of 8
AN-573
APPLICATION NOTE
+15V
MUCH LOWER SUPPLY CURRENTS
The OP07 has a quiescent current that is higher than desired in
today’s portable applications. The quiescent current of the OP777
in-amp is less than 350 μA, while the OP07 requires 4 mA for
±15 V operation. In terms of power consumption, the OP777
allows the part to be designed into many portable applications.
5V
2
R12
1MΩ
U4
U3
1
1/2
V– OP727
10V
1 F
AD680AD
2
VOUT
VIN
TEMP
1/2
OP727
V2
V+
IN4002
R48
6 10kΩ
GND
VOUT
R49
10kΩ
R14
10.1kΩ
1/4
OP747
3
V+
7.5V
1
1/4
V– OP747
10kΩ
1/4
OP747
2µF
4
R13
10.1kΩ
R15
1MΩ
10kΩ
+VS
2
3
R50
10kΩ
02380-005
3
V1
22kΩ
5V
10kΩ
10kΩ
C8
1µF
Figure 5. Single-Supply Micropower In-Amp
2.5V
CMRR = 20 × log(100/(1 − (R15 × R14)/(R13 × R12))
It is common to specify the accuracy of the resistor network in
terms of resistor-to-resistor percentage mismatch. The CMRR
equation can be rewritten to reflect this.
CMRR = 20 × log(10000/% mismatch)
The key to high CMRR is a network of resistors that is well matched
from the perspective of both resistive ratio and relative drift. The
absolute value of the resistors and their absolute drift are of no
consequence; matching is the key. CMRR is 100 dB with a 0.1%
mismatched resistor network. To maximize CMRR, one of the
resistors, such as R12, should be trimmed. Tighter matching of
two op amps in one package (OP727) offers a significant boost
in performance over the triple op amp configuration.
For this circuit, VO = 100(V2 − V1) for 0.02 mV ≤ (V1 − V2) ≤
290 mV, 2 mV ≤ VOUT ≤ 29 V.
02380-006
Figure 6. Multiple Output Tracking Voltage Reference
Figure 7 shows an example of a 5 V single-supply current monitor
that can be incorporated into the design of a voltage regulator
with foldback current limiting or a high current power supply
with crowbar protection. The design capitalizes on the commonmode range of the OP777 that extends to ground. Current is
monitored in the power supply return where a 0.1 Ω shunt resistor,
RSENSE, creates a very small voltage drop. The voltage at the inverting
terminal becomes equal to the voltage at the noninverting terminal
through the feedback of Q1, which is a 2N2222A or equivalent
NPN transistor. This makes the voltage drop across R3 equal to
the voltage drop across RSENSE. Therefore, the current through Q1
becomes directly proportional to the current through RSENSE, and
the output voltage is given by
VOUT = 5 V − (R2/R3) × RSENSE × IL)
The voltage drop across R2 increases when IL increases; therefore,
VOUT decreases when a higher supply current is sensed. For the
element values shown, VOUT is 2.5 V for a return current of 1 A.
RETURN TO
GROUND
5V
Due to its great dc accuracy and specification, the OP747 can be
used to create a multiple output tracking voltage reference from
a single source, as shown in Figure 6.
R2
2.49kΩ
VOUT
Q1
2N2222A/ZTX
RSENSE
0.1Ω
R3
100Ω
3 V+
2
1
U1
V–
OP777
Figure 7. Low-Side Current Sensing Circuit
Rev. B | Page 4 of 8
02380-007
The OP727 can be used to build an in-amp with two op amps.
A single-supply in-amp using one OP727 amplifier is shown in
Figure 5. For true difference, R14/R12 = R15/R13. The formula
for the CMRR of the circuit at dc is
1/4
OP747
APPLICATION NOTE
AN-573
+15V
Figure 8 shows the OP777 configured as a simple summing
amplifier. The output is the sum of V1 and V2.
VIN
TRIM
+15V
OP777
1
2
V1
16
15
VOUT
14
V–
10kΩ
13
12
11
02380-008
–15V
V2
10kΩ
10
9
Figure 8. Summing Amplifier
8
7
ABSENCE OF CLAMPING DIODES AT THE INPUTS
6
19
The large differential voltage capability allows for operation of
the parts in both rectifier circuits and precision comparator
applications. The need for external clamping diodes (on-board
in the OP07) is eliminated; such diodes are often needed on
precision op amps and are the bane of many comparator designs.
20
4
22
18
DB0
VOUT
DB1
DB2
GND
DB3
4
DB4
2
IOUTA
24
IOUTB
3
RFBA
23
RFBB
DB5
DB6
DB7
DB8
DB9
+5V
DB10
DB11
DAC8222
LDAC
WR
VREFA
V+
C10
0.01µF
OP777
V–
DACA
DACB
AGND DGND
TTL OUT
1N4148
2N2222A/ZTX
VIN
1N4148
R67
10kΩ
5
10kΩ
1/2
OP727
–15V
VOUT
+15V
VOUT = ±(VS) @ 1kHz
2kΩ
V+
VIN
02380-009
–VS
R3
68kΩ
1kΩ
An OP777 is used to build a precision threshold detector. In this
circuit, when VIN < VTH, the amplifier swings negative, reverse
biasing the diode. If RL = infinite, VOUT = VTH. When VIN ≥ VTH,
the feedback occurs and VOUT = VTH + (VIN − VTH)(1 + RF/RS).
C is selected to make the loop respond in a smoother fashion.
+VS
1
1
Figure 10. Programmable High Resolution Window Comparator
R61
100kΩ
2
1/2
OP727
V–
f = 1/(2R3 × C10 × ln ((R61 + R60)/R61)
3
2
10kΩ
The simple oscillator shown in Figure 9 creates a square wave
output of ±VS at 1 kHz for the values shown. Other oscillation
frequencies can be derived by using
R60
100kΩ
V+
R68
10kΩ
VREFB
1
3
02380-010
10kΩ
3
ADR01
VDD
VTH
Figure 9. Free-Running Square Wave Amplifier
RS
1kΩ
1N4148
R
VOUT = VTH + (VIN – VTH) 1+ F
RS
OP777
V–
–15V
The programmable window comparator is capable of 12-bit
accuracy. DAC8222 is used in the voltage for setting the upper
and lower thresholds.
RF
100kΩ
C
Figure 11. Precision Threshold Detector/Amplifier
Rev. B | Page 5 of 8
02380-011
3.3kΩ
21
17
V+
AN-573
APPLICATION NOTE
For VIN > 0 V and <2 kHz, there is no current flow through the
feedback resistors, and the output voltage tracks the input. For
VIN < 0 V, the output of the first amplifier goes to 0 V (that is, −VS),
which configures the second amplifier in inverting follower mode.
The output is then a full-wave rectified version of the input signal.
As can be seen from the schematic shown in Figure 12, a half-wave
rectified version of the signal is also available at the output of
the first amplifier.
A single-supply current source is shown in Figure 14. Large resistors
are used to maintain micropower operation. Output current can be
adjusted by changing the R10 resistor. Compliance voltage is
|VL| ≤ |VSAT| − |VS|; IOUT = R2/(R8 × R10) × VS;
IOUT = 1 mA to 11 mA; R2 = R10 + R7
2.7V TO 30V
VOUT (FULL-WAVE
RECTIFIED)
R6
100k
100kΩ
C1
10pF
Figure 14. Single-Supply Current Source
Figure 12. Single-Supply Half-Wave and Full-Wave Rectifier
RAIL-TO-RAIL OUTPUT
With light loads, the output can swing to within 1 mV of both
supply rails, and the parts are stable in a voltage follower
configuration. Short-circuit protection on the output protects
the devices up to 30 mA with split ±15 V supplies (10 mA with
a single 5 V supply).
02380-014
100kΩ
RLOAD
OP777
V–
1/2
OP727
02380-012
V–
1 IOUT = 1mA TO 11mA
U3
2
R9
100kΩ
2
1/2
OP727
V+
When in single-supply applications, driving motors or actuators
in two directions is often accomplished using an H-bridge (see
Figure 15). This driver is capable of driving loads from 0 V to 5 V
in both directions. To drive inductive loads in both directions, be
sure to add diode clamps to protect the bridge from inductive
kickback.
5V
NEGATIVE RAIL INPUT
5V
The amplifiers respond to signals as low as 1 mV above ground
in a single-supply arrangement. The true single-supply capability
of the OP7x7 family enables designers to operate down to the
negative supply or ground in both single- and dual-supply
applications.
3
1.67V
0V < VIN < 2.5V
2
R39
5kΩ
U3
Q4
2N2222A/ZTX
1
1/2
OP727
V–
R38
10kΩ
The high gain and low TCVOS of the OP727 ensure accurate
operation with microvolt input signals (see Figure 13). In this
circuit, the input always appears as a common-mode signal to
the op amps. The CMRR of the OP727 exceeds 120 dB, yielding
an error of less than 2 ppm.
Q3
2N2222A/ZTX
V+
VOUT
Q5
2N2907
Q6
2N2907
U3
1/2
OP727
R40
10kΩ
R37
10kΩ
02380-015
V+
1
3
C2
10pF
5V
3
R10
2.7kΩ
R8
100kΩ
VOUT (HALF-WAVE RECTIFIED)
2V p-p
R7
97.3kΩ
+15V
1
2kΩ
Figure 15. H Bridge
1/2
OP727
The current source shown in Figure 16 supplies both positive
and negative current into grounded load. Note that
0V < VOUT < 10V
2
V–
1/2
OP727
–15V
30pF
D3
1N4148
1kΩ
1kΩ
Figure 13. Precision Absolute Value Amplifier
ZOUT = R2B × ((R2A/R1) + 1)/((R2B + R2A)/R1) − R2/R5
and, for ZOUT to be infinite, (R2A + R2B)/R1 = R2/R5.
R2A
1.8kΩ
VCC
R5
2kΩ
VIN
3
2
7
R2B
200Ω
V+
6 IOUT = VIN/200Ω
U1
V–
4
R1
2kΩ
OP777
VEE
RLOAD
R2 = R2A+R2B
R2
2kΩ
Figure 16. Bilateral Current Source
Rev. B | Page 6 of 8
02380-016
D3
1N4148
V+
02380-013
VIN
3
APPLICATION NOTE
AN-573
3 V OVER THE INPUT
The PNP input stages are protected with 500 Ω current-limiting
resistors, allowing input voltages up to 3 V higher than either rail
without causing damage or phase reversals. The phase reversal
protection operates for conditions where either one or both inputs
are forced beyond their input common-mode voltage range.
VS = ±15V
AV = 1
INPUT
For designs operating at ±15 V, the OP777 is a low noise precision
amplifier available in a tiny, 8-lead MSOP package. The OP777 is
also available in an 8-lead SOIC surface-mount package.
This family is extremely useful in instrumentation, for remote
sensor acquisition, and in precision filters. The high voltage range
allows the use of the parts for single-supply current sourcing
and large range instrumentation amplifiers. Both single-supply
and dual-supply linear response bridges can also be built. The
parts are ideal for use in low-side current monitors in power
supply control circuits because the common-mode range extends
to ground in the single-supply configuration.
02380-017
VOLTAGE (5V/DIV)
OUTPUT
TIME (400µs/DIV)
Figure 17. No Phase Inversion
DESIGN REMINDERS FOR ACHIEVING HIGH
PERFORMANCES
30V
As with any application, a good ground plane is essential to
achieve optimum performance. This can significantly reduce
the undesirable effects of ground loops and I × R losses by
providing a low impedance reference point. Best results are
obtained with a multilayer board design with one layer
assigned to the ground plane.
02380-018
OP777/
OP727/
OP747
V p-p = 32V
Figure 18. Unity-Gain Follower
VS = ±15V
AV = 1
To minimize high frequency interference and prevent low
frequency ground loops, shield grounding techniques are
required when sensors are used. The cable shielding system
should include the cable end connectors.
VOLTAGE (5V/DIV)
VIN
02380-019
VOUT
TIME (400µs/DIV)
The gain characteristics, of course, are rather different at differing
rails. The inputs have a maximum, single temperature offset of
100 μV with an input offset current of 2 nA and input bias
current of only 10 nA maximum. With a single 5 V rail, the
CMRR is typically 110 dB, and the large signal voltage gain is
typically 500 V/mV with a 10 kΩ load. With ±15 V rails, the
CMRR increases, not surprisingly, by 10 dB to 120 dB, and the
large signal voltage gain increases to 2500 V/mV.
Figure 19. Input Voltage Can Exceed the Supply Voltage Without Damage
The dynamic performance and noise characteristics of the devices
are similar whether they are being used with single or dual supplies.
The slew rate with a 2 kΩ load is 200 mV/μs, and the gain bandwidth product is 700 kHz. Peak-to-peak voltage noise from 0.1 Hz
to 10 Hz is 0.4 μV, and the voltage noise density at 1 kHz is
15 nV√Hz.
Switching power supplies with high output noise is normally used
in many systems. This noise generally extends over a broad band
of frequencies and occurs as both conducted and radiated noise,
and unwanted electric and magnetic fields. The voltage output
noise of switching supplies is short-duration voltage transients
or spikes that contain frequency components easily extending
to 100 MHz or more. Although specifying switching supplies in
terms of rms noise is a common vendor practice, users should
also specify the peak (or peak-to-peak) amplitudes of the switching
spikes with the output loading of the individual system. Capacitors,
inductors, ferrite beads, and resistors are used in filters for noise
reduction. Linear post regulation can also be done and separates
the power supply circuit from sensitive analog circuits. Analog
Devices manufactures many anyCAP® low dropout linear
regulators. Examples of these devices are the ADP3300 to
ADP3310 and ADP3335 to ADP3339 for supply voltages less
than 12 V.
Rev. B | Page 7 of 8
AN-573
APPLICATION NOTE
Capacitors are probably the single most important filter component
for switchers. There are generally three classes of capacitors useful
in filters in the 10 kHz to 100 MHz frequency range suitable for
switchers. Capacitors are broadly distinguished by their generic
dielectric types: electrolytic, film, and ceramic. Background and
tutorial information on capacitors can be found in the Walter G.
Jung, Richard Marsh, Picking Capacitors, Part 1 and Part 2, AUDIO
(February, March 1980) article and many vendor catalogs.
Chip capacitors should be used for supply bypassing, with one
end of the capacitor connected to the ground plane and the other
end connected within ⅛ inch of each power pin. An additional
large tantalum electrolytic capacitor (4.7 μF to 10 μF) should be
connected in parallel. This capacitor does not need to be placed
as close to the supply pins because it provides current for fast
large signal changes at the output of the device.
Use short and wide PCB tracks to decrease voltage drops and
minimize inductance. Make track widths at least 200 mils for
every inch of track length for lowest DCR and use 1 ounce or
2 ounce copper PCB traces to further reduce IR drops and
inductance.
Be careful not to exceed the maximum junction temperature or
the maximum power dissipation rating of an amplifier. When a
capacitive load connects to the output of the amplifier, include
the power dissipation caused by the rms ac current delivered to
the load in the calculation.
Use short leads or leadless components to minimize lead
inductance. This minimizes the tendency to add excessive ESL
and/or ESR. Surface-mount packages are preferred. Use a large
area ground plane for minimum impedance. Note how components
behave over frequency, current, and temperature variations.
Make use of vendor component models for the simulation of prototype designs, and make sure that lab measurements correspond
reasonably with the simulation. SPICE modeling is a powerful
tool for predicting the performance of analog circuits. Analog
Devices provides macro models for most of its ICs. SPICE models
can be downloaded on the OP777 product page.
Because models omit many real-life effects and no model can
simulate all of the parasitic effects of discrete components and
PCB traces, build/prove prototypes before they go into production.
To ensure successful prototyping, always use a ground plane for
precision or high frequency circuits. Minimize parasitic resistance,
capacitance, and inductance. If sockets are required, use pin sockets
(cage jacks). Pay equal attention to signal routing, component
placement, grounding, and decoupling in both the prototype and
the final design. Popular prototyping techniques include Freehand
dead-bug using point-to-point wiring and solder-mount, milled
PCB from CAD layout, multilayer boards that are double-sided
with additional point-to-point wiring.
©2002–2010 Analog Devices, Inc. All rights reserved. Trademarks and
registered trademarks are the property of their respective owners.
AN02380-0-3/10(B)
Rev. B | Page 8 of 8