AN-573 APPLICATION NOTE One Technology Way • P.O. Box 9106 • Norwood, MA 02062-9106, U.S.A. • Tel: 781.329.4700 • Fax: 781.461.3113 • www.analog.com OP07 Is Still Evolving by Reza Moghimi to +2.7 V with single rail operation. The OP777/OP727/OP747 data sheet characterizes the parts with rails of +5 V and ±15 V. The OP7x7 family’s true single-supply capability enables designers to operate down to the negative supply or ground in both singleand dual-supply applications. INTRODUCTION The OP07 has been tinkered with over the years, and versions of it are still available in plastic packages. This application note highlights some of the major features that the OP7x7 brings into new designs. A number of applications using these features are presented. Figure 1 shows that the gain of the instrumentation amplifier (made up of U3 and U4) is set for 100. The AD589 establishes 1.235 V, while the U1 amplifier servos the bridge and maintains the voltage across the parallel combination of 2.55 MΩ and 6.19 kΩ to generate a 200 μA current source. This current splits evenly, flows into both halves of the bridge, eventually through RTD, and establishes an output voltage based upon its value. SINGLE-SUPPLY OPERATION One of the biggest problems with the part in today’s environment is that the OP07 requires dual supplies. This family of amplifiers from Analog Devices, Inc., addresses this problem while still giving a close replica of the original specifications. The OP777 single, OP727 dual, and OP747 quad operational amplifiers allow supplies from ±15 V down to ±1.35 V with split rails and from +30 V down 5V R4 26.7kΩ AD589 2 1 U1 N D1 V– 1/4 OP747 V2 GAIN = 100 (V2 – V1) R7 100Ω R2 200Ω RTD 100Ω V1 R5 26.7kΩ R8 2.55MΩ R9 6.19kΩ R12 1MΩ R4 10.1kΩ U3 VOUT U4 1/4 OP747 R14 10.1kΩ 1/4 OP747 R15 1MΩ Figure 1. Low Power Single-Supply RTD Amplifier Rev. B | Page 1 of 8 02380-001 R3 37.4kΩ V+ 3 AN-573 APPLICATION NOTE TABLE OF CONTENTS Introduction ...................................................................................... 1 Rail-to-Rail Output ...........................................................................6 Single-Supply Operation.................................................................. 1 Negative Rail Input ............................................................................6 Revision History ............................................................................... 2 3 V Over the Input ............................................................................7 Much Lower Supply Currents ......................................................... 4 Design Reminders for Achieving High Performances .................7 Absence of Clamping Diodes at the Inputs ................................... 5 REVISION HISTORY 3/10—Rev. A to Rev. B Changes to Format ............................................................. Universal Changes to Introduction Section and Single-Supply Operation Section ................................................................................................ 1 Changes to Figure 2 and Figure 4 ................................................... 3 Changes to Much Lower Supply Currents Section ...................... 4 Changes to Absence of Clamping Diodes at the Inputs Section and Figure 10 ..................................................................................... 5 Changes to Figure 14 and Figure 16 ............................................... 6 Changes to 3 V Over the Input Section ......................................... 7 6/03—Rev. 0 to Rev. A 11/02—Revision 0: Initial Version Rev. B | Page 2 of 8 APPLICATION NOTE AN-573 VIN 0V TO 3V R23 10kΩ R24 100kΩ R21 182kΩ R20 1.21MΩ V+ 3 2 VIN R26 100Ω OP777 1 2 R22 1kΩ 12V TO 30V T1 3 R25 220Ω V– TRIM Q1 2N1711 VOUT 5 TWIST PAIR R29 100Ω 4mA TO 20mA 6 GND C2 220pF 2 1 HP5082-2800 REF-02A/D 4 02380-002 D2 R27 100kΩ R28 100kΩ Figure 2. Self-Powered 4 mA to 20 mA Current Loop Transmitter +VS 2 VS +VS REF192 OUTPUT GND 4 6 3 V+ 1 C7 0.1µF 2 V– 1/4 OP747 R1 R1 (1 + δ) A = 300 AR1 × VREF VOUT = δ + 2.5V 2R2 VOUT R91 10.1kΩ R1 (1 + δ) 1/4 OP747 +VS R1 1/4 OP747 2 R83 1MΩ VS REF192 R2 OUTPUT R84 6 1MΩ R82 10.1kΩ 02380-003 GND R85 10kΩ 4 Figure 3. Single-Supply Linear Response Bridge 2.67kΩ As shown in Figure 2, the circuit floats up from the single-supply (12 V to 30 V) return. It consumes only 1.5 mA, leaving 2.5 mA available to the user for powering other signal conditioning circuitry. The OP7x7 is very useful in many bridge applications. Figure 3 shows a single-supply bridge circuit whose output is linearly proportional to the fractional deviation (δ) of the bridge. +3V V+ 3 2 V– 100kΩ 1µF 1 A1 1/4 OP747 1µF 2.67kΩ 2.67kΩ 2µF 1.33kΩ 2.67kΩ A2 1kΩ 1kΩ ΔR R VOUT 1/4 OP747 +3V 1MΩ To process ac signals in single-supply systems, it is often best to use a false-ground biasing scheme. In Figure 4, this is done by Amplifier A3. The user should replace the 2.67 kΩ Twin-T section with a 3.16 kΩ resistor to reject 50 Hz. Sensitivity is due to the relative matching of the capacitors and resistors in the Twin-T section. Use Mylar (5%) and 1% resistors for satisfactory results. 1µF 1MΩ 1/4 OP747 499Ω A3 0.01µF 100kΩ 1µF 02380-004 Note that δ = VIN Figure 4. 3 V Single-Supply 50 Hz/60 Hz Active Notch Filter with False Ground Rev. B | Page 3 of 8 AN-573 APPLICATION NOTE +15V MUCH LOWER SUPPLY CURRENTS The OP07 has a quiescent current that is higher than desired in today’s portable applications. The quiescent current of the OP777 in-amp is less than 350 μA, while the OP07 requires 4 mA for ±15 V operation. In terms of power consumption, the OP777 allows the part to be designed into many portable applications. 5V 2 R12 1MΩ U4 U3 1 1/2 V– OP727 10V 1 F AD680AD 2 VOUT VIN TEMP 1/2 OP727 V2 V+ IN4002 R48 6 10kΩ GND VOUT R49 10kΩ R14 10.1kΩ 1/4 OP747 3 V+ 7.5V 1 1/4 V– OP747 10kΩ 1/4 OP747 2µF 4 R13 10.1kΩ R15 1MΩ 10kΩ +VS 2 3 R50 10kΩ 02380-005 3 V1 22kΩ 5V 10kΩ 10kΩ C8 1µF Figure 5. Single-Supply Micropower In-Amp 2.5V CMRR = 20 × log(100/(1 − (R15 × R14)/(R13 × R12)) It is common to specify the accuracy of the resistor network in terms of resistor-to-resistor percentage mismatch. The CMRR equation can be rewritten to reflect this. CMRR = 20 × log(10000/% mismatch) The key to high CMRR is a network of resistors that is well matched from the perspective of both resistive ratio and relative drift. The absolute value of the resistors and their absolute drift are of no consequence; matching is the key. CMRR is 100 dB with a 0.1% mismatched resistor network. To maximize CMRR, one of the resistors, such as R12, should be trimmed. Tighter matching of two op amps in one package (OP727) offers a significant boost in performance over the triple op amp configuration. For this circuit, VO = 100(V2 − V1) for 0.02 mV ≤ (V1 − V2) ≤ 290 mV, 2 mV ≤ VOUT ≤ 29 V. 02380-006 Figure 6. Multiple Output Tracking Voltage Reference Figure 7 shows an example of a 5 V single-supply current monitor that can be incorporated into the design of a voltage regulator with foldback current limiting or a high current power supply with crowbar protection. The design capitalizes on the commonmode range of the OP777 that extends to ground. Current is monitored in the power supply return where a 0.1 Ω shunt resistor, RSENSE, creates a very small voltage drop. The voltage at the inverting terminal becomes equal to the voltage at the noninverting terminal through the feedback of Q1, which is a 2N2222A or equivalent NPN transistor. This makes the voltage drop across R3 equal to the voltage drop across RSENSE. Therefore, the current through Q1 becomes directly proportional to the current through RSENSE, and the output voltage is given by VOUT = 5 V − (R2/R3) × RSENSE × IL) The voltage drop across R2 increases when IL increases; therefore, VOUT decreases when a higher supply current is sensed. For the element values shown, VOUT is 2.5 V for a return current of 1 A. RETURN TO GROUND 5V Due to its great dc accuracy and specification, the OP747 can be used to create a multiple output tracking voltage reference from a single source, as shown in Figure 6. R2 2.49kΩ VOUT Q1 2N2222A/ZTX RSENSE 0.1Ω R3 100Ω 3 V+ 2 1 U1 V– OP777 Figure 7. Low-Side Current Sensing Circuit Rev. B | Page 4 of 8 02380-007 The OP727 can be used to build an in-amp with two op amps. A single-supply in-amp using one OP727 amplifier is shown in Figure 5. For true difference, R14/R12 = R15/R13. The formula for the CMRR of the circuit at dc is 1/4 OP747 APPLICATION NOTE AN-573 +15V Figure 8 shows the OP777 configured as a simple summing amplifier. The output is the sum of V1 and V2. VIN TRIM +15V OP777 1 2 V1 16 15 VOUT 14 V– 10kΩ 13 12 11 02380-008 –15V V2 10kΩ 10 9 Figure 8. Summing Amplifier 8 7 ABSENCE OF CLAMPING DIODES AT THE INPUTS 6 19 The large differential voltage capability allows for operation of the parts in both rectifier circuits and precision comparator applications. The need for external clamping diodes (on-board in the OP07) is eliminated; such diodes are often needed on precision op amps and are the bane of many comparator designs. 20 4 22 18 DB0 VOUT DB1 DB2 GND DB3 4 DB4 2 IOUTA 24 IOUTB 3 RFBA 23 RFBB DB5 DB6 DB7 DB8 DB9 +5V DB10 DB11 DAC8222 LDAC WR VREFA V+ C10 0.01µF OP777 V– DACA DACB AGND DGND TTL OUT 1N4148 2N2222A/ZTX VIN 1N4148 R67 10kΩ 5 10kΩ 1/2 OP727 –15V VOUT +15V VOUT = ±(VS) @ 1kHz 2kΩ V+ VIN 02380-009 –VS R3 68kΩ 1kΩ An OP777 is used to build a precision threshold detector. In this circuit, when VIN < VTH, the amplifier swings negative, reverse biasing the diode. If RL = infinite, VOUT = VTH. When VIN ≥ VTH, the feedback occurs and VOUT = VTH + (VIN − VTH)(1 + RF/RS). C is selected to make the loop respond in a smoother fashion. +VS 1 1 Figure 10. Programmable High Resolution Window Comparator R61 100kΩ 2 1/2 OP727 V– f = 1/(2R3 × C10 × ln ((R61 + R60)/R61) 3 2 10kΩ The simple oscillator shown in Figure 9 creates a square wave output of ±VS at 1 kHz for the values shown. Other oscillation frequencies can be derived by using R60 100kΩ V+ R68 10kΩ VREFB 1 3 02380-010 10kΩ 3 ADR01 VDD VTH Figure 9. Free-Running Square Wave Amplifier RS 1kΩ 1N4148 R VOUT = VTH + (VIN – VTH) 1+ F RS OP777 V– –15V The programmable window comparator is capable of 12-bit accuracy. DAC8222 is used in the voltage for setting the upper and lower thresholds. RF 100kΩ C Figure 11. Precision Threshold Detector/Amplifier Rev. B | Page 5 of 8 02380-011 3.3kΩ 21 17 V+ AN-573 APPLICATION NOTE For VIN > 0 V and <2 kHz, there is no current flow through the feedback resistors, and the output voltage tracks the input. For VIN < 0 V, the output of the first amplifier goes to 0 V (that is, −VS), which configures the second amplifier in inverting follower mode. The output is then a full-wave rectified version of the input signal. As can be seen from the schematic shown in Figure 12, a half-wave rectified version of the signal is also available at the output of the first amplifier. A single-supply current source is shown in Figure 14. Large resistors are used to maintain micropower operation. Output current can be adjusted by changing the R10 resistor. Compliance voltage is |VL| ≤ |VSAT| − |VS|; IOUT = R2/(R8 × R10) × VS; IOUT = 1 mA to 11 mA; R2 = R10 + R7 2.7V TO 30V VOUT (FULL-WAVE RECTIFIED) R6 100k 100kΩ C1 10pF Figure 14. Single-Supply Current Source Figure 12. Single-Supply Half-Wave and Full-Wave Rectifier RAIL-TO-RAIL OUTPUT With light loads, the output can swing to within 1 mV of both supply rails, and the parts are stable in a voltage follower configuration. Short-circuit protection on the output protects the devices up to 30 mA with split ±15 V supplies (10 mA with a single 5 V supply). 02380-014 100kΩ RLOAD OP777 V– 1/2 OP727 02380-012 V– 1 IOUT = 1mA TO 11mA U3 2 R9 100kΩ 2 1/2 OP727 V+ When in single-supply applications, driving motors or actuators in two directions is often accomplished using an H-bridge (see Figure 15). This driver is capable of driving loads from 0 V to 5 V in both directions. To drive inductive loads in both directions, be sure to add diode clamps to protect the bridge from inductive kickback. 5V NEGATIVE RAIL INPUT 5V The amplifiers respond to signals as low as 1 mV above ground in a single-supply arrangement. The true single-supply capability of the OP7x7 family enables designers to operate down to the negative supply or ground in both single- and dual-supply applications. 3 1.67V 0V < VIN < 2.5V 2 R39 5kΩ U3 Q4 2N2222A/ZTX 1 1/2 OP727 V– R38 10kΩ The high gain and low TCVOS of the OP727 ensure accurate operation with microvolt input signals (see Figure 13). In this circuit, the input always appears as a common-mode signal to the op amps. The CMRR of the OP727 exceeds 120 dB, yielding an error of less than 2 ppm. Q3 2N2222A/ZTX V+ VOUT Q5 2N2907 Q6 2N2907 U3 1/2 OP727 R40 10kΩ R37 10kΩ 02380-015 V+ 1 3 C2 10pF 5V 3 R10 2.7kΩ R8 100kΩ VOUT (HALF-WAVE RECTIFIED) 2V p-p R7 97.3kΩ +15V 1 2kΩ Figure 15. H Bridge 1/2 OP727 The current source shown in Figure 16 supplies both positive and negative current into grounded load. Note that 0V < VOUT < 10V 2 V– 1/2 OP727 –15V 30pF D3 1N4148 1kΩ 1kΩ Figure 13. Precision Absolute Value Amplifier ZOUT = R2B × ((R2A/R1) + 1)/((R2B + R2A)/R1) − R2/R5 and, for ZOUT to be infinite, (R2A + R2B)/R1 = R2/R5. R2A 1.8kΩ VCC R5 2kΩ VIN 3 2 7 R2B 200Ω V+ 6 IOUT = VIN/200Ω U1 V– 4 R1 2kΩ OP777 VEE RLOAD R2 = R2A+R2B R2 2kΩ Figure 16. Bilateral Current Source Rev. B | Page 6 of 8 02380-016 D3 1N4148 V+ 02380-013 VIN 3 APPLICATION NOTE AN-573 3 V OVER THE INPUT The PNP input stages are protected with 500 Ω current-limiting resistors, allowing input voltages up to 3 V higher than either rail without causing damage or phase reversals. The phase reversal protection operates for conditions where either one or both inputs are forced beyond their input common-mode voltage range. VS = ±15V AV = 1 INPUT For designs operating at ±15 V, the OP777 is a low noise precision amplifier available in a tiny, 8-lead MSOP package. The OP777 is also available in an 8-lead SOIC surface-mount package. This family is extremely useful in instrumentation, for remote sensor acquisition, and in precision filters. The high voltage range allows the use of the parts for single-supply current sourcing and large range instrumentation amplifiers. Both single-supply and dual-supply linear response bridges can also be built. The parts are ideal for use in low-side current monitors in power supply control circuits because the common-mode range extends to ground in the single-supply configuration. 02380-017 VOLTAGE (5V/DIV) OUTPUT TIME (400µs/DIV) Figure 17. No Phase Inversion DESIGN REMINDERS FOR ACHIEVING HIGH PERFORMANCES 30V As with any application, a good ground plane is essential to achieve optimum performance. This can significantly reduce the undesirable effects of ground loops and I × R losses by providing a low impedance reference point. Best results are obtained with a multilayer board design with one layer assigned to the ground plane. 02380-018 OP777/ OP727/ OP747 V p-p = 32V Figure 18. Unity-Gain Follower VS = ±15V AV = 1 To minimize high frequency interference and prevent low frequency ground loops, shield grounding techniques are required when sensors are used. The cable shielding system should include the cable end connectors. VOLTAGE (5V/DIV) VIN 02380-019 VOUT TIME (400µs/DIV) The gain characteristics, of course, are rather different at differing rails. The inputs have a maximum, single temperature offset of 100 μV with an input offset current of 2 nA and input bias current of only 10 nA maximum. With a single 5 V rail, the CMRR is typically 110 dB, and the large signal voltage gain is typically 500 V/mV with a 10 kΩ load. With ±15 V rails, the CMRR increases, not surprisingly, by 10 dB to 120 dB, and the large signal voltage gain increases to 2500 V/mV. Figure 19. Input Voltage Can Exceed the Supply Voltage Without Damage The dynamic performance and noise characteristics of the devices are similar whether they are being used with single or dual supplies. The slew rate with a 2 kΩ load is 200 mV/μs, and the gain bandwidth product is 700 kHz. Peak-to-peak voltage noise from 0.1 Hz to 10 Hz is 0.4 μV, and the voltage noise density at 1 kHz is 15 nV√Hz. Switching power supplies with high output noise is normally used in many systems. This noise generally extends over a broad band of frequencies and occurs as both conducted and radiated noise, and unwanted electric and magnetic fields. The voltage output noise of switching supplies is short-duration voltage transients or spikes that contain frequency components easily extending to 100 MHz or more. Although specifying switching supplies in terms of rms noise is a common vendor practice, users should also specify the peak (or peak-to-peak) amplitudes of the switching spikes with the output loading of the individual system. Capacitors, inductors, ferrite beads, and resistors are used in filters for noise reduction. Linear post regulation can also be done and separates the power supply circuit from sensitive analog circuits. Analog Devices manufactures many anyCAP® low dropout linear regulators. Examples of these devices are the ADP3300 to ADP3310 and ADP3335 to ADP3339 for supply voltages less than 12 V. Rev. B | Page 7 of 8 AN-573 APPLICATION NOTE Capacitors are probably the single most important filter component for switchers. There are generally three classes of capacitors useful in filters in the 10 kHz to 100 MHz frequency range suitable for switchers. Capacitors are broadly distinguished by their generic dielectric types: electrolytic, film, and ceramic. Background and tutorial information on capacitors can be found in the Walter G. Jung, Richard Marsh, Picking Capacitors, Part 1 and Part 2, AUDIO (February, March 1980) article and many vendor catalogs. Chip capacitors should be used for supply bypassing, with one end of the capacitor connected to the ground plane and the other end connected within ⅛ inch of each power pin. An additional large tantalum electrolytic capacitor (4.7 μF to 10 μF) should be connected in parallel. This capacitor does not need to be placed as close to the supply pins because it provides current for fast large signal changes at the output of the device. Use short and wide PCB tracks to decrease voltage drops and minimize inductance. Make track widths at least 200 mils for every inch of track length for lowest DCR and use 1 ounce or 2 ounce copper PCB traces to further reduce IR drops and inductance. Be careful not to exceed the maximum junction temperature or the maximum power dissipation rating of an amplifier. When a capacitive load connects to the output of the amplifier, include the power dissipation caused by the rms ac current delivered to the load in the calculation. Use short leads or leadless components to minimize lead inductance. This minimizes the tendency to add excessive ESL and/or ESR. Surface-mount packages are preferred. Use a large area ground plane for minimum impedance. Note how components behave over frequency, current, and temperature variations. Make use of vendor component models for the simulation of prototype designs, and make sure that lab measurements correspond reasonably with the simulation. SPICE modeling is a powerful tool for predicting the performance of analog circuits. Analog Devices provides macro models for most of its ICs. SPICE models can be downloaded on the OP777 product page. Because models omit many real-life effects and no model can simulate all of the parasitic effects of discrete components and PCB traces, build/prove prototypes before they go into production. To ensure successful prototyping, always use a ground plane for precision or high frequency circuits. Minimize parasitic resistance, capacitance, and inductance. If sockets are required, use pin sockets (cage jacks). Pay equal attention to signal routing, component placement, grounding, and decoupling in both the prototype and the final design. Popular prototyping techniques include Freehand dead-bug using point-to-point wiring and solder-mount, milled PCB from CAD layout, multilayer boards that are double-sided with additional point-to-point wiring. ©2002–2010 Analog Devices, Inc. All rights reserved. Trademarks and registered trademarks are the property of their respective owners. AN02380-0-3/10(B) Rev. B | Page 8 of 8