a
Four-Channel, Four-Quadrant
Analog Multiplier
MLT04
FUNCTIONAL BLOCK DIAGRAM
18-Lead Epoxy DIP (P Suffix)
18-Lead Wide Body SOIC (S Suffix)
FEATURES
Four Independent Channels
Voltage IN, Voltage OUT
No External Parts Required
8 MHz Bandwidth
Four-Quadrant Multiplication
Voltage Output; W = (X × Y)/2.5 V
0.2% Typical Linearity Error on X or Y Inputs
Excellent Temperature Stability: 0.005%
±2.5 V Analog Input Range
Operates from ±5 V Supplies
Low Power Dissipation: 150 mW typ
Spice Model Available
1
18
2
17 GND4
X1
3
16
X4
15
Y4
4
V
5
CC
Y2
6
B
SO
Av (X OR Y)
8.9MHz
–3dB
0
0
Ø (X OR Y)
–20
–40
–90
X & Y MEASUREMENTS
SUPERIMPOSED:
X = 100mV RMS, Y = 2.5V DC
Y = 100mV RMS, X = 2.5V DC
Y3
X2
7
12
X3
8
11
GND3
W2
9
10
W3
W = (X
• Y)/2.5V
The MLT04 is available in 18-pin plastic DIP, and SOIC-18
surface mount packages. All parts are offered in the extended
industrial temperature range (–40°C to +85°C).
100
VCC = +5V
V = –5V
10
THD + NOISE – %
90
Ø – Phase Degrees
O
Av GAIN – dB
20
13
Fabricated in a complementary bipolar process, the MLT04
includes four 4-quadrant multiplying cells which have been lasertrimmed for accuracy. A precision internal bandgap reference
normalizes signal computation to a 0.4 scale factor. Drift over
temperature is under 0.005%/°C. Spot noise voltage of 0.3 µV/√Hz
results in a THD + Noise performance of 0.02% (LPF = 22 kHz)
for the lower distortion Y channel. The four 8 MHz channels
consume a total of 150 mW of quiescent power.
V CC = +5V
V EE = –5V
T A = +25°C
14 V
EE
GND2
LE
APPLICATIONS
Geometry Correction in High-Resolution CRT Displays
Waveform Modulation & Generation
Voltage Controlled Amplifiers
Automatic Gain Control
Modulation and Demodulation
40
MLT-04
917
8
7
6
5
4
3
2
1
10
11
12
13
14
15
16
8
MLT04
TE
Y1
GENERAL DESCRIPTION
The MLT04 is a complete, four-channel, voltage output analog
multiplier packaged in an 18-pin DIP or SOIC-18. These complete
multipliers are ideal for general purpose applications such as voltage
controlled amplifiers, variable active filters, “zipper” noise free
audio level adjustment, and automatic gain control. Other applications include cost-effective multiple-channel power calculations
(I × V), polynomial correction generation, and low frequency
modulation. The MLT04 multiplier is ideally suited for generating
complex, high-order waveforms especially suitable for geometry
correction in high-resolution CRT display systems.
W4
W1
GND1
EE
TA = +25°C
1
LPF = 500kHz
THDX: X = 2.5VP, Y = +2.5V DC
0.1
THDY: Y = 2.5VP, X = +2.5V DC
0.01
1k
10k
100k
1M
FREQUENCY – Hz
10M
100M
Figure 1. Gain & Phase vs. Frequency Response
10
100
1k
10k
FREQUENCY – Hz
100k
1M
Figure 2. THD + Noise vs. Frequency
REV. B
Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
use, nor for any infringements of patents or other rights of third parties
which may result from its use. No license is granted by implication or
otherwise under any patent or patent rights of Analog Devices.
One Technology Way, P.O. Box 9106, Norwood. MA 02062-9106, U.S.A.
Tel: 617/329-4700
Fax: 617/326-8703
MLT04–SPECIFICATIONS (V
CC
= +5 V, VEE = –5 V, VIN = ±2.5 VP, RL = 2 kΩ, TA = +25°C unless otherwise noted.)
Parameter
Symbol
Conditions
Min
Typ
Max
Units
MULTIPLIER PERFORMANCE 1
Total Error2 X
Total Error2 Y
Linearity Error2 X
Linearity Error2 Y
Total Error Drift
Total Error Drift
Scale Factor3
Output Offset Voltage
Output Offset Drift
Offset Voltage, X
Offset Voltage, Y
EX
EY
LEX
LEY
TCEX
TCEY
K
ZOS
TCZOS
XOS
YOS
–2.5 V < X < +2.5 V, Y = +2.5 V
–2.5 V < Y < +2.5 V, X = +2.5 V
–2.5 V < X < +2.5 V, Y = +2.5 V
–2.5 V < Y < +2.5 V, X = +2.5 V
X = –2.5 V, Y = 2.5 V, TA = –40°C to +85°C
Y = –2.5 V, X = 2.5 V, TA = –40°C to +85°C
X = ± 2.5 V, Y = ± 2.5 V, TA = –40°C to +85°C
X = 0 V, Y = 0 V, TA= –40°C to +85°C
X = 0 V, Y = 0 V, TA= –40°C to +85°C
X = 0 V, Y = ± 2.5 V, TA = –40°C to +85°C
Y = 0 V, X = ± 2.5 V, TA = –40°C to +85°C
–5
–5
–1
–1
±2
±2
± 0.2
± 0.2
0.005
0.005
0.40
± 10
50
± 10.5
± 10.5
5
5
+1
+1
% FS
% FS
% FS
% FS
%/°C
%/°C
1/V
mV
µV/°C
mV
mV
DYNAMIC PERFORMANCE
Small Signal Bandwidth
Slew Rate
Settling Time
AC Feedthrough
Crosstalk @ 100 kHz
BW
SR
tS
FTAC
CTAC
VOUT = 0.1 V rms
VOUT = ± 2.5 V
VOUT = ∆2.5 V to 1% Error Band
X = 0 V, Y = 1 V rms @ f = 100 kHz
X = Y = 1 V rms Applied to Adjacent Channel
EN
EN
eN
THDX
THDY
ROUT
VPK
ISC
f = 10 Hz to 50 kHz
Noise BW = 1.9 MHz
f = 1 kHz
f = 1 kHz, LPF = 22 kHz, Y = 2.5 V
f = 1 kHz, LPF = 22 kHz, X = 2.5 V
VCC = +5 V, VEE = –5 V
± 3.0
INPUTS
Analog Input Range
Bias Current
Resistance
Capacitance
IVR
IB
RIN
CIN
GND = 0 V
X=Y=0V
–2.5
SQUARE PERFORMANCE
Total Square Error
ESQ
X=Y=1
5
POWER SUPPLIES
Positive Current
Negative Current
Power Dissipation
Supply Sensitivity
Supply Voltage Range
ICC
IEE
PDISS
PSSR
VRANGE
VCC = 5.25 V, VEE = –5.25 V
VCC = 5.25 V, VEE = –5.25 V
Calculated = 5 V × ICC + 5 V × IEE
X = Y = 0 V, VCC = ∆5% or VEE = ∆5%
For VCC & VEE
15
15
150
–50
–50
30
0.42
50
50
50
8
53
1
–65
–90
MHz
V/µs
µs
dB
dB
76
380
0.3
0.1
0.02
40
± 3.3
30
µV rms
µV rms
µV/√Hz
%
%
Ω
VP
mA
TE
B
SO
Open Loop Output Resistance
Voltage Swing
Short Circuit Current
LE
OUTPUTS
Audio Band Noise
Wide Band Noise
Spot Noise Voltage
Total Harmonic Distortion
0.38
–50
2.3
1
3
± 4.75
+2.5
10
V
µA
MΩ
pF
% FS
20
20
200
10
± 5.25
mA
mA
mW
mV/V
V
O
NOTES
1
Specifications apply to all four multipliers.
2
Error is measured as a percent of the ± 2.5 V full scale, i.e., 1% FS = 25 mV.
3
Scale Factor K is an internally set constant in the multiplier transfer equation W = K × X × Y.
Specifications subject to change without notice.
ABSOLUTE MAXIMUM RATINGS*
Supply Voltages VCC, VEE to GND
Inputs XI, YI
Outputs WI
Operating Temperature Range
Maximum Junction Temperature (T J max)
Storage Temperature
Lead Temperature (Soldering, 10 sec)
Package Power Dissipation
Thermal Resistance θJA
PDIP-18 (N-18)
SOIC-18 (SOL-18)
ORDERING INFORMATION*
±7 V
VCC, VEE
VCC, VEE
–40°C to +85°C
+150°C
–65°C to +150°C
+300°C
(TJ max–TA)/θJA
Model
Temperature
Range
Package
Description
Package
Option
MLT04GP
MLT04GS
MLT04GS-REEL
MLT04GBC
–40°C to +85°C
–40°C to +85°C
–40°C to +85°C
+25°C
18-Pin P-DIP N-18
18-Lead SOIC SOL-18
18-Lead SOIC SOL-18
Die
*For die specifications contact your local Analog sales office. The MLT04
contains 211 transistors.
74°C/W
89°C/W
*Stresses above those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. This is a stress rating only and functional operation of
the device at these or any other conditions above those indicated in the operational
section of this specification are not implied.
–2–
REV. B
MLT04
FUNCTIONAL DESCRIPTION
The MLT04 is a low cost quad, 4-quadrant analog multiplier with
single-ended voltage inputs and voltage outputs. The functional
block diagram for each of the multipliers is illustrated in Figure 3.
Due to packaging constraints, access to internal nodes for externally
adjusting scale factor, output offset voltage, or additional summing
signals is not provided.
ANALOG MULTIPLIER ERROR SOURCES
Multiplier errors consist primarily of input and output offsets, scale
factor errors, and nonlinearity in the multiplying core. An expression for the output of a real analog multiplier is given by:
V O = ( K + ∆K ){(VX + X OS )(V Y + Y OS ) + ZOS + f ( X , Y )}
where:
K
∆K
VX
XOS
VY
YOS
ZOS
ƒ(X, Y)
+VS
MLT04
X1, X2, X3, X4
0.4
G1, G2, G3, G4
W1, W2, W3, W4
=
=
=
=
=
=
=
=
Multiplier Scale Factor
Scale Factor Error
X-Input Signal
X-Input Offset Voltage
Y-Input Signal
Y-Input Offset Voltage
Multiplier Output Offset Voltage
Nonlinearity
Y1, Y2, Y3, Y4
TE
Executing the algebra to simplify the above expression yields
expressions for all the errors in an analog multiplier:
–VS
Figure 3. Functional Block Diagram of Each MLT04
Multiplier
VCC
XIN
GND
YIN
W
OUT
22k
200µA
VEE
Dependence on Input
KVXVY
True Product
Goes to Zero As Either or
Both Inputs Go to Zero
∆KVYVY
Scale-Factor Error
Goes to Zero at VX, VY = 0
VXYOS
Linear “X” Feedthrough
Due to Y-Input Offset
Proportional to VX
VYXOS
Linear “Y” Feedthrough
Due to X-Input Offset
Proportional to VY
XOSYOS
Output Offset Due to X-,
Y-Input Offsets
Independent of VX, VY
ZOS
Output Offset
Independent of VX, VY
ƒ(X, Y)
Nonlinearity
Depends on Both V X, VY.
Contains Terms Dependent
on VX, VY, Their Powers
and Cross Products
B
SO
INTERNAL
BIAS
Description
LE
Each of the MLT04’s analog multipliers is based on a Gilbert cell
multiplier configuration, a 1.23 V bandgap reference, and a unityconnected output amplifier. Multiplier scale factor is determined
through a differential pair/trimmable resistor network external to
the core. An equivalent circuit for each of the multipliers is shown
in Figure 4.
Term
22k
22k
200µA
200µA
SCALE
FACTOR
200µA
200µA
200µA
As shown in the table, the primary static errors in an analog
multiplier are input offset voltages, output offset voltage, scale
factor, and nonlinearity. Of the four sources of error, only two are
externally trimmable in the MLT04: the X- and Y-input offset
voltages. Output offset voltage in the MLT04 is factory-trimmed to
± 50 mV, and the scale factor is internally adjusted to ± 2.5% of full
scale. Input offset voltage errors can be eliminated by using the
optional trim circuit of Figure 6. This scheme then reduces the net
error to output offset, scale-factor (gain) error, and an irreducible
nonlinearity component in the multiplying core.
Figure 4. Equivalent Circuit for the MLT04
O
Details of each multiplier’s output-stage amplifier are shown in
Figure 5. The output stages idles at 200 µA, and the resistors in
series with the emitters of the output stage are 25 Ω. The output
stage can drive load capacitances up to 500 pF without oscillation.
For loads greater than 500 pF, the outputs of the MLT04 should
be isolated from the load capacitance with a 100 Ω resistor.
VCC
+VS
50kΩ
50kΩ
25Ω
I
±100mV
FOR XOS, YOS TRIM
CONNECT TO SUM
NODE OF AN EXT OP AMP
W
OUT
–VS
25Ω
Figure 6. Optional Offset Voltage Trim Configuration
VEE
Figure 5. Equivalent Circuit for MLT04 Output Stages
REV. B
–3–
MLT04
Feedthrough
In the ideal case, the output of the multiplier should be zero if
either input is zero. In reality, some portion of the nonzero input
will “feedthrough” the multiplier and appear at the output. This is
caused by the product of the nonzero input and the offset voltage of
the “zero” input. Introducing an offset equal to and opposite of the
“zero” input offset voltage will null the linear component of the
feedthrough. Residual feedthrough at the output of the multiplier
is then irreducible core nonlinearity.
VERTICAL – 5mV/DIV
100
Typical X- and Y-input feedthrough curves for the MLT04 are
shown in Figures 7 and 8, respectively. These curves illustrate
MLT04 feedthrough after “zero” input offset voltage trim.
Residual X-input feedthrough measures 0.08% of full scale,
whereas residual Y-input feedthrough is almost immeasurable.
TE
Figure 9. X-Input Nonlinearity @ Y = +2.5 V
X-INPUT: ±2.5V @ 10Hz
YOS NULLED
TA = +25°C
0%
HORIZONTAL – 0.5V/DIV
100
VERTICAL – 5mV/DIV
10
10
0%
10
0%
X-INPUT: ±2.5V @ 10Hz
Y-INPUT: –2.5V
YOS NULLED
T = +25°C
A
Figure 10. X-Input Nonlinearity @ Y = –2.5 V
B
SO
VERTICAL – 5mV/DIV
90
90
HORIZONTAL – 0.5V/DIV
Figure 7. X-Input Feedthrough with YOS Nulled
100
0%
LE
90
X-INPUT: ±2.5V @ 10Hz
Y-INPUT: +2.5V
YOS NULLED
T = +25°C
A
10
HORIZONTAL – 0.5V/DIV
Y-INPUT: ±2.5V @ 10Hz
XOS NULLED
TA = +25°C
100
VERTICAL – 5mV/DIV
VERTICAL – 5mV/DIV
100
90
90
10
0%
HORIZONTAL – 0.5V/DIV
HORIZONTAL – 0.5V/DIV
O
Y-INPUT: ±2.5V @ 10Hz
X-INPUT: +2.5V
XOS NULLED
TA = +25°C
Figure 11. Y-Input Nonlinearity @ X = +2.5 V
Figure 8. Y-Input Feedthrough with XOS Nulled
Nonlinearity
Multiplier core nonlinearity is the irreducible component of error.
It is the difference between actual performance and “best-straightline” theoretical output, for all pairs of input values. It is expressed
as a percentage of full scale with all other dc errors nulled. Typical
X- and Y-input nonlinearities for the MLT04 are shown in Figures
9 through 12. Worst-case X-input nonlinearity measured less than
0.2%, and Y-input nonlinearity measured better than 0.06%. For
modulator/demodulator or mixer applications it is, therefore,
recommended that the carrier be connected to the X-input while
the signal is applied to the Y-input.
VERTICAL – 5mV/DIV
100
90
Y-INPUT: ±2.5V @ 10Hz
X-INPUT: –2.5V
XOS NULLED
T = +25°C
A
10
0%
HORIZONTAL – 0.5V/DIV
Figure 12. Y-Input Nonlinearity @ X = –2.5 V
–4–
REV. B
Typical Performance Characteristics – MLT04
12
180
TA = +25°C
V = ±5V
NBW = 10Hz –50kHz
TA = +25°C
90
90
3
GAIN –dB
45
GAIN
0
0
–3
–45
PHASE
–6
–90
PHASE = 68.3°
@ 7.142 MHz
–9
10
0%
–12
10k
TIME = 10ms/DIV
Figure 13. Broadband Noise
–135
–180
10M
TE
100k
1M
FREQUENCY – Hz
Figure 16. X-Input Gain and Phase vs. Frequency
9
V S = ±5V
V X = +2.5V
135
6
V Y = 100mV
90
3
GAIN –dB
90
10
0%
O
–45
PHASE
–6
–90
PHASE = 68.1°
@ 8.064 MHz
–9
–12
10k
–135
–180
10M
100k
1M
FREQUENCY – Hz
Figure 17. Y-Input Gain and Phase vs. Frequency
8
6
VS = ±5V
TA = +25°C
CL= 320pF
4
CL= 560pF
CL= 220pF
Hz
2
1000
AV GAIN – dB
NOISE DENSITY – nV/
0
–3
Figure 14. Broadband Noise
100
0
–2
NO CL
CL= 100pF
–4
–6
VS = ±5V
RL = 2kΩ
TA = +25°C
–8
–10
–12
0
10
100
1k
10k
FREQUENCY – Hz
100k
1k
1M
Figure 15. Noise Density vs. Frequency
REV. B
45
GAIN
0
TIME = 10ms/DIV
10000
180
T A = +25°C
LE
NBW = 1.9MHz
TA = +25°C
100
B
SO
OUTPUT NOISE VOLTAGE – 625µV/DIV
12
PHASE – Degrees
OUTPUT NOISE VOLTAGE – 100µV/DIV
6
100
135
S
VX = 100mV
VY = +2.5V
PHASE – Degrees
9
10k
100k
1M
FREQUENCY – Hz
10M
100M
Figure 18. Amplitude Response vs. Capacitive Load
–5–
MLT04 – Typical Performance Characteristics
0
ΩX-INPUT = +2.5V
RL = 10kΩ
VS = ±5V
VX = 0V
VY = 1Vpk
–40
–60
10
TIME – 100ns/DIV
Figure 22. Y-Input Small-Signal Transient Response,
CL = 30 pF
–100
1k
10k
100k
1M
3M
TE
FREQUENCY – Hz
VERTICAL – 50mV/DIV
Figure 19. Feedthrough vs. Frequency
TA = 25°C
VS = ±5V
VX = ±2.5Vpk
–20
VY = +2.5VDC
–40
–60
ΩX-INPUT = +2.5V
RL = 10kΩ
TA = +25°C
100
90
LE
0
10
0%
TIME – 100ns/DIV
Figure 23. Y-Input Small-Signal Transient Response,
CL = 100 pF
–80
–100
–120
1k
B
SO
CROSSTALK – dB
TA = +25°C
90
VY = 0V
VX = 1Vpk
–80
10k
100k
FREQUENCY – Hz
1M
2.0
1.5
1.0
10
0%
O
ΩX-INPUT: +2.5V
RL = 10kΩ
TA = +25°C
TIME = 100ns/DIV
Figure 24. Y-Input Large-Signal Transient Response, CL = 30 pF
0
–0.5
90
ΩVS = ±5V
RL = 2kΩ
TA = +25°C
Y = 100mV RMS
X = 2.5VDC
0.5
100
10M
Figure 20. Crosstalk vs. Frequency
X = 100mV RMS
Y = 2.5VDC
–1.0
–1.5
100
VERTICAL – 1V/DIV
AV GAIN – dB
100
0%
VERTICAL – 1V/DIV
FEEDTHROUGH – dB
–20
VERTICAL – 50mV/DIV
TA = +25°C
–2.0
–2.5
–3.0
1k
10k
100k
1M
10M
100M
FREQUENCY – Hz
90
10
0%
ΩX-INPUT: +2.5V
RL = 10kΩ
TA = +25°C
Figure 21. Gain Flatness vs. Frequency
TIME = 100ns/DIV
Figure 25. Y-Input Large-Signal Transient Response,
CL = 100 pF
–6–
REV. B
MLT04
1
9
80
V = 100mV
Y
–3dB-BANDWIDTH – MHz
THD + NOISE – %
X-INPUT
Y = +2.5VDC
0.1
ΩVS = ±5V
RL = 2kΩ
T A = +25° C
fO = 1kHz
FLPF = 22kHz
0.01
8
75
–3dB BW
7
70
PHASE @ –3dB BW
6
65
PHASE @ –3dB BW – Degrees
VS = ±5V
VX = +2.5V
Y-INPUT
X = +2.5VDC
0.001
1
5
–75
10
INPUT SIGNAL LEVEL – Volts P-P
Figure 26. THD + Noise vs. Input Signal Level
LE
–0.1
B
SO
LINEARTY ERROR – %
X
0
–50
–25
0
25
50
MAXIMUM OUTPUT SWING – Volts p-p
Y
Vs = ±5V
0.1
75
100
4
3
ΩTA = +25°C
RL = 2kΩ
VS = ±5V
2
1
0
1k
PHASE @ –3dB BW
65
6
OUTPUT SWING – Volts
70
7
POSITIVE SWING
3.5
PHASE @ –3dB BW – Degrees
–3dB BW
–50
–25
0
25
50
75
100
10M
4.0
X
75
1M
4.5
V = +2.5V
8
100k
Figure 30. Maximum Output Swing vs. Frequency
80
Y
10k
FREQUENCY – Hz
V = ±5V
S
V = 100mV
O
–3dB-BANDWIDTH – MHz
60
125
1%
DISTORTION
5
3.0
2.5
2.0
NEGATIVE SWING
1.5
1.0
VS = ±5V
TA = +25°C
0.5
60
125
0
10
100
1k
10k
ΩLOAD RESISTANCE – Ω
TEMPERATURE – °C
Figure 28. X-Input Gain Bandwidth vs. Temperature
REV. B
100
6
125
Figure 27. Linearity Error vs. Temperature
5
–75
75
7
TEMPERATURE – °C
9
25
50
TEMPERATURE – °C
8
≤V = +2.5V, –2.5V ≤ V ≤ +2.5V
X
Y
V = +2.5V, –2.5V ≤ V ≤ +2.5V
0.2
–0.3
–75
0
–25
Figure 29. Y-Input Gain Bandwidth vs. Temperature
0.3
–0.2
–50
TE
0.1
Figure 31. Maximum Output Swing vs. Resistive Load
–7–
MLT04
0.407
300
TA = +25°C
V = ±5V
VS = ±5V
NO LOAD
S
250
X = ±2.5V
200
YOS @ X = ±2.5V
0.406
SCALE FACTOR – 1/V
UNITS
SS = 1000 MULTIPLIERS
XOS @ Y = ±2.5V
150
100
0.405
0.404
0.403
50
–7.5
–5 –2.5
0
2.5
5
OFFSET VOLTAGE – mV
7.5
10
0.402
–75
TE
0
–12.5 –10
12.5
–25
0
25
50
75
100
125
TEMPERATURE – °C
Figure 35. Scale Factor vs. Temperature
Figure 32. Offset Voltage Distribution
400
6
T = +25°C
VS = ±5V
A
SS = 1000
MULTIPLIERS
350
VS = ±5V
LE
4
VX = VY = 0V
300
XOS, Y = ±2.5V
2
250
UNITS
VOS – mV
–50
0
200
B
SO
150
–2
YOS, X = ±2.5V
–4
–6
–75
–50
–25
0
25
50
TEMPERATURE – °C
75
100
50
0
–15
125
–12
–9
–6
–3
0
3
6
9
12
15
OUTPUT OFFSET VOLTAGE – mV
Figure 36. Output Offset Voltage (ZOS) Distribution
Figure 33. Offset Voltage vs. Temperature
400
100
10
SS = 1000 MULTIPLIERS
300
UNITS
250
200
150
100
50
0
0.395 0.3975 0.400
0.4025 0.405
0.4075 0.410
0.4125
V = ±5V
s
OUTPUT OFFSET VOLTAGE – mV
VS = ±5V
O
350
TA = +25°C
5
0
–5
–10
–75
0.415
SCALE FACTOR – 1/V
Figure 34. Scale Factor Distribution
–50
–25
0
25
50
TEMPERATURE – °C
75
100
125
Figure 37. Output Offset Voltage (ZOS) vs. Temperature
–8–
REV.B
MLT04
17
15
12
OUTPUT VOLTAGE OFFSET – mV
SUPPLY CURRENT – mA
VS = ±5V
NO LOAD
VX = VY = 0
16
15
14
σX +3σ
9
6
3
0
X
–3
–6
–9
σX –3σ
–12
–15
–50
–25
0
25
50
75
100
0
125
Figure 38. Supply Current vs. Temperature
800
1000
0.424
TA = +25°C
LE
0.420
VS = ±5V
+PSRR
60
–PSRR
40
20
0
100
0.416
1k
10k
FREQUENCY – Hz
100k
1M
Figure 39. Power Supply Rejection vs. Frequency
1.25
σX +3σ
O
1.0
0.75
0.50
0.25
X
0
–0.25
–0.50
–0.75
σX –3σ
–1.0
–1.25
0
200
400
600
800
1000
HOURS OF OPERATION AT +125°C
Figure 40. Linearity Error (LE) Distribution Accelerated
by Burn-in
–9–
SCALE FACTOR – 1/V
80
B
SO
POWER SUPPLY REJECTION – dB
600
Figure 41. Output Voltage Offset (ZOS) Distribution
Accelerated by Burn-in
100
LINEARITY ERROR – %
400
HOURS OF OPERATION AT +125°C
TEMPERATURE – °C
REV. B
200
TE
13
–75
σX +3σ
0.412
0.408
X
0.404
0.400
0.396
σX –3σ
0.392
0.388
0.384
0
200
400
600
800
1000
HOURS OF OPERATION AT +125°C
Figure 42. Scale Factor (K) Distribution Accelerated by Burn-in
MLT04
Multiplier Connections
Figure 43 llustrates the basic connections for multiplication. Each
of the four independent multipliers has single-ended voltage inputs
(X, Y) and a low impedance voltage output (W). Also, each
multiplier has its own dedicated ground connection (GND) which
is connected to the circuit’s analog common. For best performance, circuit layout should be compact with short component
leads and well-bypassed supply voltage feeds. In applications where
fewer than four multipliers are used, all unused analog inputs must
be returned to the analog common.
The equation shows a dc term at the output which will vary
strongly with the amplitude of the input, V IN. The output dc offset
can be eliminated by capacitively coupling the MLT04’s output
with a high-pass filter. For optimal spectral performance, the
filter’s cutoff frequency should be chosen to eliminate the input
fundamental frequency.
A source of error in this configuration is the offset voltages of the X
and Y inputs. The input offset voltages produce cross products
with the input signal to distort the output waveform. To circumvent this problem, Figure 45 illustrates the use of inverting
amplifiers configured with an OP285 to provide a means by which
the X- and Y-input offsets can be trimmed.
ΩP1
50kΩ
+5V
–5V
TE
APPLICATIONS
The MLT04 is well suited for such applications as modulation/
demodulation, automatic gain control, power measurement, analog
computation, voltage-controlled amplifiers, frequency doublers,
and geometry correction in CRT displays.
XOS TRIM
ΩR5
500kΩ
1
W1
2
GND1
X1
3
X1
X4 16
X4
Y1
4
Y1
Y4 15
Y4
5
1
10
11
12
13
14
15
16
98
8
7
6
5
4
3
2
VCC 17
W1
W4 18
R1
10k
W4
GND4 17
R2
10k
2
A1
3
0.1µF
MLT04
5
VEE 14
–5V
Y2
6 Y2
Y3 13
Y3
X2
7 X2
X3 12
X3
8
W2
GND3
11
W2
ΩR6
500kΩ
–5V
W3 10
W3
W1–4 = 0.4 (X1–4
• Y1–4)
2
0.4
+
A2
0.1µF
+
7
4
1
W1
C1
100pF
VO
ΩRL
10kΩ
+
6
R3
10k
B
SO
9
GND2
A1, A2 = 1/2 OP285
LE
+5V
VIN
+
1/4 MLT04
3
1
R4
10k
YOS TRIM
ΩP2
50kΩ
+5V
Figure 45. Frequency Doubler with Input Offset Voltage
Trims
Figure 43. Basic Multiplier Connections
Squaring and Frequency Doubling
As shown in Figure 44, squaring of an input signal, V IN, is achieved
by connecting the X-and Y-inputs in parallel to produce an output
of VIN2/2.5 V. The input may have either polarity, but the output
will be positive.
+5V
Feedback Divider Connections
The most commonly used analog divider circuit is the “inverted
multiplier” configuration. As illustrated in Figure 46, an “inverted
multiplier” analog divider can be configured with a multiplier
operating in the feedback loop of an operational amplifier. The
general form of the transfer function for this circuit configuration is
given by:
O
0.1µF
VIN
X
GND
Y
1/4 MLT04
+
0.4
W
W = 0.4 VIN2
+
0.1µF
–5V
Figure 44. Connections for Squaring
R2 VIN
VO = −2.5 V ×
×
R1 VX
Here, the multiplier operates as a voltage-controlled potentiometer
that adjusts the loop gain of the op amp relative to a control signal,
VX. As the control signal to the multiplier decreases, the output of
the multiplier decreases as well. This has the effect of reducing
negative feedback which, in turn, decreases the amplifier’s loop
gain. The result is higher closed-loop gain and reduced circuit
bandwidth. As VX is increased, the output of the multiplier
increases which generates more negative feedback — closed-loop
gain drops and circuit bandwidth increases. An example of an
“inverted multiplier” analog divider frequency response is shown in
Figure 47.
When the input is a sine wave given by V IN sin ωt, the squaring
circuit behaves as a frequency doubler because of the trigonometric
identity:
(VIN sin ωt )2
V 2 1
= IN (1 − cos 2 ωt )
2.5V
2.5V 2
–10–
REV. B
MLT04
1/4 MLT04
+
3
X1
1/4 MLT04
+
W1
1
0.4
2
R2
10k
+
VX
D1
1N4148
GND1
R1
10k
V
+
VO = –2.5V •
OP113 6
3
VIN
VO =
VX
Figure 46. “Inverted-Multiplier” Configuration for
Analog Division
70
50
VX = 0.025V
40
30
VX = 0.25V
B
SO
GAIN – dB
60
10k
100k
FREQUENCY – Hz
1M
10M
In this circuit, the ratio of R2 to R1 sets the passband gain, and the
break frequency of the filter, ωLP, is given by:
Figure 47. Signal-Dependent Feedback Makes Variables
Out of Amplifier Bandwidth and Stability
R1 VX
ωLP =
R1 + R2 2.5RC
O
Although this technique works well with almost any operational
amplifier, there is one caveat: for best circuit stability, the unitygain crossover frequency of the operational amplifier should be
equal to or less than the MLT04’s 8 MHz bandwidth.
X1
+
Connection for Square Rooting
Another application of the “inverted multiplier” configuration is the
square-root function. As shown in Figure 48, both inputs of the
MLT04 are wired together and are used as the output of the
circuit. Because the circuit configuration exhibits the following
generalized transfer function:
1/4 MLT04
+
VX
W1
2
C
80pF
R
10k
1
0.4
2
A1
+
R1
10k
4
3
R2
10k
Y1
1
+
A1 = 1/2 OP285
VIN
VO
the input signal voltage is limited to the range –2.5 V ≤ VIN < 0. To
prevent circuit latchup due to positive feedback or input signal
polarity reversal, a 1N4148-type junction diode is used in series
with the output of the multiplier.
Figure 48. Connections for Square Rooting
3
GND1
R2
VO = −2.5 ×
×VIN
R1
REV. B
–2.5V • VIN
R2
1
= −
R1 R2 + R1 2.5RC
+
1
s
R1 VX
VO
VIN
VX = 2.5V
1k
O
LE
AVOL
OP113
V
+
Voltage-Controlled Low-Pass Filter
The circuit in Figure 49 illustrates how to construct a voltagecontrolled low-pass filter with an analog multiplier. The advantage
with this approach over conventional active-filter configurations is
that the overall characteristic cut-off frequency, ωO, will be directly
proportional to a multiplying input voltage. This permits the
construction of filters in which the capacitors are adjustable
(directly or inversely) by a control voltage. Hence, the frequency
scale of a filter can be manipulated by means of a single voltage
without affecting any other parameters. The general form of the
circuit’s transfer function is given by:
90
80
2
IN
VO
TE
3
100
Y1
4
OP113 6
0
X1
2
0.4
R2
10k
Y1
2
10
1
4
R1
10k
VIN
20
W1
3
VIN
fLP =
VX
π10πRC
1
=–
1+S
5RC
VX
; fLP = MAX @ VX = 2.5V
Figure 49. A Voltage-Controlled Low-Pass Filter
For example, if R1 = R2 = 10 kΩ , R = 10 kΩ , and C = 80 pF,
–11–
VO
MLT04
then the output of the circuit has a pole at frequencies from 1 kHz
to 100 kHz for VX ranging from 25 mV to 2.5 V. The performance
of this low-pass filter is illustrated in Figure 20.
OUTLINE DIMENSIONS
Dimensions shown in inches and (mm).
30
18
10
1
9
0.280 (7.11)
0.240 (6.10)
PIN 1
20
0.925 (23.49)
0.845 (21.47)
0.210
(5.33)
MAX
0.130
(3.30)
MIN
0.160 (4.06)
0.115 (2.93)
V = 0.025V
– 10
X
0.25V
2.5V
0.022 (0.558)
0.014 (0.356)
– 20
– 30
10
100
1k
0.325 (8.25)
0.300 (7.62)
0.015
(0.38)
MIN
10k
100k
FREQUENCY – Hz
1M
SEATING
PLANE
TE
GAIN – dB
10
0
10M
C1845–18–10/93
18-Lead Epoxy DIP (P Suffix)
0.100
(2.54)
BSC
0.070 (1.77)
0.045 (1.15)
15°
0°
0.015 (0.38)
0.008 (0.20)
18-Lead Wide-Body SOL (S Suffix)
LE
Figure 50. Low-Pass Cutoff Frequency vs. Control
Voltage, VX
18
10
With this approach, it is possible to construct parametric biquad
filters whose parameters (center frequency, passband gain, and Q)
can be adjusted with dc control voltages.
0.2992 (7.60)
0.2914 (7.40)
0.4193 (10.65)
0.3937 (10.00)
PIN 1
B
SO
1
0.1043 (2.65)
0.0926 (2.35)
0.4625 (11.75)
0.4469 (11.35)
0.0500 (1.27)
BSC
0.0192 (0.49)
0.0138 (0.35)
0.0125 (0.32)
0.0091 (0.23)
0.0291 (0.74)
x 45°
0.0098 (0.25)
8°
0°
0.0500 (1.27)
0.0157 (0.40)
PRINTED IN U.S.A.
O
0.0118 (0.30)
0.0040 (0.10)
9
–12–
REV. B