PDF Obsolete Data Sheets

a
450 V/ms, Precision,
Current-Feedback Op Amp
AD846
CONNECTION DIAGRAM
FEATURES
AC PERFORMANCE
Small Signal Bandwidth: 80 MHz (A V = –1)
Slew Rate: 450 V/ms
Full Power Bandwidth: 6.8 MHz at 20 V p-p,
R L = 500 V
Fast Settling: for 10 V Step: 110 ns to 0.01%,
80 ns to 0.1%
Differential Gain: <0.01% @ 4.4 MHz
Differential Phase: <0.0288 @ 4.4 MHz
Total Harmonic Distortion (THD): 0.0005% @ 100 kHz
Open-Loop Transimpedance: 200 MV
Input Voltage Noise: 2 nV/√Hz
OBS
Plastic Mini-DIP (N) Package
and
Cerdip (Q) Package
OLE
DC PERFORMANCE
Input Offset Voltage: 75 mV max (B Grade)
Input Offset Drift: 3.5 mV/8C max (B Grade)
Quiescent Supply Current: 6.5 mA max
APPLICATIONS
High Speed DAC Buffers
Multiflash ADC Error Amplifiers
Flash ADC Buffers
Coaxial Cable Drivers
High Performance Audio Circuitry
Available in Plastic Mini-DIP, Hermetic Cerdip, and
Hermetic Metal Can Packages
MIL-STD-883B Parts Available
PRODUCT DESCRIPTION
The AD846 is a monolithic, very high speed operational amplifier offering high performance. Although technically classed as a
current-feedback or transimpedance amplifier, it may be used in
much the same way as traditional op amps while providing significant performance benefits. Employing Analog Devices’ junction isolated complementary bipolar (CB) process, the AD846
achieves true “12-bit” (0.01%) precision on critical ac and dc
parameters, a level of performance unmatched by amplifiers
fabricated using either the dielectrically isolated (DI) or other
bipolar processes.
The AD846 offers significant advantages over conventional high
speed operational amplifiers. It maintains a nearly constant
bandwidth and settling time to 0.01% over a wide range of
closed-loop gains. This makes the AD846 ideal for amplifying
the residue in multiple-pass analog-to-digital converters.
TE
Other advantages include: low input errors and high open-loop
transresistance (200 MΩ) into a 500 Ω load, ensuring true
12-bit dc accuracy for closed-loop gains from –1 to gains greater
than –100. This combination of ac and dc performance makes
the AD846 an excellent choice for buffering precision high
speed DACs and flash ADCs.
The AD846 is available in three performance grades. The
AD846A and AD846B are rated over the industrial temperature
range of –40°C to +85°C. The AD846S is rated over the full
military temperature range of –55°C to +125°C and is available
processed to MIL-STD-883B, Rev C.
Extended reliability PLUS screening is available specified over
the commercial temperature range. PLUS screening includes
168 hour burn-in as well as other environmental and physical
tests. The AD846 is available in two types of 8-pin package:
plastic mini-DIP and hermetic cerdip. “A” and “S” grade chips
are also available.
PRODUCT HIGHLIGHTS
1. The AD846 achieves settling times of 110 ns to 0.01% for
gains of –1 to –10, with a 450 V/µs slew rate, while consuming only 5 mA of supply current.
2. For closed-loop gains of –1 to –100, the high speed performance of the AD846 is achieved without sacrificing full
12-bit dc precision.
3. The AD846 is well suited to line driver and video buffer
applications where the properties of low distortion and high
slew rate are required.
REV. B
Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
use, nor for any infringements of patents or other rights of third parties
which may result from its use. No license is granted by implication or
otherwise under any patent or patent rights of Analog Devices.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 617/329-4700
Fax: 617/326-8703
AD846–SPECIFICATIONS (@ +258C and 615 V dc, unless otherwise noted)
Model
INPUT OFFSET VOLTAGE1
Initial
TMIN –TMAX
vs. Temperature
vs. Supply (PSRR)
Initial
TMIN –TMAX
vs. Common Mode (CMRR)
Initial
TMIN–TMAX
INPUT BIAS CURRENT3
–Input Bias Current
Initial
TMIN –TMAX
vs. Temperature
vs. Supply
Initial
TMIN –TMAX
vs. Common Mode
Initial
TMIN –TMAX
+Input Bias Current
Initial
TMIN –TMAX
vs. Temperature
vs. Supply
Initial
TMIN –TMAX
vs. Common Mode
Initial
TMIN –TMAX
INPUT CHARACTERISTICS
Input Resistance
–Input
+Input
Input Capacitance
–Input
+Input
INPUT VOLTAGE RANGE
Common Mode
INPUT VOLTAGE NOISE
Input Current Noise
–Input
+Input
OPEN LOOP
TRANSRESISTANCE
Conditions
Min
25
50
0.8
FREQUENCY RESPONSE
Small Signal Bandwidth
(–3 dB)
Full Power Bandwidth4
Rise Time
Overshoot
Slew Rate
Settling Time
10 V Step, AV = –1
TOTAL HARMONIC
DISTORTION5
Min
200
350
5
AD846B
Typ Max
25
50
0.8
Min
75
125
3.5
AD846S
Typ Max
25
100
1
200
350
5.5
5 V–18 V2
VCM = ± 10 V
Units
µV
µV
µV/°C
110
110
125
120
120
116
125
120
110
94
125
116
dB
dB
110
110
125
120
120
116
125
120
110
94
125
116
dB
dB
OBS
OUTPUT CHARACTERISTICS
Voltage
Current
Output Resistance
AD846A
Typ Max
150
450
6
450
1200
20
100
400
6
250
750
17
150 450
1000 1500
9
20
nA
nA
nA/°C
9
11
15
20
9
11
10
15
9
11
15
25
nA/V
nA/V
5
5
10
15
3
3
5
7
5
5
10
20
nA/V
nA/V
3
4
15
15
20
80
3
4
15
5
7
45
3
5
15
15
20
80
µA
µA
nA/°C
5
5
15
20
5
5
10
15
5
5
15
20
nA/V
nA/V
5
5
15
15
3
3
10
10
5
5
15
20
nA/V
nA/V
5 V–18 V2
VCM = ± 10 V
5 V–18 V2
VCM = ± 10 V
OLE
50
10
50
10
2
2
2
2
± 10
± 10
TE
50
10
Ω
kΩ
2
2
pF
pF
± 10
F = 1 kHz
2
2
2
V
nV/√Hz
1 kHz
1 kHz
20
6
20
6
20
6
pA/√Hz
pA/√Hz
200
MΩ
MΩ
VOUT = ± 10 V
RLOAD = 500 Ω
TMIN –TMAX
RLOAD = 500 Ω
Short Circuit
Open Loop
100
50
200
610
150
75
200
610
100
50
610
65
16
65
16
65
16
V
mA
Ω
AV = –1 RF = 1k
AV = –10 RF = 875 Ω
AV = –30 RF = 875 Ω
VOUT = 20 V p-p
RI = 500 Ω
AV = –1
AV = –1
AV = –1
80
31
15
80
31
15
80
31
15
MHz
MHz
MHz
6.8
110
20
450
6.8
10
20
450
6.8
10
20
450
MHz
ns
%
V/µs
to 0.1%
to 0.01%
80
110
80
110
80
110
ns
ns
F = 100 kHz
0.0005
0.0005
0.0005
%
–2–
REV. B
AD846
Model
DIFFERENTIAL GAIN
Conditions
F = 4.4 MHz, RL = 100 Ω
DIFFERENTIAL PHASE
F = 4.4 MHz, RL = 100 Ω
POWER SUPPLY
Rated Performance
Operating Range
Quiescent Current
Min
AD846A
Typ Max
0.01
Min
0.028
±5
TMIN –TMAX
± 15
5
TRANSISTOR COUNT
AD846B
Typ Max
0.01
Min
AD846S
Typ Max
0.01
0.028
618
6.5
72
±5
± 15
5
0.028
618
6.5
65
± 15
5
72
Units
%
Degrees
618
7
V
V
mA
72
NOTES
1
Input Offset Voltage Specifications are guaranteed after 5 minutes at T A = +25°C.
2
Test Conditions: +V S = 15 V, –V S = 5 V to 18 V and +V S = 5 V to 18 V, –V S = 15 V.
3
Bias Current Specifications are guaranteed maximum after 5 minutes at T A = +25°C.
4
FPBW = Slew Rate/2 π VPEAK.
5
Total Harmonic Distortion.
All min and max specifications are guaranteed. Specifications shown in boldface are tested on all production units at final electrical test. Results from those tests are
used to calculate outgoing quality levels.
Specifications subject to change without notice.
OBS
ABSOLUTE MAXIMUM RATINGS 1
OLE
ORDERING GUIDE
Supply Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ± 18 V
Internal Power Dissipation2
Plastic Package . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1.5 W
Cerdip Package . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1.3 W
Common-Mode Input Voltage, Max Safe . . . . . . . |VS| – 3 V
Output Short Circuit Duration . . . . . . . . . . . . . . . . Indefinite
Differential Input Voltage . . . . . . . . . . . . . . . . . . . . . . . ± 1 V
Continuous Input Current
Inverting or Noninverting . . . . . . . . . . . . . . . . . . . . 2.0 mA
Storage Temperature Range (Q) . . . . . . . . –65°C to +150°C
Storage Temperature Range (N) . . . . . . . . –65°C to +125°C
Operating Temperature Range
AD846A/B . . . . . . . . . . . . . . . . . . . . . . . . –40°C to +85°C
AD846S . . . . . . . . . . . . . . . . . . . . . . . . . –55°C to +125°C
Lead Temperature Range (Soldering 60 sec) . . . . . . . +300°C
ESD Rating . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3500 V
Temperature Range
Package
Option2
AD846AN
AD846BN
AD846AQ
AD846BQ
AD846SQ
AD846SQ/883B
5962-8964601PA
–40°C to +85°C
–40°C to +85°C
–40°C to +85°C
–40°C to +85°C
–55°C to +125°C
–55°C to +125°C
–55°C to +125°C
N-8
N-8
Q-8
Q-8
Q-8
Q-8
Q-8
TE
NOTES
1
“A” and “S” grade chips are also available.
2
N = Plastic DIP Package; Q = Cerdip Package.
METALIZATION PHOTOGRAPH
Dimensions shown in inches and (mm).
Consult factory for latest dimensions.
NOTES
1
Stresses above those listed under “Absolute Maximum Ratings” may cause
permanent damage to the device. This is a stress rating only; the functional
operation of the device at these or any other conditions above those indicated in the
operational sections of this specification is not implied. Exposure to absolute
maximum rating conditions for extended periods may affect device reliability.
2
Maximum internal power dissipation is specified so that T J does not exceed
+175°C at an ambient temperature of +25°C, derate cerdip (Q) package at
8.7 mW/°C and plastic (N) package at 10 mW/°C.
Plastic Package: θJA = 100°C/Watt, θJC = 33°C/W.
Cerdip Package: θJA = 110°C/Watt, θJC = 30°C/W.
REV. B
Model1
–3–
AD846–Typical Characteristics
OBS
Figure 1. Input Voltage Swing
vs. Supply
Figure 2. Output Voltage Swing
vs. Supply
Figure 3. Quiescent Current vs.
Supply Voltage
OLE
TE
Figure 4. Quiescent Supply Current
vs. Temperature
Figure 5. Output Voltage Swing vs.
Resistive Load
Figure 6. Large Signal Frequency
Response
Figure 7. Open-Loop Transimpedance
vs. Supply
Figure 8. Positive Input Bias Current
vs. Common-Mode Voltage
Figure 9. Negative Input Bias Current
vs. Common-Mode Voltage
–4–
REV. B
AD846
OBS
Figure 10. Positive Input Bias Current
vs. Temperature
Figure 11. Negative Input Bias
Current vs. Temperature
Figure 12. Power Supply Rejection
vs. Frequency
OLE
Figure 13. Common-Mode Rejection
vs. Frequency
Figure 16. Short Circuit Current Limit
vs. Temperature
REV. B
TE
Figure 14. Input Noise Voltage
Spectral Density
Figure 17. Slew Rate vs. Temperature
–5–
Figure 15. Inverting Input Noise
Current Spectral Density
Figure 18. Slew Rate vs. Input
Error Signal
AD846–Typical Characteristics, Inverting Gain of 1
OBS
Figure 19a. Inverting Amplifier,
Gain of 1
Figure 19b. Large Signal Pulse
Response, Gain of –1
Figure 20. Normalized Output
Amplitude vs. Frequency vs. Load
OLE
TE
Figure 21. Phase Shift vs. Frequency
Figure 22. Total Harmonic Distortion
vs. Frequency
Figure 24. 3 dB Bandwidth vs.
Supply Voltage
Figure 25. Output Impedance
vs. Frequency
–6–
Figure 23. Settling Time
vs. Step Size
Figure 26. –3 dB Bandwidth
vs. Temperature
REV. B
Typical Characteristics, Inverting Gain of 10–AD846
OBS
Figure 27a. Inverting Amplifier,
Gain of 10
Figure 27b. Large Signal Pulse
Response, Gain of 10
Figure 28. Normalized Output
Amplitude vs. Frequency vs. Load
OLE
TE
Figure 29. Phase vs. Frequency
vs. Load
Figure 30. Harmonic Distortion
vs. Frequency
Figure 32. 3 dB Bandwidth vs.
Supply Voltage
Figure 33. Output Impedance
vs. Frequency
REV. B
–7–
Figure 31. Settling Time vs.
Step Size
Figure 34. –3 dB Bandwidth
vs. Temperature
AD846
POWER SUPPLY CONSIDERATIONS
The power supply connections to the AD846 must maintain a
low impedance to ground over a bandwidth of 40 MHz or more.
This is especially important when driving a significant resistive
or capacitive load, since all current delivered to the load comes
from the power supplies. Multiple high quality bypass capacitors
are recommended for each power supply line in any critical
application. A 0.1 µF ceramic and a 2.2 µF electrolytic capacitor
as shown in Figure 35 placed as close as possible to the amplifier
(with short lead lengths to power supply common) will assure
adequate high frequency bypassing, in most applications. A
minimum bypass capacitance of 0.1 µF should be used for any
application.
OBS
OLE
Figure 35. Recommended Power Supply Bypassing
THEORY OF OPERATION
Figure 37. Overload Recovery Test Circuit
The AD846 differs from conventional operational amplifiers in
that it is a transimpedance device rather than a conventional
voltage amplifier. Figure 36 is a simplified schematic of the
AD846. The input stage consists of a pair of transistors, Q1 and
Q2, which are biased by two diode-connected transistors, Q3
and Q4. Transistors Q1 and Q2 have their emitters connected
together, and this common point functions as the inverting input
of the amplifier. Correspondingly, the common connection of
the two biasing diodes acts as the noninverting input.
TE
Figure 38. Overload Recovery Time Photo
Because the input error signal developed is in the form of a current, not a voltage, the AD846 differs from conventional operational amplifiers. This also means that, unlike most operational
amplifiers which rely on negative feedback to produce a “virtual
ground” at the inverting input terminal, this terminal explicitly
has a low impedance.
A unique circuit approach allows the AD846 to realize an openloop transimpedance of close to 200 MΩ. This is nearly three
orders of magnitude greater than that of any other operational
transimpedance amplifier and results in extremely high levels of
dc precision.
As an example, the output voltage gain error is approximately
equal to the value of the feedback resistor divided by the value
of the open-loop transimpedance of the amplifier. That is, when
using a 1 kΩ feedback resistor, this error is one part in 200,000.
For a transimpedance amplifier with 1 MΩ transimpedance, this
error is only one part in 1000; such an amplifier would barely be
able to achieve 10-bit precision.
Figure 39 is a simplified three-terminal model for the AD846.
Figure 40 is a simplified three-terminal model for a conventional
voltage op amp. The action of current feedback serves to modify
the behavior of the amplifier under closed-loop conditions. The
feedback resistor, RF, is somewhat analogous to the input stage
transconductance of a conventional voltage amplifier; and therefore, if the value of RF is held constant, the closed-loop bandwidth also remains virtually constant, independent of
closed-loop voltage gain.
Figure 36. AD846 Simplified Schematic
When operated as a closed-loop amplifier, feedback error current, IIN: flows into the inverting input terminal and is conveyed
via current mirrors (transistors Q5, Q6, Q7, and Q8) to the
compensation capacitor, CCOMP. The voltage developed across
CCOMP is buffered by the output stage, consisting of transistors
Q9–Q12.
–8–
REV. B
AD846
A simple equation can, therefore, be used to determine the
bandwidth of an amplifier employing the AD846 in the inverting
configuration.
3 dB Bandwidth =
23
RF + 0.05 (1 + G )
where: The 3 dB bandwidth is in MHz
G is the closed-loop inverting gain of the AD846
RF is the feedback resistance in kΩ.
Figure 39. AD846 Three-Terminal Model
NOTE: This equation applies only for values of RF between
10 kΩ and 100 kΩ, and for RLOAD greater than 500 Ω. For RF =
1 kΩ the bandwidth should be estimated from Figure 41.
OBS
Figure 41 illustrates the closed-loop voltage gain vs. frequency
of the AD846 for various values of feedback resistor. For comparison purposes, the characteristic of a conventional amplifier
having an 80 MHz unity gain bandwidth is also shown.
OLE
TE
Figure 40. Op Amp Three-Terminal Model
A more detailed examination of the closed-loop transfer function of the AD846 results in the following equation:
Closed-Loop Gain G(s) = 
−RF
RS


 RF 
1+ CCOMP  RF + 1+ R  RIN s 
S



Compare this to the equation for a conventional op amp:
−RF
RS

C
RF  
Closed-Loop Gain G(s) =
COMP 
1+ R  s 
1+ g

M
S 

Figure 41. Closed-Loop Voltage Gain vs. Bandwidth for
Various Values of RF
For the case where RF = 1 kΩ and RS = 100 Ω (closed-loop gain
of –10), the closed-loop bandwidth is approximately 28 MHz. It
should also be noted that the use of a capacitor to shunt RF, a
normal practice for stabilizing conventional op amps, will cause
this amplifier to become unstable because the closed-loop bandwidth will increase beyond the stable operating frequency.
where: CCOMP is the internal compensation capacitor of the amplifier; gM is the input stage transconductance of the amplifier.
In the case of the voltage amplifier, the closed-loop bandwidth
decreases directly with increasing values of (1 + RF/RS), the
closed-loop gain. However, for the transimpedance amplifier,
the situation is different. At low gains, where (1 + RF/RS) RIN is
small compared to RF, the closed-loop bandwidth is controlled
by the internal compensation capacitance of 7 pF and the value
of RF, and not by the closed-loop gain. At higher gains, where (1
+ RF/RS) RIN is much larger than RF, the behavior is that of a
conventional operational amplifier in which the input stage
transconductance is equal to the inverting terminal input impedance of the transimpedance amplifier (RIN = 50 Ω).
REV. B
A similar approach can be taken to calculate the noise performance of the amplifier. A simplified noise model is shown in
Figure 42.
The equivalent mean-square output noise voltage spectral density will equal:
2
 R 
2
2
2
2
VON = ( RF I NN ) + 1+ F  [VN + ( RP INP ) + 4 kT RP ]
 RS 
R

+ 4 kT RF  F +1
R
 S 
–9–
AD846
Where:
RP is the external resistance placed in series with the noninverting input
RF is the feedback resistor
RS is the source resistor
INN is the noise current in the inverting input
INP is the noise current in the noninverting input
VN is the input noise voltage.
(RF = 1 kΩ, RS = 10 Ω) it will be 4 MHz. At gains of 3 or
greater, a small capacitor (2 pF–5 pF) connected across the
feedback resistor will help reduce overshoot; but when operating
at noninverting gains below 3, this same capacitance will cause
instability.
Typical values for these parameters (@ 1 kHz) in pA/√Hz are:
INN = 20, IPN = 6, VN = 2.
Or, referring to the signal input, the equivalent mean-square input voltage noise is:
2
R 
2 
2
= ( RF I NN ) + 1+ S  V N + RP INP
 RF  
OBS
VIN
2
(

R 
+ 4 kT RS 1+ S 
R

F
)
2
+ 4 kT RP 

OLE
Resistor RP is required for both inverting and noninverting (follower) operation, to insure stable operation. The amplifier’s
noninverting input current (flowing through RP of 100 Ω) will
typically add less than 300 µV to the AD846’s input offset voltage. This can be trimmed-out using the optional network shown
in Figure 44. The following table gives recommended values
for RP.
Supply Voltage
Gain (RF/RS)
Recommended
Value for RF
6 V to 15 V
6 V to 15 V
6 V to 15 V
5V
5V
1– 10
10–20
20–200
1– 10
10 – 200
100 Ω
47 Ω
0Ω
47 Ω
0Ω
Figure 43. AD846 Noninverting Amplifier Configuration
USING THE COMPENSATION PIN OF THE AD846
Additional compensation may be provided for the AD846 by
applying an external capacitance between Pin 5 and analog
ground (Figure 44). The nominal value of the AD846’s internal
compensation capacitor is 7 pF. For a given value of feedback
resistance (RF), any added external capacitance reduces the
amplifier’s slew rate and bandwidth proportionally.
TE
Figure 44. AD846 Inverting Amplifier Showing External
Compensation Connection, RP and Optional VOS Trim
Figure 42. Op Amp Simplified Noise Model
NONINVERTING GAIN OPERATION
In addition to providing for external compensation, Pin 5 may
be used to clamp the output of the amplifier, as shown in Figure
45. The output can be clamped anywhere within the output
range (approximately ± 10 V) of the amplifier. The input should
also be clamped as a precaution against damaging the amplifier’s
input transistors.
The AD846 can be used as a noninverting amplifier or voltage
follower, operating at gains between 1 and 200. A minimum
value of RF equal to 1 kΩ should be employed. For low gains (1
to 2), the input signal should be applied to the AD846’s noninverting input through a 100 Ω series resistor; this will help reduce peaking. The best transient response will occur when the
amplifier’s output level is below 5 V peak to peak.
At closed-loop gains of 3 or more, the input resistor is not required unless peak signals greater than 3 V will be applied. The
amplifier’s bandwidth can be determined by using the inverting
amplifier’s bandwidth equation or from Figure 41. For example,
at a gain of + 10 (RF = 1 kΩ, RS = 100 Ω) the bandwidth of the
AD846 will be approximately 33 MHz; at a gain of +100,
–10–
Figure 45. AD846 Used as a Clamped Amplifier
REV. B
AD846
This compensation node may also be used as an additional output terminal as in the precision transconductance amplifier application of Figure 46.
THE AD846 AS AN OPEN-LOOP LEVEL SHIFTER
The AD846 can also be used for open-loop level shifting. As
shown in Figure 48, resistor R S is used to develop an input current which is proportional to the input voltage, VIN. This current
flows from the compensation node (Pin 5) developing a voltage
across resistor RC (RC is equal in value to resistor RS) which,
rather than being grounded, has one end tied to reference voltage V2. The voltage appearing at Pin 5 is, therefore, voltage VIN
plus voltage V2 and will directly follow changes in VIN. By scaling resistor RC, a level shift with voltage gain can be produced.
In addition, the normal voltage output at Pin 6 is approximately
equal to the voltage at Pin 5 thus providing a low impedance,
buffered output for the level shifter.
Figure 46. A Precision Transconductance Amplifier
OBS
The AD846 can be used in either the inverting transconductance mode as shown in Figure 46, or in a noninverting mode
with RS grounded and VIN applied to the noninverting terminal.
The current output is essentially constant over a compliance
range of ± 10 V at the compensation node. The output current
(from Pin 5) is limited to about ± 1 mA due to internal saturation. Under these circumstances the normal output pin provides
a buffered version of the compensation node output voltage.
Output load impedance of 500 Ω or greater will not affect the
accuracy of the transconductance conversion.
OLE
The combination of fast settling times at high gains and low dc
errors make the AD846 ideal for use as an error amplifier in
high speed, 12-bit subranging A-D applications. In the circuit of
Figure 47, an AD842 serves as an input amplifier. First pass
conversion is accomplished, in a straightforward manner, determining the top 7 bits. The latch then holds these top 7 bits
which are applied to a 7 bit, 12-bit accurate DAC and also to
the highest 7 bits of the adder (note that a sample-and-hold
should be used ahead of this converter to minimize errors due to
its 500 ns acquisition time). In the second pass, the input
switches S1 and S2 and S3 are set to state 2. The DAC output
is then subtracted from the input signal and the resulting difference is then amplified by an AD846 gain of 32 follower. This
gain, together with a 1/64th scale offset, insures a unipolar residue which can be converted by the flash A-D. Conversion is
accomplished via switches S1, S2 and S3 in state 1. Switch S1
connects the input signal of the AD846 residue amplifier to
ground which minimized overload recovery time.
THE AD846 AS A HIGH SPEED DAC BUFFER
The AD846 will enable the AD568 12-bit DAC to develop a
10 V output step which settles to within 0.025 percent of its
final value in about 100 ns. This AD846/AD568 combination is
shown in the circuit of Figure 49. Correct power supply decoupling is essential: a 2.2 µF tantalum capacitor connected in parallel with a 0.1 µF to 0.01 µF ceramic disc capacitor is usually
sufficient. These should be placed as close to the power supply
pins as possible. Also, a ground plane should be employed; this
ensure that there is a low impedance signal path to ground
which allows the fastest possible output settling. In 12-bit systems with the AD846 operating at gains of 10 or less, inadequate supply decoupling can cause the output settling to
degrade from 100 ns to as much as 300 ns, with a 10 V output
step applied.
Figure 47. Block Diagram of a 2 MHz, 12-Bit Subranging
A/D Converter
REV. B
TE
Figure 48. AD846 Connected as a Level Shift Amplifier
THE AD846 IN A 2 MHz, 12-BIT SUBRANGING A/D
CONVERTER CIRCUIT
Figure 49. The AD846 Serving as a DAC Buffer
–11–
AD846
OUTLINE DIMENSIONS
Dimensions shown in inches and (mm).
C1155–10–9/88
Plastic
Mini-DIP (N)
Package
OBS
OLE
TE
PRINTED IN U.S.A.
Cerdip (Q) Package
–12–
REV. B