LT8301 42VIN Micropower No-Opto Isolated Flyback Converter with 65V/1.2A Switch FEATURES DESCRIPTION 2.7V to 42V Input Voltage Range n 1.2A, 65V Internal DMOS Power Switch n Low Quiescent Current: 100µA in Sleep Mode 350µA in Active Mode n Boundary Mode Operation at Heavy Load n Low-Ripple Burst Mode® Operation at Light Load n Minimum Load <0.5% (Typ) of Full Output n V OUT Set with a Single External Resistor n No Transformer Third Winding or Opto-Isolator Required for Regulation n Accurate EN/UVLO Threshold and Hysteresis n Internal Compensation and Soft-Start n Output Short-Circuit Protection n 5-Lead TSOT-23 Package The LT®8301 is a micropower isolated flyback converter. By sampling the isolated output voltage directly from the primary-side flyback waveform, the part requires no third winding or opto-isolator for regulation. The output voltage is programmed with a single external resistor. Internal compensation and soft-start further reduce external component count. Boundary mode operation provides a small magnetic solution with excellent load regulation. Low ripple Burst Mode operation maintains high efficiency at light load while minimizing the output voltage ripple. A 1.2A, 65V DMOS power switch is integrated along with all high voltage circuitry and control logic into a 5-lead ThinSOT™ package. n The LT8301 operates from an input voltage range of 2.7V to 42V and can deliver up to 6W of isolated output power. The high level of integration and the use of boundary and low ripple burst modes result in a simple to use, low component count, and high efficiency application solution for isolated power delivery. APPLICATIONS Isolated Telecom, Automotive, Industrial, Medical Power Supplies n Isolated Auxiliary/Housekeeping Power Supplies n L, LT, LTC, LTM, Linear Technology, the Linear logo and Burst Mode are registered trademarks and ThinSOT is a trademark of Linear Technology Corporation. All other trademarks are the property of their respective owners. Protected by U.S. Patents, including 5438499, 7463497, and 7471522. TYPICAL APPLICATION Efficiency vs Load Current 2.7V to 36VIN/5VOUT Micropower Isolated Flyback Converter 3:1 • 10µF 40µH VIN EN/UVLO LT8301 GND • SW 154k 4.4µH VOUT+ 5V 6mA TO 0.40A (VIN = 5V) 100µF 6mA TO 0.70A (VIN = 12V) 6mA TO 1.00A (VIN = 24V) 6mA TO 1.15A (VIN = 36V) 8301 TA01a VOUT– RFB 85 EFFICIENCY (%) VIN 2.7V TO 36V 90 80 75 70 VIN = 5V VIN = 12V VIN = 24V VIN = 36V 65 60 0 0.2 0.4 0.6 0.8 LOAD CURRENT (A) 1.0 1.2 8301 TA01b 8301f For more information www.linear.com/LT8301 1 LT8301 ABSOLUTE MAXIMUM RATINGS PIN CONFIGURATION (Note 1) TOP VIEW SW (Note 2).............................................................. 65V VIN............................................................................ 42V EN/UVLO.................................................................... VIN RFB....................................................... VIN – 0.5V to VIN Current into RFB.................................................... 200µA Operating Junction Temperature Range (Notes 3, 4) LT8301E, LT8301I............................... –40°C to 125°C LT8301H............................................. –40°C to 150°C LT8301MP.......................................... –55°C to 150°C Storage Temperature Range................... –65°C to 150°C EN/UVLO 1 5 VIN GND 2 RFB 3 4 SW S5 PACKAGE 5-LEAD PLASTIC TSOT-23 θJA = 150°C/W ORDER INFORMATION LEAD FREE FINISH TAPE AND REEL PART MARKING* PACKAGE DESCRIPTION TEMPERATURE RANGE LT8301ES5#PBF LT8301ES5#TRPBF LTGMF 5-Lead Plastic TSOT-23 –40°C to 125°C LT8301IS5#PBF LT8301IS5#TRPBF LTGMF 5-Lead Plastic TSOT-23 –40°C to 125°C LT8301HS5#PBF LT8301HS5#TRPBF LTGMF 5-Lead Plastic TSOT-23 –40°C to 150°C LT8301MPS5#PBF LT8301MPS5#TRPBF LTGMF 5-Lead Plastic TSOT-23 –55°C to 150°C Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container. Consult LTC Marketing for information on non-standard lead based finish parts. For more information on lead free part marking, go to: http://www.linear.com/leadfree/ For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/ 8301f 2 For more information www.linear.com/LT8301 LT8301 ELECTRICAL CHARACTERISTICS The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN = 5V, VEN/UVLO = VIN unless otherwise noted. SYMBOL PARAMETER VIN Input Voltage Range CONDITIONS MIN l TYP 2.7 MAX UNIT 42 V VIN UVLO Threshold Rising Falling 2.5 2.3 2.65 V V VIN Quiescent Current VEN/UVLO = 0.2V VEN/UVLO = 1.1V Sleep Mode (Switch Off) Active Mode (Switch On) 0.8 215 100 350 2 µA µA µA µA EN/UVLO Shutdown Threshold For Lowest Off IQ EN/UVLO Enable Threshold IHYS EN/UVLO Hysteresis Current fMIN Minimum Switching Frequency tON(MIN) Minimum Switch-On Time tOFF(MAX) Maximum Switch-Off Time ISW(MAX) Maximum SW Current Limit l 1.200 1.375 1.550 A ISW(MIN) Minimum SW Current Limit l 0.22 0.29 0.36 A µA IQ 0.2 0.55 Falling Hysteresis 1.204 1.228 0.014 1.248 V V VEN/UVLO = 0.2V VEN/UVLO = 1.1V VEN/UVLO = 1.3V –0.1 2.2 –0.1 0 2.5 0 0.1 2.8 0.1 µA µA µA 10 10.6 kHz l 9.4 V 170 Backup Timer ns 190 µs RDS(ON) Switch On-Resistance ISW = 500mA 0.4 ILKG Switch Leakage Current VIN = 42V, VSW = 65V 0.1 0.5 IRFB RFB Regulation Current 100 102.5 µA 0.02 0.1 %/V l RFB Regulation Current Line Regulation 2.7V ≤ VIN ≤ 42V Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Note 2: The SW pin is rated to 65V for transients. Depending on the leakage inductance voltage spike, operating waveforms of the SW pin should be derated to keep the flyback voltage spike below 65V as shown in Figure 5. Note 3: The LT8301E is guaranteed to meet performance specifications from 0°C to 125°C operating junction temperature. Specifications over the –40°C to 125°C operating junction temperature range are assured by design, characterization and correlation with statistical process controls. 97.5 Ω The LT8301I is guaranteed over the full –40°C to 125°C operating junction temperature range. The LT8301H is guaranteed over the full –40°C to 150°C operating junction temperature range. The LT8301MP is guaranteed over the full –55°C to 150°C operating junction temperature range. High junction temperatures degrade operating lifetimes. Operating lifetime is derated at junction temperature greater than 125°C. Note 4: The LT8301 includes overtemperature protection that is intended to protect the device during momentary overload conditions. Junction temperature will exceed 150°C when overtemperature protection is active. Continuous operation above the specified maximum operating junction temperature may impair device reliability. 8301f For more information www.linear.com/LT8301 3 LT8301 TYPICAL PERFORMANCE CHARACTERISTICS Output Load and Line Regulation 5.20 6 FRONT PAGE APPLICATION 5.05 5.00 4.95 4.90 VIN = 5V VIN = 12V VIN = 24V VIN = 36V 4.85 0.2 0 0.4 0.6 0.8 1.0 4 3 2 VIN = 5V VIN = 12V VIN = 24V VIN = 36V 1 0 1.2 SWITCHING FREQUENCY (kHz) OUTPUT VOLTAGE (V) OUTPUT VOLTAGE (V) 350 FRONT PAGE APPLICATION 5 5.10 0 0.2 LOAD CURRENT (A) 0.4 0.6 0.8 1.0 1.2 LOAD CURRENT (A) Boundary Mode Waveforms VOUT 50mV/DIV VSW 20V/DIV VSW 20V/DIV 8301 G04 VIN = 5V VIN = 12V VIN = 24V VIN = 36V 50 0 0.8 0.6 0.4 LOAD CURRENT (A) 0.2 1.2 1.0 8301 G03 Burst Mode Waveforms VSW 20V/DIV 8301 G05 8301 G06 20µs/DIV FRONT PAGE APPLICATION VIN = 12V ILOAD = 6mA VIN Quiescent Current, Active Mode 140 400 130 380 120 360 4 IQ (µA) 110 6 IQ (µA) IQ (µA) 100 VIN Quiescent Current, Sleep Mode TJ = 150°C TJ = 25°C TJ = –55°C 8 150 VOUT 50mV/DIV 5µs/DIV FRONT PAGE APPLICATION VIN = 12V ILOAD = 200mA VIN Shutdown Current 10 200 Discontinuous Mode Waveforms VOUT 50mV/DIV 5µs/DIV FRONT PAGE APPLICATION VIN = 12V ILOAD = 600mA 250 0 1.6 1.4 FRONT PAGE APPLICATION 300 8301 G02 8301 G01 100 90 340 320 80 2 TJ = 150°C TJ = 25°C TJ = –55°C 70 0 Switching Frequency vs Load Current Output Short-Circuit Protection 5.15 4.80 TA = 25°C, unless otherwise noted. 0 5 10 15 20 25 VIN (V) 30 35 40 45 60 0 5 10 15 20 25 30 35 40 45 VIN (V) 8301 G07 TJ = 150°C TJ = 25°C TJ = –55°C 300 280 0 5 10 15 20 25 30 35 40 45 VIN (V) 8301 G08 8301 G09 8301f 4 For more information www.linear.com/LT8301 LT8301 TYPICAL PERFORMANCE CHARACTERISTICS EN/UVLO Enable Threshold TA = 25°C, unless otherwise noted. EN/UVLO Hysteresis Current 1.245 105 5 104 1.240 102 1.225 1.220 3 IRFB (µA) 1.230 IHYS (µA) VEN/UVLO (V) 103 4 1.235 2 100 99 97 1 1.210 96 1.205 –50 –25 0 0 –50 –25 25 50 75 100 125 150 TEMPERATURE (°C) 0 8301 G10 95 –50 –25 25 50 75 100 125 150 TEMPERATURE (°C) 0 25 50 75 100 125 150 TEMPERATURE (°C) 8301 G12 8301 G11 RDS(ON) Switch Current Limit 1000 Maximum Switching Frequency 1.6 600 MAXIMUM CURRENT LIMIT 1.4 800 500 FREQUENCY (kHz) 1.2 1.0 600 ISW (A) RESISTANCE (mΩ) 101 98 1.215 400 0.8 0.6 0.4 200 MINIMUM CURRENT LIMIT 0 –50 –25 0 0 –50 –25 25 50 75 100 125 150 TEMPERATURE (°C) 0 Minimum Switching Frequency 0 –50 –25 500 400 400 300 300 TIME (ns) TIME (ns) 25 50 75 100 125 150 TEMPERATURE (°C) 25 50 75 100 125 150 TEMPERATURE (°C) Minimum Switch-Off Time 500 200 100 0 0 8301 G15 Minimum Switch-On Time 15 5 200 8301 G14 20 10 300 0 –50 –25 25 50 75 100 125 150 TEMPERATURE (°C) 8301 G13 400 100 0.2 FREQUENCY (kHz) RFB Regulation Current 0 –50 –25 200 100 0 25 50 75 100 125 150 TEMPERATURE (°C) 8301 G17 0 –50 –25 0 25 50 75 100 125 150 TEMPERATURE (°C) 8301 G18 8301 G16 8301f For more information www.linear.com/LT8301 5 LT8301 PIN FUNCTIONS EN/UVLO (Pin 1): Enable/Undervoltage Lockout. The EN/UVLO pin is used to enable the LT8301. Pull the pin below 0.2V to shut down the LT8301. This pin has an accurate 1.228V threshold and can be used to program a VIN undervoltage lockout (UVLO) threshold using a resistor divider from VIN to ground. A 2.5µA current hysteresis allows the programming of VIN UVLO hysteresis. If neither function is used, tie this pin directly to VIN. GND (Pin 2): Ground. Tie this pin directly to local ground plane. RFB (Pin 3): Input Pin for External Feedback Resistor. Connect a resistor from this pin to the transformer primary SW pin. The ratio of the RFB resistor to an internal 10k resistor, times a trimmed 1.0V reference voltage, determines the output voltage (plus the effect of any non-unity transformer turns ratio). Minimize trace area at this pin. SW (Pin 4): Drain of the 65V Internal DMOS Power Switch. Minimize trace area at this pin to reduce EMI and voltage spikes. VIN (Pin 5): Input Supply. The VIN pin supplies current to internal circuitry and serves as a reference voltage for the feedback circuitry connected to the RFB pin. Locally bypass this pin to ground with a capacitor. 8301f 6 For more information www.linear.com/LT8301 LT8301 BLOCK DIAGRAM T1 NPS:1 VIN CIN LPRI • • DOUT VOUT+ LSEC COUT RFB 5 3 VIN VOUT– 4 RFB SW BOUNDARY DETECTOR 1:4 M3 M2 OSCILLATOR – 25µA RREF 10kΩ 1.0V + – gm + S A3 R Q DRIVER M1 R1 1 – EN/UVLO 2.5µA R2 1.228V M4 + + RSENSE A2 A1 – VIN GND REFERENCE REGULATORS 2 8301 BD OPERATION The LT8301 is a current mode switching regulator IC designed specially for the isolated flyback topology. The key problem in isolated topologies is how to communicate the output voltage information from the isolated secondary side of the transformer to the primary side for regulation. Historically, opto-isolators or extra transformer windings communicate this information across the isolation boundary. Opto-isolator circuits waste output power, and the extra components increase the cost and physical size of the power supply. Opto-isolators can also cause system issues due to limited dynamic response, nonlinearity, unitto-unit variation and aging over lifetime. Circuits employing extra transformer windings also exhibit deficiencies, as using an extra winding adds to the transformer’s physical size and cost, and dynamic response is often mediocre. The LT8301 samples the isolated output voltage through the primary-side flyback pulse waveform. In this manner, neither opto-isolator nor extra transformer winding is required for regulation. Since the LT8301 operates in either boundary conduction mode or discontinuous conduction mode, the output voltage is always sampled on the SW pin when the secondary current is zero. This method improves load regulation without the need of external load compensation components. 8301f For more information www.linear.com/LT8301 7 LT8301 OPERATION The LT8301 is a simple to use micropower isolated flyback converter housed in a 5-lead TSOT-23 package. The output voltage is programmed with a single external resistor. By integrating the loop compensation and soft-start inside, the part further reduces the number of external components. As shown in the Block Diagram, many of the blocks are similar to those found in traditional switching regulators including reference, regulators, oscillator, logic, current amplifier, current comparator, driver, and power switch. The novel sections include a flyback pulse sense circuit, a sample-and-hold error amplifier, and a boundary mode detector, as well as the additional logic for boundary conduction mode, discontinuous conduction mode, and low ripple Burst Mode operation. Boundary Conduction Mode Operation The LT8301 features boundary conduction mode operation at heavy load, where the chip turns on the primary power switch when the secondary current is zero. Boundary conduction mode is a variable frequency, variable peakcurrent switching scheme. The power switch turns on and the transformer primary current increases until an internally controlled peak current limit. After the power switch turns off, the voltage on the SW pin rises to the output voltage multiplied by the primary-to-secondary transformer turns ratio plus the input voltage. When the secondary current through the output diode falls to zero, the SW pin voltage collapses and rings around VIN. A boundary mode detector senses this event and turns the power switch back on. Boundary conduction mode returns the secondary current to zero every cycle, so parasitic resistive voltage drops do not cause load regulation errors. Boundary conduction mode also allows the use of smaller transformers compared to continuous conduction mode and does not exhibit sub-harmonic oscillation. Discontinuous Conduction Mode Operation As the load gets lighter, boundary conduction mode increases the switching frequency and decreases the switch peak current at the same ratio. Running at a higher switching frequency up to several MHz increases switching and gate charge losses. To avoid this scenario, the LT8301 has an additional internal oscillator, which clamps the maximum switching frequency to be less than 430kHz (typ). Once the switching frequency hits the internal frequency clamp, the part starts to delay the switch turn-on and operates in discontinuous conduction mode. Low Ripple Burst Mode Operation Unlike traditional flyback converters, the LT8301 has to turn on and off at least for a minimum amount of time and with a minimum frequency to allow accurate sampling of the output voltage. The inherent minimum switch current limit and minimum switch-off time are necessary to guarantee the correct operation of specific applications. As the load gets very light, the LT8301 starts to fold back the switching frequency while keeping the minimum switch current limit. So the load current is able to decrease while still allowing minimum switch-off time for the sampleand-hold error amplifier. Meanwhile, the part switches between sleep mode and active mode, thereby reducing the effective quiescent current to improve light load efficiency. In this condition, the LT8301 operates in low ripple Burst Mode. The 10kHz (typ) minimum switching frequency determines how often the output voltage is sampled and also the minimum load requirement. 8301f 8 For more information www.linear.com/LT8301 LT8301 APPLICATIONS INFORMATION Output Voltage The RFB resistor as depicted in the Block Diagram is the only external resistor used to program the output voltage. The LT8301 operates similar to traditional current mode switchers, except in the use of a unique flyback pulse sense circuit and a sample-and-hold error amplifier, which sample and therefore regulate the isolated output voltage from the flyback pulse. Operation is as follows: when the power switch M1 turns off, the SW pin voltage rises above the VIN supply. The amplitude of the flyback pulse, i.e., the difference between the SW pin voltage and VIN supply, is given as: VFLBK = (VOUT + VF + ISEC • ESR) • NPS VF = Output diode forward voltage ISEC = Transformer secondary current ESR = Total impedance of secondary circuit NPS =Transformer effective primary-to-secondary turns ratio The flyback voltage is then converted to a current IRFB by the flyback pulse sense circuit (M2 and M3). This current IRFB also flows through the internal 10k RREF resistor to generate a ground-referred voltage. The resulting voltage feeds to the inverting input of the sample-and-hold error amplifier. Since the sample-and-hold error amplifier samples the voltage when the secondary current is zero, the (ISEC • ESR) term in the VFLBK equation can be assumed to be zero. An internal trimmed reference voltage,VIREF 1.0V, feeds to the non-inverting input of the sample-and-hold error amplifier. The relatively high gain in the overall loop causes the voltage across RREF resistor to be nearly equal to VIREF. The resulting relationship between VFLBK and VIREF can be expressed as: VFLBK R •RREF = VIREF FB or V VFLBK = IREF •RFB =IRFB •RFB RREF VIREF = Internal trimmed reference voltage IRFB = RFB regulation current = 100µA Combination with the previous VFLBK equation yields an equation for VOUT, in terms of the RFB resistor, transformer turns ratio, and diode forward voltage: R VOUT = 100µA • FB − VF NPS Output Temperature Coefficient The first term in the VOUT equation does not have temperature dependence, but the output diode forward voltage VF has a significant negative temperature coefficient (–1mV/°C to –2mV/°C). Such a negative temperature coefficient produces approximately 200mV to 300mV voltage variation on the output voltage across temperature. For higher voltage outputs, such as 12V and 24V, the output diode temperature coefficient has a negligible effect on the output voltage regulation. For lower voltage outputs, such as 3.3V and 5V, however, the output diode temperature coefficient does count for an extra 2% to 5% output voltage regulation. For customers requiring tight output voltage regulation across temperature, please refer to other LTC parts with integrated temperature compensation features. 8301f For more information www.linear.com/LT8301 9 LT8301 APPLICATIONS INFORMATION Selecting Actual RFB Resistor Value Output Power The LT8301 uses a unique sampling scheme to regulate the isolated output voltage. Due to the sampling nature, the scheme contains repeatable delays and error sources, which will affect the output voltage and force a re-evaluation of the RFB resistor value. Therefore, a simple two-step process is required to choose feedback resistor RFB. A flyback converter has a complicated relationship between the input and output currents compared to a buck or a boost converter. A boost converter has a relatively constant maximum input current regardless of input voltage and a buck converter has a relatively constant maximum output current regardless of input voltage. This is due to the continuous non-switching behavior of the two currents. A flyback converter has both discontinuous input and output currents which make it similar to a non-isolated buck-boost converter. The duty cycle will affect the input and output currents, making it hard to predict output power. In addition, the winding ratio can be changed to multiply the output current at the expense of a higher switch voltage. Rearrangement of the expression for VOUT in the Output Voltage section yields the starting value for RFB: RFB = ( N PS • VOUT + VF 100µA ) VOUT = Output voltage VF = Output diode forward voltage = ~0.3V NPS =Transformer effective primary-to-secondary turns ratio Power up the application with the starting RFB value and other components connected, and measure the regulated output voltage, VOUT(MEAS). The final RFB value can be adjusted to: VOUT RFB(FINAL) = •R VOUT(MEAS) FB Once the final RFB value is selected, the regulation accuracy from board to board for a given application will be very consistent, typically under ±5% when including device variation of all the components in the system (assuming resistor tolerances and transformer windings matching within ±1%). However, if the transformer or the output diode is changed, or the layout is dramatically altered, there may be some change in VOUT. The graphs in Figures 1 to 4 show the typical maximum output power possible for the output voltages 3.3V, 5V, 12V, and 24V. The maximum output power curve is the calculated output power if the switch voltage is 50V during the switch-off time. 15V of margin is left for leakage inductance voltage spike. To achieve this power level at a given input, a winding ratio value must be calculated to stress the switch to 50V, resulting in some odd ratio values. The curves below the maximum output power curve are examples of common winding ratio values and the amount of output power at given input voltages. One design example would be a 5V output converter with a minimum input voltage of 8V and a maximum input voltage of 32V. A three-to-one winding ratio fits this design example perfectly and outputs equal to 5.42W at 32V but lowers to 2.71W at 8V. The following equations calculate output power: POUT = η• VIN •D•I SW(MAX) • 0.5 η = Efficiency = 85% ( VOUT + VF ) •NPS D = DutyCycle = ( VOUT + VF ) •NPS + VIN ISW(MAX) = Maximum switch current limit = 1.2A (min) 8301f 10 For more information www.linear.com/LT8301 LT8301 APPLICATIONS INFORMATION 7 7 MAXIMUM OUTPUT CURRENT 5 N = 3:1 4 N = 2:1 3 2 N = 1:1 1 0 MAXIMUM OUTPUT CURRENT 6 N = 5:1 OUTPUT POWER (W) OUTPUT POWER (W) 6 5 N = 4:1 N = 3:1 4 N = 2:1 3 N = 1:1 2 1 0 10 20 0 40 30 0 10 INPUT VOLTAGE (V) 20 30 8301 F01 8301 F02 Figure 1. Output Power for 3.3V Output Figure 2. Output Power for 5V Output 7 7 MAXIMUM OUTPUT CURRENT 5 N = 1:1 4 N = 2:3 3 N = 1:3 2 MAXIMUM OUTPUT CURRENT 6 N = 3:2 OUTPUT POWER (W) OUTPUT POWER (W) 6 1 0 5 N = 4:5 N = 1:2 4 N = 1:3 3 N = 1:5 2 1 0 10 20 0 40 30 0 INPUT VOLTAGE (V) 10 20 Primary Inductance Requirement The LT8301 obtains output voltage information from the reflected output voltage on the SW pin. The conduction of secondary current reflects the output voltage on the primary SW pin. The sample-and-hold error amplifier needs a minimum 450ns to settle and sample the reflected output voltage. In order to ensure proper sampling, the secondary winding needs to conduct current for a minimum of 450ns. The following equation gives the minimum value for primary-side magnetizing inductance: ( I SW(MIN) 40 8301 F04 Figure 3. Output Power for 12V Output tOFF(MIN) •N PS • VOUT + VF 30 INPUT VOLTAGE (V) 8301 F03 LPRI ≥ 40 INPUT VOLTAGE (V) ) Figure 4. Output Power for 24V Output In addition to the primary inductance requirement for the minimum switch-off time, the LT8301 has minimum switch-on time that prevents the chip from turning on the power switch shorter than approximately 170ns. This minimum switch-on time is mainly for leading-edge blanking the initial switch turn-on current spike. If the inductor current exceeds the desired current limit during that time, oscillation may occur at the output as the current control loop will lose its ability to regulate. Therefore, the following equation relating to maximum input voltage must also be followed in selecting primary-side magnetizing inductance: tOFF(MIN) = Minimum switch-off time = 450ns ISW(MIN) = Minimum switch current limit = 290mA (typ) LPRI ≥ tON(MIN) • VIN(MAX) I SW(MIN) tON(MIN) = Minimum switch-on time = 170ns For more information www.linear.com/LT8301 8301f 11 LT8301 APPLICATIONS INFORMATION In general, choose a transformer with its primary magnetizing inductance about 30% larger than the minimum values calculated above. A transformer with much larger inductance will have a bigger physical size and may cause instability at light load. Linear Technology has worked with several leading magnetic component manufacturers to produce pre-designed flyback transformers for use with the LT8301. Table 1 shows the details of these transformers. Selecting a Transformer Note that when choosing the RFB resistor to set output voltage, the user has relative freedom in selecting a transformer turns ratio to suit a given application. In contrast, the use of simple ratios of small integers, e.g., 3:1, 2:1, 1:1, provides more freedom in settling total turns and mutual inductance. Transformer specification and design is perhaps the most critical part of successfully applying the LT8301. In addition to the usual list of guidelines dealing with high frequency isolated power supply transformer design, the following information should be carefully considered. Turns Ratio Table 1. Predesigned Transformers—Typical Specifications TRANSFORMER PART NUMBER 750313973 DIMENSIONS (W × L × H) (mm) LPRI (µH) LLKG (µH) NP:NS RPRI (mΩ) RSEC (mΩ) TARGET APPLICATIONS VENDOR VIN (V) VOUT (V) IOUT (A) 15.24 × 13.34 × 11.43 40 1 4:1 80 40 Würth Electronik 8 to 36 3.3 0.80 750370047 13.35 × 10.8 × 9.14 30 1 3:1:1 60 12.5 Würth Electronik 8 to 32 5 0.55 750313974 15.24 × 13.34 × 11.43 40 1 3:1 80 50 Würth Electronik 8 to 36 5 0.55 750313970 15.24 × 13.34 × 11.43 40 1 2:1 80 70 Würth Electronik 18 to 42 3.3 0.75 750310799 9.14 × 9.78 × 10.54 25 0.125 1:1:0.33 60 74 Würth Electronik 8 to 30 12 0.22 750313972 15.24 × 13.34 × 11.43 40 1 1:1 80 185 Würth Electronik 18 to 42 5 0.42 750313975 15.24 × 13.34 × 11.43 40 1 1:2 110 865 Würth Electronik 8 to 36 24 0.12 750313976 15.24 × 13.34 × 11.43 40 1 1:4 110 2300 Würth Electronik 8 to 32 48 0.05 12387-T036 15.5 × 12.5 × 11.5 40 2 4:1 160 25 Sumida 8 to 36 3.3 0.80 12387-T037 15.5 × 12.5 × 11.5 40 2 3:1 210 30 Sumida 8 to 36 5 0.55 12387-T040 15.5 × 12.5 × 11.5 40 1.5 2:1 210 50 Sumida 18 to 42 3.3 0.75 12387-T041 15.5 × 12.5 × 11.5 40 1.5 1:1 210 200 Sumida 18 to 42 5 0.42 12387-T038 15.5 × 12.5 × 11.5 40 2 1:2 220 460 Sumida 8 to 36 24 0.12 15.5 × 12.5 × 11.5 40 2 1:4 220 2200 Sumida 8 to 32 48 0.05 PA3948.003NL 12387-T039 15.24 × 13.08 × 11.45 40 1.45 4:1 210 26 Pulse Engineering 8 to 36 3.3 0.80 PA3948.004NL 15.24 × 13.08 × 11.45 40 1.95 3:1 220 29 Pulse Engineering 8 to 36 5 0.55 PA3948.001NL 15.24 × 13.08 × 11.45 40 1.45 2:1 410 70 Pulse Engineering 18 to 42 3.3 0.75 PA3948.002NL 15.24 × 13.08 × 11.45 40 1.45 1:1 405 235 Pulse Engineering 18 to 42 5 0.42 PA3948.005NL 15.24 × 13.08 × 11.45 40 1.60 1:2 220 1275 Pulse Engineering 8 to 36 24 0.12 PA3948.006NL 15.24 × 13.08 × 11.45 40 1.65 1:4 220 3350 Pulse Engineering 8 to 32 48 0.05 8301f 12 For more information www.linear.com/LT8301 LT8301 APPLICATIONS INFORMATION Typically, choose the transformer turns ratio to maximize available output power. For low output voltages (3.3V or 5V), a larger N:1 turns ratio can be used with multiple primary windings relative to the secondary to maximize the transformer’s current gain (and output power). However, remember that the SW pin sees a voltage that is equal to the maximum input supply voltage plus the output voltage multiplied by the turns ratio. In addition, leakage inductance will cause a voltage spike (VLEAKAGE) on top of this reflected voltage. This total quantity needs to remain below the 65V absolute maximum rating of the SW pin to prevent breakdown of the internal power switch. Together these conditions place an upper limit on the turns ratio, NPS, for a given application. Choose a turns ratio low enough to ensure: NPS < 65V − VIN(MAX) − VLEAKAGE Saturation Current The current in the transformer windings should not exceed its rated saturation current. Energy injected once the core is saturated will not be transferred to the secondary and will instead be dissipated in the core. When designing custom transformers to be used with the LT8301, the saturation current should always be specified by the transformer manufacturers. Winding Resistance Resistance in either the primary or secondary windings will reduce overall power efficiency. Good output voltage regulation will be maintained independent of winding resistance due to the boundary/discontinuous conduction mode operation of the LT8301. Leakage Inductance and Snubbers VOUT + VF For lower output power levels, choose a smaller N:1 turns ratio to alleviate the SW pin voltage stress. Although a 1:N turns ratio makes it possible to have very high output voltages without exceeding the breakdown voltage of the internal power switch, the multiplied parasitic capacitance through turns ratio may cause the switch turn-on current spike ringing beyond 170ns leading-edge blanking, thereby producing light load instability in certain applications. So any 1:N turns ratio should be fully evaluated before its use with the LT8301. The turns ratio is an important element in the isolated feedback scheme, and directly affects the output voltage accuracy. Make sure the transformer manufacturer specifies turns ratio accuracy within ±1%. Transformer leakage inductance on either the primary or secondary causes a voltage spike to appear on the primary after the power switch turns off. This spike is increasingly prominent at higher load currents where more stored energy must be dissipated. It is very important to minimize transformer leakage inductance. When designing an application, adequate margin should be kept for the worst-case leakage voltage spikes even under overload conditions. In most cases shown in Figure 5, the reflected output voltage on the primary plus VIN should be kept below 50V. This leaves at least 15V margin for the leakage spike across line and load conditions. A larger voltage margin will be required for poorly wound transformers or for excessive leakage inductance. In addition to the voltage spikes, the leakage inductance also causes the SW pin ringing for a while after the power switch turns off. To prevent the voltage ringing falsely triggering the boundary mode detector, the LT8301 internally blanks the boundary mode detector for approximately 350ns. Any remaining voltage ringing after 350ns may turn the power switch back on again before the secondary current falls to zero. So the leakage inductance spike ringing should be limited to less than 350ns. 8301f For more information www.linear.com/LT8301 13 LT8301 APPLICATIONS INFORMATION VSW VSW <65V VSW <65V <65V VLEAKAGE VLEAKAGE <50V VLEAKAGE <50V <50V tOFF > 450ns tOFF > 450ns tOFF > 450ns tSP < 350ns tSP < 350ns tSP < 350ns TIME TIME No Snubber TIME with DZ Snubber with RC Snubber 8301 F05 Figure 5. Maximum Voltages for SW Pin Flyback Waveform Lℓ Lℓ • Z D • C • • R 8300 F06a 8301 F06b DZ Snubber RC Snubber Figure 6. Snubber Circuits A snubber circuit is recommended for most applications. Two types of snubber circuits shown in Figure 6 that can protect the internal power switch include the DZ (diodeZener) snubber and the RC (resistor-capacitor) snubber. The DZ snubber ensures well defined and consistent clamping voltage and has slightly higher power efficiency, while the RC snubber quickly damps the voltage spike ringing and provides better load regulation and EMI performance. Figure 5 shows the flyback waveforms with the DZ and RC snubbers. For the DZ snubber, proper care must be taken when choosing both the diode and the Zener diode. Schottky diodes are typically the best choice, but some PN diodes can be used if they turn on fast enough to limit the leakage inductance spike. Choose a diode that has a reversevoltage rating higher than the maximum SW pin voltage. The Zener diode breakdown voltage should be chosen to balance power loss and switch voltage protection. The best compromise is to choose the largest voltage breakdown. Use the following equation to make the proper choice: VZENER(MAX) ≤ 65V – VIN(MAX) For an application with a maximum input voltage of 32V, choose a 20V Zener diode, the VZENER(MAX) of which is around 21V and below the 33V maximum. The power loss in the clamp will determine the power rating of the Zener diode. Power loss in the clamp is highest at maximum load and minimum input voltage. The switch current is highest at this point along with the energy stored in the leakage inductance. A 0.25W Zener will satisfy most applications when the highest VZENER is chosen. 8301f 14 For more information www.linear.com/LT8301 LT8301 APPLICATIONS INFORMATION Tables 2 and 3 show some recommended diodes and Zener diodes. Table 2. Recommended Zener Diodes VZENER (V) POWER (W) CASE CMDZ5248B 18 0.25 SOD-323 CMDZ5250B 20 0.25 SOD-323 PART VENDOR Central Semiconductor Table 3. Recommended Diodes PART IMAX (A) VREVERSE (V) CASE CMHD4448 0.25 100 SOD-123 DFLS1100 1 100 PowerDI-123 Diodes Inc. DFLS1150 1 150 PowerDI-123 Diodes Inc. VENDOR Central Semiconductor The recommended approach for designing an RC snubber is to measure the period of the ringing on the SW pin when the power switch turns off without the snubber and then add capacitance (starting with 100pF) until the period of the ringing is 1.5 to 2 times longer. The change in period will determine the value of the parasitic capacitance, from which the parasitic inductance can be determined from the initial period, as well. Once the value of the SW node capacitance and inductance is known, a series resistor can be added to the snubber capacitance to dissipate power and critically dampen the ringing. The equation for deriving the optimal series resistance using the observed periods ( tPERIOD and tPERIOD(SNUBBED)) and snubber capacitance (CSNUBBER) is: CPAR = CSNUBBER L PAR = tPERIOD 2 Undervoltage Lockout (UVLO) A resistive divider from VIN to the EN/UVLO pin implements undervoltage lockout (UVLO). The EN/UVLO pin falling threshold is set at 1.228V with 14mV hysteresis. In addition, the EN/UVLO pin sinks 2.5µA when the voltage at the pin is below 1.228V. This current provides user programmable hysteresis based on the value of R1. The programmable UVLO thresholds are: 1.242V •(R1+R2) + 2.5µA •R1 R2 1.228V •(R1+R2) VIN(UVLO−) = R2 VIN(UVLO+) = Figure 7 shows the implementation of external shutdown control while still using the UVLO function. The NMOS grounds the EN/UVLO pin when turned on, and puts the LT8301 in shutdown with quiescent current less than 2µA. VIN R1 EN/UVLO LT8301 R2 RUN/STOP CONTROL (OPTIONAL) GND 2 tPERIOD(SNUBBED) −1 t PERIOD Note that energy absorbed by the RC snubber will be converted to heat and will not be delivered to the load. In high voltage or high current applications, the snubber may need to be sized for thermal dissipation. 8301 F07 Figure 7. Undervoltage Lockout (UVLO) CPAR • 4π 2 RSNUBBER = LPAR CPAR 8301f For more information www.linear.com/LT8301 15 LT8301 APPLICATIONS INFORMATION Minimum Load Requirement The LT8301 samples the isolated output voltage from the primary-side flyback pulse waveform. The flyback pulse occurs once the primary switch turns off and the secondary winding conducts current. In order to sample the output voltage, the LT8301 has to turn on and off at least for a minimum amount of time and with a minimum frequency. The LT8301 delivers a minimum amount of energy even during light load conditions to ensure accurate output voltage information. The minimum energy delivery creates a minimum load requirement, which can be approximately estimated as: LPRI •I SW(MIN)2 • f MIN ILOAD(MIN) = 2 • VOUT LPRI = Transformer primary inductance ISW(MIN) = Minimum switch current limit = 360mA (max) fMIN = Minimum switching frequency = 10.6kHz (max) The LT8301 typically needs less than 0.5% of its full output power as minimum load. Alternatively, a Zener diode with its breakdown of 20% higher than the output voltage can serve as a minimum load if pre-loading is not acceptable. For a 5V output, use a 6V Zener with cathode connected to the output. Output Short-Circuit Protection When the output is heavily overloaded or shorted, the reflected SW pin waveform rings longer than the internal blanking time. If no protection scheme is applied, after the 450ns minimum switch-off time, the excessive ring might falsely trigger the boundary mode detector and turn the power switch back on again before the secondary current falls to zero. The part then runs into continuous conduction mode at maximum switching frequency, and the switch current may run away. To prevent the switch current from running away under this condition, the LT8301 gradually folds back both maximum switch current limit and switching frequency as the output voltage drops from regulation. As a result, the switch current remains below 1.375A (typ) maximum switch current limit. In the worst-case scenario where the output is directly shorted to ground through a long wire and the huge ring after folding back still falsely triggers the boundary mode detector, a secondary overcurrent protection ensures that the LT8301 can still function properly. Once the switch current hits 2.2A overcurrent limit, a soft-start cycle initiates and throttles back both switch current limit and switching frequency very hard. This output short protection prevents the switch current from running away and limits the average output diode current. Design Example Use the following design example as a guide to design applications for the LT8301. The design example involves designing a 5V output with a 500mA load current and an input range from 8V to 32V. VIN(MIN) = 8V, VIN(NOM) = 12V, VIN(MAX) = 32V, VOUT = 5V, IOUT = 500mA Step 1: Select the Transformer Turns Ratio. NPS < 65V − VIN(MAX) − VLEAKAGE VOUT + VF VLEAKAGE = Margin for transformer leakage spike = 15V VF = Output diode forward voltage = ~0.3V Example: NPS < 65V − 32V −15V = 3.4 5V + 0.3V The choice of transformer turns ratio is critical in determining output current capability of the converter. Table 4 shows the switch voltage stress and output current capability at different transformer turns ratio. Table 4. Switch Voltage Stress and Output Current Capability vs Turns Ratio NPS VSW(MAX) at VIN(MAX) (V) IOUT(MAX) at VIN(MIN) (mA) DUTY CYCLE (%) 1:1 37.3 330 14-40 2:1 42.6 470 25-57 3:1 47.9 540 33-67 Since only NPS = 3 can meet the 500mA output current requirement, NPS = 3 is chosen in this example. 8301f 16 For more information www.linear.com/LT8301 LT8301 APPLICATIONS INFORMATION Step 2: Determine the Primary Inductance. Example: Primary inductance for the transformer must be set above a minimum value to satisfy the minimum switch-off and switch-on time requirements: LPRI ≥ LPRI ≥ ( tOFF(MIN) •N PS • VOUT + VF I SW(MIN) ) tON(MIN) • VIN(MAX) The transformer also needs to be rated for the correct saturation current level across line and load conditions. A saturation current rating larger than 2A is necessary to work with the LT8301. The 750313974 from Würth is chosen as the flyback transformer. I SW(MIN) tOFF(MIN) = 450ns tON(MIN) = 170ns Step 3: Choose the Output Diode. ISW(MIN) = 290mA (typ) Two main criteria for choosing the output diode include forward current rating and reverse voltage rating. The maximum load requirement is a good first-order guess as the average current requirement for the output diode. A conservative metric is the maximum switch current limit multiplied by the turns ratio, Example: 450ns • 3 •(5V + 0.3V) = 25µH 290mA 170ns • 32V LPRI ≥ = 19µH 290mA LPRI ≥ Most transformers specify primary inductance with a tolerance of ±20%. With other component tolerance considered, choose a transformer with its primary inductance 30% larger than the minimum values calculated above. LPRI = 40µH is then chosen in this example. Once the primary inductance has been determined, the maximum load switching frequency can be calculated as: fSW = I SW = (5V + 0.3V)• 3 = 0.57 (5V + 0.3V)• 3+12V 5V • 0.5A • 2 = 0.86A I SW = 0.85 •12V • 0.57 fSW = 199kHz D= IDIODE(MAX) = ISW(MAX) • NPS Example: IDIODE(MAX) = 4.125A Next calculate reverse voltage requirement using maximum VIN: VREVERSE = VOUT + 1 1 = LPRI •ISW tON + tOFF LPRI •ISW + VIN NPS •(VOUT + VF ) Example: VOUT •I OUT • 2 η• VIN •D VREVERSE = 5V + VIN(MAX) NPS 32V = 15.6V 3 The CMSH5-20 (5A, 20V diode) from Central Semiconductor is chosen. 8301f For more information www.linear.com/LT8301 17 LT8301 APPLICATIONS INFORMATION Step 4: Choose the Output Capacitor. The output capacitor should be chosen to minimize the output voltage ripple while considering the increase in size and cost of a larger capacitor. Use the equation below to calculate the output capacitance: COUT = VSW(MAX) = VIN(MAX) + VZENER(MAX) 2 • VOUT • ∆VOUT Example: Design for output voltage ripple less than 1% of VOUT, i.e., 50mV. Choose a diode that is fast and has sufficient reverse voltage breakdown: VREVERSE > VSW(MAX) LPRI •I SW 2 Example: COUT = A 20V Zener with a maximum of 21V will provide optimal protection and minimize power loss. So a 20V, 0.25W Zener from Central Semiconductor (CMDZ5250B) is chosen. 40µH •(0.86A)2 = 60µF 2 • 5V • 0.05V A 100V, 0.25A diode from Central Semiconductor (CMHD4448) is chosen. Step 6: Select the RFB Resistor. Remember ceramic capacitors lose capacitance with applied voltage. The capacitance can drop to 40% of quoted capacitance at the maximum voltage rating. So a 100µF, 10V rating ceramic capacitor is chosen. Step 5: Design Snubber Circuit. The snubber circuit protects the power switch from leakage inductance voltage spike. A DZ snubber is recommended for this application because of lower leakage inductance and larger voltage margin. The Zener and the diode need to be selected. The maximum Zener breakdown voltage is set according to the maximum VIN: VZENER(MAX) ≤ 65V – VIN(MAX) VREVERSE > 53V Use the following equation to calculate the starting value for RFB: RFB = NPS •(VOUT + VF ) 100µA Example: RFB = 3 •(5V + 0.3V) = 159k 100µA Depending on the tolerance of standard resistor values, the precise resistor value may not exist. For 1% standard values, a 158k resistor should be close enough. As discussed in the Application Information section, the final RFB value should be adjusted on the measured output voltage. Example: VZENER(MAX) ≤ 65V – 32V = 33V 8301f 18 For more information www.linear.com/LT8301 LT8301 APPLICATIONS INFORMATION Step 7: Select the EN/UVLO Resistors. Step 8: Ensure minimum load. Determine the amount of hysteresis required and calculate R1 resistor value: The theoretical minimum load can be approximately estimated as: VIN(HYS) = 2.5µA • R1 ILOAD(MIN) = 40µH•(360mA)2 •10.6kHz = 5.5mA 2 • 5V Example: Choose 2V of hysteresis, Remember to check the minimum load requirement in real application. The minimum load occurs at the point where the output voltage begins to climb up as the converter delivers more energy than what is consumed at the output. The real minimum load for this application is about 6mA. In this example, a 820Ω resistor is selected as the minimum load. R1 = 806k Determine the UVLO thresholds and calculate R2 resistor value: VIN(UVLO+) = 1.242V •(R1+R2) + 2.5µA •R1 R2 Example: Set VIN UVLO rising threshold to 7.5V, R2 = 232k VIN(UVLO+) = 7.5V VIN(UVLO–) = 5.5V 8301f For more information www.linear.com/LT8301 19 LT8301 TYPICAL APPLICATIONS 2.7V to 36VIN/15VOUT Micropower Isolated Flyback Converter VIN 2.7V TO 36V 10µF VIN SW EN/UVLO LT8301 Z1 40µH D1 • T1 1:1 D2 • VOUT+ 15V 2mA TO 130mA (VIN = 5V) 10µF 2mA TO 230mA (VIN = 12V) 2mA TO 320mA (VIN = 24V) 2mA TO 370mA (VIN = 36V) 40µH 150k RFB GND 8301 TA02a VOUT– D1: CENTRAL CMHD4448 D2: CENTRAL CMMR1U-02 T1: SUMIDA 12387-T041 Z1: CENTRAL CMDZ5248B Efficiency vs Load Curent 95 EFFICIENCY (%) 90 85 80 75 VIN = 5V VIN = 12V VIN = 24V VIN = 36V 70 65 0 100 200 300 LOAD CURRENT (mA) 400 8301 TA02b 8V to 36VIN/3.3VOUT Micropower Isolated Flyback Converter VIN 8V TO 36V 4.7µF 806k 232k VIN EN/UVLO LT8301 GND SW Z1 40µH D1 • T1 4:1 D2 • 2.5µH 137k RFB VOUT+ 3.3V 8.5mA TO 0.95A (VIN = 12V) 47µF 8.5mA TO 1.30A (VIN = 24V) 8.5mA TO 1.50A (VIN = 36V) 8301 TA03 VOUT– D1: CENTRAL CMHD4448 D2: NXP PMEG2020EH T1: SUMIDA 12387-T036 Z1: CENTRAL CMDZ5250B 8301f 20 For more information www.linear.com/LT8301 LT8301 TYPICAL APPLICATIONS 8V to 36VIN/24VOUT Micropower Isolated Flyback Converter VIN 8V TO 36V D2 4.7µF T1 1:2 Z1 VIN 806k 232k D1 • 160µH • SW EN/UVLO LT8301 GND 40µH 121k VOUT+ 24V 1.2mA TO 130mA (VIN = 12V) 1.2mA TO 180mA (VIN = 24V) 4.7µF 1.2mA TO 200mA (VIN = 36V) 8301 TA04a RFB VOUT– D1: CENTRAL CMHD4448 D2: ST STPS1150A T1: WÜRTH 750313975 Z1: CENTRAL CMDZ5248B Efficiency vs Load Curent 95 EFFICIENCY (%) 90 85 80 75 VIN = 12V VIN = 24V VIN = 36V 70 65 0 50 100 150 LOAD CURRENT (mA) 200 8301 TA04b 8V to 36VIN/48VOUT Micropower Isolated Flyback Converter VIN 8V TO 36V 4.7µF 806k 232k VIN EN/UVLO LT8301 GND SW Z1 40µH D1 • T1 1:4 D2 • 640µH 118k RFB VOUT+ 48V 0.6mA TO 70mA (VIN = 12V) 1µF 0.6mA TO 90mA (VIN = 24V) 0.6mA TO 100mA (VIN = 36V) 8301 TA03 VOUT– D1: CENTRAL CMHD4448 D2: DIODES BAV21W-7-F T1: WÜRTH 750313976 Z1: CENTRAL CMDZ5252B 8301f For more information www.linear.com/LT8301 21 LT8301 TYPICAL APPLICATIONS VIN to (VIN + 10V)/(VIN – 10V) Micropower Converter D1 VIN + 10V 150mA 4.7µF VIN 2.7V TO 42V T1 1:1 10µF VIN 40µH LT8301 EN/UVLO SW Z1 VIN 150mA • 40µH • 4.7µF Z2 D2 8301 TA06 VIN – 10V 102k D1, D2: DIODES INC. DFLS160 T1: SUMIDA 12387-T041 Z1: CENTRAL CMDZ12L RFB GND 12V to 24VIN/Four 15VOUT Micropower Isolated Flyback Converter T1 D2 1:1:1:1:1 VIN 12V TO 24V 4.7µF 806k Z1 30µH VIN D1 LT8301 EN/UVLO • • 150k 2.2µF 7.5k VOUT1– VOUT2+ 15V 60mA D3 SW 232k 30µH VOUT1+ 15V 60mA • RFB 30µH 2.2µF 7.5k GND VOUT2– VOUT3+ 15V 60mA D4 D1: CENTRAL CMHD4448 D2-D5: CENTRAL CMMR1U-02 T1: SUMIDA EPH2815-ADBN-A0349 Z1: CENTRAL CMDZ5248B • 30µH 2.2µF 7.5k VOUT3– VOUT4+ 15V 60mA D5 • 30µH 2.2µF 7.5k 8301 TA07 VOUT4– 8301f 22 For more information www.linear.com/LT8301 LT8301 PACKAGE DESCRIPTION Please refer to http://www.linear.com/designtools/packaging/ for the most recent package drawings. S5 Package 5-Lead Plastic TSOT-23 (Reference LTC DWG # 05-08-1635 Rev B) 0.62 MAX 0.95 REF 2.90 BSC (NOTE 4) 1.22 REF 1.4 MIN 3.85 MAX 2.62 REF 2.80 BSC 1.50 – 1.75 (NOTE 4) PIN ONE RECOMMENDED SOLDER PAD LAYOUT PER IPC CALCULATOR 0.30 – 0.45 TYP 5 PLCS (NOTE 3) 0.95 BSC 0.80 – 0.90 0.20 BSC 0.01 – 0.10 1.00 MAX DATUM ‘A’ 0.30 – 0.50 REF 0.09 – 0.20 (NOTE 3) NOTE: 1. DIMENSIONS ARE IN MILLIMETERS 2. DRAWING NOT TO SCALE 3. DIMENSIONS ARE INCLUSIVE OF PLATING 4. DIMENSIONS ARE EXCLUSIVE OF MOLD FLASH AND METAL BURR 5. MOLD FLASH SHALL NOT EXCEED 0.254mm 6. JEDEC PACKAGE REFERENCE IS MO-193 1.90 BSC S5 TSOT-23 0302 REV B 8301f Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representaFor more information www.linear.com/LT8301 tion that the interconnection of its circuits as described herein will not infringe on existing patent rights. 23 LT8301 TYPICAL APPLICATION 8V to 36VIN/12VOUT Micropower Isolated Flyback Converter VIN 8V TO 36V D2 4.7µF T1 1:1 Z1 VIN 806k EN/UVLO LT8301 232k GND D1 40µH • VOUT+ 12V 2.5mA TO 270mA (VIN = 12V) 2.5mA TO 360mA (VIN = 24V) 10µF 2.5mA TO 400mA (VIN = 36V) 40µH • SW 118k RFB 8301 TA08a D1: CENTRAL CMHD4448 D2: DIODE INC. DFLS160 T1: WÜRTH 750313972 Z1: CENTRAL CMDZ5250B Efficiency vs Load Current Output Load and Line Regulation 12.4 95 12.3 OUTPUT VOLTAGE (V) EFFICIENCY (%) 90 85 80 75 VIN = 12V VIN = 24V VIN = 36V 70 65 VOUT– 0 100 200 300 LOAD CURRENT (mA) 400 12.2 12.1 12.0 11.9 11.8 VIN = 12V VIN = 24V VIN = 36V 11.7 11.6 0 200 100 300 LOAD CURRENT (mA) 400 8301 TA08c 8301 TA08b RELATED PARTS PART NUMBER DESCRIPTION COMMENTS LT8300 100VIN Micropower Isolated Flyback Converter with 150V/260mA Switch Low IQ Monolithic No-Opto Flybacks, 5-Lead TSOT-23 LT8302 42VIN Micropower Isolated Flyback Converter with 65V/3.6A Switch Low IQ Monolithic No-Opto Flybacks, 8-Lead SO-8E LT8309 Secondary-Side Synchronous Rectifier Driver 4.5V ≤ VCC ≤ 40V, Fast Turn-On and Turn-Off, 5-Lead TSOT-23 LT3511/LT3512 100V Isolated Flyback Converters Monolithic No-Opto Flybacks with Integrated 240mA/420mA Switch, MSOP-16(12) LT3748 100V Isolated Flyback Controller 5V ≤ VIN ≤ 100V, No Opto Flyback , MSOP-16 with High Voltage Spacing LT3798 Off-Line Isolated No Opto-Coupler Flyback Controller with Active PFC VIN and VOUT Limited Only by External Components LT3573/LT3574/LT3575 40V Isolated Flyback Converters Monolithic No-Opto Flybacks with Integrated 1.25A/0.65A/2.5A Switch LT3757A/LT3759/ LT3758 40V/100V Flyback/Boost Controllers Universal Controllers with Small Package and Powerful Gate Drive LT3957/LT3958 40V/100V Flyback/Boost Converters Monolithic with Integrated 5A/3.3A Switch LTC 3803/LTC3803-3/ LTC3803-5 200kHz/300kHz Flyback Controllers in SOT-23 VIN and VOUT Limited by External Components LTC3805/LTC3805-5 Adjustable Frequency Flyback Controllers VIN and VOUT Limited by External Components ® 8301f 24 Linear Technology Corporation 1630 McCarthy Blvd., Milpitas, CA 95035-7417 For more information www.linear.com/LT8301 (408) 432-1900 ● FAX: (408) 434-0507 ● www.linear.com/LT8301 LT 0214 • PRINTED IN USA LINEAR TECHNOLOGY CORPORATION 2014