Data Sheet No. PD 94716 IR3094PBF 3 PHASE PWM CONTROLLER FOR POINT OF LOAD DESCRIPTION The IR3094 Control IC provides a full featured, cost effective, single chip solution to offers a compact, efficient solution for high current POL converters. Control and 3 Phase Gate Drive functions are integrated into a single space-saving IC. FEATURES x x x x x x x x x x x x x x x 0.85V Reference Voltage 3A GATELX Pull Down Drive Capability Programmable 100KHz to 540KHz oscillator Programmable Voltage Positioning (can be disabled) Programmable Softstart Programmable Hiccup Over-Current Protection with Delay to prevent false triggering Simplified Powergood provides indication of proper operation and avoids false triggering Operates up to 16V converter input with 7.5V Under-Voltage Lockout 4.36V Under-Voltage Lockout threshold for gate driver voltage Adjustable Voltage, 150mA Bias Regulator provides MOSFET Drive Voltage Enable Input OVP Flag Output detects high side fet short at powerup Separate OVP sense line to sense the output voltage and latched OVP with protection Inductor DCR sensing for current sensing will support up to 5.1V output applications Available 48L MLPQ package ORDERING INFORMATION Device Order Quantity IR3094MTRPBF 3000 per Reel IR3094MPBF 100 piece strips IR3094 48LD MLPQ GATEH1 PGND1 GATEL1 VCCL1_2 5VUVL GATEL2 PGND2 GATEH2 VCCH2 VCCH3 GATEH3 PGND3 48L MLPQ (7 x 7 mm Body) o JA = 27 C/W LGND SETBIAS VCC CSINP3 CSINM3 BIASOUT PWRGD CSINP2 CSINM2 NC VCCL3 GATEL3 NC NC ROSC VOSNSOCSET VREF VDRP FB EAOUT SS/DEL SCOMP2 SCOMP3 NC NC NC NC 5VREF OVPSNS ENABLE OVP CSINP1 CSINM1 NC VCCH1 PACKAGE INFORMATION Page 1 of 29 09/26/05 IR3094PBF PIN DESCRIPTION PIN# PIN SYMBOL PIN DESCRIPTION 1 2 3 4 NC NC ROSC VOSNS- 5 OCSET 6 VREF 7 8 9 VDRP FB EAOUT 10 SS/DEL 11 SCOMP2 12 SCOMP3 13 14 15 16 17 18 19 20 21 22 23 24 25 26 27 28 29 30 31 LGND SETBIAS VCC CSINP3 CSINM3 BIASOUT PWRGD CSINP2 CSINM2 NC VCCL3 GATEL3 PGND3 GATEH3 VCCH3 VCCH2 GATEH2 PGND2 GATEL2 32 5VUVL 33 34 35 36 37 38 39 40 41 42 43 44 45 46 47 48 VCCL1_2 GATEL1 PGND1 GATEH1 VCCH1 NC CSINM1 CSINP1 OVP ENABLE OVPSNS 5VREF NC NC NC NC Not connected Not connected Connect a resistor to VOSNS- to program oscillator frequency, OCSET and STBIAS bias currents. Remote Sense Input. Connect to ground at the load. Programs the hiccup over-current threshold through an external resistor tied to VREF and an internal current source. The bias current is a function of ROSC. 0.85V Reference voltage. Current Sensing and Over Current Protection are referenced to this pin. An external RC network tied to VOSNS- is needed for the compensation. Buffered average current information. Connect an external resistor to FB to program converter output. .impedance Inverting input to the Error Amplifier. Output of the Error Amplifier. Controls Converter Softstart, Power Good, and Over-Current Timing. Connect an external capacitor to LGND to program the timing. Compensation for the Current Share control loop. Connect a capacitor to ground to set the control loop’s bandwidth. Phase 2 is forced to match phase 1’s current. Compensation for the Current Share control loop. Connect a capacitor to ground to set the control loop’s bandwidth. Phase 3 is forced to match phase 1’s current. Local Ground and IC substrate connection. External resistor to ground sets voltage at BIASOUT pin. Bias current is a function of ROSC. Power for internal circuitry and source for BIASOUT regulator. Non-inverting input to the Phase 3 Current Sense Amplifier. Inverting input to the Phase 3 Current Sense Amplifier. 150mA open-looped regulated voltage set by SETBIAS for GATE drive bias. Open Collector output that drives low during Softstart or any fault condition. Connect external pull-up. Non-inverting input to the Phase 2 Current Sense Amplifier. Inverting input to the Phase 2 Current Sense Amplifier. Not connected Power for Phase 3 Low-Side Gate Driver. Phase 3 Low-Side Gate Driver Output and input to GATEH3 non-overlap comparator. Return for Phase 3 Gate Drivers. Phase 3 High-Side Gate Driver Output and input to GATEL3 non-overlap comparator. Power for Phase 3 High-Side Gate Driver. Power for Phase 2 High-Side Gate Driver. Phase 2 High-Side Gate Driver Output and input to GATEL2 non-overlap comparator. Return for Phase 2 Gate Drivers. Phase 2 Low-Side Gate Driver Output and input to GATEH2 non-overlap comparator. Can be used to monitor the driver supply voltage or 5V supply voltage when converting from 5V. An under voltage condition initiates Soft Start. Power for Phase 1 and 2 Low-Side Gate Drivers. Phase 1 Low-Side Gate Driver Output and input to GATEH1 non-overlap comparator. Return for Phase 1 Gate Drivers. Phase 1 High-Side Gate Driver Output and input to GATEL1 non-overlap comparator. Power for Phase 1 High-Side Gate Driver. Not connected Inverting input to the Phase 1 Current Sense Amplifier. Non-inverting input to the Current Sense Amplifier. Output that drives high during an Over-Voltage condition. Enable Input. A logic low applied to this pin puts the IC into Fault mode. Dedicated output voltage sense pin for Over Voltage Protection. Decoupling for internal voltage reference rail. Not connected Not connected Not connected Not connected Page 2 of 29 09/26/05 IR3094PBF ABSOLUTE MAXIMUM RATINGS o o Operating Junction Temperature……………..0 C to 150 C o o Storage Temperature Range………………….-65 C to 150 C PIN 3 4 5 6 7 8 9 10 11 12 13 14 15 16 17 18 19 20 21 22 23 24 25 26 27 28 29 30 31 32 33 34 35 36 37 38 39 40 41 42 43 44 NAME ROSC VOSNSOCSET VDAC VDRP FB EAOUT SS/DEL SCOMP2 SCOMP3 LGND SETBIAS VCC CSINP3 CSINM3 BIASOUT PWRGD CSINP2 CSINM2 NC VCCL3 GATEL3 PGND3 GATEH3 VCCH3 VCCH2 GATEH2 PGND2 GATEL2 5VUVL VCCL1_2 GATEL1 PGND1 GATEH1 VCCH1 NC CSINM1 CSINP1 OVP ENABLE OVPSNS 5VREF Page 3 of 29 VMAX 20V 0.5V 20V 20V 20V 20V 10V 20V 20V 20V n/a 20V 20V 20V 20V 20V 20V 20V 20V n/a 20V 20V 0.3V 30V 30V 30V 30V 0.3V 20V 20V 20V 20V 0.3V 30V 30V n/a 20V 20V 20V 20V 20V 10V VMIN -0.3V -0.5V -0.3V -0.3V -0.3V -0.3V -0.3V -0.3V -0.3V -0.3V n/a -0.3V -0.3V -0.3V -0.3V -0.3V -0.3V -0.3V -0.3V n/a -0.3V -0.3V DC, -2V for 100ns -0.3V -0.3V DC, -2V for 100ns -0.3V -0.3V -0.3V DC, -2V for 100ns -0.3V -0.3V DC, -2V for 100ns -0.3V -0.3V -0.3V DC, -2V for 100ns -0.3V -0.3V DC, -2V for 100ns -0.3V n/a -0.3V -0.3V -0.3V -0.3V -0.3V -0.3V ISOURCE 1mA 10mA 1mA 1mA 25mA 1mA 5mA 1mA 1mA 1mA 50mA 1mA 1mA 1mA 1mA 450mA 1mA 1mA 1mA n/a n/a 3A for 100ns, 200mA DC 3A for 100ns, 200mA DC 3A for 100ns, 200mA DC n/a n/a 3A for 100ns, 200mA DC 3A for 100ns, 200mA DC 3A for 100ns, 200mA DC 1mA n/a 3A for 100ns, 200mA DC 3A for 100ns, 200mA DC 3A for 100ns, 200mA DC n/a n/a 1mA 1mA 1mA 1mA 1mA 10mA ISINK 1mA 1mA 1mA 1mA 5mA 1mA 10mA 1mA 1mA 1mA 1mA 1mA 500mA 1mA 1mA 1mA 20mA 1mA 1mA n/a 3A for 100ns, 200mA DC 3A for 100ns, 200mA DC n/a 3A for 100ns, 200mA DC 3A for 100ns, 200mA DC 3A for 100ns, 200mA DC 3A for 100ns, 200mA DC n/a 3A for 100ns, 200mA DC 1mA 3A for 100ns, 200mA DC 3A for 100ns, 200mA DC n/a 3A for 100ns, 200mA DC 3A for 100ns, 200mA DC n/a 1mA 1mA 1mA 1mA 1mA 20mA 09/26/05 IR3094PBF ELECTRICAL SPECIFICATIONS Unless otherwise specified, these specifications apply over: 8.0 VCC 16V, 4V VCCLX 14V, o o 4V VCCHX 28V, CGATEHX =3.3nF, CGATELX =6.8nF, 0 C TJ 125 C PARAMETER VREF Reference Sink Current Source Current System Reference Voltage TEST CONDITION RROSC = 47k95()=OCSET RROSC = 47k95()=OCSET Connect FB to EAOUT, Measure V(EAOUT)-V(VOSNS-). Applies to -0.3V<VOSNS-<0.3V. MIN TYP MAX UNIT 45 48 53 56 61 64 PA PA 0.8415 0.85 0.8585 V -5 -1 3 mV 40 1.2 90 4 90 2 100 7 1.25 430 1.1 4.9 50 150 2.5 105 uA V dB MHz 600 1.5 5.3 200 V/Ps PA mA V mV 0 125 mV 0.2 4 200 8 300 3.75 20 650 V mA 160 200 240 kHz 102 120 138 ° Error Amplifier Input Offset Voltage UVL FB Bias Current UVL Head Room DC Gain Gain-Bandwidth Product Slew Rate Source Current Sink Current Max Voltage Min Voltage Connect FB to EAOUT, Measure V(EAOUT)-V(VREF). Applies to -0.3V<VOSNS-<0.3V. Note 1 Note 1 Note 1 Note 1, 50mV FB signal 300 .75 4.5 VDRP Buffer Amplifier Positioning Offset Voltage V(VDRP) – V(REF) with CSINMX=CSINPX=0. Note 1. Output Voltage Range Source Current Sink Current -125 PA Oscillator Switching Frequency Phase Shift Page 4 of 29 RROSC = 47k Sequence: GATEH1-GATEH2GATEH3 09/26/05 IR3094PBF PARAMETER TEST CONDITION MIN TYP MAX UNIT 94 0 103 0.25 117.5 0.55 PA V 1.2 1.8 2.5 V 150 250 500 mA 0.8 1.1 1.8 V 30 3.5 25 60 6 55 90 9 70 9 10 13 PA/PA 3.8 180 4.0 245 4.2 310 V mV 170 265 350 mV -125 0 125 mV 23.5 3.9 27 29.4 PA V 7.0 6.5 400 4.05 3.92 100 7.5 7.0 500 4.36 4.17 200 8.0 7.5 700 4.60 4.40 250 V V mV V V mV 150 0 400 10 mV BIASOUT Regulator SETBIAS Bias Current Set Point Accuracy BIASOUT Dropout Voltage RROSC = 47k V(SETBIAS)-V(BIASOUT) @ 100mA I(BIASOUT)=100mA,Threshold when V(SETBIAS)-V(BIASOUT)=0.45V BIASOUT Current Limit Soft Start and Delay SS/DEL to FB Input Offset Voltage Charge Current Hiccup Discharge Current OC Discharge Current Charge/Discharge Current Ratio Charge Voltage Delay Comparator Threshold Discharge Comparator Threshold With FB = 0V, adjust V(SS/DEL) until EAOUT drives high Relative to Charge Voltage PA PA PA Over-Current Comparator Input Offset Voltage OCSET Bias Current Max OCSET Set Point V(OCSET)-V(VREF), CSINM=CSINP1=CSINP2=CSINP3, Note 1. RROSC = 47k Under-Voltage Lockout VCC Start Threshold VCC Stop Threshold VCC Hysteresis 5VUVL Start Threshold 5VUVL Stop Threshold 5VUVL Hysteresis Start – Stop Start – Stop PWRGD Output Output Voltage Leakage Current Page 5 of 29 I(PWRGD) = 4mA V(PWRGD) = 5.5V 09/26/05 PA IR3094PBF PARAMETER TEST CONDITION MIN TYP MAX UNIT 1.3 5 2.4 1.5 10 3.0 1.7 20 3.7 V k V 25 50 ns 25 50 ns 50 90 ns 30 60 ns 0 0.5V V 10 0 25 0.5V 50 V ns 10 25 50 ns 20 35 50 PA 100 150 4 0.9 65 Enable Input Threshold Input Resistance Pull-up Voltage Gate Drivers GATEH Rise Time GATEH Fall Time GATEL Rise Time GATEL Fall Time High Voltage (AC) Low Voltage (AC) GATEL low to GATEH high delay GATEH low to GATEL high delay Disable Pull-Down Current Referenced to VOSNS- VCCHX = 8V, Measure 1V to 7V transition time. Note 1. VCCHX = 8V, Measure 7V to 1V transition time. Note 1. VCCLX= 8V, Measure 1V to 7V transition time. Note 1. VCCLX= 8V, Measure 7V to 1V transition time. Note 1. Measure VCCLX– GATELX or VCCHX – GATEHX, Note 1 Measure GATELX or GATEHX, Note 1 VCCHX = VCCLX= 8V, Measure the time from GATELX falling to 1V to GATEHX rising to 1V. Note 1. VCCHX = VCCLX= 8V, Measure the time from GATEHX falling to 1V to GATELX rising to 1V. Note 1. GATHX or GATELX=2V with VCC = 0V. Measure Gate pull-down current PWM Comparator Propagation Delay Common Mode Input Range Internal Ramp Start Voltage Internal Ramp Amplitude Current Sense Amplifier CSINPX Bias Current CSINM2,3 Bias Current CSINM1 Bias Current Phase 2 and 3 Input Current Offset Ratio Phase 1 Input Current Offset Ratio Average Input Offset Voltage Offset Voltage Mismatch o Gain at TJ = 25 C o Gain at TJ = 125 C Gain Mismatch Differential Input Range Common Mode Input Range Page 6 of 29 Note1 (VDRP-VREF)/GAIN with CSINX=0. Note1 Monitor I(SCOMPX), Note1. Note 1. ns V 0.44 35 0.6 50 -1 -1 -2 0 0 -0.5 1 1 1 1 PA PA PA PA/PA 0.5 1.7 4 PA/PA -5 -5 22.5 19 -1 -25 -0.2 0 0 24 20.9 0 5 5 25.5 22 1 75 5.5 mV mV V/V V/V V/V mV V 09/26/05 V mV / %DTC IR3094PBF PARAMETER Share Adjust Error Amplifier Input Offset Voltage MAX Duty Cycle Adjust Ratio MIN Duty Cycle Adjust Ratio Transconductance SCOMPX Source/Sink Current SCOMPX Precondition and GATELX Release Threshold SCOMP precondition current Duty Cycle Match at Startup 0% Duty Cycle Comparator Threshold Voltage Propagation Delay OVP Comparator Threshold Power-up Headroom for OVP Flag OVPSNS Threshold at Powerup SS/DEL Power-up Clear Threshold Propagation Delay OVP Source Current OVP Pull Down Resistance OVP High Voltage OVPSNS Bias Current 5VREF Short Circuit Current Supply Voltage General VCC Supply Current VOSNS- Current VCCHX and VCCL3 Current VCCL1_2 Supply Current 5VUVL Supply Current Non_Sync to Sync Threshold TEST CONDITION MIN TYP MAX Note 1 Compare Duty Cycle to GATEH1 Compare Duty Cycle to GATEH1 Note 1 -5 1.5 0.6 100 16 0 2.0 0.5 200 22 5 mV 300 28 PA/V PA V(FB) 0.6 0.67 0.74 V Compare Duty Cycle to GATEHX 160 -7 360 -1 560 7 PA % Below Internal Ramp1 Start Voltage VCCLX= 8V. Step EAOUT from .8V to .3V and measure time to GATELX transition to < 7V. -25 25 75 mV 200 400 ns Compare to V(VREF) VCC=OVPSNS where V(OVP)>0.5V. Same for 5VUVL=OVPSNS. VCC=2V, V(OVP) >0.5V. Same for V(5VUVL)=2V. VCC=12V, V(OVPSNS)=1V, VREF=1.6V, where OVP<0.5V VCCLX= 8V. V(EAOUT)=0V. Step OVPSNS 540mV + V(VREF). Measure time to GATELX transition to >1V. Note 1. V(OVP)=0.5V, VCC=1.8V, 5VUVL=0V OVP to LGND I(OVP)=10uA, V(VCC) or V(5VUVL)V(OVP), VCC=1.8V 120 0.8 150 200 mV 1.1 1.8 V 0.3 0.48 0.85 V 0.35 0.60 0.95 V 150 350 650 ns 10 30 75 60 100 PA k 0.4 0.70 1.1 V -6.0 -3.0 1.5 uA I(5VREF)=0A 20 4.5 45 5 60 5.5 mA V V(VCC)=16V -0.3V VOSNS- 0.3V V(VCCHX)=28V, V(VCCL3)=14V V(VCCL1_2)=14V V(5VUVL)=5V, no OVP condition 28.5 0.6 3 6 100 35 0.8 5 10 200 40.5 1.2 7 17 400 70.6 77.7 87 mA mA mA mA uA %VRE F Note 1: Guaranteed by design, but not tested in production Note 2: VREF Output is trimmed to compensate for Error Amp input offsets errors Page 7 of 29 UNIT 09/26/05 IR3094PBF TYPICAL OPERATING CHARACTERISTICS I(VDAC) REF Sink and Source Currents vs. ROSC I(OCSET) Current vs. ROSC 90 180 140 I(VDAC) REF Source Current 120 I(VDAC) REF Sink Current 100 80 70 80 I(OCSET) 60 uA uA 160 50 40 30 60 20 40 10 20 0 0 10 20 30 40 50 10 60 70 80 90 100 110 120 130 ROSC in Kohms 20 30 450 350 300 uA Frequency (kHz) 400 250 200 150 100 50 0 30 40 50 60 70 80 90 60 70 80 90 100 110 120 90 100 110 120 I(SETBIAS) vs. ROSC Oscillator freq vs. ROSC 20 50 ROSC (kOhm) 500 10 40 100 110 120 320 300 280 260 240 220 200 180 160 140 120 100 80 60 40 20 10 20 30 40 50 ROSC (kOhm) 60 70 80 ROSC (kOhm) Peak High side Gate drive current vs. Laod oad capacitance Frequency and Bias Current Accuracy vs. ROSC (includes temperature) 2.000 6 +/-3 Sigm a Variation (% ) 5 Frequency 4 VREF Sink VREF Source 3 OCSET SETBIAS 2 I(GATEHX) in Amps 1.900 1.800 1.700 1.600 1.500 1.400 1.300 I(RISE) I(FALL) 1.200 1.100 1.000 1 1 10 20 30 40 50 60 ROSC (kOhm) Page 8 of 29 70 80 90 100 2 3 4 5 6 7 8 9 C(GATEHX) in nF 09/26/05 10 IR3094PBF Peak Low side Gate drive current vs. Laod oad capacitance 3.250 I(GATELX) in Amps 3.000 2.750 2.500 2.250 2.000 1.750 I(RISE) I(FALL) 1.500 1.250 1.000 1 2 3 4 5 6 7 8 9 10 C(GATELX) in nF Error Amplifier Frequency Response 180 100 0 93dB DC gain 88° Phase Margin 3.1MHz Crossover -100 -180 1.0Hz 10Hz DB(V(comp)) 100Hz 1.0KHz P(V(comp)) Page 9 of 29 10KHz 100KHz 1.0MHz 10MHz 100MHz Frequency 09/26/05 EAOUT STARTUP OVP Comparator OVPSNS + 75U - ON VCC LGND UVL VCCH1 60k 5VUVL DRIVE 0.48V RESET DOMINANT - 4 X IROSC OVP Comparator + 1.243 UVL 5VUVL + 4.36V START 4.17V STOP S VDAC CLK1 CLK1 CLK2 CLK2 CLK3 CLK3 Q PWM COMPARATOR OVP LATCH 150mV IROSC BIASOUT OL_IN S - - 7.5V START 7.0V STOP + SETBIAS CLK1 R IN Error_Amp VCCH2 0% DUTY CYCLE DRIVE RESET DOMINANT CLK2 + GATEHI OL_OUT PGND PGND2 GateHI QB GATEH2 RSFF SYNC LATCH DELAY S 0.6V + - 245mV R - Sof tStart_Clamp OFF + - SS QB SET DOMINANT VOSN S- + 60U IROSC/2 Q CO1 CO2 + PRESET DRIVE OL_IN IN 9p 0 TO IROSC*3/4 Share Adjust Error Amp GATELO PGND 0.6V - OL_OUT SCOMP2 H FORCES IROSC/2 VCCH3 6U 55U FAULT LATCH ON U37 S RESET DOMINANT Q CLK3 Discharge Comparator - SET DOMINANT + + CO3 0.265V summer REF BUFFER ENABLE PGND3 CO1 + - PRESET 9p 0.6V IN PGND OL_OUT GATELO SCOMP3 INTERNAL REFERENCE 5VREF VOSNS- VREF CSINM3 CSINP3 CSINM2 CSINP2 CSINM1 GATEL3 GateLO + - VCCL3 DRIVE OL_IN 0 TO IROSC*3/4 Share Adjust Error Amp X23.5 + X23.5 + V O S N S- OL_OUT GateHI + VOSN S- - 0.85V 1.5V GATEH3 CSINP1 VDRP IR3094PBF VD AC 09/26/05 X23.5 PGND GATEHI IROSC CO3 IROSC + OL_IN QB RSFF summer VD AC + R IN Q CO1 summer VD AC - CO2 - 10k - R OR4 DRIVE S PWM COMPARATOR IAVE 3V GATEL2 GateLO 0.75*VDAC 1.1V - 4V R + + IROSC + Figure 1 – IR3094 Block Diagram IAVE Q - - IN OL_IN S PWM COMPARATOR OVER CURRENT - GATEL1 - DISABLE 0.575V OCSET GATELO PGND 0.6V OL_OUT + + DRIVE OL_IN GateLO + PWRGD PGND1 VCCL1_2 IROSC/2 QB 9p - GATEH1 RSFF Q SET DOMINANT Oscillator GATEHI OL_OUT PGND GateHI QB R + IN - Page 10 of 29 IR O SC OVP 75U IR3094 THEORY OF OPERATION FB ROSC VCC IR3094PBF PWM Operation The IR3094 is a fully integrated 3 phase interleaved PWM control IC which uses voltage mode control with trailing edge modulation. A high-gain wide-bandwidth voltage type Error Amplifier in the Control IC is used for the voltage control loop. The PWM block diagram of the IR3094 is shown in Figure 2. U30 IROSC RSFF CLK1 S CLK2 CLK2 CLK3 CLK3 Q PWM COMPARATOR QB R RESET DOMINANT + OSCBLOCK VIN OVP LATCH OVP SET S OVP RESET VREF GATEH1 IROSC/2 Q 1 GATEL1 CCS1 ERROR AMPLIFIER + VOSNS- + RDAC RCS1 0.6V 9p CDAC VDAC 2 QB R - 0% DUTY CYCLE FB CCOMP 0.575V CLK2 IROSC + EAOUT - RFB 2 VOUT+ GATEL2 COUT RCS2 CCS2 VOUT- + VDRP VOUT SENSE+ 1 IROSC VDRP BUFFER RDRP GATEH2 Q QB R RESET DOMINANT - RCOMP VIN RSFF S PWM COMPARATOR VOUT SENSE- Share Adjust Error Amp + - 0.6V 9p 0 TO IROSC*3/4 SCOMP2 CSC2 CLK3 RSC2 PWM COMPARATOR EAOUT - + VIN RSFF S GATEH3 Q QB R RESET DOMINANT 1 2 GATEL3 RCS3 IROSC CCS3 Share Adjust Error Amp + - 0 TO IROSC*3/4 9p 0.6V SCOMP3 VDAC CSC3 RSC3 X23.5 CSINM3 - CSINP3 + VDAC X23.5 CSINM2 - CSINP2 + VDAC X23.5 CSINM1 - CSINP1 + Figure 2 – PWM Block Diagram Refer to Figure 3. Upon receiving a clock pulse, the RSFF is set, the internal PWM ramp voltage begins to increase, the low side driver is turned off, and the high side driver is then turned on. For phase 1, an internal 9pf capacitor is charged by a current source that proportional to the switching frequency resulting in a ramp rate of 50mV per percent duty cycle. For example, if the steady-state operating switch node duty cycle is 10%, then the internal ramp amplitude is typically 500mV from the starting point (or floor) to the crossing of the EAOUT control voltage. When the PWM ramp voltage exceeds the Error Amplifier’s output voltage, the RSFF is reset. This turns off the high side driver, turns on the low side driver, and discharges the PWM ramp to 0.6V until the next clock pulse. Page 11 of 29 09/26/05 IR3094PBF 50% INTERNAL OSCILLATOR RAMP DUTY CYCLE CLK1 CLK2 CLK3 RAMP3 MIN DUTY CYCLE ADJUST EAOUT RAMP3 FIXED RAMP1 RAMP3 MAX DUTY CYCLE ADJUST RAMP2 0.6V RAMP1 SLOPE = 50mV / % DC THE SHARE ADJUST ERROR AMPLIFIER CAN CHANGE THE PULSE WIDTH OF RAMPS 2 & 3 FROM 0.5 x RAMP1 TO 2.0 X RAMP1 TO FORCE CURRENT SHARING. Figure 3 – 3 Phase Oscillator and PWM Waveforms The RSFF is reset dominant allowing both phases to go to zero duty cycle within a few tens of nanoseconds in response to a load step decrease. Phases can overlap and go to 100% duty cycle in response to a load step increase with turn-on gated by the clock pulses. An Error Amplifier output voltage greater than the common mode input range of the PWM comparator results in 100% duty cycle regardless of the voltage of the PWM ramp. This arrangement guarantees the Error Amplifier is always in control and can demand 0 to 100% duty cycle as required. It also favors response to a load step decrease which is appropriate given the low output to input voltage ratio of most systems. The inductor current will increase much more rapidly than decrease in response to load transients. This control method is designed to provide “single cycle transient response” where the inductor current changes in response to load transients within a single switching cycle maximizing the effectiveness of the power train and minimizing the output capacitor requirements. Body Braking TM In a conventional synchronous buck converter, the minimum time required to reduce the current in the inductor in response to a load step decrease is; TSLEW = [L x (IMAX - IMIN)] / Vout (1) The slew rate of the inductor current can be significantly increased by turning off the synchronous rectifier in response to a load step decrease. The switch node voltage is then forced to decrease until conduction of the synchronous rectifier’s body diode occurs. This increases the voltage across the inductor from Vout to Vout + VBODY DIODE. The minimum time required to reduce the current in the inductor in response to a load transient decrease is now; TSLEW = [L x (IMAX - IMIN)] / (Vout + VBODY DIODE) Page 12 of 29 (2) 09/26/05 IR3094PBF Since the voltage drop in the body diode is often higher than output voltage, the inductor current slew rate can be increased by 2X or more. This patent pending technique is referred to as “body braking” and is accomplished through the “0% Duty Cycle Comparator”. If the Error Amplifier’s output voltage drops below 0.575V, this comparator turns off the low side gate driver. Figure 4 depicts PWM operating waveforms under various conditions CLK1 PULSE EAOUT PWM Ramp1 0.6V 0.575V GATEH1 GATEL1 STEADY-STATE OPERATION DUTY CYCLE INCREASE DUE TO LOAD INCREASE DUTY CYCLE DECREASE DUE TO LOAD DECREASE (BODY BRAKING) OR FAULT STEADY-STATE OPERATION Figure 4 – PWM Operating Waveforms Current Sense Amplifier A high speed differential current sense amplifier is shown in Figure 5. Its gain decreases with increasing temperature and is nominally 24 at 25ºC and 20.9 at 125ºC (-1400 ppm/ºC). This reduction of gain tends to compensate the 3850 ppm/ºC increase in inductor DCR. Since in most designs the IR3094 IC junction is hotter than the inductors these two effects tend to cancel such that no additional temperature compensation of the load line is required. The current sense amplifier can accept positive differential input up to 75mV and negative up to -25mV before clipping. The output of the current sense amplifier is summed with the VREF voltage which is used for over current protection, voltage positioning and current sharing. vL iL CO CSA L RL Rs Cs Vo Co vc Figure 5 – Inductor Current Sensing and Current Sense Amplifier Page 13 of 29 09/26/05 IR3094PBF Power-up in Non-Synchronous Mode The SYNC LATCH is set by either a UVLO or a Low Enable fault at the beginning of the power-up cycle, keeping all three low side gate drivers low. The SYNC LATCH is then reset once the FB pin exceeds 78% of VREF to release the low side gate drive control to the Error-Amp. SCOMP preconditioning is also released at this time. NonSynchronous startup helps preventing negative inductor current until current sharing is stabilized. VCC Under Voltage Lockout (UVLO) The VCC UVLO function monitors the IR3094’s VCC supply pin and ensures enough voltage is available to power the internal circuitry. During power-up the fault latch is reset when VCC exceeds 7.5V and all other faults are cleared. The fault latch is set when VCC drops below 7.0V and SS/DEL is below 3.75V. 5VUVL Under Voltage Lockout (5VUVL) The 5VUVL function is provided for converters using a separate voltage supply other than VCC for gate driver bias. The 5VUVL comparator prevents operation by discharging SS/DEL below 3.75V to force EAOUT low. The 5VUVL comparator has an OK threshold of 4.36V ensuring adequate gate drive voltage is present and a fault threshold of 4.17V. Power Good Output The PWRGD pin is an open-collector output and should be pulled up to a voltage source through a resistor. During soft start, the PWRGD remains low until the output voltage is in regulation and SS/DEL is above 3.75V. The PWRGD pin becomes low if the fault latch is set. A high level at the PWRGD pin indicates that the converter is in operation and has no fault, but does not ensure the output voltage is within the specification. Output voltage regulation within the design limits can logically be assured however, assuming no component failure in the system. Tri-State Gate Drivers The GATELX drivers can pull down up to 3.5A peak current and source up to 1.5A. The GATEHX drivers can source and sink up to 1.5A peak current. An adaptive non-overlap circuit monitors the voltage on the GATEHX and GATELX pins to prevent MOSFET shoot-through current while minimizing body diode conduction. The Error Amplifier output of the Control IC drives low in response to any fault condition such as VCC input under voltage or output overload. The 0% duty cycle comparator detects this and drives both gate outputs low. This tristate operation prevents negative inductor current and negative output voltage during power-down. The Gate Drivers revert to a high impedance “off” state at VCCLX and VCCHX supply voltages below the normal operating range. An 80kUHVLVWRULVFRQQHFWHGDFURVVWKH*$7(;DQG3*1';SLQVWRSUHYHQWWKH*$7(;YROWDJH from rising due to leakage or other cause under these conditions. Over Voltage Protection (OVP) The output Over-Voltage Protection comparator monitors the output voltage through the OVPSNS pin, the positive remote sense point. If OVPSNS exceeds VREF plus 150mV, the OVP LATCH will be set. This will set the fault latch immediately pulling the Error Amplifier’s output low, reset the PWM latch to fully turn-off the high side MOSFETs and turn-on the low side MOSFETs within approximately 350ns. The low side MOSFETs will remain ON until the OVP LATCH is reset by recycling VCC. OVPSNS exceeding VREF by 150mV also activates 75uA sources on the OVP pin. The lower MOSFETs alone can not clamp the output voltage however an SCR or MOSFET could be triggered with the OVP pin to prevent processor damage. If powering up with a high side MOSFET short, the OVP flag is activated and the OVP LATCH is set with as little VCC supply voltage as possible. The OVPSNS pin is compared against both VCC and 5VUVL for OVP conditions at power-up. VCC is monitored for conversion off 12V, 5VUVL is monitored for conversion off 5V. The OVP pin flags a voltage greater than 0.48V with supply voltages as low as 1.0V. This headroom voltage varies inversely with temperature. An external comparator can be used to disable the silver box, activate a crowbar, or supply source. Page 14 of 29 09/26/05 IR3094PBF APPLICATIONS INFORMATION OVP ENABLE VIN VOUT+ CCS1 RCS1 C5VREF CBST1 1 2 NC NC NC NC 5VREF OVPSNS ENABLE OVP CSINP1 CSINM1 NC VCCH1 L1 RROSC RFB NC NC ROSC VOSNSOCSET VREF VDRP FB EAOUT SS/DEL SCOMP2 SCOMP3 RREF CREF ROCSET RDRP CSC3 CSC2 CSS VOUT SENSE+ CBST2 RSC3 1 2 CCS2 VOUT+ COUT GND VIN CBIAS RSET L2 RCS2 Cosns- RSC2 VIN 48LD MLPQ VIN LGND SETBIAS VCC CSINP3 CSINM3 BIASOUT PWRGD CSINP2 CSINM2 NC VCCL3 GATEL3 RCOMP CCOMP IR3094 GATEH1 PGND1 GATEL1 VCCL1_2 5VUVL GATEL2 PGND2 GATEH2 VCCH2 VCCH3 GATEH3 PGND3 VIN CBST3 1 L3 VOUT SENSE2 CVCC CVIN RCS3 CCS3 GND PGOOD Figure 6 – System Diagram Oscillator Resistor RROSC The oscillator frequency is programmable from 100kHz to 540kHz with an external resistor RROSC as shown in Figure 6. The Oscillator generates an internal 50% duty cycle sawtooth signal (Figure 3.) that is used to generate 120° out-of-phase timing pulses to set Phase 1,2 and 3 RS flip-flops. Once the switching frequency is chosen, RROSC can be determined from the curve in the Typical Operating Characteristics Section. Soft Start, Over-Current Fault Delay, and Hiccup Mode The IR3094 has a programmable soft-start function to limit the surge current during converter power-up. A capacitor connected between the SS/DEL and LGND pins controls soft start timing as well as over-current protection delay and hiccup mode timing. Figure 8 depicts the various operating modes of the SS/DEL function. Under a no fault condition, the SS/DEL capacitor will charge. The SS/DEL charge soft-start duration is controlled by a 60uA charge current which charges CSS up to 4.0V. The Error Amplifier output is clamped low until SS/DEL reaches 1.1V. The Error Amplifier will then regulate the converter’s output voltage to match the SS/DEL voltage less the 1.1V offset until it reaches the level determined by the VREF voltage. The PWRGD signal is asserted once the SS/DEL voltage exceeds 3.75V. Four different faults will immediately cause SS/DEL to begin discharging and set the Fault Latch once SS/DEL is below 3.75V; Page 15 of 29 09/26/05 IR3094PBF 1. 2. 3. 4. VCC Under Voltage Lock Out 5VUVL Under Voltage Lock Out Low Enable pin Over Current condition. A delay is included if any of the four fault conditions occurs after a successful soft start sequence. This is required since momentary faults can occur as part of normal operation due to load transients such as exciting an overcurrent condition. If any fault occurs during normal operation, the SS/DEL capacitor will discharge through a 55uA current sink but will not set the fault latch immediately. If the fault condition persists long enough for the SS/DEL capacitor to discharge below the 3.75V threshold of the delay comparator, the Fault latch will be set pulling the Error Amplifier’s output low, inhibiting switching and de-asserting the PWRGD signal. The SS/DEL capacitor is then discharged through a 6uA discharge current resulting in a long hiccup duration. The SS/DEL capacitor will continue to discharge until it reaches 0.265V where the fault latch is reset allowing a normal soft start to occur. If a fault condition is again encountered during the soft start cycle, the fault latch will be set without any delay and hiccup mode will begin. During hiccup mode the 10 to 1 charge to discharge ratio results in a 9.1% hiccup mode duty cycle regardless of at what point a fault condition occurs. OVP fault immediately sets the fault latch causing SS/DEL to begin to discharge and this fault can only be cleared by cycling power to the IR3094 on and off. If SS/DEL pin is pulled below 0.8V, the converter can be disabled. 7.0V UVLO VCC (12V) 4.36V 5VUVL SS/DEL 3.75V 1.1V VOUT PWRGD OCP THRESHOLD IOUT START-UP (5VUVL GATES FAULT MODE) NORMAL OPERATION (VOUT CHANGES DUE TO LOAD AND VID CHANGES) OCP DELAY HICCUP OVER-CURRENT PROTECTION RE-START AFTER OCP CLEARS POWER-DOWN (VCC GATES FAULT MODE) Figure 7 – Operating Waveforms Soft-start delay time tSSDEL is the time SS/DEL charged up to 1.1V. After that the error amplifier output is released to allow the soft start. The soft start time tSS represents the time during which converter output voltage rises from zero to VO. tSS can be programmed by CSS using equation (3). C SS Page 16 of 29 I CHG * t SS VO 60 *10 6 * t SS VO (3) 09/26/05 IR3094PBF Once CSS is chosen, the soft start delay time tSSDEL, the over-current fault latch delay time tOCDEL, and the delay time tVccPG from output voltage (VO) in regulation to Power Good are fixed and shown in equation (4), (5) and (6) respectively. t SSDEL C SS * 'V I CHG C SS *1.1 60 *10 6 (4) t OCDEL C SS * 'V I DISCHG C SS * 0.25 61*10 6 (5) tVccPG C SS * 'V I CHG C SS * (3.75 VO 1.1) 60 *10 6 (6) VREF Compensation Network RREF and CREF A RC network tied between VREF pin and VOSENS- is needed to compensate VREF circuit. VREF should come up earlier than SS/DEL pin charged up to 3.75V. For save estimation, use half of the soft start time that is 0.5*tSS as the VREF voltage establishing time. Use equation (7) and (8) to determine RREF and CREF where VREF source current ISOURCE is determained by RROSC and can be found using the curve in the TYPICAL OPERATING CHARACTERISTICS section. C REF I SOURCE * 0.5 * t SS VREF (7) RREF 3.2 10 15 0.5 2 C REF (8) Over Current Protection (OCP) The current limit threshold is set by a resistor connected between the OCSET and VREF pins. If the average Current Sense Amplifier output plus VREF voltage exceeds the OCSET voltage, the over-current protection is triggered. A delay is included if an over-current condition occurs after a successful soft-start sequence. This is required since over-current conditions can occur as part of normal operation due to load transients. If an over-current fault occurs during normal operation, the Over Current Comparator will initiate the discharge of the capacitor at SS/DEL but will not set the fault latch immediately. If the over-current condition persists long enough for the SS/DEL capacitor to discharge below the 245mV offset of the delay comparator, the Fault latch will be set pulling the Error Amplifier’s output low inhibiting switching in the phase ICs and de-asserting the PWRGD signal. The hiccup mode duty cycle of over current protection is determined by the fixed 10:1 ratio of the charge to discharge current. The inductor DC resistance RL is utilized to sense the inductor current. The current limit threshold is set by a resistor ROCSET connected between the OCSET and VREF pins, as shown in Fig6. ILIMIT is the required over current limit. IOCSET, the bias current of OCSET pin, is set by RROSC and is determined by the curve in the Typical Operating Characteristics Section. OCP need to satisfy the high temperature condition. RL_MAX and RL_ROOM are the inductor DCR at maximum temperature TL_MAX and room temperature T_ROOM respectively, the maximum inductor DCR can be calculated from Equation (9) Page 17 of 29 09/26/05 IR3094PBF RL _ MAX RL _ ROOM [1 3850 *10 6 (TL _ MAX TROOM )] (9) The current sense amplifier gain of IR3094 decreases with temperature at the rate of 1400 PPM, which compensates part of the inductor DCR increase. The minimum current sense amplifier gain at the maximum IC temperature TIC_MAX is calculated from Equation (10). GCS _ MIN GCS _ ROOM [1 1400 *10 6 (TIC _ MAX TROOM )] (10) ROCSET can be calculated by the following equation (11). ROCSET I LIMIT RL _ MAX ) GCS _ MIN / I OCSET 3 ( (11) Output Voltage Droop In some of the applications, output voltage droop is needed to minimize output voltage deviations during load transients and reduce power dissipation of the load when it is drawing maximum current. The voltage at the VDRP pin is an average of three phase Current Sense Amplifiers and represents the sum of the VREF voltage and the average inductor current of all the phases. The VDRP pin is connected to the FB pin through the RDRP resistor, see figure 6. The Error Amplifier forces the voltage on the FB pin to equal VREF through the power supply loop therefore the current through RDRP is equal to (VDRP-VREF) / RDRP. As the load current increases, the VDRP voltage increases accordingly which results in an increase in RFB current, positioning the output regulated voltage lower thus making the output voltage reduction proportional to an increase in load current. The droop impedance or output impedance of the converter can thus be programmed by the resistor RDRP. The offset and slope of the converter output impedance are independent of the VREF voltage. The VDRP pin voltage represents the average current of the converter plus the 0.84V reference voltage. The load current can be retrieved by subtracting the VREF voltage from the VDRP voltage. The converter voltage will be lowered by RO*IO, where RO is the required output impedance of the converter. RDRP is determined by Equation (12) R DRP R FB R L _ MAX GCS _ MIN n RO (12) Lossless Average Inductor Current Sensing Inductor current can be sensed by connecting a series resistor and a capacitor network in parallel with the inductor and measuring the voltage across the capacitor. The equation of the sensing network is, vC ( s ) v L ( s) 1 1 sRS C S i L ( s) R L sL 1 sRS C S (13) Usually the resistor Rcs and capacitor Ccs are chosen so that the time constant of Rcs and Ccs equals the time constant of the inductor which is the inductance L over the inductor DCR. If the two time constants match, the voltage across Ccs is proportional to the current through L, and the sense circuit can be treated as if only a sense resistor with the value of RL was used. The mismatch of the time constants does not affect the measurement of inductor DC current, but affects the AC component of the inductor current. Page 18 of 29 09/26/05 IR3094PBF The advantage of sensing the inductor current versus high side or low side sensing is that actual output current being delivered to the load is obtained rather than peak or sampled information about the switch currents. The output voltage can be positioned to meet a load line based on real time information. Except for a sense resistor in series with the inductor, this is the only sense method that can support a single cycle transient response. Other methods provide no information during either load increase (low side sensing) or load decrease (high side sensing). An additional problem associated with peak or valley current mode control for voltage positioning is that they suffer from peak-to-average errors. These errors will show in many ways but one example is the effect of frequency variation. If the frequency of a particular unit is 10% low, the peak to peak inductor current will be 10% larger and the output impedance of the converter will drop by about 10%. Variations in inductance, current sense amplifier bandwidth, PWM prop delay, any added slope compensation, input voltage, and output voltage are all additional sources of peak-to-average errors. Measure the inductance L and the inductor DC resistance RL. Pre-select the capacitor CCS and calculate RCS as follows. RCS L RL CCS (14) Inductor DCR Temperature Correction If the Current Sense Amplifier temperature dependent gain is not adequate to compensate the inductor DCR TC, a negative temperature coefficient (NTC) thermistor can be added. The thermistor should be placed close to the inductor and connected in parallel with the feedback resistor, as shown in Figure 8. The resistor in series with the thermistor is used to reduce the nonlinearity of the thermistor. Figure 8 - Temperature compensation of inductor DCR Remote Voltage Sensing To compensate for impedance in the ground plane, the VOSNS- pin is used for remote sensing and connects directly to the load. The VREF voltage is referenced to VOSNS- to avoid additional error terms or delay related to a separate differential amplifier. The capacitor connecting the VREF and VOSNS- pins ensure that high speed transients are fed directly into the Error Amplifier without delay. Page 19 of 29 09/26/05 IR3094PBF Master-Slave Current Share Loop Current sharing between phases of the converter is achieved by a Master-Slave current share loop topology. The output of the Phase 1 Current Sense Amplifier sets the reference for the Share Adjust Error Amplifiers. Each Share Adjust Error Amplifier adjusts the duty cycle of its respective PWM Ramp and to force its input error to zero compared to the master Phase 1, resulting in accurate current sharing. The maximum and minimum duty cycle adjust range of Ramps 2 & 3 compared to Ramp1 has been limited to a minimum of 0.5x and a maximum of 2.0x typical (see Figure 3.). The crossover frequency of the current share loop can be programmed with a capacitor at the SCOMPX pin so that the share loop does not interact with the output voltage loop. The SCOMPX capacitor is driven by a trans-conductance stage capable of sourcing and sinking 22uA. The duty cycle of Ramps 2 & 3 inversely tracks the voltage on their SCOMPX pin; if V(SCOMP2) increases, Ramp2’s slope will increase and the effective duty cycle will decrease resulting in a reduction in Phase 2’s output current. Due to the limited 22uA source current, an SCOMPX pre-conditon circuit has been included to pre-condition V(SCOMPX) so that the duty cycle of Ramps 2 & 3 are equal to Ramp1 prior to any GATEHX high pulses. The pre-condition circuit can source/sink 360uA. The SYNC LATCH (see Figure 1) releases the pre-condition circuit once FB reaches 78% of VREF. Set BIASOUT voltage BIASOUT pin provides a 150mA open-loop regulated voltage for GATE drive bias. The voltage is set by SETBIAS through an external resistor Rset connecting between SETBIAS pin and ground. Bias current ISETBIAS is a function of ROSC. Rset is chosen by equation (15). VFD in the equation is the forward voltage drop across the Bootstrap diode. RSET V BIASOUT V FD I SETBIAS (15) Compensation of the Current Share Loop The crossover frequency of the current share loop should be at least one decade lower than that of the voltage loop in order to eliminate the interaction between the two loops. A 22nF capacitor from SCOMP to LGND is good for most of the applications. If necessary have a resistor in series with the Csc to make the current loop a little bit faster. Compensation of Voltage Loop The selection of compensation types depends on the output capacitors used in the converter. For the applications using Electrolytic, Polymer or AL-Polymer capacitors and running at lower frequency, type II compensation shown in Figure 9(a) is usually enough. While for the applications using only ceramic capacitors and running at higher frequency, type III compensation shown in Figure 9(b) is preferred. For applications without voltage droop, the compensation is the same as for the regular voltage mode control. For converter using Polymer, AL-Polymer, and ceramic capacitors, which have much higher ESR zero frequency, type III compensation is required as shown in Figure 9(b) with RDRP and CDRP removed. Page 20 of 29 09/26/05 IR3094PBF CCP1 CCP1 RFB VO+ RCP VO+ RFB FB CCP EAOUT VDRP RDRP RFB1 VREF VDAC EAOUT CFB RDRP VDRP RCP FB VREF VDAC CCP EAOUT EAOUT + CDRP + (a) Type II compensation (b) Type III compensation Figure 9. Voltage loop compensation networks Type II Compensation for Voltage Droop Applications Determine the compensation at no load, the worst case condition. Choose the crossover frequency fc between 1/10 and 1/5 of the switching frequency per phase. Assume the time constant of the resistor and capacitor across the output inductors matches that of the inductor, and determine RCP and CCP from (16) and (17), where LE and CE are the equivalent inductance of output inductors and the equivalent capacitance of output capacitors respectively. RCP C CP ( 2S f C ) 2 L E C E R FB 5 V I * 1 (2S * f C * C * RC ) 2 10 L E C E (16) (17) RCP CCP1 is optional and may be needed in some applications to reduce the jitter caused by the high frequency noise. A ceramic capacitor between 10pF and 220pF is usually enough. Type III Compensation for Voltage Droop Applications Determine the compensation at no load, the worst case condition. Assume the time constant of the resistor and capacitor across the output inductors matches that of the inductor, the crossover frequency and phase margin of the voltage loop can be estimated by (18) and (19), where RLE is the equivalent resistance of inductor DCR. f C1 R DRP 2S * C E GCS * R FB R LE T C1 90 A tan(0.5) 180 S (18) (19) Choose the desired crossover frequency fc around fc1 estimated by (18) or choose fc between 1/10 and 1/5 of the switching frequency per phase, and select the components to ensure the slope of close loop gain is -20dB /Dec around the crossover frequency. Choose resistor RFB1 according to (20), and determine CFB and CDRP from (21) and (22). Page 21 of 29 09/26/05 IR3094PBF R FB1 1 R FB 2 R FB1 to 2 R FB 3 1 4S f C R FB1 C FB C DRP (20) (21) ( R FB R FB1 ) C FB R DRP (22) RCP and CCP have limited effect on the crossover frequency, and are used only to fine tune the crossover frequency and transient load response. Determine RCP and CCP from (23) and (24). ( 2S f C ) 2 L E C E R FB 5 VI RCP C CP 10 L E C E RCP (23) (24) CCP1 is optional and may be needed in some applications to reduce the jitter caused by the high frequency noise. A ceramic capacitor between 10pF and 220pF is usually enough. Type III Compensation for No Droop Applications Resistor RDRP and capacitor CDRP are not needed. Choose the crossover frequency fc between 1/10 and 1/5 of the switching frequency per phase and select the desired phasHPDUJLQ F&DOFXODWH.IDFWRUIURP25), and determine the component values based on (26) to (30), K Page 22 of 29 S T tan[ ( C 1.5)] 4 180 ( 2S L E C E f C ) 2 5 (25) RCP R FB C CP K 2S f C RCP (27) C CP1 1 2S f C K RCP (28) VI K (26) C FB K 2S f C R FB (29) R FB1 1 2S f C K C FB (30) 09/26/05 IR3094PBF MathCAD file to estimate the power dissipation of the IC The full featured Control IC IR3094 contain both Control and 3 phase Gate Drive functions. It also has the adjustable voltage bias regulator inside to provide MOSFET Drive Voltage. For the thermal consideration, this Mathcad file step by step shows how to estimate the power dissipation of IR3094 . Initial Conditions: n 3 No.of Phases: IC Supply Voltage: Vcc 12 ( V) Icq 35 ( mA) , IC Supply Current(quiescent): Total High side Driver VCCH supply current(quiescent): Iqh 5 n ( mA) Total Low side Driver VCCL supply Current(quiescent): Biasout Voltage: Vbias 7.5 ( V) Iql 5 n ( mA) fsw 450 ( kHz) Switching Frequency per phase: Thermal Impedance of IC: T JA 27 (o C/W) The data from the selected MOSFETs: nc 1 ControI FET IR6637, Number of Control FET per phase: Control FET total gate charge: Qgc 15 ( nC) Synchronous FET IR6612, Number of sync. FET per phase: Sync FET total gate charge: Qgs 45 ( nC) ns 1 Power Dissipation: The IC will have less power dissipation if using external gate driver supply. For the worst case estimation, assuming using the bias regulator for all the gate drive supply voltage. 1. Quiescent Power dissipation Total Quiescent Power Dissipation: 3 Pq ( Icq Iqh Iql) Vcc 10 Pq (W ) 2. The Power Loss to drive the gate of the MOSFETs With the assumption of the low MOSFET gate resistances, most gate drive losses are dissipated in the driver circuit. Pdrv Vbias fsw 10 n ª¬( nc Qgc ns Qgs ) 10 9º 3 ¼ Pdrv (W ) Where the Ig fsw 103 n ( nc Qgc ns Qgs ) 10 9 term in the equation gives the total average bias current required to drive all the MOSFETs. 3. The bias regulator Power Loss to supply driving the MOSFETs Preg ( Vcc Vbias) Ig Preg (W ) 4. Total Power Dissipation of the IC: Pdiss Pq Pdrv Preg And the total Junction temperature rising is: Page 23 of 29 Pdiss Pdiss T JA (W ) (oC) 09/26/05 IR3094PBF LAYOUT GUIDELINES The following layout guidelines are recommended to reduce the parasitic inductance and resistance of the PCB layout, therefore minimizing the noise coupled to the IC. Refer to the schematic in Figure 6 – System Diagram. x Dedicate at least one inner layer of the PCB as power ground plane (PGND). x The center pad of IC must be connected to ground plane (PGND) using the recommended via pattern shown in “Package Dimensions”. x The IC’s PGND1, 2, 3 and LGND should connect to the center pad under IC. x The following components must be grounded directly to the LGND pin on the IC using a ground plane on the component side of PCB: CSS, RSC2, RSC3, RSET, CVCC and C5VREF. The LGND should only be connected to ground plan on the center pad under IC x Place the decoupling capacitors CVCC and CBIAS as close as possible to the VCC and VCCL1_2, VCCL3 pins. The ground side of CBIAS should not be connected to LGND and it should directly ground through vias. x The following components should be placed as close as possible to the respective pins on the IC: RROSC, ROCSET, CREF, RREF, CSS, CSC2, RSC2, CSC3, RSC3, RSET. x Place current sense capacitors CCS1, 2, 3 and resistors RCS1, 2, 3 as close as possible to CSINP1, 2, 3 pins of IC and route the two current sense signals in pairs connecting to the IC. The current sense signals should be located away from gate drive signals and switch nodes. x Use Kelvin connections to route the current sense traces to each individual phase inductor, in order to achieve good current share between phases. x Place the input decoupling capacitors closer to the drain of top MOSFET and the source of the bottom MOSFET. If possible, Use multiple smaller value ceramic caps instead of one big cap, or use low inductance type of ceramic cap, to reduce the parasitic inductance. x Route the high current paths using wide and short traces or polygons. Use multiple vias for connections between layers. x - The symmetry of the following connections from phase to phase is important for proper operation: The Kelvin connections of the current sense signals to inductors. The gate drive signals from the IC to the MOSFETS. The polygon shape of switching nodes. Page 24 of 29 09/26/05 IR3094PBF PCB AND STENCIL DESIGN METHODOLOGY x x x 7x7 48 Lead 0.5mm pitch MLPQ See Figures 10-12. PCB Metal Design (0.5mm Pitch Leads) 1. Lead land width should be equal to nominal part lead width. The minimum lead to lead spacing should be PPWRPLQLPL]HVKRUWLQJ 2. Lead land length should be equal to maximum part lead length + 0.2 mm outboard extension + 0.05mm inboard extension. The outboard extension ensures a large and inspectable toe fillet, and the inboard extension will accommodate any part misalignment and ensure a fillet. 3. Center pad land length and width should be = maximum part pad length and width. However, the minimum metal to metal spacing should be PPR]&RSSHUPPIRUR]&RSSHU and PPIRUR]&RSSHU 4. Sixteen 0.30mm diameter vias shall be placed in the pad land spaced at 1.2mm, and connected to ground to minimize the noise effect on the IC, and to transfer heat to the PCB. PCB Solder Resist Design (0.5mm Pitch Leads) 1. Lead lands. The solder resist should be pulled away from the metal lead lands by a minimum of 0.060mm. The solder resist mis-alignment is a maximum of 0.050mm and it is recommended that the lead lands are all NSMD. Therefore pulling the S/R 0.060mm will always ensure NSMD pads. 2. The minimum solder resist width is 0.13mm, therefore it is recommended that the solder resist is completely removed from between the lead lands forming a single opening for each “group” of lead lands. 3. At the inside corner of the solder resist where the lead land groups meet, it is recommended to provide a fillet so a solder resist width of PPUHPDLQV 4. Land Pad. The land pad should be SMD, with a minimum overlap of the solder resist onto the copper of 0.060mm to accommodate solder resist mis-alignment. In 0.5mm pitch cases it is allowable to have the solder resist opening for the land pad to be smaller than the part pad. 5. Ensure that the solder resist in-between the lead lands and the pad land is PPGXHWR the high aspect ratio of the solder resist strip separating the lead lands from the pad land. 6. The single via in the land pad should be tented with solder resist 0.4mm diameter, or 0.1mm larger than the diameter of the via. Stencil Design (0.5mm Pitch Leads) 1. The stencil apertures for the lead lands should be approximately 80% of the area of the lead lands. Reducing the amount of solder deposited will minimize the occurrence of lead shorts. Since for 0.5mm pitch devices the leads are only 0.25mm wide, the stencil apertures should not be made narrower; openings in stencils < 0.25mm wide are difficult to maintain repeatable solder release. 2. The stencil lead land apertures should therefore be shortened in length by 80% and centered on the lead land. 3. The center land pad aperture should be striped with 0.25mm wide openings and spaces to deposit approximately 50% area of solder on the center pad. If too much solder is deposited on the center land pad the part will float and the lead lands will be open. 4. The maximum length and width of the center land pad stencil aperture should be equal to the solder resist opening minus an annular 0.2mm pull back to decrease the incidence of shorting the center land to the lead lands when the part is pushed into the solder paste. Page 25 of 29 09/26/05 IR3094PBF Figure 10. PCB metal and solder resist. Page 26 of 29 09/26/05 IR3094PBF Figure 11. PCB metal and component placement. Page 27 of 29 09/26/05 IR3094PBF Figure 12. Stencil design. Page 28 of 29 09/26/05 IR3094PBF PACKAGE DIMENSIONS Data and specifications subject to change without notice. This product has been designed and qualified for the Consumer market. Qualification Standards can be found on IR’s Web site. IR WORLD HEADQUARTERS: 233 Kansas St., El Segundo, California 90245, USA Tel: (310) 252-7105 TAC Fax: (310) 252-7903 Visit us at www.irf.com for sales contact information www.irf.com Page 29 of 29 09/26/05