LT1424-9 Isolated Flyback Switching Regulator with 9V Output U FEATURES ■ ■ ■ ■ ■ ■ ■ DESCRIPTIO No Transformer “Third Winding” or Optoisolator Required Application Circuit Meets PCMCIA Type II Height Requirement Fixed, Application Specific 9V Output Voltage Regulation Maintained Well into Discontinuous Mode (Light Load) Load Compensation Provides Excellent Load Regulation Available in 8-Pin PDIP and SO Packages Operating Frequency: 285kHz U APPLICATIO S ■ The LT ®1424-9 is a monolithic high power switching regulator specifically designed for the isolated flyback topology. No “third winding” or optoisolator is required; the integrated circuit senses the isolated output voltage directly from the primary side flyback waveform. A high current, high efficiency switch is included on the die along with all oscillator, control and protection circuitry. The LT1424-9 operates with input supply voltages from 3V to 20V and draws only 7mA quiescent current. It can deliver up to 200mA at 9V with no external power devices. By utilizing current mode switching techniques, it provides excellent AC and DC line regulation. The LT1424-9 has a number of features not found on other switching regulator ICs. Its unique control circuitry can maintain regulation well into discontinuous mode. Load compensation circuitry allows for improved load regulation. An externally activated shutdown mode reduces total supply current to 20µA typical for standby operation. Ethernet Isolated 5V to – 9V Converter , LTC and LT are registered trademarks of Linear Technology Corporation. U TYPICAL APPLICATIO – 9V PCMCIA Type II Isolated LAN Supply (2.41mm Maximum Component Height) ISOLATION BARRIER R2 75Ω C6 220pF 5V C1 10µF 25V C2 10µF 25V D1 1N5248 0.1µF C5 220pF LT1424-9 2 1 100k 47pF 1000pF 3 4 VC RCCOMP SHDN VIN SYNC VSW SGND PGND 8 OUT COM 1 3 T1 D2 MBR0540T4 7 2 6 5 C3 10µF 25V C4 10µF 25V 1.8k 4 1:1 0.1µF INPUT COM MBRS130LT3 R1 75Ω 1424 TA01 –9V 200mA C1, C2, C3, C4: MARCON THCS50E1E106Z CERAMIC CAPACITOR, SIZE 1812. (847) 696-2000 T1: COILTRONICS CTX02-13483 1 LT1424-9 W U U U W W W ABSOLUTE MAXIMUM RATINGS PACKAGE/ORDER INFORMATION (Note 1) Supply Voltage ........................................................ 20V Switch Voltage ......................................................... 35V SHDN, SYNC Pin Voltage ........................................... 7V Operating Junction Temperature Range Commercial .......................................... 0°C to 125°C Industrial ......................................... – 40°C to 125°C Storage Temperature Range ................. – 65°C to 150°C Lead Temperature (Soldering, 10 sec).................. 300°C ORDER PART NUMBER TOP VIEW SHDN 1 VC 2 8 RCCOMP SYNC 3 6 VSW SGND 4 5 PGND N8 PACKAGE 8-LEAD PDIP LT1424CN8-9 LT1424CS8-9 LT1424IN8-9 LT1424IS8-9 7 VIN S8 PACKAGE 8-LEAD PLASTIC SO S8 PART MARKING TJMAX = 145°C, θJA = 130°C/ W (N) TJMAX = 145°C, θJA = 110°C/ W (S) 14249 14249I Consult factory for Military grade parts. ELECTRICAL CHARACTERISTICS The ● denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN = 5V, VSW Open, VC = 1.4V, unless otherwise specified. SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS Power Supply VIN(MIN) Minimum Input Voltage ● 2.8 3.1 V ICC Supply Current ● 7.0 9.5 mA Shutdown Mode Supply Current ● 15 40 µA Shutdown Mode Threshold ● 0.3 0.9 1.3 V ● 9.00 8.90 9.15 9.15 9.30 9.40 V V ● 400 1000 1600 µmho ● 30 50 80 Feedback Amplifier VREF gm Reference Voltage Feedback Amplifier Transconductance Measured at VSW Pin (Note 2) ∆IC = ±10µA (Note 2) ISOURCE, ISINK Feedback Amplifier Source or Sink Current VCL Feedback Amplifier Clamp Voltage 1.9 µA V Reference Voltage/Current Line Regulation 5V ≤ VIN ≤ 18V Voltage Gain (Note 3) BV Output Switch Breakdown Voltage IC = 5mA ● V(VSW) Output Switch ON Voltage ISW = 1A ● 0.55 0.85 V ILIM Switch Current Limit Duty Cycle = 50%, 0°C ≤ TJ ≤ 125°C Duty Cycle = 50%, – 40°C ≤ TJ ≤ 125°C Duty Cycle = 80% ● ● 1.35 1.20 1.6 1.6 1.3 1.95 1.95 A A A ● 0.95 0.85 1.2 1.2 1.3 1.4 V V 0.01 ● 0.04 %/V 500 V/V 50 V Output Switch 35 Current Amplifier Control Pin Threshold Control Voltage to Switch Transconductance 2 Duty Cycle = Minimum 2 A/V LT1424-9 ELECTRICAL CHARACTERISTICS The ● denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN = 5V, VSW Open, VC = 1.4V, unless otherwise specified. SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS 260 240 285 285 300 320 kHz kHz 200 260 Timing f Switching Frequency ● tON Minimum Switch ON Time tED Flyback Enable Delay Time 150 ns tEN Minimum Flyback Enable Time 180 ns 90 % 1.5 Ω Maximum Switch Duty Cycle 170 ● 85 ns Load Compensation ∆VREF /∆ISW SYNC Function Minimum SYNC Amplitude ● Synchronization Range ● SYNC Pin Input Resistance Note 1: Absolute Maximum Ratings are those values beyond which the life of a device may be impaired. Note 2: VREF is a parameter which is measured at the VSW pin. It differs from the output voltage because it accounts for output diode drop, transformer leakage inductance, etc. Nominal output voltage is 9V in the intended application circuit. 1.5 330 2.2 V 450 kHz 40 kΩ Note 3: Feedback amplifier transconductance is RREF referred. Note 4: Voltage gain is RREF referred. 3 LT1424-9 U W TYPICAL PERFOR A CE CHARACTERISTICS Switch Saturation Voltage vs Switch Current Switch Current Limit vs Duty Cycle Minimum Input Voltage vs Temperature 3.1 2.0 1.2 3.0 125°C 0.8 25°C 0.6 –55°C 0.4 1.5 INPUT VOLTAGE (V) 1.0 SWITCH CURRENT LIMIT (A) SWITCH SATURATION VOLTAGE (V) TA = 25°C 1.0 0.5 0.2 0 0 0.2 1.0 0.4 0.6 0.8 SWITCH CURRENT (A) 1.2 0 1.4 2.6 0 9.20 9.15 9.10 9.05 50 25 75 0 TEMPERATURE (°C) 100 125 20 0 –20 –40 –60 275 270 50 25 75 0 TEMPERATURE (°C) 1.75 1.50 100 125 1424-9 G07 VC THRESHOLD 1.25 1.00 –80 7.5 8.0 8.5 9.0 9.5 10.0 FLYBACK VOLTAGE (V) 0.75 –50 –25 10.5 11.0 50 25 75 0 TEMPERATURE (°C) 100 SHDN Pin Input Current vs Voltage 1 2.50 TA = 25°C 2.25 2.00 1.75 1.50 1.25 1.00 0.75 –50 –25 125 1424-9 G06 SHDN PIN INPUT CURRENT (µA) 280 VC HIGH CLAMP 2.00 1424-9 G05 MINIMUM SYNCHRONIZATION VOLTAGE (VP-P) 285 265 – 50 – 25 2.25 Minimum Synchronization Voltage vs Temperature 290 125 2.50 25°C 125°C –55°C 40 Switching Frequency vs Temperature 295 100 VC Pin Threshold and High Clamp Voltage vs Temperature 60 1424-9 G04 300 50 25 75 0 TEMPERATURE (°C) 1424-9 G03 VC PIN VOLTAGE (V) FEEDBACK AMPLIFIER OUTPUT CURRENT (µA) 9.25 9.00 – 50 – 25 2.4 –50 –25 10 20 30 40 50 60 70 80 90 100 DUTY CYCLE (%) Feedback Amplifier Output Current vs Flyback Voltage 9.30 REFERENCE VOLTAGE (V) 2.7 1424-9 G02 Reference Voltage vs Temperature SWITCHING FREQUENCY (kHz) 2.8 2.5 1424-9 G01 4 2.9 0 –1 –2 –3 –4 50 25 75 0 TEMPERATURE (°C) 100 125 1424-9 G08 0 1 3 4 2 SHDN PIN VOLTAGE (V) 5 1424-9 G09 LT1424-9 U W TYPICAL PERFOR A CE CHARACTERISTICS Flyback Enable Delay Time vs Temperature Minimum Flyback Enable Time vs Temperature 250 275 250 225 250 200 225 225 200 175 150 125 ENABLE TIME (ns) 275 ENABLE DELAY TIME (ns) SWITCH ON TIME (ns) Minimum Switch On Time vs Temperature 175 150 125 100 100 – 50 – 25 50 25 75 0 TEMPERATURE (°C) 100 125 75 – 50 – 25 200 175 150 125 50 25 75 0 TEMPERATURE (°C) 1424-9 G10 100 125 1424-9 G11 100 – 50 – 25 50 25 75 0 TEMPERATURE (°C) 100 125 1424-9 G12 U U U PIN FUNCTIONS SHDN (Pin 1): Shutdown. This pin is used to turn off the regulator and reduce VIN input current to a few tens of microamperes. The SHDN pin can be left floating when unused. PGND (Pin 5): Power Ground. This pin is the emitter of the power switch device and has large currents flowing through it. It should be connected directly to a good quality ground plane. VC (Pin 2): Control Voltage. This pin is the output of the feedback amplifier and the input of the current comparator. Frequency compensation of the overall loop is effected by placing a capacitor between this node and ground. VSW (Pin 6): This is the collector node of the output switch and has large currents flowing through it. Keep the traces to the switching components as short as possible to minimize electromagnetic radiation and voltage spikes. SYNC (Pin 3): Pin to synchronize internal oscillator to external frequency reference. It is directly logic compatible and can be driven with any signal between 10% and 90% duty cycle. If unused, this pin should be tied to ground. VIN (Pin 7): Supply Voltage. Bypass input supply pin with 10µF or more. The part goes into undervoltage lockout when VIN drops below 2.8V. Undervoltage lockout stops switching and pulls the VC pin low. SGND (Pin 4): Signal Ground. This pin is a clean ground. The internal reference and feedback amplifier are referred to it. Keep the ground path connection to the VC compensation capacitor free of large ground currents. RCCOMP (Pin 8): Pin for the External Filter Capacitor for Load Compensation Function. A common 0.1µF ceramic capacitor will suffice. 5 LT1424-9 W BLOCK DIAGRAM VIN 2.6V REGULATOR SHDN VSW FLYBACK ERROR AMPLIFIER 285kHz OSCILLATOR SYNC LOGIC DRIVER RCCOMP LOAD COMPENSATION COMP SGND VC + CURRENT AMPLIFIER GND IS OMITTED FOR CLARITY – RSENSE PGND 1424BD W W FLYBACK ERROR A PLIFIER DIAGRA D1 VIN T1 + • VSW + ISOLATED VOUT C1 • – VIN D2 Q4 RFB IM IFXD VC Q1 ENABLE Q2 Q3 CEXT VBG RREF I IM 1424 EA 6 LT1424-9 WU W TI I G DIAGRA VSW VOLTAGE COLLAPSE DETECT VFLBK 0.80× VFLBK VIN GND SWITCH STATE OFF OFF ON MINIMUM tON FLYBACK AMP STATE ON ENABLE DELAY DISABLED MINIMUM ENABLE TIME ENABLED DISABLED 1424 TD 7 LT1424-9 U OPERATION The LT1424-9 is a current mode switching regulator IC that has been designed specifically for the isolated flyback topology. The special problem normally encountered in such circuits is that information relating to the output voltage on the isolated secondary side of the transformer must be communicated to the primary side in order to maintain regulation. Historically, this has been done with optoisolators or extra transformer windings. Optoisolator circuits waste output power and the extra components they require increase the cost and physical volume of the power supply. Optoisolators can also exhibit trouble due to limited dynamic response (temporal), nonlinearity, unit-to-unit variation and aging over life. Circuits employing extra transformer windings also exhibit deficiencies. The extra winding adds to the transformer’s physical size and cost. Dynamic response is often mediocre. There is usually no method for maintaining load regulation versus load. The LT1424-9 derives its information about the isolated output voltage by examining the primary side flyback pulse waveform. In this manner no optoisolator nor extra transformer winding is required. This IC is a quantum improvement over previous approaches because: target output voltage is directly resistor programmable, regulation is maintained well into discontinuous mode and optional load compensation is available. The Block Diagram shows an overall view of the system. Many of the blocks are similar to those found in traditional designs including: internal bias regulator, oscillator, logic, current amplifier and comparator, driver and output switch. The novel sections include a special flyback error amplifier and a load compensation mechanism. Also, due to the special dynamic requirements of flyback control, the logic system contains additional functionality not found in conventional designs. The RREF, RRFB and ROCOMP resistors in the Block Diagram are application-specific thin-film resistors internal to the LT1424-9. The capacitor connected to the RCCOMP pin is external. The LT1424-9 operates much the same as traditional current mode switchers, the major difference being a different type of error amplifier which derives its feedback 8 information from the flyback pulse. Due to space constraints, this discussion will not reiterate the basics of current mode switcher/controllers and isolated flyback converters. A good source of information on these topics is LTC’s Application Note 19. ERROR AMPLIFIER—PSEUDO DC THEORY Please refer to the simplified diagram of the Flyback Error Amplifier. Operation is as follows: when output switch Q4 turns off, its collector voltage rises above the VIN rail. The amplitude of this flyback pulse, i.e., the difference between it and VIN, is given as: V + VF + (ISEC)(ESR) VFLBK = OUT NSP VF = D1 forward voltage ISEC = Transformer secondary current ESR = Total impedance of secondary circuit NSP = Transformer effective secondary-to-primary turns ratio The flyback voltage is then converted to a current by the action of RFB and Q1. Nearly all of this current flows through resistor RREF to form a ground-referred voltage. This is then compared to the internal bandgap reference by the differential transistor pair Q2/Q3. The collector current from Q2 is mirrored around and subtracted from fixed current source IFXD at the VC pin. An external capacitor integrates this net current to provide the control voltage to set the current mode trip point. The relatively high gain in the overall loop will then cause the voltage at the RREF resistor to be nearly equal to the bandgap reference VBG. The relationship between VFLBK and VBG may then be expressed as: V V α FLBK = BG or, RFB RREF VFLBK = VBG ) )) ) RFB RREF 1 α α = Ratio of Q1 IC to IE VBG = Internal bandgap reference LT1424-9 U OPERATION Combination with the previous VFLBK expression yields an expression for VOUT, in terms of the internal reference, programming resistors, transformer turns ratio and diode forward voltage drop: VOUT = VBG ) )) ) RFB RREF NSP α – VF – ISEC (ESR) Additionally, it includes the effect of nonzero secondary output impedance. See Load Compensation for details. The practical aspects of applying this equation for VOUT are found in the Applications Information section. So far, this has been a pseudo-DC treatment of flyback error amplifier operation. But the flyback signal is a pulse, not a DC level. Provision must be made to enable the flyback amplifier only when the flyback pulse is present. This is accomplished by the dashed line connections to the block labeled “ENABLE”. Timing signals are then required to enable and disable the flyback amplifier. ERROR AMPLIFIER—DYNAMIC THEORY There are several timing signals that are required for proper LT1424-9 operation. Please refer to the Timing Diagram. Minimum Output Switch ON Time The LT1424-9 effects output voltage regulation via flyback pulse action. If the output switch is not turned on at all, there will be no flyback pulse, and output voltage information is no longer available. This would cause irregular loop response and start-up/latchup problems. The solution chosen is to require the output switch to be on for an absolute minimum time per each oscillator cycle. This in turn establishes a minimum load requirement to maintain regulation. See Applications Information section for further details. Enable Delay When the output switch shuts off, the flyback pulse appears. However, it takes a finite time until the transformer primary side voltage waveform approximately rep- resents the output voltage. This is partly due to rise time on the VSW node, but more importantly due to transformer leakage inductance. The latter causes a voltage spike on the primary side not directly related to output voltage. (Some time is also required for internal settling of the feedback amplifier circuitry.) In order to maintain immunity to these phenomena, a fixed delay is introduced between the switch turn-off command and the enabling of the feedback amplifier. This is termed “enable delay”. In certain cases where the leakage spike is not sufficiently settled by the end of the enable delay period, regulation error may result. See Applications Information section for further details. Collapse Detect Once the feedback amplifier is enabled, some mechanism is then required to disable it. This is accomplished by a collapse detect comparator, that compares the flyback voltage (RREF referred) to a fixed reference, nominally 80% of VBG. When the flyback waveform drops below this level, the feedback amplifier is disabled. This action accommodates both continuous and discontinuous mode operation. Minimum Enable Time The feedback amplifier, once enabled, stays enabled for a fixed minimum time period termed “minimum enable time”. This prevents lock-up, especially when the output voltage is abnormally low, e.g., during start-up. The minimum enable time period ensures that the VC node is able to “pump up” and increase the current mode trip point to the level where the collapse detect system exhibits proper operation. The “minimum enable time” often determines the low load level at which output voltage regulation is lost. See Applications Information section for details. Effects of Variable Enable Period It should now be clear that the flyback amplifier is enabled only during a portion of the cycle time. This can vary from the fixed “minimum enable time” described to a maximum of roughly the OFF switch time minus the enable delay 9 LT1424-9 U OPERATION time. Certain parameters of flyback amp behavior will then be directly affected by the variable enable period. These include effective transconductance and VC node slew rate. LOAD COMPENSATION THEORY The LT1424-9 uses the flyback pulse to obtain information about the isolated output voltage. A potential error source is caused by transformer secondary current flow through the real life nonzero impedances of the output rectifier, transformer secondary and output capacitor. This has been represented previously by the expression (ISEC)(ESR). However, it is generally more useful to convert this expression to an effective output impedance. Because the secondary current only flows during the off portion of the duty cycle, the effective output impedance equals the lumped secondary impedance times the inverse of the OFF duty cycle. That is, ROUT = ESR ) ) 1 where, DC OFF ROUT = Effective supply output impedance ESR = Lumped secondary impedance DC OFF = OFF duty cycle Expressing this in terms of the ON duty cycle, remembering DC OFF = 1 – DC, ROUT = ESR ) ) 1 1 – DC DC = ON duty cycle In less critical applications, or if output load current remains relatively constant, this output impedance error may be judged acceptable and the external RFB resistor value adjusted to compensate for nominal expected error. In more demanding applications, output impedance error may be minimized by the use of the load compensation function. To implement the load compensation function, a voltage is developed that is proportional to average output switch current. This voltage is then impressed across the external ROCOMP resistor and the resulting current is then sub- 10 tracted from the RFB node. As output loading increases, average switch current increases to maintain rough output voltage regulation. This causes an increase in ROCOMP resistor current subtracted from the RFB node, through which feedback loop action causes a corresponding increase in target output voltage. Assuming a relatively fixed power supply efficiency, Eff Power Out = (Eff)(Power In) (VOUT)(IOUT) = (Eff)(VIN)(IIN) Average primary side current may be expressed in terms of output current as follows: IIN = ) ) VOUT I (VIN)(Eff) OUT combining the efficiency and voltage terms in a single variable, IIN = K1(IOUT) where, K1 = ) VOUT (VIN)(Eff) ) Switch current is converted to voltage by a sense resistor and amplified by the current sense amplifier with associated gain G. This voltage is then impressed across the external ROCOMP resistor to form a current that is subtracted from the RFB node. So the effective change in VOUT target is: ∆VOUT = K1(∆IOUT) ) ) (RSENSE)(G) RFB ROCOMP Expressing the product of RSENSE and G as the data sheet value of ∆VRCCOMP/∆ISW, ROUT = K1 ) )) ) ∆VRCCOMP RFB and, ∆ISW ROCOMP ROCOMP = K1 ) )) ) ∆VRCCOMP RFB where, ROUT ∆ISW K1 = Dimensionless variable related to VIN, VOUT and efficiency as above LT1424-9 U OPERATION ) ) ∆VRCCOMP = Data sheet value for RCCOMP pin ∆ISW action vs switch current RFB = External “feedback” resistor value ROUT = Uncompensated output impedance ) )) ∆VOUT ∆VRCCOMP RFB = K1 ∆IOUT ∆ISW ROCOMP ) Nominal output impedance cancellation is obtained by equating this expression with ROUT. U W U U APPLICATIONS INFORMATION The LT1424-X is an application-specific 8-pin part which implements an isolated flyback switcher/controller. Three on-chip thin-film resistors are used to “program” the part for a specific application including mainly desired output voltage, transformer turns ratio and secondary circuit ESR behavior. As of Initial Release, the LT1424-9 is available which implements the “– 9V PCMCIA II Isolated LAN Supply” as described in the Typical Application section. Potential users with a high volume requirement for other applications are advised as follows: general experimentation/breadboarding may be done with the LT1425. This is a general purpose 16-pin part whose functionality is similar to the LT1424-X, with the exception that the three application resistors are external user-supplied components. Application information relating to the proper selection of these resistor values is contained within the LT1425 data sheet. Once technical feasibility is demonstrated, the potential user may discuss the possibility of an additional LT1424-X version with the factory. OUTPUT VOLTAGE ERROR SOURCES Conventional nonisolated switching power supply ICs typically have only two substantial sources of output voltage error— the internal or external resistor divider network that connects to VOUT and the internal IC reference. The LT1424-9, which senses the output voltage in both a dynamic and an isolated manner, exhibits additional potential error sources to contend with. Some of these errors are proportional to output voltage, others are fixed in an absolute millivolt sense. Here is a list of possible error sources and their effective contribution: Internal Voltage Reference The internal bandgap voltage reference is, of course, imperfect. Its error, both at 25°C and over temperature is already included in the specifications for Reference Voltage. Schottky Diode Drop The LT1424-9 senses the output voltage from the transformer primary side during the flyback portion of the cycle. This sensed voltage therefore includes the forward drop, VF, of the rectifier (usually a Schottky diode). Lot-to-lot and ambient temperature variations will show up as output voltage shift/drift. Secondary Leakage Inductance Leakage inductance on the transformer secondary reduces the effective primary-to-secondary turns ratio (NP/NS) from its ideal value. This increases the output voltage target by a similar percentage and has been nominally taken into account in the design of the LT1424-9. To the extent that secondary leakage inductance varies from part-to-part, the output voltage will be affected. Output Impedance Error The LT1424-9 contains a load compensation function to provide a nominal, first-order cancellation of the effects of secondary circuit ESR. Unit-to-unit variation plus some inherent nonlinearity in the cancellation results in some residual VOUT variation with load. 11 LT1424-9 U U W U APPLICATIONS INFORMATION MINIMUM LOAD CONSIDERATIONS The LT1424-9 generally provides better low load performance than previous generation switcher/controllers utilizing indirect output voltage sensing techniques. Specifically, it contains circuitry to detect flyback pulse “collapse,” thereby supporting operation well into discontinuous mode. In general, there are two possible constraints to ultimate low load operation, minimum switch ON time which sets a minimum level of delivered power, and minimum flyback enable time, which deals with the ability of the feedback system to derive valid output voltage information from the flyback pulse. In the application for which the LT1424-9 is designed, the minimum flyback enable time is more restrictive. The LT1424-9 derives its output voltage information from the flyback pulse. If the internal minimum enable time pulse extends beyond the flyback pulse, loss of regulation will occur. The onset of this condition can be determined by setting the width of the flyback pulse equal to the sum of the flyback enable delay, tED, plus the minimum enable time, tEN. Minimum power delivered to the load is then: ) )) ) 1 f [V • (t + t )]2 2 LSEC OUT EN ED = (VOUT)(IOUT) Min Power = Which yields a minimum output constraint: IOUT(MIN) = ) )) ) 1 f(VOUT) (tED + tEN)2, where LSEC 2 f = Switching frequency (nominally 285kHz) LSEC = Transformer secondary side inductance VOUT = Output voltage tED = Enable delay time tEN = Minimum enable time In reality, the previously derived expression is a conservative one, as it assumes perfectly “square” waveforms, which is not the case at light load. Furthermore, the equation was set up to yield just the onset of control error. In other words, while the equation suggests a minimum load current of perhaps 7mA, laboratory observations 12 suggest operation down to 2mA to 3mA before significant output voltage rise is observed. Nevertheless, this situation is addressed in the application by the use of a fixed 1.8k load resistor, which preloads the supply with a nominal 5mA. MAXIMUM LOAD/SHORT-CIRCUIT CONSIDERATIONS The LT1424-9 is a current mode controller. It uses the VC node voltage as an input to a current comparator which turns off the output switch on a cycle-by-cycle basis as this peak current is reached. The internal clamp on the VC node, nominally 1.9V, then acts as an output switch peak current limit. This action becomes the switch current limit specification. The maximum available output power is then determined by the switch current limit, which is somewhat duty cycle dependent due to internal slope compensation action. Short-circuit conditions are handled by the same mechanism. The output switch turns on, peak current is quickly reached and the switch is turned off. Because the output switch is only on for a small fraction of the available period, internal power dissipation is controlled. (The LT1424-9 contains an internal overtemperature shutdown circuit, that disables switch action, just in case.) THERMAL CONSIDERATIONS Care should be taken to ensure that the worst-case input voltage and load current conditions do not cause excessive die temperatures. The packages are rated at 110°C/W for SO-8 and 130°C/W for N8. Average supply current (including driver current) is: ) ) I IIN = 7mA + DC SW where, 35 ISW = Switch current DC = On switch duty cycle Switch power dissipation is given by: PSW = (ISW)2(RSW)(DC) RSW = Output switch ON resistance LT1424-9 U W U U APPLICATIONS INFORMATION Total power dissipation of the die is the sum of supply current times supply voltage plus switch power: PCB LAYOUT CONSIDERATIONS For maximum efficiency, switch rise and fall times are made as short as practical. To prevent radiation and high frequency resonance problems, proper layout of the components connected to the IC is essential, especially the power paths (primary and secondary). B field (magnetic) radiation is minimized by keeping output diode, switch pin and output bypass capacitor leads as short as possible. E field radiation is kept low by minimizing the length and area of all traces connected to the switch pin. A ground plane should always be used under the switcher circuitry to prevent interplane coupling. PD(TOTAL) = (IIN • VIN) + PSW FREQUENCY COMPENSATION Loop frequency compensation is performed by connecting a capacitor from the output of the error amplifier (VC pin) to ground. An additional series resistor, often required in traditional current mode switcher controllers is usually not required; and can even prove detrimental. The phase margin improvement traditionally offered by this extra resistor will usually be already accomplished by the nonzero secondary circuit impedance, which adds a “zero” to the loop response. The high speed switching current paths are shown schematically in Figure 1. Minimum lead length in these paths are essential to ensure clean switching and minimal EMI. The path containing the input capacitor, transformer primary, output switch, the path containing the transformer secondary, output diode and output capacitor are the only ones containing nanosecond rise and fall times. Keep these paths as short as possible. In further contrast to traditional current mode switchers, VC pin ripple is generally not an issue with the LT1424-9. The dynamic nature of the clamped feedback amplifier forms an effective track/hold type response, whereby the VC voltage changes during the flyback pulse, but is then “held” during the subsequent “switch ON” portion of the next cycle. This action naturally holds the VC voltage stable during the current comparator sense action (current mode switching). VOUT • VIN HIGH FREQUENCY CIRCULATING PATH • HIGH FREQUENCY CIRCULATING PATH ISOLATED LOAD F 1424 F01 Figure 1 13 LT1424-9 U PACKAGE DESCRIPTION Dimensions in inches (millimeters) unless otherwise noted. N8 Package 8-Lead PDIP (Narrow 0.300) (LTC DWG # 05-08-1510) 0.400* (10.160) MAX 8 7 6 5 1 2 3 4 0.255 ± 0.015* (6.477 ± 0.381) 0.300 – 0.325 (7.620 – 8.255) 0.009 – 0.015 (0.229 – 0.381) ( +0.035 0.325 –0.015 8.255 +0.889 –0.381 ) 0.045 – 0.065 (1.143 – 1.651) 0.065 (1.651) TYP 0.100 ± 0.010 (2.540 ± 0.254) *THESE DIMENSIONS DO NOT INCLUDE MOLD FLASH OR PROTRUSIONS. MOLD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.010 INCH (0.254mm) 14 0.130 ± 0.005 (3.302 ± 0.127) 0.125 (3.175) 0.020 MIN (0.508) MIN 0.018 ± 0.003 (0.457 ± 0.076) N8 1197 LT1424-9 U PACKAGE DESCRIPTION Dimensions in inches (millimeters) unless otherwise noted. S8 Package 8-Lead Plastic Small Outline (Narrow 0.150) (LTC DWG # 05-08-1610) 0.189 – 0.197* (4.801 – 5.004) 8 7 6 5 0.150 – 0.157** (3.810 – 3.988) 0.228 – 0.244 (5.791 – 6.197) 1 0.010 – 0.020 × 45° (0.254 – 0.508) 0.008 – 0.010 (0.203 – 0.254) 0.053 – 0.069 (1.346 – 1.752) 0°– 8° TYP 0.016 – 0.050 0.406 – 1.270 0.014 – 0.019 (0.355 – 0.483) 2 3 4 0.004 – 0.010 (0.101 – 0.254) 0.050 (1.270) TYP *DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE **DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. SO8 0996 15 LT1424-9 U TYPICAL APPLICATION – 9V PCMCIA Type II Isolated LAN Supply ISOLATION BARRIER R2 75Ω C6 220pF 5V C1 10µF 25V C2 10µF 25V D1 1N5248 0.1µF C5 220pF LT1424-9 2 1 100k 47pF 1000pF 3 4 VC RCCOMP SHDN VIN SYNC VSW SGND PGND 8 MBRS130LT3 R1 75Ω OUT COM 1 3 T1 D2 MBR0540T4 7 2 6 5 C3 10µF 25V C4 10µF 25V 4 1424 TA01 1:1 0.1µF INPUT COM 1.8k –9V 200mA C1, C2, C3, C4: MARCON THCS50E1E106Z CERAMIC CAPACITOR, SIZE 1812. (847) 696-2000 T1: COILTRONICS CTX02-13483 Transformer T1 COILTRONICS CTX02-13483 LPRI RATIO ISOLATION (L × W × H) IOUT EFFICIENCY 27µH 1:1 500VAC 14 × 14 × 2.2mm 200mA 70% RELATED PARTS PART NUMBER DESCRIPTION COMMENTS LT1105 Off-Line Switching Regulator Built-In Isolated Regulation Without Optoisolator LTC 1145/46 Isolated Digital Data Transceivers Up to 200kbps Data Rate, UL Listed LT1170/71/72 5A/3A/1.25A Flyback Regulators Isolated Flyback Mode for Higher Currents LT1370/71 6A/3A Flyback Regulators Uses Small Magnetics LT1372/77 500kHz/1MHz Boost/Flyback Regulators Uses Ultrasmall Magnetics LT1424-5 Isolated Flyback Switching Regulator Same as LT1424-9 But with 5V Output LT1425 Isolated Flyback Switching Regulator General Purpose with External Application Resistors ® 16 Linear Technology Corporation 1630 McCarthy Blvd., Milpitas, CA 95035-7417 ● (408) 432-1900 FAX: (408) 434-0507● TELEX: 499-3977 ● www.linear-tech.com 14249f LT/TP 0599 4K • PRINTED IN USA LINEAR TECHNOLOGY CORPORATION 1998