Triple, 1.5 GHz Op Amp AD8003 POWER DOWN 2 +IN 2 –IN 2 FEEDBACK 2 +VS2 CONNECTION DIAGRAM 24 23 22 21 20 19 2 17 FEEDBACK 3 –IN 1 3 16 –IN 3 4 15 POWER DOWN 1 5 14 POWER DOWN 3 –VS1 6 13 –VS3 7 8 9 10 11 12 +IN 3 05721-001 +IN 1 OUT 3 FEEDBACK 1 NC +VS3 OUT 2 18 NC 1 OUT 1 +VS1 NC High speed 1650 MHz (G = +1) 730 MHz (G = +2, VO = 2 V p-p) 4300 V/μs (G = +2, 4 V step) Settling time 12 ns to 0.1%, 2 V step Excellent for QXGA resolution video Gain flatness 0.1 dB to 190 MHz 0.05% differential gain error, RL = 150 Ω 0.01° differential phase error, RL = 150 Ω Low voltage offset: 0.7 mV (typical) Low input bias current: 7 μA (typical) Low noise: 1.8 nV/√Hz Low distortion over wide bandwidth: SFDR −73 dBc @ 20 MHz High output drive: 100 mA output load drive Supply operation: +5 V to ±5 V voltage supply Supply current: 9.5 mA/amplifier –VS2 FEATURES Figure 1. 24-Lead, 4 mm × 4 mm LFCSP_VQ (CP-24) APPLICATIONS High resolution video graphics Professional video Consumer video High speed instrumentation The AD8003 has excellent video specifications with a frequency response that remains flat out to 190 MHz and 0.1% settling within 12 ns to ensure that even the most demanding video systems maintain excellent fidelity. For applications that use NTSC video, as well as high speed video, the amplifier provides a differential gain of 0.05% and a differential gain of 0.01°. The AD8003 has very low spurious-free dynamic range (SFDR) (−73 dBc @ 20 MHz) and noise (1.8 nV/√Hz). With a supply range between 5 V and 11 V and ability to source 100 mA of output current, the AD8003 is ideal for a variety of applications. The AD8003 amplifier is available in a compact 4 mm × 4 mm, 24-lead LFCSP_VQ. The AD8003 is rated to work over the industrial temperature range of −40°C to +85°C. 3 VS = ±5V 2 G = +1, RF = 432Ω G = +2, +5, RF = 464Ω RL = 150Ω 1 VOUT = 2V p-p G = +1 G = +2 0 –1 –2 G = +5 –3 –4 –5 05721-009 The AD8003 is a triple ultrahigh speed current feedback amplifier. Using ADI’s proprietary eXtra Fast Complementary Bipolar (XFCB) process, the AD8003 achieves a bandwidth of 1.5 GHz and a slew rate of 4300 V/μs. Additionally, the amplifier provides excellent dc precision with an input bias current of 50 μA maximum and a dc input voltage of 0.7 mV. The AD8003 operates on only 9.5 mA of supply current per amplifier. The independent power-down function of the AD8003 reduces the quiescent current even further to 1.6 mA. NORMALIZED CLOSED-LOOP GAIN (dB) GENERAL DESCRIPTION –6 –7 1 10 100 1000 FREQUENCY (MHz) Figure 2. Large Signal Frequency Response for Various Gains Rev. A Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties that may result from its use. Specifications subject to change without notice. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices. Trademarks and registered trademarks are the property of their respective owners. One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A. Tel: 781.329.4700 www.analog.com Fax: 781.461.3113 ©2006 Analog Devices, Inc. All rights reserved. AD8003 TABLE OF CONTENTS Features .............................................................................................. 1 Gain Configurations .................................................................. 12 Applications....................................................................................... 1 RGB Video Driver ...................................................................... 12 Connection Diagram ....................................................................... 1 Printed Circuit Board Layout ....................................................... 13 General Description ......................................................................... 1 Low Distortion Pinout............................................................... 13 Revision History ............................................................................... 2 Signal Routing............................................................................. 13 Specifications with ±5 V Supply ..................................................... 3 Exposed Paddle........................................................................... 13 Specifications with +5 V Supply ..................................................... 4 Power Supply Bypassing ............................................................ 13 Absolute Maximum Ratings............................................................ 5 Grounding ................................................................................... 14 Thermal Resistance ...................................................................... 5 Outline Dimensions ....................................................................... 15 ESD Caution.................................................................................. 5 Ordering Guide .......................................................................... 15 Typical Performance Characteristics ............................................. 6 Applications..................................................................................... 12 REVISION HISTORY 2/06—Rev. 0 to Rev. A Changes to Figure 34...................................................................... 11 10/05—Revision 0: Initial Version Rev. A | Page 2 of 16 AD8003 SPECIFICATIONS WITH ±5 V SUPPLY TA = 25°C, VS = ±5 V, RL = 150 Ω, Gain = +2, RF = 464 Ω, unless otherwise noted. Table 1. Parameter DYNAMIC PERFORMANCE –3 dB Bandwidth Conditions Min Typ Max Unit G = +1, Vo = 0.2 V p-p, RF = 432 Ω G = +2, Vo = 2 V p-p 1650 MHz 730 MHz G = +10, Vo = 0.2 V p-p G = +5, Vo = 2 V p-p 290 MHz 330 MHz Bandwidth for 0.1 dB Flatness Vo = 2 V p-p 190 MHz Slew Rate G = +2, Vo = 2 V step, RL = 150 Ω 3800 V/μs Settling Time to 0.1% G = +2, Vo = 2 V step 12 ns 30/40 ns G = +1, Vo = 2 V p-p 76/97 dBc G = +1, Vo = 2 V p-p f = 1 MHz f = 1 MHz NTSC, G = +2, RL = 150 Ω NTSC, G = +2, RL = 150 Ω 79/73 dBc 1.8 36/3 0.05 0.01 nV/√Hz pA/√Hz % Degree Overload Recovery Input/Output NOISE/HARMONIC PERFORMANCE Second/Third Harmonic @ 5 MHz Second/Third Harmonic @ 20 MHz Input Voltage Noise Input Current Noise (I−/I+) Differential Gain Error Differential Phase Error DC PERFORMANCE Input Offset Voltage −9.3 +IB/−IB TMIN − TMAX (+IB/−IB) −19/−40 Vo = ±2.5 V 400 +0.7 1.08 7.4 −7/−7 −3.8/+29.5 ±14.2 600 VCM = ±2.5 V −51 1.6/3 ±3.6 −48 RL = 150 Ω VO = 2 V p-p, second harmonic < −50 dBc 40% over shoot ±3.85 TMIN − TMAX Input Offset Voltage Drift Input Bias Current Input Offset Current Transimpedance INPUT CHARACTERISTICS Noninverting Input Impedance Input Common-Mode Voltage Range Common-Mode Rejection Ratio OUTPUT CHARACTERISTICS Output Voltage Swing Linear Output Current Capacitive Load Drive POWER DOWN PINS Power-Down Input Voltage Turn-Off Time Turn-On Time Input Current Enabled Power-Down POWER SUPPLY Operating Range Quiescent Current per Amplifier Quiescent Current per Amplifier Power Supply Rejection Ratio (+PSRR/−PSRR) Power down Enable 50% of power-down voltage to 10% of VOUT final, VIN = 0.5 V p-p 50% of power-down voltage to 90% of VOUT final, VIN = 0.5 V p-p Enabled Power down Rev. A | Page 3 of 16 ±3.9 100 27 +9.3 1100 mV mV μV/°C μA μA μA kΩ −46 MΩ/pF V dB +4/+50 ±3.92 V mA pF <VS − 2.5 >VS − 2.5 40 V V ns 130 ns −365 0.1 −235 −85 μA μA 4.5 8.1 1.2 −59/−57 9.5 1.4 −57/−53 10 10.2 1.6 −55/−50 V mA mA dB AD8003 SPECIFICATIONS WITH +5 V SUPPLY TA = 25°C, VS = 5 V, RL = 150 Ω, Gain = +2, RF = 464 Ω, unless otherwise noted. Table 2. Parameter DYNAMIC PERFORMANCE –3 dB Bandwidth Conditions Min Typ Max Unit G = +1, Vo = 0.2 V p-p, RF = 432 Ω G = +2, Vo = 2 V p-p 1050 MHz 590 MHz G = +10, Vo = 0.2 V p-p G = +5, Vo = 2 V p-p 290 MHz 310 MHz Bandwidth for 0.1 dB Flatness Vo = 2 V p-p 83 MHz Slew Rate G = +2, Vo = 2 V step, RL = 150 Ω 2860 V/μs Settling Time to 0.1% G = +2, Vo = 2 V step 12 ns 40/60 ns 75/78 dBc Overload Recovery Input/Output NOISE/HARMONIC PERFORMANCE Second/Third Harmonic @ 5 MHz Second/Third Harmonic @ 20 MHz Input Voltage Noise Input Current Noise (I−/I+) Differential Gain Error Differential Phase Error DC PERFORMANCE Input Offset Voltage G = +1, Vo = 2 V p-p G = +1, Vo = 2 V p-p 66/61 dBc f = 1 MHz f = 1 MHz NTSC, G = +2, RL = 150 Ω NTSC, G = +2, RL = 150 Ω 1.8 36/3 0.04 0.01 nV/√Hz pA/√Hz % Degree −6.5 300 +2.7 2.06 14.2 −7.7/−2.3 −4/−27.8 ±5.4 530 −50 1.6/3 1.3 to 3.7 −48 TMIN − TMAX Input Offset Voltage Drift Input Bias Current (+IB/−IB) −21/−50 TMIN − TMAX (+IB/−IB) Input Offset Current Transimpedance INPUT CHARACTERISTICS Noninverting Input Impedance Input Common-Mode Voltage Range Common-Mode Rejection Ratio OUTPUT CHARACTERISTICS Output Voltage Swing Linear Output Current Capacitive Load Drive POWER DOWN PINS Power-Down Input Voltage Turn-Off Time Turn-On Time Input Current Enabled Power-Down POWER SUPPLY Operating Range Quiescent Current per Amplifier Quiescent Current per Amplifier Power Supply Rejection Ratio (+PSRR/−PSRR) RL = 150 Ω VO = 2 V p-p, second harmonic < −50 dBc 45% over shoot ±1.52 Power down Enable 50% of power-down voltage to 10% of VOUT final, VIN = 0.5 V p-p 50% of power-down voltage to 90% of VOUT final, VIN = 0.5 V p-p Enabled Power down Rev. A | Page 4 of 16 ±1.57 70 27 +11 1500 mV mV μV/°C μA μA μA kΩ −45 MΩ/pF V dB +5/+48 ±1.62 V mA pF <VS − 2.5 >VS − 2.5 125 V V ns 80 ns −160 0.1 −43 +80 μA μA 4.5 6.3 0.8 −59/−56 7.9 0.9 −57/−53 10 9.4 1.1 −55/−50 V mA mA dB AD8003 ABSOLUTE MAXIMUM RATINGS Rating 11 V See Figure 3 −VS − 0.7 V to +VS + 0.7 V ±VS −VS −65°C to +125°C −40°C to +85°C 300°C 150°C Stresses above those listed under Absolute Maximum Ratings may cause permanent damage to the device. This is a stress rating only; functional operation of the device at these or any other conditions above those indicated in the operational section of this specification is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability. THERMAL RESISTANCE θJA is specified for the worst-case conditions, that is, θJA is specified for device soldered in circuit board for surface-mount packages. Table 4. Thermal Resistance Package Type 24-Lead LFCSP_VQ θJA 70 Unit °C/W Maximum Power Dissipation The maximum safe power dissipation for the AD8003 is limited by the associated rise in junction temperature (TJ) on the die. At approximately 150°C, which is the glass transition temperature, the plastic changes its properties. Even temporarily exceeding this temperature limit may change the stresses that the package exerts on the die, permanently shifting the parametric performance of the AD8003. Exceeding a junction temperature of 175°C for an extended period can result in changes in silicon devices, potentially causing degradation or loss of functionality. The power dissipated in the package (PD) is the sum of the quiescent power dissipation and the power dissipated in the die due to the AD8003 drive at the output. The quiescent power is the voltage between the supply pins (VS) times the quiescent current (IS). PD = Quiescent Power + (Total Drive Power – Load Power) ⎛V V PD = (VS × I S ) + ⎜⎜ S × OUT RL ⎝ 2 ⎞ VOUT 2 ⎟– ⎟ RL ⎠ RMS output voltages should be considered. If RL is referenced to −VS, as in single-supply operation, the total drive power is VS × IOUT. If the rms signal levels are indeterminate, consider the worst case, when VOUT = VS/4 for RL to midsupply. PD = (VS × I S ) + (VS / 4 )2 RL In single-supply operation with RL referenced to −VS, worst case is VOUT = VS/2. Airflow increases heat dissipation, effectively reducing θJA. In addition, more metal directly in contact with the package leads and exposed paddle from metal traces, through holes, ground, and power planes reduce θJA. Figure 3 shows the maximum safe power dissipation in the package vs. the ambient temperature for the exposed paddle, 4 mm × 4 mm LFCSP_VQ (70°C/W) package on a JEDEC standard 4-layer board. θJA values are approximations. 3.0 2.5 2.0 1.5 1.0 0.5 0 05721-037 Parameter Supply Voltage Power Dissipation Common-Mode Input Voltage Differential Input Voltage Exposed Paddle Voltage Storage Temperature Range Operating Temperature Range Lead Temperature (Soldering 10 sec) Junction Temperature MAXIMUM POWER DISSIPATION (W) Table 3. –55 –35 –15 5 25 45 65 85 AMBIENT TEMPERATURE (°C) 105 125 Figure 3. Maximum Power Dissipation vs. Temperature for a 4-Layer Board ESD CAUTION ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily accumulate on the human body and test equipment and can discharge without detection. Although this product features proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high energy electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance degradation and loss of functionality. Rev. A | Page 5 of 16 AD8003 TYPICAL PERFORMANCE CHARACTERISTICS 3 0 –1 –2 –3 G = –1 –4 –5 G = –2 –6 –7 1 10 100 1 –1 –2 G = +10 –3 –4 –5 –6 1 10 NORMALIZED CLOSED-LOOP GAIN (dB) 0 VS = ±5V –1 –2 –3 –4 VS = +5V –5 05721-004 NORMALIZED CLOSED-LOOP GAIN (dB) 3 G = +2 RL = 150Ω VOUT = 200mV p-p 1 –6 –7 1 10 100 1 0 –2 –3 –4 T = +25°C –5 T = –40°C –6 1 10 3 RF = 357Ω 0 RF = 432Ω –1 –2 RF = 464Ω –3 –4 –5 –6 –7 1 10 100 1 RF = 392Ω RF = 357Ω 0 –1 RF = 432Ω –2 RF = 464Ω –3 –4 –5 –6 –7 1000 G = +2 VS = ±5V RL = 150Ω VOUT = 2V p-p 2 05721-008 RF = 392Ω NORMALIZED CLOSED-LOOP GAIN (dB) 1 1000 Figure 8. Small Signal Frequency Response for Various Temperatures 05721-007 NORMALIZED CLOSED-LOOP GAIN (dB) 2 100 FREQUENCY (MHz) Figure 5. Small Signal Frequency Response for Various Supplies G = +2 VS = ±5V RL = 150Ω VOUT = 200mV p-p T = +105°C –1 –7 1000 G = +2 VS = ±5V RL = 150Ω VOUT = 200mV p-p 2 FREQUENCY (MHz) 3 1000 Figure 7. Small Signal Frequency Response for Various Gains Figure 4. Small Signal Frequency Response for Various Gains 2 100 FREQUENCY (MHz) FREQUENCY (MHz) 3 G = +1 0 –7 1000 G = +2 05721-003 1 VS = ±5V G = +1, RF = 432Ω G = +2, +10, RF = 464Ω RL = 150Ω VOUT = 200mV p-p 2 05721-005 2 NORMALIZED CLOSED-LOOP GAIN (dB) VS = ±5V RF = 464Ω RL = 150Ω VOUT = 200mV p-p 05721-002 NORMALIZED CLOSED-LOOP GAIN (dB) 3 1 10 100 1000 FREQUENCY (MHz) FREQUENCY (MHz) Figure 9. Large Signal Feedback Resistor (RF) Optimization Figure 6. Small Signal Feedback Resistor (RF) Optimization Rev. A | Page 6 of 16 AD8003 0.3 NORMALIZED CLOSED-LOOP GAIN (dB) RS = 0Ω RS = 25Ω 0 RS = 50Ω –3 –6 –9 1 10 100 FREQUENCY (MHz) 1000 0 VS = ±5V –0.1 –0.2 –0.3 –0.4 –0.5 –0.6 –0.7 –0.8 –0.9 10000 1 G = +1 G = +2 0 –1 –2 G = +5 –4 –5 –6 –7 1 10 100 1 T = +105°C T = –40°C 0 T = +25°C –1 –2 –3 –4 –5 –6 –7 1000 VS = ±5V G = +2 RL = 150Ω VOUT = 2V p-p 2 1 10 FREQUENCY (MHz) G = +1 R = 100Ω –40 V L = 2V p-p OUT –30 G = +2 R = 150Ω –40 V L = 2V p-p OUT VS = ±5V VS = +5V DISTORTION (dBc) DISTORTION (dBc) VS = ±5V VS = +5V –50 –60 SECOND –70 –80 THIRD –60 –70 –90 –110 –110 05721-017 –100 1 10 FREQUENCY (MHz) –120 0.1 100 Figure 12. Harmonic Distortion vs. Frequency for Various Supplies SECOND –80 –100 –120 0.1 1000 Figure 14. Large Signal Frequency Response for Various Temperatures –50 –90 100 FREQUENCY (MHz) Figure 11. Large Signal Frequency Response for Various Gains –30 1000 05721-010 NORMALIZED CLOSED-LOOP GAIN (dB) 3 05721-009 NORMALIZED CLOSED-LOOP GAIN (dB) 3 –3 10 100 FREQUENCY (MHz) Figure 13. 0.1 dB Flatness Response Figure 10. G = +1 Series Resistor (RS) Optimization VS = ±5V 2 G = +1, RF = 432Ω G = +2, +5, RF = 464Ω RL = 150Ω 1 VOUT = 2V p-p VS = +5V THIRD 05721-018 –12 G = +2 0.2 RL = 150Ω VOUT = 2V p-p 0.1 05721-016 G = +1 VS = ±5V RL = 150Ω 3 VOUT = 200mV p-p 05721-006 NORMALIZED CLOSED-LOOP GAIN (dB) 6 1 10 FREQUENCY (MHz) 100 Figure 15. Harmonic Distortion vs. Frequency for Various Supplies Rev. A | Page 7 of 16 AD8003 0.20 VS = ±5V VS = +5V 0.15 OUTPUT VOLTAGE (V) SECOND –40 –50 –60 THIRD –70 –90 05721-019 –80 10 12 14 16 18 20 RL (Ω) 22 24 26 28 2.60 VS = ±5V 0.05 2.55 0 2.50 –0.05 2.45 –0.10 2.40 –0.15 2.35 0 Figure 16. Harmonic Distortion vs. RL 1.5 –1.5 1.0 –2.0 0.5 10 11 12 13 14 15 4 5 6 7 8 9 TIME (ns) OUTPUT VOLTAGE (V) OUTPUT VOLTAGE (V) –1.0 6 7 8 9 TIME (ns) 10 11 12 13 14 15 2.30 0.1 CL = 15pF 0 CL = 0pF –0.1 G = +2 RL = 150Ω VS = ±5V VOUT = 200mV p-p –0.2 05721-012 2.0 –0.3 0 5 10 CL = 27pF 15 20 TIME (ns) 25 30 35 Figure 17. Large Signal Pulse Response for Various Supplies Figure 20. Small Signal Pulse Response for Various Capacitive Loads 2.8 1.5 2.7 1.0 G = +2 VS = ±5V RL = 150Ω 0.3 VOUT 0.2 VIN 2.5 CL = 0pF 2.4 G = +2 RL = 150Ω VS = 5V VOUT = 200mV p-p 2.3 2.2 0 5 10 CL = 27pF 15 20 TIME (ns) 25 30 35 0.5 0.1 0 0 VSETTLE –0.5 –0.1 –1.0 –1.5 Figure 18. Small Signal Pulse Response for Various Capacitive Loads –0.2 –5 0 5 10 15 20 25 TIME (ns) 30 35 40 Figure 21. Short-Term 0.1% Settling Time Rev. A | Page 8 of 16 45 –0.3 05721-021 CL = 15pF SETTLING (%) AMPLITUDE (V) 2.6 05721-022 OUTPUT VOLTAGE (V) OUTPUT VOLTAGE (V) 3.0 –0.5 3 5 0.2 3.5 2.5 2 4 0.3 0 1 3 4.0 VS = ±5V 0 2 4.5 VS = +5V 0.5 1 Figure 19. Small Signal Pulse Response for Various Supplies 2.0 G = +2 RL = 150Ω 1.5 VOUT = 2V p-p 1.0 2.65 0.10 –0.20 30 VS = +5V 05721-020 DISTORTION (dBc) –30 2.70 G = +2 RL = 150Ω VOUT = 200mV p-p OUTPUT VOLTAGE (V) G = +2 VOUT = 2V p-p –20 f = 5MHz C 05721-011 –10 AD8003 6000 G = +2 RL = 150Ω 5 RISE FALL VS = ±5V G = +1 VS = ±5V RL = 150Ω INPUT 4 5000 3 AMPLITUDE (V) 3000 VS = +5V 2000 1 0 –1 –2 –3 1000 –4 05721-013 0 1 0 2 3 4 VOUT p-p (V) 5 6 05721-023 SLEW RATE (V/µs) OUTPUT 2 4000 –5 7 0 0.1 1000 G = +2 VS = ±5V RL = 150Ω INPUT × 2 4 0.3 0.4 0.5 0.6 TIME (µs) 0.7 0.8 0.9 1.0 Figure 25. Input Overdrive Recovery Figure 22. Slew Rate vs. Output Voltage 5 0.2 3 G = +1/+2 VS = ±5V 100 OUTPUT IMPEDANCE (Ω) AMPLITUDE (V) 2 1 0 –1 –2 10 1 –5 0 0.1 0.2 0.3 0.4 0.5 0.6 TIME (µs) 0.7 0.8 0.9 0.1 0.1 1.0 0 POWER SUPPLY REJECTION (dB) G=0 VS = ±5V RL = 150Ω –20 –30 –40 –50 –60 0.1 05721-026 COMMON-MODE REJECTION (dB) –10 1 10 FREQUENCY (MHz) 10 FREQUENCY (MHz) 100 1000 Figure 26. Output Impedance vs. Frequency Figure 23. Output Overdrive Recovery 0 1 –10 G = +2 VS = ±5V RL = 150Ω –20 –30 PSR– –40 –50 PSR+ –60 –70 0.1 100 Figure 24. Common-Mode Rejection vs. Frequency 05721-025 –4 05721-027 05721-024 –3 1 10 FREQUENCY (MHz) 100 Figure 27. Power Supply Rejection vs. Frequency Rev. A | Page 9 of 16 1000 AD8003 80 20 15 60 VS = ±5V VS = +5V 40 VS = +5V VS = ±5V 10 IB (µA) VOS (mV) 5 20 0 0 –5 –20 –10 –60 –5 –4 –3 –2 –1 0 1 2 3 4 –20 –5 5 05721-032 –15 05721-031 –40 –4 –3 –2 –1 VCM (V) 0 1 2 3 4 5 VCM (V) Figure 31. Noninverting Input Bias Current vs. Common-Mode Range Figure 28. Offset Voltage vs. Input Common-Mode Range 10 6 VS = +5V 8 6 5 VS = ±5V 4 AMPLITUDE (V) 0 –2 –4 VDIS (VS = +5V) 3 VOUT (VS = +5V) 2 VOUT (VS = ±5V) –6 –4 –3 –2 –1 0 1 2 3 4 VOUT (VS = ±5V) 5 0 VOUT (V) Figure 29. Inverting Input Bias Current Linearity 10 150 9 50 0 IDIS 5 –50 4 –100 3 –150 ICC 2 –200 POSITIVE SUPPLY CURRENT (mA) 100 7 POWER DOWN PIN CURRENT (µA) 0.7 0.8 1.0 0.9 Figure 30. POWER DOWN Pin Current and Supply Current vs. POWER DOWN Pin Voltage 30 20 ICC 10 IDIS –10 4 –20 3 –30 2 –40 –50 0 5 40 0 1 4 0.4 0.5 0.6 TIME (µs) 5 –300 –3 –2 –1 0 1 2 3 POWER DOWN PIN VOLTAGE (VDIS (V)) 0.3 6 0 –4 0.2 G = +2 RL = 150Ω VS = 5V 7 –250 –5 0.1 8 1 05721-028 POSITIVE SUPPLY CURRENT (mA) 200 8 6 0 Figure 32. Disable Switching Time for Various Supplies G = +2 RL = 150Ω VS = ±5V 9 05721-014 –10 –5 VOUT (VS = +5V) 1 05721-033 –8 0 0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 POWER DOWN PIN VOLTAGE (VDIS (V)) 4.5 5.0 –60 Figure 33. POWER DOWN Pin Current and Supply Current vs. POWER DOWN Pin Voltage Rev. A | Page 10 of 16 POWER DOWN PIN CURRENT (µA) IB (μA) 2 05721-029 4 10 G = +2 RL = 150Ω VIN = 0.5V dc VDIS (VS = ±5V) AD8003 10000 100 100 I– 10 I+ 100 1k 10k 100k 1M 1 10 10M 100 1k 100k 1M 10M Figure 36. Input Current Noise vs. Frequency Figure 34. Input Voltage Noise vs. Frequency 200 1M 0 G = +2 RL = 150Ω –10 DRIVING: CH1 AND CH3 RECEIVING: CH2 –20 180 VS = ±5V 160 100k MAGNITUDE (Ω) VS = +5V –40 –50 –60 120 10k 100 80 60 –70 1k 40 –80 –90 –100 0.1 PHASE (Degrees) 140 –30 20 05721-015 NORMALIZED CLOSED-LOOP GAIN (dB) 10k FREQUENCY (Hz) FREQUENCY (Hz) 1 10 FREQUENCY (MHz) 100 100 1k 1000 10k 100k 1M 10M FREQUENCY (Hz) Figure 35. Worst-Case Crosstalk Figure 37. Transimpedance Rev. A | Page 11 of 16 100M 1G 0 05721-030 1 10 1000 05721-034 10 VS = ±5V 05721-035 VS = ±5V RF = 1kΩ INPUT CURRENT NOISE (pA/√Hz) INPUT VOLTAGE NOISE (nV/√Hz) 1000 AD8003 APPLICATIONS GAIN CONFIGURATIONS RGB VIDEO DRIVER Unlike conventional voltage feedback amplifiers, the feedback resistor has a direct impact on the closed-loop bandwidth and stability of the current feedback op amp circuit. Reducing the resistance below the recommended value can make the amplifier response peak and can even become unstable. Increasing the size of the feedback resistor reduces the closed-loop bandwidth. Figure 40 shows a typical RGB driver application using bipolar supplies. The gain of the amplifier is set at +2 , where RF = RG = 464 Ω. The amplifier inputs are terminated with shunt 75 Ω resistors, and the outputs have series 75 Ω resistors for proper video matching. In Figure 40, the POWER DOWN pins are not shown connected to any signal source for simplicity. If the power-down function is not used, it is recommended that the POWER DOWN pins be tied to the positive supply and not be left floating (not connected). Table 5 provides a convenient reference for quickly determining the feedback and gain set resistor values, and the small and large signal bandwidths for common gain configurations. The feedback resistors in Table 5 have been optimized for 0.1 dB flatness frequency response. Table 5. Recommended Values and Frequency Response1 RF (Ω) 300 432 464 300 300 RG (Ω) 300 N/A 464 75 33.2 RS (Ω) 0 24.9 0 0 0 Large Signal −3 dB BW 668 822 730 558 422 Large Signal 0.1 dB BW --190 165 170 PD3 PD2 PD1 5 23 +VS 14 10µF 1 0.1µF RIN 4 RG 75Ω 464Ω RF 464Ω 1 Conditions: VS = ±5 V, TA = 25°C, RL = 150 Ω. Figure 38 and Figure 39 show the typical noninverting and inverting configurations and recommended bypass capacitor values. 75Ω –VS 3 6 0.1µF 2 +VS +VS 10µF 10µF AD8003 RF RG VIN RS 0.1µF FB 0.1µF 75Ω 75Ω AD8003 VO + –V 19 22 GIN +V – ROUT 10µF RG VO RL 464Ω RF 464Ω 0.1µF 10µF –VS 21 GOUT 10µF 24 0.1µF 20 +VS 10µF 18 05721-038 –VS 0.1µF Figure 38. Noninverting Gain RG 10µF 464Ω RF 464Ω RF 0.1µF FB AD8003 + VO 16 10µF Figure 39. Inverting Gain Rev. A | Page 12 of 16 BOUT 10µF 13 0.1µF Figure 40. RGB Video Driver VO RL –V 0.1µF –VS –VS 17 +V – 05721-039 RG 75Ω 75Ω +VS VIN 15 BIN 05721-036 Gain −1 +1 +2 +5 +10 −3 dB SS BW (MHz) 734 1650 761 567 446 In applications that require a fixed gain of +2, as previously mentioned, the designer may consider the ADA4862-3. The ADA4862-3 is another high performance triple current feedback amplifier. The ADA4862-3 has integrated feedback and gain set resistors that reduce board area and simplify designs. AD8003 PRINTED CIRCUIT BOARD LAYOUT Printed circuit board (PCB) layout is usually one of the last steps in the design process and often proves to be one of the most critical. A high performance design can be rendered mediocre due to poor or sloppy layout. Because the AD8003 can operate into the RF frequency spectrum, high frequency board layout considerations must be taken into account. The PCB layout, signal routing, power supply bypassing, and grounding must all be addressed to ensure optimal performance. LOW DISTORTION PINOUT The AD8003 LFCSP features ADI’s low distortion pinout. The pinout lowers the second harmonic distortion and simplifies the circuit layout. The close proximity of the noninverting input and the negative supply pin creates a source of second harmonic distortion. Physical separation of the noninverting input pin and the negative power supply pin reduces this distortion. By providing an additional output pin, the feedback resistor can be connected directly between the feedback pin and the inverting input. This greatly simplifies the routing of the feedback resistor and allows a more compact circuit layout, which reduces its size and helps to minimize parasitics and increase stability. SIGNAL ROUTING To minimize parasitic inductances, ground planes should be used under high frequency signal traces. However, the ground plane should be removed from under the input and output pins to minimize the formation of parasitic capacitors, which degrades phase margin. Signals that are susceptible to noise pickup should be run on the internal layers of the PCB, which can provide maximum shielding. EXPOSED PADDLE The AD8003 features an exposed paddle, which lowers the thermal resistance by approximately 40% compared to a standard SOIC plastic package. The paddle can be soldered directly to the ground plane of the board. Thermal vias or heat pipes can also be incorporated into the design of the mounting pad for the exposed paddle. These additional vias improve the thermal transfer from the package to the PCB. Using a heavier weight copper also reduces the overall thermal resistance path to ground. POWER SUPPLY BYPASSING Power supply bypassing is a critical aspect of the PCB design process. For best performance, the AD8003 power supply pins need to be properly bypassed. Each amplifier has its own supply pins brought out for the utmost flexibility. Supply pins can be commoned together or routed to a dedicated power plane. Commoned supply connections can also reduce the need for bypass capacitors on each supply line. The exact number and values of the bypass capacitors are dictated by the design specifications of the actual circuit. A parallel combination of different value capacitors from each of the power supply pins to ground tends to work the best. Paralleling different values and sizes of capacitors helps to ensure that the power supply pins see a low ac impedance across a wide band of frequencies. This is important for minimizing the coupling of noise into the amplifier. Starting directly at the power supply pins, the smallest value and physical-sized component should be placed on the same side of the board as the amplifier, and as close as possible to the amplifier, and connected to the ground plane. This process should be repeated for the next largest capacitor value. It is recommended that a 0.1 μF ceramic 0508 case be used for the AD8003. The 0508 offers low series inductance and excellent high frequency performance. The 0.1 μF case provides low impedance at high frequencies. A 10 μF electrolytic capacitor should be placed in parallel with the 0.1 μF. The 10 μF capacitor provides low ac impedance at low frequencies. Smaller values of electrolytic capacitors can be used depending on the circuit requirements. Additional smaller value capacitors help provide a low impedance path for unwanted noise out to higher frequencies but are not always necessary. Placement of the capacitor returns (grounds), where the capacitors enter into the ground plane, is also important. Returning the capacitor grounds close to the amplifier load is critical for distortion performance. Keeping the capacitors distance short, but equal from the load, is optimal for performance. In some cases, bypassing between the two supplies can help improve PSRR and maintain distortion performance in crowded or difficult layouts. Designers should note this as another option for improving performance. Rev. A | Page 13 of 16 AD8003 Minimizing the trace length and widening the trace from the capacitors to the amplifier reduces the trace inductance. A series inductance with the parallel capacitance can form a tank circuit, which can introduce high frequency ringing at the output. This additional inductance can also contribute to increased distortion due to high frequency compression at the output. The use of vias should be minimized in the direct path to the amplifier power supply pins because vias can introduce parasitic inductance, which can lead to instability. When required, use multiple large diameter vias because this lowers the equivalent parasitic inductance. GROUNDING The use of ground and power planes is encouraged as a method of proving low impedance returns for power supply and signal currents. Ground and power planes can also help to reduce stray trace inductance and provide a low thermal path for the amplifier. Ground and power planes should not be used under any of the pins of the AD8003. The mounting pads and the ground or power planes can form a parasitic capacitance at the amplifiers input. Stray capacitance on the inverting input and the feedback resistor form a pole, which degrades the phase margin, leading to instability. Excessive stray capacitance on the output also forms a pole, which degrades phase margin. Rev. A | Page 14 of 16 AD8003 OUTLINE DIMENSIONS 0.60 MAX 4.00 BSC SQ PIN 1 INDICATOR 0.60 MAX TOP VIEW 0.50 BSC 3.75 BSC SQ 0.50 0.40 0.30 1.00 0.85 0.80 0.80 MAX 0.65 TYP 12° MAX 0.30 0.23 0.18 SEATING PLANE PIN 1 INDICATOR 24 1 19 18 2.25 2.10 SQ 1.95 EXPOSED PAD (BOTTOM VIEW) 13 12 7 6 0.25 MIN 2.50 REF 0.05 MAX 0.02 NOM 0.20 REF COPLANARITY 0.08 COMPLIANT TO JEDEC STANDARDS MO-220-VGGD-2 Figure 41. 24-Lead Lead Frame Chip Scale Package [LFCSP_VQ] 4 mm × 4 mm Body, Very Thin Quad (CP-24-1) Dimensions shown in millimeters ORDERING GUIDE Model AD8003ACPZ-R2 1 AD8003ACPZ-REEL1 AD8003ACPZ-REEL71 1 Temperature Range –40°C to +85°C –40°C to +85°C –40°C to +85°C Package Description 24-Lead LFCSP_VQ 24-Lead LFCSP_VQ 24-Lead LFCSP_VQ Z = Pb-free part. Rev. A | Page 15 of 16 Package Option CP-24-1 CP-24-1 CP-24-1 Ordering Quantity 250 5,000 1,500 AD8003 NOTES ©2006 Analog Devices, Inc. All rights reserved. Trademarks and registered trademarks are the property of their respective owners. D05721-0-2/06(A) Rev. A | Page 16 of 16