LINER LT3755

LT3791
60V 4-Switch Synchronous
Buck-Boost LED Driver
Controller
DESCRIPTION
FEATURES
4-Switch Single Inductor Architecture Allows VIN
Above, Below or Equal to VOUT
n Wide V Range: 4.7V to 60V
IN
n Wide V
OUT Range: 0V to 60V (55V LED)
n±2% Output Voltage Accuracy
n Synchronous Switching: Up to 98.5% Efficiency
n ±6% LED Current Accuracy: 0V ≤ V
OUT < 60V
n V
Disconnected
from
V
During
Shutdown
OUT
IN
n Accurate Rail-to-Rail LED Current Sense with
Monitor Output
n Input Current Sense with Monitor Output
n PWM and Analog Dimming
n Capable of 100W or greater per IC
n38-Lead TSSOP with Exposed Pad
The LT®3791 is a synchronous 4-switch buck-boost LED
driver and voltage regulator controller. The controller
operates from input voltages above, below, or equal to
the output voltage. The LT3791 has a wide 4.7V to 60V
input and 0V to 60V output range along with seamless
transitions between operating modes. A ground reference
voltage FB pin serves as the input for several LED protection features and also makes it possible for the converter
to operate as a constant-voltage source. The LT3791 is
ideal for a wide variety of applications.
n
The LT3791 runs in forced continuous mode, which is
ideal for systems with stringent EMI requirements. Fault
protection is provided to survive and report an open or
shorted LED condition. A timer allows the LT3791 to
continue to run, latch off or restart when a fault occurs.
APPLICATIONS
L, LT, LTC, LTM, Linear Technology and the Linear logo are registered and True Color PWM
is a trademark of Linear Technology Corporation. All other trademarks are the property of their
respective owners.
Automotive Headlamps/Running Lamps
n General Purpose Lighting
n
TYPICAL APPLICATION
98.5% Efficient 100W (33.3V 3A) Buck-Boost LED Driver
2.2µF
100V
×5
0.003Ω
VIN
INTVCC
4.7µF
1µF
IVINN
BST1
470nF
28k
TG1
IVINP
499k
15.8k
SWI
EN/UVLO
OVLO
INTVCC
200k
Efficiency vs VIN
BST2
50Ω
BG1
LT3791
200k
0.1µF
0.1µF
10µH
1M
4.7µF
50V
×5
98
34.2k
SNSP
0.004Ω
SHORTLED
OPENLED
0.033Ω
SNSN
PGND
PWM
IVINMON
ISMON
CLKOUT
BUCK
BUCK-BOOST
96
94
90
ISP
10
20
40
30
INPUT VOLTAGE (V)
50
60
3791 TA01b
ISN
0.1µF
3A, 100W
LED POWER
BOOST
92
BG2
SW2
TG2
FB
VREF
CTRL
100
EFFICIENCY (%)
VIN
15V TO 58V
PWMOUT
SS SYNC VC
RT
2.2k
10nF
SGND
86.6k
300kHz
10nF
3791 TA01a
3791f
1
LT3791
ABSOLUTE MAXIMUM RATINGS
(Note 1)
PIN CONFIGURATION
Supply Voltages
Input Supply (VIN)......................................................60V
SW1, SW2.......................................................–1V to 60V
OPENLED, SHORTLED................................................15V
EN/UVLO, IVINP, IVINN, ISP, ISN...............................60V
INTVCC, (BST1-SW1), (BST2-SW2)..............................6V
TEST2, SYNC, RT, CTRL, OVLO, PWM........................6V
IVINMON, ISMON, FB, SS, VC, VREF............................6V
IVINP-IVINN, ISP-ISN, SNSP-SNSN........................±0.5V
SNSP, SNSN............................................................±0.3V
Operating Junction Temperature (Notes 2, 3)
LT3791E/LT3791I................................ –40°C to 125°C
LT3791H............................................. –40°C to 150°C
LT3791MP.......................................... –55°C to 150°C
Storage Temperature Range................... –65°C to 150°C
Lead Temperature (Soldering, 10 sec).................... 300°C
TOP VIEW
CTRL
1
38 OVLO
SS
2
37 FB
PWM
3
36 VC
OPENLED
4
35 RT
SHORTLED
5
34 SYNC
VREF
6
33 CLKOUT
ISMON
7
32 TEST2
IVINMON
8
31 PWMOUT
EN/UVLO
9
IVINP 10
30 SGND
39
SGND
29 TEST1
IVINN 11
28 SNSN
VIN 12
27 SNSP
INTVCC 13
26 ISN
TG1 14
25 ISP
BST1 15
24 TG2
SW1 16
23 NC
PGND 17
22 BST2
BG1 18
21 SW2
BG2 19
20 PGND
FE PACKAGE
38-LEAD PLASTIC TSSOP
TJMAX = 150°C, θJA = 28°C/W
EXPOSED PAD (PIN 39) IS SGND, MUST BE SOLDERED TO PCB
ORDER INFORMATION
LEAD FREE FINISH
TAPE AND REEL
PART MARKING*
PACKAGE DESCRIPTION
TEMPERATURE RANGE
LT3791EFE#PBF
LT3791EFE#TRPBF
LT3791FE
38-Lead Plastic TSSOP
–40°C to 125°C
LT3791IFE#PBF
LT3791IFE#TRPBF
LT3791FE
38-Lead Plastic TSSOP
–40°C to 125°C
LT3791HFE#PBF
LT3791HFE#TRPBF
LT3791FE
38-Lead Plastic TSSOP
–40°C to 150°C
LT3791MPFE#PBF
LT3791MPFE#TRPBF
LT3791FE
38-Lead Plastic TSSOP
–55°C to 150°C
Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
ELECTRICAL
CHARACTERISTICS
The l denotes the specifications which apply over the full operating
junction temperature range, otherwise specifications are at TA = 25°C (Note 2). VIN = 12V, VEN/UVLO = 12V unless otherwise noted.
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
Input
VIN Operating Voltage
4.7
60
V
VIN Shutdown IQ
VEN/UVLO = 0V
0.1
1
µA
VIN Operating IQ (Not Switching)
FB = 1.3V, RT = 59.0k
3.0
4
mA
3791f
2
LT3791
ELECTRICAL
CHARACTERISTICS
The l denotes the specifications which apply over the full operating
junction temperature range, otherwise specifications are at TA = 25°C (Note 2). VIN = 12V, VEN/UVLO = 12V unless otherwise noted.
PARAMETER
CONDITIONS
MIN
TYP
MAX
1.16
1.2
1.24
UNITS
Logic Inputs
EN/UVLO Falling Threshold
l
EN/UVLO Rising Hysteresis
15
EN/UVLO Input Low Voltage
IVIN Drops Below 1µA
EN/UVLO Pin Bias Current Low
VEN/UVLO = 1V
EN/UVLO Pin Bias Current High
CTRL Input Bias Current
V
mV
0.3
V
3
4
µA
VEN/UVLO = 1.6V
10
100
nA
VCTRL = 1V
20
50
nA
2
CTRL Latch-Off Threshold
175
OVLO Rising Shutdown Voltage
l
2.85
OVLO Falling Hysteresis
3
mV
3.15
75
V
mV
Regulation
VREF Voltage
VREF Line Regulation
4.7V < VIN < 60V
V(ISP-ISN) Threshold
VCTRL = 2V
l
1.96
l
97.5
94
2.04
V
0.002
0.04
%/V
100
100
102.5
106
mV
mV
l
87
84
90
90
93
96
mV
mV
l
47.5
46
50
50
52.5
54
mV
mV
l
6.5
5
10
10
13.5
15
mV
mV
VCTRL = 1100mV
VCTRL = 700mV
VCTRL = 300mV
ISP Bias Current
2.00
110
ISN Bias Current
µA
20
LED Current Sense Common Mode Range
0
LED Current Sense Amplifier gm
µA
60
890
µS
ISMON Monitor Voltage
V(ISP-ISN) = 100mV
l
0.96
1
1.04
Input Current Sense Threshold V(IVINP-IVINN)
3V ≤ VIVINP ≤ 60V
l
46.5
50
54
IVINP Bias Current
90
IVINN Bias Current
3
Input Current Sense Amplifier gm
IVINMON Monitor Voltage
0.96
1
1.04
V
l
1.194
1.176
1.2
1.2
1.206
1.220
V
V
0.002
0.025
%/V
4.7V < VIN < 60V
FB Amplifier gm
FB Pin Input Bias Current
565
FB in Regulation
VC Standby Input Bias Current
PWM = 0V
VSENSE(MAX) (VSNSP-SNSN)
Boost
Buck
V
mS
l
FB Regulation Voltage
FB Line Regulation
mV
µA
60
2.12
V(IVINP-IVINN) = 50mV
V
µA
20
Input Current Sense Common Mode Range
V
150
20
nA
51
–47.5
60
–39
mV
mV
–20
l
l
42
–56
µS
100
nA
Fault
SS Pull-Up Current
VSS = 0V
SS Discharge Current
FB Overvoltage Rising Threshold
Open LED Rising Threshold (VFB)
V(ISP-ISN) = 0V
l
14
µA
1.4
µA
1.22
1.25
1.127
1.15
V
1.173
V
3791f
3
LT3791
ELECTRICAL
CHARACTERISTICS
The l denotes the specifications which apply over the full operating
junction temperature range, otherwise specifications are at TA = 25°C (Note 2). VIN = 12V, VEN/UVLO = 12V unless otherwise noted.
PARAMETER
CONDITIONS
MIN
TYP
MAX
1.078
1.1
1.122
5
10
15
mV
380
400
450
mV
1.1
2.0
kΩ
SHORTLED Pin Output Impedance
1.1
2.0
kΩ
SS Latch-Off Threshold
1.75
V
SS Reset Threshold
0.2
V
Open LED Falling Threshold (VFB)
Open LED Falling Threshold (V(ISP-ISN))
l
VFB = 1.2V
Short LED Falling Threshold (VFB)
OPENLED Pin Output Impedance
UNITS
V
Oscillator
Switching Frequency
RT = 147k
RT = 59.0k
RT = 29.1k
SYNC Frequency
190
380
665
200
400
700
200
SYNC Pin Resistance to GND
210
420
735
kHz
kHz
kHz
700
kHz
90
SYNC Threshold Voltage
0.3
kΩ
1.5
V
Internal VCC Regulator
INTVCC Regulation Voltage
Dropout (VIN – INTVCC)
4.8
IINTVCC = –10mA, VIN = 5V
INTVCC Undervoltage Lockout
INTVCC Current Limit
3.1
VINTVCC = 4V
5
5.2
V
240
350
mV
3.5
3.9
67
V
mA
PWM
PWM Threshold Voltage
0.3
1.5
V
PWM Pin Resistance to GND
90
PWMOUT Pull-Up Resistance
10
20
kΩ
Ω
PWMOUT Pull-Down Resistance
5
10
Ω
NMOS Drivers
TG1, TG2 Gate Driver On-Resistance
Gate Pull-Up
Gate Pull-Down
VBST – VSW = 5V
BG1, BG2 Gate Driver On-Resistance
Gate Pull-Up
Gate Pull-Down
VINTVCC = 5V
TG Off to BG On Delay
BG Off to TG On Delay
TG1, TG2, tOFF(MIN)
2.6
1.7
Ω
Ω
3
1.2
Ω
Ω
CL = 3300pF
60
ns
CL = 3300pF
60
RT = 59.0k
220
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: The LT3791E is guaranteed to meet performance from 0°C
to 125°C junction temperature. Specification over the -40°C to
125°C operating junction temperature range are assured by design,
characterization and correlation with statistical process controls.
The LT3791I is guaranteed to meet performance specifications over the
–40°C to 125°C operating junction temperature range. The LT3791H is
guaranteed to meet performance specifications over the –40°C to 150°C
ns
260
ns
operating junction temperature range. The LT3791MP is guaranteed to
meet performance specifications over the –55°C to 150°C operating
junction temperature range. High junction temperatures degrade operating
lifetimes. Operating lifetime is derated for junction temperatures greater
than 125°C.
Note 3: The LT3791 includes overtemperature protection that is intended
to protect the device during momentary overload conditions. Junction
temperature will exceed the maximum operating junction temperature
when overtemperature protection is active. Continuous operation above
the specified absolute maximum operating junction temperature may
impair device reliability.
3791f
4
LT3791
TYPICAL PERFORMANCE CHARACTERISTICS
INTVCC Dropout Voltage
vs Current, Temperature
VIN-VINTVCC (V)
90
5.20
TA = 150°C
TA = 25°C
TA = –50°C
2.0
INTVCC Current Limit
vs Temperature
INTVCC Voltage vs Temperature
80
5.15
INTVCC CURRENT LIMIT (mA)
2.5
TA = 25°C, unless otherwise noted.
5.10
INTVCC (V)
1.5
1.0
5.05
VIN = 60V
5.00
VIN = 12V
4.95
4.90
0.5
4.85
0
20
10
0
30
0
LDO CURRENT (mA)
50
40
30
20
0
–50 –25
25 50 75 100 125 150
TEMPERATURE (°C)
3791 G01
VREF Voltage vs Temperature
VREF Load Regulation
5.75
2.03
2.15
5.50
2.02
2.10
5.25
2.01
2.05
2.00
4.75
1.99
4.50
1.98
4.25
VREF (V)
2.20
VREF (V)
2.04
4.00
10
20
30
40
50
60
1.90
1.96
–50 –25
70
ILOAD (mA)
VIN = 60V
VIN = 12V
VIN = 4.7V
0
V(ISP-ISN) Threshold vs VCTRL
60
50
40
30
20
10
0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0
VCTRL (V)
3791 G07
50
100 150 200 250 300 350 400
IREF (µA)
V(ISP-ISN) Threshold
vs Temperature
108
108
106
106
104
104
V(ISP-ISN) (mV)
V(ISP-ISN) (mV)
V(ISP-ISN) (mV)
80
70
0
3791 G06
V(ISP-ISN) Threshold vs VISP
90
0
1.80
3791 G05
100
0
1.85
25 50 75 100 125 150
TEMPERATURE (°C)
3791 G04
110
2.00
1.95
1.97
0
25 50 75 100 125 150
TEMPERATURE (°C)
3791 G03
6.00
5.00
0
3791 G02
INTVCC Load Regulation
INTVCC (V)
60
10
4.80
–50 –25
40
70
102
100
98
102
100
98
96
96
94
94
92
0
10
20
40
30
VISP (V)
50
60
3791 G08
VIN = 12V
92
–50 –25
VISP = 60V
VISP = 12V
VISP = 0V
0
25 50 75 100 125 150
TEMPERATURE (°C)
3791 G09
3791f
5
LT3791
TYPICAL PERFORMANCE CHARACTERISTICS
120
V(ISP-ISN) Threshold vs VFB
TA = 25°C, unless otherwise noted.
ISMON Voltage vs Temperature
1.04
VIN = 12V
V(ISP-ISN) = 100mV
1.03
100
ISMON Voltage vs V(ISP-ISN)
1.0
0.9
0.8
60
40
0.7
1.01
VISMON (V)
80
VISMON (V)
V(ISP-ISN) (mV)
1.02
1.00
0.99
1.19
1.18
1.20 1.21
VFB (V)
1.22
0.1
0.96
–50 –25
1.23
0
V(IVINP-IVINN) Threshold
vs VIVINP
IVINMON Voltage vs Temperature
54
51.5
1.03
51.0
1.02
50.5
1.01
50
VIVINP = 3V
48
46
44
42
–50 –25
0
50.0
49.5
48.5
0.97
10
20
30
40
50
50
40
SHORTLED Threshold
vs Temperature
1.24
0.500
1.23
0.475
1.22
0.450
FB VOLTAGE (V)
60
VFB (V)
1.21
1.20
1.19
20
VIN = 60V
VIN = 12V
VIN = 4.7V
1.17
1.19
1.20 1.21
VFB (V)
1.22
1.23
3791 G16
1.16
–50 –25
RISING
0.425
0.400
FALLING
0.375
0.350
1.18
10
25 50 75 100 125 150
TEMPERATURE (°C)
3791 G15
FB Regulation Voltage
vs Temperature
30
0
3791 G14
V(IVINP-IVINN) Threshold vs VFB
1.18
0.96
–50 –25
60
VIVINP (V)
3791 G13
0
1.17
0.99
0.98
0
VIVINP = 12V
V(IVINP-VINN) = 50mV
1.00
49.0
48.0
25 50 75 100 125 150
TEMPERATURE (°C)
VIVINMON (V)
1.04
V(IVINP-IVINN) (mV)
52.0
VIVINP = 60V
10 20 30 40 50 60 70 80 90 100
V(ISP-ISN) (mV)
3791 G12
56
52
0
3791 G11
V(IVINP-IVINN) Threshold
vs Temperature
V(IVINP-IVINN) (mV)
0
25 50 75 100 125 150
TEMPERATURE (°C)
3791 G10
V(IVINP-IVINSN) (mV)
0.4
0.2
0.97
0
1.17
0.5
0.3
0.98
20
0.6
0
25 50 75 100 125 150
TEMPERATURE (°C)
3791 G17
0.325
0.300
–50 –25
0
25 50 75 100 125 150
TEMPERATURE (°C)
3791 G18
3791f
6
LT3791
TYPICAL PERFORMANCE CHARACTERISTICS
OPENLED Threshold
vs Temperature
OVLO Threshold vs Temperature
1.125
FALLING
1.100
1.075
1.050
1.025
1.000
–50 –25
0
16
3.2
14
3.1
12
RISING
3.0
2.9
FALLING
2.7
4
2.6
2
0
Supply Current vs Input Voltage
7
IQ (mA)
1.5
1.0
TA = 150°C
TA = 25°C
TA = –50°C
0
0
10
20
30
VIN (V)
40
50
60
VEN/ULO = 1V
1.28
6
5
4
3
2
TG1, TG2 MINIMUM ON-TIME (ns)
SWITCHING FREQUENCY (kHz)
600
RT = 59.0k
RT = 147k
200
100
0
–50 –25
0
25 50 75 100 125 150
TEMPERATURE (°C)
3791 G25
FALLING
1.16
1.14
1.12
1.10
–50 –25
0
100
300
1.18
25 50 75 100 125 150
TEMPERATURE (°C)
0
25 50 75 100 125 150
TEMPERATURE (°C)
3791 G24
TG1, TG2 Minimum On-Time
vs Temperature
RT = 29.1k
RISING
1.20
3791 G23
800
400
1.22
0
–50 –25
Oscillator Frequency
vs Temperature
500
1.26
1.24
1
3791 G22
700
EN/UVLO Threshold Voltage
1.30
EN/UVLO THRESHOLD (V)
EN/UVLO PIN CURRENT (µA)
3.5
2.0
25 50 75 100 125 150
TEMPERATURE (°C)
3791 G21
EN/UVLO Pin Current
8
2.5
0
3791 G20
4.0
3.0
DISCHARGING
0
–50 –25
25 50 75 100 125 150
TEMPERATURE (°C)
3791 G19
0.5
8
6
2.5
–50 –25
25 50 75 100 125 150
TEMPERATURE (°C)
CHARGING
10
2.8
TG1, TG2 Minimum Off-Time
vs Temperature
350
TG2
90
TG1, TG2 MINIMUM OFF-TIME (ns)
FB VOLTAGE (V)
1.150
OVLO THRESHOLD (V)
RISING
Soft-Start Current vs Temperature
3.3
ISS (µA)
1.200
1.175
TA = 25°C, unless otherwise noted.
TG1
80
70
60
50
40
30
20
–50 –25
0
25 50 75 100 125 150
TEMPERATURE (°C)
3791 G26
300
fSW = 200kHz
250
fSW = 400kHz
200
fSW = 700kHz
150
100
50
0
–50 –25
0
25 50 75 100 125 150
TEMPERATURE (˚C)
3791 G27
3791f
7
LT3791
TYPICAL PERFORMANCE CHARACTERISTICS
BG1, BG2 Driver On-Resistance
vs Temperature
4.5
3.8
4.0
RISING
3.7
3.6
3.5
3.4
FALLING
3.3
3.2
4.0
3.5
PULL-UP
3.5
3.0
2.5
2.0
PULL-DOWN
1.5
1.0
0
0
–50 –25
25 50 75 100 125 150
TEMPERATURE (°C)
0
1.0
14
1.6
12
1.4
25 50 75 100 125 150
TEMPERATURE (°C)
V(SNSP-SNSN) Buck Threshold
vs VC
PULL-UP
40
BG2
1.0
VC (V)
60
V(SNSP-SNSN) = 0V
1.2
PULL-DOWN
0
3791 G30
V(SNSP-SNSN) (mV)
PWMOUT RESISTANCE (Ω)
1.5
VC Voltage vs Duty Cycle
8
PULL-DOWN
2.0
3791 G29
PWMOUT On-Resistance
vs Temperature
6
2.5
0
–50 –25
25 50 75 100 125 150
TEMPERATURE (°C)
3791 G28
10
PULL-UP
3.0
0.5
0.5
3.1
–50 –25
BG1
0.8
0.6
4
20
0
–20
0.4
2
–40
0.2
0
–50 –25
0
25
50
0
75
100 125 150
TEMPERATURE (˚C)
0
20
40
60
DUTY CYCLE (%)
80
3791 G31
V(SNSP-SNSN) (mV)
20
0
–20
VC(MAX)
–60
–50 –25
0
3791 G34
1.2
VC (V)
1.4
1.6
60
60
40
40
20
0
–20
–40
–80
0.6
0.8
1.0
1.2
VC (V)
1.4
1.6
1.8
V(SNSP-SNSN) Boost Threshold
vs Temperature
–60
25 50 75 100 125 150
TEMPERATURE (°C)
1.0
0.8
3791 G33
V(SNSP-SNSN) THRESHOLD (mV)
VC(MIN)
–40
0.6
V(SNSP-SNSN) Boost Threshold
vs VC
60
40
–60
100
3791 G32
V(SNSP-SNSN) Buck Threshold
vs Temperature
V(SNSP-SNSN) THRESHOLD (mV)
TG1, TG2 Driver On-Resistance
vs Temperature
TG1, TG2 RESISTANCE (Ω)
3.9
BG1, BG2 RESISTANCE (Ω)
V(BST1-SW1), V(BST2,SW2) (V)
V(BST1-SW1), V(BST2-SW2) UVLO
vs Temperature
TA = 25°C, unless otherwise noted.
1.8
3791 G35
VC(MAX)
20
0
–20
–40
–60
–80
–50 –25
VC(MIN)
0
25 50 75 100 125 150
TEMPERATURE (˚C)
3791 G36
3791f
8
LT3791
PIN FUNCTIONS
CTRL (Pin 1): Current Sense Threshold Adjustment Pin for
Analog Dimming. Regulating threshold V(ISP-ISN) is 1/10th
of (VCTRL – 200mV). CTRL linear range is from 200mV
to 1.1V. For VCTRL > 1.3V, the current sense threshold is
constant at the full-scale value of 100mV. For 1.1V < VCTRL
< 1.3V, the dependence of the current sense threshold
upon VCTRL transitions from a linear function to a constant value, reaching 98% of full scale by VCTRL = 1.2V.
Connect CTRL to VREF for the 100mV default threshold.
Force less than 175mV (typical) to stop switching. Do not
leave this pin open.
SS (Pin 2): Soft-start reduces the input power sources
surge current by gradually increasing the controller’s current limit. A minimum value of 10nF is recommended on
this pin. SS is used as a timer when an open or shorted
LED condition occurs. A 500k resistor placed from SS to
VREF will latch the part off in the event of a fault. A 100k
resistor to VREF will allow the part to keep running in a
fault. If left open, a 1.4µA current source pulls down on
SS and the part restarts in a fault.
PWM (Pin 3): A signal low turns off switches, idles switching and disconnects the VC pin from all external loads. The
PWMOUT pin follows the PWM pin. PWM has an internal
100k pull-down resistor. If not used, connect to INTVCC.
OPENLED (Pin 4): An open-drain pull-down on OPENLED
asserts if FB is greater than 1.15V (typical) and V(ISP-ISN)
is less than 10mV (typical). To function, the pin requires
an external pull-up resistor.
SHORTLED (Pin 5): An open-drain pull-down on
SHORTLED asserts if FB is less than 400mV (typical).
To function, the pin requires an external pull-up resistor.
VREF (Pin 6): Voltage Reference Output Pin, Typically 2V.
This pin drives a resistor divider for the CTRL pin, either
for analog dimming or for temperature limit/compensation of the LED load. Can supply up to 200µA of current.
ISMON (Pin 7): Monitor pin that produces a voltage that
is ten times the voltage V(ISP-ISN). ISMON will equal 1V
when V(ISP-ISN) = 100mV.
IVINMON (Pin 8): Monitor pin that produces a voltage
that is twenty times the voltage V(IVINP-IVINN). IVINMON
will equal 1V when V(IVINP-IVINN) = 50mV.
EN/UVLO (Pin 9): Enable Control Pin. Forcing an accurate
1.2V falling threshold with an externally programmable
hysteresis is generated by the external resistor divider
and a 3µA pull-down current. Above the 1.2V (typical)
threshold (but below 6V), EN/UVLO input bias current is
sub-µA. Below the falling threshold, a 3µA pull-down current is enabled so the user can define the hysteresis with
the external resistor selection. An undervoltage condition
resets soft-start. Tie to 0.3V, or less, to disable the device
and reduce VIN quiescent current below 1µA.
IVINP (Pin 10): Positive Input for the Input Current Limit
and Monitor. Input bias current for this pin is typically 90µA.
IVINN (Pin 11): Negative Input for the Input Current Limit
and Monitor. The input bias current for this pin is typically
20µA.
VIN (Pin 12): Main Input Supply. Bypass this pin to PGND
with a capacitor.
INTVCC (Pin 13): Internal 5V Regulator Output. The driver
and control circuits are powered from this voltage. Bypass
this pin to PGND with a minimum 4.7µF ceramic capacitor.
TG1 (Pin 14): Top Gate Drive. Drives the top N-channel
MOSFET with a voltage equal to INTVCC superimposed on
the switch node voltage SW1.
BST1 (Pin 15): Bootstrapped Driver Supply. The BST1 pin
swings from a diode voltage below INTVCC up to a diode
voltage below VIN + INTVCC.
SW1 (Pin 16): Switch Node. SW1 pin swings from a diode
voltage drop below ground up to VIN.
PGND (Pins 17, 20): Power Ground. Connect these pins
closely to the source of the bottom N-channel MOSFET.
BG1 (Pin 18): Bottom Gate Drive. Drives the gate of the
bottom N-channel MOSFET between ground and INTVCC.
BG2 (Pin 19): Bottom Gate Drive. Drives the gate of the
bottom N-channel MOSFET between ground and INTVCC.
SW2 (Pin 21): Switch Node. SW2 pin swings from a diode
voltage drop below ground up to VOUT.
BST2 (Pin 22): Bootstrapped Driver Supply. The BST2 pin
swings from a diode voltage below INTVCC up to a diode
voltage below VOUT + INTVCC.
3791f
9
LT3791
PIN FUNCTIONS
NC (Pin 23): No Connect Pin. Leave this pin floating.
TG2 (Pin 24): Top Gate Drive. Drives the top N-channel
MOSFET with a voltage equal to INTVCC superimposed on
the switch node voltage SW2.
ISP (Pin 25): Connection Point for the Positive Terminal
of the Output Current Feedback Resistor.
ISN (Pin 26): Connection Point for the Negative Terminal
of the Output Current Feedback Resistor.
SNSP (Pin 27): The Positive Input to the Current Sense
Comparator. The VC pin voltage and controlled offsets
between the SNSP and SNSN pins, in conjunction with a
resistor, set the current trip threshold.
SNSN (Pin 28): The Negative Input to the Current Sense
Comparator.
TEST1 (Pin 29): This pin is used for testing purposes only
and must be connected to SGND for the part to operate
properly.
SGND (Pin 30, Exposed Pad Pin 39): Signal Ground.
All small-signal components and compensation should
connect to this ground, which should be connected to
PGND at a single point. Solder the exposed pad directly
to the ground plane.
PWMOUT (Pin 31): Buffered Version of PWM Signal for
Driving LED Load Disconnect N-Channel MOSFET. The
PWMOUT pin is driven from INTVCC. Use of a MOSFET
with a gate cutoff voltage higher than 1V is recommended.
TEST2 (Pin 32): This pin is used for testing purposes only
and must be connected to INTVCC (Pin 13) for the part to
operate properly.
CLKOUT (Pin 33): Clock Output Pin. An in-phase clock is
provided at the oscillator frequency to allow for synchronizing two devices for extending output power capability.
SYNC (Pin 34): External Synchronization Input Pin. This
pin is internally terminated to GND with a 90k resistor.
The rising edge will be synchronized with the rising edge
of the SYNC signal.
RT (Pin 35): Frequency Set Pin. Place a resistor to GND
to set the internal frequency. The range of oscillation is
200kHz to 700kHz.
VC (Pin 36): Current Control Threshold and Error Amplifier
Compensation Point. The current comparator threshold
increases with this control voltage. The voltage ranges
from 0.7V to 1.9V.
FB (Pin 37): Voltage Loop Feedback Pin. FB is intended
for constant-voltage regulation or for LED protection of
an open or shorted LED. The internal transconductance
amplifier with output VC will regulate FB to 1.2V (typical)
through the DC/DC converter. If the FB input is regulating
the loop and V(ISP-ISN) < 10mV, the OPENLED pull-down is
asserted. If the FB pin is less than 400mV, the SHORTLED
pull-down is asserted.
OVLO (Pin 38): Overvoltage Input Pin. This pin is used for
OVLO, if OVLO > 3V then SS is pulled low, the part stops
switching and resets. Do not leave this pin open.
3791f
10
LT3791
BLOCK DIAGRAM
26
–
+
A = 10
7
8
9
11
IVINP
+
A2
10
ISN
ISP
A1
A = 10 A = 20
12
6
13
VREF
VIN
IVINN
INTVCC
–
25
A = 24
REGS
SHDN_INT
ISMON
ISMON_INT
TSD
IVINMON_INT
BST1
+
IVINMON
A13
A3
EN/UVLO
A4
1.2V
SW1
–
–
3µA
TG1
SHDN_INT
SHDN_INT
SS_RESET
SS LATCH
PWM
+
35
34
33
A14
BG1
A15
17
19
BOOST
LOGIC
SYNC
CLKOUT
SW2
+
A16
TG2
BST2
–
SHORTLED
+
SNSP
–
SNSN
A10
+
0.4V
–
A11
FB
+
OPENLED
+
Q
R
S
VREF
0.2V
A8
A6
–
1.15V
+
SS LATCH
IVINMON_INT
FB
21
24
22
27
28
37
1.2V
A12
SS RESET
–
–
+
14µA
–
+
+
–
ISMON_INT
+
3V
CTRL
1
A9
–
1.75V
INTVCC
PWM
18
INTVCC
RT
A5
3
BG2
SLOPE_COMP_BOOST
SLOPE_COMP_BUCK
4
16
INTVCC
A7
5
14
BUCK
LOGIC
PGND
OSC
15
A18
A17
PWMOUT
31
SGND
30, 39
1.4µA
–
38
VC
SS
2
OVLO
36
3791 BD
3791f
11
LT3791
OPERATION
The LT3791 is a current mode controller that provides an
output voltage above, equal to or below the input voltage.
The LTC proprietary topology and control architecture uses
a current sensing resistor in buck or boost operation. The
sensed inductor current is controlled by the voltage on
the VC pin, which is the output of the feedback amplifiers
A11 and A12. The VC pin is controlled by three inputs,
one input from the output current loop, one input from the
input current loop, and the third input from the feedback
loop. Whichever feedback input is higher takes precedence,
forcing the converter into either a constant-current or a
constant-voltage mode.
The LT3791 is designed to transition cleanly between
the two modes of operation. Current sense amplifier A1
senses the voltage between the IVINP and IVINN pins and
provides a pre-gain to amplifier A11. When the voltage
between IVINP and IVINN reaches 50mV, the output of A1
provides IVINMON_INT to the inverting input of A11 and
the converter is in constant-current mode. If the current
sense voltage exceeds 50mV, the output of A1 increases
causing the output of A11 to decrease, thus reducing the
amount of current delivered to the output. In this manner
the current sense voltage is regulated to 50mV.
The output current amplifier works similar to the input
current amplifier but with a 100mV voltage instead of
50mV. The output current sense level is also adjustable
by the CTRL pin. Forcing CTRL to less than 1.2V forces
ISMON_INT to the same level as CTRL, thus providing
current-level control. The output current amplifier provides
rail-to-rail operation. Similarly if the FB pin goes above
1.2V the output of A11 decreases to reduce the current
level and regulate the output (constant-voltage mode).
The LT3791 provides monitoring pins IVINMON and ISMON
that are proportional to the voltage across the input and
output current amplifiers respectively.
The main control loop is shut down by pulling the EN/
UVLO pin low. When the EN/UVLO pin is higher than 1.2V,
an internal 14µA current source charges soft-start capacitor CSS at the SS pin. The VC voltage is then clamped a
diode voltage higher than the SS voltage while the CSS is
slowly charged during start-up. This “soft-start” clamping
prevents abrupt current from being drawn from the input
power supply. The SS can also be used as a fault timer
whenever an open or shorted LED is detected.
The top MOSFET drivers are biased from floating bootstrap capacitors C1 and C2, which are normally recharged
through an external diode when the top MOSFET is turned
off. Schottky diodes across the synchronous switch M4
and synchronous switch M2 are not required, but they do
provide a lower drop during the dead time. The addition
of the Schottky diode typically improves peak efficiency
by 1% to 2% at 500kHz.
Power Switch Control
Figure 1 shows a simplified diagram of how the four
power switches are connected to the inductor, VIN, VOUT
and GND. Figure 2 shows the regions of operation for the
LT3791 as a function of duty cycle D. The power switches
are properly controlled so the transfer between regions is
continuous. When VIN approaches VOUT, the buck-boost
region is reached.
VOUT
VIN
TG1
M1
L1
SW1
BG1
TG2
M4
M2
SW2
M3
BG2
RSENSE
3791 F01
Figure 1. Simplified Diagram of the Output Switches
DMAX
BOOST
(BG2)
DMIN
BOOST
DMAX
BUCK
(TG1)
DMIN
BUCK
BOOST REGION
BUCK-BOOST REGION
BUCK REGION
M1 ON, M2 OFF
PWM M3, M4 SWITCHES
4-SWITCH PWM
M4 ON, M3 OFF
PWM M2, M1 SWITCHES
3791 F02
Figure 2. Operating Regions vs Duty Cycle
3791f
12
LT3791
OPERATION
Buck Region (VIN > VOUT)
where D(BUCK-BOOST) is the duty cycle of the buck-boost
switch range:
Switch M4 is always on and switch M3 is always off during
this mode. At the start of every cycle, synchronous switch
M2 is turned on first. Inductor current is sensed when
synchronous switch M2 is turned on. After the sensed
inductor current falls below the reference voltage, which
is proportional to VC, synchronous switch M2 is turned off
and switch M1 is turned on for the remainder of the cycle.
Switches M1 and M2 will alternate, behaving like a typical
synchronous buck regulator. The duty cycle of switch M1
increases until the maximum duty cycle of the converter
in buck operation reaches DMAX(BUCK, TG1), given by:
D(BUCK-BOOST) = 8%
Figure 3 shows typical buck operation waveforms. If VIN
approaches VOUT, the buck-boost region is reached.
Buck-Boost (VIN ~ VOUT)
When VIN is close to VOUT, the controller is in buck-boost
operation. Figure 4 and Figure 5 show typical waveforms
in this mode. Every cycle the controller turns on switches
M2 and M4, then M1 and M4 are turned on until 180° later
when switches M1 and M3 turn on, and then switches
M1 and M4 are turned on for the remainder of the cycle.
DMAX(BUCK,TG1) = 100% – D(BUCK-BOOST)
M2 + M4
M2 + M4
M1 + M4
M2 + M4
M1 + M4
M1 + M4
3791 F03
Figure 3. Buck Operation (VIN > VOUT)
M1 + M4
M1+ M3
M1 + M4
M2 + M4
M1+ M3
M1 + M4
M1 + M4
M2 + M4
M1+ M3
M1 + M4
M2 + M4
M1 + M4
3791 F04
Figure 4. Buck-Boost Operation (VIN ≤ VOUT)
M1 + M4
M2 + M4
M1 + M3
M1 + M4
M1 + M4
M2 + M4
M1 + M3
M1 + M4
M1 + M4
M2 + M4
M1 + M3
M1 + M4
3791 F05
Figure 5. Buck-Boost Operation (VIN ≥ VOUT)
3791f
13
LT3791
OPERATION
Boost Region (VIN < VOUT)
Low Current Operation
Switch M1 is always on and synchronous switch M2 is
always off in boost operation. Every cycle switch M3 is
turned on first. Inductor current is sensed when synchronous switch M3 is turned on. After the sensed inductor
current exceeds the reference voltage which is proportional
to VC, switch M3 turns off and synchronous switch M4
is turned on for the remainder of the cycle. Switches M3
and M4 alternate, behaving like a typical synchronous
boost regulator.
The LT3791 runs in forced continuous mode. In this mode
the controller behaves as a continuous, PWM current
mode synchronous switching regulator. In boost operation, switch M1 is always on, switch M3 and synchronous
switch M4 are alternately turned on to maintain the output
voltage independent of the direction of inductor current.
In buck operation, synchronous switch M4 is always on,
switch M1 and synchronous switch M2 are alternately
turned on to maintain the output voltage independent of
the direction of inductor current. In this mode, the output
can source or sink current.
The duty cycle of switch M3 decreases until the minimum
duty cycle of the converter in boost operation reaches
DMIN(BOOST,BG2), given by:
DMIN(BOOST,BG2) = D(BUCK-BOOST)
where D(BUCK-BOOST) is the duty cycle of the buck-boost
switch range:
D(BUCK-BOOST) = 8%
Figure 6 shows typical boost operation waveforms. If VIN
approaches VOUT, the buck-boost region is reached.
M1 + M3
M1 + M4
M1 + M3
M1 + M4
M1 + M3
M1 + M4
3791 F06
Figure 6. Boost Operation (VIN < VOUT)
3791f
14
LT3791
APPLICATIONS INFORMATION
The Typical Application on the front page is a basic LT3791
application circuit. External component selection is driven
by the load requirement, and begins with the selection of
RSENSE and the inductor value. Next, the power MOSFETs
are selected. Finally, CIN and COUT are selected. This circuit
can operate up to an input voltage of 60V.
Programming The Switching Frequency
The RT frequency adjust pin allows the user to program the
switching frequency from 200kHz to 700kHz to optimize
efficiency/performance or external component size. Higher
frequency operation yields smaller component size but
increases switching losses and gate driving current, and
may not allow sufficiently high or low duty cycle operation.
Lower frequency operation gives better performance at the
cost of larger external component size. For an appropriate
RT resistor value see Table 1. An external resistor from
the RT pin to GND is required; do not leave this pin open.
The rising edge of CLK_OUT corresponds to the rising edge
of SYNC thus allowing paralleling converters. The falling
edge of CLK_OUT turns on switch M3 and the rising edge
of CLK_OUT turns on switch M2.
Inductor Selection
The operating frequency and inductor selection are interrelated in that higher operating frequencies allow the use
of smaller inductor and capacitor values. The inductor
value has a direct effect on ripple current. The maximum
inductor current ripple ΔIL can be seen in Figure 7. This
is the maximum ripple that will prevent subharmonic
oscillation and also regulate with zero load. The ripple
should be less than this to allow proper operation over
all load currents. For a given ripple the inductance terms
in continuous mode are as follows:
LBUCK >
Table 1. Switching Frequency vs RT Value
fOSC (kHz)
RT (kΩ)
200
147
300
84.5
400
59.0
500
45.3
600
35.7
700
29.4
Frequency Synchronization
The LT3791 switching frequency can be synchronized
to an external clock using the SYNC pin. Driving SYNC
with a 50% duty cycle waveform is always a good choice,
otherwise maintain the duty cycle between 10% and 90%.
(
)
VOUT • VIN(MAX) – VOUT • 100
LBOOST >
f •ILED • %Ripple • VIN(MAX)
(
)
VIN(MIN)2 • VOUT – VIN(MIN) • 100
f •ILED • %Ripple • VOUT 2
where:
f is operating frequency
% ripple is allowable inductor current ripple
VIN(MIN) is minimum input voltage
VIN(MAX) is maximum input voltage
VOUT is output voltage
ILED is current through the LEDs
3791f
15
LT3791
APPLICATIONS INFORMATION
where ΔIL is peak-to-peak inductor ripple current. In buck
operation, the maximum average load current is:
200
180
∆IL/ISENSE(MAX) (%)
160
140
120
100
BOOST ∆IL/
ISENSE(MAX) LIMIT
The maximum current sensing RSENSE value for the boost
operation is:
80
60
40
BUCK ∆IL/
ISENSE(MAX) LIMIT
20
0
 47.5mV ∆IL 
IOUT(MAX _ BUCK) = 
+
2 
 RSENSE
50 55 60 65 70 75 80 85 90 95 100
BG1, BG2 DUTY CYCLE (%)
3791 F07
Figure 7. Maximum Peak-to-Peak Ripple vs Duty Cycle
For high efficiency, choose an inductor with low core
loss. Also, the inductor should have low DC resistance to
reduce the I2R losses, and must be able to handle the peak
inductor current without saturating. To minimize radiated
noise, use a shielded inductor.
RSENSE(MAX) =
2 • 51mV• VIN(MIN)
2 •ILED • VOUT + ∆IL(BOOST) • VIN(MIN)
The maximum current sensing RSENSE value for the buck
operation is:
RSENSE(MAX) =
2 • 47.5mV
2 •ILED – ∆IL(BUCK)
The final RSENSE value should be lower than the calculated
RSENSE(MAX) in both the boost and buck operation. A 20%
to 30% margin is usually recommended.
RSENSE Selection and Maximum Output Current
CIN and COUT Selection
RSENSE is chosen based on the required output current. The
current comparator threshold sets the peak of the inductor current in boost operation and the maximum inductor
valley current in buck operation. In boost operation, the
maximum average load current at VIN(MIN) is:
In boost operation, input current is continuous. In buck
operation, input current is discontinuous. In buck operation, the selection of input capacitor, CIN, is driven by the
need to filter the input square wave current. Use a low ESR
capacitor sized to handle the maximum RMS current. For
buck operation, the input RMS current is given by:
 51mV ∆IL  VIN(MIN)
IOUT(MAX _ BOOST) = 
–
•
 RSENSE 2  VOUT
IRMS = ILED2 •D+
∆IL 2
•D
12
3791f
16
LT3791
APPLICATIONS INFORMATION
The formula has a maximum at VIN = 2VOUT. Note that
ripple current ratings from capacitor manufacturers are
often based on only 2000 hours of life which makes it
advisable to derate the capacitor.
In boost operation, the discontinuous current shifts
from the input to the output, so COUT must be capable
of reducing the output voltage ripple. The effects of ESR
(equivalent series resistance) and the bulk capacitance
must be considered when choosing the right capacitor
for a given output ripple voltage. The steady ripple due to
charging and discharging the bulk capacitance is given by:
∆VRIPPLE (BOOST _ CAP ) =
(
ILED • VOUT – VIN(MIN)
)
C OUT • VOUT • f
∆IL
∆VRIPPLE (BUCK _ CAP ) ≈
8 • f •C
Programming VIN UVLO and OVLO
The falling UVLO value can be accurately set by the resistor
divider R1 and R2. A small 3µA pull-down current is active
when the EN/UVLO is below the threshold. The purpose
of this current is to allow the user to program the rising
hysteresis. The following equations should be used to
determine the resistor values:
VIN(UVLO –) = 1.2 •
VIN(OVLO + ) = 3 •
where COUT is the output filter capacitor.
ΔVBOOST(ESR) = ILED • ESR
VIN(UVLO + ) = 3µA •R1+1.215 •
R3+R4
R4
VIN(OVLO –) = 2.925 •
R3+R4
R4
VIN
ΔVBUCK(ESR) = ILED • ESR
Multiple capacitors placed in parallel may be needed to
meet the ESR and RMS current handling requirements.
Output capacitors are also used for stability for the LT3791.
A good starting point for output capacitors is seen in the
Typical Applications circuits. Ceramic capacitors have
excellent low ESR characteristics but can have a high
voltage coefficient and are recommended for applications
less than 100W. Capacitors available with low ESR and
high ripple current ratings, such as OS-CON and POSCAP
may be needed for applications greater than 100W.
R1+R2
R2
The rising OVLO value can be accurately set by the resistor divider R3 and R4. The following equations should be
used to determine the resistor values:
OUT
The steady ripple due to the voltage drop across the ESR
is given by:
R1+R2
R2
LT3791
R1
R3
R2
R4
OVLO
EN/UVLO
3791 F08
Figure 8. Resistor Connection to Set VIN UVLO and
OVLO Thresholds
3791f
17
LT3791
APPLICATIONS INFORMATION
Programming LED Current
The LED current is programmed by placing an appropriate
value current sense resistor, RLED, in series with the LED
string. The voltage drop across RLED is (Kelvin) sensed
by the ISP and ISN pins. The CTRL pin should be tied to
a voltage higher than 1.2V to get the full-scale 100mV
(typical) threshold across the sense resistor. The CTRL
pin can also be used to dim the LED current, although
relative accuracy decreases with the decreasing sense
threshold. When the CTRL pin voltage is less than 1V,
the LED current is:
ILED =
VCTRL – 200mV
RLED • 10
When the CTRL pin voltage is between 1.1V and 1.3V
the LED current varies with VCTRL, but departs from the
equation above by an increasing amount as VCTRL voltage
increases. Ultimately, when VCTRL > 1.3V the LED current
no longer varies. The typical V(ISP-ISN) threshold vs VCTRL
is listed in Table 2.
Table 2. V(ISP-ISN) Threshold vs CTRL
VCTRL (V)
V(ISP-ISN) (mV)
1.1
90
1.15
94.5
1.2
98
1.25
99.5
1.3
100
When VCTRL is higher than 1.3V, the LED current is
regulated to:
ILED =
100mV
RLED
The CTRL pin should not be left open (tie to VREF if not
used). The CTRL pin can also be used in conjunction with
a thermistor to provide overtemperature protection for
the LED load, or with a resistor divider to VIN to reduce
output power and switching current when VIN is low.
The presence of a time varying differential voltage signal
(ripple) across ISP and ISN at the switching frequency
is expected. The amplitude of this signal is increased by
high LED load current, low switching frequency and/or a
smaller value output filter capacitor. Some level of ripple
signal is acceptable: the compensation capacitor on the
VC pin filters the signal so the average difference between
ISP and ISN is regulated to the user-programmed value.
Ripple voltage amplitude (peak-to-peak) in excess of
20mV should not cause mis-operation, but may lead to
noticeable offset between the average value and the userprogrammed value.
ISMON
The ISMON pin provides a linear indication of the current flowing through the LEDs. The equation for VISMON
is V(ISP–ISN) • 10. This pin is suitable for driving an ADC
input, however, the output impedance of this pin is 12.5kΩ
so care must be taken not to load this pin.
Programming Input Current Limit
The LT3791 has a standalone current sense amplifier. It
can be used to limit the input current. The input current
limit is calculated by the following equation:
IIN =
50mV
RIN
3791f
18
LT3791
APPLICATIONS INFORMATION
For loop stability a lowpass RC filter is needed. For
most applications, a 50Ω resistor and 470nF capacitor
is sufficient.
VOUT
FB
Table 3
R6
RIN (mΩ)
20
15
12
10
6
5
4
3
2
ILIMIT (A)
2.5
3.3
4.2
5.0
8.3
10.0
12.5
16.7
25
IVINMON
The IVINMON pin provides a linear indication of the current
flowing through the input. The equation for VIVINMON is
V(IVINP-IVINN) • 20. This pin is suitable for driving an ADC
input, however, the output impedance of this pin is 12.5kΩ
so care must be taken not to load this pin.
Programming Output Voltage (Constant Voltage
Regulation) or Open LED/Overvoltage Threshold
For a voltage regulator, the output voltage can be set by
selecting the values of R5 and R6 (see Figure 9) according
to the following equation:
R5
LT3791
VOUT = 1.2 •
R5+R6
R6
For an LED driver application, set the resistor from the
output to the FB pin such that the expected VFB during
normal operation does not exceed 1.1V. Once VFB is higher
than its overvoltage threshold, 1.25V (typical), the LT3791
stops switching.
3791 F09
Figure 9. Resistor Connection for Open LED Threshold
and Constant Output Voltage Regulation
Dimming Control
There are two methods to control the current source for
dimming using the LT3791. One method uses the CTRL
pin to adjust the current regulated in the LEDs. A second
method uses the PWM pin to modulate the current source
between zero and full current to achieve a precisely programmed average current. To make PWM dimming more
accurate, the switch demand current is stored on the VC
node during the quiescent phase when PWM is low. This
feature minimizes recovery time when the PWM signal goes
high. To further improve the recovery time a disconnect
switch may be used in the LED current path to prevent the
ISP node from discharging during the PWM signal low
phase. The minimum PWM on- or off-time is affected by
choice of operating frequency and external component
selection. The best overall combination of PWM and
analog dimming capabilities is available if the minimum
PWM pulse is at least six switching cycles and the PWM
pulse is synchronized to the SYNC signal.
SHORTLED Pin
The LT3791 provides an open-drain status pin,
SHORTLED, which pulls low when the FB pin is below
400mV. The only time the FB pin will be below 400mV
is during start-up or if the LEDs are shorted. During
3791f
19
LT3791
APPLICATIONS INFORMATION
start-up the LT3791 ignores the voltage on the FB pin
until the soft-start capacitor reaches 1.75V. To prevent
false tripping after startup, a large enough soft-start
capacitor must be used to allow the output to get up to
approximately 40% to 50% of the final value.
OPENLED Pin
The LT3791 provides an open-drain status pin, OPENLED,
which pulls low when the FB pin is above 1.15V and the
voltage across V(ISP-ISN) is less than 10mV. If the open
LED clamp voltage is programmed correctly using the FB
pin, then the FB pin should never exceed 1.1V when the
LEDs are connected. Therefore, the only way for the FB
pin to exceed 1.15V is for an open LED event to occur.
Soft-Start, Fault Function
Soft-start reduces the input power sources’ surge currents
by gradually increasing the controller’s current limit (proportional to an internally buffered clamped equivalent of
VC). The soft-start interval is set by the soft-start capacitor
selection according to the following equation
t SS =
1.2V
•C
14µA SS
Make sure CSS is large enough when there is loading
during start-up.
The SS pin is also used as a fault timer. Once an open
LED or a shorted LED fault is detected, a 1.4µA pull-down
current source is activated. With a 100k pull-up resistor
to VREF on the SS pin, the LT3791 will continue to switch
normally. With a 500k pull-up resistor to VREF on the SS
pin, the LT3791 will latch off until the EN/UVLO pin is
toggled. Without any resistor to VREF the SS pin enters
a hiccup mode operation. The 1.4µA pulls SS down until
0.2V is reached, at which point the 14µA pull-up current
source turns on. If the fault condition hasn’t been removed
when SS reaches 1.75V, then the 1.4µA pull-down current source turns on again initiating a new cycle. This will
continue until the fault is removed.
Loop Compensation
The LT3791 uses an internal transconductance error amplifier whose VC output compensates the control loop. The
external inductor, output capacitor and the compensation
resistor and capacitor determine the loop stability.
The inductor and output capacitor are chosen based on
performance, size and cost. The compensation resistor
and capacitor at VC are set to optimize control loop response and stability. For typical LED applications, a 10nF
compensation capacitor at VC is adequate, and a series
resistor should always be used to increase the slew rate
on the VC pin to maintain tighter regulation of LED current
during fast transients on the input supply of the converter.
3791f
20
LT3791
APPLICATIONS INFORMATION
Power MOSFET Selections and Efficiency
Considerations
The LT3791 requires four external N-channel power MOSFETs, two for the top switches (switch M1 and M4, shown in
Figure 1) and two for the bottom switches (switch M2 and
M3 shown in Figure 1). Important parameters for the power
MOSFETs are the breakdown voltage, VBR(DSS), threshold
voltage, VGS(TH), on-resistance, RDS(ON), reverse transfer
capacitance, CRSS, and maximum current, IDS(MAX).
The drive voltage is set by the 5V INTVCC supply. Consequently, logic-level threshold MOSFETs must be used
in LT3791 applications. If the input voltage is expected
to drop below the 5V, then sub-logic threshold MOSFETs
should be considered.
In order to select the power MOSFETs, the power dissipated by the device must be known. For switch M1, the
maximum power dissipation happens in boost operation,
when it remains on all the time. Its maximum power dissipation at maximum output current is given by:
2
I •V 
PM1(BOOST) =  LED OUT  • ρT •RDS(ON)
VIN


where ρT is a normalization factor (unity at 25°C)
accounting for the significant variation in on-resistance
with temperature, typically 0.4%/°C as shown in Figure 10.
For a maximum junction temperature of 125°C, using a
value of ρT = 1.5 is reasonable.
Switch M2 operates in buck operation as the synchronous
rectifier. Its power dissipation at maximum output current
is given by:
PM2(BUCK) =
VIN – VOUT
•ILED2 • ρT •RDS(ON)
VIN
Switch M3 operates in boost operation as the control
switch. Its power dissipation at maximum current is
given by:
PM3(BOOST) =
( VOUT – VIN ) • VOUT •I
VIN2
+ k • VOUT 3 •
LED
2 •ρ
T
•RDS(ON)
ILED
•C
•f
VIN ROSS
where CRSS is usually specified by the MOSFET manufacturers. The constant k, which accounts for the loss caused
by reverse-recovery current, is inversely proportional to
the gate drive current and has an empirical value of 1.7.
For switch M4, the maximum power dissipation happens
in boost operation, when its duty cycle is higher than
50%. Its maximum power dissipation at maximum output
current is given by:
V
PM4(BOOST) = IN
VOUT
2
I •V 
•  LED OUT  • ρT •RDS(ON)
VIN


For the same output voltage and current, switch M1 has
the highest power dissipation and switch M2 has the lowest power dissipation unless a short occurs at the output.
3791f
21
LT3791
APPLICATIONS INFORMATION
From a known power dissipated in the power MOSFET, its
junction temperature can be obtained using the following
formula:
TJ = TA + P • RTH(JA)
The RTH(JA) to be used in the equation normally includes
the RTH(JC) for the device plus the thermal resistance from
the case to the ambient temperature (RTH(JC)). This value
of TJ can then be compared to the original, assumed value
used in the iterative calculation process.
ρT NORMALIZED ON-RESISTANCE (Ω)
2.0
1.5
1.0
0.5
0
–50
50
100
0
JUNCTION TEMPERATURE (°C)
150
3791 F10
Figure 10. Normalized RDS(ON) vs Temperature
Optional Schottky Diode (D3, D4) Selection
The Schottky diodes D3 and D4 shown in the Typical Applications section conduct during the dead time between
the conduction of the power MOSFET switches. They
are intended to prevent the body diode of synchronous
switches M2 and M4 from turning on and storing charge
during the dead time. In particular, D4 significantly reduces
reverse-recovery current between switch M4 turn-off and
switch M3 turn-on, which improves converter efficiency
and reduces switch M3 voltage stress. In order for the
diode to be effective, the inductance between it and the
synchronous switch must be as small as possible, mandating that these components be placed adjacently.
INTVCC Regulator
An internal P-channel low dropout regulator produces 5V
at the INTVCC pin from the VIN supply pin. INTVCC powers
the drivers and internal circuitry within the LT3791. The
INTVCC pin regulator can supply a peak current of 67mA
and must be bypassed to ground with a minimum of 4.7µF
ceramic capacitor or low ESR electrolytic capacitor. An
additional 0.1µF ceramic capacitor placed directly adjacent
to the INTVCC and PGND IC pins is highly recommended.
Good bypassing is necessary to supply the high transient
current required by MOSFET gate drivers.
Higher input voltage applications in which large MOSFETs
are being driven at high frequencies may cause the maximum junction temperature rating for the LT3791 to be
exceeded. The system supply current is normally dominated
by the gate charge current. Additional external loading of
the INTVCC also needs to be taken into account for the
power dissipation calculations. Power dissipation for the
IC in this case is VIN • IINTVCC, and overall efficiency is
lowered. The junction temperature can be estimated by
using the equations given
TJ = TA + (PD • θJA)
where θJA (in °C/W) is the package thermal impedance.
For example, a typical application operating in continuous
current operation might draw 24mA from a 24V supply:
TJ = 70°C + 24mA • 24V • 28°C/W = 86°C
3791f
22
LT3791
APPLICATIONS INFORMATION
To prevent maximum junction temperature from being
exceeded, the input supply current must be checked
operating in continuous mode at maximum VIN.
Top Gate (TG) MOSFET Driver Supply (C1, D1, C2, D2)
The external bootstrap capacitors C1 and C2 connected
to the BST1 and BST2 pins supply the gate drive voltage
for the topside MOSFET switches M1 and M4. When the
top MOSFET switch M1 turns on, the switch node SW1
rises to VIN and the BST1 pin rises to approximately VIN +
INTVCC. When the bottom MOSFET switch M2 turns on, the
switch node SW1 drops low and the bootstrap capacitor
C1 is charged through D1 from INTVCC. When the bottom
MOSFET switch M3 turns on, the switch node SW2 drops
low and the bootstrap capacitor C2, is charged through D2
from INTVCC. The bootstrap capacitors C1 and C2 need to
store about 100 times the gate charge required by the top
MOSFET switch M1 and M4. In most applications a 0.1µF
to 0.47µF, X5R or X7R ceramic capacitor is adequate.
Efficiency Considerations
The power efficiency of a switching regulator is equal to
the output power divided by the input power times 100%.
It is often useful to analyze individual losses to determine
what is limiting the efficiency and which change would
produce the most improvement. Although all dissipative
elements in circuits produce losses, four main sources
account for most of the losses in LT3791 circuits:
1. DC I2R losses. These arise from the resistances of the
MOSFETs, sensing resistor, inductor and PC board
traces and cause the efficiency to drop at high output
currents.
2. Transition loss. This loss arises from the brief amount
of time switch M1 or switch M3 spends in the saturated
region during switch node transitions. It depends upon
the input voltage, load current, driver strength and
MOSFET capacitance, among other factors. The loss
is significant at input voltages above 20V and can be
estimated from:
Transition Loss ≈ 2.7 • VIN2 • IOUT • CRSS • f
where CRSS is the reverse-transfer capacitance.
3. INTVCC current. This is the sum of the MOSFET driver
and control currents.
4. CIN and COUT loss. The input capacitor has the difficult
job of filtering the large RMS input current to the regulator in buck operation. The output capacitor has the
difficult job of filtering the large RMS output current
in boost operation. Both CIN and COUT are required to
have low ESR to minimize the AC I2R loss and sufficient
capacitance to prevent the RMS current from causing
additional upstream losses in fuses or batteries.
5. Other losses. Schottky diode D3 and D4 are responsible for conduction losses during dead time and light
load conduction periods. Inductor core loss occurs
predominately at light loads. Switch M3 causes reverse
recovery current loss in boost operation.
When making adjustments to improve efficiency, the input
current is the best indicator of changes in efficiency. If you
make a change and the input current decreases, then the
efficiency has increased. If there is no change in the input
current, then there is no change in efficiency.
3791f
23
LT3791
APPLICATIONS INFORMATION
PC Board Layout Checklist
The basic PC board layout requires a dedicated ground
plane layer. Also, for high current, a multilayer board
provides heat sinking for power components.
The PGND ground plane layer should not have any traces
and it should be as close as possible to the layer with
power MOSFETs.
n
Place CIN, switch M1, switch M2 and D1 in one compact
area. Place COUT, switch M3, switch M4 and D2 in one
compact area.
n
Use immediate vias to connect the components (including the LT3791’s SGND and PGND pins) to the ground
plane. Use several large vias for each power component.
n
The path formed by switch M1, switch M2, D1 and the
CIN capacitor should have short leads and PC trace
lengths. The path formed by switch M3, switch M4, D2
and the COUT capacitor also should have short leads
and PC trace lengths.
n
The output capacitor (–) terminals should be connected
as close as possible to the (–) terminals of the input
capacitor.
n
Connect the top driver bootstrap capacitor, C1, closely
to the BST1 and SW1 pins. Connect the top driver
bootstrap capacitor, C2, closely to the BST2 and SW2
pins.
n
Connect the input capacitors, CIN, and output capacitors,
COUT, closely to the power MOSFETs. These capacitors carry the MOSFET AC current in boost and buck
operation.
n
Use planes for VIN and VOUT to maintain good voltage
filtering and to keep power losses low.
n
Flood all unused areas on all layers with copper. Flooding
with copper will reduce the temperature rise of power
components. Connect the copper areas to any DC net
(VIN or PGND).
n
Separate the signal and power grounds. All small-signal
components should return to the SGND pin at one point,
which is then tied to the PGND pin close to the sources
of switch M2 and switch M3.
n
Place switch M2 and switch M3 as close to the controller as possible, keeping the PGND, BG and SW traces
short.
n
Keep the high dV/dT SW1, SW2, BST1, BST2, TG1 and
TG2 nodes away from sensitive small-signal nodes.
n
Route SNSN and SNSP leads together with minimum
PC trace spacing. Avoid sense lines pass through noisy
areas, such as switch nodes. Ensure accurate current
sensing with Kelvin connections at the SENSE resistor.
n
Connect the VC pin compensation network close to the
IC, between VC and the signal ground pins. The capacitor helps to filter the effects of PCB noise and output
voltage ripple voltage from the compensation loop.
n
Connect the INTVCC bypass capacitor, CVCC, close to the
IC, between the INTVCC and the power ground pins. This
capacitor carries the MOSFET drivers’ current peaks. An
additional 0.1µF ceramic capacitor placed immediately
next to the INTVCC and PGND pins can help improve
noise performance substantially.
n
3791f
24
LT3791
TYPICAL APPLICATIONS
98% Efficient 50W (25V 2A) Buck-Boost LED Driver
VIN
4.7V TO 58V
RIN
0.003Ω
VIN
C3
R7 1µF
50Ω
IVINN
BST1
TG1
IVINP
R3
1M
BG1
OVLO
INTVCC
R9
200k
R4
54.9k
LT3791
R10
200k
CSS
10nF
BG2
SW2
TG2
FB
ISN
PWMOUT
RT
SGND
R8
RC
86.6k
2.2k
300kHz
CC
10nF
D1, D2: NXP BAT46WJ
L1: COOPER HC9-100-R 10µH
M1, M2: RENESAS RJK0651DPB 60VDS
M3, M4: RENESAS RJK0451DPB 40VDS
M5: VISHAY Si2318CDS 40VDS
M5
3791 TA02a
100Hz 50:1 PWM Dimming (VIN = 12V)
100
EFFICIENCY (%)
RLED
0.05Ω
25V LED
2A
Efficiency vs VIN
96
R6
44.2k
ISP
CTRL
TEST1
SS
SYNC VC
98
M3
COUT
4.7µF
50V
×4
SNSN
PGND
VREF
R12
237k
L1 10µH
M4
R5
1M
RSENSE
0.004Ω
SHORTLED
PWM
IVINMON
ISMON
CLKOUT
C8
0.1µF
M2
C2
0.1µF
C1
0.1µF
CVCC
4.7µF
SNSP
OPENLED
R11
1M
M1
SWI
EN/UVLO
R2
121k
D1 D2
TEST2
BST2
C7
470nF
R1
332k
INTVCC
CIN
2.2µF
100V
×4
PWM
5V/DIV
BOOST
94
BUCK
BUCK-BOOST
IL1
2A/DIV
92
90
ILED
2A/DIV
88
86
50µs/DIV
84
3791 TA02c
82
80
0
10
30
40
20
INPUT VOLTAGE (V)
50
60
3791 TA02b
3791f
25
LT3791
TYPICAL APPLICATIONS
98% Efficient 60W (12V 5A) Voltage Regulator Runs Down to 3V VIN
VIN
3V TO 55V
CIN
4.7µF
100V
×4
R10
200k
INTVCC
TEST2
SHORTLED
D5
BST2
OPENLED
BST1
VIN
C3
1µF
TG1
BG1
IVINP
D6
R1
866k
TEST1
SNSP
LT3791
IVINMON
ISMON
CLKOUT
R3
1M
EN/UVLO
OVLO
R2
576k
R4
57.6k
C8
0.1µF
L1 6.8µH
SS SYNC VC
RSENSE
0.004Ω
SNSN
PGND
R5
732k
BG2
R6
80.6k
SGND PWMOUT
RT
RC
2.2k
CC
22nF
R8
86.6k
300kHz
3791 TA03a
D1, D2: NXP BAT46WJ
D3: IRF 10BQ060
D4: IRF 10BQ040
D5, D6: DIODES INC. BAT46W
L1: WURTH ELEKTRONIK WE-HCI 7443556680
M1, M2: RENASAS RJK0651DPB 60VDS
M3, M4: VISHAY SiR424DP 40VDS
6
BUCK
96
EFFICIENCY (%)
Maximum Output Current vs VIN
IOUT = 5A
BOOST
BUCK-BOOST
94
COUT
10µF
25V
×3
VOUT
12V
5A
M3
D3
MAXIMUM OUTPUT CURRENT (A)
98
COUT2
100µF
25V ROUT
0.015Ω
D4 M4
Efficiency vs VIN
100
+
ISP
ISN
CTRL
CSS
10nF
M2
C2
0.1µF
C1
0.1µF
SW2
TG2
FB
PWM
VREF
RFAULT
100k
M1
SWI
IVINN
CVCC
4.7µF
D1 D2
92
90
88
86
84
5
4
3
2
1
82
80
0
10
30
40
20
INPUT VOLTAGE (V)
50
60
3791 TA03b
0
3
4
5
6
7 8 9 10 20 30 40 50 60
INPUT VOLTAGE (V)
3791 TA03c
3791f
26
LT3791
PACKAGE DESCRIPTION
Please refer to http://www.linear.com/designtools/packaging/ for the most recent package drawings.
FE Package
38-Lead Plastic TSSOP (4.4mm)
(Reference LTC DWG # 05-08-1772 Rev C)
Exposed Pad Variation AA
4.75 REF
38
9.60 – 9.80*
(.378 – .386)
4.75 REF
(.187)
20
6.60 ±0.10
4.50 REF
2.74 REF
SEE NOTE 4
6.40
2.74
REF (.252)
(.108)
BSC
0.315 ±0.05
1.05 ±0.10
0.50 BSC
RECOMMENDED SOLDER PAD LAYOUT
4.30 – 4.50*
(.169 – .177)
0.50 – 0.75
(.020 – .030)
0.09 – 0.20
(.0035 – .0079)
NOTE:
1. CONTROLLING DIMENSION: MILLIMETERS
2. DIMENSIONS ARE IN MILLIMETERS
(INCHES)
3. DRAWING NOT TO SCALE
1
0.25
REF
19
1.20
(.047)
MAX
0° – 8°
0.50
(.0196)
BSC
0.17 – 0.27
(.0067 – .0106)
TYP
0.05 – 0.15
(.002 – .006)
FE38 (AA) TSSOP REV C 0910
4. RECOMMENDED MINIMUM PCB METAL SIZE
FOR EXPOSED PAD ATTACHMENT
*DIMENSIONS DO NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.150mm (.006") PER SIDE
3791f
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
27
LT3791
TYPICAL APPLICATION
120W (24V 5A) Buck-Boost Voltage Regulator
VIN
12V TO 58V
R10
200k
INTVCC
TEST2
SHORTLED
BST2
OPENLED
BST1
VIN
C3
1µF
TG1
BG1
IVINP
TEST1
R1
499k
EN/UVLO
OVLO
R2
56.2k
R4
27.4k
C8
0.1µF
C2
0.1µF
C1
0.1µF
L1
10µH
SS SYNC VC
RT
COUT
4.7µF
50V
×6
M4
RC
1.1k
CC
22nF
CSS
10nF
R8
147k
200kHz
VOUT
24V
5A
M3
SNSN
PGND
R4
732k
BG2
R5
18.7k
SGND PWMOUT
CIN2
47µF
100V
ROUT
0.015Ω
ISP
ISN
CTRL
+
RSENSE
0.004Ω
SW2
TG2
FB
PWM
VREF
RFAULT
100k
M2
SNSP
LT3791
IVINMON
ISMON
CLKOUT
R3
499k
M1
SWI
IVINN
CVCC
4.7µF
D1 D2
CIN
2.2µF
100V
3791 TA04
D1, D2: NXP BAT46WJ
L1: WURTH ELEKTRONICS 74435571100 10µH
M1, M2: RENESAS RJK0651DPB 60VDS
M3, M4: RENESAS RJK0451DPB 40VDS
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
®
LTC 3780
High Efficiency, Synchronous, 4-Switch Buck-Boost
Controller
VIN: 4V to 36V, VOUT Range: 0.8V to 30V, ISD < 55µA, SSOP-24, QFN-32
Packages
LTC3789
High Efficiency, Synchronous, 4-Switch Buck-Boost
Controller
VIN: 4V to 38V, VOUT Range: 0.8V to 38V, ISD < 40µA, 4mm × 5mm QFN-28,
SSOP-28 Packages
LT3755/LT3755-1 High Side 60V, 1MHz LED Controller with True Color
LT3755-2
3000:1 PWM Dimming
VIN: 4.5V to 40V, VOUT Range: 5V to 60V, 3000:1 True Color PWM™, Analog,
ISD < 1µA, 3mm × 3mm QFN-16, MSOP-16E Packages
LT3756/LT3756-1 High Side 100V, 1MHz LED Controller with True Color VIN: 6V to 100V, VOUT Range: 5V to 100V, 3000:1 True Color PWM, Analog,
ISD < 1µA, 3mm × 3mm QFN-16, MSOP-16E Packages
LT3756-2
3000:1 PWM Dimming
LT3596
60V, 300mA Step-Down LED Driver
VIN: 6V to 60V, VOUT Range: 5V to 55V, 10000:1 True Color PWM, Analog,
ISD < 1µA, 5mm × 8mm QFN-52 Package
LT3743
Synchronous Step-Down 20A LED Driver with
Thee-State LED Current Control
VIN: 5.5V to 36V, VOUT Range: 5.5V to 35V, 3000:1 True Color PWM, Analog,
ISD < 1µA, 4mm × 5mm QFN-28, TSSOP-28E Packages
3791f
28 Linear Technology Corporation
LT 0312 • PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 ● FAX: (408) 434-0507
●
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 LINEAR TECHNOLOGY CORPORATION 2012