LT3791 60V 4-Switch Synchronous Buck-Boost LED Driver Controller DESCRIPTION FEATURES 4-Switch Single Inductor Architecture Allows VIN Above, Below or Equal to VOUT n Wide V Range: 4.7V to 60V IN n Wide V OUT Range: 0V to 60V (55V LED) n±2% Output Voltage Accuracy n Synchronous Switching: Up to 98.5% Efficiency n ±6% LED Current Accuracy: 0V ≤ V OUT < 60V n V Disconnected from V During Shutdown OUT IN n Accurate Rail-to-Rail LED Current Sense with Monitor Output n Input Current Sense with Monitor Output n PWM and Analog Dimming n Capable of 100W or greater per IC n38-Lead TSSOP with Exposed Pad The LT®3791 is a synchronous 4-switch buck-boost LED driver and voltage regulator controller. The controller operates from input voltages above, below, or equal to the output voltage. The LT3791 has a wide 4.7V to 60V input and 0V to 60V output range along with seamless transitions between operating modes. A ground reference voltage FB pin serves as the input for several LED protection features and also makes it possible for the converter to operate as a constant-voltage source. The LT3791 is ideal for a wide variety of applications. n The LT3791 runs in forced continuous mode, which is ideal for systems with stringent EMI requirements. Fault protection is provided to survive and report an open or shorted LED condition. A timer allows the LT3791 to continue to run, latch off or restart when a fault occurs. APPLICATIONS L, LT, LTC, LTM, Linear Technology and the Linear logo are registered and True Color PWM is a trademark of Linear Technology Corporation. All other trademarks are the property of their respective owners. Automotive Headlamps/Running Lamps n General Purpose Lighting n TYPICAL APPLICATION 98.5% Efficient 100W (33.3V 3A) Buck-Boost LED Driver 2.2µF 100V ×5 0.003Ω VIN INTVCC 4.7µF 1µF IVINN BST1 470nF 28k TG1 IVINP 499k 15.8k SWI EN/UVLO OVLO INTVCC 200k Efficiency vs VIN BST2 50Ω BG1 LT3791 200k 0.1µF 0.1µF 10µH 1M 4.7µF 50V ×5 98 34.2k SNSP 0.004Ω SHORTLED OPENLED 0.033Ω SNSN PGND PWM IVINMON ISMON CLKOUT BUCK BUCK-BOOST 96 94 90 ISP 10 20 40 30 INPUT VOLTAGE (V) 50 60 3791 TA01b ISN 0.1µF 3A, 100W LED POWER BOOST 92 BG2 SW2 TG2 FB VREF CTRL 100 EFFICIENCY (%) VIN 15V TO 58V PWMOUT SS SYNC VC RT 2.2k 10nF SGND 86.6k 300kHz 10nF 3791 TA01a 3791f 1 LT3791 ABSOLUTE MAXIMUM RATINGS (Note 1) PIN CONFIGURATION Supply Voltages Input Supply (VIN)......................................................60V SW1, SW2.......................................................–1V to 60V OPENLED, SHORTLED................................................15V EN/UVLO, IVINP, IVINN, ISP, ISN...............................60V INTVCC, (BST1-SW1), (BST2-SW2)..............................6V TEST2, SYNC, RT, CTRL, OVLO, PWM........................6V IVINMON, ISMON, FB, SS, VC, VREF............................6V IVINP-IVINN, ISP-ISN, SNSP-SNSN........................±0.5V SNSP, SNSN............................................................±0.3V Operating Junction Temperature (Notes 2, 3) LT3791E/LT3791I................................ –40°C to 125°C LT3791H............................................. –40°C to 150°C LT3791MP.......................................... –55°C to 150°C Storage Temperature Range................... –65°C to 150°C Lead Temperature (Soldering, 10 sec).................... 300°C TOP VIEW CTRL 1 38 OVLO SS 2 37 FB PWM 3 36 VC OPENLED 4 35 RT SHORTLED 5 34 SYNC VREF 6 33 CLKOUT ISMON 7 32 TEST2 IVINMON 8 31 PWMOUT EN/UVLO 9 IVINP 10 30 SGND 39 SGND 29 TEST1 IVINN 11 28 SNSN VIN 12 27 SNSP INTVCC 13 26 ISN TG1 14 25 ISP BST1 15 24 TG2 SW1 16 23 NC PGND 17 22 BST2 BG1 18 21 SW2 BG2 19 20 PGND FE PACKAGE 38-LEAD PLASTIC TSSOP TJMAX = 150°C, θJA = 28°C/W EXPOSED PAD (PIN 39) IS SGND, MUST BE SOLDERED TO PCB ORDER INFORMATION LEAD FREE FINISH TAPE AND REEL PART MARKING* PACKAGE DESCRIPTION TEMPERATURE RANGE LT3791EFE#PBF LT3791EFE#TRPBF LT3791FE 38-Lead Plastic TSSOP –40°C to 125°C LT3791IFE#PBF LT3791IFE#TRPBF LT3791FE 38-Lead Plastic TSSOP –40°C to 125°C LT3791HFE#PBF LT3791HFE#TRPBF LT3791FE 38-Lead Plastic TSSOP –40°C to 150°C LT3791MPFE#PBF LT3791MPFE#TRPBF LT3791FE 38-Lead Plastic TSSOP –55°C to 150°C Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container. For more information on lead free part marking, go to: http://www.linear.com/leadfree/ For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/ ELECTRICAL CHARACTERISTICS The l denotes the specifications which apply over the full operating junction temperature range, otherwise specifications are at TA = 25°C (Note 2). VIN = 12V, VEN/UVLO = 12V unless otherwise noted. PARAMETER CONDITIONS MIN TYP MAX UNITS Input VIN Operating Voltage 4.7 60 V VIN Shutdown IQ VEN/UVLO = 0V 0.1 1 µA VIN Operating IQ (Not Switching) FB = 1.3V, RT = 59.0k 3.0 4 mA 3791f 2 LT3791 ELECTRICAL CHARACTERISTICS The l denotes the specifications which apply over the full operating junction temperature range, otherwise specifications are at TA = 25°C (Note 2). VIN = 12V, VEN/UVLO = 12V unless otherwise noted. PARAMETER CONDITIONS MIN TYP MAX 1.16 1.2 1.24 UNITS Logic Inputs EN/UVLO Falling Threshold l EN/UVLO Rising Hysteresis 15 EN/UVLO Input Low Voltage IVIN Drops Below 1µA EN/UVLO Pin Bias Current Low VEN/UVLO = 1V EN/UVLO Pin Bias Current High CTRL Input Bias Current V mV 0.3 V 3 4 µA VEN/UVLO = 1.6V 10 100 nA VCTRL = 1V 20 50 nA 2 CTRL Latch-Off Threshold 175 OVLO Rising Shutdown Voltage l 2.85 OVLO Falling Hysteresis 3 mV 3.15 75 V mV Regulation VREF Voltage VREF Line Regulation 4.7V < VIN < 60V V(ISP-ISN) Threshold VCTRL = 2V l 1.96 l 97.5 94 2.04 V 0.002 0.04 %/V 100 100 102.5 106 mV mV l 87 84 90 90 93 96 mV mV l 47.5 46 50 50 52.5 54 mV mV l 6.5 5 10 10 13.5 15 mV mV VCTRL = 1100mV VCTRL = 700mV VCTRL = 300mV ISP Bias Current 2.00 110 ISN Bias Current µA 20 LED Current Sense Common Mode Range 0 LED Current Sense Amplifier gm µA 60 890 µS ISMON Monitor Voltage V(ISP-ISN) = 100mV l 0.96 1 1.04 Input Current Sense Threshold V(IVINP-IVINN) 3V ≤ VIVINP ≤ 60V l 46.5 50 54 IVINP Bias Current 90 IVINN Bias Current 3 Input Current Sense Amplifier gm IVINMON Monitor Voltage 0.96 1 1.04 V l 1.194 1.176 1.2 1.2 1.206 1.220 V V 0.002 0.025 %/V 4.7V < VIN < 60V FB Amplifier gm FB Pin Input Bias Current 565 FB in Regulation VC Standby Input Bias Current PWM = 0V VSENSE(MAX) (VSNSP-SNSN) Boost Buck V mS l FB Regulation Voltage FB Line Regulation mV µA 60 2.12 V(IVINP-IVINN) = 50mV V µA 20 Input Current Sense Common Mode Range V 150 20 nA 51 –47.5 60 –39 mV mV –20 l l 42 –56 µS 100 nA Fault SS Pull-Up Current VSS = 0V SS Discharge Current FB Overvoltage Rising Threshold Open LED Rising Threshold (VFB) V(ISP-ISN) = 0V l 14 µA 1.4 µA 1.22 1.25 1.127 1.15 V 1.173 V 3791f 3 LT3791 ELECTRICAL CHARACTERISTICS The l denotes the specifications which apply over the full operating junction temperature range, otherwise specifications are at TA = 25°C (Note 2). VIN = 12V, VEN/UVLO = 12V unless otherwise noted. PARAMETER CONDITIONS MIN TYP MAX 1.078 1.1 1.122 5 10 15 mV 380 400 450 mV 1.1 2.0 kΩ SHORTLED Pin Output Impedance 1.1 2.0 kΩ SS Latch-Off Threshold 1.75 V SS Reset Threshold 0.2 V Open LED Falling Threshold (VFB) Open LED Falling Threshold (V(ISP-ISN)) l VFB = 1.2V Short LED Falling Threshold (VFB) OPENLED Pin Output Impedance UNITS V Oscillator Switching Frequency RT = 147k RT = 59.0k RT = 29.1k SYNC Frequency 190 380 665 200 400 700 200 SYNC Pin Resistance to GND 210 420 735 kHz kHz kHz 700 kHz 90 SYNC Threshold Voltage 0.3 kΩ 1.5 V Internal VCC Regulator INTVCC Regulation Voltage Dropout (VIN – INTVCC) 4.8 IINTVCC = –10mA, VIN = 5V INTVCC Undervoltage Lockout INTVCC Current Limit 3.1 VINTVCC = 4V 5 5.2 V 240 350 mV 3.5 3.9 67 V mA PWM PWM Threshold Voltage 0.3 1.5 V PWM Pin Resistance to GND 90 PWMOUT Pull-Up Resistance 10 20 kΩ Ω PWMOUT Pull-Down Resistance 5 10 Ω NMOS Drivers TG1, TG2 Gate Driver On-Resistance Gate Pull-Up Gate Pull-Down VBST – VSW = 5V BG1, BG2 Gate Driver On-Resistance Gate Pull-Up Gate Pull-Down VINTVCC = 5V TG Off to BG On Delay BG Off to TG On Delay TG1, TG2, tOFF(MIN) 2.6 1.7 Ω Ω 3 1.2 Ω Ω CL = 3300pF 60 ns CL = 3300pF 60 RT = 59.0k 220 Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Note 2: The LT3791E is guaranteed to meet performance from 0°C to 125°C junction temperature. Specification over the -40°C to 125°C operating junction temperature range are assured by design, characterization and correlation with statistical process controls. The LT3791I is guaranteed to meet performance specifications over the –40°C to 125°C operating junction temperature range. The LT3791H is guaranteed to meet performance specifications over the –40°C to 150°C ns 260 ns operating junction temperature range. The LT3791MP is guaranteed to meet performance specifications over the –55°C to 150°C operating junction temperature range. High junction temperatures degrade operating lifetimes. Operating lifetime is derated for junction temperatures greater than 125°C. Note 3: The LT3791 includes overtemperature protection that is intended to protect the device during momentary overload conditions. Junction temperature will exceed the maximum operating junction temperature when overtemperature protection is active. Continuous operation above the specified absolute maximum operating junction temperature may impair device reliability. 3791f 4 LT3791 TYPICAL PERFORMANCE CHARACTERISTICS INTVCC Dropout Voltage vs Current, Temperature VIN-VINTVCC (V) 90 5.20 TA = 150°C TA = 25°C TA = –50°C 2.0 INTVCC Current Limit vs Temperature INTVCC Voltage vs Temperature 80 5.15 INTVCC CURRENT LIMIT (mA) 2.5 TA = 25°C, unless otherwise noted. 5.10 INTVCC (V) 1.5 1.0 5.05 VIN = 60V 5.00 VIN = 12V 4.95 4.90 0.5 4.85 0 20 10 0 30 0 LDO CURRENT (mA) 50 40 30 20 0 –50 –25 25 50 75 100 125 150 TEMPERATURE (°C) 3791 G01 VREF Voltage vs Temperature VREF Load Regulation 5.75 2.03 2.15 5.50 2.02 2.10 5.25 2.01 2.05 2.00 4.75 1.99 4.50 1.98 4.25 VREF (V) 2.20 VREF (V) 2.04 4.00 10 20 30 40 50 60 1.90 1.96 –50 –25 70 ILOAD (mA) VIN = 60V VIN = 12V VIN = 4.7V 0 V(ISP-ISN) Threshold vs VCTRL 60 50 40 30 20 10 0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0 VCTRL (V) 3791 G07 50 100 150 200 250 300 350 400 IREF (µA) V(ISP-ISN) Threshold vs Temperature 108 108 106 106 104 104 V(ISP-ISN) (mV) V(ISP-ISN) (mV) V(ISP-ISN) (mV) 80 70 0 3791 G06 V(ISP-ISN) Threshold vs VISP 90 0 1.80 3791 G05 100 0 1.85 25 50 75 100 125 150 TEMPERATURE (°C) 3791 G04 110 2.00 1.95 1.97 0 25 50 75 100 125 150 TEMPERATURE (°C) 3791 G03 6.00 5.00 0 3791 G02 INTVCC Load Regulation INTVCC (V) 60 10 4.80 –50 –25 40 70 102 100 98 102 100 98 96 96 94 94 92 0 10 20 40 30 VISP (V) 50 60 3791 G08 VIN = 12V 92 –50 –25 VISP = 60V VISP = 12V VISP = 0V 0 25 50 75 100 125 150 TEMPERATURE (°C) 3791 G09 3791f 5 LT3791 TYPICAL PERFORMANCE CHARACTERISTICS 120 V(ISP-ISN) Threshold vs VFB TA = 25°C, unless otherwise noted. ISMON Voltage vs Temperature 1.04 VIN = 12V V(ISP-ISN) = 100mV 1.03 100 ISMON Voltage vs V(ISP-ISN) 1.0 0.9 0.8 60 40 0.7 1.01 VISMON (V) 80 VISMON (V) V(ISP-ISN) (mV) 1.02 1.00 0.99 1.19 1.18 1.20 1.21 VFB (V) 1.22 0.1 0.96 –50 –25 1.23 0 V(IVINP-IVINN) Threshold vs VIVINP IVINMON Voltage vs Temperature 54 51.5 1.03 51.0 1.02 50.5 1.01 50 VIVINP = 3V 48 46 44 42 –50 –25 0 50.0 49.5 48.5 0.97 10 20 30 40 50 50 40 SHORTLED Threshold vs Temperature 1.24 0.500 1.23 0.475 1.22 0.450 FB VOLTAGE (V) 60 VFB (V) 1.21 1.20 1.19 20 VIN = 60V VIN = 12V VIN = 4.7V 1.17 1.19 1.20 1.21 VFB (V) 1.22 1.23 3791 G16 1.16 –50 –25 RISING 0.425 0.400 FALLING 0.375 0.350 1.18 10 25 50 75 100 125 150 TEMPERATURE (°C) 3791 G15 FB Regulation Voltage vs Temperature 30 0 3791 G14 V(IVINP-IVINN) Threshold vs VFB 1.18 0.96 –50 –25 60 VIVINP (V) 3791 G13 0 1.17 0.99 0.98 0 VIVINP = 12V V(IVINP-VINN) = 50mV 1.00 49.0 48.0 25 50 75 100 125 150 TEMPERATURE (°C) VIVINMON (V) 1.04 V(IVINP-IVINN) (mV) 52.0 VIVINP = 60V 10 20 30 40 50 60 70 80 90 100 V(ISP-ISN) (mV) 3791 G12 56 52 0 3791 G11 V(IVINP-IVINN) Threshold vs Temperature V(IVINP-IVINN) (mV) 0 25 50 75 100 125 150 TEMPERATURE (°C) 3791 G10 V(IVINP-IVINSN) (mV) 0.4 0.2 0.97 0 1.17 0.5 0.3 0.98 20 0.6 0 25 50 75 100 125 150 TEMPERATURE (°C) 3791 G17 0.325 0.300 –50 –25 0 25 50 75 100 125 150 TEMPERATURE (°C) 3791 G18 3791f 6 LT3791 TYPICAL PERFORMANCE CHARACTERISTICS OPENLED Threshold vs Temperature OVLO Threshold vs Temperature 1.125 FALLING 1.100 1.075 1.050 1.025 1.000 –50 –25 0 16 3.2 14 3.1 12 RISING 3.0 2.9 FALLING 2.7 4 2.6 2 0 Supply Current vs Input Voltage 7 IQ (mA) 1.5 1.0 TA = 150°C TA = 25°C TA = –50°C 0 0 10 20 30 VIN (V) 40 50 60 VEN/ULO = 1V 1.28 6 5 4 3 2 TG1, TG2 MINIMUM ON-TIME (ns) SWITCHING FREQUENCY (kHz) 600 RT = 59.0k RT = 147k 200 100 0 –50 –25 0 25 50 75 100 125 150 TEMPERATURE (°C) 3791 G25 FALLING 1.16 1.14 1.12 1.10 –50 –25 0 100 300 1.18 25 50 75 100 125 150 TEMPERATURE (°C) 0 25 50 75 100 125 150 TEMPERATURE (°C) 3791 G24 TG1, TG2 Minimum On-Time vs Temperature RT = 29.1k RISING 1.20 3791 G23 800 400 1.22 0 –50 –25 Oscillator Frequency vs Temperature 500 1.26 1.24 1 3791 G22 700 EN/UVLO Threshold Voltage 1.30 EN/UVLO THRESHOLD (V) EN/UVLO PIN CURRENT (µA) 3.5 2.0 25 50 75 100 125 150 TEMPERATURE (°C) 3791 G21 EN/UVLO Pin Current 8 2.5 0 3791 G20 4.0 3.0 DISCHARGING 0 –50 –25 25 50 75 100 125 150 TEMPERATURE (°C) 3791 G19 0.5 8 6 2.5 –50 –25 25 50 75 100 125 150 TEMPERATURE (°C) CHARGING 10 2.8 TG1, TG2 Minimum Off-Time vs Temperature 350 TG2 90 TG1, TG2 MINIMUM OFF-TIME (ns) FB VOLTAGE (V) 1.150 OVLO THRESHOLD (V) RISING Soft-Start Current vs Temperature 3.3 ISS (µA) 1.200 1.175 TA = 25°C, unless otherwise noted. TG1 80 70 60 50 40 30 20 –50 –25 0 25 50 75 100 125 150 TEMPERATURE (°C) 3791 G26 300 fSW = 200kHz 250 fSW = 400kHz 200 fSW = 700kHz 150 100 50 0 –50 –25 0 25 50 75 100 125 150 TEMPERATURE (˚C) 3791 G27 3791f 7 LT3791 TYPICAL PERFORMANCE CHARACTERISTICS BG1, BG2 Driver On-Resistance vs Temperature 4.5 3.8 4.0 RISING 3.7 3.6 3.5 3.4 FALLING 3.3 3.2 4.0 3.5 PULL-UP 3.5 3.0 2.5 2.0 PULL-DOWN 1.5 1.0 0 0 –50 –25 25 50 75 100 125 150 TEMPERATURE (°C) 0 1.0 14 1.6 12 1.4 25 50 75 100 125 150 TEMPERATURE (°C) V(SNSP-SNSN) Buck Threshold vs VC PULL-UP 40 BG2 1.0 VC (V) 60 V(SNSP-SNSN) = 0V 1.2 PULL-DOWN 0 3791 G30 V(SNSP-SNSN) (mV) PWMOUT RESISTANCE (Ω) 1.5 VC Voltage vs Duty Cycle 8 PULL-DOWN 2.0 3791 G29 PWMOUT On-Resistance vs Temperature 6 2.5 0 –50 –25 25 50 75 100 125 150 TEMPERATURE (°C) 3791 G28 10 PULL-UP 3.0 0.5 0.5 3.1 –50 –25 BG1 0.8 0.6 4 20 0 –20 0.4 2 –40 0.2 0 –50 –25 0 25 50 0 75 100 125 150 TEMPERATURE (˚C) 0 20 40 60 DUTY CYCLE (%) 80 3791 G31 V(SNSP-SNSN) (mV) 20 0 –20 VC(MAX) –60 –50 –25 0 3791 G34 1.2 VC (V) 1.4 1.6 60 60 40 40 20 0 –20 –40 –80 0.6 0.8 1.0 1.2 VC (V) 1.4 1.6 1.8 V(SNSP-SNSN) Boost Threshold vs Temperature –60 25 50 75 100 125 150 TEMPERATURE (°C) 1.0 0.8 3791 G33 V(SNSP-SNSN) THRESHOLD (mV) VC(MIN) –40 0.6 V(SNSP-SNSN) Boost Threshold vs VC 60 40 –60 100 3791 G32 V(SNSP-SNSN) Buck Threshold vs Temperature V(SNSP-SNSN) THRESHOLD (mV) TG1, TG2 Driver On-Resistance vs Temperature TG1, TG2 RESISTANCE (Ω) 3.9 BG1, BG2 RESISTANCE (Ω) V(BST1-SW1), V(BST2,SW2) (V) V(BST1-SW1), V(BST2-SW2) UVLO vs Temperature TA = 25°C, unless otherwise noted. 1.8 3791 G35 VC(MAX) 20 0 –20 –40 –60 –80 –50 –25 VC(MIN) 0 25 50 75 100 125 150 TEMPERATURE (˚C) 3791 G36 3791f 8 LT3791 PIN FUNCTIONS CTRL (Pin 1): Current Sense Threshold Adjustment Pin for Analog Dimming. Regulating threshold V(ISP-ISN) is 1/10th of (VCTRL – 200mV). CTRL linear range is from 200mV to 1.1V. For VCTRL > 1.3V, the current sense threshold is constant at the full-scale value of 100mV. For 1.1V < VCTRL < 1.3V, the dependence of the current sense threshold upon VCTRL transitions from a linear function to a constant value, reaching 98% of full scale by VCTRL = 1.2V. Connect CTRL to VREF for the 100mV default threshold. Force less than 175mV (typical) to stop switching. Do not leave this pin open. SS (Pin 2): Soft-start reduces the input power sources surge current by gradually increasing the controller’s current limit. A minimum value of 10nF is recommended on this pin. SS is used as a timer when an open or shorted LED condition occurs. A 500k resistor placed from SS to VREF will latch the part off in the event of a fault. A 100k resistor to VREF will allow the part to keep running in a fault. If left open, a 1.4µA current source pulls down on SS and the part restarts in a fault. PWM (Pin 3): A signal low turns off switches, idles switching and disconnects the VC pin from all external loads. The PWMOUT pin follows the PWM pin. PWM has an internal 100k pull-down resistor. If not used, connect to INTVCC. OPENLED (Pin 4): An open-drain pull-down on OPENLED asserts if FB is greater than 1.15V (typical) and V(ISP-ISN) is less than 10mV (typical). To function, the pin requires an external pull-up resistor. SHORTLED (Pin 5): An open-drain pull-down on SHORTLED asserts if FB is less than 400mV (typical). To function, the pin requires an external pull-up resistor. VREF (Pin 6): Voltage Reference Output Pin, Typically 2V. This pin drives a resistor divider for the CTRL pin, either for analog dimming or for temperature limit/compensation of the LED load. Can supply up to 200µA of current. ISMON (Pin 7): Monitor pin that produces a voltage that is ten times the voltage V(ISP-ISN). ISMON will equal 1V when V(ISP-ISN) = 100mV. IVINMON (Pin 8): Monitor pin that produces a voltage that is twenty times the voltage V(IVINP-IVINN). IVINMON will equal 1V when V(IVINP-IVINN) = 50mV. EN/UVLO (Pin 9): Enable Control Pin. Forcing an accurate 1.2V falling threshold with an externally programmable hysteresis is generated by the external resistor divider and a 3µA pull-down current. Above the 1.2V (typical) threshold (but below 6V), EN/UVLO input bias current is sub-µA. Below the falling threshold, a 3µA pull-down current is enabled so the user can define the hysteresis with the external resistor selection. An undervoltage condition resets soft-start. Tie to 0.3V, or less, to disable the device and reduce VIN quiescent current below 1µA. IVINP (Pin 10): Positive Input for the Input Current Limit and Monitor. Input bias current for this pin is typically 90µA. IVINN (Pin 11): Negative Input for the Input Current Limit and Monitor. The input bias current for this pin is typically 20µA. VIN (Pin 12): Main Input Supply. Bypass this pin to PGND with a capacitor. INTVCC (Pin 13): Internal 5V Regulator Output. The driver and control circuits are powered from this voltage. Bypass this pin to PGND with a minimum 4.7µF ceramic capacitor. TG1 (Pin 14): Top Gate Drive. Drives the top N-channel MOSFET with a voltage equal to INTVCC superimposed on the switch node voltage SW1. BST1 (Pin 15): Bootstrapped Driver Supply. The BST1 pin swings from a diode voltage below INTVCC up to a diode voltage below VIN + INTVCC. SW1 (Pin 16): Switch Node. SW1 pin swings from a diode voltage drop below ground up to VIN. PGND (Pins 17, 20): Power Ground. Connect these pins closely to the source of the bottom N-channel MOSFET. BG1 (Pin 18): Bottom Gate Drive. Drives the gate of the bottom N-channel MOSFET between ground and INTVCC. BG2 (Pin 19): Bottom Gate Drive. Drives the gate of the bottom N-channel MOSFET between ground and INTVCC. SW2 (Pin 21): Switch Node. SW2 pin swings from a diode voltage drop below ground up to VOUT. BST2 (Pin 22): Bootstrapped Driver Supply. The BST2 pin swings from a diode voltage below INTVCC up to a diode voltage below VOUT + INTVCC. 3791f 9 LT3791 PIN FUNCTIONS NC (Pin 23): No Connect Pin. Leave this pin floating. TG2 (Pin 24): Top Gate Drive. Drives the top N-channel MOSFET with a voltage equal to INTVCC superimposed on the switch node voltage SW2. ISP (Pin 25): Connection Point for the Positive Terminal of the Output Current Feedback Resistor. ISN (Pin 26): Connection Point for the Negative Terminal of the Output Current Feedback Resistor. SNSP (Pin 27): The Positive Input to the Current Sense Comparator. The VC pin voltage and controlled offsets between the SNSP and SNSN pins, in conjunction with a resistor, set the current trip threshold. SNSN (Pin 28): The Negative Input to the Current Sense Comparator. TEST1 (Pin 29): This pin is used for testing purposes only and must be connected to SGND for the part to operate properly. SGND (Pin 30, Exposed Pad Pin 39): Signal Ground. All small-signal components and compensation should connect to this ground, which should be connected to PGND at a single point. Solder the exposed pad directly to the ground plane. PWMOUT (Pin 31): Buffered Version of PWM Signal for Driving LED Load Disconnect N-Channel MOSFET. The PWMOUT pin is driven from INTVCC. Use of a MOSFET with a gate cutoff voltage higher than 1V is recommended. TEST2 (Pin 32): This pin is used for testing purposes only and must be connected to INTVCC (Pin 13) for the part to operate properly. CLKOUT (Pin 33): Clock Output Pin. An in-phase clock is provided at the oscillator frequency to allow for synchronizing two devices for extending output power capability. SYNC (Pin 34): External Synchronization Input Pin. This pin is internally terminated to GND with a 90k resistor. The rising edge will be synchronized with the rising edge of the SYNC signal. RT (Pin 35): Frequency Set Pin. Place a resistor to GND to set the internal frequency. The range of oscillation is 200kHz to 700kHz. VC (Pin 36): Current Control Threshold and Error Amplifier Compensation Point. The current comparator threshold increases with this control voltage. The voltage ranges from 0.7V to 1.9V. FB (Pin 37): Voltage Loop Feedback Pin. FB is intended for constant-voltage regulation or for LED protection of an open or shorted LED. The internal transconductance amplifier with output VC will regulate FB to 1.2V (typical) through the DC/DC converter. If the FB input is regulating the loop and V(ISP-ISN) < 10mV, the OPENLED pull-down is asserted. If the FB pin is less than 400mV, the SHORTLED pull-down is asserted. OVLO (Pin 38): Overvoltage Input Pin. This pin is used for OVLO, if OVLO > 3V then SS is pulled low, the part stops switching and resets. Do not leave this pin open. 3791f 10 LT3791 BLOCK DIAGRAM 26 – + A = 10 7 8 9 11 IVINP + A2 10 ISN ISP A1 A = 10 A = 20 12 6 13 VREF VIN IVINN INTVCC – 25 A = 24 REGS SHDN_INT ISMON ISMON_INT TSD IVINMON_INT BST1 + IVINMON A13 A3 EN/UVLO A4 1.2V SW1 – – 3µA TG1 SHDN_INT SHDN_INT SS_RESET SS LATCH PWM + 35 34 33 A14 BG1 A15 17 19 BOOST LOGIC SYNC CLKOUT SW2 + A16 TG2 BST2 – SHORTLED + SNSP – SNSN A10 + 0.4V – A11 FB + OPENLED + Q R S VREF 0.2V A8 A6 – 1.15V + SS LATCH IVINMON_INT FB 21 24 22 27 28 37 1.2V A12 SS RESET – – + 14µA – + + – ISMON_INT + 3V CTRL 1 A9 – 1.75V INTVCC PWM 18 INTVCC RT A5 3 BG2 SLOPE_COMP_BOOST SLOPE_COMP_BUCK 4 16 INTVCC A7 5 14 BUCK LOGIC PGND OSC 15 A18 A17 PWMOUT 31 SGND 30, 39 1.4µA – 38 VC SS 2 OVLO 36 3791 BD 3791f 11 LT3791 OPERATION The LT3791 is a current mode controller that provides an output voltage above, equal to or below the input voltage. The LTC proprietary topology and control architecture uses a current sensing resistor in buck or boost operation. The sensed inductor current is controlled by the voltage on the VC pin, which is the output of the feedback amplifiers A11 and A12. The VC pin is controlled by three inputs, one input from the output current loop, one input from the input current loop, and the third input from the feedback loop. Whichever feedback input is higher takes precedence, forcing the converter into either a constant-current or a constant-voltage mode. The LT3791 is designed to transition cleanly between the two modes of operation. Current sense amplifier A1 senses the voltage between the IVINP and IVINN pins and provides a pre-gain to amplifier A11. When the voltage between IVINP and IVINN reaches 50mV, the output of A1 provides IVINMON_INT to the inverting input of A11 and the converter is in constant-current mode. If the current sense voltage exceeds 50mV, the output of A1 increases causing the output of A11 to decrease, thus reducing the amount of current delivered to the output. In this manner the current sense voltage is regulated to 50mV. The output current amplifier works similar to the input current amplifier but with a 100mV voltage instead of 50mV. The output current sense level is also adjustable by the CTRL pin. Forcing CTRL to less than 1.2V forces ISMON_INT to the same level as CTRL, thus providing current-level control. The output current amplifier provides rail-to-rail operation. Similarly if the FB pin goes above 1.2V the output of A11 decreases to reduce the current level and regulate the output (constant-voltage mode). The LT3791 provides monitoring pins IVINMON and ISMON that are proportional to the voltage across the input and output current amplifiers respectively. The main control loop is shut down by pulling the EN/ UVLO pin low. When the EN/UVLO pin is higher than 1.2V, an internal 14µA current source charges soft-start capacitor CSS at the SS pin. The VC voltage is then clamped a diode voltage higher than the SS voltage while the CSS is slowly charged during start-up. This “soft-start” clamping prevents abrupt current from being drawn from the input power supply. The SS can also be used as a fault timer whenever an open or shorted LED is detected. The top MOSFET drivers are biased from floating bootstrap capacitors C1 and C2, which are normally recharged through an external diode when the top MOSFET is turned off. Schottky diodes across the synchronous switch M4 and synchronous switch M2 are not required, but they do provide a lower drop during the dead time. The addition of the Schottky diode typically improves peak efficiency by 1% to 2% at 500kHz. Power Switch Control Figure 1 shows a simplified diagram of how the four power switches are connected to the inductor, VIN, VOUT and GND. Figure 2 shows the regions of operation for the LT3791 as a function of duty cycle D. The power switches are properly controlled so the transfer between regions is continuous. When VIN approaches VOUT, the buck-boost region is reached. VOUT VIN TG1 M1 L1 SW1 BG1 TG2 M4 M2 SW2 M3 BG2 RSENSE 3791 F01 Figure 1. Simplified Diagram of the Output Switches DMAX BOOST (BG2) DMIN BOOST DMAX BUCK (TG1) DMIN BUCK BOOST REGION BUCK-BOOST REGION BUCK REGION M1 ON, M2 OFF PWM M3, M4 SWITCHES 4-SWITCH PWM M4 ON, M3 OFF PWM M2, M1 SWITCHES 3791 F02 Figure 2. Operating Regions vs Duty Cycle 3791f 12 LT3791 OPERATION Buck Region (VIN > VOUT) where D(BUCK-BOOST) is the duty cycle of the buck-boost switch range: Switch M4 is always on and switch M3 is always off during this mode. At the start of every cycle, synchronous switch M2 is turned on first. Inductor current is sensed when synchronous switch M2 is turned on. After the sensed inductor current falls below the reference voltage, which is proportional to VC, synchronous switch M2 is turned off and switch M1 is turned on for the remainder of the cycle. Switches M1 and M2 will alternate, behaving like a typical synchronous buck regulator. The duty cycle of switch M1 increases until the maximum duty cycle of the converter in buck operation reaches DMAX(BUCK, TG1), given by: D(BUCK-BOOST) = 8% Figure 3 shows typical buck operation waveforms. If VIN approaches VOUT, the buck-boost region is reached. Buck-Boost (VIN ~ VOUT) When VIN is close to VOUT, the controller is in buck-boost operation. Figure 4 and Figure 5 show typical waveforms in this mode. Every cycle the controller turns on switches M2 and M4, then M1 and M4 are turned on until 180° later when switches M1 and M3 turn on, and then switches M1 and M4 are turned on for the remainder of the cycle. DMAX(BUCK,TG1) = 100% – D(BUCK-BOOST) M2 + M4 M2 + M4 M1 + M4 M2 + M4 M1 + M4 M1 + M4 3791 F03 Figure 3. Buck Operation (VIN > VOUT) M1 + M4 M1+ M3 M1 + M4 M2 + M4 M1+ M3 M1 + M4 M1 + M4 M2 + M4 M1+ M3 M1 + M4 M2 + M4 M1 + M4 3791 F04 Figure 4. Buck-Boost Operation (VIN ≤ VOUT) M1 + M4 M2 + M4 M1 + M3 M1 + M4 M1 + M4 M2 + M4 M1 + M3 M1 + M4 M1 + M4 M2 + M4 M1 + M3 M1 + M4 3791 F05 Figure 5. Buck-Boost Operation (VIN ≥ VOUT) 3791f 13 LT3791 OPERATION Boost Region (VIN < VOUT) Low Current Operation Switch M1 is always on and synchronous switch M2 is always off in boost operation. Every cycle switch M3 is turned on first. Inductor current is sensed when synchronous switch M3 is turned on. After the sensed inductor current exceeds the reference voltage which is proportional to VC, switch M3 turns off and synchronous switch M4 is turned on for the remainder of the cycle. Switches M3 and M4 alternate, behaving like a typical synchronous boost regulator. The LT3791 runs in forced continuous mode. In this mode the controller behaves as a continuous, PWM current mode synchronous switching regulator. In boost operation, switch M1 is always on, switch M3 and synchronous switch M4 are alternately turned on to maintain the output voltage independent of the direction of inductor current. In buck operation, synchronous switch M4 is always on, switch M1 and synchronous switch M2 are alternately turned on to maintain the output voltage independent of the direction of inductor current. In this mode, the output can source or sink current. The duty cycle of switch M3 decreases until the minimum duty cycle of the converter in boost operation reaches DMIN(BOOST,BG2), given by: DMIN(BOOST,BG2) = D(BUCK-BOOST) where D(BUCK-BOOST) is the duty cycle of the buck-boost switch range: D(BUCK-BOOST) = 8% Figure 6 shows typical boost operation waveforms. If VIN approaches VOUT, the buck-boost region is reached. M1 + M3 M1 + M4 M1 + M3 M1 + M4 M1 + M3 M1 + M4 3791 F06 Figure 6. Boost Operation (VIN < VOUT) 3791f 14 LT3791 APPLICATIONS INFORMATION The Typical Application on the front page is a basic LT3791 application circuit. External component selection is driven by the load requirement, and begins with the selection of RSENSE and the inductor value. Next, the power MOSFETs are selected. Finally, CIN and COUT are selected. This circuit can operate up to an input voltage of 60V. Programming The Switching Frequency The RT frequency adjust pin allows the user to program the switching frequency from 200kHz to 700kHz to optimize efficiency/performance or external component size. Higher frequency operation yields smaller component size but increases switching losses and gate driving current, and may not allow sufficiently high or low duty cycle operation. Lower frequency operation gives better performance at the cost of larger external component size. For an appropriate RT resistor value see Table 1. An external resistor from the RT pin to GND is required; do not leave this pin open. The rising edge of CLK_OUT corresponds to the rising edge of SYNC thus allowing paralleling converters. The falling edge of CLK_OUT turns on switch M3 and the rising edge of CLK_OUT turns on switch M2. Inductor Selection The operating frequency and inductor selection are interrelated in that higher operating frequencies allow the use of smaller inductor and capacitor values. The inductor value has a direct effect on ripple current. The maximum inductor current ripple ΔIL can be seen in Figure 7. This is the maximum ripple that will prevent subharmonic oscillation and also regulate with zero load. The ripple should be less than this to allow proper operation over all load currents. For a given ripple the inductance terms in continuous mode are as follows: LBUCK > Table 1. Switching Frequency vs RT Value fOSC (kHz) RT (kΩ) 200 147 300 84.5 400 59.0 500 45.3 600 35.7 700 29.4 Frequency Synchronization The LT3791 switching frequency can be synchronized to an external clock using the SYNC pin. Driving SYNC with a 50% duty cycle waveform is always a good choice, otherwise maintain the duty cycle between 10% and 90%. ( ) VOUT • VIN(MAX) – VOUT • 100 LBOOST > f •ILED • %Ripple • VIN(MAX) ( ) VIN(MIN)2 • VOUT – VIN(MIN) • 100 f •ILED • %Ripple • VOUT 2 where: f is operating frequency % ripple is allowable inductor current ripple VIN(MIN) is minimum input voltage VIN(MAX) is maximum input voltage VOUT is output voltage ILED is current through the LEDs 3791f 15 LT3791 APPLICATIONS INFORMATION where ΔIL is peak-to-peak inductor ripple current. In buck operation, the maximum average load current is: 200 180 ∆IL/ISENSE(MAX) (%) 160 140 120 100 BOOST ∆IL/ ISENSE(MAX) LIMIT The maximum current sensing RSENSE value for the boost operation is: 80 60 40 BUCK ∆IL/ ISENSE(MAX) LIMIT 20 0 47.5mV ∆IL IOUT(MAX _ BUCK) = + 2 RSENSE 50 55 60 65 70 75 80 85 90 95 100 BG1, BG2 DUTY CYCLE (%) 3791 F07 Figure 7. Maximum Peak-to-Peak Ripple vs Duty Cycle For high efficiency, choose an inductor with low core loss. Also, the inductor should have low DC resistance to reduce the I2R losses, and must be able to handle the peak inductor current without saturating. To minimize radiated noise, use a shielded inductor. RSENSE(MAX) = 2 • 51mV• VIN(MIN) 2 •ILED • VOUT + ∆IL(BOOST) • VIN(MIN) The maximum current sensing RSENSE value for the buck operation is: RSENSE(MAX) = 2 • 47.5mV 2 •ILED – ∆IL(BUCK) The final RSENSE value should be lower than the calculated RSENSE(MAX) in both the boost and buck operation. A 20% to 30% margin is usually recommended. RSENSE Selection and Maximum Output Current CIN and COUT Selection RSENSE is chosen based on the required output current. The current comparator threshold sets the peak of the inductor current in boost operation and the maximum inductor valley current in buck operation. In boost operation, the maximum average load current at VIN(MIN) is: In boost operation, input current is continuous. In buck operation, input current is discontinuous. In buck operation, the selection of input capacitor, CIN, is driven by the need to filter the input square wave current. Use a low ESR capacitor sized to handle the maximum RMS current. For buck operation, the input RMS current is given by: 51mV ∆IL VIN(MIN) IOUT(MAX _ BOOST) = – • RSENSE 2 VOUT IRMS = ILED2 •D+ ∆IL 2 •D 12 3791f 16 LT3791 APPLICATIONS INFORMATION The formula has a maximum at VIN = 2VOUT. Note that ripple current ratings from capacitor manufacturers are often based on only 2000 hours of life which makes it advisable to derate the capacitor. In boost operation, the discontinuous current shifts from the input to the output, so COUT must be capable of reducing the output voltage ripple. The effects of ESR (equivalent series resistance) and the bulk capacitance must be considered when choosing the right capacitor for a given output ripple voltage. The steady ripple due to charging and discharging the bulk capacitance is given by: ∆VRIPPLE (BOOST _ CAP ) = ( ILED • VOUT – VIN(MIN) ) C OUT • VOUT • f ∆IL ∆VRIPPLE (BUCK _ CAP ) ≈ 8 • f •C Programming VIN UVLO and OVLO The falling UVLO value can be accurately set by the resistor divider R1 and R2. A small 3µA pull-down current is active when the EN/UVLO is below the threshold. The purpose of this current is to allow the user to program the rising hysteresis. The following equations should be used to determine the resistor values: VIN(UVLO –) = 1.2 • VIN(OVLO + ) = 3 • where COUT is the output filter capacitor. ΔVBOOST(ESR) = ILED • ESR VIN(UVLO + ) = 3µA •R1+1.215 • R3+R4 R4 VIN(OVLO –) = 2.925 • R3+R4 R4 VIN ΔVBUCK(ESR) = ILED • ESR Multiple capacitors placed in parallel may be needed to meet the ESR and RMS current handling requirements. Output capacitors are also used for stability for the LT3791. A good starting point for output capacitors is seen in the Typical Applications circuits. Ceramic capacitors have excellent low ESR characteristics but can have a high voltage coefficient and are recommended for applications less than 100W. Capacitors available with low ESR and high ripple current ratings, such as OS-CON and POSCAP may be needed for applications greater than 100W. R1+R2 R2 The rising OVLO value can be accurately set by the resistor divider R3 and R4. The following equations should be used to determine the resistor values: OUT The steady ripple due to the voltage drop across the ESR is given by: R1+R2 R2 LT3791 R1 R3 R2 R4 OVLO EN/UVLO 3791 F08 Figure 8. Resistor Connection to Set VIN UVLO and OVLO Thresholds 3791f 17 LT3791 APPLICATIONS INFORMATION Programming LED Current The LED current is programmed by placing an appropriate value current sense resistor, RLED, in series with the LED string. The voltage drop across RLED is (Kelvin) sensed by the ISP and ISN pins. The CTRL pin should be tied to a voltage higher than 1.2V to get the full-scale 100mV (typical) threshold across the sense resistor. The CTRL pin can also be used to dim the LED current, although relative accuracy decreases with the decreasing sense threshold. When the CTRL pin voltage is less than 1V, the LED current is: ILED = VCTRL – 200mV RLED • 10 When the CTRL pin voltage is between 1.1V and 1.3V the LED current varies with VCTRL, but departs from the equation above by an increasing amount as VCTRL voltage increases. Ultimately, when VCTRL > 1.3V the LED current no longer varies. The typical V(ISP-ISN) threshold vs VCTRL is listed in Table 2. Table 2. V(ISP-ISN) Threshold vs CTRL VCTRL (V) V(ISP-ISN) (mV) 1.1 90 1.15 94.5 1.2 98 1.25 99.5 1.3 100 When VCTRL is higher than 1.3V, the LED current is regulated to: ILED = 100mV RLED The CTRL pin should not be left open (tie to VREF if not used). The CTRL pin can also be used in conjunction with a thermistor to provide overtemperature protection for the LED load, or with a resistor divider to VIN to reduce output power and switching current when VIN is low. The presence of a time varying differential voltage signal (ripple) across ISP and ISN at the switching frequency is expected. The amplitude of this signal is increased by high LED load current, low switching frequency and/or a smaller value output filter capacitor. Some level of ripple signal is acceptable: the compensation capacitor on the VC pin filters the signal so the average difference between ISP and ISN is regulated to the user-programmed value. Ripple voltage amplitude (peak-to-peak) in excess of 20mV should not cause mis-operation, but may lead to noticeable offset between the average value and the userprogrammed value. ISMON The ISMON pin provides a linear indication of the current flowing through the LEDs. The equation for VISMON is V(ISP–ISN) • 10. This pin is suitable for driving an ADC input, however, the output impedance of this pin is 12.5kΩ so care must be taken not to load this pin. Programming Input Current Limit The LT3791 has a standalone current sense amplifier. It can be used to limit the input current. The input current limit is calculated by the following equation: IIN = 50mV RIN 3791f 18 LT3791 APPLICATIONS INFORMATION For loop stability a lowpass RC filter is needed. For most applications, a 50Ω resistor and 470nF capacitor is sufficient. VOUT FB Table 3 R6 RIN (mΩ) 20 15 12 10 6 5 4 3 2 ILIMIT (A) 2.5 3.3 4.2 5.0 8.3 10.0 12.5 16.7 25 IVINMON The IVINMON pin provides a linear indication of the current flowing through the input. The equation for VIVINMON is V(IVINP-IVINN) • 20. This pin is suitable for driving an ADC input, however, the output impedance of this pin is 12.5kΩ so care must be taken not to load this pin. Programming Output Voltage (Constant Voltage Regulation) or Open LED/Overvoltage Threshold For a voltage regulator, the output voltage can be set by selecting the values of R5 and R6 (see Figure 9) according to the following equation: R5 LT3791 VOUT = 1.2 • R5+R6 R6 For an LED driver application, set the resistor from the output to the FB pin such that the expected VFB during normal operation does not exceed 1.1V. Once VFB is higher than its overvoltage threshold, 1.25V (typical), the LT3791 stops switching. 3791 F09 Figure 9. Resistor Connection for Open LED Threshold and Constant Output Voltage Regulation Dimming Control There are two methods to control the current source for dimming using the LT3791. One method uses the CTRL pin to adjust the current regulated in the LEDs. A second method uses the PWM pin to modulate the current source between zero and full current to achieve a precisely programmed average current. To make PWM dimming more accurate, the switch demand current is stored on the VC node during the quiescent phase when PWM is low. This feature minimizes recovery time when the PWM signal goes high. To further improve the recovery time a disconnect switch may be used in the LED current path to prevent the ISP node from discharging during the PWM signal low phase. The minimum PWM on- or off-time is affected by choice of operating frequency and external component selection. The best overall combination of PWM and analog dimming capabilities is available if the minimum PWM pulse is at least six switching cycles and the PWM pulse is synchronized to the SYNC signal. SHORTLED Pin The LT3791 provides an open-drain status pin, SHORTLED, which pulls low when the FB pin is below 400mV. The only time the FB pin will be below 400mV is during start-up or if the LEDs are shorted. During 3791f 19 LT3791 APPLICATIONS INFORMATION start-up the LT3791 ignores the voltage on the FB pin until the soft-start capacitor reaches 1.75V. To prevent false tripping after startup, a large enough soft-start capacitor must be used to allow the output to get up to approximately 40% to 50% of the final value. OPENLED Pin The LT3791 provides an open-drain status pin, OPENLED, which pulls low when the FB pin is above 1.15V and the voltage across V(ISP-ISN) is less than 10mV. If the open LED clamp voltage is programmed correctly using the FB pin, then the FB pin should never exceed 1.1V when the LEDs are connected. Therefore, the only way for the FB pin to exceed 1.15V is for an open LED event to occur. Soft-Start, Fault Function Soft-start reduces the input power sources’ surge currents by gradually increasing the controller’s current limit (proportional to an internally buffered clamped equivalent of VC). The soft-start interval is set by the soft-start capacitor selection according to the following equation t SS = 1.2V •C 14µA SS Make sure CSS is large enough when there is loading during start-up. The SS pin is also used as a fault timer. Once an open LED or a shorted LED fault is detected, a 1.4µA pull-down current source is activated. With a 100k pull-up resistor to VREF on the SS pin, the LT3791 will continue to switch normally. With a 500k pull-up resistor to VREF on the SS pin, the LT3791 will latch off until the EN/UVLO pin is toggled. Without any resistor to VREF the SS pin enters a hiccup mode operation. The 1.4µA pulls SS down until 0.2V is reached, at which point the 14µA pull-up current source turns on. If the fault condition hasn’t been removed when SS reaches 1.75V, then the 1.4µA pull-down current source turns on again initiating a new cycle. This will continue until the fault is removed. Loop Compensation The LT3791 uses an internal transconductance error amplifier whose VC output compensates the control loop. The external inductor, output capacitor and the compensation resistor and capacitor determine the loop stability. The inductor and output capacitor are chosen based on performance, size and cost. The compensation resistor and capacitor at VC are set to optimize control loop response and stability. For typical LED applications, a 10nF compensation capacitor at VC is adequate, and a series resistor should always be used to increase the slew rate on the VC pin to maintain tighter regulation of LED current during fast transients on the input supply of the converter. 3791f 20 LT3791 APPLICATIONS INFORMATION Power MOSFET Selections and Efficiency Considerations The LT3791 requires four external N-channel power MOSFETs, two for the top switches (switch M1 and M4, shown in Figure 1) and two for the bottom switches (switch M2 and M3 shown in Figure 1). Important parameters for the power MOSFETs are the breakdown voltage, VBR(DSS), threshold voltage, VGS(TH), on-resistance, RDS(ON), reverse transfer capacitance, CRSS, and maximum current, IDS(MAX). The drive voltage is set by the 5V INTVCC supply. Consequently, logic-level threshold MOSFETs must be used in LT3791 applications. If the input voltage is expected to drop below the 5V, then sub-logic threshold MOSFETs should be considered. In order to select the power MOSFETs, the power dissipated by the device must be known. For switch M1, the maximum power dissipation happens in boost operation, when it remains on all the time. Its maximum power dissipation at maximum output current is given by: 2 I •V PM1(BOOST) = LED OUT • ρT •RDS(ON) VIN where ρT is a normalization factor (unity at 25°C) accounting for the significant variation in on-resistance with temperature, typically 0.4%/°C as shown in Figure 10. For a maximum junction temperature of 125°C, using a value of ρT = 1.5 is reasonable. Switch M2 operates in buck operation as the synchronous rectifier. Its power dissipation at maximum output current is given by: PM2(BUCK) = VIN – VOUT •ILED2 • ρT •RDS(ON) VIN Switch M3 operates in boost operation as the control switch. Its power dissipation at maximum current is given by: PM3(BOOST) = ( VOUT – VIN ) • VOUT •I VIN2 + k • VOUT 3 • LED 2 •ρ T •RDS(ON) ILED •C •f VIN ROSS where CRSS is usually specified by the MOSFET manufacturers. The constant k, which accounts for the loss caused by reverse-recovery current, is inversely proportional to the gate drive current and has an empirical value of 1.7. For switch M4, the maximum power dissipation happens in boost operation, when its duty cycle is higher than 50%. Its maximum power dissipation at maximum output current is given by: V PM4(BOOST) = IN VOUT 2 I •V • LED OUT • ρT •RDS(ON) VIN For the same output voltage and current, switch M1 has the highest power dissipation and switch M2 has the lowest power dissipation unless a short occurs at the output. 3791f 21 LT3791 APPLICATIONS INFORMATION From a known power dissipated in the power MOSFET, its junction temperature can be obtained using the following formula: TJ = TA + P • RTH(JA) The RTH(JA) to be used in the equation normally includes the RTH(JC) for the device plus the thermal resistance from the case to the ambient temperature (RTH(JC)). This value of TJ can then be compared to the original, assumed value used in the iterative calculation process. ρT NORMALIZED ON-RESISTANCE (Ω) 2.0 1.5 1.0 0.5 0 –50 50 100 0 JUNCTION TEMPERATURE (°C) 150 3791 F10 Figure 10. Normalized RDS(ON) vs Temperature Optional Schottky Diode (D3, D4) Selection The Schottky diodes D3 and D4 shown in the Typical Applications section conduct during the dead time between the conduction of the power MOSFET switches. They are intended to prevent the body diode of synchronous switches M2 and M4 from turning on and storing charge during the dead time. In particular, D4 significantly reduces reverse-recovery current between switch M4 turn-off and switch M3 turn-on, which improves converter efficiency and reduces switch M3 voltage stress. In order for the diode to be effective, the inductance between it and the synchronous switch must be as small as possible, mandating that these components be placed adjacently. INTVCC Regulator An internal P-channel low dropout regulator produces 5V at the INTVCC pin from the VIN supply pin. INTVCC powers the drivers and internal circuitry within the LT3791. The INTVCC pin regulator can supply a peak current of 67mA and must be bypassed to ground with a minimum of 4.7µF ceramic capacitor or low ESR electrolytic capacitor. An additional 0.1µF ceramic capacitor placed directly adjacent to the INTVCC and PGND IC pins is highly recommended. Good bypassing is necessary to supply the high transient current required by MOSFET gate drivers. Higher input voltage applications in which large MOSFETs are being driven at high frequencies may cause the maximum junction temperature rating for the LT3791 to be exceeded. The system supply current is normally dominated by the gate charge current. Additional external loading of the INTVCC also needs to be taken into account for the power dissipation calculations. Power dissipation for the IC in this case is VIN • IINTVCC, and overall efficiency is lowered. The junction temperature can be estimated by using the equations given TJ = TA + (PD • θJA) where θJA (in °C/W) is the package thermal impedance. For example, a typical application operating in continuous current operation might draw 24mA from a 24V supply: TJ = 70°C + 24mA • 24V • 28°C/W = 86°C 3791f 22 LT3791 APPLICATIONS INFORMATION To prevent maximum junction temperature from being exceeded, the input supply current must be checked operating in continuous mode at maximum VIN. Top Gate (TG) MOSFET Driver Supply (C1, D1, C2, D2) The external bootstrap capacitors C1 and C2 connected to the BST1 and BST2 pins supply the gate drive voltage for the topside MOSFET switches M1 and M4. When the top MOSFET switch M1 turns on, the switch node SW1 rises to VIN and the BST1 pin rises to approximately VIN + INTVCC. When the bottom MOSFET switch M2 turns on, the switch node SW1 drops low and the bootstrap capacitor C1 is charged through D1 from INTVCC. When the bottom MOSFET switch M3 turns on, the switch node SW2 drops low and the bootstrap capacitor C2, is charged through D2 from INTVCC. The bootstrap capacitors C1 and C2 need to store about 100 times the gate charge required by the top MOSFET switch M1 and M4. In most applications a 0.1µF to 0.47µF, X5R or X7R ceramic capacitor is adequate. Efficiency Considerations The power efficiency of a switching regulator is equal to the output power divided by the input power times 100%. It is often useful to analyze individual losses to determine what is limiting the efficiency and which change would produce the most improvement. Although all dissipative elements in circuits produce losses, four main sources account for most of the losses in LT3791 circuits: 1. DC I2R losses. These arise from the resistances of the MOSFETs, sensing resistor, inductor and PC board traces and cause the efficiency to drop at high output currents. 2. Transition loss. This loss arises from the brief amount of time switch M1 or switch M3 spends in the saturated region during switch node transitions. It depends upon the input voltage, load current, driver strength and MOSFET capacitance, among other factors. The loss is significant at input voltages above 20V and can be estimated from: Transition Loss ≈ 2.7 • VIN2 • IOUT • CRSS • f where CRSS is the reverse-transfer capacitance. 3. INTVCC current. This is the sum of the MOSFET driver and control currents. 4. CIN and COUT loss. The input capacitor has the difficult job of filtering the large RMS input current to the regulator in buck operation. The output capacitor has the difficult job of filtering the large RMS output current in boost operation. Both CIN and COUT are required to have low ESR to minimize the AC I2R loss and sufficient capacitance to prevent the RMS current from causing additional upstream losses in fuses or batteries. 5. Other losses. Schottky diode D3 and D4 are responsible for conduction losses during dead time and light load conduction periods. Inductor core loss occurs predominately at light loads. Switch M3 causes reverse recovery current loss in boost operation. When making adjustments to improve efficiency, the input current is the best indicator of changes in efficiency. If you make a change and the input current decreases, then the efficiency has increased. If there is no change in the input current, then there is no change in efficiency. 3791f 23 LT3791 APPLICATIONS INFORMATION PC Board Layout Checklist The basic PC board layout requires a dedicated ground plane layer. Also, for high current, a multilayer board provides heat sinking for power components. The PGND ground plane layer should not have any traces and it should be as close as possible to the layer with power MOSFETs. n Place CIN, switch M1, switch M2 and D1 in one compact area. Place COUT, switch M3, switch M4 and D2 in one compact area. n Use immediate vias to connect the components (including the LT3791’s SGND and PGND pins) to the ground plane. Use several large vias for each power component. n The path formed by switch M1, switch M2, D1 and the CIN capacitor should have short leads and PC trace lengths. The path formed by switch M3, switch M4, D2 and the COUT capacitor also should have short leads and PC trace lengths. n The output capacitor (–) terminals should be connected as close as possible to the (–) terminals of the input capacitor. n Connect the top driver bootstrap capacitor, C1, closely to the BST1 and SW1 pins. Connect the top driver bootstrap capacitor, C2, closely to the BST2 and SW2 pins. n Connect the input capacitors, CIN, and output capacitors, COUT, closely to the power MOSFETs. These capacitors carry the MOSFET AC current in boost and buck operation. n Use planes for VIN and VOUT to maintain good voltage filtering and to keep power losses low. n Flood all unused areas on all layers with copper. Flooding with copper will reduce the temperature rise of power components. Connect the copper areas to any DC net (VIN or PGND). n Separate the signal and power grounds. All small-signal components should return to the SGND pin at one point, which is then tied to the PGND pin close to the sources of switch M2 and switch M3. n Place switch M2 and switch M3 as close to the controller as possible, keeping the PGND, BG and SW traces short. n Keep the high dV/dT SW1, SW2, BST1, BST2, TG1 and TG2 nodes away from sensitive small-signal nodes. n Route SNSN and SNSP leads together with minimum PC trace spacing. Avoid sense lines pass through noisy areas, such as switch nodes. Ensure accurate current sensing with Kelvin connections at the SENSE resistor. n Connect the VC pin compensation network close to the IC, between VC and the signal ground pins. The capacitor helps to filter the effects of PCB noise and output voltage ripple voltage from the compensation loop. n Connect the INTVCC bypass capacitor, CVCC, close to the IC, between the INTVCC and the power ground pins. This capacitor carries the MOSFET drivers’ current peaks. An additional 0.1µF ceramic capacitor placed immediately next to the INTVCC and PGND pins can help improve noise performance substantially. n 3791f 24 LT3791 TYPICAL APPLICATIONS 98% Efficient 50W (25V 2A) Buck-Boost LED Driver VIN 4.7V TO 58V RIN 0.003Ω VIN C3 R7 1µF 50Ω IVINN BST1 TG1 IVINP R3 1M BG1 OVLO INTVCC R9 200k R4 54.9k LT3791 R10 200k CSS 10nF BG2 SW2 TG2 FB ISN PWMOUT RT SGND R8 RC 86.6k 2.2k 300kHz CC 10nF D1, D2: NXP BAT46WJ L1: COOPER HC9-100-R 10µH M1, M2: RENESAS RJK0651DPB 60VDS M3, M4: RENESAS RJK0451DPB 40VDS M5: VISHAY Si2318CDS 40VDS M5 3791 TA02a 100Hz 50:1 PWM Dimming (VIN = 12V) 100 EFFICIENCY (%) RLED 0.05Ω 25V LED 2A Efficiency vs VIN 96 R6 44.2k ISP CTRL TEST1 SS SYNC VC 98 M3 COUT 4.7µF 50V ×4 SNSN PGND VREF R12 237k L1 10µH M4 R5 1M RSENSE 0.004Ω SHORTLED PWM IVINMON ISMON CLKOUT C8 0.1µF M2 C2 0.1µF C1 0.1µF CVCC 4.7µF SNSP OPENLED R11 1M M1 SWI EN/UVLO R2 121k D1 D2 TEST2 BST2 C7 470nF R1 332k INTVCC CIN 2.2µF 100V ×4 PWM 5V/DIV BOOST 94 BUCK BUCK-BOOST IL1 2A/DIV 92 90 ILED 2A/DIV 88 86 50µs/DIV 84 3791 TA02c 82 80 0 10 30 40 20 INPUT VOLTAGE (V) 50 60 3791 TA02b 3791f 25 LT3791 TYPICAL APPLICATIONS 98% Efficient 60W (12V 5A) Voltage Regulator Runs Down to 3V VIN VIN 3V TO 55V CIN 4.7µF 100V ×4 R10 200k INTVCC TEST2 SHORTLED D5 BST2 OPENLED BST1 VIN C3 1µF TG1 BG1 IVINP D6 R1 866k TEST1 SNSP LT3791 IVINMON ISMON CLKOUT R3 1M EN/UVLO OVLO R2 576k R4 57.6k C8 0.1µF L1 6.8µH SS SYNC VC RSENSE 0.004Ω SNSN PGND R5 732k BG2 R6 80.6k SGND PWMOUT RT RC 2.2k CC 22nF R8 86.6k 300kHz 3791 TA03a D1, D2: NXP BAT46WJ D3: IRF 10BQ060 D4: IRF 10BQ040 D5, D6: DIODES INC. BAT46W L1: WURTH ELEKTRONIK WE-HCI 7443556680 M1, M2: RENASAS RJK0651DPB 60VDS M3, M4: VISHAY SiR424DP 40VDS 6 BUCK 96 EFFICIENCY (%) Maximum Output Current vs VIN IOUT = 5A BOOST BUCK-BOOST 94 COUT 10µF 25V ×3 VOUT 12V 5A M3 D3 MAXIMUM OUTPUT CURRENT (A) 98 COUT2 100µF 25V ROUT 0.015Ω D4 M4 Efficiency vs VIN 100 + ISP ISN CTRL CSS 10nF M2 C2 0.1µF C1 0.1µF SW2 TG2 FB PWM VREF RFAULT 100k M1 SWI IVINN CVCC 4.7µF D1 D2 92 90 88 86 84 5 4 3 2 1 82 80 0 10 30 40 20 INPUT VOLTAGE (V) 50 60 3791 TA03b 0 3 4 5 6 7 8 9 10 20 30 40 50 60 INPUT VOLTAGE (V) 3791 TA03c 3791f 26 LT3791 PACKAGE DESCRIPTION Please refer to http://www.linear.com/designtools/packaging/ for the most recent package drawings. FE Package 38-Lead Plastic TSSOP (4.4mm) (Reference LTC DWG # 05-08-1772 Rev C) Exposed Pad Variation AA 4.75 REF 38 9.60 – 9.80* (.378 – .386) 4.75 REF (.187) 20 6.60 ±0.10 4.50 REF 2.74 REF SEE NOTE 4 6.40 2.74 REF (.252) (.108) BSC 0.315 ±0.05 1.05 ±0.10 0.50 BSC RECOMMENDED SOLDER PAD LAYOUT 4.30 – 4.50* (.169 – .177) 0.50 – 0.75 (.020 – .030) 0.09 – 0.20 (.0035 – .0079) NOTE: 1. CONTROLLING DIMENSION: MILLIMETERS 2. DIMENSIONS ARE IN MILLIMETERS (INCHES) 3. DRAWING NOT TO SCALE 1 0.25 REF 19 1.20 (.047) MAX 0° – 8° 0.50 (.0196) BSC 0.17 – 0.27 (.0067 – .0106) TYP 0.05 – 0.15 (.002 – .006) FE38 (AA) TSSOP REV C 0910 4. RECOMMENDED MINIMUM PCB METAL SIZE FOR EXPOSED PAD ATTACHMENT *DIMENSIONS DO NOT INCLUDE MOLD FLASH. MOLD FLASH SHALL NOT EXCEED 0.150mm (.006") PER SIDE 3791f Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. 27 LT3791 TYPICAL APPLICATION 120W (24V 5A) Buck-Boost Voltage Regulator VIN 12V TO 58V R10 200k INTVCC TEST2 SHORTLED BST2 OPENLED BST1 VIN C3 1µF TG1 BG1 IVINP TEST1 R1 499k EN/UVLO OVLO R2 56.2k R4 27.4k C8 0.1µF C2 0.1µF C1 0.1µF L1 10µH SS SYNC VC RT COUT 4.7µF 50V ×6 M4 RC 1.1k CC 22nF CSS 10nF R8 147k 200kHz VOUT 24V 5A M3 SNSN PGND R4 732k BG2 R5 18.7k SGND PWMOUT CIN2 47µF 100V ROUT 0.015Ω ISP ISN CTRL + RSENSE 0.004Ω SW2 TG2 FB PWM VREF RFAULT 100k M2 SNSP LT3791 IVINMON ISMON CLKOUT R3 499k M1 SWI IVINN CVCC 4.7µF D1 D2 CIN 2.2µF 100V 3791 TA04 D1, D2: NXP BAT46WJ L1: WURTH ELEKTRONICS 74435571100 10µH M1, M2: RENESAS RJK0651DPB 60VDS M3, M4: RENESAS RJK0451DPB 40VDS RELATED PARTS PART NUMBER DESCRIPTION COMMENTS ® LTC 3780 High Efficiency, Synchronous, 4-Switch Buck-Boost Controller VIN: 4V to 36V, VOUT Range: 0.8V to 30V, ISD < 55µA, SSOP-24, QFN-32 Packages LTC3789 High Efficiency, Synchronous, 4-Switch Buck-Boost Controller VIN: 4V to 38V, VOUT Range: 0.8V to 38V, ISD < 40µA, 4mm × 5mm QFN-28, SSOP-28 Packages LT3755/LT3755-1 High Side 60V, 1MHz LED Controller with True Color LT3755-2 3000:1 PWM Dimming VIN: 4.5V to 40V, VOUT Range: 5V to 60V, 3000:1 True Color PWM™, Analog, ISD < 1µA, 3mm × 3mm QFN-16, MSOP-16E Packages LT3756/LT3756-1 High Side 100V, 1MHz LED Controller with True Color VIN: 6V to 100V, VOUT Range: 5V to 100V, 3000:1 True Color PWM, Analog, ISD < 1µA, 3mm × 3mm QFN-16, MSOP-16E Packages LT3756-2 3000:1 PWM Dimming LT3596 60V, 300mA Step-Down LED Driver VIN: 6V to 60V, VOUT Range: 5V to 55V, 10000:1 True Color PWM, Analog, ISD < 1µA, 5mm × 8mm QFN-52 Package LT3743 Synchronous Step-Down 20A LED Driver with Thee-State LED Current Control VIN: 5.5V to 36V, VOUT Range: 5.5V to 35V, 3000:1 True Color PWM, Analog, ISD < 1µA, 4mm × 5mm QFN-28, TSSOP-28E Packages 3791f 28 Linear Technology Corporation LT 0312 • PRINTED IN USA 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408) 432-1900 ● FAX: (408) 434-0507 ● www.linear.com LINEAR TECHNOLOGY CORPORATION 2012