LTC3717 Wide Operating Range, No RSENSETM Step-Down Controller for DDR/QDR Memory Termination U DESCRIPTIO FEATURES ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ VOUT = 1/2 VIN (Supply Splitter) Adjustable and Symmetrical Sink/Source Current Limit up to 20A ±0.65% Output Voltage Accuracy Up to 97% Efficiency No Sense Resistor Required Ultrafast Transient Response True Current Mode Control 2% to 90% Duty Cycle at 200kHz tON(MIN) ≤ 100ns Stable with Ceramic COUT Dual N-Channel MOSFET Synchronous Drive Power Good Output Voltage Monitor Wide VCC Range: 4V to 36V Adjustable Switching Frequency up to 1.5MHz Output Overvoltage Protection Optional Short-Circuit Shutdown Timer Available in a 16-Pin Narrow SSOP Package The LTC®3717 is a synchronous step-down switching regulator controller for double data rate (DDR) and Quad Data RateTM (QDRTM) memory termination. The controller uses a valley current control architecture to deliver very low duty cycles without requiring a sense resistor. Operating frequency is selected by an external resistor and is compensated for variations in VIN. Forced continuous operation reduces noise and RF interference. Output voltage is internally set to half of VREF, which is user programmable. Fault protection is provided by an output overvoltage comparator and optional short-circuit shutdown timer. Soft-start capability for supply sequencing is accomplished using an external timing capacitor. The regulator current limit level is symmetrical and user programmable. Wide supply range allows operation from 4V to 36V at the VCC input. U APPLICATIO S ■ ■ Bus Termination: DDR and QDR Memory, SSTL, HSTL, ... Notebook Computers, Desktop Servers Tracking Power Supply U ■ , LTC and LT are registered trademarks of Linear Technology Corporation. No RSENSE is a trademark of Linear Technology Corporation. QDR RAMs and Quad Data Rate RAMs comprise a new family of products developed by Cypress Semiconductor, Hitachi, IDT, Micron Technology, Inc. and Samsung. TYPICAL APPLICATIO 1µF RUN/SS RC 20k + VDD = 2.5V M1 Si7840DP TG CSS 0.1µF D2 B320A SW ITH CB 0.22µF BOOST INTVCC BG PGOOD PGND L1 0.68µH DB CMDSH-3 LTC3717 SGND VIN 2.5V TO 5.5V ION VREF CC 470pF Efficiency vs Load Current RON 715k VCC + M2 Si7840DP CVCC 4.7µF + 100 VOUT 1.25V COUT ±10A 180µF 4V ×2 D1 B320A VIN = 5V 80 70 VIN = 2.5V 60 50 40 30 20 10 0 VFB VOUT = 1.25V 90 CIN 150µF 6.3V ×2 EFFICIENCY (%) VCC 5V TO 28V 3717 F01a 0 2 4 6 8 10 LOAD CURRENT (A) 12 14 3717 F01b Figure 1. High Efficiency DDR Memory Termination Supply sn3717 3717fs 1 LTC3717 W W W AXI U U ABSOLUTE RATI GS U U W PACKAGE/ORDER I FOR ATIO (Note 1) Input Supply Voltage (VCC, ION) .................36V to – 0.3V Boosted Topside Driver Supply Voltage (BOOST) ................................................... 42V to – 0.3V SW Voltage .................................................. 36V to – 5V EXTVCC, (BOOST – SW), RUN/SS, PGOOD Voltages ....................................... 7V to – 0.3V VREF, VRNG Voltages ...............(INTVCC + 0.3V) to – 0.3V ITH, VFB Voltages...................................... 2.7V to – 0.3V TG, BG, INTVCC, EXTVCC Peak Currents .................... 2A TG, BG, INTVCC, EXTVCC RMS Currents .............. 50mA Operating Ambient Temperature Range (Note 4) ................................... – 40°C to 85°C Junction Temperature (Note 2) ............................ 125°C Storage Temperature Range ................. – 65°C to 150°C Lead Temperature (Soldering, 10 sec).................. 300°C ORDER PART NUMBER TOP VIEW RUN/SS 1 16 BOOST PGOOD 2 15 TG VRNG 3 14 SW ITH 4 SGND 5 13 PGND 12 BG ION 6 11 INTVCC VFB 7 10 VCC VREF 8 LTC3717EGN 9 GN PART MARKING EXTVCC 3717 GN PACKAGE 16-LEAD PLASTIC SSOP TJMAX = 125°C, θJA = 130°C/ W Consult LTC Marketing for parts specified with wider operating temperature ranges. ELECTRICAL CHARACTERISTICS The ● denotes specifications which apply over the full operating temperature range, otherwise specifications are TA = 25°C. VCC = 15V unless otherwise noted. SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS 1000 15 2000 30 µA µA 0.65 % Main Control Loop IQ Input DC Supply Current Normal Shutdown Supply Current VRUN/SS = 0V VFB Feedback Voltage Accuracy ITH = 1.2V (Note 3), VREF = 2.4V ∆VFB(LINEREG) Feedback Voltage Line Regulation VCC= 4V to 36V, ITH = 1.2V (Note 3) ∆VFB(LOADREG) Feedback Voltage Load Regulation ITH = 0.5V to 1.9V (Note 3) – 0.05 – 0.3 % gm(EA) Error Amplifier Transconductance ITH = 1.2V (Note 3) 0.93 1.13 1.33 mS tON On-Time ION = 30µA ION = 60µA 186 95 233 115 280 135 ns ns tON(MIN) Minimum On-Time ION = 180µA 50 100 ns tOFF(MIN) Minimum Off-Time ION = 30µA 300 400 ns VSENSE(MAX) Maximum Current Sense Threshold (Source) VPGND – VSW VRNG = 1V, VFB = VREF/2 – 50mV VRNG = 0V, VFB = VREF/2 – 50mV VRNG = INTVCC, VFB = VREF/2 – 50mV ● ● ● 108 76 148 135 95 185 162 114 222 mV mV mV VSENSE(MIN) Minimum Current Sense Threshold (Sink) VPGND – VSW VRNG = 1V, VFB = VREF/2 + 50mV VRNG = 0V, VFB = VREF/2 + 50mV VRNG = INTVCC, VFB = VREF/2 + 50mV ● ● ● – 140 – 97 – 200 – 165 – 115 – 235 – 190 – 133 – 270 mV mV mV ∆VFB(OV) Output Overvoltage Fault Threshold 8 10 12 % ∆VFB(UV) Output Undervoltage Fault Threshold VRUN/SS(ON) RUN Pin Start Threshold VRUN/SS(LE) RUN Pin Latchoff Enable Threshold RUN/SS Pin Rising VRUN/SS(LT) RUN Pin Latchoff Threshold RUN/SS Pin Falling IRUN/SS(C) Soft-Start Charge Current – 0.5 IRUN/SS(D) Soft-Start Discharge Current 0.8 1.8 3 – 0.65 0.002 ● %/V – 25 ● 0.8 % 1.5 2 V 4 4.5 V 3.5 4.2 V – 1.2 –3 µA µA sn3717 3717fs 2 LTC3717 ELECTRICAL CHARACTERISTICS The ● denotes specifications which apply over the full operating temperature range, otherwise specifications are TA = 25°C. VCC = 15V unless otherwise noted. SYMBOL PARAMETER CONDITIONS VCC(UVLO) Undervoltage Lockout Threshold VCC Falling VCC(UVLOR) Undervoltage Lockout Threshold VCC Rising TG RUP TG Driver Pull-Up On Resistance TG High TG RDOWN TG Driver Pull-Down On Resistance BG RUP BG RDOWN MIN TYP MAX UNITS ● 3.4 3.9 ● 3.5 4 V 2 3 Ω TG Low 2 3 Ω BG Driver Pull-Up On Resistance BG High 3 4 Ω BG Driver Pull-Down On Resistance BG Low 1 2 Ω TG tr TG Rise Time CLOAD = 3300pF 20 ns TG tf TG Fall Time CLOAD = 3300pF 20 ns BG tr BG Rise Time CLOAD = 3300pF 20 ns BG tf BG Fall Time CLOAD = 3300pF 20 ns V Internal VCC Regulator VINTVCC Internal VCC Voltage 6V < VCC < 30V, VEXTVCC = 4V ∆VLDO(LOADREG) Internal VCC Load Regulation ICC = 0mA to 20mA, VEXTVCC = 4V VEXTVCC EXTVCC Switchover Voltage ICC = 20mA, VEXTVCC Rising ∆VEXTVCC EXTVCC Switch Drop Voltage ICC = 20mA, VEXTVCC = 5V ∆VEXTVCC(HYS) EXTVCC Switchover Hysteresis ● ● 4.7 4.5 5 5.3 V – 0.1 ±2 % 300 mV 4.7 150 V 200 mV PGOOD Output ∆VFBH PGOOD Upper Threshold VFB Rising (0% = 1/3 VREF) 8 10 12 % ∆VFBL PGOOD Lower Threshold VFB Falling (0% = 1/3 VREF) –8 – 10 – 12 % ∆VFB(HYS) PGOOD Hysteresis VFB Returning (0% = 1/3 VREF) 1 2 % VPGL PGOOD Low Voltage IPGOOD = 5mA 0.15 0.4 V Note 1: Absolute Maximum Ratings are those values beyond which the life of a device may be impaired. Note 2: TJ is calculated from the ambient temperature TA and power dissipation PD as follows: LTC3717EGN: TJ = TA + (PD • 130°C/W) Note 3: The LTC3717 is tested in a feedback loop that adjusts VFB to achieve a specified error amplifier output voltage (ITH). Note 4: The LTC3717E is guaranteed to meet performance specifications from 0°C to 70°C. Specifications over the –40°C to 85°C operating temperature range are assured by design, characterization and correlation with statistical process controls. sn3717 3717fs 3 LTC3717 U W TYPICAL PERFOR A CE CHARACTERISTICS VOUT/VIN Tracking Ratio vs Input Voltage Efficiency vs Load Current 100 VIN = 2.5V VOUT = 1.25V 90 400 LOAD = 0A 49.95 80 LOAD = 10A 350 70 60 50 40 30 FREQUENCY (kHz) 49.90 VOUT/VIN (%) EFFICIENCY (%) Frequency vs Input Voltage 450 50.00 LOAD = 1A 49.85 LOAD = 10A 49.80 49.75 20 300 250 LOAD = 0A 200 150 100 10 49.70 FIGURE 1 CIRCUIT 0 0.01 0.1 1 10 LOAD CURRENT (A) 49.65 1.5 100 VOUT = 1.25V FIGURE 1 CIRCUIT 50 FIGURE 1 CIRCUIT 0 1.7 1.9 2.1 2.3 2.5 INPUT VOLTAGE (V) 2.7 1.5 2.9 1.7 1.9 2.1 2.3 2.5 INPUT VOLTAGE (V) 2.7 2.9 3717 G06 3717 G08 3717 G07 Load Regulation Start-Up Response Load-Step Transient 0 VIN = 2.5V VOUT = 1.25V –0.1 VOUT 1V/DIV VOUT 200mV/DIV IL 2A/DIV IL 5A/DIV ∆VOUT/VOUT (%) –0.2 –0.3 –0.4 –0.5 VIN = 2.5V 4ms/DIV VOUT = 1.25V LOAD = 0.2Ω FIGURE 1 CIRCUIT FIGURE 1 CIRCUIT –0.6 0 1 2 3 4 5 6 7 LOAD CURRENT (A) 8 9 10 VIN = 2.5V 20µs/DIV VOUT = 1.25V LOAD = 500mA TO 10A STEP FIGURE 1 CIRCUIT 3718 G09.eps 3718 G10.eps 3717 G09 On-Time vs VON Voltage 1000 On-Time vs Temperature 300 IION = 30µA IION = 30µA 250 600 400 200 ON-TIME (ns) ON-TIME (ns) 800 ON-TIME (ns) On-Time vs ION Current 10k 150 100 200 0 1k 100 50 0 2 1 VON VOLTAGE (V) 3 3717 G11 0 –50 –25 10 50 25 75 0 TEMPERATURE (°C) 100 125 3717 G12 1 10 ION CURRENT (µA) 100 3717 G13 sn3717 3717fs 4 LTC3717 U W TYPICAL PERFOR A CE CHARACTERISTICS –0.1 2 –0.2 –0.3 5.0 RUN/SS THRESHOLD (V) 3 FCB PIN CURRENT (µA) ∆INTVCC (%) 0 PULL-DOWN CURRENT 1 0 PULL-UP CURRENT LATCHOFF ENABLE 4.0 3.5 0 10 30 40 20 INTVCC LOAD CURRENT (mA) LATCHOFF THRESHOLD –2 –50 –25 50 50 25 0 75 TEMPERATURE (°C) 100 Undervoltage Lockout Threshold vs Temperature 3.0 2.5 75 0 25 50 TEMPERATURE (C) 100 125 300 250 200 150 100 50 0 0.50 0.75 1.00 1.25 1.50 VRNG (V) 1.75 3717 G17 100 1.50 160 1.40 140 1.30 120 60 40 20 0 –50 –30 –10 10 30 50 70 90 110 130 3717 G20 TEMPERATURE (°C) 120 100 80 60 40 20 0 2.0 2.2 2.4 2.6 2.8 3.0 RUN/SS (V) 3.2 3.4 3.6 3717 G19 1.20 gm (ms) 80 140 Error Amplifier gm vs Temperature 180 100 2.00 125 160 3717 G18 Maximum Current Sense Threshold vs Temperature, VRNG = 1V MAXIMUM CURRENT SENSE THRESHOLD (mV) 75 0 25 50 TEMPERATURE (°C) Maximum Current Sense Threshold vs RUN/SS Voltage, VRNG = 1V MAXIMUM CURRENT SENSE THRESHOLD (mV) MAXIMUM CURRENT SENSE THRESHOLD (mV) 3.5 –25 3717 G16 Maximum Current Sense Threshold vs VRNG Voltage 4.0 2.0 –50 –25 3.0 –50 125 3717 G15 3717 G14 UNDERVOLTAGE LOCKOUT THRESHOLD (V) 4.5 –1 –0.4 –0.5 RUN/SS Latchoff Thresholds vs Temperature RUN/SS Latchoff Thresholds vs Temperature INTVCC Load Regulation 1.10 1.00 0.90 0.80 0.70 –50 –30 –10 10 30 50 70 90 110 130 3717 G21 TEMPERATURE (°C) sn3717 3717fs 5 LTC3717 U U U PI FU CTIO S RUN/SS (Pin 1): Run Control and Soft-Start Input. A capacitor to ground at this pin sets the ramp time to full output current (approximately 3s/µF) and the time delay for overcurrent latchoff (see Applications Information). Forcing this pin below 0.8V shuts down the device. PGOOD (Pin 2): Power Good Output. Open drain logic output that is pulled to ground when the output voltage is not within ±10% of the regulation point. VRNG (Pin 3): Sense Voltage Range Input. The voltage at this pin is ten times the nominal sense voltage at maximum output current and can be set from 0.5V to 2V by a resistive divider from INTVCC. The nominal sense voltage defaults to 70mV when this pin is tied to ground, 140mV when tied to INTVCC. ITH (Pin 4): Current Control Threshold and Error Amplifier Compensation Point. The current comparator threshold increases with this control voltage. The voltage ranges from 0V to 2.4V with 0.8V corresponding to zero sense voltage (zero current). SGND (Pin 5): Signal Ground. All small-signal components and compensation components should connect to this ground, which in turn connects to PGND at one point. ION (Pin 6): On-Time Current Input. Tie a resistor from VIN to this pin to set the one-shot timer current and thereby set the switching frequency. VFB (Pin 7): Error Amplifier Feedback Input. This pin connects to VOUT and divides its voltage to 2/3 • VFB through precision internal resistors before it is applied to the input of the error amplifier. Do not apply more than 1.5V on VFB. For higher output voltages, attach an external resistor R2 (1/2 • R1 at VREF) from VOUT to VFB. VREF (Pin 8): Positive Input of Internal Error Amplifier. This pin connects to an external reference and divides its voltage to 1/3 VREF through precision internal resisters before it is applied to the positive input of the error amplifier. Reference voltage for output voltage, power good threshold, and short-circuit shutdown threshold. Do not apply more than 3V on VREF. If higher voltages are used, connect an external resistor (R1 ≥ 160k) from voltage reference to VREF. EXTVCC (Pin 9): External VCC Input. When EXTVCC exceeds 4.7V, an internal switch connects this pin to INTVCC and shuts down the internal regulator so that controller and gate drive power is drawn from EXTVCC. Do not exceed 7V at this pin and ensure that EXTVCC < VCC. VCC (Pin 10): Bias Input Supply. 4V to 36V operating range. Decouple this pin to PGND with an RC filter (1Ω, 0.1µF). INTVCC (Pin 11): Internal 5V Regulator Output. The driver and control circuits are powered from this voltage. Decouple this pin to power ground with a minimum of 4.7µF low ESR tantalum or ceramic capacitor. BG (Pin 12): Bottom Gate Drive. Drives the gate of the bottom N-channel MOSFET between ground and INTVCC. PGND (Pin 13): Power Ground. Connect this pin closely to the source of the bottom N-channel MOSFET, the (–) terminal of CVCC and the (–) terminal of CIN. SW (Pin 14): Switch Node. The (–) terminal of the bootstrap capacitor CB connects here. This pin swings from a diode voltage drop below ground up to a diode voltage drop above VIN. TG (Pin 15): Top Gate Drive. Drives the top N-channel MOSFET with a voltage swing equal to INTVCC superimposed on the switch node voltage SW. BOOST (Pin 16): Boosted Floating Driver Supply. The (+) terminal of the bootstrap capacitor CB connects here. This pin swings from a diode voltage drop below INTVCC up to VIN + INTVCC. sn3717 3717fs 6 LTC3717 W FU CTIO AL DIAGRA U U RON VIN 6 ION 10 VCC 9 EXTVCC + 4.7V CIN + – 1.192V BNGP 5V REG BOOST tON = 0.7V (10pF) IION R S TG Q FCNT 14 L1 SWITCH LOGIC IREV VOUT DB – – M1 SW + ICMP CB 15 ON 20k + 16 INTVCC INTVCC 11 SHDN 1.4V BG OV 12 + COUT CVCC M2 VRNG PGND 3 × 13 PGOOD 0.7V 2 5.7µA 1 240k + Q2 3 V 10 REF ITHB Q1 Q5 – 20k + 40k OV – – SS + – – + 8 40k 5 SGND 1.2µA 6V 0.6V 80k 11 V 30 REF RUN SHDN + EA VREF 7 VFB UV 4 ITH RC CC1 VREF 4 1 RUN/SS CSS 3717 FD sn3717 3717fs 7 LTC3717 U OPERATIO Main Control Loop The LTC3717 is a current mode controller for DC/DC step-down converters. In normal operation, the top MOSFET is turned on for a fixed interval determined by a one-shot timer OST. When the top MOSFET is turned off, the bottom MOSFET is turned on until the current comparator ICMP trips, restarting the one-shot timer and initiating the next cycle. Inductor current is determined by sensing the voltage between the PGND and SW pins using the bottom MOSFET on-resistance . The voltage on the ITH pin sets the comparator threshold corresponding to inductor valley current. The error amplifier EA adjusts this ITH voltage by comparing 2/3 of the feedback signal VFB from the output voltage with a reference equal to 1/3 of the VREF voltage. If the load current increases, it causes a drop in the feedback voltage relative to the reference. The ITH voltage then rises until the average inductor current again matches the load current. As a result in normal DDR operation VOUT is equal to 1/2 of the VREF voltage. The operating frequency is determined implicitly by the top MOSFET on-time and the duty cycle required to maintain regulation. The one-shot timer generates an ontime that is proportional to the ideal duty cycle, thus holding frequency approximately constant with changes in VIN. The nominal frequency can be adjusted with an external resistor RON. Overvoltage and undervoltage comparators OV and UV pull the PGOOD output low if the output feedback voltage exits a ±10% window around the regulation point. Furthermore, in an overvoltage condition, M1 is turned off and M2 is turned on and held on until the overvoltage condition clears. Pulling the RUN/SS pin low forces the controller into its shutdown state, turning off both M1 and M2. Releasing the pin allows an internal 1.2µA current source to charge up an external soft-start capacitor CSS. When this voltage reaches 1.5V, the controller turns on and begins switching, but with the ITH voltage clamped at approximately 0.6V below the RUN/SS voltage. As CSS continues to charge, the soft-start current limit is removed. INTVCC/EXTVCC Power Power for the top and bottom MOSFET drivers and most of the internal controller circuitry is derived from the INTVCC pin. The top MOSFET driver is powered from a floating bootstrap capacitor CB. This capacitor is recharged from INTVCC through an external Schottky diode DB when the top MOSFET is turned off. When the EXTVCC pin is grounded, an internal 5V low dropout regulator supplies the INTVCC power from VCC. If EXTVCC rises above 4.7V, the internal regulator is turned off, and an internal switch connects EXTVCC to INTVCC. This allows a high efficiency source connected to EXTVCC, such as an external 5V supply or a secondary output from the converter, to provide the INTVCC power. Voltages up to 7V can be applied to EXTVCC for additional gate drive. If the VCC voltage is low and INTVCC drops below 3.4V, undervoltage lockout circuitry prevents the power switches from turning on. U W U U APPLICATIO S I FOR ATIO The basic LTC3717 application circuit is shown in Figure 1. External component selection is primarily determined by the maximum load current and begins with the selection of the sense resistance and power MOSFET switches. The LTC3717 uses the on-resistance of the synchronous power MOSFET for determining the inductor current. The desired amount of ripple current and operating frequency largely determines the inductor value. Finally, CIN is selected for its ability to handle the large RMS current into the converter and COUT is chosen with low enough ESR to meet the output voltage ripple and transient specification. Maximum Sense Voltage and VRNG Pin Inductor current is determined by measuring the voltage across a sense resistance that appears between the PGND and SW pins. The maximum sense voltage is set by the voltage applied to the VRNG pin and is equal to approximately (0.13)VRNG for sourcing current and (0.17)VRNG for sinking current. The current mode control loop will not allow the inductor current valleys to exceed (0.13)VRNG/ RSENSE for sourcing and (0.17)VRNG/RSENSE for sinking. In practice, one should allow some margin for variations in sn3717 3717fs 8 LTC3717 U W U U APPLICATIO S I FOR ATIO RSENSE = VRNG 10 • IOUT (MAX) An external resistive divider from INTVCC can be used to set the voltage of the VRNG pin between 0.5V and 2V resulting in nominal sense voltages of 50mV to 200mV. Additionally, the VRNG pin can be tied to SGND or INTVCC in which case the nominal sense voltage defaults to 70mV or 140mV, respectively. The maximum allowed sense voltage is about 1.3 times this nominal value for positive output current and 1.7 times the nominal value for negative output current. Power MOSFET Selection The LTC3717 requires two external N-channel power MOSFETs, one for the top (main) switch and one for the bottom (synchronous) switch. Important parameters for the power MOSFETs are the breakdown voltage V(BR)DSS, threshold voltage V(GS)TH, on-resistance RDS(ON), reverse transfer capacitance CRSS and maximum current IDS(MAX). The gate drive voltage is set by the 5V INTVCC supply. Consequently, logic-level threshold MOSFETs must be used in LTC3717 applications. If the input voltage is expected to drop below 5V, then sub-logic level threshold MOSFETs should be considered. When the bottom MOSFET is used as the current sense element, particular attention must be paid to its onresistance. MOSFET on-resistance is typically specified with a maximum value RDS(ON)(MAX) at 25°C. In this case, additional margin is required to accommodate the rise in MOSFET on-resistance with temperature: the load current. During LTC3717’s normal operation, the duty cycles for the MOSFETs are: VOUT VIN V –V = IN OUT VIN D TOP = DBOT The resulting power dissipation in the MOSFETs at maximum output current are: PTOP = DTOP IOUT(MAX)2 ρT(TOP) RDS(ON)(MAX) + k VIN2 IOUT(MAX) CRSS f PBOT = DBOT IOUT(MAX)2 ρT(BOT) RDS(ON)(MAX) Both MOSFETs have I2R losses and the top MOSFET includes an additional term for transition losses, which are largest at high input voltages. The constant k = 1.7A–1 can be used to estimate the amount of transition loss. The bottom MOSFET losses are greatest when the bottom duty cycle is near 100%, during a short-circuit or at high input voltage. 2.0 ρT NORMALIZED ON-RESISTANCE the LTC3717 and external component values and a good guide for selecting the sense resistance is: 1.5 1.0 0.5 0 – 50 50 100 0 JUNCTION TEMPERATURE (°C) 150 3717 F02 RDS(ON)(MAX) R = SENSE ρT The ρT term is a normalization factor (unity at 25°C) accounting for the significant variation in on-resistance with temperature, typically about 0.4%/°C as shown in Figure 2. For a maximum junction temperature of 100°C, using a value ρT = 1.3 is reasonable. The power dissipated by the top and bottom MOSFETs strongly depends upon their respective duty cycles and Figure 2. RDS(ON) vs. Temperature Operating Frequency The choice of operating frequency is a tradeoff between efficiency and component size. Low frequency operation improves efficiency by reducing MOSFET switching losses but requires larger inductance and/or capacitance in order to maintain low output ripple voltage. The operating frequency of LTC3717 applications is determined implicitly by the one-shot timer that controls the sn3717 3717fs 9 LTC3717 U W U U APPLICATIO S I FOR ATIO on-time tON of the top MOSFET switch. The on-time is set by the current into the ION pin according to: tON = (0.7V) (10pF ) IION Tying a resistor RON from VIN to the ION pin yields an ontime inversely proportional to VIN. For a step-down converter, this results in approximately constant frequency operation as the input supply varies: f= VOUT [HZ ] (0.7V) RON (10pF ) Because the voltage at the ION pin is about 0.7V, the current into this pin is not exactly inversely proportional to VIN, especially in applications with lower input voltages. A more exact equation taking in account the 0.7V drop on the ION pin is: f= VOUT (VIN – 0.7V) [HZ ] (0.7V) RON (10pF )VIN To correct for this error, an additional resistor RON2 connected from the ION pin to the 5V INTVCC supply will further stabilize the frequency. RON2 = 5V RON 0.7V Inductor Selection Given the desired input and output voltages, the inductor value and operating frequency determine the ripple current: V V ∆IL = OUT 1 − OUT VIN f L Lower ripple current reduces core losses in the inductor, ESR losses in the output capacitors and output voltage ripple. Highest efficiency operation is obtained at low frequency with small ripple current. However, achieving this requires a large inductor. There is a tradeoff between component size, efficiency and operating frequency. A reasonable starting point is to choose a ripple current that is about 40% of IOUT(MAX). The largest ripple current occurs at the highest VIN. To guarantee that ripple current does not exceed a specified maximum, the inductance should be chosen according to: VOUT V L= 1 − OUT f ∆IL(MAX) VIN(MAX) Once the value for L is known, the type of inductor must be selected. High efficiency converters generally cannot afford the core loss found in low cost powdered iron cores, forcing the use of more expensive ferrite, molypermalloy or Kool Mµ® cores. A variety of inductors designed for high current, low voltage applications are available from manufacturers such as Sumida, Panasonic, Coiltronics, Coilcraft and Toko. Schottky Diode D1 Selection The Schottky diode D1 shown in Figure 1 conducts during the dead time between the conduction of the power MOSFET switches. It is intended to prevent the body diode of the bottom MOSFET from turning on and storing charge during the dead time, which can cause a modest (about 1%) efficiency loss. The diode can be rated for about one half to one fifth of the full load current since it is on for only a fraction of the duty cycle. In order for the diode to be effective, the inductance between it and the bottom MOSFET must be as small as possible, mandating that these components be placed adjacently. The diode can be omitted if the efficiency loss is tolerable. CIN and COUT Selection The input capacitance CIN is required to filter the square wave current at the drain of the top MOSFET. Use a low ESR capacitor sized to handle the maximum RMS current. IRMS ≅ IOUT (MAX) VOUT VIN VIN –1 VOUT This formula has a maximum at VIN = 2VOUT, where IRMS = IOUT(MAX) / 2. This simple worst-case condition is commonly used for design because even significant deviations do not offer much relief. Note that ripple Kool Mµ is a registered trademark of Magnetics, Inc. sn3717 3717fs 10 LTC3717 U W U U APPLICATIO S I FOR ATIO current ratings from capacitor manufacturers are often based on only 2000 hours of life which makes it advisable to derate the capacitor. The selection of COUT is primarily determined by the ESR required to minimize voltage ripple and load step transients. The output ripple ∆VOUT is approximately bounded by: ∆VOUT 1 ≤ ∆IL ESR + 8fC OUT Since ∆IL increases with input voltage, the output ripple is highest at maximum input voltage. Typically, once the ESR requirement is satisfied, the capacitance is adequate for filtering and has the necessary RMS current rating. Multiple capacitors placed in parallel may be needed to meet the ESR and RMS current handling requirements. Dry tantalum, special polymer, aluminum electrolytic and ceramic capacitors are all available in surface mount packages. Special polymer capacitors offer very low ESR but have lower capacitance density than other types. Tantalum capacitors have the highest capacitance density but it is important to only use types that have been surge tested for use in switching power supplies. Aluminum electrolytic capacitors have significantly higher ESR, but can be used in cost-sensitive applications providing that consideration is given to ripple current ratings and long term reliability. Ceramic capacitors have excellent low ESR characteristics but can have a high voltage coefficient and audible piezoelectric effects. The high Q of ceramic capacitors with trace inductance can also lead to significant ringing. When used as input capacitors, care must be taken to ensure that ringing from inrush currents and switching does not pose an overvoltage hazard to the power switches and controller. To dampen input voltage transients, add a small 5µF to 50µF aluminum electrolytic capacitor with an ESR in the range of 0.5Ω to 2Ω. High performance through-hole capacitors may also be used, but an additional ceramic capacitor in parallel is recommended to reduce the effect of their lead inductance. Top MOSFET Driver Supply (CB, DB) An external bootstrap capacitor CB connected to the BOOST pin supplies the gate drive voltage for the topside MOSFET. This capacitor is charged through diode DB from INTVCC when the switch node is low. When the top MOSFET turns on, the switch node rises to VIN and the BOOST pin rises to approximately VIN + INTVCC. The boost capacitor needs to store about 100 times the gate charge required by the top MOSFET. In most applications 0.1µF to 0.47µF, X5R or X7R dielectric capacitor is adequate. Fault Condition: Current Limit The maximum inductor current is inherently limited in a current mode controller by the maximum sense voltage. In the LTC3717, the maximum sense voltage is controlled by the voltage on the VRNG pin. With valley current control, the maximum sense voltage and the sense resistance determine the maximum allowed inductor valley current. The corresponding output current limits are: ILIMIT POSITIVE = VSNS(MAX) 1 + ∆IL RDS(ON) ρT 2 ILIMIT NEGATIVE = VSNS(MIN) 1 – ∆IL RDS(ON) ρT 2 The current limit value should be checked to ensure that ILIMIT(MIN) > IOUT(MAX). The minimum value of current limit generally occurs with the largest VIN at the highest ambient temperature, conditions that cause the largest power loss in the converter. Note that it is important to check for self-consistency between the assumed MOSFET junction temperature and the resulting value of ILIMIT which heats the MOSFET switches. Caution should be used when setting the current limit based upon the RDS(ON) of the MOSFETs. The maximum current limit is determined by the minimum MOSFET onresistance. Data sheets typically specify nominal and maximum values for RDS(ON), but not a minimum. A reasonable assumption is that the minimum RDS(ON) lies the same amount below the typical value as the maximum lies above it. Consult the MOSFET manufacturer for further guidelines. Minimum Off-time and Dropout Operation The minimum off-time tOFF(MIN) is the smallest amount of sn3717 3717fs 11 LTC3717 U W U U APPLICATIO S I FOR ATIO time that the LTC3717 is capable of turning on the bottom MOSFET, tripping the current comparator and turning the MOSFET back off. This time is generally about 300ns. The minimum off-time limit imposes a maximum duty cycle of tON/(tON + tOFF(MIN)). If the maximum duty cycle is reached, due to a dropping input voltage for example, then the output will drop out of regulation. The minimum input voltage to avoid dropout is: VIN(MIN) = VOUT tON + tOFF(MIN) tON 1. EXTVCC grounded. INTVCC is always powered from the internal 5V regulator. 2. EXTVCC connected to an external supply. A high efficiency supply compatible with the MOSFET gate drive requirements (typically 5V) can improve overall efficiency. 3. EXTVCC connected to an output derived boost network. The low voltage output can be boosted using a charge pump or flyback winding to greater than 4.7V. The system will start-up using the internal linear regulator until the boosted output supply is available. INTVCC Regulator An internal P-channel low dropout regulator produces the 5V supply that powers the drivers and internal circuitry within the LTC3717. The INTVCC pin can supply up to 50mA RMS and must be bypassed to ground with a minimum of 4.7µF tantalum or other low ESR capacitor. Good bypassing is necessary to supply the high transient currents required by the MOSFET gate drivers. Applications using large MOSFETs with a high input voltage and high frequency of operation may cause the LTC3717 to exceed its maximum junction temperature rating or RMS current rating. Most of the supply current drives the MOSFET gates unless an external EXTVCC source is used. In continuous mode operation, this current is IGATECHG = f(Qg(TOP) + Qg(BOT)). The junction temperature can be estimated from the equations given in Note 2 of the Electrical Characteristics. For example, the LTC3717CGN is limited to less than 14mA from a 30V supply: TJ = 70°C + (14mA)(30V)(130°C/W) = 125°C For larger currents, consider using an external supply with the EXTVCC pin. EXTVCC Connection The EXTVCC pin can be used to provide MOSFET gate drive and control power from the output or another external source during normal operation. Whenever the EXTVCC pin is above 4.7V the internal 5V regulator is shut off and an internal 50mA P-channel switch connects the EXTVCC pin to INTVCC. INTVCC power is supplied from EXTVCC until this pin drops below 4.5V. Do not apply more than 7V to the EXTVCC pin and ensure that EXTVCC ≤ VCC. The following list summarizes the possible connections for EXTVCC: 12 External Gate Drive Buffers The LTC3717 drivers are adequate for driving up to about 60nC into MOSFET switches with RMS currents of 50mA. Applications with larger MOSFET switches or operating at frequencies requiring greater RMS currents will benefit from using external gate drive buffers such as the LTC1693. Alternately, the external buffer circuit shown in Figure 4 can be used. Note that the bipolar devices reduce the signal swing at the MOSFET gate, and benefit from an increased EXTVCC voltage of about 6V. INTVCC BOOST 10Ω TG Q1 FMMT619 GATE OF M1 Q2 FMMT720 SW Q3 FMMT619 GATE OF M2 Q4 FMMT720 10Ω BG PGND 3717 F04 Figure 4. Optional External Gate Driver Soft-Start and Latchoff with the RUN/SS Pin The RUN/SS pin provides a means to shut down the LTC3717 as well as a timer for soft-start and overcurrent latchoff. Pulling the RUN/SS pin below 0.8V puts the LTC3717 into a low quiescent current shutdown (IQ < 30µA). Releasing the pin allows an internal 1.2µA current source to charge up the external timing capacitor CSS. If RUN/SS has been pulled all the way to ground, there is a delay before starting of about: tDELAY = 1.5V C SS = (1.3s/µF)C SS 1.2µA sn3717 3717fs LTC3717 U W U U APPLICATIO S I FOR ATIO When the voltage on RUN/SS reaches 1.5V, the LTC3717 begins operating with a clamp on ITH of approximately 0.9V. As the RUN/SS voltage rises to 3V, the clamp on ITH is raised until its full 2.4V range is available. This takes an additional 1.3s/µF. The pin can be driven from logic as shown in Figure 5. Diode D1 reduces the start delay while allowing CSS to charge up slowly for the soft-start function. After the controller has been started and given adequate time to charge up the output capacitor, CSS is used as a short-circuit timer. After the RUN/SS pin charges above 4V, if the output voltage falls below 75% of its regulated value, then a short-circuit fault is assumed. A 1.8µA current then begins discharging CSS. If the fault condition persists until the RUN/SS pin drops to 3.5V, then the controller turns off both power MOSFETs, shutting down the converter permanently. The RUN/SS pin must be actively pulled down to ground in order to restart operation. The overcurrent protection timer requires that the softstart timing capacitor CSS be made large enough to guarantee that the output is in regulation by the time CSS has reached the 4V threshold. In general, this will depend upon the size of the output capacitance, output voltage and load current characteristic. A minimum soft-start capacitor can be estimated from: CSS > COUT VOUT RSENSE (10 – 4 [F/V s]) Generally 0.1µF is more than sufficient. Overcurrent latchoff operation is not always needed or desired. The feature can be overridden by adding a pullup current greater than 5µA to the RUN/SS pin. The additional current prevents the discharge of C SS during a fault and also shortens the soft-start period. Using a resistor to VIN as shown in Figure 5a is simple, but slightly increases shutdown current. Connecting a resistor to INTVCC as shown in Figure 5b eliminates the additional shutdown current, but requires a diode to isolate CSS . Any pull-up network must be able to pull RUN/SS above the 4.5V maximum threshold that arms the latchoff circuit and overcome the 4µA maximum discharge current. INTVCC RSS* VIN 3.3V OR 5V D1 RUN/SS RSS* D2* RUN/SS 2N7002 CSS CSS 3717 F06 *OPTIONAL TO OVERRIDE OVERCURRENT LATCHOFF (5a) (5b) Figure 5. RUN/SS Pin Interfacing with Latchoff Defeated Efficiency Considerations The percent efficiency of a switching regulator is equal to the output power divided by the input power times 100%. It is often useful to analyze individual losses to determine what is limiting the efficiency and which change would produce the most improvement. Although all dissipative elements in the circuit produce losses, four main sources account for most of the losses in LTC3717 circuits: 1. DC I2R losses. These arise from the resistances of the MOSFETs, inductor and PC board traces and cause the efficiency to drop at high output currents. In continuous mode the average output current flows through L, but is chopped between the top and bottom MOSFETs. If the two MOSFETs have approximately the same RDS(ON), then the resistance of one MOSFET can simply be summed with the resistances of L and the board traces to obtain the DC I2R loss. For example, if RDS(ON) = 0.01Ω and RL = 0.005Ω, the loss will range from 15mW to 1.5W as the output current varies from 1A to 10A. 2. Transition loss. This loss arises from the brief amount of time the top MOSFET spends in the saturated region during switch node transitions. It depends upon the input voltage, load current, driver strength and MOSFET capacitance, among other factors. The loss is significant at input voltages above 20V and can be estimated from: Transition Loss ≅ (1.7A–1) VIN2 IOUT CRSS f 3. INTVCC current. This is the sum of the MOSFET driver and control currents. This loss can be reduced by supplying INTVCC current through the EXTVCC pin from a high sn3717 3717fs 13 LTC3717 U W U U APPLICATIO S I FOR ATIO efficiency source, such as an output derived boost network or alternate supply if available. 4. CIN loss. The input capacitor has the difficult job of filtering the large RMS input current to the regulator. It must have a very low ESR to minimize the AC I2R loss and sufficient capacitance to prevent the RMS current from causing additional upstream losses in fuses or batteries. Other losses, including COUT ESR loss, Schottky diode D1 conduction loss during dead time and inductor core loss generally account for less than 2% additional loss. When making adjustments to improve efficiency, the input current is the best indicator of changes in efficiency. If you make a change and the input current decreases, then the efficiency has increased. If there is no change in input current, then there is no change in efficiency. Checking Transient Response The regulator loop response can be checked by looking at the load transient response. Switching regulators take several cycles to respond to a step in load current. When a load step occurs, VOUT immediately shifts by an amount equal to ∆ILOAD (ESR), where ESR is the effective series resistance of COUT. ∆ILOAD also begins to charge or discharge COUT generating a feedback error signal used by the regulator to return VOUT to its steady-state value. During this recovery time, VOUT can be monitored for overshoot or ringing that would indicate a stability problem. The ITH pin external components shown in Figure 6 will provide adequate compensation for most applications. For a detailed explanation of switching control loop theory see Application Note 76. Design Example As a design example, take a supply with the following specifications: VIN = VREF = 2.5V, VEXTVCC = 5V, VOUT = 1.25V ±5%, IOUT(MAX) = 10A, f = 250kHz. First, calculate the timing resistor with VON = VOUT: RON = 1.25V(2.5V – 0.7V) = 514kΩ (0.7V)(250kHz)(10pF )2.5V and choose the inductor for about 40% ripple current at the maximum VIN: L= 1.25V 1.25V 1− = 0.63µH (250kHz)(0.4)(10A) 2.5V Selecting a standard value of 0.68µH results in a maximum ripple current of: ∆IL = 1.25V 1.25V 1– = 3.7A (250kHz)(0.68µH) 2.5V Next, choose the synchronous MOSFET switch. Choosing a Si4874 (RDS(ON) = 0.0083Ω (NOM) 0.010Ω (MAX), θJA = 40°C/W) yields a nominal sense voltage of: VSNS(NOM) = (10A)(1.3)(0.0083Ω) = 108mV Tying VRNG to 1.1V will set the current sense voltage range for a nominal value of 110mV with current limit occurring at 143mV. To check if the current limit is acceptable, assume a junction temperature of about 40°C above a 70°C ambient with ρ110°C = 1.4: ILIMIT ≥ 143mV 1 + (3.7A) = 12.1A (1.4)(0.010Ω) 2 and double check the assumed TJ in the MOSFET: PBOT = 2.5V – 1.25V (12.1A)2 (1.4)(0.010Ω) = 1.02 W 2.5V TJ = 70°C + (1.02W)(40°C/W) = 111°C Because the top MOSFET is on roughly the same amount of time as the bottom MOSFET, the same Si4874 can be used as the synchronous MOSFET. The junction temperatures will be significantly less at nominal current, but this analysis shows that careful attention to heat sinking will be necessary in this circuit. CIN is chosen for an RMS current rating of about 5A at 85°C. The output capacitors are chosen for a low ESR of 0.013Ω to minimize output voltage changes due to inductor ripple current and load steps. For current sinking applications where current flows back to the input through the top transistor, output capacitors with a similar amount of bulk C and ESR should be placed on the input as well. sn3717 3717fs 14 LTC3717 U W U U APPLICATIO S I FOR ATIO (This is typically the case, since VIN is derived from another DC/DC converter.) The ripple voltage will be only: ∆VOUT(RIPPLE) = ∆IL(MAX) (ESR) = (4A) (0.013Ω) = 52mV However, a 0A to 10A load step will cause an output change of up to: ∆VOUT(STEP) = ∆ILOAD (ESR) = (10A) (0.013Ω) = 130mV An optional 22µF ceramic output capacitor is included to minimize the effect of ESL in the output ripple. The complete circuit is shown in Figure 6. PC Board Layout Checklist When laying out a PC board follow one of the two suggested approaches. The simple PC board layout requires a dedicated ground plane layer. Also, for higher currents, it is recommended to use a multilayer board to help with heat sinking power components. 1 R3 11k R4 39k CC1 470pF RPG 100k 2 3 RC 20k 4 CC2 100pF 5 CON 0.01µF 6 7 8 RON 511k (OPT) 0.1µF LTC3717 RUN/SS BOOST PGOOD TG VRNG SW PGND ITH BG SGND ION INTVCC VFB VCC VREF EXTVCC 16 15 • Place CIN, COUT, MOSFETs, D1 and inductor all in one compact area. It may help to have some components on the bottom side of the board. • Place LTC3717 chip with pins 9 to 16 facing the power components. Keep the components connected to pins 1 to 8 close to LTC3717 (noise sensitive components). • Use an immediate via to connect the components to ground plane including SGND and PGND of LTC3717. Use several bigger vias for power components. • Use compact plane for switch node (SW) to improve cooling of the MOSFETs and to keep EMI down. • Use planes for VIN and VOUT to maintain good voltage filtering and to keep power losses low. DB CMDSH-3 CB 0.22µF M1 Si4874 14 13 M2 Si4874 CIN 22µF 6.3V X7R D2 B320A L1 0.68µH D1 B320A + COUT1-2 270µF 2V ×2 + VIN = 2.5V CIN 180µF 4V ×2 VOUT 1.25V ±10A COUT3 22µF 6.3V X7R 12 11 + CSS 0.1µF • The ground plane layer should not have any traces and it should be as close as possible to the layer with power MOSFETs. CVCC 4.7µF 10 RF 1Ω 9 CF 0.1µF VEXT 5V 10Ω 3717 F06a CIN, COUT1-2: CORNELL DUBILIER ESRE181E04B L1: SUMIDA CEP125-0R68MC-H Figure 6. Design Example: 1.25V/±10A at 250kHz sn3717 3717fs 15 LTC3717 U W U U APPLICATIO S I FOR ATIO • Flood all unused areas on all layers with copper. Flooding with copper will reduce the temperature rise of power component. You can connect the copper areas to any DC net (VIN, VOUT, GND or to any other DC rail in your system). • Connect the input capacitor(s) CIN close to the power MOSFETs. This capacitor carries the MOSFET AC current. When laying out a printed circuit board, without a ground plane, use the following checklist to ensure proper operation of the controller. These items are also illustrated in Figure 7. • Connect the INTVCC decoupling capacitor CVCC closely to the INTVCC and PGND pins. • Segregate the signal and power grounds. All small signal components should return to the SGND pin at one point which is then tied to the PGND pin close to the source of M2. • Keep the high dV/dT SW, BOOST and TG nodes away from sensitive small-signal nodes. • Connect the top driver boost capacitor CB closely to the BOOST and SW pins. • Connect the VCC pin decoupling capacitor CF closely to the VCC and PGND pins. • Place M2 as close to the controller as possible, keeping the PGND, BG and SW traces short. CSS 2 3 CC1 CB LTC3717 1 RC 4 CC2 5 RUN/SS BOOST PGOOD TG VRNG SW ITH PGND SGND BG L 16 15 DB 14 + M1 13 12 CION M2 D2 D1 VIN CIN CVCC ION INTVCC VFB VCC CFB 7 8 VREF EXTVCC – 11 + 6 10 9 – VOUT COUT CF + RF RON BOLD LINES INDICATE HIGH CURRENT PATHS 3717F07 Figure 7. LTC3717 Layout Diagram sn3717 3717fs 16 LTC3717 U TYPICAL APPLICATIO S 1.5V/±10A at 300kHz from 5V to 28V Input CSS 0.1µF 1 RR1 11k RR2 39k CC1 680pF RPG 100k 2 3 RC 20k 4 CC2 100pF 5 CON 0.01µF 6 7 8 VREF 3V 10µF 6.3V X7R LTC3717 RUN/SS BOOST PGOOD TG VRNG SW ITH SGND PGND BG ION INTVCC VFB VCC VREF EXTVCC 16 15 DB CMDSH-3 CB 0.22µF M1 IRF7811W 14 B320A L1 1.2µH 13 M2 IRF7822 D1 B320A + VIN CIN 5V TO 28V 10µF 35V VOUT ×3 1.5V ±10A COUT 270µF 2V ×2 12 11 CVCC 4.7µF 10 9 RON 510k COUT: CORNELL DUBILIER ESRE271M02B 3717 TA01 sn3717 3717fs 17 LTC3717 U TYPICAL APPLICATIO S High Voltage Half (VIN) Power Supply CSS 0.1µF LTC3717 1 RUN/SS BOOST RPG 100k 2 CC1 470pF 3 RC 20k 4 CC2 100pF 5 CON 0.01µF 6 7 8 RON 510k R2 1M TG PGOOD SW VRNG PGND ITH BG SGND ION INTVCC VFB VCC VREF EXTVCC 16 15 DB CMDSH-3 CB 0.22µF 14 CIN 10µF 25V ×2 M1 FDS6680S L1 1.8µH 13 M2 FDS6680S + COUT1 270µF 16V VIN 5V TO 25V VOUT VIN/2 ±6A COUT2 10µF 15V 12 11 CVCC 4.7µF 10 RF 1Ω 9 CF 0.1µF R1 2M C2 2200pF 3717 TA02 CIN: TAIYO YUDEN TMK432BJ106MM COUT1: SANYO, OS-CON 16SP270 COUT2: TAIYO YUDEN JMK316BJ106ML L1: TOKO 919AS-1R8N sn3717 3717fs 18 LTC3717 U PACKAGE DESCRIPTIO GN Package 16-Lead Plastic SSOP (Narrow .150 Inch) (Reference LTC DWG # 05-08-1641) .189 – .196* (4.801 – 4.978) .045 ±.005 16 15 14 13 12 11 10 9 .254 MIN .009 (0.229) REF .150 – .165 .229 – .244 (5.817 – 6.198) .0165 ± .0015 .150 – .157** (3.810 – 3.988) .0250 TYP RECOMMENDED SOLDER PAD LAYOUT 1 .015 ± .004 × 45° (0.38 ± 0.10) .007 – .0098 (0.178 – 0.249) 2 3 4 5 6 7 .053 – .068 (1.351 – 1.727) 8 .004 – .0098 (0.102 – 0.249) 0° – 8° TYP .016 – .050 (0.406 – 1.270) NOTE: 1. CONTROLLING DIMENSION: INCHES INCHES 2. DIMENSIONS ARE IN (MILLIMETERS) .008 – .012 (0.203 – 0.305) .0250 (0.635) BSC 3. DRAWING NOT TO SCALE *DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE **DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE GN16 (SSOP) 0502 sn3717 3717fs Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. 19 LTC3717 U TYPICAL APPLICATIO Typical Application 1.25V/±3A at 1.4MHz CSS 0.1µF 1 RPG 100k 2 3 CC1 470pF RC 33k 4 CC2 100pF CON, 0.01µF 5 6 7 8 LTC3717 RUN/SS BOOST PGOOD VRNG ITH SGND TG SW 15 CB 0.22µF + M1 1/2 Si9802 L1 0.7µH BG INTVCC VFB VCC EXTVCC CIN 120µF 4V 14 + PGND ION VREF 16 DB CMDSH-3 13 M2 1/2 Si9802 VIN 2.5V VOUT 1.25V ±3A COUT 120µF 4V 12 11 10 CVCC 4.7µF 5V 9 1µF RON 92k 3717 TA03 CIN, COUT: CORNELL DUBILIER ESRD121M04B L1: TOKO A921CY-0R7M RELATED PARTS PART NUMBER DESCRIPTION COMMENTS TM LTC1625/LTC1775 No RSENSE Current Mode Synchronous Step-Down Controller 97% Efficiency; No Sense Resistor; 99% Duty Cycle LTC1628-PG Dual, 2-Phase Synchronous Step-Down Controller Power Good Output; Minimum Input/Output Capacitors; 3.5V ≤ VIN ≤ 36V LTC1628-SYNC Dual, 2-Phase Synchronous Step-Down Controller Synchronizable 150kHz to 300kHz LTC1709-7 High Efficiency, 2-Phase Synchronous Step-Down Controller with 5-Bit VID Up to 42A Output; 0.925V ≤ VOUT ≤ 2V LTC1709-8 High Efficiency, 2-Phase Synchronous Step-Down Controller Up to 42A Output; VRM 8.4; 1.3V ≤ VOUT ≤ 3.5V LTC1735 High Efficiency, Synchronous Step-Down Controller Burst ModeTM Operation; 16-Pin Narrow SSOP; 3.5V ≤ VIN ≤ 36V LTC1736 High Efficiency, Synchronous Step-Down Controller with 5-Bit VID Mobile VID; 0.925V ≤ VOUT ≤ 2V; 3.5V ≤ VIN ≤ 36V LTC1772 SOT-23 Step-Down Controller Current Mode; 550kHz; Very Small Solution Size LTC1773 Synchronous Step-Down Controller Up to 95% Efficiency, 550kHz, 2.65V ≤ VIN ≤ 8.5V, 0.8V ≤ VOUT ≤ VIN, Synchronizable to 750kHz LTC1778/LTC3778 Wide Operating Range, No RSENSE Step-Down Synchronous Controllers 4V ≤ VIN ≤ 36V, True Current Mode Control, 2% to 90% Duty Cycle LTC1874 Dual, Step-Down Controller Current Mode; 550kHz; Small 16-Pin SSOP, VIN < 9.8V LTC1876 2-Phase, Dual Synchronous Step-Down Controller with Step-Up Regulator 2.6V ≤ VIN ≤ 36V, Power Good Output, 300kHz Operation LTC3413 Monolithic DDR Memory Termination Regulator 90% Efficiency, ±3A Output, 2MHz Operation Burst Mode is a registered trademark of Linear Technology Corporation. sn3717 3717fs 20 Linear Technology Corporation LT/TP 0103 2K • PRINTED IN USA 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408) 432-1900 ● FAX: (408) 434-0507 ● www.linear.com LINEAR TECHNOLOGY CORPORATION 2001