L6725 Low cost adjustable step-down controller Features ■ Input voltage range from 1.8V to 14V ■ Supply voltage range from 4.5V to 14V ■ Adjustable output voltage down to 0.6V with ±0.8% accuracy over line voltage and temperature (0°C~125°C) ■ Fixed frequency voltage mode control ■ 0% to 100% duty cycle ■ External input voltage reference ■ Soft-start and inhibit ■ High current embedded drivers ■ Predictive anti-cross conduction control ■ Programmable high-side and low-side RDS(on) sense over-current-protection Applications ■ Sink current capability ■ Low voltage distributed DC-DC ■ Selectable switching frequency 250KHz/ 500KHz ■ Graphic cards ■ Pre-bias start up capability ■ Over voltage protection ■ Thermal shut-down ■ Package: SO16N SO16N (Narrow) Order Codes June 2006 Part number Package Packing L6725 SO16N Tube L6725TR SO16N Tape & Reel Rev 3 1/27 www.st.com 27 L6725 Contents Contents 1 Summary description . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 1.1 2 Functional description . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 Electrical data . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5 2.1 Maximum rating . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5 2.2 Thermal data . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5 3 Pin connections and functions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 4 Electrical characteristics . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 5 Device description . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10 2/27 5.1 Oscillator . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10 5.2 Internal LDO . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10 5.3 Bypassing the LDO to avoid the voltage drop with low Vcc . . . . . . . . . . . . . 11 5.4 Internal and external references . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11 5.5 Error amplifier . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12 5.6 Soft start . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12 5.7 Driver section . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 14 5.8 Monitoring and protections . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 14 5.9 Hiccup mode . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15 5.10 Thermal shutdown . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15 L6725 6 7 Contents Application details . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16 6.1 Inductor design . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16 6.2 Output capacitors . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17 6.3 Input capacitors . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17 6.4 Compensation network . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18 L6725 demoboard . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20 7.1 Description . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20 8 Package mechanical data . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24 9 Revision history . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 26 3/27 Summary description 1 L6725 Summary description The device is a low cost pwm controller dedicated for low voltage distributed DC-DC. The input voltage can range from 1.8V to 14V, while the supply voltage can range from 4.5V to 14V. The output voltage is adjustable down to 0.6V. High peak current gate drivers provide for fast switching to the external power section, and the output current can be in excess of 20A. The device is capable to manage minimum on-times (TON) shorter than 100ns making possible conversions with very low duty cycle and very high switching frequency. In order to guarantee a real overcurrent protection, also with very narrow TON, the current sense is realized both on the high-side and low-side MOSFETs. When necessary, two different current limit protections can be externally set through an external resistor. The device can sink current after the soft-start phase while, during the soft-start, the sink mode capability is disabled in order to allow a proper start-up also in pre-biased output voltage conditions. Other features are over-voltage-protection and thermal shutdown. 1.1 Figure 1. 4/27 Functional description Block diagram L6725 Electrical data 2 Electrical data 2.1 Maximum rating Table 1. Absolute maximum ratings Symbol Parameter VCC Value Unit -0.3 to 18 V 0 to 6 \ 0 to VBOOT - VPHASE V BOOT -0.3 to 24 V PHASE -1 to 18 VCC to GND and PGND, OCH VBOOT - VPHASE Boot Voltage VHGATE - VPHASE VBOOT VPHASE PHASE Spike, transient < 50ns (FSW = 500KHz) SS, FB, EAREF, OCL, LGATE, COMP, VCCDR OCH Pin OTHER PINS 2.2 Maximum Withstanding Voltage Range Test Condition: CDF-AEC-Q100-002 "Human Body Model" Acceptance Criteria: "Normal Performance" -3 V +24 -0.3 to 6 V ±1500 V ±2000 Thermal data Table 2. Thermal data Symbol Value Unit 50 °C/W Storage temperature range -40 to 150 °C TJ Junction operating temperature range -40 to 125 °C TA Ambient operating temperature range -40 to +85 °C RthJA(1) TSTG Description Max. Thermal Resistance Junction to ambient 1. Package mounted on demoboard 5/27 L6725 Pin connections and functions 3 Pin connections and functions Figure 2. Pins connection (Top view) COMP 1 16 FB SS/INH 2 15 SGND EAREF 3 14 N.C. OCL 4 13 N.C. OCH 5 12 VCC PHASE 6 11 VCCDR HGATE 7 10 BOOT 8 9 LGATE PGND SO16N Table 3. Pin functions Pin n. Name Function 15 SGND 16 FB 1 COMP This pin is connected to the error amplifier output and is used to compensate the voltage control feedback loop. 2 SS/INH The soft-start time is programmed connecting an external capacitor from this pin to GND. The internal current generator forces a current of 10µA through the capacitor. When the voltage at this pin is lower than 0.5V the device is disabled. All the internal references are referred to this pin. This pin is connected to the error amplifier inverting input. Connect it to VOUT through the compensation network. This pin is also used to sense the output voltage in order to manage the over voltage protection. By setting the voltage at this pin is possible to select the internal/external reference and the switching frequency: VEAREF 0-80% of VCCDR -> External Reference/FSW = 250KHz 3 EAREF VEAREF = 80% - 95% of VCCDR -> VREF = 0.6V/FSW = 500KHz VEAREF = 95% - 100% of VCCDR ->VREF = 0.6V/FSW = 250KHz An internal clamp limits the maximum VEAREF at 2.5V (typ.). The device captures the analog value present at this pin at the start-up when VCC meets the UVLO threshold. 6/27 L6725 Table 3. 4 Pin connections and functions Pin functions OCL A resistor connected from this pin to ground sets the valley- current-limit. The valley current is sensed through the low-side MOSFET(s). The internal current generator sources a current of 100µA (IOCL) from this pin to ground through the external resistor (ROCL). The over-current threshold is given by the following equation: I 5 OCH VALLEY = IOCL ⋅ R OCL 2 ⋅ RDSonLS A resistor connected from this pin and the high-side MOSFET(s) drain sets the peakcurrent-limit. The peak current is sensed through the high-side MOSFET(s). The internal 100µA current generator (IOCH) sinks a current from the drain through the external resistor (ROCH). The over-current threshold is given by the following equation: IPEAK = IOCH ⋅ R OCH RDSonHS 6 PHASE This pin is connected to the source of the high-side MOSFET(s) and provides the return path for the high-side driver. This pin monitors the drop across both the upper and lower MOSFET(s) for the current limit together with OCH and OCL. 7 HGATE This pin is connected to the high-side MOSFET(s) gate. 8 BOOT Through this pin is supplied the high-side driver. Connect a capacitor from this pin to the PHASE pin and a diode from VCCDR to this pin (cathode versus BOOT). 9 PGND This pin has to be connected closely to the low-side MOSFET(s) source in order to reduce the noise injection into the device. 10 LGATE This pin is connected to the low-side MOSFET(s) gate. 11 VCCDR 5V internally regulated voltage. It is used to supply the internal drivers. Filter it to ground with a 1uF ceramic cap. 12 VCC Supply voltage pin. The operative supply voltage range is from 4.5V to 14V. 7/27 L6725 Electrical characteristics 4 Electrical characteristics VCC = 12V, TA = 25°C unless otherwise specified. Table 4. Electrical characteristics Symbol Parameter Test Condition Min. Typ. Max. 7 9 8.5 10 Unit VCC SUPPLY CURRENT ICC VCC Stand By current SS to GND VCC quiescent current HG = open, LG = open, PH=open Turn-ON VCC threshold VOCH = 1.7V 4.0 4.2 4.4 V Turn-OFF VCC threshold VOCH = 1.7V 3.6 3.8 4.0 V Turn-ON VOCH threshold 1.1 1.25 1.47 V Turn-OFF VOCH threshold 0.9 1.05 1.27 V 4.5 5 5.5 V SS = 2V 7 10 13 SS = 0 to 0.5V 20 30 45 237 250 263 KHz 450 500 550 KHz mA Power-ON VCC VIN OK VCCDR Regulation VCCDR voltage VCC =5.5V to 14V IDR = 1mA to 100mA Soft Start and Inhibit ISS Soft Start Current µA Oscillator fOSC ∆VOSC Accuracy Ramp Amplitude 2.1 V Output Voltage VFB 8/27 Output Voltage 0.597 0.6 0.603 V L6725 Table 4. Electrical characteristics Electrical characteristics Symbol Parameter Test Condition Min. Typ. Max. Unit 70 100 150 kΩ 0.290 0.5 µA Error Amplifier REAREF IFB EAREF Input Resistance Vs. GND I.I. bias current VFΒ = 0V Ext Ref Clamp VOFFSET 2.3 V Error amplifier offset Vref = 0.6V GV Open Loop Voltage Gain Guaranteed by design 100 dB GBWP Gain-Bandwidth Product Guaranteed by design 10 MHz Slew-Rate COMP = 10pF Guaranteed by design 5 V/µs High Side Source Resistance VBOOT - VPHASE = 5V 1.7 Ω RHGATE_OFF High Side Sink Resistance VBOOT - VPHASE = 5V 1.12 Ω RLGATE_ON VCCDR = 5V 1.15 Ω VCCDR = 5V 0.6 Ω SR -5 +5 mV Gate Drivers RHGATE_ON Low Side Source Resistance RLGATE_OFF Low Side Sink Resistance Protections IOCH OCH Current Source IOCL OCL Current Source VOCH = 1.7V VFB Rising OVP Over Voltage Trip (VFB / VEAREF) VEAREF = 0.6V VFB Falling VEAREF = 0.6V 90 100 110 µΑ 90 100 110 µΑ 120 % 117 % 9/27 L6725 5 Device description Device description The controller provides complete control logic and protection for flexible and cost-effective DCDC converters. It is designed to drive N-Channel MOSFETs in a synchronous rectified buck topology. The output voltage of the converter can be precisely regulated down to 600mV with a maximum tolerance of ±0.8%, when the internal reference is used. The device allows also using an external reference (0V to 2.5V) for the regulation. The device provides voltage-mode control. The switching frequency can be set at two different values: 250KHz or 500KHz. The error amplifier features a 10MHz gain-bandwidth-product and 5V/µs slew-rate that permits to realize high converter bandwidth for fast transient response. The PWM duty cycle can range from 0% to 100%. The device protects against over current conditions providing a constantcurrent-protection during the soft-start phase and entering in HICCUP mode in all the other conditions. The device monitors the current by using the RDS(ON) of both the high-side and lowside MOSFET(s), eliminating the need for a current sensing resistor and guaranteeing an effective over-current-protection in all the application conditions. Other features are overvoltage-protection and thermal shutdown. The device is available in SO16N package. 5.1 Oscillator The switching frequency can be fixed to two values: 250KHz or 500KHz by setting the proper voltage at the EAREF pin (see Table 3. Pins function and section 4.3 Internal and external reference). 5.2 Internal LDO An internal LDO supplies the internal circuitry of the device. The input of this stage is the VCC pin and the output (5V) is the VCCDR pin. The LDO can be by-passed, providing directly a 5V voltage to VCCDR. In this case VCC and VCCDR pins must be shorted together as shown in Figure 3. VCCDR pin must be filtered with a 1uF capacitor to sustain the internal LDO during the recharge of the bootstrap capacitor. 10/27 L6725 5.3 Device description Bypassing the LDO to avoid the voltage drop with low Vcc If VCC≈ 5V the internal LDO works in dropout with an output resistance of about 1Ω. The maximum LDO output current is about 100mA and so the output voltage drop is 100mV, to avoid this the LDO can be bypassed. Figure 3. 5.4 Bypassing the LDO Internal and external references It is possible to set the internal/external reference and the switching frequency by setting the proper voltage at the EAREF pin. The maximum value of the external reference depends on the VCC: with VCC = 4V the clamp operates at about 2V (typ.), while with VCC greater than 5V the maximum external reference is 2.5V (typ.). ● VEAREF from 0% to 80% of VCCDR -> External reference/Fsw=250KHz ● VEAREF from 80% to 95% of VCCDR -> VREF = 0.6V/Fsw=500KHz ● VEAREF from 95% to 100% of VCCDR -> VREF = 0.6V/Fsw=250KHz Providing an external reference from 0V to 450mV the output voltage will be regulated but some restrictions must be considered: ● The minimum OVP threshold is set at 300mV; ● The under-voltage-protection doesn't work; To set the resistor divider it must be considered that a 100K pull-down resistor is integrated into the device (see Figure 4.). Finally it must be taken into account that the voltage at the EAREF pin is captured by the device at the start-up when VCC is about 4V. 11/27 L6725 Device description 5.5 Figure 4. 5.6 Error amplifier Error amplifier reference Soft start When both VCC and VIN are above their turn-ON thresholds (VIN is monitored by the OCH pin) the start-up phase takes place. Otherwise the SS pin is internally shorted to GND. At start-up, a ramp is generated charging the external capacitor CSS with an internal current generator. The initial value for this current is 35µA and charges the capacitor up to 0.5V. After that it becomes 10µA until the final charge value of approximately 4V (see Figure 5.). Figure 5. Soft-Start phase. Vcc Vin 4.2V 1.25V t Vss 4V 0.5V t 12/27 L6725 Device description The output of the error amplifier is clamped with this voltage (VSS) until it reaches the programmed value. No switching activity is observable if VSS is lower than 0.5V and both MOSFETs are OFF. When VSS is between 0.5V and 1.1V the low-side MOSFET is turned ON. As VSS reaches 1.1V (i.e. the oscillator triangular wave inferior limit) even the high-side MOSFET begins to switch and the output voltage starts to increase. During the soft-start phase the current can’t be reversed in order to allow pre-biased start-up (see Figure 6. and Figure 7.). Figure 6. Start-up without pre-bias LGate VOUT IL VSS Figure 7. Start-up with pre-bias LGate VOUT IL VSS If an over current is detected during the soft-start phase, the device provides a constantcurrent-protection. In this way, in case of short soft-start time and/or small inductor value and/or high output capacitors value, the converter can start in any case, limiting the current (Chapter 5.8: Monitoring and protections on page 14). The soft-start phase ends when Vss reaches 3.5V. After that the over-current-protection triggers the HICCUP mode. 13/27 L6725 Device description 5.7 Driver section The high-side and low-side drivers allow using different types of power MOSFETs (also multiple MOSFETs to reduce the RDSON), maintaining fast switching transitions. The low-side driver is supplied by VCCDR while the high-side driver is supplied by the BOOT pin. A predictive dead time control avoids MOSFETs cross-conduction maintaining very short dead time duration in the range of 20ns. The control monitors the phase node in order to sense the low-side body diode recirculation. If the phase node voltage is less than a certain threshold (-350mV typ.) during the dead time, it will be reduced in the next PWM cycle. The predictive dead time control doesn’t work when the high-side body diode is conducting because the phase node doesn’t go negative. This situation happens when the converter is sinking current for example and, in this case, an adaptive dead time control operates. 5.8 Monitoring and protections The output voltage is monitored by means of pin FB. The device provides over-voltageprotection: when the voltage sensed on FB pin reaches a value 20% (typ.) greater than the reference the low-side driver is turned on as long as the over voltage is detected (see Figure 8). Figure 8. OVP LGate FB The device realizes the over-current-protection (OCP) sensing the current both on the high-side MOSFET(s) and the low-side MOSFET(s) and so 2 current limit thresholds can be set (see OCH pin and OCL pin in Table 3: Pin functions): ● Peak Current Limit ● Valley Current Limit The Peak Current Protection is active when the high-side MOSFET(s) is turned on, after a masking time of 100ns. The valley-current-protection is enabled when the low-side MOSFET(s) is turned on after a masking time of 500ns. If, when the soft-start phase is completed, an over current event occurs during the on time (peak-current-protection) or during the off time (valleycurrent-protection) the device enters in HICCUP mode: the high-side and low-side MOSFET(s) are turned off, the soft-start capacitor is discharged with a constant current of 10µA and when the voltage at the SS pin reaches 0.5V the soft-start phase restarts (see Figure 9). 14/27 L6725 5.9 Figure 9. Device description Hiccup mode Constant current and Hiccup Mode during an OCP. VSS VCOMP IL During the soft-start phase the OCP provides a constant-current-protection. If during the TON the OCH comparator triggers an over current the high-side MOSFET(s) is immediately turned OFF (after the masking time and the internal delay) and returned on at the next pwm cycle. The limit of this protection is that the TON cannot be less than masking time plus propagation delay, because during the masking time the peak-current-protection is disabled. In case of very hard short circuit, even with this short TON, the current could escalate. The valley-current-protection is very helpful in this case to limit the current. If during the off-time the OCL comparator triggers an over current, the high-side MOSFET(s) is not turned on until the current is over the valleycurrent-limit. This implies that, if it is necessary, some pulses of the high-side MOSFET(s) will be skipped, guaranteeing a maximum current due to the following formula: I MAX = IVALLEY + 5.10 Vin − Vout ⋅ TON , MIN L (1) Thermal shutdown When the junction temperature reaches 150°C ±10°C the device enters in thermal shutdown. Both MOSFETs are turned OFF and the soft-start capacitor is rapidly discharged with an internal switch. The device does not restart until the junction temperature goes down to 120°C and, in any case, until the voltage at the soft-start pin reaches 500mV. 15/27 L6725 Application details 6 Application details 6.1 Inductor design The inductance value is defined by a compromise between the transient response time, the efficiency, the cost and the size. The inductor has to be calculated to sustain the output and the input voltage variation to maintain the ripple current (∆IL) between 20% and 30% of the maximum output current. The inductance value can be calculated with the following relationship: L≅ Vin − Vout Vout ⋅ Fsw ⋅ ∆I L Vin (2) Where FSW is the switching frequency, VIN is the input voltage and VOUT is the output voltage. Figure 10 shows the ripple current vs. the output voltage for different values of the inductor, with VIN = 5V and VIN = 12V at a switching frequency of 500KHz. INDUCT O R CURRE NT RIP P L Figure 10. Inductor current ripple. 8 7 V in = 1 2 V , L = 1 u H 6 5 4 3 V in = 1 2 V , L = 2 u H 2 V in = 5 V , L = 5 0 0 n H 1 V in = 5 V , L = 1 .5 u H 0 0 1 2 3 4 O UT P UT V O L T AG E (V ) Increasing the value of the inductance reduces the ripple current but, at the same time, increases the converter response time to a load transient. If the compensation network is well designed, during a load transient the device is able to set the duty cycle to 100% or to 0%. When one of these conditions is reached, the response time is limited by the time required to change the inductor current. During this time the output current is supplied by the output capacitors. Minimizing the response time can minimize the output capacitor size. 16/27 L6725 6.2 Application details Output capacitors The output capacitors are basic components for the fast transient response of the power supply. For example, during a positive load transient, they supply the current to the load until the converter reacts. The controller recognizes immediately the load transient and sets the duty cycle at 100%, but the current slope is limited by the inductor value. The output voltage has a first drop due to the current variation inside the capacitor (neglecting the effect of the ESL): ∆Vout ESR = ∆Iout ⋅ ESR (3) Moreover, there is an additional drop due to the effective capacitor discharge that is given by: ∆VoutCOUT = ∆Iout 2 ⋅ L 2 ⋅ Cout ⋅ (Vin, min⋅ D max − Vout ) (4) Where DMAX is the maximum duty cycle value that in the L6725 is 100%. Usually the voltage drop due to the ESR is the biggest one while the drop due to the capacitor discharge is almost negligible. Moreover the ESR value also affects the voltage static ripple, that is: ∆Vout = ESR ⋅ ∆I L 6.3 (5) Input capacitors The input capacitors have to sustain the RMS current flowing through them, that is: Irms = Iout ⋅ D ⋅ (1 − D) (6) Where D is the duty cycle. The equation reaches its maximum value, IOUT /2 with D = 0.5. The losses in worst case are: P = ESR ⋅ (0.5 ⋅ Iout ) 2 (7) 17/27 L6725 Application details 6.4 Compensation network The loop is based on a voltage mode control (Figure 11). The output voltage is regulated to the internal/external reference voltage and scaled by the external resistor divider. The error amplifier output VCOMP is then compared with the oscillator triangular waveform to provide a pulse-width modulated (PWM) with an amplitude of VIN at the PHASE node. This waveform is filtered by the output filter. The modulator transfer function is the small signal transfer function of VOUT/VCOMP. This function has a double pole at frequency FLC depending on the L-COUT resonance and a zero at FESR depending on the output capacitor’s ESR. The DC Gain of the modulator is simply the input voltage VIN divided by the peak-to-peak oscillator voltage: VOSC. Figure 11. Compensation Network The compensation network consists in the internal error amplifier, the impedance networks ZIN (R3, R4 and C20) and ZFB (R5, C18 and C19). The compensation network has to provide a closed loop transfer function with the highest 0dB crossing frequency to have fastest transient response (but always lower than fSW/10) and the highest gain in DC conditions to minimize the load regulation error. A stable control loop has a gain crossing the 0dB axis with -20dB/decade slope and a phase margin greater than 45°. To locate poles and zeroes of the compensation networks, the following suggestions may be used: ● Modulator singularity frequencies: ω LC = ● (8) ω ESR = 1 ESR ⋅ Cout (9) Compensation network singularity frequencies: ω P1 = ωZ 1 = 18/27 1 L ⋅ Cout 1 (10) ⎛ C18 ⋅ C19 ⎞ ⎟⎟ R5 ⋅ ⎜⎜ ⎝ C18 + C19 ⎠ 1 R5 ⋅ C19 (12) ωP 2 = ωZ 2 = 1 R4 ⋅ C20 1 C20 ⋅ (R3 + R4 ) (11) (13) L6725 Application details ● Compensation network design: – Put the gain R5/R3 in order to obtain the desired converter bandwidth: ϖC = R5 Vin ⋅ ⋅ϖ LC (14) R3 ∆Vosc – Place ωZ1 before the output filter resonance ωLC; – Place ωZ2 at the output filter resonance ωLC; – Place ωP1 at the output capacitor ESR zero ωESR; – Place ωP2 at one half of the switching frequency; – Check the loop gain considering the error amplifier open loop gain. Figure 12. Asymptotic Bode plot of Converter's open loop gain 19/27 L6725 L6725 demoboard 7 L6725 demoboard 7.1 Description L6725 demoboard realizes in a four layer PCB a step-down DC/DC converter and shows the operation of the device in a general purpose application. The input voltage can range from 4.5V to 14V and the output voltage is at 3.3V. The module can deliver an output current in excess of 20A. The switching frequency is set at 250 KHz (controller free-running FSW) but it can be set to 500KHz acting on the EAREF pin. Figure 13. Demoboard schematic VIN R9 D1 J1 C11 C12-C13 C10 R5 VCCDR GIN R6 C9 BOOT 8 11 EAREF J2 OCH R11 5 HGATE 7 3 L1 Q4-6 EXT REF VCC VCC C8 R8 C7 GND PHASE 6 C5 LGATE 10 12 U1 L6725 15 VOUT R10 R12 D3 C15 Q1-3 9 PGND C16-C19 GOUT SS 2 C4 4 OCL 1 16 VFB COMP R3 R7 C2 R4 R1 C1 C3 R2 Table 5. 20/27 Demoboard part list Reference Value Manufacturer Package Supplier R1 1kΩ Neohm SMD 0603 IFARCAD R2 1kΩ Neohm SMD 0603 IFARCAD R3 4K7 R4 2k7 Neohm SMD 0603 IFARCAD R5 0Ω Neohm SMD 0603 IFARCAD R6 N.C. Neohm SMD 0603 IFARCAD R7 2K Neohm SMD 0603 IFARCAD R8 10Ω Neohm SMD 0603 IFARCAD R9 1K5 Neohm SMD 0603 IFARCAD R10 2.2Ω Neohm SMD 0603 IFARCAD L6725 L6725 demoboard Table 5. Demoboard part list R11 2.2Ω Neohm SMD 0603 IFARCAD R12 N.C. Neohm SMD 0603 IFARCAD C1 4.7nF Kemet SMD 0603 IFARCAD C2 47nF Kemet SMD 0603 IFARCAD C3 1nF Kemet SMD 0603 IFARCAD C4 100nF Kemet SMD 0603 IFARCAD C5 100nF Kemet SMD 0603 IFARCAD C6 N.C. / / / C7 100nF Kemet SMD 0603 IFARCAD C8 4.7uF 20V AVX SMA6032 IFARCAD C9 1nF Kemet SMD 0603 IFARCAD C10 1uF Kemet SMD 0603 IFARCAD C11 220nF Kemet SMD 0603 IFARCAD C12-13 3X 15uF / / ST (TDK) C15 N.C. / / / C16-19 2X 330µF / / ST (poscap) L1 1.8µH Panasonic SMD ST D1 STPS1L30M ST DO216AA ST D3 N.C. / / / Q1-Q2 STS12NH3LL ST SO8 ST Q4-Q5 STS25NH3LL ST SO8 ST U1 L6725 ST SO16N ST Table 6. Other inductor manufacturer Manufacturer Series Inductor Value (µH) Saturation Current (A) WURTH ELEKTRONIC 744318180 1.8 20 SUMIDA CDEP134-2R7MC-H 2.7 15 EPCOS HPI_13 T640 1.4 22 TDK SPM12550T-1R0M220 1 22 TOKO FDA1254 2.2 14 HCF1305-1R0 1.15 22 HC5-1R0 1.3 27 COILTRONICS 21/27 L6725 L6725 demoboard Table 7. Other capacitor manufacturer Manufacturer Series Capacitor value(µF) Rated voltage (V) C4532X5R1E156M 15 25 C3225X5R0J107M 100 6.3 NIPPON CHEMI-CON 25PS100MJ12 100 25 PANASONIC ECJ4YB0J107M 100 6.3 TDK Figure 14. Demoboard efficiency Fs F w ==4500KHz 00K H z SW E F F IC IE N 95.00% 90.00% VIN = 5V 85.00% 80.00% VIN = 12V 75.00% 1 3 5 7 9 I o u t (A ) Figure 15. PCB Layout: Top Layer L6725 Figure 16. PCB Layout: Power Ground Layer 22/27 11 13 15 L6725 L6725 demoboard Figure 17. PCB Layout: Signal-Ground Layer Figure 18. PCB Layout: Bottom Layer 23/27 Package mechanical data 8 L6725 Package mechanical data In order to meet environmental requirements, ST offers these devices in ECOPACK® packages. These packages have a Lead-free second level interconnect . The category of second Level Interconnect is marked on the package and on the inner box label, in compliance with JEDEC Standard JESD97. The maximum ratings related to soldering conditions are also marked on the inner box label. ECOPACK is an ST trademark. ECOPACK specifications are available at: www.st.com. 24/27 L6725 Package mechanical data Table 8. SO16N mechanical data Dim. mm MIN. TYP. A a1 inch MAX. MIN. TYP. 1.75 0.1 0.069 0.25 a2 0.004 0.009 1.6 b 0.35 b1 0.19 C 0.063 0.46 0.014 0.25 0.007 0.5 c1 MAX. 0.018 0.010 0.020 45° (typ.) D(1) 9.8 10 0.386 0.394 E 5.8 6.2 0.228 0.244 e 1.27 0.050 e3 8.89 0.350 (1) 3.8 4.0 0.150 0.157 G 4.60 5.30 0.181 0.208 L 0.4 1.27 0.150 0.050 F M S 0.62 0.024 8 °(max.) 1. "D" and "F" do not include mold flash or protrusions -Mold flash or protrusions shall not exceed 0.15mm (.006inc.) Figure 19. Package dimensions 25/27 L6725 Revision history 9 Revision history Table 9. 26/27 Revision history Date Revision Changes 20-Dec-2005 1 Initial release. 30-May-2006 2 New template, thermal data updated 26-Jun-2006 3 Note page 5 deleted L6725 Please Read Carefully: Information in this document is provided solely in connection with ST products. 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