LINER LT1720IMS8

LT1720/LT1721
Dual/Quad, 4.5ns, Single
Supply 3V/5V Comparators
with Rail-to-Rail Outputs
DESCRIPTION
FEATURES
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UltraFast: 4.5ns at 20mV Overdrive
7ns at 5mV Overdrive
Low Power: 4mA per Comparator
Optimized for 3V and 5V Operation
Pinout Optimized for High Speed Ease of Use
Input Voltage Range Extends 100mV
Below Negative Rail
TTL/CMOS Compatible Rail-to-Rail Outputs
Internal Hysteresis with Specified Limits
Low Dynamic Current Drain; 15μA/(V-MHz),
Dominated by Load In Most Circuits
Tiny 3mm × 3mm × 0.75mm DFN Package (LT1720)
APPLICATIONS
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High Speed Differential Line Receiver
Crystal Oscillator Circuits
Window Comparators
Threshold Detectors/Discriminators
Pulse Stretchers
Zero-Crossing Detectors
High Speed Sampling Circuits
The LT®1720/LT1721 are UltraFastTM dual/quad comparators optimized for single supply operation, with a supply
voltage range of 2.7V to 6V. The input voltage range extends
from 100mV below ground to 1.2V below the supply voltage. Internal hysteresis makes the LT1720/LT1721 easy to
use even with slow moving input signals. The rail-to-rail
outputs directly interface to TTL and CMOS. Alternatively,
the symmetric output drive can be harnessed for analog
applications or for easy translation to other single supply
logic levels.
The LT1720 is available in three 8-pin packages; three pins
per comparator plus power and ground. In addition to SO
and MSOP packages, a 3mm × 3mm low profile (0.8mm)
dual fine pitch leadless package (DFN) is available for space
limited applications. The LT1721 is available in the 16-pin
SSOP and S packages.
The pinouts of the LT1720/LT1721 minimize parasitic
effects by placing the most sensitive inputs (inverting)
away from the outputs, shielded by the power rails. The
LT1720/LT1721 are ideal for systems where small size and
low power are paramount.
L, LT, LTC and LTM are registered trademarks of Linear Technology Corporation. UltaFast is
a trademark of Linear Technology Corporation. All other trademarks are the property of their
respective owners.
TYPICAL APPLICATION
Propagation Delay vs Overdrive
2.7V to 6V Crystal Oscillator with TTL/CMOS Output
8
2.7V TO 6V
7
1MHz TO 10MHz
CRYSTAL (AT-CUT)
2k
+
DELAY (ns)
620Ω
GROUND
CASE
C1
1/2 LT1720
RISING EDGE
(tPDLH)
6
220Ω
25°C
VSTEP = 100mV
VCC = 5V
CLOAD = 10pF
5
4
FALLING EDGE
(tPDHL)
3
OUTPUT
2
–
1
2k
17201 TA01
0
0.1μF
1.8k
0
10
20
30
OVERDRIVE (mV)
40
50
17201 TA02
17201fc
1
LT1720/LT1721
ABSOLUTE MAXIMUM RATINGS
(Note 1)
Supply Voltage, VCC to GND ........................................7V
Input Current....................................................... ±10mA
Output Current (Continuous) ............................. ±20mA
Junction Temperature .......................................... 150°C
(DD Package) .................................................... 125°C
Lead Temperature (Soldering, 10 sec) .................. 300°C
Storage Temperature Range................... –65°C to 150°C
(DD Package) ..................................... –65°C to 125°C
Operating Temperature Range
C Grade ................................................... 0°C to 70°C
I Grade ............................................... –40°C to 85°C
PIN CONFIGURATION
TOP VIEW
+IN A
1
8
VCC
–IN A
2
7
OUT A
–IN B
3
6
OUT B
+IN B
4
5
GND
9
TOP VIEW
+IN A
–IN A
–IN B
+IN B
8
7
6
5
1
2
3
4
VCC
OUT A
OUT B
GND
MS8 PACKAGE
8-LEAD PLASTIC MSOP
DD PACKAGE
8-LEAD (3mm s 3mm) PLASTIC DFN
TJMAX = 150°C, θJA = 230°C/W
TJMAX = 125°C, θJA = 160°C/W
UNDERSIDE METAL INTERNALLY
CONNECTED TO GND
TOP VIEW
TOP VIEW
+IN A 1
8
VCC
–IN A 2
7
OUT A
–IN B 3
6
OUT B
+IN B 4
5
GND
S8 PACKAGE
8-LEAD PLASTIC SO
TJMAX = 150°C, θJA = 200°C/W
–IN A
1
16 –IN D
+IN A
2
15 +IN D
GND
3
14 VCC
OUT A
4
13 OUT D
OUT B
5
12 OUT C
GND
6
11 VCC
+IN B
7
10 +IN C
–IN B
8
9
GN PACKAGE
16-LEAD NARROW
PLASTIC SSOP
–IN C
S PACKAGE
16-LEAD PLASTIC SO
TJMAX = 150°C, θJA = 135°C/W (GN)
TJMAX = 150°C, θJA = 115°C/W (S)
17201fc
2
LT1720/LT1721
ORDER INFORMATION
LEAD FREE FINISH
TAPE AND REEL
PART MARKING*
PACKAGE DESCRIPTION
TEMPERATURE RANGE
LT1720CDD#PBF
LT1720CDD#TRPBF
LAAV
8-Lead (3mm × 3mm) Plastic DFN
0°C to 70°C
LT1720IDD#PBF
LT1720IDD#TRPBF
LAAV
8-Lead (3mm × 3mm) Plastic DFN
–40°C to 85°C
LT1720CMS8#PBF
LT1720CMS8#TRPBF
LTDS
8-Lead Plastic MSOP
0°C to 70°C
LT1720IMS8#PBF
LT1720IMS8#TRPBF
LTACW
8-Lead Plastic MSOP
–40°C to 85°C
LT1720CS8#PBF
LT1720CS8#TRPBF
1720
8-Lead Plastic SO
0°C to 70°C
LT1720IS8#PBF
LT1720IS8#TRPBF
1720I
8-Lead Plastic SO
–40°C to 85°C
LT1721CGN#PBF
LT1721CGN#TRPBF
1721
16-Lead Narrow Plastic SSOP
0°C to 70°C
LT1721IGN#PBF
LT1721IGN#TRPBF
1721I
16-Lead Narrow Plastic SSOP
–40°C to 85°C
LT1721CS#PBF
LT1721CS#TRPBF
1721
16-Lead Plastic SO
0°C to 70°C
LT1721IS#PBF
LT1721IS#TRPBF
1721I
16-Lead Plastic SO
–40°C to 85°C
Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container.
Consult LTC Marketing for information on non-standard lead based finish parts.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
ELECTRICAL CHARACTERISTICS
The l denotes specifications that apply over the full operating temperature
range, otherwise specifications are at TA = 25°C. VCC = 5V, VCM = 1V, COUT = 10pF, VOVERDRIVE = 20mV, unless otherwise specified.
SYMBOL
PARAMETER
CONDITIONS
VCC
Supply Voltage
ICC
Supply Current (Per Comparator)
VCC = 5V
VCC = 3V
l
l
VCMR
Common Mode Voltage Range
(Note 2)
l
–0.1
VCC – 1.2
VTRIP+
Input Trip Points
(Note 3)
l
–2.0
–3.0
5.5
6.5
mV
mV
VTRIP–
Input Trip Points
(Note 3)
l
–5.5
–6.5
2.0
3.0
mV
mV
VOS
Input Offset Voltage
(Note 3)
1.0
3.0
4.5
mV
mV
VHYST
Input Hysteresis Voltage
(Note 3)
3.5
7.0
mV
ΔVOS/ΔT
Input Offset Voltage Drift
l
IB
Input Bias Current
l
IOS
Input Offset Current
l
CMRR
Common Mode Rejection Ratio
(Note 4)
l
55
70
dB
PSRR
Power Supply Rejection Ratio
(Note 5)
l
65
80
dB
AV
Voltage Gain
(Note 6)
Output High Voltage
l
VOL
Output Low Voltage
ISOURCE = 4mA, VIN = VTRIP+ + 10mV
ISINK = 10mA, VIN = VTRIP– – 10mV
tPD20
Propagation Delay
VOVERDRIVE = 20mV (Note 7)
tPD5
Propagation Delay
VOVERDRIVE = 5mV (Notes 7, 8)
VOH
MIN
2.7
4
3.5
l
l
TYP
2.0
MAX
6
V
7
6
mA
mA
10
–6
UNITS
V
μV/°C
0
0.6
μA
μA
∞
l
VCC – 0.4
V
l
l
l
0.4
V
4.5
6.5
8.0
ns
ns
7
10
13
ns
ns
17201fc
3
LT1720/LT1721
ELECTRICAL CHARACTERISTICS
The l denotes specifications that apply over the full operating temperature
range, otherwise specifications are at TA = 25°C. VCC = 5V, VCM = 1V, COUT = 10pF, VOVERDRIVE = 20mV, unless otherwise specified.
SYMBOL
PARAMETER
CONDITIONS
ΔtPD
Differential Propagation Delay
(Note 9) Between Channels
0.3
1.0
ns
tSKEW
Propagation Delay Skew
(Note 10) Between tPDLH/tPDHL
0.5
1.5
ns
tr
Output Rise Time
10% to 90%
2.5
ns
tf
Output Fall Time
90% to 10%
2.2
ns
tJITTER
Output Timing Jitter
VIN = 1.2VP-P (6dBm), ZIN = 50Ω
VCM = 2V, f = 20MHz
15
11
psRMS
psRMS
fMAX
Maximum Toggle Frequency
VOVERDRIVE = 50mV, VCC = 3V
VOVERDRIVE = 50mV, VCC = 5V
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: If one input is within these common mode limits, the other input
can go outside the common mode limits and the output will be valid.
Note 3: The LT1720/LT1721 comparators include internal hysteresis.
The trip points are the input voltage needed to change the output state in
each direction. The offset voltage is defined as the average of VTRIP+ and
VTRIP–, while the hysteresis voltage is the difference of these two.
Note 4: The common mode rejection ratio is measured with VCC = 5V
and is defined as the change in offset voltage measured from VCM = –0.1V
to VCM = 3.8V, divided by 3.9V.
Note 5: The power supply rejection ratio is measured with VCM = 1V and
is defined as the change in offset voltage measured from VCC = 2.7V to
VCC = 6V, divided by 3.3V.
MIN
tPDLH
tPDHL
TYP
MAX
UNITS
70.0
62.5
MHz
MHz
Note 6: Because of internal hysteresis, there is no small-signal region
in which to measure gain. Proper operation of internal circuity is ensured
by measuring VOH and VOL with only 10mV of overdrive.
Note 7: Propagation delay measurements made with 100mV steps.
Overdrive is measured relative to VTRIP±.
Note 8: tPD cannot be measured in automatic handling equipment with
low values of overdrive. The LT1720/LT1721 are 100% tested with a
100mV step and 20mV overdrive. Correlation tests have shown that
tPD limits can be guaranteed with this test, if additional DC tests are
performed to guarantee that all internal bias conditions are correct.
Note 9: Differential propagation delay is defined as the larger of the two:
ΔtPDLH = tPDLH(MAX) – tPDLH(MIN)
ΔtPDHL = tPDHL(MAX) – tPDHL(MIN)
where (MAX) and (MIN) denote the maximum and minimum values
of a given measurement across the different comparator channels.
Note 10: Propagation Delay Skew is defined as:
tSKEW = |tPDLH – tPDHL|
TYPICAL PERFORMANCE CHARACTERISTICS
Input Offset and Trip Voltages
vs Supply Voltage
Input Offset and Trip Voltages
vs Temperature
1
VOS
0
–1
–3
2.5
VTRIP–
25°C
VCM = 1V
3.0
5.0
3.5 4.0 4.5
SUPPLY VOLTAGE (V)
5.5
6.0
17201 G01
VCC = 5V
+
VTRIP
2
COMMON MODE INPUT VOLTAGE (V)
2
–2
4.2
3
VTRIP+
VOS AND TRIP POINT VOLTAGE (mV)
VOS AND TRIP POINT VOLTAGE (mV)
3
Input Common Mode Limits
vs Temperature
1
VOS
0
–1
VTRIP–
–2
–3
–50
–25
25
0
50
75
TEMPERATURE (°C)
100
125
17201 G02
4.0
3.8
3.6
0.2
0
–0.2
–0.4
–50 –25
50
25
75
0
TEMPERATURE (°C)
100
125
17201 G03
17201fc
4
LT1720/LT1721
2
25°C
VCC = 5V
–2
–3
–4
–5
–6
–7
–5 –4 –3 –2 –1 0 1 2 3 4
DIFFERENTIAL INPUT VOLTAGE (V)
5
6.0
5.5
5.0
4.5
VCC = 5V
4.0
VCC = 3V
3.5
3.0
2.5
2.0
–50 –25
75
50
25
TEMPERATURE (˚C)
0
100
Propagation Delay
vs Load Capacitance
7
RISING EDGE
(tPDLH)
DELAY (ns)
6
FALLING EDGE
(tPDHL)
5
4
3
2
VCC = 3V
7.0
VCC = 5V
6.5
OVERDRIVE = 5mV
6.0
5.5
1
4
3
2
5
SUPPLY VOLTAGE (V)
1
5.0
25°C
VSTEP = 100mV
OVERDRIVE = 20mV
CLOAD = 10pF
RISING EDGE
(tPDLH)
4.5
VCC = 5V
FALLING EDGE
(tPDHL)
OVERDRIVE = 20mV
0
10
20
40
30
OUTPUT LOAD CAPACITANCE (pF)
VCC = 3V
4.0
75
0
50
–50 –25
25
TEMPERATURE (°C)
50
100
OUTPUT VOLTAGE RELATIVE TO VCC (V)
0.4
125°C
125°C
VCC = 2.7V
25°C
0.3
–55°C
0.2
0
4
12
16
8
OUTPUT SINK CURRENT (mA)
20
17201 G10
5.0
3.5 4.0 4.5
SUPPLY VOLTAGE (V)
3.0
5.5
Supply Current vs Frequency
VCC = 5V
VCM = 1V
VIN = 15mV
125°C
–0.2
–55°C
25°C
–0.4
–0.6
–0.8
25°C
VCC = 2.7V
–1.0
0
6.0
17201 G09
Output High Voltage
vs Load Current
0.0
VCC = 5V
VCM = 1V
VIN = –15mV
4.0
2.5
125
17201 G08
0.5
7
6
17201 G06
5.0
Output Low Voltage
vs Load Current
OUTPUT VOLTAGE (V)
2
0
tPDLH
VCM = 1V
VSTEP = 100mV
CLOAD = 10pF
17201 G07
0.1
–55°C
3
4.5
1
0
25°C
4
Propagation Delay
vs Supply Voltage
7.5
PROPAGATION DELAY (ns)
8
5
Propagation Delay
vs Temperature
8.0
25°C
VSTEP = 100mV
OVERDRIVE = 20mV
VCC = 5V
125°C
6
17201 G05
17201 G04
9
7
0
125
SUPPLY CURRENT PER COMPARATOR (mA)
INPUT CURRENT (μA)
0
–1
Quiescent Supply Current
vs Supply Voltage
DELAY (ns)
1
Quiescent Supply Current
vs Temperature
SUPPLY CURRENT PER COMPARATOR (mA)
Input Current
vs Differential Input Voltage
QUIESCENT SUPPLY CURRENT PER COMPARATOR (mA)
TYPICAL PERFORMANCE CHARACTERISTICS
4
12
16
8
OUTPUT SOURCE CURRENT (mA)
20
10
25°C
VCC = 5V
9
8
CLOAD = 20pF
7
6
NO LOAD
5
4
3
0
10
20
30
40
FREQUENCY (MHz)
17201 G11
17201 G12
17201fc
5
LT1720/LT1721
PIN FUNCTIONS
LT1720
LT1721
+IN A (Pin 1): Noninverting Input of Comparator A.
–IN A (Pin 1): Inverting Input of Comparator A.
–IN A (Pin 2): Inverting Input of Comparator A.
+IN A (Pin 2): Noninverting Input of Comparator A.
–IN B (Pin 3): Inverting Input of Comparator B.
GND (Pins 3, 6): Ground.
+IN B (Pin 4): Noninverting Input of Comparator B.
OUT A (Pin 4): Output of Comparator A.
GND (Pin 5): Ground.
OUT B (Pin 5): Output of Comparator B.
OUT B (Pin 6): Output of Comparator B.
+IN B (Pin 7): Noninverting Input of Comparator B.
OUT A (Pin 7): Output of Comparator A.
–IN B (Pin 8): Inverting Input of Comparator B.
VCC (Pin 8): Positive Supply Voltage.
–IN C (Pin 9): Inverting Input of Comparator C.
+IN C (Pin 10): Noninverting Input of Comparator C.
VCC (Pins 11, 14): Positive Supply Voltage.
OUT C (Pin 12): Output of Comparator C.
OUT D (Pin 13): Output of Comparator D.
+IN D (Pin 15): Noninverting Input of Comparator D.
–IN D (Pin 16): Inverting Input of Comparator D.
17201fc
6
LT1720/LT1721
TEST CIRCUITS
±VTRIP Test Circuit
15VP-P
BANDWIDTH-LIMITED
TRIANGLE WAVE
~1kHz
LTC203
14
VCC
15
3
2
0.1μF
1000 s VTRIP+
50k
1μF
10nF
+
16
1
9
8
10k
50Ω
50Ω
VCM
–
200k
1/2 LT1112
–
DUT
1/2 LT1720 OR
1/4 LT1721
11
10
6
+
1000 s VHYST
7
1000 s VOS
10k
LTC203
3
1/2 LT1638
+
100k
15
1000 s VTRIP–
1μF
10nF
+
1
16
8
9
2.4k
–
100k
14
100k
–
100k
2
1/2 LT1638
1/2 LT1112
–
0.15μF
6
7
11
+
10
17201 TC01
NOTES: LT1638, LT1112, LTC203s ARE POWERED FROM p15V.
200kW PULL-DOWN PROTECTS LTC203 LOGIC INPUTS
WHEN DUT IS NOT POWERED
Response Time Test Circuit
+VCC – VCM
0V
DUT
1/2 LT1720 OR
1/4 LT1721
–100mV
25Ω
25Ω
50k
0.01μF
+
10 s SCOPE PROBE
(CIN ≈ 10pF)
–
0.01μF
0.1μF
0V
PULSE
IN
V1*
130Ω
50Ω
2N3866
1N5711
–VCM
–3V
50Ω
400Ω
–5V
750Ω
*V1 = –1000 • (OVERDRIVE VTRIP+)
NOTE: RISING EDGE TEST SHOWN.
FOR FALLING EDGE, REVERSE LT1720 INPUTS
17201 TC02
17201fc
7
LT1720/LT1721
APPLICATIONS INFORMATION
Input Voltage Considerations
Input Protection
The LT1720/LT1721 are specified for a common mode range
of –100mV to 3.8V when used with a single 5V supply. In
general the common mode range is 100mV below ground
to 1.2V below VCC. The criterion for this common mode
limit is that the output still responds correctly to a small
differential input signal. Also, if one input is within the
common mode limit, the other input signal can go outside
the common mode limits, up to the absolute maximum
limits (a diode drop past either rail at 10mA input current)
and the output will retain the correct polarity.
The input stage is protected against damage from large
differential signals, up to and beyond a differential voltage
equal to the supply voltage, limited only by the absolute
maximum currents noted. External input protection circuitry is only needed if currents would otherwise exceed
these absolute maximums. The internal catch diodes can
conduct current up to these rated maximums without
latchup, even when the supply voltage is at the absolute
maximum rating.
When either input signal falls below the negative common
mode limit, the internal PN diode formed with the substrate
can turn on, resulting in significant current flow through
the die. An external Schottky clamp diode between the
input and the negative rail can speed up recovery from
negative overdrive by preventing the substrate diode from
turning on.
When both input signals are below the negative common
mode limit, phase reversal protection circuitry prevents
false output inversion to at least –400mV common mode.
However, the offset and hysteresis in this mode will increase
dramatically, to as much as 15mV each. The input bias
currents will also increase.
When both input signals are above the positive common
mode limit, the input stage will become debiased and
the output polarity will be random. However, the internal
hysteresis will hold the output to a valid logic level, and
because the biasing of each comparator is completely
independent, there will be no impact on any other comparator. When at least one of the inputs returns to within
the common mode limits, recovery from this state will
take as long as 1μs.
The propagation delay does not increase significantly when
driven with large differential voltages. However, with low
levels of overdrive, an apparent increase may be seen with
large source resistances due to an RC delay caused by the
2pF typical input capacitance.
The LT1720/LT1721 input stage has general purpose
internal ESD protection for the human body model. For
use as a line receiver, additional external protection may
be required. As with most integrated circuits, the level
of immunity to ESD is much greater when residing on a
printed circuit board where the power supply decoupling
capacitance will limit the voltage rise caused by an ESD
pulse.
Unused Inputs
The inputs of any unused compartor should be tied off in
a way that defines the output logic state. The easiest way
to do this is to tie IN+ to VCC and IN– to GND.
Input Bias Current
Input bias current is measured with both inputs held at 1V.
As with any PNP differential input stage, the LT1720/LT1721
bias current flows out of the device. With a differential
input voltage of even just 100mV or so, there will be zero
bias current into the higher of the two inputs, while the
current flowing out of the lower input will be twice the
measured bias current. With more than two diode drops
of differential input voltage, the LT1720/LT1721’s input
protection circuitry activates, and current out of the lower
input will increase an additional 30% and there will be a
small bias current into the higher of the two input pins,
of 4μA or less. See the Typical Performance curve “Input
Current vs Differential Input Voltage.”
17201fc
8
LT1720/LT1721
APPLICATIONS INFORMATION
High Speed Design Considerations
Application of high speed comparators is often plagued
by oscillations. The LT1720/LT1721 have 4mV of internal
hysteresis, which will prevent oscillations as long as
parasitic output to input feedback is kept below 4mV.
However, with the 2V/ns slew rate of the LT1720/LT1721
outputs, a 4mV step can be created at a 100Ω input source
with only 0.02pF of output to input coupling. The pinouts
of the LT1720/LT1721 have been arranged to minimize
problems by placing the most sensitive inputs (inverting) away from the outputs, shielded by the power rails.
The input and output traces of the circuit board should
also be separated, and the requisite level of isolation is
readily achieved if a topside ground plane runs between
the outputs and the inputs. For multilayer boards where
the ground plane is internal, a topside ground or supply
trace should be run between the inputs and outputs, as
illustrated in Figure 1.
Although both VCC pins are electrically shorted internal to
the LT1721, they must be shorted together externally as
well in order for both to function as shields. The same is
true for the two GND pins.
The supply bypass should include an adjacent 10nF ceramic capacitor and a 2.2μF tantalum capacitor no farther
than 5cm away; use more capacitance if driving more
than 4mA loads. To prevent oscillations, it is helpful to
balance the impedance at the inverting and noninverting
inputs; source impedances should be kept low, preferably
1kΩ or less.
The outputs of the LT1720/LT1721 are capable of very
high slew rates. To prevent overshoot, ringing and other
problems with transmission line effects, keep the output
traces shorter than 10cm, or be sure to terminate the lines
to maintain signal integrity. The LT1720/LT1721 can drive
DC terminations of 250Ω or more, but lower characteristic
impedance traces can be driven with series termination
or AC termination topologies.
Hysteresis
(b)
17201 F01
Figure 1. Typical Topside Metal for Multilayer PCB Layouts
Figure 1a shows a typical topside layout of the LT1720
on such a multilayer board. Shown is the topside metal
etch including traces, pin escape vias, and the land pads
for an SO-8 LT1720 and its adjacent X7R 10nF bypass
capacitor in a 1206 case.
The input-output transfer characteristic is illustrated in
Figure 2 showing the definitions of VOS and VHYST based
upon the two measurable trip points. The hysteresis band
makes the LT1720/LT1721 well behaved, even with slowly
moving inputs.
VOUT
(a)
The LT1720/LT1721 include internal hysteresis, which
makes them easier to use than many other comparable
speed comparators.
VOH
The ground trace from Pin 5 runs under the device up to
the bypass capacitor, shielding the inputs from the outputs. Note the use of a common via for the LT1720 and
the bypass capacitor, which minimizes interference from
high frequency energy running around the ground plane
or power distribution traces.
Figure 1b shows a typical topside layout of the LT1721
on a multilayer board. In this case, the power and ground
traces have been extended to the bottom of the device
solely to act as high frequency shields between input and
output traces.
VHYST
(= VTRIP+ – VTRIP–)
VHYST/2
VOL
$VIN = VIN+ – VIN–
0
–
VTRIP+
VTRIP
++V
V
VOS = TRIP
–
TRIP
2
17201 F02
Figure 2. Hysteresis I/O Characteristics
17201fc
9
LT1720/LT1721
APPLICATIONS INFORMATION
The exact amount of hysteresis will vary from part to part
as indicated in the specifications table. The hysteresis level
will also vary slightly with changes in supply voltage and
common mode voltage. A key advantage of the LT1720/
LT1721 is the significant reduction in these effects, which
is important whenever an LT1720/LT1721 is used to detect
a threshold crossing in one direction only. In such a case,
the relevant trip point will be all that matters, and a stable
offset voltage with an unpredictable level of hysteresis,
as seen in competing comparators, is of little value. The
LT1720/LT1721 are many times better than prior comparators in these regards. In fact, the CMRR and PSRR tests are
performed by checking for changes in either trip point to
the limits indicated in the specifications table. Because the
offset voltage is the average of the trip points, the CMRR
and PSRR of the offset voltage is therefore guaranteed to
be at least as good as those limits. This more stringent
test also puts a limit on the common mode and power
supply dependence of the hysteresis voltage.
Additional hysteresis may be added externally. The
rail-to-rail outputs of the LT1720/LT1721 make this more
predictable than with TTL output comparators due to the
LT1720/LT1721’s small variability of VOH (output high
voltage).
To add additional hysteresis, set up positive feedback
by adding additional external resistor R3 as shown in
Figure 3. Resistor R3 adds a portion of the output to the
threshold set by the resistor string. The LT1720/LT1721
pulls the outputs to the supply rail and ground to within
200mV of the rails with light loads, and to within 400mV
with heavy loads. For the load of most circuits, a good
model for the voltage on the right side of R3 is 300mV or
VCC – 300mV, for a total voltage swing of (VCC – 300mV)
– 300mV = VCC – 600mV.
With this in mind, calculation of the resistor values needed
is a two-step process. First, calculate the value of R3 based
on the additional hysteresis desired, the output voltage
swing, and the impedance of the primary bias string:
R3 = (R1 || R2)(VCC – 0.6V)/(additional hysteresis)
Additional hysteresis is the desired overall hysteresis less
the internal 3.5mV hysteresis.
The second step is to recalculate R2 to set the same average threshold as before. The average threshold before
was set at VTH = (VREF)(R1)/(R1 + R2). The new R2 is
calculated based on the average output voltage (VCC/2)
and the simplified circuit model in Figure 4. To assure
that the comparator’s noninverting input is, on average,
the same VTH as before:
R2′ = (VREF – VTH)/(VTH/R1 + (VTH – VCC/2)/R3)
For additional hysteresis of 10mV or less, it is not
uncommon for R2′ to be the same as R2 within 1%
resistor tolerances.
This method will work for additional hysteresis of up to
a few hundred millivolts. Beyond that, the impedance of
R3 is low enough to effect the bias string, and adjustment of R1 may also be required. Note that the currents
through the R1/R2 bias string should be many times the
input currents of the LT1720/LT1721. For 5% accuracy,
the current must be at least 120μA(6μA IB ÷ 0.05); more
for higher accuracy.
VREF
VREF
R3
R2a
VTH
R3
R2
+
R1
V
VAVERAGE = CC
2
R1
+
1/2 LT1720
–
1/2 LT1720
–
INPUT
17201 F03
Figure 3. Additional External Hysteresis
17201 F04
Figure 4. Model for Additional Hysteresis Calculations
17201fc
10
LT1720/LT1721
APPLICATIONS INFORMATION
Interfacing the LT1720/LT1721 to ECL
The LT1720/LT1721 comparators can be used in high
speed applications where Emitter-Coupled Logic (ECL) is
deployed. To interface the outputs of the LT1720/LT1721
to ECL logic inputs, standard TTL/CMOS to ECL level
translators such as the 10H124, 10H424 and 100124
can be used. These components come at a cost of a few
nanoseconds additional delay as well as supply currents
of 50mA or more, and are only available in quads. A faster,
simpler and lower power translator can be constructed
with resistors as shown in Figure 5.
Finally, Figure 5d shows the case of driving standard, negative-rail, ECL with the LT1720/LT1721. Resistor values are
given for both ECL interface types and for both a 5V and 3V
LT1720/LT1721 supply rail. Again, a fourth resistor, R4 is
needed to prevent the low state current from flowing out of
the LT1720/LT1721, turning on the internal ESD/substrate
diodes. Not only can the output stage functionality and
speed suffer, but in this case the substrate is common to
all the comparators in the LT1720/LT1721, so operation
of the other comparator(s) in the same package could
also be affected. Resistor R4 again prevents this with the
minimum additional power dissipation.
Figure 5a shows the standard TTL to Positive ECL (PECL)
resistive level translator. This translator cannot be used for
the LT1720/LT1721, or with CMOS logic, because it depends
on the 820Ω resistor to limit the output swing (VOH) of
the all-NPN TTL gate with its so-called totem-pole output.
The LT1720/LT1721 are fabricated in a complementary
bipolar process and their output stage has a PNP driver
that pulls the output nearly all the way to the supply rail,
even when sourcing 10mA.
For all the dividers shown, the output impedance is about
110Ω. This makes these fast, less than a nanosecond,
with most layouts. Avoid the temptation to use speedup
capacitors. Not only can they foul up the operation of
the ECL gate because of overshoots, they can damage
the ECL inputs, particularly during power-up of separate
supply configurations.
Figure 5b shows a three resistor level translator for interfacing the LT1720/LT1721 to ECL running off the same supply
rail. No pull-down on the output of the LT1720/LT1721
is needed, but pull-down R3 limits the VIH seen by the
PECL gate. This is needed because ECL inputs have both
a minimum and maximum VIH specification for proper
operation. Resistor values are given for both ECL interface
types; in both cases it is assumed that the LT1720/LT1721
operates from the same supply rail.
Figure 5c shows the case of translating to PECL from an
LT1720/LT1721 powered by a 3V supply rail. Again, resistor values are given for both ECL interface types. This time
four resistors are needed, although with 10KH/E, R3 is not
needed. In that case, the circuit resembles the standard TTL
translator of Figure 5a, but the function of the new resistor,
R4, is much different. R4 loads the LT1720/LT1721 output
when high so that the current flowing through R1 doesn’t
forward bias the LT1720/LT1721’s internal ESD clamp diode.
Although this diode can handle 20mA without damage,
normal operation and performance of the output stage can
be impaired above 100μA of forward current. R4 prevents
this with the minimum additional power dissipation.
The level translator designs assume one gate load. Multiple
gates can have significant IIH loading, and the transmission line routing and termination issues also make this
case difficult.
ECL, and particularly PECL, is valuable technology for high
speed system design, but it must be used with care. With
less than a volt of swing, the noise margins need to be
evaluated carefully. Note that there is some degradation of
noise margin due to the ±5% resistor selections shown.
With 10KH/E, there is no temperature compensation of the
logic levels, whereas the LT1720/LT1721 and the circuits
shown give levels that are stable with temperature. This
will degrade the noise margin over temperature. In some
configurations it is possible to add compensation with
diode or transistor junctions in series with the resistors
of these networks.
For more information on ECL design, refer to the ECLiPS
data book (DL140), the 10KH system design handbook
(HB205) and PECL design (AN1406), all from ON
Semiconductor (www.onsemi.com).
17201fc
11
LT1720/LT1721
APPLICATIONS INFORMATION
5V
5V
180Ω
LSTTL
DO NOT USE FOR LT1720/LT1721
LEVEL TRANSLATION. SEE TEXT
270Ω
10KH/E
820Ω
(a) STANDARD TTL TO PECL TRANSLATOR
VCC
R2
R1
1/2 LT1720
10KH/E
100K/E
R3
VCC
R1
R2
R3
5V OR 5.2V 510Ω 180Ω 750Ω
4.5V
620Ω 180Ω 510Ω
(b) LT1720/LT1721 OUTPUT TO PECL TRANSLATOR
VCC
3V
R1
R2
1/2 LT1720
R4
10KH/E
100K/E
R3
R4
VCC
R1
R2
R3
5V OR 5.2V 300Ω 180Ω OMIT 560Ω
4.5V
330Ω 180Ω 1500Ω 1000Ω
(c) 3V LT1720/LT1721 OUTPUT TO PECL TRANSLATOR
VCC
R4
R1
1/2 LT1720
R2
ECL FAMILY
VEE
10KH/E
–5.2V
100K/E
–4.5V
R3
VCC
5V
3V
5V
3V
R1
560Ω
270Ω
680Ω
330Ω
R2
270Ω
510Ω
270Ω
390Ω
R3
R4
330Ω 1200Ω
300Ω 330Ω
300Ω 1500Ω
270Ω 430Ω
17201 F05
VEE
(d) LT1720/LT1721 OUTPUT TO STANDARD ECL TRANSLATOR
Figure 5
17201fc
12
LT1720/LT1721
APPLICATIONS INFORMATION
Circuit Description
The block diagram of one comparator in the LT1720/LT1721
is shown in Figure 6. There are differential inputs (+IN/–IN),
an output (OUT), a single positive supply (VCC) and ground
(GND). All comparators are completely independent, sharing only the power and ground pins. The circuit topology
consists of a differential input stage, a gain stage with
hysteresis and a complementary common-emitter output
stage. All of the internal signal paths utilize low voltage
swings for high speed at low power.
The input stage topology maximizes the input dynamic
range available without requiring the power, complexity and die area of two complete input stages such as
are found in rail-to-rail input comparators. With a 2.7V
supply, the LT1720/LT1721 still have a respectable 1.6V
of input common mode range. The differential input voltage range is rail-to-rail, without the large input currents
found in competing devices. The input stage also features
phase reversal protection to prevent false outputs when
the inputs are driven below the –100mV common mode
voltage limit.
The internal hysteresis is implemented by positive, nonlinear feedback around a second gain stage. Until this point,
the signal path has been entirely differential. The signal
path is then split into two drive signals for the upper and
lower output transistors. The output transistors are connected common emitter for rail-to-rail output operation.
The Schottky clamps limit the output voltages at about
300mV from the rail, not quite the 50mV or 15mV of Linear
Technology’s rail-to-rail amplifiers and other products. But
the output of a comparator is digital, and this output stage
can drive TTL or CMOS directly. It can also drive ECL, as
described earlier, or analog loads as demonstrated in the
applications to follow.
The bias conditions and signal swings in the output stages
are designed to turn their respective output transistors off
faster than on. This nearly eliminates the surge of current
from VCC to ground that occurs at transitions, keeping
the power consumption low even with high output-toggle
frequencies.
The low surge current is what keeps the power consumption low at high output-toggle frequencies. The frequency
dependence of the supply current is shown in the Typical
Performance Characteristics. Just 20pF of capacitive load
on the output more than triples the frequency dependent
rise. The slope of the no-load curve is just 32μA/MHz. With
a 5V supply, this current is the equivalent of charging and
discharging just 6.5pF. The slope of the 20pF load curve is
133μA/MHz, an addition of 101μA/MHz, or 20μA/MHz-V,
units that are equivalent to picoFarads.
The LT1720/LT1721 dynamic current can be estimated
by adding the external capacitive loading to an internal
equivalent capacitance of 5pF to 15pF, multiplied by the
toggle frequency and the supply voltage. Because the
capacitance of routing traces can easily approach these
values, the dynamic current is dominated by the load in
most circuits.
VCC
NONLINEAR STAGE
+
–
+
+IN
+
–IN
–
+
3
AV1
+
AV2
3
OUT
–
+
–
GND
17201 F06
Figure 6. LT1720/LT1721 Block Diagram
17201fc
13
LT1720/LT1721
APPLICATIONS INFORMATION
Speed Limits
The LT1720/LT1721 comparators are intended for high
speed applications, where it is important to understand a
few limitations. These limitations can roughly be divided
into three categories: input speed limits, output speed
limits, and internal speed limits.
There are no significant input speed limits except the shunt
capacitance of the input nodes. If the 2pF typical input
nodes are driven, the LT1720/LT1721 will respond.
The output speed is constrained by two mechanisms,
the first of which is the slew currents available from the
output transistors. To maintain low power quiescent operation, the LT1720/LT1721 output transistors are sized
to deliver 25mA to 45mA typical slew currents. This is
sufficient to drive small capacitive loads and logic gate
inputs at extremely high speeds. But the slew rate will
slow dramatically with heavy capacitive loads. Because
the propagation delay (tPD) definition ends at the time the
output voltage is halfway between the supplies, the fixed
slew current actually makes the LT1720/LT1721 faster at
3V than 5V with 20mV of input overdrive.
Another manifestation of this output speed limit is skew,
the difference between tPDLH and tPDHL. The slew currents
of the LT1720/LT1721 vary with the process variations of
the PNP and NPN transistors, for rising edges and falling
edges respectively. The typical 0.5ns skew can have either
polarity, rising edge or falling edge faster. Again, the skew
will increase dramatically with heavy capacitive loads.
The skews of comparators in a single package are correlated, but not identical. Besides some random variability,
there is a small (100ps to 200ps) systematic skew due to
physical parasitics of the packages. For the LT1720 SO-8,
comparator A, whose output is adjacent to the VCC pin,
will have a relatively faster rising edge than comparator
B. Likewise, comparator B, by virtue of an output adjacent
to the ground pin will have a relatively faster falling edge.
Similar dependencies occur in the LT1721 S16, while the
systemic skews in the smaller MSOP and SSOP packages
are half again as small. Of course, if the capacitive loads on
the two comparators of a single package are not identical,
the differential timing will degrade further.
The second output speed limit is the clamp turnaround.
The LT1720/LT1721 output is optimized for fast initial
response, with some loss of turnaround speed, limiting
the toggle frequency. The output transistors are idled in a
low power state once VOH or VOL is reached by detecting
the Schottky clamp action. It is only when the output has
slewed from the old voltage to the new voltage, and the
clamp circuitry has settled, that the idle state is reached
and the output is fully ready to transition again. This clamp
turnaround time is typically 8ns for each direction, resulting
in a maximum toggle frequency of 62.5MHz, or a 125MB
data rate. With higher frequencies, dropout and runt pulses
can occur. Increases in capacitive load will increase the time
needed for slewing due to the limited slew currents and
the maximum toggle frequency will decrease further. For
higher toggle frequency applications, refer to the LT1715,
whose output stage can toggle at 150MHz typical.
The internal speed limits manifest themselves as dispersion. All comparators have some degree of dispersion,
defined as a change in propagation delay versus input
overdrive. The propagation delay of the LT1720/LT1721
will vary with overdrive, from a typical of 4.5ns at 20mV
overdrive to 7ns at 5mV overdrive (typical). The LT1720/
LT1721’s primary source of dispersion is the hysteresis
stage. As a change of polarity arrives at the gain stage,
the positive feedback of the hysteresis stage subtracts
from the overdrive available. Only when enough time has
elapsed for a signal to propagate forward through the gain
stage, backwards through the hysteresis stage and forward
through the gain stage again, will the output stage receive
the same level of overdrive that it would have received in
the absence of hysteresis.
With 5mV of overdrive, the LT1720/LT1721 are faster with
a 5V supply than with a 3V supply, the opposite of what
is true with 20mV overdrive. This is due to the internal
speed limit, because the gain stage is faster at 5V than 3V
due primarily to the reduced junction capacitances with
higher reverse voltage bias.
In many applications, as shown in the following examples,
there is plenty of input overdrive. Even in applications
providing low levels of overdrive, the LT1720/LT1721
are fast enough that the absolute dispersion of 2.5ns
(= 7 – 4.5) is often small enough to ignore.
17201fc
14
LT1720/LT1721
APPLICATIONS INFORMATION
The gain and hysteresis stage of the LT1720/LT1721 is
simple, short and high speed to help prevent parasitic
oscillations while adding minimum dispersion. This
internal “self-latch” can be usefully exploited in many
applications because it occurs early in the signal chain, in
a low power, fully differential stage. It is therefore highly
immune to disturbances from other parts of the circuit,
either in the same comparator, on the supply lines, or from
the other comparator(s) in the same package. Once a high
speed signal trips the hysteresis, the output will respond,
after a fixed propagation delay, without regard to these
external influences that can cause trouble in nonhysteretic
comparators.
±VTRIP Test Circuit
The input trip points are tested using the circuit shown in
the Test Circuits section that precedes this Applications
Information section. The test circuit uses a 1kHz triangle
wave to repeatedly trip the comparator being tested. The
LT1720/LT1721 output is used to trigger switched capacitor sampling of the triangle wave, with a sampler for each
direction. Because the triangle wave is attenuated 1000:1
and fed to the LT1720/LT1721’s differential input, the
sampled voltages are therefore 1000 times the input trip
voltages. The hysteresis and offset are computed from
the trip points as shown.
As the power is applied, the circuit remains off until the
LT1720/LT1721 bias circuits activate, at a typical VCC of
2V to 2.2V (25°C), at which point the desired frequency
output is generated.
The output duty cycle for this circuit is roughly 50%, but
it is affected by resistor tolerances and, to a lesser extent,
by comparator offsets and timings. If a 50% duty cycle is
required, the circuit of Figure 7 creates a pair of complementary outputs with a forced 50% duty cycle. Crystals are
narrow-band elements, so the feedback to the noninverting
input is a filtered analog version of the square wave output.
Changing the noninverting reference level can therefore
vary the duty cycle. C1 operates as in the previous example,
whereas C2 creates a complementary output by comparing the same two nodes with the opposite input polarity.
A1 compares band-limited versions of the outputs and
biases C1’s negative input. C1’s only degree of freedom to
respond is variation of pulse width; hence the outputs are
forced to 50% duty cycle. Again, the circuit operates from
2.7V to 6V, and the skew between the edges of the two
outputs are shown in Figure 8. There is a slight duty cycle
dependence on comparator loading, so equal capacitive
and resistive loading should be used in critical applications.
This circuit works well because of the two matched delays
and rail-to-rail style outputs of the LT1720.
VCC
2.7V TO 6V
Crystal Oscillators
A simple crystal oscillator using one comparator of an
LT1720/LT1721 is shown on the first page of this data
sheet. The 2k-620Ω resistor pair set a bias point at the
comparator’s noninverting input. The 2k-1.8k-0.1μF path
sets the inverting input node at an appropriate DC average level based on the output. The crystal’s path provides
resonant positive feedback and stable oscillation occurs.
Although the LT1720/LT1721 will give the correct logic
output when one input is outside the common mode range,
additional delays may occur when it is so operated, opening the possibility of spurious operating modes. Therefore,
the DC bias voltages at the inputs are set near the center
of the LT1720/LT1721’s common mode range and the
220Ω resistor attenuates the feedback to the noninverting input. The circuit will operate with any AT-cut crystal
from 1MHz to 10MHz over a 2.7V to 6V supply range.
1MHz TO 10MHz
CRYSTAL (AT-CUT)
2k
220Ω
620Ω
GROUND
CASE
+
C1
1/2 LT1720
OUTPUT
–
100k
2k
0.1μF
+
A1
LT1636
1.8k
0.1μF
–
1k
+
C2
1/2 LT1720
0.1μF
100k
OUTPUT
–
17201 F07
Figure 7. Crystal Oscillator with Complementary
Outputs and 50% Duty Cycle
17201fc
15
LT1720/LT1721
APPLICATIONS INFORMATION
The optional A1 feedback network shown can be used to
force identical output duty cycles. The steady state duty
cycles of both outputs will be 44%. Note, though, that
the addition of this network only adjusts the percentage
of time each output is high to be the same, which can be
important in switching circuits requiring identical settling
times. It cannot adjust the relative phases between the two
outputs to be exactly 180° apart, because the signal at the
input node driven by the crystal is not a pure sinusoid.
1000
OUTPUT SKEW (ps)
800
600
400
200
0
2.5
3.0
3.5 4.0 4.5 5.0
SUPPLY VOLTAGE (V)
5.5
Q0
2V/DIV
6.0
1720/21 F08
Figure 8. Timing Skew of Figure 7’s Circuit
Q1
2V/DIV
The circuit in Figure 9 shows a crystal oscillator circuit
that generates two nonoverlapping clocks by making full
use of the two independent comparators of the LT1720.
C1 oscillates as before, but with a lower reference level,
C2’s output will toggle at different times. The resistors set
the degree of separation between the output’s high pulses.
With the values shown, each output has a 44% high and
56% low duty cycle, sufficient to allow 2ns between the
high pulses. Figure 10 shows the two outputs.
20ns/DIV
17201 F10
Figure 10. Nonoverlapping Outputs of Figure 9’s Circuit
VCC
2.7V TO 6V
10MHz
CRYSTAL (AT-CUT)
2k
220Ω
620Ω
+
GROUND
CASE
C1
1/2 LT1720
OUTPUT 0
–
2k
1.3k
OPTIONAL—
SEE TEXT
A1
LT1636
0.1μF
2.2k
+
C2
1/2 LT1720
100k
+
0.1μF
–
1k
0.1μF
100k
OUTPUT 1
–
17201 F09
Figure 9. Crystal-Based Nonoverlapping 10MHz Clock Generator
17201fc
16
LT1720/LT1721
APPLICATIONS INFORMATION
Timing Skews
For a number of reasons, the LT1720/LT1721’s superior
timing specifications make them an excellent choice for
applications requiring accurate differential timing skew.
The comparators in a single package are inherently well
matched, with just 300ps ΔtPD typical. Monolithic construction keeps the delays well matched vs supply voltage and
temperature. Crosstalk between the comparators, usually a
disadvantage in monolithic duals and quads, has minimal
effect on the LT1720/LT1721 timing due to the internal
hysteresis, as described in the Speed Limits section.
The circuits of Figure 11 show basic building blocks for
differential timing skews. The 2.5k resistance interacts with
the 2pF typical input capacitance to create at least ±4ns
delay, controlled by the potentiometer setting. A differential
and a single-ended version are shown. In the differential
configuration, the output edges can be smoothly scrolled
through Δt = 0 with negligible interaction.
3ns Delay Detector
It is often necessary to measure comparative timing of
pulse edges in order to determine the true synchronicity
of clock and control signals, whether in digital circuitry
or in high speed instrumentation. The circuit in Figure 12
is a delay detector which will output a pulse when signals
X and Y are out of sync (specifically, when X is high and
Y is low). Note that the addition of an identical circuit to
detect the opposite situation (X low and Y high) allows
for full skew detection.
Comparators U1A and U1B clean up the incoming signals
and render the circuit less sensitive to input levels and
slew rates. The resistive divider network provides level
shifting for the downstream comparator’s common mode
input range, as well as offset to keep the output low except
during a decisive event. When the upstream comparator’s
outputs can overcome the resistively generated offset (and
hysteresis), comparator U1C performs a Boolean “X*_Y”
function and produces an output pulse (see Figure 13).
The circuit will give full output response with input delays
down to 3ns and partial output response with input delays
down to 1.8ns. Capacitor C1 helps ensure that an imbalance of parasitic capacitances in the layout will not cause
common mode excursions to result in differential mode
signal and false outputs.1
1
Make sure the input levels at X and Y are not too close to the 0.5V threshold set by the R8–R9
divider. If you are still getting false outputs, try increasing C1 to 10pF or more. You can also look
for the problem in the impedance balance (R5 || R6 = R7) at the inputs of U1C. Increasing the
offset by lowering R5 will help reject false outputs, but R7 should also be lowered to maintain
impedance balance. For ease of design and parasitic matching, R7 can be replaced by two parallel
resistors equal to R5 and R6.
LT1720
LT1720
CIN
CIN
+
+
INPUT
–
INPUT
–
2.5k
CIN
CIN
0ns TO 4ns
SINGLE-ENDED
DELAY
DIFFERENTIAL p4ns
RELATIVE SKEW
2.5k
CIN
CIN
–
+
+
CIN
CIN
VREF
–
VREF
17201 F11
Figure 11. Building Blocks for Timing Skew Generation with the LT1720
17201fc
17
LT1720/LT1721
APPLICATIONS INFORMATION
DELAY DETECTOR
5V
R5
1.82k*
+
Y
U1A
1/4 LT1721
51Ω*
301Ω*
OPTIONAL LOGARITHMIC PULSE STRETCHER (SEE TEXT)
R6
301Ω*
5V
–
CAPTURE
–
R8*
4.53k
5V
0.1μF
R9
487Ω*
301Ω*
C1
5.6pF
U1C
1/4 LT1721
+
–
U1B
1/4 LT1721
Z
1N5711
VIN
0.33μF
R1
499Ω*
+
VC
U1D
1/4 LT1721
C2
540pF
**
301Ω*
475Ω*
+
R7
261Ω*
51Ω*
X
1V
R2
1k*
DECAY
0V
Y
–
5V
+
X
L
R3
1Ω*
VOFF
–
R4
30Ω*
1V
0V
17201 F12
Z
* 1% METAL FILM RESISTOR
** 270pF s2 FOR REDUCED LEAD INDUCTANCE
5V
0V
RESULT OF X AND NOT Y
Figure 12. 3ns Delay Detector with Logarithmic Pulse Stretcher
Figure 13. Output Pulse Due to Delay of Y Input Pulse
17201fc
18
LT1720/LT1721
APPLICATIONS INFORMATION
Optional Logarithmic Pulse Stretcher
The fourth comparator of the quad LT1721 can be put to
work as a logarithmic pulse stretcher. This simple circuit
can help tremendously if you don’t have a fast enough
oscilloscope (or control circuit) to easily capture 3ns
pulse widths (or faster). When an input pulse occurs, C2
is charged up with a 180ns capture2 time constant. The
hysteresis and 10mV offset across R3 are overcome within
the first nanosecond3, switching the comparator output
high. When the input pulse subsides, C2 discharges with
a 540ns time constant, keeping the comparator on until
the decay overrides the 10mV offset across R3 minus
hysteresis. Because of this exponential decay, the output
pulse width will be proportional to the logarithm of the
input pulse width. It is important to bypass the circuit’s
VCC well to avoid coupling into the resistive divider. R4
keeps the quiescent input voltage in a range where forward
leakage of the diode due to the 0.4V VOL of the driving
comparator is not a problem.
Neglecting some effects4, the output pulse is related to
the input pulse as:
tOUT = τ2 • ln {VCH • [1 – exp (–tP/τ1)]/(VOFF – VH/2)}
– τ1 • ln [VCH/(VCH – VOFF – VH/2)]
+ tP
(1)
where
tP = input pulse width
tOUT = output pulse width
τ1 = R1 || R2 • C2
the capture time constant
τ2 = R2 • C2
the decay time constant
VOFF = 10mV
the voltage drop across R1
VH = 3.5mV
LT1721 hysteresis
VC = VIN – VFDIODE
the input pulse voltage after
the diode drop
VCH = VC • R2/(R1 + R2)
the effective source voltage
for the charge
For simplicity, with tP < τ1, and neglecting the very slight
delay in turn-on due to offset and hysteresis, the equation
can be approximated by:
tOUT = τ2 • ln [(VCH • tP/τ1)/(VOFF – VH/2)]
(2)
For example, an 8ns input pulse gives a 1.67μs output
pulse. Doubling the input pulse to 16ns lengthens the
output pulse by 0.37μs. Doubling the input pulse again
to 32ns adds another 0.37μs to the output pulse, and so
on. The rate of 0.37μs per octave falls out of the above
equation as:
ΔtOUT/octave = τ2 • ln(2)
(3)
There is ±0.01μs jitter5 in the output pulse which gives an
uncertainty referred to the input pulse of less than 2% (60ps
resolution on a 3ns pulse with a 60MHz oscilloscope—not
bad!). The beauty of this circuit is that it gives resolution
precisely where it’s hardest to get. The jitter is due to a
combination of the slow decay of the last few millivolts
on C2 and the 4nV/√Hz noise and 400MHz bandwidth of
the LT1721 input stage. Increasing the offset across R3
or decreasing τ2 will decrease this jitter at the expense of
dynamic range.
The circuit topology itself is extremely fast, limited theoretically only by the speed of the diode, the capture time
constant τ1 and the pulse source impedance. Figure 14
shows results achieved with the implementation shown,
compared to a plot of Equation (1). The low end is limited
by the delivery time of the upstream comparators. As the
input pulse width is increased, the log function is constrained by the asymptotic RC response but, rather than
becoming clamped, becomes time linear. Thus, for very
long input pulses the third term of Equation (1) dominates
and the circuit becomes a 3μs pulse stretcher.
2 So
called because the very fast input pulse is “captured,” for later examination, as a charge on
the capacitor.
3 Assuming the input pulse slew rate at the diode is infinite. This effective delay constant, about
0.4% of τ1 or 0.8ns, is the second term of equation 1, below. Driven by the 2.5ns slew-limited
LT1721, this effective delay will be 2ns.
4 V is dependent on the LT1721 output voltage and nonlinear diode characteristics. Also, the Thevenin
C
equivalent charge voltage seen by C2 is boosted slightly by R2 being terminated above ground.
5 Output jitter increases with inputs pulse widths below ~3ns.
17201fc
19
LT1720/LT1721
APPLICATIONS INFORMATION
in the two output pulse widths is the per-octave response
of your circuit (see Equation (3)). Shorter cable length differences can be used to get a plot of circuit performance
down to 1.5ns (if any), which can then later be used as a
lookup reference when you have moved from quantifying the
circuit to using the circuit. (Note there is a slight aberration
in performance below 10ns. See Figure 14.) As a final check,
feed the circuit with identical cable lengths and check that
it is not producing any output pulses.
14
tOUT STRETCHED (μs)
12
10
8
6
MEASURED
4
EQUATION 1
2
0
1
10
100
tPULSE (ns)
1000
10ns Triple Overlap Generator
10000
17201 F14
Figure 14. Log Pulse Stretcher Output Pulse vs Input Pulse
NANOSECOND
INPUT RANGE
MICROSECOND
OUTPUT RANGE
X
1 FOOT CABLE
L
Y
CIRCUIT OF
FIGURE 12
2V
tOUT
(SEE TEXT)
SPLITTER
0V
n FOOT CABLE
17201 F15
Figure 15. RG-58 Cable with Velocity of Propogation = 66%;
Delay at Y = (n – 1) • 1.54ns
You don’t need expensive equipment to confirm the actual
overall performance of this circuit. All you need is a respectable waveform generator (capable of >~100kHz), a splitter, a
variety of cable lengths and a 20MHz or 60MHz oscilloscope.
Split a single pulse source into different cable lengths and
then into the delay detector, feeding the longer cable into
the Y input (see Figure 15). A 6 foot cable length difference
will create a ~9.2ns delay (using 66% propagation speed
RG-58 cable), and should result in easily measured 1.70μs
output pulses. A 12 foot cable length difference will result
in ~18.4ns delay and 2.07μs output pulses. The difference
The circuit of Figure 16 utilizes an LT1721 to generate three
overlapping outputs whose pulse edges are separated by
10ns as shown. The time constant is set by the RC network on the output of comparator A. Comparator B and D
trip at fixed percentages of the exponential voltage decay
across the capacitor. The 4.22kΩ feed-forward to the C
comparator’s inverting input keeps the delay differences
the same in each direction despite the exponential nature
of the RC network’s voltage.
There is a 15ns delay to the first edge in both directions,
due to the 4.5ns delay of two LT1721 comparators, plus 6ns
delay in the RC network. This starting delay is shortened
somewhat if the pulse was shorter than 40ns because the
RC network will not have fully settled; however, the 10ns
edge separations stay constant.
The values shown utilize only the lowest 75% of the supply
voltage span, which allows it to work down to 2.7V supply.
The delay differences grow a couple nanoseconds from
5V to 2.7V supply due to the fixed VOL /VOH drops which
grow as a percentage at low supply voltage. To keep this
effect to a minimum, the 1kΩ pull-up on comparator A
provides equal loading in either state.
Fast Waveform Sampler
Figure 17 uses a diode-bridge-type switch for clean, fast
waveform sampling. The diode bridge, because of its
inherent symmetry, provides lower AC errors than other
semiconductor-based switching technologies. This circuit
features 20dB of gain, 10MHz full power bandwidth and
100μV/°C baseline uncertainty. Switching delay is less
than 15ns and the minimum sampling window width for
full power response is 30ns.
17201fc
20
LT1720/LT1721
APPLICATIONS INFORMATION
VCC
VCC
OUTPUTS
+
U1B
1/4 LT1721
1k
VCC
–
VCC
INPUT
909Ω
+
U1A
1/4 LT1721
–
VREF
10ns
10ns
750Ω
1.37k
215Ω
+
681Ω
U1C
1/4 LT1721
100pF
–
10ns
10ns
681Ω
4.22k
+
U1D
1/4 LT1721
–
453Ω
17201 F16
Figure 16. 10ns Triple Overlap Generator
5V
2.2k
2.2k
INPUT
p100mV FULL SCALE
+
1k
LT1227
–
OUTPUT
p1V FULL SCALE
909Ω
= 1N5711
AC BALANCE
= CA3039 DIODE ARRAY
(SUBSTRATE TO –5V)
100Ω
3pF
5V
1.5k
3.6k
1.1k
0.1μF
+
CIN
1/2 LT1720
1.1k
–
SAMPLE
COMMAND
2k
10pF
1.1k
1.1k
MRF501
MRF501
SKEW
COMP
DC BALANCE
2.5k
+
500Ω
11
1/2 LT1720
–
2k
820Ω
8
9
CIN
680Ω
820Ω
6
LM3045
10
13
51Ω
7
51Ω
17201 F17
–5V
Figure 17. Fast Waveform Sampler Using the LT1720 for Timing-Skew Compensation
17201fc
21
LT1720/LT1721
APPLICATIONS INFORMATION
The input waveform is presented to the diode bridge switch,
the output of which feeds the LT1227 wideband amplifier.
The LT1720 comparators, triggered by the sample command, generate phase-opposed outputs. These signals are
level shifted by the transistors, providing complementary
bipolar drive to switch the bridge. A skew compensation
trim ensures bridge-drive signal simultaneity within 1ns.
The AC balance corrects for parasitic capacitive bridge imbalances. A DC balance adjustment trims bridge offset.
The trim sequence involves grounding the input via
50Ω and applying a 100kHz sample command. The
DC balance is adjusted for minimal bridge ON vs OFF
variation at the output. The skew compensation and AC
balance adjustments are then optimized for minimum AC
disturbance in the output. Finally, unground the input and
the circuit is ready for use.
Voltage-Controlled Clock Skew Generator
It is sometimes necessary to generate pairs of identical
clock signals that are phase skewed in time. Further, it is
desirable to be able to set the amount of time skew via a
tuning voltage. Figure 18’s circuit does this by utilizing the
LT1720 to digitize phase information from a varactor-tuned
time domain bridge. A 0V to 2V control signal provides
≈±10ns of output skew. This circuit operates from a 2.7V
to 6V supply.
CLOCK
INPUT
VCC
2.7V TO 6V
+
C1
1/2 LT1720
–
Q
FIXED
OUTPUT
Qa
SKEWED
OUTPUT
VCC
2k
2.5k
2.5k*
14k
2k*
2.5k
“FIXED”
+
“SKEWED”
–
MV-209
VARACTOR
DIODE
1M
0.1μF
0.005μF
INPUT
0V TO 2V ≈
p10ns
SKEW
= 1N4148
C2
1/2 LT1720
36pF†
12pF†
1M
L1**
+
VCC
A1
LT1077
–
47μF
+
2.2μF
VIN
6.2M*
= 74HC04
* 1% FILM RESISTOR
** SUMIDA CD43-100
† POLYSTYRENE, 5%
+
10ns
TRIM
SW
LT1317
VC
1.82M*
1.1M
FB
GND
100k
200pF
17201 F18
Figure 18. Voltage-Controlled Clock Skew
17201fc
22
LT1720/LT1721
APPLICATIONS INFORMATION
Coincidence Detector
High speed comparators are especially suited for interfacing pulse-output transducers, such as particle detectors,
to logic circuitry. The matched delays of a monolithic dual
are well suited for those cases where the coincidence of
two pulses needs to be detected. The circuit of Figure 19
is a coincidence detector that uses an LT1720 and discrete
components as a fast AND gate.
The reference level is set to 1V, an arbitrary threshold. Only
when both input signals exceed this will a coincidence
be detected. The Schottky diodes from the comparator
outputs to the base of the MRF-501 form the AND gate,
while the other two Schottkys provide for fast turn-off.
A logic AND gate could instead be used, but would add
considerably more delay than the 300ps contributed by
this discrete stage.
This circuit can detect coincident pulses as narrow as 3ns.
For narrower pulses, the output will degrade gracefully,
responding, but with narrow pulses that don’t rise all the
way to “high” before starting to fall. The decision delay is
4.5ns with input signals 50mV or more above the reference level. This circuit creates a TTL compatible output
but it can typically drive CMOS as well.
For a more detailed description of the operation of this
circuit, see Application Note 75, pages 10 and 11.
5V
300Ω
+
51Ω
3.9k
5V
MRF501
1/2 LT1720
–
5V
OUTPUT
1k
0.1μF
GROUND
CASE LEAD
–
1/2 LT1720
+
4s 1N5711
51Ω
300Ω
17201 F19
COINCIDENCE COMPARATORS
300ps AND GATE
Figure 19. A 3ns Coincidence Detector
17201fc
23
LT1720/LT1721
GND
150Ω
+IN
–IN
VCC
150Ω
17201 SS
OUTPUT
SIMPLIFIED SCHEMATIC
17201fc
24
LT1720/LT1721
PACKAGE DESCRIPTION
DD Package
8-Lead Plastic DFN (3mm × 3mm)
(Reference LTC DWG # 05-08-1698)
R = 0.115
TYP
5
0.38 p 0.10
8
0.675 p0.05
3.00 p0.10
(4 SIDES)
3.5 p0.05
1.65 p0.05
2.15 p0.05 (2 SIDES)
PACKAGE
OUTLINE
1.65 p 0.10
(2 SIDES)
PIN 1
TOP MARK
(NOTE 6)
(DD) DFN 1203
4
0.25 p 0.05
0.75 p0.05
0.200 REF
0.25 p 0.05
1
0.50 BSC
0.50
BSC
2.38 p0.05
(2 SIDES)
2.38 p0.10
(2 SIDES)
0.00 – 0.05
BOTTOM VIEW—EXPOSED PAD
NOTE:
1. DRAWING TO BE MADE A JEDEC PACKAGE OUTLINE M0-229 VARIATION OF (WEED-1)
2. DRAWING NOT TO SCALE
3. ALL DIMENSIONS ARE IN MILLIMETERS
4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE
MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE
5. EXPOSED PAD SHALL BE SOLDER PLATED
6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION
ON TOP AND BOTTOM OF PACKAGE
RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS
S8 Package
8-Lead Plastic Small Outline (Narrow .150 Inch)
(Reference LTC DWG # 05-08-1610)
.189 – .197
(4.801 – 5.004)
NOTE 3
.045 p.005
.050 BSC
8
.245
MIN
7
6
5
.160 p.005
.150 – .157
(3.810 – 3.988)
NOTE 3
.228 – .244
(5.791 – 6.197)
.030 p.005
TYP
1
RECOMMENDED SOLDER PAD LAYOUT
.010 – .020
s 45o
(0.254 – 0.508)
.008 – .010
(0.203 – 0.254)
0o– 8o TYP
.016 – .050
(0.406 – 1.270)
NOTE:
1. DIMENSIONS IN
.053 – .069
(1.346 – 1.752)
.014 – .019
(0.355 – 0.483)
TYP
INCHES
(MILLIMETERS)
2. DRAWING NOT TO SCALE
3. THESE DIMENSIONS DO NOT INCLUDE MOLD FLASH OR PROTRUSIONS.
MOLD FLASH OR PROTRUSIONS SHALL NOT EXCEED .006" (0.15mm)
2
3
4
.004 – .010
(0.101 – 0.254)
.050
(1.270)
BSC
SO8 0303
17201fc
25
LT1720/LT1721
PACKAGE DESCRIPTION
MS8 Package
8-Lead Plastic MSOP
(Reference LTC DWG # 05-08-1660)
0.889 p 0.127
(.035 p .005)
5.23
(.206)
MIN
3.20 – 3.45
(.126 – .136)
0.42 p 0.038
(.0165 p .0015)
TYP
3.00 p 0.102
(.118 p .004)
(NOTE 3)
0.65
(.0256)
BSC
8
7 6 5
0.52
(.0205)
REF
RECOMMENDED SOLDER PAD LAYOUT
3.00 p 0.102
(.118 p .004)
(NOTE 4)
4.90 p 0.152
(.193 p .006)
DETAIL “A”
0.254
(.010)
0o – 6o TYP
GAUGE PLANE
1
0.53 p 0.152
(.021 p .006)
2 3
4
1.10
(.043)
MAX
DETAIL “A”
0.86
(.034)
REF
0.18
(.007)
SEATING
PLANE
0.22 – 0.38
(.009 – .015)
TYP
0.1016 p 0.0508
(.004 p .002)
0.65
(.0256)
BSC
MSOP (MS8) 0307 REV F
NOTE:
1. DIMENSIONS IN MILLIMETER/(INCH)
2. DRAWING NOT TO SCALE
3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS.
MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS.
INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX
S Package
16-Lead Plastic Small Outline (Narrow .150 Inch)
(Reference LTC DWG # 05-08-1610)
.386 – .394
(9.804 – 10.008)
NOTE 3
.045 p.005
.050 BSC
16
N
15
14
13
12
11
10
9
N
.245
MIN
.160 p.005
.150 – .157
(3.810 – 3.988)
NOTE 3
.228 – .244
(5.791 – 6.197)
1
.030 p.005
TYP
2
3
N/2
N/2
RECOMMENDED SOLDER PAD LAYOUT
1
.010 – .020
s 45o
(0.254 – 0.508)
.008 – .010
(0.203 – 0.254)
2
3
4
5
.053 – .069
(1.346 – 1.752)
NOTE:
1. DIMENSIONS IN
.014 – .019
(0.355 – 0.483)
TYP
7
8
.004 – .010
(0.101 – 0.254)
0o – 8o TYP
.016 – .050
(0.406 – 1.270)
6
.050
(1.270)
BSC
S16 0502
INCHES
(MILLIMETERS)
2. DRAWING NOT TO SCALE
3. THESE DIMENSIONS DO NOT INCLUDE MOLD FLASH OR PROTRUSIONS.
MOLD FLASH OR PROTRUSIONS SHALL NOT EXCEED .006" (0.15mm)
17201fc
26
LT1720/LT1721
PACKAGE DESCRIPTION
GN Package
16-Lead Plastic SSOP (Narrow .150 Inch)
(Reference LTC DWG # 05-08-1641)
.189 – .196*
(4.801 – 4.978)
.045 p.005
16 15 14 13 12 11 10 9
.254 MIN
.009
(0.229)
REF
.150 – .165
.229 – .244
(5.817 – 6.198)
.0165 p.0015
.150 – .157**
(3.810 – 3.988)
.0250 BSC
RECOMMENDED SOLDER PAD LAYOUT
1
.015 p .004
s 45o
(0.38 p 0.10)
.007 – .0098
(0.178 – 0.249)
.0532 – .0688
(1.35 – 1.75)
2 3
4
5 6
7
8
.004 – .0098
(0.102 – 0.249)
0o – 8o TYP
.016 – .050
(0.406 – 1.270)
NOTE:
1. CONTROLLING DIMENSION: INCHES
INCHES
2. DIMENSIONS ARE IN
(MILLIMETERS)
.008 – .012
(0.203 – 0.305)
TYP
.0250
(0.635)
BSC
GN16 (SSOP) 0204
3. DRAWING NOT TO SCALE
*DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE
**DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD
FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE
17201fc
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
27
LT1720/LT1721
TYPICAL APPLICATION
Pulse Stretcher
capacitor C now begins charging through R and, at the
end of 100ns, C2 resets low. The output of C1 also goes
low, latching both outputs low. A new pulse at the input
of C1 can now restart the process. Timing capacitor C can
be increased without limit for longer output pulses.
For detecting short pulses from a single sensor, a pulse
stretcher is often required. The circuit of Figure 20 acts as
a one-shot, stretching the width of an incoming pulse to a
consistent 100ns. Unlike a logic one-shot, this LT1720-based
circuit requires only 100pV-s of stimulus to trigger.
This circuit has an ultimate sensitivity of better than
14mV with 5ns to 10ns input pulses. It can even detect
an avalanche generated test pulse of just 1ns duration
with sensitivity better than 100mV.6 It can detect short
events better than the coincidence detector of Figure 14
because the one-shot is configured to catch just 100mV of
upward movement from C1’s VOL, whereas the coincidence
detector’s 3ns specification is based on a full, legitimate
logic high, without the help of a regenerative one-shot.
The circuit works as follows: Comparator C1 functions as
a threshold detector, whereas comparator C2 is configured
as a one-shot. The first comparator is prebiased with a
threshold of 8mV to overcome comparator and system
offsets and establish a low output in the absence of an
input signal. An input pulse sends the output of C1 high,
which in turn latches C2’s output high. The output of C2
is fed back to the input of the first comparator, causing
regeneration and latching both outputs high. Timing
6 See Linear Technology Application Note 47, Appendix B. This circuit can detect the output of the
pulse generator described after 40dB attenuation.
5V
0.01μF
15k
–
OUTPUT
C1
1/2 LT1720
PULSE SOURCE
50Ω
+
100ns
24Ω
51Ω
R
1k
6.8k
C
100pF
–
1N5711
C2
1/2 LT1720
+
2k
2k
17201 F20
2k
Figure 20. A 1ns Pulse Stretcher
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
LT1016
UltraFast Precision Comparator
Industry Standard 10ns Comparator
LT1116
12ns Single Supply Ground-Sensing Comparator
Single Supply Version of LT1016
LT1394
7ns, UltraFast, Single Supply Comparator
6mA Single Supply Comparator
LT1671
60ns, Low Power, Single Supply Comparator
450μA Single Supply Comparator
LT1715
4ns, 150MHz Dual Comparator
Similar to the LT1720 with Independent Input/Output Supplies
LT1719
4.5ns Single Supply 3V/5V Comparator
Single Comparator Similar to the LT1720/LT1721
17201fc
28 Linear Technology Corporation
LT 0908 REV C • PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 ● FAX: (408) 434-0507
●
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© LINEAR TECHNOLOGY CORPORATION 1998