LTC3859AL
Triple Output,
Buck/Buck/Boost Synchronous
Controller with 28µA Burst Mode IQ
Description
Features
Low Operating IQ: 28μA (One Channel On)
n Dual Buck Plus Single Boost Synchronous Controllers
n Outputs Remain in Regulation Through Cold Crank
Down to 2.5V
n Wide Bias Input Voltage Range: 4.5V to 38V
n Buck Output Voltage Range: 0.8V ≤ V
OUT ≤ 24V
n Boost Output Voltage Up to 60V
n R
SENSE or DCR Current Sensing
n100% Duty Cycle for Boost Synchronous MOSFET
Even in Burst Mode® Operation
n Phase-Lockable Frequency (75kHz to 850kHz)
n Programmable Fixed Frequency (50kHz to 900kHz)
n Selectable Continuous, Pulse-Skipping or Low Ripple
Burst Mode Operation at Light Loads
n Very Low Buck Dropout Operation: 99% Duty Cycle
n Adjustable Output Voltage Soft-Start or Tracking
n Low Shutdown I : 10μA
Q
n Small 38-Lead 5mm × 7mm QFN and TSSOP Packages
n
Applications
n
n
n
n
Automotive Always-On and Start-Stop Systems
Battery Operated Digital Devices
Distributed DC Power Systems
Multioutput Buck-Boost Applications
The LTC®3859AL is a high performance triple output (buck/
buck/boost) synchronous DC/DC switching regulator
controller that drives all N-channel power MOSFET stages.
Constant frequency current mode architecture allows
a phase-lockable switching frequency of up to 850kHz.
The LTC3859AL operates from a wide 4.5V to 38V input
supply range. When biased from the output of the boost
converter or another auxiliary supply, the LTC3859AL can
operate from an input supply as low as 2.5V after start-up.
The 28μA no-load quiescent current extends operating
runtime in battery powered systems. OPTI-LOOP compensation allows the transient response to be optimized
over a wide range of output capacitance and ESR values.
The LTC3859AL features a precision 0.8V reference for
the bucks, 1.2V reference for the boost and a power good
output indicator. The PLLIN/MODE pin selects among
Burst Mode operation, pulse-skipping mode, or continuous inductor current mode at light loads.
Compared to the LTC3859 and the LTC3859A, the
LTC3859AL is fully pin-compatible, but with lower Burst
Mode operation IQ (28µA with one channel on).
L, LT, LTC, LTM, Burst Mode, OPTI-LOOP and µModule are registered trademarks and
No RSENSE is a trademark of Linear Technology Corporation. All other trademarks are the
property of their respective owners. Protected by U.S. Patents including 5481178, 5705919,
5929620, 6144194, 6177787, 6580258.
Typical Application
220µF
Efficiency and Power Loss vs
Output Current
1µF
499k
VFB3
VBIAS
TG1
68.1k
100
4.9µH
2mΩ
SW1
TG3
1.2µH
SW3
220µF
BG3
SENSE3–
SENSE3+
INTVCC
4.7µF
BOOST1, 2, 3
SW1, 2, 3
0.1µF
BG1
LTC3859AL
SENSE1+
SENSE1–
VFB1
RUN1, 2, 3
EXTVCC
357k
220µF
90
80
VOUT1 = BURST
EFFICIENCY
1000
70
60
100
50
40
30
20
VOUT1
10000
VIN = 12V
10
VOUT1 = BURST LOSS
1
10
TG2
SW2
BG2
ITH1, 2, 3
0.1µF
68.1k
VOUT1
5V
5A
POWER LOSS (mW)
VIN
2.5V TO 38V
(START-UP ABOVE 5V)
6mΩ
EFFICIENCY (%)
VOUT3
REGULATED AT 10V WHEN VIN < 10V
FOLLOWS VIN WHEN VIN > 10V
6.5µH
VOUT2
8.5V
3A
0
0.0001
0.001
0.1
0.01
1
OUTPUT CURRENT (A)
10
0.1
3859al TA01b
SENSE2+
SENSE2–
VFB2
TRACK/SS1, 2
SS3
PGND SGND
8mΩ
68.1k
649k
68µF
3859al TA01a
3859alf
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1
LTC3859AL
Absolute Maximum Ratings
(Note 1)
Bias Input Supply Voltage (VBIAS)............... –0.3V to 40V
Buck Top Side Driver Voltages
(BOOST1, BOOST2) .............................. –0.3V to 46V
Boost Top Side Driver Voltages
(BOOST3) ............................................. –0.3V to 76V
Buck Switch Voltage (SW1, SW2) ................. –5V to 40V
Boost Switch Voltage (SW3) ......................... –5V to 70V
INTVCC, (BOOST1–SW1),
(BOOST2–SW2), (BOOST3–SW3)............ –0.3V to 6V
RUN1, RUN2, RUN3 ..................................... –0.3V to 8V
Maximum Current Sourced Into Pin
from Source >8V...............................................100µA
SENSE1+, SENSE2 +, SENSE1–
SENSE2 – Voltages...................................... –0.3V to 28V
SENSE3 +, SENSE3– Voltages...................... –0.3V to 40V
FREQ Voltages.......................................–0.3V to INTVCC
EXTVCC....................................................... –0.3V to 14V
ITH1, ITH2, ITH3, VFB1, VFB2, VFB3 Voltages..... –0.3V to 6V
PLLIN/MODE, PGOOD1, OV3 Voltages ......... –0.3V to 6V
TRACK/SS1, TRACK/SS2, SS3 Voltages ...... –0.3V to 6V
Operating Junction Temperature Range (Notes 2, 3)
LTC3859ALE, LTC3859ALI................. –40°C to 125°C
LTC3859ALH...................................... –40°C to 150°C
LTC3859ALMP.................................... –55°C to 150°C
Storage Temperature Range............... –65°C to 150°C
Pin Configuration
TOP VIEW
4
35 SW1
34 BOOST1
SS3 3
29 BG1
SENSE3+ 4
28 SW3
SENSE3– 5
27 TG3
SS3
7
32 SW3
SENSE3+
8
31 TG3
SENSE3–
9
30 BOOST3
29 BG3
ITH3 7
28 VBIAS
SGND 12
27 EXTVCC
RUN1 13
26 INTVCC
RUN2 14
25 BG2
RUN3 15
24 BOOST2
SENSE2– 16
23 SW2
SENSE2+ 17
22 TG2
VFB2 18
21 OV3
ITH2 19
20 TRACK/SS2
26 BOOST3
39
PGND
25 BG3
SGND 8
24 VBIAS
RUN1 9
23 EXTVCC
RUN2 10
22 INTVCC
RUN3 11
21 BG2
SENSE2– 12
20 BOOST2
FE PACKAGE
38-LEAD PLASTIC TSSOP
TJMAX = 150°C, qJA = 25°C/W
EXPOSED PAD (PIN 39) IS PGND, MUST BE SOLDERED TO PCB
SW2
TG2
13 14 15 16 17 18 19
OV3
ITH3 11
39
PGND
VFB3 6
TRACK/SS2
VFB3 10
30 BOOST1
33 BG1
ITH2
6
31 SW1
PLLIN/MODE 2
VFB2
PLLIN/MODE
5
38 37 36 35 34 33 32
FREQ 1
SENSE2+
FREQ
TG1
36 TG1
SENSE1–
PGOOD1
37 PGOOD1
3
TRACK/SS1
2
ITH1
VFB1
SENSE1+
VFB1
38 TRACK/SS1
SENSE1+
1
SENSE1–
TOP VIEW
ITH1
UHF PACKAGE
38-LEAD (5mm × 7mm) PLASTIC QFN
TJMAX = 150°C, qJA = 34.7°C/W
EXPOSED PAD (PIN 39) IS PGND, MUST BE SOLDERED TO PCB
3859alf
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LTC3859AL
Order Information
LEAD FREE FINISH
TAPE AND REEL
PART MARKING*
PACKAGE DESCRIPTION
TEMPERATURE RANGE
LTC3859ALEFE#PBF
LTC3859ALEFE#TRPBF
LTC3859ALFE
38-Lead Plastic TSSOP
–40°C to 125°C
LTC3859ALIFE#PBF
LTC3859ALIFE#TRPBF
LTC3859ALFE
38-Lead Plastic TSSOP
–40°C to 125°C
LTC3859ALHFE#PBF
LTC3859ALHFE#TRPBF
LTC3859ALFE
38-Lead Plastic TSSOP
–40°C to 150°C
LTC3859ALMPFE#PBF
LTC3859ALMPFE#TRPBF
LTC3859ALFE
38-Lead Plastic TSSOP
–55°C to 150°C
LTC3859ALEUHF#PBF
LTC3859ALEUHF#TRPBF
3859AL
38-Lead (5mm × 7mm) Plastic QFN
–40°C to 125°C
LTC3859ALIUHF#PBF
LTC3859ALIUHF#TRPBF
3859AL
38-Lead (5mm × 7mm) Plastic QFN
–40°C to 125°C
LTC3859ALHUHF#PBF
LTC3859ALHUHF#TRPBF
3859AL
38-Lead (5mm × 7mm) Plastic QFN
–40°C to 150°C
LTC3859ALMPUHF#PBF
LTC3859ALMPUHF#TRPBF
3859AL
38-Lead (5mm × 7mm) Plastic QFN
–55°C to 150°C
Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
Electrical Characteristics
The l denotes the specifications which apply over the specified operating
junction temperature range, otherwise specifications are at TA = 25°C. VBIAS = 12V, VRUN1,2,3 = 5V, EXTVCC = 0V unless otherwise
noted. (Note 2)
SYMBOL
PARAMETER
VBIAS
Bias Input Supply Operating Voltage
Range
VFB1,2
Buck Regulated Feedback Voltage
VFB3
Boost Regulated Feedback Voltage
CONDITIONS
MAX
UNITS
38
V
(Note 4); ITH1,2 Voltage = 1.2V
0°C to 85°C, All Grades
LTC3859ALE, LTC3859ALI
LTC3859ALH, LTC3859ALMP
l
l
0.792
0.788
0.786
0.800
0.800
0.800
0.808
0.812
0.812
V
V
V
(Note 4); ITH3 Voltage = 1.2V
0°C to 85°C, All Grades
LTC3859ALE, LTC3859ALI
LTC3859ALH, LTC3859ALMP
l
l
1.183
1.181
1.176
1.200
1.200
1.200
1.214
1.218
1.218
V
V
V
IFB1,2,3
Feedback Current
(Note 4)
Reference Voltage Line Regulation
(Note 4); VIN = 4V to 38V
VLOADREG
Output Voltage Load Regulation
(Note 4)
Transconductance Amplifier gm
TYP
4.5
VREFLNREG
gm1,2,3
MIN
–2
±50
nA
0.002
0.02
%/V
Measured in Servo Loop;
DITH Voltage = 1.2V to 0.7V
l
0.01
0.1
%
Measured in Servo Loop;
DITH Voltage = 1.2V to 2V
l
–0.01
–0.1
%
(Note 4); ITH1,2,3 = 1.2V;
Sink/Source 5µA
2
mmho
3859alf
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3
LTC3859AL
Electrical
Characteristics
The l denotes the specifications which apply over the specified operating
junction temperature range, otherwise specifications are at TA = 25°C. VBIAS = 12V, VRUN1,2,3 = 5V, EXTVCC = 0V unless otherwise
noted. (Note 2)
SYMBOL
PARAMETER
CONDITIONS
IQ
Input DC Supply Current
(Note 5)
Pulse-Skipping or
Forced Continuous Mode
(One Channel On)
RUN1 = 5V and RUN2,3 = 0V or
RUN2 = 5V and RUN1,3 = 0V or
RUN3 = 5V and RUN1,2 = 0V
VFB1, 2 ON = 0.83V (No Load)
VFB3 = 1.25V
Pulse-Skipping or
Forced Continuous Mode
(All Channels On)
RUN1,2,3 = 5V,
VFB1,2 = 0.83V (No Load)
VFB3 = 1.25V
Sleep Mode
(One Channel On, Buck)
RUN1 = 5V and RUN2,3 = 0V or
RUN2 = 5V and RUN1,3 = 0V
VFB,ON = 0.83V (No Load)
Sleep Mode
(One Channel On, Boost)
UVLO
MIN
TYP
MAX
UNITS
1.5
mA
3
mA
28
35
48
59
µA
RUN3 = 5V and RUN1,2 = 0V
VFB3 = 1.25V
33
53
µA
Sleep Mode
(Buck and Boost Channel On)
RUN1 = 5V and RUN2 = 0V or
RUN2 = 5V and RUN1 = 0V
RUN3 = 5V
VFB1,2 = 0.83V (No Load)
VFB3 = 1.25V
33
40
46
59
µA
Sleep Mode
(All Three Channels On)
RUN1,2,3 = 5V,
VFB1,2 = 0.83V (No Load)
VFB3 = 1.25V
38
56
µA
Shutdown
RUN1,2,3 = 0V
10
20
µA
Undervoltage Lockout
INTVCC Ramping Up
l
INTVCC Ramping Down
l
l
l
4.15
4.5
V
3.5
3.8
4.0
V
7
10
13
%
±1
µA
VOVL1,2
Buck Feedback Overvoltage Protection
Measured at VFB1,2 Relative to
Regulated VFB1,2
ISENSE1,2+
SENSE+ Pin Current
Bucks (Channels 1 and 2)
ISENSE3+
SENSE+ Pin Current
Boost (Channel 3)
170
ISENSE1,2–
SENSE– Pin Current
Bucks (Channels 1 and 2)
VOUT1,2 < VINTVCC – 0.5V
VOUT1,2 > VINTVCC + 0.5V
700
µA
±2
µA
µA
±1
µA
ISENSE3 –
SENSE– Pin Current
Boost (Channel 3)
VSENSE3+, VSENSE3 – = 12V
DFMAX,TG
Maximum Duty Factor for TG
Bucks (Channels 1,2) in Dropout, FREQ = 0V
Boost (Channel 3) in Overvoltage
DFMAX,BG
Maximum Duty Factor for BG
Bucks (Channels 1,2) in Overvoltage
Boost (Channel 3)
ITRACK/SS1,2
Soft-Start Charge Current
VTRACK/SS1,2 = 0V
3
5
8
µA
ISS3
Soft-Start Charge Current
VSS3 = 0V
3
5
8
µA
VRUN1 ON
VRUN2,3 ON
RUN1 Pin Threshold
RUN2,3 Pin Threshold
VRUN1 Rising
VRUN2,3 Rising
1.18
1.21
1.24
1.27
1.32
1.33
V
V
VRUN1,2,3 Hyst
RUN Pin Hysteresis
l
l
99
100
%
%
100
96
%
%
70
VSENSE1,2,3(MAX) Maximum Current Sense Threshold
VSENSE3(CM)
98
VFB1,2 = 0.7V, VSENSE1,2– = 3.3V
VFB1,2,3 = 1.1V, VSENSE3 + = 12V
SENSE3 Pins Common Mode Range
(BOOST Converter Input Supply Voltage)
l
43
2.5
50
mV
57
mV
38
V
3859alf
4
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LTC3859AL
Electrical
Characteristics
The l denotes the specifications which apply over the specified operating
junction temperature range, otherwise specifications are at TA = 25°C. VBIAS = 12V, VRUN1,2,3 = 5V, EXTVCC = 0V unless otherwise
noted. (Note 2)
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
Gate Driver
TG1,2
Pull-Up On-Resistance
Pull-Down On-Resistance
2.5
1.5
Ω
Ω
BG1,2
Pull-Up On-Resistance
Pull-Down On-Resistance
2.4
1.1
Ω
Ω
TG3
Pull-Up On-Resistance
Pull-Down On-Resistance
1.2
1.0
Ω
Ω
BG3
Pull-Up On-Resistance
Pull-Down On-Resistance
1.2
1.0
Ω
Ω
TG1,2,3 tr
TG1,2,3 tf
TG Transition Time:
Rise Time
Fall Time
(Note 6)
CLOAD = 3300pF
CLOAD = 3300pF
25
16
ns
ns
BG1,2,3 tr
BG1,2,3 tf
BG Transition Time:
Rise Time
Fall Time
(Note 6)
CLOAD = 3300pF
CLOAD = 3300pF
28
13
ns
ns
TG/BG t1D
Top Gate Off to Bottom Gate On Delay
Synchronous Switch-On Delay Time
CLOAD = 3300pF Each Driver Bucks (Channels 1, 2)
Boost (Channel 3)
30
70
ns
ns
BG/TG t1D
Bottom Gate Off to Top Gate On Delay
Top Switch-On Delay Time
CLOAD = 3300pF Each Driver Bucks (Channels 1, 2)
Boost (Channel 3)
30
70
ns
ns
tON(MIN)1,2
Buck Minimum On-Time
(Note 7)
95
ns
tON(MIN)3
Boost Minimum On-Time
(Note 7)
120
ns
INTVCC Linear Regulator
VINTVCCVBIAS
Internal VCC Voltage
6V < VBIAS < 38V, VEXTVCC = 0V, IINTVCC = 0mA
VLDOVBIAS
INTVCC Load Regulation
ICC = 0mA to 50mA, VEXTVCC = 0V
VINTVCCEXT
Internal VCC Voltage
6V < VEXTVCC < 13V, IINTVCC = 0mA
VLDOEXT
INTVCC Load Regulation
ICC = 0mA to 50mA, VEXTVCC = 8.5V
VEXTVCC
EXTVCC Switchover Voltage
EXTVCC Ramping Positive
VLDOHYS
EXTVCC Hysteresis
5.0
5.0
4.5
5.4
5.6
V
0.7
2
%
5.4
5.6
V
0.7
2
%
4.7
V
200
mV
Oscillator and Phase-Locked Loop
f25k
Programmable Frequency
RFREQ = 25k; PLLIN/MODE = DC Voltage
115
kHz
f65k
Programmable Frequency
RFREQ = 65k; PLLIN/MODE = DC Voltage
440
kHz
f105k
Programmable Frequency
RFREQ = 105k; PLLIN/MODE = DC Voltage
fLOW
Low Fixed Frequency
VFREQ = 0V PLLIN/MODE = DC Voltage
320
350
380
kHz
fHIGH
High Fixed Frequency
VFREQ = INTVCC; PLLIN/MODE = DC Voltage
485
535
585
kHz
fSYNC
Synchronizable Frequency
PLLIN/MODE = External Clock
850
kHz
835
l
75
kHz
PGOOD1 Output
VPGL1
PGOOD1 Voltage Low
IPGOOD1 = 2mA
IPGOOD1
PGOOD1 Leakage Current
VPGOOD1 = 5V
VPG1
PGOOD1 Trip Level
VFB1 with Respect to Set Regulated Voltage
VFB1 Ramping Negative
0.2
–13
Hysteresis
VFB1 Ramping Positive
Hysteresis
–10
0.4
V
±1
µA
–7
2.5
7
10
2.5
%
%
13
%
%
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5
LTC3859AL
Electrical Characteristics
The l denotes the specifications which apply over the specified operating
junction temperature range, otherwise specifications are at TA = 25°C. VBIAS = 12V, VRUN1,2,3 = 5V, EXTVCC = 0V unless otherwise
noted. (Note 2)
SYMBOL
PARAMETER
TPG1
Delay For Reporting a Fault
CONDITIONS
MIN
TYP
MAX
40
UNITS
µs
OV3 Boost Overvoltage Indicator Output
VOV3L
OV3 Voltage Low
IOV3 = 2mA
IOV3
OV3 Leakage Current
VOV3 = 5V
0.2
VOV
OV3 Trip Level
VFB3 Ramping Positive with Respect to Set
Regulated Voltage
6
10
0.4
V
±1
µA
13
%
Hysteresis
1.5
%
VBOOST3 = 16V; VSW3 = 12V;
Forced Continuous Mode
65
µA
BOOST3 Charge Pump
IBST3
BOOST3 Charge Pump Available Output
Current
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: The LTC3859AL is tested under pulsed load conditions such that
TJ ≈ TA. The LTC3859ALE is guaranteed to meet performance
specifications from 0°C to 85°C. Specifications over the –40°C to
125°C operating junction temperature range are assured by design,
characterization and correlation with statistical process controls. The
LTC3859ALI is guaranteed over the –40°C to 125°C operating junction
temperature range, the LTC3859ALH is guaranteed over the –40°C to
150°C operating junction temperature range and the LTC3859ALMP
is tested and guaranteed over the –55°C to 150°C operating junction
temperature range. High junction temperatures degrade operating
lifetimes; operating lifetime is derated for junction temperatures greater
than 125°C. Note that the maximum ambient temperature consistent with
these specifications is determined by specific operating conditions in
conjunction with board layout, the rated package thermal impedance and
other environmental factors. TJ is calculated from the ambient temperature
TA and power dissipation PD according to the following formula: TJ = TA +
(PD • θJA), where θJA = 34.7°C/W for the QFN package and θJA = 25°C/W
for the TSSOP package.
Note 3: This IC includes overtermperature protection that is intended to
protect the device during momentary overload conditions. The maximum
rated junction temperature will be exceeded when this protection is active.
Continuous operation above the specified absolute maximum operating
junction temperature may impair device reliability or permanently damage
the device.
Note 4: The LTC3859AL is tested in a feedback loop that servos VITH1,2,3
to a specified voltage and measures the resultant VFB. The specification at
85°C is not tested in production and is assured by design, characterization
and correlation to production testing at other temperatures (125°C for the
LTC3859ALE/LTC3859ALI, 150°C for the LTC3859ALH/LTC3859ALMP).
For the LTC3859ALI and LTC3859ALH, the specification at 0°C is not
tested in production and is assured by design, characterization and
correlation to production testing at –40°C. For the LTC3859ALMP, the
specification at 0°C is not tested in production and is assured by design,
characterization and correlation to production testing at –55°C.
Note 5: Dynamic supply current is higher due to the gate charge being
delivered at the switching frequency. See the Applications Information
section.
Note 6: Rise and fall times are measured using 10% and 90% levels. Delay
times are measured using 50% levels.
Note 7: The minimum on-time condition is specified for an inductor
peak-to-peak ripple current ≥ 40% of IMAX (See the Minimum On-Time
Considerations in the Applications Information section).
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LTC3859AL
Typical Performance Characteristics
Efficiency and Power Loss
vs Output Current (Buck)
Efficiency
vs Output Current (Buck)
10000
90
90
60
100
50
FCM EFFICIENCY
10
PULSE-SKIPPING
EFFICIENCY
BURST LOSS
20
BURST EFFICIENCY 1
FCM LOSS
10
PULSE-SKIPPING
LOSS
0.1
0
0.0001 0.001
0.1
0.01
1
10
OUTPUT CURRENT (A)
3859al G01
FIGURE 12 CIRCUIT
VIN = 10V, VOUT = 5V
40
30
EFFICIENCY (%)
70
FIGURE 12 CIRCUIT
99 VOUT = 5V
ILOAD = 4A
VIN = 10V
VIN = 20V
98
70
60
50
40
30
97
96
95
94
20
93
10 FIGURE 12 CIRCUIT
VOUT = 5V
0
0.0001 0.001
0.1
0.01
1
OUTPUT CURRENT (A)
92
10
0
5
IL
2A//DIV
3859al G04
Inductor Current at Light Load
(Buck)
15 20 25 30
INPUT VOLTAGE (V)
35
40
3859al G03
Load Step (Buck)
Pulse-Skipping Mode
VOUT
100mV/DIV
AC-COUPLED
10
3859al G02
Load Step (Buck)
Burst Mode Operation
50µs/DIV
VIN = 12V
VOUT = 5V
FIGURE 12 CIRCUIT
100
80
1000
POWER LOSS (mW)
EFFICIENCY (%)
80
Efficiency vs Input Voltage (Buck)
100
EFFICIENCY (%)
100
Load Step (Buck)
Forced Continuous Mode
VOUT
100mV/DIV
AC-COUPLED
VOUT
100mV/DIV
AC-COUPLED
IL
2A//DIV
IL
2A//DIV
50µs/DIV
VIN = 12V
VOUT = 5V
FIGURE 12 CIRCUIT
50µs/DIV
VIN = 12V
VOUT = 5V
FIGURE 12 CIRCUIT
3859al G05
3859al G06
Buck Regulated Feedback Voltage
vs Temperature
Soft Start-Up (Buck)
FORCED
CONTINUOUS
MODE
VOUT2
2V/DIV
Burst Mode
OPERATION
1A/DIV
VOUT1
2V/DIV
RUN1/
RUN2
5V/DIV
PULSESKIPPING
MODE
2µs/DIV
VIN = 10V
VOUT = 5V
ILOAD = 1mA
FIGURE 12 CIRCUIT
3859al G07
5ms/DIV
VIN = 12V
FIGURE 12 CIRCUIT
3859al G08
REGULATED FEEDBACK VOLTAGE (mV)
808
806
804
802
800
798
796
794
792
–75 –50 –25
0 25 50 75 100 125 150
TEMPERATURE (°C)
3859al G09
3859alf
For more information www.linear.com/3859AL
7
LTC3859AL
Typical Performance Characteristics
Efficiency and Power Loss
vs Output Current (Boost)
Efficiency
vs Output Current (Boost)
100
10000
90
90
100
50
FCM EFFICIENCY
PULSE-SKIPPING 10
40
EFFICIENCY
BURST LOSS
30
BURST
20
1
EFFICIENCY
FCM LOSS
10
PULSE-SKIPPING
LOSS
0.1
0
0.0001 0.001
0.1
0.01
1
10
OUTPUT CURRENT (A)
3859al G10
FIGURE 12 CIRCUIT
VIN = 5V, VOUT = 10V, VBIAS = VIN
EFFICIENCY (%)
60
100
VIN = 5V
80
1000
70
POWER LOSS (mW)
EFFICIENCY (%)
80
VIN = 8V
70
EFFICIENCY (%)
100
Efficiency vs Input Voltage
(Boost)
60
50
40
30
FIGURE 12 CIRCUIT
99 VBIAS = VIN
V
= 10V
98 OUT
ILOAD = 2A
97
96
95
94
93
92
20
FIGURE 12 CIRCUIT
10 VBIAS = VIN
VOUT = 10V
0
0.0001 0.001
0.1
0.01
1
OUTPUT CURRENT (A)
91
90
10
Load Step (Boost)
Burst Mode Operation
3
4
5
6
7
8
INPUT VOLTAGE (V)
Load Step (Boost)
Pulse-Skipping Mode
VOUT
100mV/DIV
AC-COUPLED
IL
5A/DIV
IL
5A/DIV
IL
5A/DIV
200µs/DIV
VOUT = 10V
VIN = 5V
FIGURE 12 CIRCUIT
Inductor Current at Light Load
(Boost)
10
Load Step (Boost)
Forced Continuous Mode
VOUT
100mV/DIV
AC-COUPLED
3859al G13
9
3859al G12
VOUT
100mV/
DIV
ACCOUPLED
200µs/DIV
VOUT = 10V
VIN = 5V
FIGURE 12 CIRCUIT
2
3859al G11
200µs/DIV
VOUT = 10V
VIN = 5V
FIGURE 12 CIRCUIT
3859al G14
3859al G15
Boost Regulated Feedback
Voltage vs Temperature
Soft Start-Up (Boost)
VOUT3
2V/DIV
Burst Mode
OPERATION
5A/DIV
RUN3
5V/DIV
PULSESKIPPING
MODE
GND
2µs/DIV
VOUT = 10V
VIN = 7V
ILOAD = 1mA
FIGURE 12 CIRCUIT
3859al G16
5ms/DIV
VIN = 5V
FIGURE 12 CIRCUIT
3859al G17
REGULATED FEEDBACK VOLTAGE (V)
1.212
FORCED
CONTINUOUS
MODE
1.209
1.206
1.203
1.200
1.197
1.194
1.191
1.188
–75 –50 –25
0 25 50 75 100 120 150
TEMPERATURE (°C)
3859al G18
3859alf
8
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LTC3859AL
Typical Performance Characteristics
5.6
6.0
EXTVCC = 0V
5.4
5.4
5.3
5.2
5.8
EXTVCC = 8.5V
5.2
INTVCC VOLTAGE (V)
INTVCC VOLTAGE (V)
EXTVCC Switchover and INTVCC
Voltages vs Temperature
EXTVCC AND INTVCC VOLTAGE (V)
5.5
INTVCC and EXTVCC
vs Load Current
INTVCC Line Regulation
5.0
4.8
EXTVCC = 5V
4.6
4.4
5.1
4.2
5.0
0
5
10
15 20 25 30
INPUT VOLTAGE (V)
35
4.0
40
20
60
80
40
LOAD CURRENT (mA)
3859al G19
SENSE CURRENT (µA)
SENSE CURRENT (µA)
500
400
300
SENSE3 PIN
800
180
VOUT > INTVCC + 0.5V
500
400
300
BOOST
50
BUCK
40
30
20
10
0
0
10 20 30 40 50 60 70 80 90 100
DUTY CYCLE (%)
3859al G25
100
80
60
0
–75 –50 –25
0 25 50 75 100 125 150
TEMPERATURE (°C)
SENSE3– PIN
0 25 50 75 100 125 150
TEMPERATURE (°C)
60
3859al G24
Maximum Current Sense
Threshold vs ITH Voltage
TRACK/SS Pull-Up Current
vs Temperature
6.00
50
5.75
40
TRACK/SS CURRENT (µA)
MAXIMUM CURRENT SENSE VOLTAGE (mV)
MAXIMUM CURRENT SENSE VOLTAGE (mV)
60
120
3859al G23
Maximum Current Sense
Threshold vs Duty Cycle
70
140
20
VOUT < INTVCC – 0.5V
3859al G22
80
SENSE3+ PIN
40
0
–75 –50 –25
40
VIN = 12V
160
200
5
10 15 20 25 30 35
VSENSE COMMON MODE VOLTAGE (V)
0 25 50 75 100 125 150
TEMPERATURE (°C)
3859al G21
200
100
0
EXTVCC FALLING
4.4
Boost SENSE Pin Total Input
Current vs Temperature
600
100
0
4.6
900
700
600
EXTVCC RISING
4.8
Buck SENSE– Pin Input Bias
Current vs Temperature
SENSE1, 2 PINS
200
5.0
3859al G20
SENSE Pins Total Input Current
vs VSENSE Voltage
700
5.2
4.0
–75 –50 –25
100
SENSE CURRENT (µA)
800
INTVCC
5.4
4.2
VBIAS = 12V
0
5.6
30
20
10
0
–10
PULSE-SKIPPING
FORCED CONTINUOUS
Burst Mode OPERATION
–20
–30
0
0.2
0.4
0.6 0.8
ITH (V)
1
1.2
1.4
3859al G26
5.50
5.25
5.00
4.75
4.50
4.25
4.00
–75 –50 –25
0 25 50 75 100 125 150
TEMPERATURE (°C)
3859al G27
3859alf
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9
LTC3859AL
Typical Performance Characteristics
Shutdown Current
vs Input Voltage
Shutdown Current vs Temperature
20
Quiescent Current vs Temperature
25
VBIAS = 12V
80
70
16
14
12
10
8
20
QUIESCENT CURRENT (µA)
SHUTDOWN CURRENT (µA)
SHUTDOWN CURRENT (µA)
18
15
10
5
6
0
0 25 50 75 100 125 150
TEMPERATURE (°C)
5
3859al G28
10
15
20
25
30
VBIAS INPUT VOLTAGE (V)
35
30
CHANNEL 1 ON
20
0
–75 –50 –25
40
0 25 50 75 100 125 150
TEMPERATURE (°C)
3859al G29
3859al G30
Undervoltage Lockout Threshold
vs Temperature
600
4.4
4.3
FREQ = INTVCC
500
450
400
FREQ = GND
350
RISING
4.2
INTVCC VOLTAGE (V)
550
FREQUENCY (kHz)
4.1
4.0
3.9
FALLING
3.8
3.7
3.6
3.5
0
300
–75 –50 –25
100 200 300 400 500 600 700 800
FEEDBACK VOLTAGE (mV)
3859al G31
1.40
RUN2,3 RISING
RUN1 RISING
1.20
1.15
1.10
RUN2,3 FALLING
RUN1 FALLING
1.05
1.00
–75 –50 –25
Charge Pump Charging Current
vs Switch Voltage
100
0 25 50 75 100 125 150
TEMPERATURE (°C)
3859al G34
CHARGE PUMP CHARGING CURRENT (µA)
1.25
3859al G33
Charge Pump Charging Current
vs Operating Frequency
1.35
0 25 50 75 100 125 150
TEMPERATURE (°C)
3859al G32
Shutdown (RUN) Threshold
vs Temperature
1.30
3.4
–75 –50 –25
0 25 50 75 100 125 150
TEMPERATURE (°C)
100
VBOOST3 = 16V
90 VSW3 = 12V
–55°C
80
70
CHARGE PUMP CHARGING CURRENT (µA)
MAXIMUM CURRENT SENSE VOLTAGE (mV)
ALL CHANNELS ON
40
Oscillator Frequency
vs Temperature
Buck Foldback Current Limit
RUN PIN VOLTAGE (V)
50
10
4
–75 –50 –25
70
65
60
55
50
45
40
35
30
25
20
15
10
5
0
60
25°C
60
50
40
150°C
30
20
10
0
100
200 300 400 500 600 700
OPERATING FREQUENCY (kHz)
800
3859al G35
VBOOST3 – VSW3 = 4V
90
FREQ = 0V
80
70
FREQ = INTVCC
60
50
40
30
20
10
0
5
10
15
20
25
30
SWITCH VOLTAGE (V)
35
40
3859al G36
3859alf
10
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LTC3859AL
Pin Functions
(QFN/TSSOP)
FREQ (Pin 1/Pin 5): The Frequency Control Pin for the
Internal VCO. Connecting the pin to GND forces the VCO
to a fixed low frequency of 350kHz. Connecting the pin
to INTVCC forces the VCO to a fixed high frequency of
535kHz. Other frequencies between 50kHz and 900kHz can
be programmed using a resistor between FREQ and GND.
The resistor and an internal 20µA source current create a
voltage used by the internal oscillator to set the frequency.
PLLIN/MODE (Pin 2/Pin 6): External Synchronization
Input to Phase Detector and Forced Continuous Mode
Input. When an external clock is applied to this pin, the
phase-locked loop will force the rising TG1 signal to be
synchronized with the rising edge of the external clock,
and the regulators operate in forced continuous mode.
When not synchronizing to an external clock, this input,
which acts on all three controllers, determines how the
LTC3859AL operates at light loads. Pulling this pin to
ground selects Burst Mode operation. An internal 100k
resistor to ground also invokes Burst Mode operation
when the pin is floated. Tying this pin to INTVCC forces
continuous inductor current operation. Tying this pin to
a voltage greater than 1.2V and less than INTVCC – 1.3V
selects pulse-skipping operation. This can be done by
connecting a 100k resistor from this pin to INTVCC.
SGND (Pin 8/Pin 12): Small Signal Ground common to
all three controllers, must be routed separately from high
current grounds to the common (–) terminals of the CIN
capacitors.
RUN1, RUN2, RUN3 (Pins 9, 10, 11/Pins 13, 14, 15):
Digital Run Control Inputs for Each Controller. Forcing
RUN1 below 1.17V and RUN2/RUN3 below 1.20V shuts
down that controller. Forcing all of these pins below 0.7V
shuts down the entire LTC3859AL, reducing quiescent
current to approximately 10µA.
OV3 (Pin 17/Pin 21): Overvoltage Open-Drain Logic
Output for the Boost Regulator. OV3 is pulled to ground
when the voltage on the VFB3 pin is under 110% of its set
point, and becomes high impedance when VFB3 goes over
110% of its set point.
INTVCC (Pin 22/Pin 26): Output of the Internal Linear
Low Dropout Regulator. The driver and control circuits
are powered from this voltage source. Must be decoupled
to PGND with a minimum of 4.7µF ceramic or tantalum
capacitor.
EXTVCC (Pin 23/Pin 27): External Power Input to an Internal LDO Connected to INTVCC. This LDO supplies INTVCC
power, bypassing the internal LDO powered from VBIAS
whenever EXTVCC is higher than 4.7V. See EXTVCC Connection in the Applications Information section. Do not
float or exceed 14V on this pin.
VBIAS (Pin 24/Pin 28): Main Bias Input Supply Pin. A
bypass capacitor should be tied between this pin and the
SGND pin.
BG1, BG2, BG3 (Pins 29, 21, 25/Pins 33, 25, 29): High
Current Gate Drives for Bottom (Synchronous) N-Channel
MOSFETs. Voltage swing at these pins is from ground to
INTVCC.
BOOST1, BOOST2, BOOST3 (Pins 30, 20, 26/Pins 34,
24, 30): Bootstrapped Supplies to the Top Side Floating
Drivers. Capacitors are connected between the BOOST and
SW pins and Schottky diodes are tied between the BOOST
and INTVCC pins. Voltage swing at the BOOST pins is from
INTVCC to (VIN + INTVCC).
SW1, SW2, SW3 (Pins 31, 19, 28/Pins 35, 23, 32):
Switch Node Connections to Inductors.
TG1, TG2, TG3 (Pins 32, 18, 27/Pins 36, 22, 31): High
Current Gate Drives for Top N-Channel MOSFETs. These are
the outputs of floating drivers with a voltage swing equal
to INTVCC superimposed on the switch node voltage SW.
PGOOD1 (Pin 33/Pin 37): Open-Drain Logic Output.
PGOOD1 is pulled to ground when the voltage on the VFB1
pin is not within ±10% of its set point.
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11
LTC3859AL
pin functions
(QFN/TSSOP)
TRACK/SS1, TRACK/SS2, SS3 (Pins 34, 16, 3/Pins 38,
20, 7): External Tracking and Soft-Start Input. For the buck
channels, the LTC3859AL regulates the VFB1,2 voltage to
the smaller of 0.8V, or the voltage on the TRACK/SS1,2
pin. For the boost channel, the LTC3859AL regulates the
VFB3 voltage to the smaller of 1.2V, or the voltage on the
SS3 pin. An internal 5µA pull-up current source is connected to this pin. A capacitor to ground at this pin sets the
ramp time to final regulated output voltage. Alternatively, a
resistor divider on another voltage supply connected to the
TRACK/SS pins of the buck channels allow the LTC3859AL
buck outputs to track the other supply during start-up.
ITH1, ITH2, ITH3 (Pins 35, 15, 7/Pins 1, 19, 11): Error
Amplifier Outputs and Switching Regulator Compensation
Points. Each associated channel’s current comparator trip
point increases with this control voltage.
VFB1, VFB2, VFB3 (Pins 36, 14, 6/Pins 2, 18, 10): Receives
the remotely sensed feedback voltage for each controller
from an external resistive divider across the output.
SENSE1+, SENSE2+, SENSE3+ (Pins 37, 13, 4/Pins 3, 17, 8):
The (+) Input to the Differential Current Comparators.
The ITH pin voltage and controlled offsets between the
SENSE– and SENSE+ pins in conjunction with RSENSE
set the current trip threshold. For the boost channel, the
SENSE3+ pin supplies current to the current comparator.
SENSE1–, SENSE2–, SENSE3– (Pins 38, 12, 5/Pins 4,
16, 9): The (–) Input to the Differential Current Comparators. When SENSE1,2– for the buck channels is greater
than INTVCC, then SENSE1,2– pin supplies current to the
current comparator.
PGND (Exposed Pad Pin 39): Driver Power Ground. Connects to the sources of bottom N-channel MOSFETs and the
(–) terminal(s) of CIN. The exposed pad must be soldered
to the PCB for rated electrical and thermal performance.
3859alf
12
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For more information www.linear.com/3859AL
4.7V
EXTVCC
VBIAS
FREQ
+
–
LDO
LDO
EN
5.4V
SGND
SYNC
DET
CLP
VCO
5.4V
EN
100k
20µA
+
–
+
–
CLK1
CLK2
INTVCC
PFD
0.72V
VFB1
0.88V
6.8V
RUN
11V
SHDN
RST
2(VFB)
7µA CH1
160nA CH2
+
–
BOT
FOLDBACK
SLEEP
SHDN
TOPON
+
– –+
SLOPE COMP
2.8V
0.65V
Q
R
ICMP
Q
DROPOUT
DET
S
BUCK CHANNELS 1 AND 2
3mV
–+
PGOOD1
OV
+
–
SHDN
+
–
–
EA +
+
IR
SWITCHING
LOGIC
5µA
0.88V
PGND
BG
SW
TG
TRACK/SS
ITH
VFB
SENSE–
SENSE+
INTVCC
0.80V
TRACK/SS
BOT
TOP
3859al BD
BOOST
CSS
CB
DB
INTVCC
L
RB
D
CC2
CC
RA
VIN1,2
RC
RSENSE
COUT
CIN
VOUT1,2
LTC3859AL
Functional Diagram
3859alf
13
14
OV3
+
–
VFB3
1.32V
PLLIN/MODE
CLK1
RUN3
Q
For more information www.linear.com/3859AL
11V
160nA
SHDN
+
–
SNSLO
5µA
OV
+
–
2mV
1.32V
+
–
SS3
ITH3
VFB3
SENSE3+
SENSE3–
PGND
BG3
SW3
TG3
3859al BD
BOOST3
INTVCC
1.2V
SS3
BOT
TOP
–
EA +
+
2V
IR
SWITCHING
LOGIC
+
+– –
SLEEP
SHDN
BOTON
SNSLO
+
– –+
ICMP
0.425V
Q
S
R
SLOPE COMP
2.8V
0.7V
BOOST CHANNEL 3
CSS
CB
DB
INTVCC
L
RB
CC2
CC
RA
VOUT3
RC
RSENSE
CIN
COUT
VOUT3
VIN3
LTC3859AL
functional diagram
3859alf
LTC3859AL
Operation
(Refer to Functional Diagram)
Main Control Loop
The LTC3859AL uses a constant frequency, current mode
step-down architecture. The two buck controllers, channels 1 and 2, operate 180 degrees out of phase with each
other. The boost controller, channel 3, operates in phase
with channel 1. During normal operation, the external
top MOSFET for the buck channels (the external bottom
MOSFET for the boost channel) is turned on when the
clock for that channel sets the RS latch, and is turned off
when the main current comparator, ICMP, resets the RS
latch. The peak inductor current at which ICMP trips and
resets the latch is controlled by the voltage on the ITH pin,
which is the output of the error amplifier EA. The error
amplifier compares the output voltage feedback signal at
the VFB pin, (which is generated with an external resistor
divider connected across the output voltage, VOUT, to
ground) to the internal 0.800V reference voltage for the
bucks (1.2V reference voltage for the boost). When the
load current increases, it causes a slight decrease in VFB
relative to the reference, which causes the EA to increase
the ITH voltage until the average inductor current matches
the new load current.
After the top MOSFET for the bucks (the bottom MOSFET
for the boost) is turned off each cycle, the bottom MOSFET
is turned on (the top MOSFET for the boost) until either
the inductor current starts to reverse, as indicated by the
current comparator IR, or the beginning of the next clock
cycle.
INTVCC/EXTVCC Power
Power for the top and bottom MOSFET drivers and most
other internal circuitry is derived from the INTVCC pin.
When the EXTVCC pin is left open or tied to a voltage less
than 4.7V, the VBIAS LDO (low dropout linear regulator)
supplies 5.4V from VBIAS to INTVCC. If EXTVCC is taken
above 4.7V, the VBIAS LDO is turned off and an EXTVCC
LDO is turned on. Once enabled, the EXTVCC LDO supplies
5.4V from EXTVCC to INTVCC. Using the EXTVCC pin allows
the INTVCC power to be derived from a high efficiency
external source such as one of the LTC3859AL switching
regulator outputs.
Each top MOSFET driver is biased from the floating bootstrap capacitor CB, which normally recharges during each
cycle through an external diode when the switch voltage
goes low.
For buck channels 1 and 2, if the buck’s input voltage
decreases to a voltage close to its output, the loop may
enter dropout and attempt to turn on the top MOSFET continuously. The dropout detector detects this and forces the
top MOSFET off for about one twelfth of the clock period
every tenth cycle to allow CB to recharge.
Shutdown and Start-Up (RUN1, RUN2, RUN3 and
TRACK/SS1, TRACK/SS2, SS3 Pins)
The three channels of the LTC3859AL can be independently
shut down using the RUN1, RUN2 and RUN3 pins. Pulling
RUN1 below 1.17V and RUN2/RUN3 below 1.20V shuts
down the main control loop for that channel. Pulling all
three pins below 0.7V disables all controllers and most
internal circuits, including the INTVCC LDOs. In this state,
the LTC3859AL draws only 10µA of quiescent current.
Releasing a RUN pin allows a small internal current to pull
up the pin to enable that controller. The RUN1 pin has a
7µA pull-up current while the RUN2 and RUN3 pins have
a smaller 160nA. The 7µA current on RUN1 is designed
to be large enough so that the RUN1 pin can be safely
floated (to always enable the controller) without worry
of condensation or other small board leakage pulling the
pin down. This is ideal for always-on applications where
one or more controllers are enabled continuously and
never shut down.
Each RUN pin may also be externally pulled up or driven
directly by logic. When driving a RUN pin with a low impedance source, do not exceed the absolute maximum rating
of 8V. Each RUN pin has an internal 11V voltage clamp
that allows the RUN pin to be connected through a resistor
to a higher voltage (for example, VBIAS), so long as the
maximum current in the RUN pin does not exceed 100µA.
The start-up of each channel’s output voltage VOUT is controlled by the voltage on the TRACK/SS pin (TRACK/SS1 for
channel 1, TRACK/SS2 for channel 2, SS3 for channel 3).
When the voltage on the TRACK/SS pin is less than the
3859alf
For more information www.linear.com/3859AL
15
LTC3859AL
operation
0.8V internal reference for the bucks and the 1.2V internal
reference for the boost, the LTC3859AL regulates the VFB
voltage to the TRACK/SS pin voltage instead of the corresponding reference voltage. This allows the TRACK/SS
pin to be used to program a soft-start by connecting an
external capacitor from the TRACK/SS pin to SGND. An
internal 5µA pull-up current charges this capacitor creating
a voltage ramp on the TRACK/SS pin. As the TRACK/SS
voltage rises linearly from 0V to 0.8V/1.2V (and beyond
up to INTVCC), the output voltage VOUT rises smoothly
from zero to its final value.
Alternatively the TRACK/SS pins for buck channels 1 and 2
can be used to cause the start-up of VOUT to track that of
another supply. Typically, this requires connecting to the
TRACK/SS pin an external resistor divider from the other
supply to ground (see the Applications Information section).
Light Load Current Operation (Burst Mode Operation,
Pulse-Skipping, or Continuous Conduction)
(PLLIN/MODE Pin)
The LTC3859AL can be enabled to enter high efficiency
Burst Mode operation, constant frequency pulse-skipping
mode or forced continuous conduction mode at low load
currents. To select Burst Mode operation, tie the PLLIN/
MODE pin to ground. To select forced continuous operation, tie the PLLIN/MODE pin to INTVCC. To select pulseskipping mode, tie the PLLIN/MODE pin to a DC voltage
greater than 1.2V and less than INTVCC – 1.3V.
When a controller is enabled for Burst Mode operation, the
minimum peak current in the inductor is set to approximately 25% of the maximum sense voltage (30% for the
boost) even though the voltage on the ITH pin indicates a
lower value. If the average inductor current is higher than
the load current, the error amplifier EA will decrease the
voltage on the ITH pin. When the ITH voltage drops below
0.425V, the internal sleep signal goes high (enabling sleep
mode) and both external MOSFETs are turned off. The ITH
pin is then disconnected from the output of the EA and
parked at 0.450V.
In sleep mode, much of the internal circuitry is turned off,
reducing the quiescent current that the LTC3859AL draws.
If channel 1 is in sleep mode and the other two are shut
down, the LTC3859AL draws only 28µA of quiescent cur-
rent. If channels 1 and 3 are in sleep mode and channel 2
is shut down, it draws only 33µA of quiescent current. If all
three controllers are enabled in sleep mode, the LTC3859AL
draws only 38µA of quiescent. In sleep mode, the load
current is supplied by the output capacitor. As the output
voltage decreases, the EA’s output begins to rise. When the
output voltage drops enough, the ITH pin is reconnected
to the output of the EA, the sleep signal goes low, and the
controller resumes normal operation by turning on the top
external MOSFET on the next cycle of the internal oscillator.
When a controller is enabled for Burst Mode operation,
the inductor current is not allowed to reverse. The reverse
current comparator (IR) turns off the bottom external
MOSFET (the top external MOSFET for the boost) just
before the inductor current reaches zero, preventing it
from reversing and going negative. Thus, the controller
operates in discontinuous operation.
In forced continuous operation or clocked by an external
clock source to use the phase-locked loop (see the Frequency Selection and Phase-Locked Loop section), the
inductor current is allowed to reverse at light loads or
under large transient conditions. The peak inductor current is determined by the voltage on the ITH pin, just as
in normal operation. In this mode, the efficiency at light
loads is lower than in Burst Mode operation. However,
continuous operation has the advantage of lower output
voltage ripple and less interference to audio circuitry. In
forced continuous mode, the output ripple is independent
of load current.
When the PLLIN/MODE pin is connected for pulse-skipping
mode, the LTC3859AL operates in PWM pulse-skipping
mode at light loads. In this mode, constant frequency
operation is maintained down to approximately 1% of
designed maximum output current. At very light loads, the
current comparator ICMP may remain tripped for several
cycles and force the external top MOSFET to stay off for
the same number of cycles (i.e., skipping pulses). The
inductor current is not allowed to reverse (discontinuous
operation). This mode, like forced continuous operation,
exhibits low output ripple as well as low audio noise and
reduced RF interference as compared to Burst Mode
operation. It provides higher low current efficiency than
forced continuous mode, but not nearly as high as Burst
Mode operation.
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LTC3859AL
Operation
Frequency Selection and Phase-Locked Loop
(FREQ and PLLIN/MODE Pins)
The selection of switching frequency is a tradeoff between
efficiency and component size. Low frequency operation increases efficiency by reducing MOSFET switching
losses, but requires larger inductance and/or capacitance
to maintain low output ripple voltage.
The switching frequency of the LTC3859AL’s controllers
can be selected using the FREQ pin.
If the PLLIN/MODE pin is not being driven by an external
clock source, the FREQ pin can be tied to SGND, tied to
INTVCC, or programmed through an external resistor.
Tying FREQ to SGND selects 350kHz while tying FREQ to
INTVCC selects 535kHz. Placing a resistor between FREQ
and SGND allows the frequency to be programmed between
50kHz and 900kHz.
A phase-locked loop (PLL) is available on the LTC3859AL
to synchronize the internal oscillator to an external clock
source that is connected to the PLLIN/MODE pin. The
LTC3859AL’s phase detector adjusts the voltage (through
an internal lowpass filter) of the VCO input to align the
turn-on of controller 1’s external top MOSFET to the rising edge of the synchronizing signal. Thus, the turn-on
of controller 2’s external top MOSFET is 180 degrees out
of phase to the rising edge of the external clock source.
The VCO input voltage is pre-biased to the operating
frequency set by the FREQ pin before the external clock
is applied. If prebiased near the external clock frequency,
the PLL loop only needs to make slight changes to the
VCO input in order to synchronize the rising edge of the
external clock’s to the rising edge of TG1. The ability to
pre-bias the loop filter allows the PLL to lock in rapidly
without deviating far from the desired frequency.
The typical capture range of the LTC3859AL’s phaselocked loop is from approximately 55kHz to 1MHz, with a
guarantee over all manufacturing variations to be between
75kHz and 850kHz. In other words, the LTC3859AL’s PLL
is guaranteed to lock to an external clock source whose
frequency is between 75kHz and 850kHz.
The typical input clock thresholds on the PLLIN/MODE
pin are 1.6V (rising) and 1.2V (falling).
Boost Controller Operation When VIN > VOUT
When the input voltage to the boost channel rises above
its regulated VOUT voltage, the controller can behave differently depending on the mode, inductor current and
VIN voltage. In forced continuous mode, the loop works
to keep the top MOSFET on continuously once VIN rises
above VOUT. An internal charge pump delivers current to
the boost capacitor from the BOOST3 pin to maintain a
sufficiently high TG voltage. (The amount of current the
charge pump can deliver is characterized by two curves
in the Typical Performance Characteristics section.)
In pulse-skipping mode, if VIN is between 100% and
110% of the regulated VOUT voltage, TG3 turns on if the
inductor current rises above approximately 3% of the
programmed ILIM current. If the part is programmed in
Burst Mode operation under this same VIN window, then
TG3 turns on at the same threshold current as long as
the chip is awake (one of the buck channels is awake and
switching). If both buck channels are asleep or shut down
in this VIN window, then TG3 will remain off regardless of
the inductor current.
If VIN rises above 110% of the regulated VOUT voltage in
any mode, the controller turns on TG3 regardless of the
inductor current. In Burst Mode operation, however, the
internal charge pump turns off if the entire chip is asleep
(the two buck channels are asleep or shut down). With
the charge pump off, there would be nothing to prevent
the boost capacitor from discharging, resulting in an
insufficient TG voltage needed to keep the top MOSFET
completely on. The charge pump turns back on when the
chip wakes up, and it remains on as long as one of the
buck channels is actively switching.
Boost Controller at Low SENSE Pin Common Voltage
The current comparator of the boost controller is powered
directly from the SENSE3+ pin and can operate to voltages
as low as 2.5V. Since this is lower than the VBIAS UVLO of
the chip, VBIAS can be connected to the output of the boost
controller, as illustrated in the typical application circuit
in Figure 12. This allows the boost controller to handle
input voltage transients down to 2.5V while maintaining
output voltage regulation. If the SENSE3+ rises back
above 2.5V, the SS3 pin will be released initiating a new
soft-start sequence.
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LTC3859AL
operation
Buck Controller Output Overvoltage Protection
Buck Foldback Current
The two buck channels have an overvoltage comparator
that guards against transient overshoots as well as other
more serious conditions that may overvoltage their outputs.
When the VFB1,2 pin rises by more than 10% above its
regulation point of 0.800V, the top MOSFET is turned off
and the bottom MOSFET is turned on until the overvoltage
condition is cleared.
When the buck output voltage falls to less than 70% of
its nominal level, foldback current limiting is activated,
progressively lowering the peak current limit in proportion
to the severity of the overcurrent or short-circuit condition.
Foldback current limiting is disabled during the soft-start
interval (as long as the VFB voltage is keeping up with
the TRACK/SS1,2 voltage). There is no foldback current
limiting for the boost channel.
Channel 1 Power Good (PGOOD1)
Channel 1 has a PGOOD1 pin that is connected to an open
drain of an internal N-channel MOSFET. The MOSFET
turns on and pulls the PGOOD1 pin low when the VFB1 pin
voltage is not within ±10% of the 0.8V reference voltage
for the buck channel. The PGOOD1 pin is also pulled low
when the RUN1 pin is low (shut down). When the VFB1
pin voltage is within the ±10% requirement, the MOSFET
is turned off and the pin is allowed to be pulled up by an
external resistor to a source no greater than 6V.
Boost Overvoltage Indicator (OV3)
The OV3 pin is an overvoltage indicator that signals whether
the output voltage of the channel 3 boost controller goes
over its programmed regulated voltage. The pin is connected to an open drain of an internal N-channel MOSFET.
The MOSFET turns on and pulls the OV3 pin low when the
VFB3 pin voltage is less than 110% of the 1.2V reference
voltage for the boost channel. The OV3 pin is also pulled
low when the RUN3 pin is low (shut down). When the VFB3
pin voltage goes higher than 110% of the 1.2V reference,
the MOSFET is turned off and the pin is allowed to be pulled
up by an external resistor to a source no greater than 6V.
THEORY AND BENEFITS OF 2-PHASE OPERATION
Why the need for 2-phase operation? Up until the 2-phase
family, constant-frequency dual switching regulators
operated both channels in phase (i.e., single-phase
operation). This means that both switches turned on at
the same time, causing current pulses of up to twice the
amplitude of those for one regulator to be drawn from the
input capacitor and battery. These large amplitude current
pulses increased the total RMS current flowing from the
input capacitor, requiring the use of more expensive input
capacitors and increasing both EMI and losses in the input
capacitor and battery.
With 2-phase operation, the two buck controllers of the
LTC3859AL are operated 180 degrees out of phase. This
effectively interleaves the current pulses drawn by the
switches, greatly reducing the overlap time where they add
together. The result is a significant reduction in total RMS
input current, which in turn allows less expensive input
capacitors to be used, reduces shielding requirements for
EMI and improves real world operating efficiency.
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LTC3859AL
Operation
5V SWITCH
20V/DIV
3.3V SWITCH
20V/DIV
INPUT CURRENT
5A/DIV
INPUT VOLTAGE
500mV/DIV
IIN(MEAS) = 2.53ARMS
IIN(MEAS) = 1.55ARMS
3859al F01a
(a)
3859al F01b
(b)
Figure 1. Input Waveforms Comparing Single-Phase (a) and 2-Phase (b) Operation for Dual Switching
Regulators Converting 12V to 5V and 3.3V at 3A Each. The Reduced Input Ripple with the 2-Phase Regulator
Allows Less Expensive Input Capacitors, Reduces Shielding Requirements for EMI and Improves Efficiency
Of course, the improvement afforded by 2-phase operation is a function of the dual switching regulator’s relative
duty cycles which, in turn, are dependent upon the input
voltage VIN (Duty Cycle = VOUT/VIN). Figure 2 shows how
the RMS input current varies for single-phase and 2-phase
operation for 3.3V and 5V regulators over a wide input
voltage range.
It can readily be seen that the advantages of 2-phase operation are not just limited to a narrow operating range,
for most applications is that 2-phase operation will reduce
the input capacitor requirement to that for just one channel operating at maximum current and 50% duty cycle.
The schematic on the first page is a basic LTC3859AL
application circuit. External component selection is driven
by the load requirement, and begins with the selection of
RSENSE and the inductor value. Next, the power MOSFETs
are selected. Finally, CIN and COUT are selected.
3.0
SINGLE-PHASE
DUAL CONTROLLER
2.5
INPUT RMS CURRENT (A)
Figure 1 compares the input waveforms for a representative
single-phase dual switching regulator to the 2-phase dual
buck controllers of the LTC3859AL. An actual measurement of the RMS input current under these conditions
shows that 2-phase operation dropped the input current
from 2.53ARMS to 1.55ARMS. While this is an impressive
reduction in itself, remember that the power losses are
proportional to IRMS2, meaning that the actual power wasted
is reduced by a factor of 2.66. The reduced input ripple
voltage also means less power is lost in the input power
path, which could include batteries, switches, trace/connector resistances and protection circuitry. Improvements
in both conducted and radiated EMI also directly accrue
as a result of the reduced RMS input current and voltage.
2.0
1.5
2-PHASE
DUAL CONTROLLER
1.0
0.5
0
VO1 = 5V/3A
VO2 = 3.3V/3A
0
10
20
30
INPUT VOLTAGE (V)
40
3859al F02
Figure 2. RMS Input Current Comparison
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LTC3859AL
Applications Information
The Typical Application on the first page is a basic
LTC3859AL application circuit. LTC3859AL can be configured to use either DCR (inductor resistance) sensing or low
value resistor sensing. The choice between the two current
sensing schemes is largely a design trade-off between
cost, power consumption, and accuracy. DCR sensing
is becoming popular because it saves expensive current
sensing resistors and is more power efficient, especially
in high current applications. However, current sensing
resistors provide the most accurate current limits for the
controller. Other external component selection is driven
by the load requirement, and begins with the selection of
RSENSE (if RSENSE is used) and inductor value. Next, the
power MOSFETs and Schottky diodes are selected. Finally,
input and output capacitors are selected.
Filter components mutual to the sense lines should be
placed close to the LTC3859AL, and the sense lines should
run close together to a Kelvin connection underneath the
current sense element (shown in Figure 3). Sensing current elsewhere can effectively add parasitic inductance
and capacitance to the current sense element, degrading
the information at the sense terminals and making the
programmed current limit unpredictable. If DCR sensing
is used (Figure 4b), sense resistor R1 should be placed
close to the switching node, to prevent noise from coupling
into sensitive small-signal nodes.
TO SENSE FILTER
NEXT TO THE CONTROLLER
SENSE+ and SENSE– Pins
CURRENT FLOW
The SENSE+ and SENSE– pins are the inputs to the current comparators.
Buck Controllers (SENSE1+/SENSE1–,SENSE2+/SENSE2–):
The common mode voltage range on these pins is 0V to
28V (absolute maximum), enabling the LTC3859AL to
regulate buck output voltages up to a nominal 24V (allowing margin for tolerances and transients). The SENSE+
pin is high impedance over the full common mode range,
drawing at most ±1µA. This high impedance allows the
current comparators to be used in inductor DCR sensing.
The impedance of the SENSE– pin changes depending on
the common mode voltage. When SENSE– is less than
INTVCC –0.5V, a small current of less than 1µA flows out
of the pin. When SENSE– is above INTVCC +0.5V, a higher
current (≈700µA) flows into the pin. Between INTVCC –0.5V
and INTVCC +0.5V, the current transitions from the smaller
current to the higher current.
Boost Controller (SENSE3+/SENSE3–): The common mode
input range for these pins is 2.5V to 38V, allowing the boost
converter to operate from inputs over this full range. The
SENSE3+ pin also provides power to the current comparator and draws about 170µA during normal operation (when
not shut down or asleep in Burst Mode operation). There is
a small bias current of less than 1µA that flows out of the
SENSE3– pin. This high impedance on the SENSE3– pin allows
the current comparator to be used in inductor DCR sensing.
INDUCTOR OR RSENSE
3859al F03
Figure 3. Sense Lines Placement with Inductor or Sense Resistor
Low Value Resistor Current Sensing
A typical sensing circuit using a discrete resistor is shown
in Figure 4a. RSENSE is chosen based on the required
output current.
The current comparators have a maximum threshold
VSENSE(MAX) of 50mV. The current comparator threshold
sets the peak of the inductor current, yielding a maximum
average output current, IMAX, equal to the peak value less
half the peak-to-peak ripple current, DIL. To calculate the
sense resistor value, use the equation:
RSENSE =
VSENSE(MAX)
∆I
IMAX + L
2
When using the buck controllers in very low dropout
conditions, the maximum output current level will be
reduced due to the internal compensation required to
meet stability criterion for buck regulators operating at
greater than 50% duty factor. A curve is provided in the
Typical Performance Characteristics section to estimate
this reduction in peak output current level depending upon
the operating duty factor.
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LTC3859AL
Applications Information
VIN1,2
(VOUT3)
INTVCC
BOOST
TG
LTC3859AL
SW
RSENSE
VOUT1,2
(VIN3)
BG
SENSE1,2+
(SENSE3–)
SENSE1, 2–
(SENSE3+)
CAP
PLACED NEAR SENSE PINS
across the external capacitor is equal to the drop across
the inductor DCR multiplied by R2/(R1 + R2). R2 scales the
voltage across the sense terminals for applications where
the DCR is greater than the target sense resistor value.
To properly dimension the external filter components, the
DCR of the inductor must be known. It can be measured
using a good RLC meter, but the DCR tolerance is not
always the same and varies with temperature; consult
the manufacturers’ data sheets for detailed information.
Using the inductor ripple current value from the Inductor
Value Calculation section, the target sense resistor value is:
SGND
R(EQUIV) =
3859al F04a
4a. Using a Resistor to Sense Current
VIN1,2
(VOUT3)
INTVCC
BOOST
INDUCTOR
TG
LTC3859AL
SW
L
DCR
VOUT1,2
(VIN3)
SENSE1, 2–
(SENSE3+)
R1
C1*
R2
SGND
To scale the maximum inductor DCR to the desired sense
resistor value, use the divider ratio:
3859al F04b
*PLACE C1 NEAR SENSE PINS
To ensure that the application will deliver full load current over the full operating temperature range, determine
RSENSE(EQUIV), keeping in mind that the maximum current
sense threshold (VSENSE(MAX)) for the LTC3859AL is fixed
at 50mV.
Next, determine the DCR of the inductor. Where provided,
use the manufacturer’s maximum value, usually given at
20°C. Increase this value to account for the temperature
coefficient of resistance, which is approximately 0.4%/°C.
A conservative value for TL(MAX) is 100°C.
BG
SENSE1, 2+
(SENSE3–)
VSENSE(MAX)
∆I
IMAX + L
2
(R1||R2) • C1 = L/DCR
RSENSE(EQ) = DCR(R2/(R1+R2))
4b. Using the Inductor DCR to Sense Current
Figure 4. Current Sensing Methods
Inductor DCR Sensing
For applications requiring the highest possible efficiency at
high load currents, the LTC3859AL is capable of sensing the
voltage drop across the inductor DCR, as shown in Figure 4b.
The DCR of the inductor represents the small amount of
DC winding resistance of the copper, which can be less
than 1mΩ for today’s low value, high current inductors.
In a high current application requiring such an inductor,
conduction loss through a sense resistor would cost several
points of efficiency compared to DCR sensing.
If the external R1||R2 • C1 time constant is chosen to be
exactly equal to the L/DCR time constant, the voltage drop
RD =
RSENSE(EQUIV)
DCRMAX at TL(MAX)
C1 is usually selected to be in the range of 0.1µF to 0.47µF.
This forces R1||R2 to around 2k, reducing error that might
have been caused by the SENSE+ pin’s ±1µA current.
The equivalent resistance R1||R2 is scaled to the room
temperature inductance and maximum DCR:
L
R1 R2 =
(DCR at 20°C) • C1
The sense resistor values are:
R1=
R1 R2
R1• RD
; R2 =
RD
1− RD
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LTC3859AL
applications information
The maximum power loss in R1 is related to duty cycle.
For the buck controllers, the maximum power loss will
occur in continuous mode at the maximum input voltage:
PLOSS R1=
(VIN(MAX) − VOUT ) • VOUT
R1
For the boost controller, the maximum power loss in R1
will occur in continuous mode at VIN = 1/2•VOUT :
PLOSS R1=
(VOUT(MAX) − VIN ) • VIN
R1
Ensure that R1 has a power rating higher than this value.
If high efficiency is necessary at light loads, consider this
power loss when deciding whether to use DCR sensing or
sense resistors. Light load power loss can be modestly
higher with a DCR network than with a sense resistor,
due to the extra switching losses incurred through R1.
However, DCR sensing eliminates a sense resistor, reduces
conduction losses and provides higher efficiency at heavy
loads. Peak efficiency is about the same with either method.
Inductor Value Calculation
The operating frequency and inductor selection are interrelated in that higher operating frequencies allow the use
of smaller inductor and capacitor values. So why would
anyone ever choose to operate at lower frequencies with
larger components? The answer is efficiency. A higher
frequency generally results in lower efficiency because
of MOSFET gate charge losses. In addition to this basic
trade-off, the effect of inductor value on ripple current and
low current operation must also be considered.
The inductor value has a direct effect on ripple current.
The inductor ripple current DIL decreases with higher
inductance or frequency. For the buck controllers, DIL
increases with higher VIN:
∆IL =
Accepting larger values of DIL allows the use of low
inductances, but results in higher output voltage ripple
and greater core losses. A reasonable starting point for
setting ripple current is DIL = 0.3(IMAX). The maximum
DIL occurs at the maximum input voltage for the bucks
and VIN = 1/2•VOUT for the boost.
The inductor value also has secondary effects. The transition to Burst Mode operation begins when the average
inductor current required results in a peak current below
25% of the current limit (30% for the boost) determined
by RSENSE. Lower inductor values (higher DIL) will cause
this to occur at lower load currents, which can cause a dip
in efficiency in the upper range of low current operation. In
Burst Mode operation, lower inductance values will cause
the burst frequency to decrease.
Inductor Core Selection
Once the value for L is known, the type of inductor must
be selected. High efficiency converters generally cannot
afford the core loss found in low cost powdered iron cores,
forcing the use of more expensive ferrite or molypermalloy
cores. Actual core loss is independent of core size for a
fixed inductor value, but it is very dependent on inductance
selected. As inductance increases, core losses go down.
Unfortunately, increased inductance requires more turns
of wire and therefore copper losses will increase.
Ferrite designs have very low core loss and are preferred
at high switching frequencies, so design goals can concentrate on copper loss and preventing saturation. Ferrite
core material saturates “hard,” which means that inductance collapses abruptly when the peak design current is
exceeded. This results in an abrupt increase in inductor
ripple current and consequent output voltage ripple. Do
not allow the core to saturate!
V
1
VOUT 1− OUT
(f)(L)
VIN
For the boost controller, the inductor ripple current DIL
increases with higher VOUT:
∆IL =
22
1
V
VIN 1− IN
(f)(L) VOUT
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Applications Information
Power MOSFET and Schottky Diode
(Optional) Selection
The MOSFET power dissipations at maximum output
current are given by:
Two external power MOSFETs must be selected for each
controller in the LTC3859AL: one N-channel MOSFET for
the top switch (main switch for the buck, synchronous
for the boost), and one N-channel MOSFET for the bottom switch (main switch for the boost, synchronous for
the buck).
2
VOUT
IOUT(MAX) (1+ δ) RDS(ON) +
VIN
IOUT(MAX)
(VIN )2
(RDR )(CMILLER ) •
2
1
1
+
(f)
VINTVCC − VTHMIN VTHMIN
2
V −V
PSYNC _ BUCK = IN OUT IOUT(MAX) (1+ δ) RDS(ON)
VIN
The peak-to-peak drive levels are set by the INTVCC voltage.
This voltage is typically 5.4V during start-up (see EXTVCC
Pin Connection). Consequently, logic-level threshold
MOSFETs must be used in most applications. Pay close
attention to the BVDSS specification for the MOSFETs as
well; many of the logic level MOSFETs are limited to 30V
or less.
Selection criteria for the power MOSFETs include the
on-resistance RDS(ON), Miller capacitance CMILLER, input
voltage and maximum output current. Miller capacitance,
CMILLER, can be approximated from the gate charge curve
usually provided on the MOSFET manufacturers’ data
sheet. CMILLER is equal to the increase in gate charge
along the horizontal axis while the curve is approximately
flat divided by the specified change in VDS. This result is
then multiplied by the ratio of the application applied VDS
to the gate charge curve specified VDS. When the IC is
operating in continuous mode the duty cycles for the top
and bottom MOSFETs are given by:
V
Buck Main Switch Duty Cycle = OUT
VIN
V − VOUT
Buck Sync Switch Duty Cycle = IN
VIN
− VIN
V
Boost Main Switch Duty Cycle = OUT
VOUT
Boost Sync Switch Duty Cycle =
VIN
VOUT
(
PMAIN _ BUCK =
)
(
PMAIN _ BOOST =
)
( VOUT − VIN ) VOUT
VIN2
(IOUT(MAX) )
2
•
VOUT2 IOUT(MAX)
(1+ δ )RDS(ON) +
•
2
VIN
1
1
+
(f)
INTVCC − VTHMIN VTHMIN
2
V
= IN IOUT(MAX) (1+ δ )RDS(ON)
VOUT
(RDR )(CMILLER ) • V
PSYNC _ BOOST
(
)
where z is the temperature dependency of RDS(ON) and
RDR (approximately 2Ω) is the effective driver resistance
at the MOSFET’s Miller threshold voltage. VTHMIN is the
typical MOSFET minimum threshold voltage.
Both MOSFETs have I2R losses while the main N-channel
equations for the buck and boost controllers include an
additional term for transition losses, which are highest at
high input voltages for the bucks and low input voltages for
the boost. For VIN < 20V (high VIN for the boost) the high
current efficiency generally improves with larger MOSFETs,
while for VIN > 20V (low VIN for the boost) the transition
losses rapidly increase to the point that the use of a higher
RDS(ON) device with lower CMILLER actually provides higher
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LTC3859AL
applications information
efficiency. The synchronous MOSFET losses for the buck
controllers are greatest at high input voltage when the top
switch duty factor is low or during a short-circuit when
the synchronous switch is on close to 100% of the period.
The synchronous MOSFET losses for the boost controller are greatest when the input voltage approaches the
output voltage or during an overvoltage event when the
synchronous switch is on 100% of the period.
The term (1+ z) is generally given for a MOSFET in the
form of a normalized RDS(ON) vs Temperature curve, but
z = 0.005/°C can be used as an approximation for low
voltage MOSFETs.
In a boost converter, the output has a discontinuous current,
so COUT must be capable of reducing the output voltage
ripple. The effects of ESR (equivalent series resistance) and
the bulk capacitance must be considered when choosing
the right capacitor for a given output ripple voltage. The
steady ripple due to charging and discharging the bulk
capacitance is given by:
Ripple =
(
I OUT(MAX) • VOUT − VIN(MIN)
COUT • VOUT • f
)V
where COUT is the output filter capacitor.
The steady ripple due to the voltage drop across the ESR
is given by:
The optional Schottky diodes D4, D5, and D6 shown in
Figure 13 conduct during the dead-time between the
conduction of the two power MOSFETs. This prevents
the body diode of the synchronous MOSFET from turning
on, storing charge during the dead-time and requiring a
reverse recovery period that could cost as much as 3%
in efficiency at high VIN. A 1A to 3A Schottky is generally
a good compromise for both regions of operation due to
the relatively small average current. Larger diodes result
in additional transition losses due to their larger junction
capacitance.
Multiple capacitors placed in parallel may be needed to
meet the ESR and RMS current handling requirements.
Dry tantalum, special polymer, aluminum electrolytic
and ceramic capacitors are all available in surface mount
packages. Ceramic capacitors have excellent low ESR
characteristics but can have a high voltage coefficient.
Capacitors are now available with low ESR and high ripple
current ratings such as OS-CON and POSCAP.
Boost CIN, COUT Selection
Buck CIN, COUT Selection
The input ripple current in a boost converter is relatively
low (compared with the output ripple current), because
this current is continuous. The boost input capacitor CIN
voltage rating should comfortably exceed the maximum
input voltage. Although ceramic capacitors can be relatively
tolerant of overvoltage conditions, aluminum electrolytic
capacitors are not. Be sure to characterize the input voltage
for any possible overvoltage transients that could apply
excess stress to the input capacitors.
The selection of CIN for the two buck controllers is simplified
by the 2-phase architecture and its impact on the worstcase RMS current drawn through the input network (battery/fuse/capacitor). It can be shown that the worst-case
capacitor RMS current occurs when only one controller
is operating. The controller with the highest (VOUT)(IOUT)
product needs to be used in the formula shown in Equation (1) to determine the maximum RMS capacitor current
requirement. Increasing the output current drawn from
the other controller will actually decrease the input RMS
ripple current from its maximum value. The out-of-phase
technique typically reduces the input capacitor’s RMS
ripple current by a factor of 30% to 70% when compared
to a single phase power supply solution.
The value of CIN is a function of the source impedance, and
in general, the higher the source impedance, the higher the
required input capacitance. The required amount of input
capacitance is also greatly affected by the duty cycle. High
output current applications that also experience high duty
cycles can place great demands on the input supply, both
in terms of DC current and ripple current.
DVESR = IL(MAX) • ESR
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LTC3859AL
Applications Information
In continuous mode, the source current of the top MOSFET
is a square wave of duty cycle (VOUT)/(VIN). To prevent
large voltage transients, a low ESR capacitor sized for the
maximum RMS current of one channel must be used. The
maximum RMS capacitor current is given by:
CIN Required IRMS ≈
IMAX
VIN
( VOUT ) ( VIN − VOUT )1/ 2 (1)
This formula has a maximum at VIN = 2VOUT, where IRMS
= IOUT/2. This simple worst-case condition is commonly
used for design because even significant deviations do not
offer much relief. Note that capacitor manufacturers’ ripple
current ratings are often based on only 2000 hours of life.
This makes it advisable to further derate the capacitor, or
to choose a capacitor rated at a higher temperature than
required. Several capacitors may be paralleled to meet
size or height requirements in the design. Due to the high
operating frequency of the LTC3859AL, ceramic capacitors
can also be used for CIN. Always consult the manufacturer
if there is any question.
The benefit of the LTC3859AL 2-phase operation can be
calculated by using Equation (1) for the higher power controller and then calculating the loss that would have resulted
if both controller channels switched on at the same time.
The total RMS power lost is lower when both controllers
are operating due to the reduced overlap of current pulses
required through the input capacitor’s ESR. This is why
the input capacitor’s requirement calculated above for the
worst-case controller is adequate for the dual controller
design. Also, the input protection fuse resistance, battery
resistance, and PC board trace resistance losses are also
reduced due to the reduced peak currents in a 2-phase
system. The overall benefit of a multiphase design will
only be fully realized when the source impedance of the
power supply/battery is included in the efficiency testing.
The drains of the top MOSFETs should be placed within
1cm of each other and share a common CIN (s). Separating the drains and CIN may produce undesirable voltage
and current resonances at VIN.
A small (0.1µF to 1µF) bypass capacitor between the chip
VIN pin and ground, placed close to the LTC3859AL, is also
suggested. A small (1Ω to 10Ω) resistor placed between
CIN (C1) and the VIN pin provides further isolation between
the two channels.
The selection of COUT is driven by the effective series
resistance (ESR). Typically, once the ESR requirement
is satisfied, the capacitance is adequate for filtering. The
output ripple (DVOUT) is approximated by:
1
∆VOUT ≈ ∆IL ESR +
8fCOUT
where f is the operating frequency, COUT is the output
capacitance and DIL is the ripple current in the inductor.
The output ripple is highest at maximum input voltage
since DIL increases with input voltage.
Setting Output Voltage
The LTC3859AL output voltages are each set by an external
feedback resistor divider carefully placed across the output,
as shown in Figure 5. The regulated output voltages are
determined by:
R
VOUT, BUCK = 0.8V 1+ B
RA
R
VOUT, BOOST = 1.2V 1+ B
RA
To improve the frequency response, a feedforward capacitor, CFF, may be used. Great care should be taken to
route the VFB line away from noise sources, such as the
inductor or the SW line.
VOUT
1/3 LTC3859AL
RB
CFF
VFB
3859al F05
RA
Figure 5. Setting Output Voltage
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LTC3859AL
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Tracking and Soft-Start
(TRACK/SS1, TRACK/SS2, SS3 Pins)
1/3 LTC3859AL
TRACK/SS
0.8V
5µA
1.2V
= CSS •
5µA
tSS _ BUCK = CSS •
tSS _ BOOST
Figure 6. Using the TRACK/SS Pin to Program Soft-Start
VX(MASTER)
VOUT(SLAVE)
TIME
3859al F07a
7a. Coincident Tracking
VX(MASTER)
VOUT(SLAVE)
TIME
Alternatively, the TRACK/SS1 and TRACK/SS2 pins for the
two buck controllers can be used to track two (or more) supplies during start-up, as shown qualitatively in Figures 7a
and 7b. To do this, a resistor divider should be connected
from the master supply (VX) to the TRACK/SS pin of the
slave supply (VOUT), as shown in Figure 8. During start-up
VOUT will track VX according to the ratio set by the resistor divider:
3859al F06
OUTPUT (VOUT)
Soft-start is enabled by simply connecting a capacitor
from the TRACK/SS pin to ground, as shown in Figure 6.
An internal 5µA current source charges the capacitor,
providing a linear ramping voltage at the TRACK/SS pin.
The LTC3859AL will regulate the VFB pin (and hence VOUT)
according to the voltage on the TRACK/SS pin, allowing
VOUT to rise smoothly from 0V to its final regulated value.
The total soft-start time will be approximately:
SGND
OUTPUT (VOUT)
The start-up of each VOUT is controlled by the voltage on
the respective TRACK/SS pin (TRACK/SS1 for channel 1,
TRACK/SS2 for channel 2, SS3 for channel 3). When the
voltage on the TRACK/SS pin is less than the internal
0.8V reference (1.2V reference for the boost channel), the
LTC3859AL regulates the VFB pin voltage to the voltage
on the TRACK/SS pin instead of the internal reference.
Likewise, the TRACK/SS pin for the buck channels can be
used to program an external soft-start function or to allow
VOUT to track another supply during start-up.
CSS
3859al F07b
7b. Radiometric Tracking
Figure 7. Two Different Modes of Output Voltage Tracking
VOUT
RB
LTC3859AL
VFB1,2
+ RTRACKB
R
RA
VX
=
• TRACKA
VOUT RTRACKA
RA + RB
VX
RA
RTRACKB
For coincident tracking (VOUT = VX during start-up),
TRACK/SS1,2
RA = RTRACKA
RTRACKA
RB = RTRACKB
3859al F08
Figure 8. Using the TRACK/SS Pin for Tracking
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INTVCC Regulators
The LTC3859AL features two separate internal P-channel
low dropout linear regulators (LDO) that supply power
at the INTVCC pin from either the VBIAS supply pin or the
EXTVCC pin depending on the connection of the EXTVCC
pin. INTVCC powers the gate drivers and much of the
LTC3859AL’s internal circuitry. The VBIAS LDO and the
EXTVCC LDO regulate INTVCC to 5.4V. Each of these must
be bypassed to ground with a minimum of 4.7µF ceramic
capacitor. No matter what type of bulk capacitor is used, an
additional 1µF ceramic capacitor placed directly adjacent
to the INTVCC and PGND IC pins is highly recommended.
Good bypassing is needed to supply the high transient
currents required by the MOSFET gate drivers and to
prevent interaction between the channels.
High input voltage applications in which large MOSFETs
are being driven at high frequencies may cause the maximum junction temperature rating for the LTC3859AL to be
exceeded. The INTVCC current, which is dominated by the
gate charge current, may be supplied by either the VBIAS
LDO or the EXTVCC LDO. When the voltage on the EXTVCC
pin is less than 4.7V, the VBIAS LDO is enabled. Power
dissipation for the IC in this case is highest and is equal to
VBIAS • IINTVCC. The gate charge current is dependent
on operating frequency as discussed in the Efficiency
Considerations section. The junction temperature can be
estimated by using the equations given in Note 2 of the
Electrical Characteristics. For example, the LTC3859AL
INTVCC current is limited to less than 40mA from a 40V
supply when not using the EXTVCC supply at a 70°C ambient temperature in the QFN package:
TJ = 70°C + (40mA)(40V)(34°C/W) = 125°C
To prevent the maximum junction temperature from being
exceeded, the input supply current must be checked while
operating in continuous conduction mode (PLLIN/MODE
= INTVCC) at maximum VIN.
When the voltage applied to EXTVCC rises above 4.7V, the
VBIAS LDO is turned off and the EXTVCC LDO is enabled.
The EXTVCC LDO remains on as long as the voltage applied
to EXTVCC remains above 4.5V. The EXTVCC LDO attempts
to regulate the INTVCC voltage to 5.4V, so while EXTVCC
is less than 5.4V, the LDO is in dropout and the INTVCC
voltage is approximately equal to EXTVCC. When EXTVCC
is greater than 5.4V, up to an absolute maximum of 14V,
INTVCC is regulated to 5.4V.
Using the EXTVCC LDO allows the MOSFET driver and
control power to be derived from one of the LTC3859AL’s
switching regulator outputs (4.7V ≤ VOUT ≤ 14V) during normal operation and from the VBIAS LDO when the
output is out of regulation (e.g., startup, short-circuit). If
more current is required through the EXTVCC LDO than
is specified, an external Schottky diode can be added
between the EXTVCC and INTVCC pins. In this case, do
not apply more than 6V to the EXTVCC pin and make sure
that EXTVCC ≤ VBIAS.
Significant efficiency and thermal gains can be realized
by powering INTVCC from the buck output, since the VIN
current resulting from the driver and control currents will
be scaled by a factor of (Duty Cycle)/(Switcher Efficiency).
For 5V to 14V regulator outputs, this means connecting
the EXTVCC pin directly to VOUT. Tying the EXTVCC pin to
a 8.5V supply reduces the junction temperature in the
previous example from 125°C to:
TJ = 70°C + (40mA)(8.5V)(34°C/W) = 82°C
However, for 3.3V and other low voltage outputs, additional
circuitry is required to derive INTVCC power from the output.
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LTC3859AL
Applications Information
The following list summarizes the four possible connections for EXTVCC:
1.EXTVCC grounded. This will cause INTVCC to be powered
from the internal 5.4V regulator resulting in an efficiency
penalty of up to 10% at high input voltages.
2.EXTVCC connected directly to the output voltage of one
of the buck regulators. This is the normal connection
for a 5V to 14V regulator and provides the highest efficiency.
3.EXTVCC connected to an external supply. If an external
supply is available in the 5V to 14V range, it may be
used to power EXTVCC providing it is compatible with
the MOSFET gate drive requirements. Ensure that
EXTVCC < VBIAS.
4.EXTVCC connected to an output-derived boost network
off one of the buck regulators. For 3.3V and other low
voltage buck regulators, efficiency gains can still be
realized by connecting EXTVCC to an output-derived
voltage that has been boosted to greater than 4.7V. This
can be done with the capacitive charge pump shown in
Figure 9. Ensure that EXTVCC < VBIAS.
VIN1,2
C1
LTC3859AL
BAT85
BAT85
MTOP
BAT85
TG
EXTVCC
L
SW
RSENSE
VOUT1,2
MBOT
BG
PGND
3859al F09
Figure 9. Capacitive Charge Pump for EXTVCC
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Topside MOSFET Driver Supply (CB, DB)
External bootstrap capacitors CB connected to the BOOST
pins supply the gate drive voltages for the topside MOSFETs. Capacitor CB in the Functional Diagram is charged
though external diode DB from INTVCC when the SW pin
is low. When one of the topside MOSFETs is to be turned
on, the driver places the CB voltage across the gate-source
of the desired MOSFET. This enhances the MOSFET and
turns on the topside switch. The switch node voltage,
SW, rises to VIN for the buck channels (VOUT for the boost
channel) and the BOOST pin follows. With the topside
MOSFET on, the boost voltage is above the input supply:
VBOOST = VIN + VINTVCC (VBOOST = VOUT + VINTVCC for the
boost controller). The value of the boost capacitor CB
needs to be 100 times that of the total input capacitance
of the topside MOSFET(s). The reverse breakdown of the
external Schottky diode must be greater than VIN(MAX) for
the buck channels and VOUT(MAX) for the boost channel.
The external diode DB can be a Schottky diode or silicon
diode, but in either case it should have low leakage and fast
recovery. Pay close attention to the reverse leakage at high
temperatures where it generally increases substantially.
The topside MOSFET driver for the boost channel includes
an internal charge pump that delivers current to the
bootstrap capacitor from the BOOST3 pin. This charge
current maintains the bias voltage required to keep the
top MOSFET on continuously during dropout/overvoltage conditions. The Schottky/silicon diode selected for
the boost topside driver should have a reverse leakage
less than the available output current the charge pump
can supply. Curves displaying the available charge pump
current under different operating conditions can be found
in the Typical Performance Characteristics section.
A leaky diode DB in the boost converter can not only
prevent the top MOSFET from fully turning on but it can
also completely discharge the bootstrap capacitor CB and
create a current path from the input voltage to the BOOST3
pin to INTVCC. This can cause INTVCC to rise if the diode
leakage exceeds the current consumption on INTVCC . This
is particularly a concern in Burst Mode operation where
the load on INTVCC can be very small. There is an internal
voltage clamp on INTVCC that prevents the INTVCC voltage
from running away, but this clamp should be regarded
as a failsafe only. The external Schottky or silicon diode
should be carefully chosen such that INTVCC never gets
charged up much higher than its normal regulation voltage.
Care should also be taken when choosing the external
diode DB for the buck converters. A leaky diode not only
increases the quiescent current of the buck converter, but
it can also cause a similar leakage path to INTVCC from
VOUT for applications with output voltages greater than
the INTVCC voltage (~5.4V).
1000
900
FREQUENCY (kHz)
800
700
600
500
400
300
200
100
0
15 25 35 45 55 65 75 85 95 105 115 125
FREQ PIN RESISTOR (kΩ)
3859al F10
Figure 10. Relationship Between Oscillator
Frequency and Resistor Value at the FREQ Pin
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LTC3859AL
Applications Information
Fault Conditions: Buck Current Limit and Current
Foldback
The LTC3859AL includes current foldback for the buck
channels to help limit load current when the output is
shorted to ground. If the buck output falls below 70% of
its nominal output level, then the maximum sense voltage is progressively lowered from 100% to 40% of its
maximum selected value. Under short-circuit conditions
with very low duty cycles, the buck channel will begin
cycle skipping in order to limit the short-circuit current.
In this situation the bottom MOSFET will be dissipating
most of the power but less than in normal operation. The
short-circuit ripple current is determined by the minimum
on-time tON(MIN) of the LTC3859AL (≈95ns), the input
voltage and inductor value:
DIL(SC) = tON(MIN) (VIN/L)
The resulting average short-circuit current is:
1
I = 40% • I
LIM(MAX) − ∆IL(SC)
SC
2
Fault Conditions: Buck Overvoltage Protection
(Crowbar)
The overvoltage crowbar is designed to blow a system
input fuse when the output voltage of the one of the buck
regulators rises much higher than nominal levels. The
crowbar causes huge currents to flow, that blow the fuse
to protect against a shorted top MOSFET if the short occurs while the controller is operating.
A comparator monitors the buck output for overvoltage
conditions. The comparator detects faults greater than
10% above the nominal output voltage. When this condition is sensed, the top MOSFET of the buck controller is
turned off and the bottom MOSFET is turned on until the
overvoltage condition is cleared. The bottom MOSFET
remains on continuously for as long as the overvoltage
condition persists; if VOUT returns to a safe level, normal
operation automatically resumes.
A shorted top MOSFET for the buck channel will result in
a high current condition which will open the system fuse.
The switching regulator will regulate properly with a leaky
top MOSFET by altering the duty cycle to accommodate
the leakage.
Fault Conditions: Over Temperature Protection
At higher temperatures, or in cases where the internal
power dissipation causes excessive self heating on chip
(such as INTVCC short to ground), the over temperature
shutdown circuitry will shut down the LTC3859AL. When
the junction temperature exceeds approximately 170°C, the
over temperature circuitry disables the INTVCC LDO, causing the INTVCC supply to collapse and effectively shutting
down the entire LTC3859AL chip. Once the junction temperature drops back to approximately 155°C, the INTVCC
LDO turns back on. Long term overstress (TJ > 125°C)
should be avoided as it can degrade the performance or
shorten the life of the part.
Phase-Locked Loop and Frequency Synchronization
The LTC3859AL has an internal phase-locked loop (PLL)
comprised of a phase frequency detector, a lowpass filter,
and a voltage-controlled oscillator (VCO). This allows the
turn-on of the top MOSFET of controller 1 to be locked to
the rising edge of an external clock signal applied to the
PLLIN/MODE pin. The turn-on of controller 2’s top MOSFET
is thus 180 degrees out of phase with the external clock.
The phase detector is an edge sensitive digital type that
provides zero degrees phase shift between the external
and internal oscillators. This type of phase detector does
not exhibit false lock to harmonics of the external clock.
If the external clock frequency is greater than the internal oscillator’s frequency, fOSC, then current is sourced
continuously from the phase detector output, pulling up
the VCO input. When the external clock frequency is less
than fOSC, current is sunk continuously, pulling down the
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VCO input. If the external and internal frequencies are the
same but exhibit a phase difference, the current sources
turn on for an amount of time corresponding to the phase
difference. The voltage at the VCO input is adjusted until
the phase and frequency of the internal and external oscillators are identical. At the stable operating point, the
phase detector output is high impedance and the internal
filter capacitor, CLP, holds the voltage at the VCO input.
Note that the LTC3859AL can only be synchronized to
an external clock whose frequency is within range of the
LTC3859AL’s internal VCO, which is nominally 55kHz to
1MHz. This is guaranteed to be between 75kHz and 850kHz.
Typically, the external clock (on PLLIN/MODE pin) input
high threshold is 1.6V, while the input low threshold is 1.2V.
Rapid phase-locking can be achieved by using the FREQ
pin to set a free-running frequency near the desired
synchronization frequency. The VCO’s input voltage is
prebiased at a frequency correspond to the frequency
set by the FREQ pin. Once prebiased, the PLL only needs
to adjust the frequency slightly to achieve phase-lock
and synchronization. Although it is not required that the
free-running frequency be near external clock frequency,
doing so will prevent the operating frequency from passing through a large range of frequencies as the PLL locks.
Table 1 summarizes the different states in which the FREQ
pin can be used.
Minimum On-Time Considerations
Minimum on-time tON(MIN) is the smallest time duration that
the LTC3859AL is capable of turning on the top MOSFET
(bottom MOSFET for the boost controller). It is determined
by internal timing delays and the gate charge required to
turn on the top MOSFET. Low duty cycle applications may
approach this minimum on-time limit and care should be
taken to ensure that
tON(MIN)_ BUCK <
VOUT
VIN (f)
tON(MIN)_ BOOST <
VOUT − VIN
VOUT (f)
If the duty cycle falls below what can be accommodated
by the minimum on-time, the controller will begin to skip
cycles. The output voltage will continue to be regulated,
but the ripple voltage and current will increase.
The minimum on-time for the LTC3859AL is approximately
95ns for the bucks and 120ns for the boost. However, as
the peak sense voltage decreases the minimum on-time
gradually increases up to about 130ns. This is of particular concern in forced continuous applications with low
ripple current at light loads. If the duty cycle drops below
the minimum on-time limit in this situation, a significant
amount of cycle skipping can occur with correspondingly
larger current and voltage ripple.
Table 1
FREQ PIN
PLLIN/MODE PIN
FREQUENCY
0V
DC Voltage
350kHz
INTVCC
DC Voltage
535kHz
Resistor to SGND
DC Voltage
50kHz to 900kHz
Any of the Above
External Clock
Phase-Locked to
External Clock
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Efficiency Considerations
The percent efficiency of a switching regulator is equal to
the output power divided by the input power times 100%.
It is often useful to analyze individual losses to determine
what is limiting the efficiency and which change would
produce the most improvement. Percent efficiency can
be expressed as:
%Efficiency = 100% – (L1 + L2 + L3 + ...)
where L1, L2, etc. are the individual losses as a percentage of input power.
Although all dissipative elements in the circuit produce
losses, four main sources usually account for most of
the losses in LTC3859AL circuits: 1) IC VBIAS current,
2) INTVCC regulator current, 3) I2R losses, 4) Topside
MOSFET transition losses.
1.The VBIAS current is the DC supply current given in the
Electrical Characteristics table, which excludes MOSFET driver and control currents. VBIAS current typically
results in a small (<0.1%) loss.
2.INTVCC current is the sum of the MOSFET driver and
control currents. The MOSFET driver current results
from switching the gate capacitance of the power
MOSFETs. Each time a MOSFET gate is switched from
low to high to low again, a packet of charge, dQ, moves
from INTVCC to ground. The resulting dQ/dt is a current
out of INTVCC that is typically much larger than the
control circuit current. In continuous mode, IGATECHG
= f(QT + QB), where QT and QB are the gate charges of
the topside and bottom side MOSFETs.
Supplying INTVCC from an output-derived source power
through EXTVCC will scale the VIN current required for
the driver and control circuits by a factor of (Duty Cycle)/
(Efficiency). For example, in a 20V to 5V application,
10mA of INTVCC current results in approximately 2.5mA
of VIN current. This reduces the mid-current loss from
10% or more (if the driver was powered directly from
VIN) to only a few percent.
3.I2R losses are predicted from the DC resistances of the
fuse (if used), MOSFET, inductor, current sense resistor, and input and output capacitor ESR. In continuous
mode the average output current flows through L and
RSENSE, but is “chopped” between the topside MOSFET
and the synchronous MOSFET. If the two MOSFETs have
approximately the same RDS(ON), then the resistance
of one MOSFET can simply be summed with the resistances of L, RSENSE and ESR to obtain I2R losses. For
example, if each RDS(ON) = 30mΩ, RL = 50mΩ, RSENSE
= 10mΩ and RESR = 40mΩ (sum of both input and
output capacitance losses), then the total resistance
is 130mΩ. This results in losses ranging from 3% to
13% as the output current increases from 1A to 5A for
a 5V output, or a 4% to 20% loss for a 3.3V output.
Efficiency varies as the inverse square of VOUT for the
same external components and output power level. The
combined effects of increasingly lower output voltages
and higher currents required by high performance digital
systems is not doubling but quadrupling the importance
of loss terms in the switching regulator system!
4.Transition losses apply only to the top MOSFET(s) (bottom MOSFET for the boost), and become significant only
when operating at high input voltages (typically 15V or
greater). Transition losses can be estimated from:
Transition Loss = (1.7)VIN2 • IO(MAX) • CRSS • f
Other hidden losses such as copper trace and internal
battery resistances can account for an additional 5%
to 10% efficiency degradation in portable systems. It is
very important to include these “system” level losses
during the design phase. The internal battery and fuse
resistance losses can be minimized by making sure that
CIN has adequate charge storage and very low ESR at
the switching frequency. A 25W supply will typically
require a minimum of 20µF to 40µF of capacitance
having a maximum of 20mΩ to 50mΩ of ESR. The
LTC3859AL 2-phase architecture typically halves this
input capacitance requirement over competing solutions. Other losses including Schottky conduction losses
during dead-time and inductor core losses generally
account for less than 2% total additional loss.
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Checking Transient Response
The regulator loop response can be checked by looking at
the load current transient response. Switching regulators
take several cycles to respond to a step in DC (resistive)
load current. When a load step occurs, VOUT shifts by an
amount equal to DILOAD(ESR), where ESR is the effective
series resistance of COUT. DILOAD also begins to charge or
discharge COUT generating the feedback error signal that
forces the regulator to adapt to the current change and
return VOUT to its steady-state value. During this recovery
time VOUT can be monitored for excessive overshoot or
ringing, which would indicate a stability problem. OPTILOOP compensation allows the transient response to be
optimized over a wide range of output capacitance and
ESR values. The availability of the ITH pin not only allows
optimization of control loop behavior, but it also provides
a DC coupled and AC filtered closed loop response test
point. The DC step, rise time and settling at this test
point truly reflects the closed loop response. Assuming a
predominantly second order system, phase margin and/
or damping factor can be estimated using the percentage
of overshoot seen at this pin. The bandwidth can also be
estimated by examining the rise time at the pin. The ITH
external components shown in Figure 16 will provide an
adequate starting point for most applications.
The ITH series RC-CC filter sets the dominant pole-zero
loop compensation. The values can be modified slightly
(from 0.5 to 2 times their suggested values) to optimize
transient response once the final PC layout is done and
the particular output capacitor type and value have been
determined. The output capacitors need to be selected
because the various types and values determine the loop
gain and phase. An output current pulse of 20% to 80%
of full-load current having a rise time of 1µs to 10µs will
produce output voltage and ITH pin waveforms that will
give a sense of the overall loop stability without breaking
the feedback loop.
Placing a power MOSFET directly across the output capacitor and driving the gate with an appropriate signal
generator is a practical way to produce a realistic load step
condition. The initial output voltage step resulting from
the step change in output current may not be within the
bandwidth of the feedback loop, so this signal cannot be
used to determine phase margin. This is why it is better
to look at the ITH pin signal which is in the feedback loop
and is the filtered and compensated control loop response.
The gain of the loop will be increased by increasing RC
and the bandwidth of the loop will be increased by decreasing CC. If RC is increased by the same factor that
CC is decreased, the zero frequency will be kept the same,
thereby keeping the phase shift the same in the most
critical frequency range of the feedback loop. The output
voltage settling behavior is related to the stability of the
closed-loop system and will demonstrate the actual overall
supply performance.
A second, more severe transient is caused by switching
in loads with large (>1µF) supply bypass capacitors. The
discharged bypass capacitors are effectively put in parallel
with COUT, causing a rapid drop in VOUT. No regulator can
alter its delivery of current quickly enough to prevent this
sudden step change in output voltage if the load switch
resistance is low and it is driven quickly. If the ratio of
CLOAD to COUT is greater than 1:50, the switch rise time
should be controlled so that the load rise time is limited
to approximately 25 • CLOAD. Thus a 10µF capacitor would
require a 250µs rise time, limiting the charging current
to about 200mA.
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Applications Information
Buck Design Example
As a design example for one of the buck channels channel,
assume VIN = 12V(NOMINAL), VIN = 22V(MAX), VOUT = 3.3V,
IMAX = 6A, VSENSE(MAX) = 50mV, and f = 350kHz.
The inductance value is chosen first based on a 30% ripple
current assumption. The highest value of ripple current
occurs at the maximum input voltage. Tie the FREQ pin
to GND, generating 350kHz operation. The minimum
inductance for 30% ripple current is:
∆IL =
VOUT
VOUT
1−
(f)(L) VIN(NOMINAL)
A 3.9µH inductor will produce 29% ripple current. The
peak inductor current will be the maximum DC value plus
one half the ripple current, or 6.88A. Increasing the ripple
current will also help ensure that the minimum on-time
of 95ns is not violated. The minimum on-time occurs at
maximum VIN:
tON(MIN) =
VOUT
VIN(MAX)(f)
=
3.3V
= 429ns
22V(350kHz)
The RSENSE resistor value can be calculated by using the
minimum value for the maximum current sense threshold
(43mV):
RSENSE ≤
43mV
= 0.006Ω
6.88A
Choosing 1% resistors: RA = 25k and RB = 80.6k yields
an output voltage of 3.33V.
The power dissipation on the top side MOSFET can be easily
estimated. Choosing a Fairchild FDS6982S dual MOSFET
results in: RDS(ON) = 0.035Ω/0.022Ω, CMILLER = 215pF.
At maximum input voltage with T(estimated) = 50°C:
3.3V
(6A)2 {1+ (0.005)(50°C − 25°C)}
22V
5A
(0.035Ω) + (22V)2 6 (2.5Ω)(215pF) •
2
1
1
+
(350kHz) = 433mW
5V − 2.3V 2.3V
PMAIN =
A short-circuit to ground will result in a folded back current of:
ISC =
20 mV 1 95ns(22V)
−
= 3.07A
0.006Ω 2 3.9µH
with a typical value of RDS(ON) and z = (0.005/°C)(25°C)
= 0.125. The resulting power dissipated in the bottom
MOSFET is:
2
PSYNC = (2.23A) (1.125)(0.022Ω) = 233mW
which is less than under full-load conditions.
The input capacitor to the buck regulator CIN is chosen
for an RMS current rating of at least 3A at temperature
assuming only this channel is on. COUT is chosen with an
ESR of 0.02Ω for low output ripple. The output ripple in
continuous mode will be highest at the maximum input voltage. The output voltage ripple due to ESR is approximately:
VORIPPLE = RESR (DIL) = 0.02Ω(1.75A) = 35mVP-P
3859alf
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For more information www.linear.com/3859AL
LTC3859AL
Applications Information
PC Board Layout Checklist
When laying out the printed circuit board, the following
checklist should be used to ensure proper operation of
the IC. These items are also illustrated graphically in the
layout diagram of Figure 11. Figure 12 illustrates the current
waveforms present in the various branches of the 2-phase
synchronous buck regulators operating in the continuous
mode. Check the following in your layout:
1.Are the top N-channel MOSFETs MTOP1 and MTOP2
located within 1cm of each other with a common drain
connection at CIN? Do not attempt to split the input
decoupling for the two channels as it can cause a large
resonant loop.
2.Are the signal and power grounds kept separate? The
combined IC signal ground pin and the ground return
of CINTVCC must return to the combined COUT (–) terminals. The path formed by the top N-channel MOSFET,
Schottky diode and the CIN capacitor should have short
leads and PC trace lengths. The output capacitor (–)
terminals should be connected as close as possible
to the (–) terminals of the input capacitor by placing
the capacitors next to each other and away from the
Schottky loop described above.
3.Do the LTC3859AL VFB pins’ resistive dividers connect to the (+) terminals of COUT? The resistive divider
must be connected between the (+) terminal of COUT
and signal ground. The feedback resistor connections
should not be along the high current input feeds from
the input capacitor(s).
4.Are the SENSE– and SENSE+ leads routed together with
minimum PC trace spacing? The filter capacitor between
SENSE+ and SENSE– should be as close as possible
to the IC. Ensure accurate current sensing with Kelvin
connections at the sense resistor.
5.Is the INTVCC decoupling capacitor connected close
to the IC, between the INTVCC and the power ground
pins? This capacitor carries the MOSFET drivers’ current peaks. An additional 1µF ceramic capacitor placed
immediately next to the INTVCC and PGND pins can help
improve noise performance substantially.
6.Keep the switching nodes (SW1, SW2, SW3), top gate
nodes (TG1, TG2, TG3), and boost nodes (BOOST1,
BOOST2, BOOST3) away from sensitive small-signal
nodes, especially from the opposites channel’s voltage
and current sensing feedback pins. All of these nodes
have very large and fast moving signals and therefore
should be kept on the output side of the LTC3859AL
and occupy minimum PC trace area.
7.Use a modified star ground technique: a low impedance,
large copper area central grounding point on the same
side of the PC board as the input and output capacitors
with tie-ins for the bottom of the INTVCC decoupling
capacitor, the bottom of the voltage feedback resistive
divider and the SGND pin of the IC.
3859alf
For more information www.linear.com/3859AL
35
LTC3859AL
Applications Information
PC Board Layout Debugging
Start with one controller on at a time. It is helpful to use
a DC-50MHz current probe to monitor the current in the
inductor while testing the circuit. Monitor the output switching node (SW pin) to synchronize the oscilloscope to the
internal oscillator and probe the actual output voltage as
well. Check for proper performance over the operating
voltage and current range expected in the application.
The frequency of operation should be maintained over the
input voltage range down to dropout and until the output
load drops below the low current operation threshold—
typically 25% of the maximum designed current level in
Burst Mode operation.
The duty cycle percentage should be maintained from cycle
to cycle in a well-designed, low noise PCB implementation.
Variation in the duty cycle at a subharmonic rate can suggest noise pickup at the current or voltage sensing inputs
or inadequate loop compensation. Overcompensation of
the loop can be used to tame a poor PC layout if regulator
bandwidth optimization is not required. Only after each
controller is checked for its individual performance should
both controllers be turned on at the same time. A particularly
difficult region of operation is when one controller channel
is nearing its current comparator trip point when the other
channel is turning on its top MOSFET. This occurs around
50% duty cycle on either channel due to the phasing of
the internal clocks and may cause minor duty cycle jitter.
Reduce VIN from its nominal level to verify operation of
the regulator in dropout. Check the operation of the undervoltage lockout circuit by further lowering VIN while
monitoring the outputs to verify operation.
Investigate whether any problems exist only at higher output currents or only at higher input voltages. If problems
coincide with high input voltages and low output currents,
look for capacitive coupling between the BOOST, SW, TG,
and possibly BG connections and the sensitive voltage
and current pins. The capacitor placed across the current
sensing pins needs to be placed immediately adjacent to
the pins of the IC. This capacitor helps to minimize the
effects of differential noise injection due to high frequency
capacitive coupling. If problems are encountered with
high current output loading at lower input voltages, look
for inductive coupling between CIN, Schottky and the top
MOSFET components to the sensitive current and voltage
sensing traces. In addition, investigate common ground
path voltage pickup between these components and the
SGND pin of the IC.
An embarrassing problem, which can be missed in an
otherwise properly working switching regulator, results
when the current sensing leads are hooked up backwards.
The output voltage under this improper hookup will still
be maintained but the advantages of current mode control
will not be realized. Compensation of the voltage loop will
be much more sensitive to component selection. This
behavior can be investigated by temporarily shorting out
the current sensing resistor—don’t worry, the regulator
will still maintain control of the output voltage.
3859alf
36
For more information www.linear.com/3859AL
LTC3859AL
Applications Information
SW1
L1
RSENSE1
D1
VOUT1
COUT1
RL1
VIN
RIN
CIN
SW2
BOLD LINES INDICATE
HIGH SWITCHING
CURRENT. KEEP LINES
TO A MINIMUM LENGTH.
L2
RSENSE2
D2
VOUT2
COUT2
RL2
3859al F11
Figure 11. Branch Current Waveforms for Bucks
3859alf
For more information www.linear.com/3859AL
37
LTC3859AL
Typical Applications
VOUT1
RB1
357k
OPT
RA1
68.1k
VFB1
LTC3859AL
SENSE1–
C1
1nF
CITH1A
100pF
SENSE1+
RITH1
15k
CITH1
1500pF
CSS1
0.1µF
ITH1
PGOOD1
100k
MTOP1
TG1
TRACK/SS1
FREQ
PLLIN/MODE
SW1
RUN1
RA2
68.1k
RB2
649k
10pF
RUN3
VOUT1
5V
5A
COUT1
220µF
D1
VBIAS
CBIAS
10µF
PGND
VFB2
CITH2
2.2nF
CINT1
4.7µF
RITH2
15k
ITH2
CITH2A
68pF
C2
10µF
INTVCC
D2
TG2
CSS2
0.1µF
TRACK/SS2
MTOP2
CB2
0.1µF
BOOST2
L2
6.5µH
RSENSE2
8mΩ
SW2
VOUT3
RA3
68.1k
RSENSE1
6mΩ
MBOT1
BG1
RUN2
VOUT2
L1
4.9µH
CB1
0.1µF
BOOST1
SGND
C1
10µF
RB3
499k
OPT
COUT2
68µF
MBOT2
BG2
VOUT2
8.5V
3A
VFB3
CITH3
0.01µF
RITH3
3.6k
SENSE2+
C2
1nF
ITH3
CITH3A
820pF
SENSE2–
CSS3
0.1µF
SS3
VOUT1
EXTVCC
MTOP3
TG3
SW3
OV3
MTOP1, MTOP2: BSZ097NO4LS
MBOT1, MBOT2: BSZ097NO4LS
MTOP3: BSC027NO4LS
MBOT3: BSCO1BN04LS
L1: WÜRTH 744314490
L2: WÜRTH 744314650
L3: WÜRTH 744325120
COUT1: SANYO 6TPB220ML
COUT2: SANYO 10TPC68M
CIN, COUT3: SANYO 50CE220LX
D1, D2: CMDH-4E
D3: BAS140W
VOUT3
10V*
D3
BOOST3
L3
1.2µH
COUT3
220µF
RSENSE2
2mΩ
VIN
2.5V TO 38V
(START-UP ABOVE 5V)
CB3
0.1µF
BG3
MBOT3
SENSE3–
CIN
220µF
C3
1nF
SENSE3+
3859al F12
* VOUT3 IS 10V WHEN VIN < 10V,
FOLLOWS VIN WHEN VIN > 10V
Figure 12. High Efficiency Wide Input Range Dual 5V/8.5V Converter
3859alf
38
For more information www.linear.com/3859AL
LTC3859AL
typical Applications
VOUT1
RB1
475k
33pF
RA1
34k
VFB1
LTC3859AL
SENSE1–
C1
1nF
CITH1A
100pF
SENSE1+
RITH1
10k
CITH1
680pF
ITH1
CSS1
0.1µF
PGOOD1
FREQ
PLLIN/MODE
SW1
RUN1
RB2
215k
15pF
RUN3
VOUT1
12V
3A
COUT1
47µF
D1
VBIAS
CBIAS
10µF
PGND
VFB2
CITH2
820pF
CINT1
4.7µF
RITH2
15k
ITH2
CITH2A
150pF
C2
10µF
INTVCC
D2
TG2
CSS2
0.1µF
TRACK/SS2
MTOP2
CB2
0.1µF
BOOST2
L2
3.2µH
RSENSE2
6mΩ
SW2
VOUT3
RA3
68.1k
RSENSE1
9mΩ
MBOT1
BG1
RUN2
VOUT2
L1
8.8µH
CB1
0.1µF
BOOST1
SGND
C1
10µF
MTOP1
TG1
TRACK/SS1
RA2
68.1k
100k
RB3
787k
OPT
COUT2
150µF
MBOT2
BG2
VOUT2
3.3V
5A
VFB3
CITH3
0.01µF
RITH3
3.6k
SENSE2+
C2
1nF
ITH3
CITH3A
820pF
SENSE2–
CSS3
0.1µF
VOUT3
15V*
D3
INTVCC
MTOP1, MTOP2: VISHAY Si7848DP
MBOT1, MBOT2: VISHAY Si7848DP
MTOP3: BSC027NO4LS
MBOT3: BSCO1BN04LS
L1: SUMIDA CDEP105-8R8M
L2: SUMIDA CDEP105-3R2M
L3: WÜRTH 744325120
COUT1: KEMET T525D476MO16E035
COUT2: SANYO 4TPE150M
CIN, COUT3: SANYO 50CE220LX
D1, D2: CMDH-4E
D3: BAS140W
100k
SS3
TG3
OV3
SW3
BOOST3
EXTVCC
MTOP3
L3
1.2µH
COUT3
220µF
RSENSE2
2mΩ
VIN
2.5V TO 38V
(START-UP ABOVE 5V)
CB3
0.1µF
BG3
MBOT3
SENSE3–
CIN
220µF
C3
1nF
SENSE3+
3859al F13
* VOUT3 IS 15V WHEN VIN < 15V,
FOLLOWS VIN WHEN VIN > 15V
Figure 13. High Efficiency Wide Input Range Dual 12V/3.3V Converter
3859alf
For more information www.linear.com/3859AL
39
LTC3859AL
typical Applications
VOUT1
RA1
115k
RB1
28.7k
56pF
CITH1A
200pF
LTC3859AL
SENSE1–
C1
1nF
SENSE1+
RITH1
3.93k
CITH1
1000pF
VFB1
ITH1
CSS1
0.01µF
PGOOD1
FREQ
PLLIN/MODE
SW1
RUN1
RB2
57.6k
56pF
RUN3
VOUT1
1V
8A
COUT1
220µF
×2
D1
VBIAS
CBIAS
10µF
PGND
VFB2
CITH2
1000pF
CINT1
4.7µF
RITH2
3.93k
ITH2
CITH2A
200pF
C2
10µF
INTVCC
D2
TG2
CSS2
0.01µF
TRACK/SS2
MTOP2
CB2
0.1µF
BOOST2
L2
0.47µH
RSENSE2
3.5mΩ
VOUT2
1.2V
8A
SW2
VOUT3
RA3
12.1k
RSENSE1
3.5mΩ
MBOT1
BG1
RUN2
VOUT2
L1
0.47µH
CB1
0.1µF
BOOST1
SGND
C1
10µF
MTOP1
TG1
TRACK/SS1
RA2
115k
100k
RB3
232k
OPT
COUT2
220µF
×2
MBOT2
BG2
VFB3
CITH3
15nF
RITH3
8.66k
CITH3A
220pF
CSS3
0.01µF
MTOP1, MTOP2: RENESAS RJK0305
MBOT1, MBOT2: RENESAS RJK0328
MTOP3, MBOT3: RENESAS HAT2169H
L1, L2: SUMIDA CDEP105-0R4
L3: PULSE PA1494.362NL
COUT1, COUT2: SANYO 2R5TPE220M
CIN, COUT3: SANYO 50CE220AX
D1, D2: CMDH-4E
D3: BAS140W
SENSE2+
C2
1nF
ITH3
SENSE2–
D3
SS3
TG3
OV3
SW3
BOOST3
EXTVCC
MTOP3
L3
3.3µH
RSENSE2
4mΩ
CB3
0.1µF
BG3
MBOT3
COUT3
220µF
VOUT3
24V
5A
VIN
12V
CIN
220µF
SENSE3–
C3
1nF
SENSE3+
3859al F14
Figure 14. High Efficiency Triple 24V/1V/1.2V Converter from 12V VIN
3859alf
40
For more information www.linear.com/3859AL
LTC3859AL
typical Applications
VOUT1
RB1
57.6k
RA1
115k
CITH1A
100pF
LTC3859AL
SENSE1–
C1
1nF
SENSE1+
RITH1
5.6k
CITH1
2.2nF
VFB1
ITH1
CSS1
0.1µF
PGOOD1
100k
MTOP1
TG1
TRACK/SS1
FREQ
PLLIN/MODE
SW1
RUN1
RB2
357k
RA2
115k
RUN3
RSENSE1
9mΩ
VOUT1
1.2V
3A
COUT1
220µF
MBOT1
BG1
D1
VBIAS
CBIAS
10µF
RUN2
VOUT2
L1
2.2µH
CB1
0.1µF
BOOST1
SGND
C1
10µF
PGND
VFB2
CITH2
3.3nF
CINT1
4.7µF
RITH2
9.1k
ITH2
CITH2A
100pF
C2
10µF
INTVCC
D2
TG2
CSS2
0.1µF
TRACK/SS2
MTOP2
CB2
0.1µF
BOOST2
L2
6.5µH
RSENSE2
9mΩ
VOUT2
3.3V
3A
SW2
VOUT3
RB3
887k
RA3
115k
COUT2
220µF
MBOT2
BG2
VFB3
CITH3
100nF
RITH3
13k
SENSE2+
C2
1nF
ITH3
CITH3A
10pF
SENSE2–
CSS3
0.1µF
TG3
OV3
SW3
COUT3
270µF
•
SS3
D3
EXTVCC
BOOST3
RSENSE2
9mΩ
•
MTOP1, MTOP2: BSZ097NO4LS
MBOT1, MBOT2: BSZ097NO4LS
MBOT3: BSZ097NO4L
L1: WURTH 744311220
L2: WURTH 744314650
L3: COOPER BUSSMANN DRQ125-100
COUT1: SANYO 2R5TPE220MAFB
COUT2: SANYO 4TPE220MAZB
COUT3: SANYO SVPC270M
CIN: SANYO 50CE220LX
D1, D2: CMDH-4E
D3: DIODES INC B360A-13-F
C3
10µF
50V
BG3
MBOT3
L3
10µH
SENSE3–
VOUT3
10.5V
1.2A
VIN
5.8V TO 34V
CIN
220µF
C3
1nF
SENSE3+
3859al F15
Figure 15. High Efficiency 1.2V/3.3V Step-Down Converter with 10.5V SEPIC Converter
3859alf
For more information www.linear.com/3859AL
41
LTC3859AL
Package Description
Please refer to http://www.linear.com/designtools/packaging/ for the most recent package drawings.
FE Package
38-Lead Plastic TSSOP (4.4mm)
(Reference LTC DWG # 05-08-1772 Rev C)
Exposed Pad Variation AA
4.75 REF
38
9.60 – 9.80*
(.378 – .386)
4.75 REF
(.187)
20
6.60 ±0.10
4.50 REF
2.74 REF
SEE NOTE 4
6.40
2.74
REF (.252)
(.108)
BSC
0.315 ±0.05
1.05 ±0.10
0.50 BSC
RECOMMENDED SOLDER PAD LAYOUT
4.30 – 4.50*
(.169 – .177)
0.09 – 0.20
(.0035 – .0079)
0.50 – 0.75
(.020 – .030)
NOTE:
1. CONTROLLING DIMENSION: MILLIMETERS
2. DIMENSIONS ARE IN MILLIMETERS
(INCHES)
3. DRAWING NOT TO SCALE
1
0.25
REF
19
1.20
(.047)
MAX
0° – 8°
0.50
(.0196)
BSC
0.17 – 0.27
(.0067 – .0106)
TYP
0.05 – 0.15
(.002 – .006)
FE38 (AA) TSSOP REV C 0910
4. RECOMMENDED MINIMUM PCB METAL SIZE
FOR EXPOSED PAD ATTACHMENT
*DIMENSIONS DO NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.150mm (.006") PER SIDE
3859alf
42
For more information www.linear.com/3859AL
LTC3859AL
Package Description
Please refer to http://www.linear.com/designtools/packaging/ for the most recent package drawings.
UHF Package
38-Lead Plastic QFN (5mm × 7mm)
(Reference LTC DWG # 05-08-1701 Rev C)
0.70 ±0.05
5.50 ±0.05
5.15 ±0.05
4.10 ±0.05
3.00 REF
3.15 ±0.05
PACKAGE
OUTLINE
0.25 ±0.05
0.50 BSC
5.5 REF
6.10 ±0.05
7.50 ±0.05
RECOMMENDED SOLDER PAD LAYOUT
APPLY SOLDER MASK TO AREAS THAT ARE NOT SOLDERED
5.00 ±0.10
0.75 ±0.05
PIN 1 NOTCH
R = 0.30 TYP OR
0.35 × 45° CHAMFER
3.00 REF
37
0.00 – 0.05
38
0.40 ±0.10
PIN 1
TOP MARK
(SEE NOTE 6)
1
2
5.15 ±0.10
5.50 REF
7.00 ±0.10
3.15 ±0.10
(UH) QFN REF C 1107
0.200 REF 0.25 ±0.05
R = 0.125
TYP
0.50 BSC
R = 0.10
TYP
BOTTOM VIEW—EXPOSED PAD
NOTE:
1. DRAWING CONFORMS TO JEDEC PACKAGE
OUTLINE M0-220 VARIATION WHKD
2. DRAWING NOT TO SCALE
3. ALL DIMENSIONS ARE IN MILLIMETERS
4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE
MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.20mm ON ANY SIDE
5. EXPOSED PAD SHALL BE SOLDER PLATED
6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION
ON THE TOP AND BOTTOM OF PACKAGE
3859alf
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection
of its circuits
as described
herein will not infringe on existing patent rights.
For more
information
www.linear.com/3859AL
43
LTC3859AL
Typical Application
High Efficiency Wide Input Range Dual 3.3V/8.5V Converter
VOUT1
RA1
68.1k
RB1
215k
15pF
VFB1
LTC3859AL
SENSE1–
CITH1A
150pF
SENSE1+
RITH1
15k
CITH1
820pF
ITH1
CSS1
0.1µF
PGOOD1
C1
1nF
100k
TRACK/SS1
FREQ
PLLIN/MODE
SW1
RUN1
RA2
68.1k
RB2
649k
10pF
RUN3
VOUT1
3.3V
5A
COUT1
150µF
D1
VBIAS
CBIAS
10µF
PGND
VFB2
CITH2
2.2nF
CINT1
4.7µF
RITH2
15k
ITH2
CITH2A
68pF
C2
10µF
INTVCC
D2
TG2
CSS2
0.1µF
TRACK/SS2
MTOP2
CB2
0.1µF
BOOST2
L2
6.5µH
RSENSE2
8mΩ
SW2
VOUT3
RA3
68.1k
RSENSE1
6mΩ
MBOT1
BG1
RUN2
VOUT2
L1
3.2µH
CB1
0.1µF
BOOST1
SGND
C1
10µF
MTOP1
TG1
RB3
499k
OPT
COUT2
68µF
MBOT2
BG2
VOUT2
8.5V
3A
VFB3
CITH3
0.01µF
RITH3
3.6k
SENSE2+
CITH3A
820pF
SENSE2–
CSS3
0.1µF
MTOP1, MTOP2: VISHAY Si7848DP
MBOT1, MBOT2: BSZ097NO4LS
MTOP3: BSC027NO4LS
MBOT3: BSCO1BN04LS
L1: SUMIDA CDEP105-3R2M
L2: WÜRTH 744314650
L3: WÜRTH 744325120
COUT1: SANYO 6TPB220ML
COUT2: SANYO 4TPE150M
CIN, COUT3: SANYO 50CE220LX
D1, D2: CMDH-4E
D3: BAS140W
C2
1nF
ITH3
VOUT3
10V*
D3
SS3
SW3
OV3
VOUT2
EXTVCC
MTOP3
TG3
BOOST3
L3
1.2µH
RSENSE2
2mΩ
CB3
0.1µF
BG3
MBOT3
SENSE3–
COUT3
220µF
VIN
2.5V TO 38V
(START-UP ABOVE 5V)
CIN
220µF
C3
1nF
SENSE3+
* VOUT3 IS 10V WHEN VIN < 10V,
FOLLOWS VIN WHEN VIN > 10V
3859al TA02
Related Parts
PART NUMBER DESCRIPTION
COMMENTS
LTC3786
Low IQ Synchronous Step-Up DC/DC Controller
4.5V (Down to 2.5V after Start-Up) ≤ VIN ≤ 38V, VOUT Up to 60V, IQ = 55µA
PLL Fixed Frequency 50kHz to 900kHz, 3mm × 3mm QFN-16, MSOP-16E
LTC3787
Low IQ, Multiphase, Dual Channel Single Output
Synchronous Step-Up DC/DC Controller
4.5V (Down to 2.5V after Start-Up) ≤ VIN ≤ 38V, VOUT up to 60V, PLL Fixed
Frequency 50kHz to 900kHz, IQ = 135µA
LTC3826/
LTC3826-1
Low IQ, Dual Output 2-Phase Synchronous Step-Down
DC/DC Controllers with 99% Duty Cycle
PLL Fixed Frequency 50kHz to 900kHz, 4V≤ VIN ≤ 36V, 0.8V ≤ VOUT ≤ 10V,
IQ = 30µA
LTC3890/
LTC3890-1/
LTC3890-2
60V, Low IQ, Dual 2-Phase Synchronous Step-Down DC/DC PLL Fixed Frequency 50kHz to 900kHz, 4V ≤ VIN ≤ 60V, 0.8V ≤ VOUT ≤ 24V,
Controller with 99% Duty Cycle
IQ = 50µA
LTC3891
60V, Low IQ, Synchronous Step-Down DC/DC Controller
with 99% Duty Cycle
PLL Fixed Frequency 50kHz to 900kHz, 4V ≤ VIN ≤ 60V, 0.8V ≤ VOUT ≤ 24V,
IQ = 50µA
LTC3864
60V, Low IQ, High Voltage DC/DC Controller with 100%
Duty Cycle
Fixed Frequency 50kHz to 850kHz, 3.5V≤ VIN ≤ 60V, 0.8V ≤ VOUT ≤ VIN,
IQ = 40µA, MSOP-12E, 3mm × 4mm DFN-12
LTC3789
4-Switch High Efficiency Buck-Boost DC/DC Controller
4V≤ VIN ≤ 38V, 0.8V ≤ VOUT ≤ 38V
3859alf
44 Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
For more information www.linear.com/3859AL
(408) 432-1900 ● FAX: (408) 434-0507
●
www.linear.com/3859AL
LT 0113 • PRINTED IN USA
© LINEAR TECHNOLOGY CORPORATION 2013