LT3474/LT3474-1 Step-Down 1A LED Driver FEATURES DESCRIPTION n The LT®3474/LT3474-1 are fixed frequency step-down DC/DC converters designed to operate as constant-current sources. An internal sense resistor monitors the output current allowing accurate current regulation, ideal for driving high current LEDs. High output current accuracy is maintained over a wide current range, from 35mA to 1A, allowing a wide dimming range. n n n n n n n n n n n True Color PWM™ Delivers Constant Color with 400:1 Dimming Range Wide Input Range: 4V to 36V Up to 1A LED Current Adjustable 200kHz–2MHz Switching Frequency Adjustable Control of LED Current Integrated Boost Diode High Output Current Accuracy is Maintained Over a Wide Range from 35mA to 1A Open LED (LT3474) and Short-Circuit Protection High Side Sense Allows Grounded Cathode Connection Uses Small Inductors and Ceramic Capacitors LT3474-1 Drives LED Strings Up to 26V Compact 16-Lead TSSOP Thermally Enhanced Surface Mount Package Unique PWM circuitry allows a dimming range of 400:1, avoiding the color shift normally associated with LED current dimming. The high switching frequency offers several advantages, permitting the use of small inductors and ceramic capacitors. Small inductors combined with the 16-lead TSSOP surface mount package save space and cost versus alternative solutions. The constant switching frequency combined with low-impedance ceramic capacitors result in low, predictable output ripple. APPLICATIONS ■ ■ ■ ■ Automotive and Avionic Lighting Architectural Detail Lighting Display Backlighting Constant Current Sources L, LT, LTC and LTM are registered trademarks of Linear Technology Corporation. True Color PWM is a trademark of Linear Technology Corporation. All other trademarks are the property of their respective owners. Patent Pending With their wide input range of 4V to 36V, the LT3474/ LT3474-1 regulate a broad array of power sources, from 5V logic rails to unregulated wall transformers, lead acid batteries and distributed power supplies. A current mode PWM architecture provides fast transient response and cycle-by-cycle current limiting. Frequency foldback and thermal shutdown provide additional protection. TYPICAL APPLICATION Step-Down 1A LED Driver Efficiency 95 VIN 5V TO 36V 0.22μF BOOST SHDN 10μH LT3474 RT 80.6k BIAS REF OUT VADJ PWM 0.1μF DIMMING* CONTROL 2.2μF LED VC TWO SERIES CONNECTED WHITE 1A LEDS 85 SW EFFICIENCY (%) VIN 2.2μF VIN = 12V 90 80 ONE WHITE 1A LED 75 70 65 60 GND LED1 55 *SEE APPLICATIONS SECTION FOR DETAILS 3474 TA01a 0 200 400 800 600 LED CURRENT (mA) 1000 3474 G02 3474fd 1 LT3474/LT3474-1 ABSOLUTE MAXIMUM RATINGS PIN CONFIGURATION (Note 1) VIN Pin ........................................................(–0.3V), 36V BIAS Pin....................................................................25V BOOST Pin Voltage ...................................................51V BOOST above SW Pin ...............................................25V OUT, LED Pins (LT3474) ............................................15V OUT, LED Pins (LT3474-1).........................................26V PWM Pin ...................................................................10V VADJ Pin .....................................................................6V VC, REF, RT Pins ..........................................................3V SHDN Pin ...................................................................VIN BIAS Pin Current .........................................................1A Maximum Junction Temperature (Note 2)............. 125°C Operating Temperature Range (Note 3) LT3474E, LT3474E-1 ............................ –40°C to 85°C LT3474I, LT3474I-1 ............................ –40°C to 125°C Storage Temperature Range................... –65°C to 150°C Lead Temperature (Soldering, 10 sec) .................. 300°C TOP VIEW DNC* 1 16 DNC* OUT 2 15 GND LED 3 VIN 4 14 PWM 17 13 VADJ SW 5 12 VC BOOST 6 11 REF BIAS 7 10 SHDN GND 8 9 RT FE PACKAGE 16-LEAD PLASTIC TSSOP θJC = 8°C/W, θJA = 40°C/W EXPOSED PAD (PIN 17) IS GND, MUST BE SOLDERED TO PCB *DO NOT CONNECT EXTERNAL CIRCUITRY TO THESE PINS. ORDER INFORMATION LEAD FREE FINISH TAPE AND REEL PART MARKING PACKAGE DESCRIPTION TEMPERATURE RANGE LT3474EFE#PBF LT3474EFE#TRPBF 3474EFE 16-Lead TSSOP –40°C to 85°C LT3474IFE#PBF LT3474IFE#TRPBF 3474IFE 16-Lead TSSOP –40°C to 125°C LT3474EFE-1#PBF LT3474EFE-1#TRPBF 3474EFE-1 16-Lead TSSOP –40°C to 85°C LT3474IFE-1#PBF LT3474IFE-1#TRPBF 3474IFE-1 16-Lead TSSOP –40°C to 125°C Consult LTC Marketing for parts specified with wider operating temperature ranges. Consult LTC Marketing for information on non-standard lead based finish parts. For more information on lead free part marking, go to: http://www.linear.com/leadfree/ For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/ ELECTRICAL CHARACTERISTICS The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN = 12V, VBOOST = 16V, VOUT = 4V unless otherwise noted (Note 3). PARAMETER CONDITIONS MIN l Minimum Input Voltage TYP MAX 3.5 4 UNITS V Input Quiescent Current Not Switching 2.6 4 mA Shutdown Current SHDN = 0.3V, VBOOST = 0V, VOUT = 0V 0.01 2 μA LED Pin Current VADJ Tied to VREF 1 1.02 1.025 0.207 0.210 A A A A 1.265 V VADJ Tied to VREF/5 REF Voltage l 0.98 0.968 0.193 0.186 l 1.23 l 0.2 1.25 3474fd 2 LT3474/LT3474-1 ELECTRICAL CHARACTERISTICS The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN = 12V, VBOOST = 16V, VOUT = 4V unless otherwise noted (Note 3). PARAMETER CONDITIONS Reference Voltage Line Regulation 5V < VIN < 36V Reference Voltage Load Regulation 0 < IREF < 250μA MIN l VADJ Pin Bias Current (Note 4) Switching Frequency RT = 80.6k Maximum Duty Cycle RT = 80.6k RT = 10k RT = 232k Foldback Frequency RT = 80.6k, VOUT = 0V MAX UNITS 0.01 %/V 0.0002 %/μA 20 400 nA 530 540 kHz kHz l 470 450 500 l 90 95 76 98 % % % 70 kHz SHDN Threshold (to Switch) SHDN Pin Current (Note 5) TYP VSHDN = SHDN Threshold PWM Threshold 2.6 2.65 2.7 V 8.3 10.3 12.3 μA 0.4 0.9 1.2 V VC Switching Threshold 0.8 V VC Source Current VC = 1V 100 μA VC Sink Current VC = 1V 100 μA 1.5 μA/mA 1 V/mA 2 A/V LED to VC Current Gain LED to VC Transresistance VC to Switch Current Gain VC Clamp Voltage VC Pin Current in PWM Mode 1.9 l VC = 1V, VPWM = 0.3V OUT Pin Clamp Voltage (LT3474) OUT Pin Current in PWM Mode Switch Current Limit (Note 6) 13.2 VOUT = 4V, VPWM = 0.3V l –40°C to 85°C LT3474I, LT3474I-1 at 125°C l 1.6 1.5 V 0.01 1 μA 13.8 14.5 V 0.1 10 μA 2.1 3.2 3.2 A A Switch VCESAT ISW = 1A 380 500 mV Boost Pin Current ISW = 1A 30 50 mA Switch Leakage Current 0.01 1 μA Minimum Boost Voltage (Note 7) 1.9 2.5 V Boost Diode Forward Voltage IDIO = 100mA Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Note 2: This IC includes overtemperature protection that is intended to protect the device during momentary overload conditions. Junction temperature will exceed 125°C when overtemperature protection is active. Continuous operation above the specified maximum operating junction temperature may impair device reliability. Note 3: The LT3474E and LT3474E-1 are guaranteed to meet performance specifications from 0°C to 70°C. Specifications over the –40°C to 85°C 600 mV operating temperature range are assured by design, characterization and correlation with statistical process controls. The LT3474I and LT3474I-1 are guaranteed to meet performance specifications over the –40°C to 125°C operating temperature range. Note 4: Current flows out of pin. Note 5: Current flows into pin. Note 6: Current limit is guaranteed by design and/or correlation to static test. Slope compensation reduces current limit at higher duty cycles. Note 7: This is the minimum voltage across the boost capacitor needed to guarantee full saturation of the internal power switch. 3474fd 3 LT3474/LT3474-1 TYPICAL PERFORMANCE CHARACTERISTICS LED Current vs VADJ LED Current vs Temperature 700 TA = 25°C 600 600 400 200 800 600 400 VADJ = VREF/5 200 0 0.25 0.75 0.5 VADJ (V) 1 50 25 75 0 TEMPERATURE (°C) 100 3474 GO3 CURRENT LIMIT (A) CURRENT LIMIT (A) TYPICAL MINIMUM (85°C) MINIMUM (125°C) 1 60 40 DUTY CYCLE (%) 80 Current Limit vs Output Voltage 2.5 2 2 TA = 25°C 1.5 1 0 –25 50 25 0 75 TEMPERATURE (°C) 100 125 OSCILLATOR FREQUENCY (kHz) OSCILLATOR FREQUENCY (kHz) 3474 G09 4 6 VOUT (V) 10 8 550 500 450 –25 12 600 RT = 80.6k 400 –50 RT (kΩ) 2 Oscillator Frequency Foldback 600 100 0 3474 G08 Oscillator Frequency vs Temperature TA = 25°C OSCILLATOR FREQUENCY (kHz) 1 3474 G07 Oscillator Frequency vs RT 10 1.5 0.5 3474 G06 100 1500 3474 G05 2.5 0 –50 100 1000 1000 500 SWITCH CURRENT (mA) 0 0.5 0.5 20 200 0 125 CURRENT LIMIT (A) 2 0 300 Switch Current Limit vs Temperature 2.5 0 400 3474 G04 Current Limit vs Duty Cycle 1.5 500 100 0 –50 –25 1.25 SWITCH VOLTAGE DROP (mV) 1000 LED CURRENT (mA) LED CURRENT (mA) TA = 25°C VADJ = VREF 800 0 Switch Voltage Drop 1200 1000 75 0 25 50 TEMPERATURE (°C) 100 125 3474 G10 TA = 25°C RT = 80.6k 500 400 300 200 100 0 0 0.5 1 1.5 VOUT (V) 2 2.5 3474 G11 3474fd 4 LT3474/LT3474-1 TYPICAL PERFORMANCE CHARACTERISTICS Boost Pin Current Quiescent Current TA = 25°C TA = 25°C 2.5 30 20 2.0 1.250 1.5 1.240 0.5 10 0 0 250 500 750 1000 1250 SWITCH CURRENT (mA) 1500 0 6 12 18 VIN (V) 24 30 Schottky Reverse Leakage –25 75 0 25 50 TEMPERATURE (°C) 100 125 50 400 300 200 100 0 0 3474 G15 6 40 5 LT3474 30 4 3 LT3474-1 20 LT3474 1000 2 OUTPUT VOLTAGE 1 0 0 10 20 VIN (V) 30 3474 G19 10 TA = 25°C 40 3474 G16 Minimum Input Voltage, Two Series Connected White Luxeon III Stars TA = 25°C TO START 5 7 LT3474-1 0 200 600 800 400 FORWARD VOLTAGE (mV) Minimum Input Voltage, One White Luxeon III Star 6 INPUT CURRENT 10 9 TO RUN 4 LED VOLTAGE VIN (V) 0 –50 8 TA = 25°C OUTPUT VOLTAGE (V) FORWARD CURRENT (mA) 5 125 60 TA = 25°C 10 100 Open-Circuit Output Voltage and Input Current 500 15 50 25 0 75 TEMPERATURE (°C) –25 3474 G14 Schottky Forward Voltage Drop VR = 5V VIN (V) REVERSE CURRENT (μA) 1.235 –50 36 3474 G13 3473 G12 20 1.245 1.0 INPUT CURRENT (mA) 40 1.255 VREF (V) INPUT CURRENT (mA) BOOST PIN CURRENT (mA) 50 0 Reference Voltage 1.260 3.0 60 3 8 TO START 7 TO RUN LED VOLTAGE 2 6 1 0 0 200 400 600 800 LED CURRENT (mA) 1000 3474 G17 5 0 200 600 800 400 LED CURRENT (mA) 1000 3474 G18 3474fd 5 LT3474/LT3474-1 PIN FUNCTIONS DNC (Pins 1, 16): Do not connect external circuitry to these pins, or tie them to GND. Leave the DNC pins floating. OUT (Pin 2): The OUT pin is the input to the current sense resistor. Connect this pin to the inductor and the output capacitor. LED (Pin 3): The LED pin is the output of the current sense resistor. Connect the anode of the LED here. SHDN (Pin 10): The SHDN pin is used to shut down the switching regulator and the internal bias circuits. The 2.6V switching threshold can function as an accurate under-voltage lockout. Pull below 0.3V to shut down the LT3474/LT3474-1. Pull above 2.65V to enable the LT3474/ LT3474-1. Tie to VIN if the SHDN function is unused. VIN (Pin 4): The VIN pin supplies current to the internal circuitry and to the internal power switch and must be locally bypassed. REF (Pin 11): The REF pin is the buffered output of the internal reference. Either tie the REF pin to the VADJ pin for a 1A output current, or use a resistor divider to generate a lower voltage at the VADJ pin. Leave this pin unconnected if unused. SW (Pin 5): The SW pin is the output of the internal power switch. Connect this pin to the inductor and switching diode. VC (Pin 12): The Vc pin is the output of the internal error amp. The voltage on this pin controls the peak switch current. Use this pin to compensate the control loop. BOOST (Pin 6): The BOOST pin is used to provide a drive voltage, higher than the input voltage, to the internal bipolar NPN power switch. VADJ (Pin 13): The VADJ pin is the input to the internal voltage to current amplifier. Connect the VADJ pin to the REF pin for a 1A output current. For lower output currents, program the VADJ pin using the following formula: ILED = 1A • VADJ/1.25V. BIAS (Pin 7): The BIAS pin connects through a Schottky diode to BOOST. Tie to OUT. GND (Pins 8, 15, Exposed Pad Pin 17): Ground. Tie both GND pins and the Exposed Pad directly to the ground plane. The Exposed Pad metal of the package provides both electrical contact to ground and good thermal contact to the printed circuit board. It must be soldered to the circuit board for proper operation. PWM (Pin 14): The PWM pin controls the connection of the VC pin to the internal circuitry. When the PWM pin is low, the VC pin is disconnected from the internal circuitry and draws minimal current. If the PWM feature is unused, leave this pin unconnected. RT (Pin 9): The RT pin is used to set the internal oscillator frequency. Tie an 80.6k resistor from RT to GND for a 500kHz switching frequency. 3474fd 6 LT3474/LT3474-1 BLOCK DIAGRAM VIN 4 VIN CIN BIAS 10 SHDN INT REG AND UVLO BOOST ∑ SLOPE COMP C1 9 7 RT R Q S Q 6 C1 Q1 DRIVER SW OSC L1 5 RT D1 FREQUENCY FOLDBACK OUT 2 – C2 100Ω 2V LED + 1.25V 11 0.1Ω 3 DLED1 gm REF PWM 14 13 PWM VADJ VC USE WITH PWM DIMMING 12 Q2 CC1 CC2 RC 1.25k GND 8 3474 BD Figure 1. Block Diagram 3474fd 7 LT3474/LT3474-1 APPLICATIONS INFORMATION Operation The LT3474 is a constant frequency, current mode regulator with an internal power switch capable of generating a constant 1A output. Operation can be best understood by referring to the Block Diagram. If the SHDN pin is tied to ground, the LT3474 is shut down and draws minimal current from the input source tied to VIN. If the SHDN pin exceeds 1.5V, the internal bias circuits turn on, including the internal regulator, reference, and oscillator. The switching regulator will only begin to operate when the SHDN pin exceeds 2.65V. The switcher is a current mode regulator. Instead of directly modulating the duty cycle of the power switch, the feedback loop controls the peak current in the switch during each cycle. Compared to voltage mode control, current mode control improves loop dynamics and provides cycle-bycycle current limit. A pulse from the oscillator sets the RS flip-flop and turns on the internal NPN bipolar power switch. Current in the switch and the external inductor begins to increase. When this current exceeds a level determined by the voltage at VC, current comparator C1 resets the flip-flop, turning off the switch. The current in the inductor flows through the external Schottky diode and begins to decrease. The cycle begins again at the next pulse from the oscillator. In this way, the voltage on the VC pin controls the current through the inductor to the output. The internal error amplifier regulates the output current by continually adjusting the VC pin voltage. The threshold for switching on the VC pin is 0.8V, and an active clamp of 1.9V limits the output current. The voltage on the VADJ pin sets the current through the LED pin. The NPN Q2 pulls a current proportional to the voltage on the VADJ pin through the 100Ω resistor. The gm amplifier servos the VC pin to set the current through the 0.1Ω resistor and the LED pin. When the voltage drop across the 0.1Ω resistor is equal to the voltage drop across the 100Ω resistor, the servo loop is balanced. Tying the REF pin to the VADJ pin sets the LED pin current to 1A. Tying a resistor divider to the REF pin allows the programming of LED pin currents of less than 1A. LED pin current can also be programmed by tying the VADJ pin directly to a voltage source up to 1.25V. An LED can be dimmed with pulse width modulation using the PWM pin and an external NFET. If the PWM pin is unconnected or pulled high, the part operates nominally. If the PWM pin is pulled low, the VC pin is disconnected from the internal circuitry and draws minimal current from the compensation capacitor. Circuitry drawing current from the OUT pin is also disabled. This way, the VC pin and the output capacitor store the state of the LED pin current until PWM is pulled high again. This leads to a highly linear relationship between pulse width and output light, allowing for a large and accurate dimming range. The RT pin allows programming of the switching frequency. For applications requiring the smallest external components possible, a fast switching frequency can be used. If very low or very high input voltages are required, a slower switching frequency can be programmed. During startup VOUT will be at a low voltage. The NPN Q2 can only operate correctly with sufficient voltage at VOUT, around 1.7V. A comparator senses VOUT and forces the VC pin high until VOUT rises above 2V, and Q2 is operating correctly. The switching regulator performs frequency foldback during overload conditions. An amplifier senses when VOUT is less than 2V and begins decreasing the oscillator frequency down from full frequency to 20% of the nominal frequency when VOUT = 0V. The OUT pin is less than 2V during startup, short circuit, and overload conditions. Frequency foldback helps limit switch current under these conditions. 3474fd 8 LT3474/LT3474-1 APPLICATIONS INFORMATION The switch driver operates either from VIN or from the BOOST pin. An external capacitor and internal Schottky diode are used to generate a voltage at the BOOST pin that is higher than the input supply. This allows the driver to saturate the internal bipolar NPN power switch for efficient operation. An internal comparator will force the part into shutdown when VIN falls below 3.5V. If an adjustable UVLO threshold is required, the SHDN pin can be used. The threshold voltage of the SHDN pin comparator is 2.65V. A internal resistor pulls 10.3μA to ground from the SHDN pin at the UVLO threshold. Choose resistors according to the following formula: Open Circuit Protection The LT3474 has internal open circuit protection. If the LED is absent or fails open, the LT3474 clamps the voltage on the LED pin at 14V. The switching regulator then skips cycles to limit the input current. The LT3474-1 has no internal open circuit protection. With the LT3474-1, be careful not to violate the ABSMAX voltage of the BOOST pin; if VIN > 25V, external open circuit protection circuitry (as shown in Figure 2) may be necessary. The output voltage during an open LED condition is shown in the Typical Performance Characteristics section. R2 = VTH = UVLO Threshold Example: Switching should not start until the input is above 8V. VTH = 8V R1 = 100k R2 = Undervoltage Lockout Undervoltage lockout (UVLO) is typically used in situations where the input supply is current limited, or has high source resistance. A switching regulator draws constant power from the source, so the source current increases as the source voltage drops. This looks like a negative resistance load to the source and can cause the source to current limit or latch low under low source voltage conditions. UVLO prevents the regulator from operating at source voltages where these problems might occur. 2.65V VTH – 2.65V – 10.3μA R1 2.65V = 61.9k 8V – 2.65V – 10.3μA 100k Keep the connections from the resistors to the SHDN pin short and make sure the coupling to the SW and BOOST pins is minimized. If high resistance values are used, the SHDN pin should be bypassed with a 1nF capacitor to prevent coupling problems from switching nodes. LT3474 VIN OUT VIN 2.65V R1 10k VC SHDN 27V VC C1 R2 10.3μA GND 100k 3474 F03 3474 F02 Figure 2. External Overvoltage Protection Circuitry for the LT3474-1. Figure 3. Undervoltage Lockout 3474fd 9 LT3474/LT3474-1 APPLICATIONS INFORMATION Setting the Switching Frequency The LT3474 uses a constant frequency architecture that can be programmed over a 200kHz to 2MHz range with a single external timing resistor from the RT pin to ground. The current that flows into the timing resistor is used to charge an internal oscillator capacitor. A graph for selecting the value of RT for a given operating frequency is shown in the Typical Performance Characteristics section. Table 1 shows suggested RT selections for a variety of switching frequencies. Table 1. Switching Frequencies SWITCHING FREQUENCY (MHz) RT (kΩ) 2 10 1.5 18.7 1 33.2 0.7 52.3 0.5 80.6 0.3 147 0.2 232 Operating Frequency Selection The choice of operating frequency is determined by several factors. There is a tradeoff between efficiency and component size. Higher switching frequency allows the use of smaller inductors at the cost of increased switching losses and decreased efficiency. Another consideration is the maximum duty cycle. In certain applications, the converter needs to operate at a high duty cycle in order to work at the lowest input voltage possible. The LT3474 has a fixed oscillator off-time and a variable on-time. As a result, the maximum duty cycle increases as the switching frequency is decreased. Input Voltage Range The minimum operating voltage is determined either by the LT3474’s undervoltage lockout of 4V, or by its maximum duty cycle. The duty cycle is the fraction of time that the internal switch is on and is determined by the input and output voltages: DC = ( VOUT + VF ) ( VIN – VSW + VF ) where VF is the forward voltage drop of the catch diode (~0.4V) and VSW is the voltage drop of the internal switch (~0.4V at maximum load). This leads to a minimum input voltage of: VIN(MIN) = VOUT + VF – VF + VSW DCMAX with DCMAX = 1–tOFF(MIN) • f where t0FF(MIN) is equal to 200ns and f is the switching frequency. Example: f = 500kHz, VOUT = 4V DCMAX = 1− 200ns • 500kHz = 0.90 4V + 0.4V – 0.4V + 0.4V = 4.9 V VIN(MIN) = 0.99 The maximum operating voltage is determined by the absolute maximum ratings of the VIN and BOOST pins, and by the minimum duty cycle. VIN(MAX ) = VOUT + VF – VF + VSW DCMIN with DCMIN = tON(MIN) • f where tON(MIN) is equal to 160ns and f is the switching frequency. Example: f = 500kHz, VOUT = 2.5V DCMIN = 160ns • 500kHz = 0.08 2.5V + 0.4V – 0.4V + 0.4V = 36 V VIN(MAX ) = 0.008 The minimum duty cycle depends on the switching frequency. Running at a lower switching frequency might allow a higher maximum operating voltage. Note that this is a restriction on the operating input voltage; the circuit will tolerate transient inputs up to the Absolute Maximum Rating. 3474fd 10 LT3474/LT3474-1 APPLICATIONS INFORMATION Inductor Selection and Maximum Output Current A good first choice for the inductor value is 900kHz f where VF is the voltage drop of the catch diode (~0.4V), f is the switching frequency and L is in μH. With this value the maximum load current will be 1.1A, independent of input voltage. The inductor’s RMS current rating must be greater than the maximum load current and its saturation current should be at least 30% higher. For highest efficiency, the series resistance (DCR) should be less than 0.2Ω. Table 2 lists several vendors and types that are suitable. For robust operation at full load and high input voltages (VIN > 30V), use an inductor with a saturation current higher than 2.5A. L = ( VOUT + VF ) • Table 2. Inductors VALUE (μH) IRMS (A) DCR (Ω) HEIGHT (mm) CR43-3R3 3.3 1.44 0.086 3.5 CR43-4R7 4.7 1.15 0.109 3.5 PART NUMBER Sumida CDRH4D16-3R3 3.3 1.1 0.063 1.8 CDRH4D28-3R3 3.3 1.57 0.049 3 CDRH4D28-4R7 4.7 1.32 0.072 3 CDRH5D28-100 10 1.3 0.048 3 CDRH5D28-150 15 1.1 0.076 3 CDRH73-100 10 1.68 0.072 3.4 CDRH73-150 15 1.33 0.13 3.4 DO1606T-332 3.3 1.3 0.1 2 DO1606T-472 4.7 1.1 0.12 2 DO1608C-332 3.3 2 0.08 2.9 DO1608C-472 4.7 1.5 0.09 2.9 MOS6020-332 3.3 1.8 0.046 2 MOS6020-472 10 1.5 0.05 2 Coilcraft The optimum inductor for a given application may differ from the one indicated by this simple design guide. A larger value inductor provides a higher maximum load current, and reduces the output voltage ripple. If your load is lower than the maximum load current, then you can relax the value of the inductor and operate with higher ripple current. This allows you to use a physically smaller inductor, or one with a lower DCR resulting in higher efficiency. Be aware that if the inductance differs from the simple rule above, then the maximum load current will depend on input voltage. In addition, low inductance may result in discontinuous mode operation, which further reduces maximum load current. For details of maximum output current and discontinuous mode operation, see Linear Technology’s Application Note 44. Finally, for duty cycles greater than 50% (VOUT/VIN > 0.5), a minimum inductance is required to avoid sub-harmonic oscillations. See Application Note 19. The current in the inductor is a triangle wave with an average value equal to the load current. The peak switch current is equal to the output current plus half the peak-to-peak inductor ripple current. The LT3474 limits its switch current in order to protect itself and the system from overload faults. Therefore, the maximum output current that the LT3474 will deliver depends on the switch current limit, the inductor value, and the input and output voltages. When the switch is off, the potential across the inductor is the output voltage plus the catch diode drop. This gives the peak-to-peak ripple current in the inductor ΔIL = (1– DC) ( VOUT + VF ) (L • f) where f is the switching frequency of the LT3474 and L is the value of the inductor. The peak inductor and switch current is ISW (PK ) =IL (PK ) =IOUT + ΔIL 2 3474fd 11 LT3474/LT3474-1 APPLICATIONS INFORMATION To maintain output regulation, this peak current must be less than the LT3474’s switch current limit ILIM. For SW1, ILIM is at least 1.6A (1.5A at 125°C) at low duty cycles and decreases linearly to 1.15A (1.08A at 125°C) at DC = 0.8. The maximum output current is a function of the chosen inductor value: ΔI IOUT (MAX ) = ILIM – L 2 =1.6A • (1 – 0.35 •DC) – at the LT3474 input and to force this switching current into a tight local loop, minnimizing EMI. The input capacitor must have low impedance at the switching frequency to do this effectively, and it must have an adequate ripple current rating. The RMS input is: CINRMS = IOUT • ΔIL 2 Choosing an inductor value so that the ripple current is small will allow a maximum output current near the switch current limit. One approach to choosing the inductor is to start with the simple rule given above, look at the available inductors, and choose one to meet cost or space goals. Then use these equations to check that the LT3474 will be able to deliver the required output current. Note again that these equations assume that the inductor current is continuous. Discontinuous operation occurs when IOUT is less than ΔIL/2. Input Capacitor Selection Bypass the input of the LT3474 circuit with a 2.2μF or higher ceramic capacitor of X7R or X5R type. A lower value or a less expensive Y5V type will work if there is additional bypassing provided by bulk electrolytic capacitors or if the input source impedance is low. The following paragraphs describe the input capacitor considerations in more detail. ( VOUT VIN – VOUT VIN ) < IOUT 2 and is largest when VIN = 2VOUT (50% duty cycle). Considering that the maximum load current is 1A, RMS ripple current will always be less than 0.5A The high switching frequency of the LT3474 reduces the energy storage requirements of the input capacitor, so that the capacitance required is less than 10μF. The combination of small size and low impedance (low equivalent series resistance or ESR) of ceramic capacitors makes them the preferred choice. The low ESR results in very low voltage ripple. Ceramic capacitors can handle larger magnitudes of ripple current than other capacitor types of the same value. Use X5R and X7R types. An alternative to a high value ceramic capacitor is a lower value ceramic along with a larger electrolytic capacitor. The electrolytic capacitor likely needs to be greater than 10μF in order to meet the ESR and ripple current requirements. The input capacitor is likely to see high surge currents when the input source is applied. Tantalum capacitors can fail due to an over-surge of current. Only use tantalum capacitors with the appropriate surge current rating. The manufacturer may also recommend operation below the rated voltage of the capacitor. Step-down regulators draw current from the input supply in pulses with very fast rise and fall times. The input capacitor is required to reduce the resulting voltage ripple 3474fd 12 LT3474/LT3474-1 APPLICATIONS INFORMATION A final caution is in order regarding the use of ceramic capacitors at the input. A ceramic input capacitor can combine with stray inductance to form a resonant tank circuit. If power is applied quickly (for example by plugging the circuit into a live power source), this tank can ring, doubling the input voltage and damaging the LT3474. The solution is to either clamp the input voltage or dampen the tank circuit by adding a lossy capacitor in parallel with the ceramic capacitor. For details, see Application Note 88. Output Capacitor Selection For most LEDs, a 2.2μF 6.3V ceramic capacitor (X5R or X7R) at the output results in very low output voltage ripple and good transient response. Other types and values will also work; the following discusses tradeoffs in output ripple and transient performance. The output capacitor filters the inductor current to generate an output with low voltage ripple. It also stores energy in order to satisfy transient loads and stabilizes the LT3474’s control loop. Because the LT3474 operates at a high frequency, minimal output capacitance is necessary. In addition, the control loop operates well with or without the presence of output capacitor series resistance (ESR). Ceramic capacitors, which achieve very low output ripple and small circuit size, are therefore an option. You can estimate output ripple with the following equation: VRIPPLE = ΔIL (8 • f •COUT ) for ceramic capacitors where ΔIL is the peak-to-peak ripple current in the inductor. The RMS content of this ripple is very low so the RMS current rating of the output capacitor is usually not of concern. It can be estimated with the formula: IC (RMS) = ΔIL 12 The low ESR and small size of ceramic capacitors make them the preferred type for LT3474 applications. Not all ceramic capacitors are the same, however. Many of the higher value capacitors use poor dielectrics with high temperature and voltage coefficients. In particular, Y5V and Z5U types lose a large fraction of their capacitance with applied voltage and at temperature extremes. Because loop stability and transient response depend on the value of COUT, this loss may be unacceptable. Use X7R and X5R types. Table 3 lists several capacitor vendors. Table 3. Low-ESR Surface Mount Capacitors VENDOR TYPE SERIES Taiyo-Yuden Ceramic X5R, X7R AVX Ceramic X5R, X7R TDK Ceramic X5R, X7R 3474fd 13 LT3474/LT3474-1 APPLICATIONS INFORMATION Diode Selection The catch diode (D1 from Figure 1) conducts current only during switch off time. Average forward current in normal operation can be calculated from: ID( AVG) = IOUT ( VIN – VOUT ) VIN The only reason to consider a diode with a larger current rating than necessary for nominal operation is for the worst-case condition of shorted output. The diode current will then increase to one half the typical peak switch current. Table 4 lists several Schottky diodes and their manufacturers. Table 4. Schottky Diodes VR (V) I AVE (A) VF at 0.5A (mV) MBR0520L 20 0.5 385 MBR0540 40 0.5 510 MBRM120E 20 1 530 MBRM140 40 1 550 B0530W 30 0.5 PART NUMBER VF at 1A (mV) On Semiconductor 620 Diodes Inc. 430 B120 20 1 500 Peak reverse voltage is equal to the regulator input voltage. Use a diode with a reverse voltage rating greater than the input voltage. B130 30 1 500 B140 HB 40 1 530 If using the PWM mode of the LT3474, select a diode with low reverse leakage. 10BQ030 30 1 420 International Rectifier 3474fd 14 LT3474/LT3474-1 APPLICATIONS INFORMATION BOOST and BIAS Pin Considerations The capacitor and internal diode tied to the BOOST pin generate a voltage that is higher than the input voltage. In most cases, a 0.22μF capacitor will work well. Figure 4 shows three ways to arrange the boost circuit. The BOOST pin must be more than 2.5V above the SW pin for full efficiency. For outputs of 2.8V or higher, the standard circuit (Figure 4a) is best. For lower output voltages, the BIAS pin BIAS BOOST LT3474 VIN SW VIN C3 VOUT Programming LED Current GND 3474 F04a VBOOST – VSW ≈ VOUT MAX VBOOST ≈ VIN + VOUT (4a) BIAS BOOST LT3474 VIN SW VIN can be tied to the input (Figure 4b). The circuit in Figure 4a is more efficient because the BOOST pin current comes from a lower voltage source. The BIAS pin can be tied to another source that is at least 3V (Figure 4c). For example, if a 3.3V source is on whenever the LED is on, the BIAS pin can be connected to the 3.3V output. For LT3474-1 applications with higher output voltages, an additional Zener diode may be necessary (Figure 4d) to maintain the BOOST pin voltage below the absolute maximum. In any case, be sure that the maximum voltage at the BOOST pin is both less than 51V and the voltage difference between the BOOST and SW pins is less than 25V. C3 The LED current can be set by adjusting the voltage on the VADJ pin. For a 1A LED current, either tie VADJ to REF or to a 1.25V source. For lower output currents, program the VADJ using the following formula: ILED = VOUT GND 3474 F04b 1A • VADJ 1.25V Voltages less than 1.25V can be generated with a voltage divider from the REF pin, as shown in Figure 5. VBOOST – VSW ≈ VIN MAX VBOOST ≈ 2VIN (4b) REF R1 VIN2 > 3V VIN BIAS BOOST LT3474 VIN SW C3 R2 3474 F04 Figure 5. Setting VADJ with a Resistor Divider 3474 F04c VBOOST – VSW ≈ VIN2 MAX VBOOST ≈ VIN2 + VIN MINIMUM VALUE FOR VIN2 = 3V In order to have accurate LED current, precision resistors are preferred (1% or better is recommended). Note that the VADJ pin sources a small amount of bias current, so use the following formula to choose resistors: (4c) BIAS BOOST LT3474 VIN SW GND VOUT GND VIN LT3474 VADJ C3 VOUT GND 3474 F04d R2 = VADJ 1.25V – VADJ + 50nA R1 VBOOST – VSW ≈ VOUT – VZ MAX VBOOST ≈ VIN + VOUT – VZ (4d) Figure 4. Generating the Boost Voltage 3474fd 15 LT3474/LT3474-1 APPLICATIONS INFORMATION To minimize the error from variations in VADJ pin current, use resistors with a parallel resistance of less than 4k. Use resistors with a series resistance of 5.11k or greater so as not to exceed the 250μA current limit on the REF pin. Dimming Control There are several different types of dimming control circuits. One dimming control circuit (Figure 6) changes the voltage on the VADJ pin by tying a low on-resistance FET to the resistor divider string. This allows the selection of two different LED currents. For reliable operation, program an LED current of no less than 35mA. The maximum current dimming ratio (IRATIO) can be calculated from the maximum LED current (IMAX) and the minimum LED current (IMIN) as follows: the C-RC string (tied to the VC pin) shown in Figure 7 for proper operation during start-up. When the PWM pin goes high again, the LED current returns rapidly to its previous on state since the compensation and output capacitors are at the correct voltage. This fast settling time allows The LT3474 to maintain diode current regulation with PWM pulse widths as short as 40μs. If the NFET is omitted and the cathode of the LED is instead tied to GND, use PWM pulse widths of 1ms or greater. The maximmum PWM dimming ratio (PWMRATIO) can be calculated from the maximum PWM period (tMAX) and minimum PWM pulse width (tMIN) as follows: tMAX = PWMRATIO tMIN Total dimming ratio (DIMRATIO) is the product of the PWM dimming ratio and the current dimming ratio. IMAX = IRATIO IMIN Another dimming control circuit (Figure 7) uses the PWM pin and an external NFET tied to the cathode of the LED. When the PWM signal goes low, the NFET turns off, turning off the LED and leaving the output capacitor charged. The PWM pin is pulled low as well, which disconnects the VC pin, storing the voltage in the capacitor tied there. Use Example: IMAX = 1A, IMIN = 0.1A, tMAX = 12ms, tMIN = 40μs 1A =10:1 0.1A 12ms PWMRATIO = = 300:1 40μs DIMRATIO =10 • 300 = 3000:1 IRATIO = REF R1 LT3474 PWM 60Hz TO 10kHz VADJ R2 DIM GND PWM LT3474 3474 F05 LED GND 3.3nF 10k 0.1μF 3474 F06 Figure 6. Dimming with an NFET and Resistor Divider Figure 7. Dimming Using PWM Signal 3474fd 16 LT3474/LT3474-1 APPLICATIONS INFORMATION LED Voltage Range The LT3474 can drive LED voltages from 2.4V to 12V. The LT3474-1 can drive LED voltages from 2.4V to 30V. Be careful not to exceed the ABSMAX rating of the OUT, LED, or BOOST pins of the LT3474-1 since the internal output clamp is disabled. See the Typical Application section for an example of adding an external output clamp. If the LED voltage can drift below 2.4V due to temperature or component variation, add extra series resistance to bring the overall voltage above 2.4V. as short as possible. To prevent electromagnetic interference (EMI) problems, proper layout of the high frequency switching path is essential. The voltage signal of the SW and BOOST pins have sharp rise and fall edges. Minimize the area of all traces connected to the BOOST and SW pins and always use a ground plane under the switching regulator to minimize interplane coupling. In addition, the ground connection for frequency setting resistor RT (refer to Figure 1) should be tied directly to the GND pin and not shared with any other component, ensuring a clean, noise-free connection. Layout Hints As with all switching regulators, careful attention must be paid to the PCB layout and component placement. To maximize efficiency, switch rise and fall times are made PWM SHDN VIN GND VIA TO LOCAL GND PLANE VIA TO OUT Figure 8. Recommended Component Placement 3474fd 17 LT3474/LT3474-1 TYPICAL APPLICATIONS Step-Down 1A LED Driver with PWM Dimming LED Current in PWM Mode VIN 6V TO 36V C1 2.2μF 50V VIN BOOST SHDN SW RT V(PWM) 5V/DIV D1 LT3474 R1 80.6k ILED1 500mA/DIV C3 0.22μF L1 10μH 6.3V BIAS REF OUT VADJ PWM VC LED C4 3.3nF 1ms/DIV C2 2.2μF 6.3V GND LED1 R2 10k M1 C5 0.1μF PWM 3474 TA01 D1: B140HB C1 TO C3: X5R OR X7R M1: Si2302ADS Step-Down 1A LED Driver with Two Series Connected LED Output Efficiency, Two LED Output 95 90 C1 2.2μF 50V VIN C3 0.22μF L1 10μH 10V BOOST SHDN SW D1 LT3474 R1 33.2k C4 0.1μF RT BIAS REF OUT VADJ PWM VC C2 2.2μF 10V LED GND LED1 LED2 D1: MBRM 140 C1 TO C3: X5R OR X7R VIN = 24V 80 75 70 65 60 55 1A LED CURRENT VIN = 12V 85 EFFICIENCY (%) VIN 12V TO 36V 0 200 400 800 600 LED CURRENT (mA) 1000 3474 G01 3474 TA02 3474fd 18 LT3474/LT3474-1 PACKAGE DESCRIPTION FE Package 16-Lead Plastic TSSOP (4.4mm) (Reference LTC DWG # 05-08-1663) Exposed Pad Variation BA 4.90 – 5.10* (.193 – .201) 2.74 (.108) 2.74 (.108) 16 1514 13 12 1110 6.60 ±0.10 9 2.74 (.108) 4.50 ±0.10 2.74 6.40 (.108) (.252) BSC SEE NOTE 4 0.45 ±0.05 1.05 ±0.10 0.65 BSC 1 2 3 4 5 6 7 8 RECOMMENDED SOLDER PAD LAYOUT 4.30 – 4.50* (.169 – .177) 0.09 – 0.20 (.0035 – .0079) 0.50 – 0.75 (.020 – .030) NOTE: 1. CONTROLLING DIMENSION: MILLIMETERS MILLIMETERS 2. DIMENSIONS ARE IN (INCHES) 3. DRAWING NOT TO SCALE 0.25 REF 1.10 (.0433) MAX 0° – 8° 0.65 (.0256) BSC 0.195 – 0.30 (.0077 – .0118) TYP 0.05 – 0.15 (.002 – .006) FE16 (BA) TSSOP 0204 4. RECOMMENDED MINIMUM PCB METAL SIZE FOR EXPOSED PAD ATTACHMENT *DIMENSIONS DO NOT INCLUDE MOLD FLASH. MOLD FLASH SHALL NOT EXCEED 0.150mm (.006") PER SIDE 3474fd Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. 19 LT3474/LT3474-1 TYPICAL APPLICATION Step-Down 1A LED Driver with Four Series Connected LED Output VIN 21V TO 36V C1 2.2μF 50V SHDN SW LT3474-1 BIAS RT REF C4 0.1μF D2 L1 47μH D1 OUT VADJ R1 80.6k C3 0.22μF 16V BOOST VIN PWM LED VC C2 2.2μF 25V R2 10k GND D3 12V TO 18V LED VOLTAGE 1A LED CURRENT Q1 R3 100k fSW = 500kHz 3474 TA02a D1: MBRM 140 D2: 7.5V Zener Diode D3: 22V Zener Diode Q1: MMBT3904 C1 TO C3: X5R OR X7R RELATED PARTS PART NUMBER DESCRIPTION COMMENTS LT1618 Constant Current, 1.4MHz, 1.5A Boost Converter VIN: 1.6V to 18V, VOUT(MAX) = 36V, IQ = 1.8mA, ISD = <1μA, MS10 Package LT1766 60V, 1.2A (IOUT ), 200kHz, High Efficiency Step-Down DC/DC Converter VIN: 5.5V to 60V, VOUT(MAX) = 1.20V, IQ = 2.5mA, ISD = 25μA, TSSOP16/E Packages LT1956 60V, 1.2A (IOUT ), 500kHz, High Efficiency Step-Down DC/DC Converter VIN: 5.5V to 60V, VOUT(MAX) = 1.20V, IQ = 2.5mA, ISD = 25μA, TSSOP16/E Packages LT1961 1.5A (ISW ), 1.25MHz, High Efficiency Step-Up DC/DC Converter VIN: 3V to 25V, VOUT(MAX) = 35V, IQ = 0.9mA, ISD = 6μA, MS8E Package LT1976/LT1977 60V, 1.2A (IOUT ), 200kHz/500kHz, High Efficiency Step-Down DC/DC Converters with BurstMode ® Operation VIN: 3.3V to 60V, VOUT(MAX) = 1.20V, IQ = 100μA, ISD = <1μA, TSSOP16E Package LT3430/LT3431 60V, 2.5A (IOUT ), 200kHz, High Efficiency Step-Down DC/DC Converters VIN: 5.5V to 60V, VOUT(MAX) = 1.20V, IQ = 2.5μA, ISD = <25μA, TSSOP16/E Packages LT3433 60V, 400mA (IOUT ), 200kHz, High Efficiency Step-Up/Step-Down DC/DC Converters with Burst Mode Operation VIN: 4V to 60V, VOUT: 3.3V to 20V, IQ = 100μA, ISD = <1μA, TSSOP16E Package LT3434/LT3435 60V, 2.5A (IOUT ), 200kHz/500kHz, High Efficiency Step-Down DC/DC Converters with Burst Mode Operation VIN: 3.3V to 60V, VOUT(MAX) = 1.20V, IQ = 100μA, ISD = <1μA, TSSOP16E Package LTC3453 VIN: 2.7V to 5.5V, VOUT(MAX) = 5.5V, IQ = 2.5mA, ISD = <6μA, QFN Package 1MHz, 800mA Synchronous Buck-Boost High Power LED Driver LT3467/LT3467A 1.1A (ISW), 1.3MHz/2.1MHz, High Efficiency Step-Up DC/DC Converters with Integrated Soft-Start VIN: 2.4V to 16V, VOUT(MAX) = 40V, IQ = 1.2mA, ISD = <1μA, ThinSOT™ Package LT3477 3A, 42V, 3MHz Step-Up Regulator with Dual Rail to Rail Current Sense VIN: 2.5V to 2.5V, VOUT(MAX) = 40V, IQ = 5mA, ISD = <1μA, QFN, TSSOP16E Packages LT3479 3A, Full Featured DC/DC Converter with Soft-Start and Inrush Current Protection VIN: 2.5V to 24V, VOUT(MAX) = 40V, IQ = 6.5mA, ISD = <1μA, DFN and TSSOP Packages Burst Mode is a registered trademark of Linear Technology Corporation. ThinSOT is a trademark of Linear Technology Corporation. 3474fd 20 Linear Technology Corporation LT 1008 REV D • PRINTED IN USA 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408) 432-1900 ● FAX: (408) 434-0507 ● www.linear.com © LINEAR TECHNOLOGY CORPORATION 2005